JP6519384B2 - SAW sensor resonance frequency detection device - Google Patents

Description

  The present invention relates to a resonance frequency detection device of a surface acoustic wave (SAW) sensor.

  When a surface acoustic wave sensor (hereinafter referred to as a SAW sensor) receives a change in the physical quantity of the object to be measured due to temperature or strain, the propagation speed of the SAW changes accordingly and the electrical characteristics change. Among such systems, a system for detecting a physical quantity in response to detection of a resonant frequency of a SAW sensor has been developed (see, for example, Non-Patent Document 1). According to the technology described in Non-Patent Document 1, the resonant frequency is detected by detecting the frequency at which the level of the reflection characteristic S11 of the SAW is minimum (see, for example, FIG. 3 of Non-Patent Document 1).

J. Beckley, V. Kalinin, M. Lee, K. Voliansky, "Non-contact torque sensors based on SAW resonators", Frequency Control Symposium and PDA Exhibition, 2002 IEEE International, vol, no, pp. 202-213

Japanese Patent Application Publication No. 2005-528595

  When the inventor examines the technique described in Non-Patent Document 1 and mixes the received signal with the signal of the carrier frequency and down-converts, the carrier frequency is in the same tendency as the low frequency range in the high frequency range beyond a certain frequency. It was derived that the output value was obtained, so that the high frequency range and the low frequency range beyond a certain frequency could not be distinguished uniquely. When detecting the resonant frequency of the SAW sensor within a monotonically changing frequency range, it is not possible to widen the detectable frequency range (locking range), and it has been found that the degree of freedom in design is inferior.

  An object of the present invention is to provide a resonant frequency detector for a SAW sensor, which can detect the resonant frequency of the SAW sensor while keeping the locking range as wide as possible.

  According to the first aspect of the present invention, the transmitter transmits the narrowband frequency-modulated signal to the carrier to the SAW sensor including the resonant SAW element, and the receiver transmits the transmission signal transmitted to the SAW sensor by the transmitter. The signal is received via the SAW element of the SAW sensor. The resonance frequency detection device detects the resonance frequency of the SAW element of the SAW sensor based on the phase component detected according to the reception signal of the receiver.

  At this time, in the auxiliary circuit, the frequency characteristic of the amplitude product of the transmission signal and the reception signal when the transmission signal of the transmitter and the reception signal of the receiver are mixed has a minimum value at the amplitude resonance frequency and a frequency region lower than the amplitude resonance frequency. The characteristic is monotonically decreasing according to the frequency increase and monotonously increasing according to the frequency increase in the frequency domain higher than the amplitude resonance frequency, and the phase difference between the transmission signal and the reception signal becomes maximum at the phase difference resonance frequency. It is configured to have a characteristic of monotonously increasing in response to an increase in frequency in a frequency range lower than the phase difference resonance frequency and decreasing monotonically in response to a frequency increase in a frequency range higher than the phase difference resonance frequency.

  The receiver receives the transmission signal transmitted to the SAW sensor by the transmitter through the auxiliary circuit and the SAW sensor, converts the reception signal from the SAW sensor, and generates the frequency component of the modulation signal and the modulation signal The phase difference component according to the phase difference with the modulation signal of the unit is fed back to the transmitter and superimposed on the modulation signal of the modulation signal generation unit. The transmitter oscillates a signal using a signal in which the phase difference component is superimposed on the modulation signal as a control signal, and causes the carrier frequency to converge on the resonance frequency of the SAW sensor. As a result, the resonance frequency of the SAW sensor can be detected with high accuracy.

A block diagram schematically showing an example of a conceptual electrical configuration of a resonance frequency detecting device of a SAW sensor according to the first embodiment A block diagram schematically showing an example of a specific electrical configuration of a resonance frequency detection device of a SAW sensor Diagram schematically showing the configuration of a SAW sensor and an image of signal propagation Characteristic diagram schematically showing the frequency dependency of the amplitude product Characteristic diagram schematically showing the frequency dependence of the phase difference cosine value Characteristic diagram schematically showing frequency dependency of phase difference Diagram schematically showing an example of characteristic change when the parameter of the auxiliary circuit is changed Characteristic diagram schematically showing frequency dependency of amplitude product and phase difference in comparative example Characteristic diagram schematically showing frequency dependency of amplitude product and phase difference in the first embodiment A block diagram schematically showing an example of a conceptual electrical configuration of a resonant frequency detection device of a SAW sensor in a second embodiment Diagram schematically showing the image of the configuration of the SAW sensor and the signal propagation characteristics A block diagram schematically showing an example of a conceptual electrical configuration of a resonant frequency detection device of a SAW sensor in a third embodiment A circuit diagram showing an example of an electrical configuration of an envelope detection unit A block diagram schematically showing an example of a conceptual electrical configuration of a resonant frequency detection device for a SAW sensor in a fourth embodiment A block diagram schematically showing an example of a conceptual electrical configuration of a resonant frequency detection device for a SAW sensor in a fifth embodiment A block diagram schematically showing an example of a conceptual electrical configuration of a resonance frequency detecting device of a SAW sensor in a sixth embodiment A block diagram schematically showing an example of a conceptual electrical configuration of a resonant frequency detection device for a SAW sensor in a seventh embodiment

  Hereinafter, some embodiments of a resonant frequency detection device of a SAW (Surface Acoustic Wave) sensor will be described with reference to the drawings. In each embodiment described below, the same or similar reference numerals are assigned to components performing the same or similar operations, and the description will be omitted as necessary.

First Embodiment
1 to 9 show an explanatory view of the first embodiment. The sensing system 1 shown in FIG. 1 includes a SAW sensor 2 disposed so as to be deformed according to an externally applied strain, and a resonant frequency detection device 3 connected to the SAW sensor 2. FIG. 2 shows a specific example of the electrical configuration of the sensing system 1 shown in FIG.

  The resonance frequency detection device 3 includes a transmitter 7 including a modulation signal generator (corresponding to a modulation signal generation unit) 4, an adder 5, and an oscillator 6, an amplifier 8, 9, a down conversion unit 10, a filter 11, a phase comparator ( And a receiver 15 including a filter 13 and a calculator 14. An auxiliary circuit 16 is connected between the transmitter 7 and the receiver 15. The amplifiers 8 and 9 may be provided as needed, and may not be added if the signal level is sufficient.

  As shown in FIG. 2, the oscillator 6 is constituted by, for example, a VCO 6z (Voltage-Controlled Oscillator). The down conversion unit 10 includes, for example, a mixer 10 z. The filter 11 is configured of, for example, a BPF 11z (Band-Pass Filter). The phase comparator 12 is composed of, for example, a mixer 12z. The filter 13 is configured by, for example, an LPF 13 z (Low-Pass Filter). The auxiliary circuit 16 is constituted by, for example, a passive circuit (capacitive circuit or inductive circuit), and is constituted by using, for example, an auxiliary element 16z (eg, a capacitor element or an inductor element) of a single element.

The modulation signal generator 4 of the transmitter 7 outputs a modulation signal (AC signal) of a predetermined modulation frequency for narrow band FM modulation, for example. The adder 5 superimposes the modulation signal (for example, AC voltage) of the modulation signal generator 4 and the output signal (for example, DC voltage) of the computing unit 14 and outputs the result as a control signal _Vctrl to the oscillator 6. The oscillator 6 inputs the signal input from the adder 5 as a control signal _Vctrl to control the carrier, outputs the narrowband frequency-modulated signal after the control to the amplifier 8, and the SAW sensor through the auxiliary circuit 16. 2 and the amplifier 9.

Oscillator 6 changes the carrier frequency f _c in accordance with the DC component the DC component output (corresponding phase difference component) is changed from the receiver 15 to be described later.
The input impedances of the two amplifiers 8 and 9 are set to a value (several tens of ohms) identical or close to the characteristic impedance (for example, 50 ohms) of a transmission line such as the node N1 transmitting the output signal of the oscillator 6. Assuming that nodes of input terminals of the two amplifiers 8 and 9 are a first node N1 and a second node N2, respectively, an auxiliary circuit 16 is connected between the first node N1 and the second node N2. The auxiliary circuit 16 is configured using, for example, a passive circuit.

As schematically shown in FIG. 3 the peripheral configuration and signal propagation image of the SAW sensor 2, the SAW sensor 2 is connected via an antenna 17 (not shown in FIG. 1) connected to the node N2 side. It is done. The SAW sensor 2 includes a resonant SAW element 19 configured by connecting an antenna 18. The SAW element 19 has an interdigital transducer 20 disposed on a piezoelectric substrate (not shown), and is formed in the same electrode film as the interdigital transducer 20 on both sides of the interdigital transducer 20. The reflector 21 is disposed. The SAW element 19 inputs the transmission signal V _TX input through the antenna 18, and simultaneously outputs a propagation signal to the side of the node N2 through the antennas 18 and 17. The antennas 17 and 18 can be configured using, for example, an inductive antenna or a resonant antenna.

The amplitude and the phase of the reception signal V _RX reflected from the SAW sensor 2 change at each frequency in the frequency component of the transmission signal V _TX according to the transmission / reception transfer function of the SAW sensor 2. Since the amplitude product and phase difference of the transmission signal _VTX and the reception signal _VRX depend on the resonance frequency, the receiver 15 described later calculates the resonance frequency using a circuit. Thereby, the distortion (physical quantity) applied to the SAW sensor 2 can be detected.

The two amplifiers 8 and 9 are amplifiers in which the same amplification factor or the amplification factor is preset to a predetermined ratio, and output the amplified signal to the down conversion unit 10. The down conversion unit 10 down-converts the reception signal V _RX from the SAW sensor 2 by the transmission signal V _TX by multiplying the signal amplified by these two amplifiers 8 and 9, and the component of the carrier frequency f _c Cancel each other out. The filter 11 performs a filtering process to pass the component of the modulation frequency f _mod with respect to the signal down-converted by the down conversion unit 10, and unnecessary components (for example, a DC component, a component of the modulation frequency f _{mod , the} modulation frequency Cut the component twice the f _mod , etc.).

The phase comparator 12 outputs a signal obtained by comparing the phase of the output signal of the modulation signal generator 4 and the phase of the output signal of the filter 11 to the filter 13. The phase comparator 12 _cancels the components of the modulation frequency f _mod . The filter 13 passes the DC component, cuts an unnecessary component (a component of twice the modulation frequency), outputs it to the computing unit 14, and feeds back the DC component to the adder 5. The computing unit 14 has a function of controlling the DC voltage of the filter 11 within the voltage range within the locking range of the oscillator 6. The adder 5 adds the DC component corresponding to the phase difference to the output signal of the modulation signal generator 4 and outputs the result as the control signal V _ctrl to the oscillator 6. The oscillator 6 adjusts the carrier frequency f _c according to the control signal V _ctrl and outputs it to the two amplifiers 8 and 9. These series of processes, the carrier frequency f _c of the carrier oscillator 6 outputs is repeated until it converges to the resonance frequency f _r of the SAW sensor 2.

The significance of the above configuration will be described using mathematical formulas. First, the output signal C (t) of the oscillator 6 is defined as in equation (1). Here, f _c represents a carrier frequency, and φ represents an initial phase angle.

このとき、変調信号生成器４の出力信号の変調周波数をｆ_ｍｏｄ、変調指数をｍf、信号波をＳ（ｔ）＝ｃｏｓ（２πｆ_ｍｏｄ・ｔ）とすれば、ＦＭ変調波を下式のように表すことができる。そして、例えば狭帯域ＦＭ変調であると仮定してΔｆを最大周波数偏移としたときのβ＝Δｆ／ｆ_ｍｏｄを１より十分小さい値と仮定し、さらに、初期位相角φを０と仮定し、（１）式を用いて展開すると（２）式のように定義できる。

At this time, assuming that the modulation frequency of the output signal of the modulation signal generator 4 is f _mod , the modulation index is m f, and the signal wave is S (t) = cos (2πf _mod · t), the FM modulation wave is It can be expressed in Then, assuming that .beta. =. DELTA.f / f _mod when .DELTA.f is the maximum frequency shift assuming that it is narrow band FM modulation, for example, it is assumed that the initial phase angle .phi. When expanded using the equation (1), it can be defined as the equation (2).

さらに、β＜＜１であるため（３−１）式及び（３−２）式が成立する。

Furthermore, since β << 1, the equations (3-1) and (3-2) hold true.

このため、ＦＭ変調波を表す（２）式は（４）式のように展開できる。

Therefore, the equation (2) representing the FM modulated wave can be expanded as the equation (4).

すなわち、狭帯域ＦＭ変調波は、（４）式の第１項のキャリア周波数ｆ_ｃの成分（以下、成分Ｓ０と称す）、第２項の（キャリア周波数ｆ_ｃ−変調周波数ｆ_ｍｏｄ）成分（以下、成分Ｓ１と称す）、第３項の（キャリア周波数ｆ_ｃ＋変調周波数ｆ_ｍｏｄ）成分（以下、成分Ｓ２と称す）、の３つの周波数成分によって近似して表すことができる。そこで、ＳＡＷセンサ２に送信するＦＭ変調波の送信信号Ｖ_ＴＸ（＝Ｖ_{ＦＭ＿Ｔｘ}（ｔ））を（５）式のように定義する。ここで、成分Ｓ０の振幅をＡ_ＬＯ０、成分Ｓ１の振幅をＡ_ＬＯ１、成分Ｓ２の振幅をＡ_ＬＯ２とし、成分Ｓ０の初期位相差をθ_ＬＯ０、成分Ｓ１の初期位相差をθ_ＬＯ１、成分Ｓ２の初期位相差をθ_ＬＯ２として一般化している。

That is, the narrow-band FM modulation wave has a component (hereinafter referred to as component S0) of the carrier frequency f _c of the first term of the equation (4) and a (carrier frequency f _c -modulation frequency f _mod ) component of the second term Hereinafter, it can be approximated by three frequency components of component S1) and the third term (carrier frequency f _c + modulation frequency f _mod ) component (hereinafter referred to as component S2). Therefore, the transmission signal V _TX (= V FM — _Tx (t)) of the FM modulation wave to be transmitted to the SAW sensor 2 is defined as shown in equation (5). Here, the amplitude of component S0 is A _LO0 , the amplitude of component S1 is A _LO1 , the amplitude of component S2 is A _LO2 , the initial phase difference of component S0 is θ _LO0 , and the initial phase difference of component S1 is θ _LO1 , component S2 It is a generalization of the initial phase difference as theta _LO2.

また、ＳＡＷセンサ２を通じて受信されるＦＭ変調波の受信信号Ｖ_ＲＸ（＝Ｖ_{ＦＭ＿Ｒｘ}（ｔ））を（６）式のように定義する。ここで、成分Ｓ０の振幅をＡ_ＲＦ０、成分Ｓ１の振幅をＡ_ＲＦ１、成分Ｓ２の振幅をＡ_ＲＦ２とし、成分Ｓ０の位相差をθ_ＲＦ０、成分Ｓ１の位相差をθ_ＲＦ１、成分Ｓ２の位相差をθ_ＲＦ２として一般化している。

Further, the reception signal V _RX (= V FM — _Rx (t)) of the FM modulation wave received through the SAW sensor 2 is defined as in the equation (6). Here, the amplitude of the component S0 is A _RF0 , the amplitude of the component S1 is A _RF1 , the amplitude of the component S2 is A _RF2 , the phase difference of the component S0 is θ _RF0 , the phase difference of the component S1 is θ _RF1 , the position of the component S2 The phase difference is generalized as θ _RF2 .

ダウンコンバージョン部１０は、これらの（５）式の信号と（６）式の信号とをミキシングする（Ｖ_{ＦＭ＿Ｒｘ＿Ｃｏｎｖ}（ｔ）を示す（７−１）式参照）。この結果、周波数を低域周波数側に変換処理できる。途中の計算処理を省略するが、フィルタ１１を通過した後の変調周波数ｆ_ｍｏｄの成分は（７−２）式に示すように表すことができる。

The down conversion unit 10 mixes the signal of the equation (5) and the signal of the equation (6) (see equation (7-1) indicating V _{FM_Rx_Conv} (t)). As a result, the frequency can be converted to the low frequency side. Although calculation processing in the middle is omitted, the component of the modulation frequency f _mod after passing through the filter 11 can be expressed as shown in the equation (7-2).

フィルタ１１は（７−２）式に示される変調周波数ｆ_ｍｏｄの成分を取得して位相比較器１２に出力する。位相比較器１２は、送信用の変調周波数ｆ_ｍｏｄの信号と、この受信信号Ｖ_{ＦＭ＿Ｒｘ＿Ｃｏｎｖ}（ｔ）の変調周波数ｆ_ｍｏｄの成分とを混合することで周波数変換処理する。ここで位相比較器１２に入力される変調周波数ｆ_ｍｏｄの成分は（７−２）式で表される。また、変調周波数ｆ_ｍｏｄの成分はＳ（ｔ）＝ｃｏｓ（２πｆ_ｍｏｄ・ｔ）であるため、位相比較器１２が例えばミキシングしてダウンコンバートすることで下記の（８）式のように表される信号を得る（Ｖ_{ＦＭ＿Ｒｘ＿Ｃｏｎｖ＿Ｃｏｎｖ}（ｔ）参照）。

The filter 11 obtains the component of the modulation frequency f _{mod represented} by the equation (7-2) and outputs the component to the phase comparator 12. Phase comparator 12, a signal of the modulation frequency _{f mod} for transmission, frequency conversion by mixing the components of the modulation frequency _{f mod} of the received signal _{V FM_Rx_Conv} (t). Here, the component of the modulation frequency f _mod input to the phase comparator 12 is expressed by equation (7-2). Further, since the component of the modulation frequency f _mod is S (t) = cos (2πf _mod · t), the phase comparator 12 performs mixing and down conversion, for example, and is expressed as in the following equation (8) Signal (see V _{FM_Rx_Conv_Conv} (t)).

フィルタ１３は、このうちＤＣ成分を通過し変調周波数ｆ_ｍｏｄの２倍（２×ｆ_ｍｏｄ）の成分をカットする。ＤＣ成分は（９）式のように表されるが、第１項＋第２項を（９−１）式とし、第３項＋第４項を（９−２）式とする。

The filter 13 passes the DC component and cuts the component of twice the modulation frequency f _mod (2 × f _mod ). The DC component is expressed as in equation (9), but the first term + the second term is the equation (9-1), and the third term + the fourth term is the equation (9-2).

この（９）式に示されるＤＣ成分が、演算器１４を通じて加算器５にフィードバックされる。（９）式のＤＣ成分では、振幅積（−Ａ_ＲＦ１・Ａ_ＬＯ０）／４と位相差（θ_ＲＦ１−θ_ＬＯ０）の余弦値との乗算値と、振幅積（Ａ_ＲＦ２・Ａ_ＬＯ０）／４と位相差（θ_ＲＦ２−θ_ＬＯ０）の余弦値との乗算値と、を加算した成分を（９−１）式として示している。また、振幅積（−Ａ_ＲＦ０・Ａ_ＬＯ１）／４と位相差（θ_ＲＦ０−θ_ＬＯ１）の余弦値との乗算値と、振幅積（Ａ_ＲＦ０・Ａ_ＬＯ２）／４と位相差（θ_ＲＦ０−θ_ＬＯ２）の余弦値との乗算値と、を加算した成分を（９−２）式として示している。

The DC component shown in the equation (9) is fed back to the adder 5 through the calculator 14. In the DC component of the equation (9), the product of the product of the amplitude product (−A _RF1 · A _LO0 ) / 4 and the phase difference (θ _RF1 −θ _LO0 ) multiplied by the cosine value and the amplitude product (A _RF2 · A _LO0 ) / A component obtained by adding 4 and the _product of the cosine value of the phase difference (θ _RF2 −θ _LO0 ) is added as Expression (9-1). Also, the product of the amplitude product (-A _RF0 · A _LO1 ) / 4 and the cosine value of the phase difference (θ _RF0- θ _LO1 ), the amplitude product (A _RF0 · A _LO2 ) / 4 and the phase difference (θ _RF0 A component obtained by adding the _product of the cosine value of -.theta. _LO2 ) and the product is shown as equation (9-2).

The tendency of the frequency dependency of each element when the equation (9) is decomposed into each element is shown in FIG. 4 and FIG. FIG. 4 shows the tendency of the frequency dependency of the absolute value of the amplitude product of each term, and FIG. 5 shows the frequency dependency of the cosine value of the phase difference. Further, FIG. 6 shows the frequency dependency of the phase difference θ _r between the transmission signal V _TX and the reception signal V _RX . These frequency characteristics indicate frequency characteristics within a predetermined frequency range obtained when an appropriate element (for example, a single element (for example, a capacitor)) or a circuit (for example, a capacitive circuit) is selected as an auxiliary circuit.

Within the frequency range shown in FIG. 4, the amplitude product is a minimum value in the amplitude resonance frequency f _{R_Amp,} monotonically decreases in accordance with the frequency increased in the frequency region lower than the amplitude resonance frequency f _{R_Amp,} than the amplitude resonance frequency f _{R_Amp} In the high frequency region, the characteristic is monotonically increasing according to the frequency increase. As shown in FIG. 5, the cosine value of the phase difference has a value at the phase difference resonance frequency _{fr_p-p} as the minimum value, and monotonously decreases in the frequency region lower than the phase difference resonance frequency _{fr_p-p} according to the frequency increase. In the frequency range higher than the phase difference resonance frequency _{fr_p-p,} the characteristic is monotonically increasing according to the frequency increase.

By selecting an appropriate auxiliary circuit 16, in the frequency range shown in FIG. 6, the phase difference θ _r between the transmission signal V _TX and the reception signal V _RX has a maximum value at the phase resonance frequency f _{r_Phase} . Phase difference theta _r is monotonically increased according to the frequency increase in the frequency region lower than the phase resonance frequency f _{R_Phase,} a monotonically decreasing characteristic according to the frequency increase in the frequency region higher than the phase resonance frequency _f r_Phase.

Here, since the component of the equation (9-1) has a component including the amplitude A _LO0 of the carrier frequency of the transmission signal V _TX as the main component, the component of the equation (9-2) is significantly larger than the component of the equation (9-2). Become. For this reason, below, it demonstrates using the numerical formula of the component of (9-1) Formula.

First, when an appropriate circuit is selected as the auxiliary circuit 16, the cosine value of the phase difference of the first term of the equation (9-1) is changed with the same sign within the locking range where the carrier frequency fc is predetermined. It is desirable to select so that the cosine value of the phase difference of the second term is changed with the same sign. Furthermore, in order to make the transfer characteristic of the cosine value of the phase difference V-shaped according to the frequency change as described later, 0 to 90 [deg], 90 to 180 [deg], -90 to 0 [deg],- It is desirable to suppress the change in the phase difference in the range of 90 to -180 [deg]. Although these ranges may be exceeded, when this range is exceeded, either one or both of the transmission signal V _TX and the reception signal V _RX may be phase-shifted (if necessary, an amplifier, a filter, etc.) It is desirable to shift the phase difference by providing the same and to set the phase difference related to the transmission signal V _TX and the reception signal V _RX in the above-mentioned range.

Now, under such conditions, when the amplitude product satisfies the relationship of the following equation (10-1) and the cosine value of the phase difference satisfies the relationship of the following equation (10-2), (9-1) The negative value of the first term of equation (1) exceeds the positive value of the second term, and the DC component becomes a negative value. At this time, on the characteristics shown in FIG. 4 and FIG. 5, it operates in the region shown by the amplitude product FA1 of FIG. 4 and the cosine value FT1 of FIG. 5 and approximately amplitude resonance frequency _{fr_Amp} , phase difference resonance frequency _{fr_p-p} Operate at lower frequencies. The negative DC component is fed back to the adder 5 through the computing unit 14 so that the carrier frequency fc is adjusted to the high frequency side.

振幅積が下記の（１１−１）式の関係を満たすと共に、位相差の余弦値が下記の（１１−２）式の関係を満たすときには、（９−１）式の第１項の負の値が第２項の正の値を下回ることになり、ＤＣ成分は正の値となる。このとき、図４、図５に示す特性上では、図４のＦＡ２、図５の余弦値ＦＴ２に示す領域で動作し、振幅共振周波数ｆ_{ｒ＿Ａｍｐ}、位相差共振周波数ｆ_{ｒ＿ｐ−ｐ}より高い周波数で動作する。正のＤＣ成分は、演算器１４を通じて加算器５にフィードバックされることになり、キャリア周波数ｆｃが低域側に調整される。

When the amplitude product satisfies the relationship of the following equation (11-1) and the cosine value of the phase difference satisfies the relationship of the following equation (11-2), the negative of the first term of the equation (9-1) The value falls below the positive value of the second term, and the DC component becomes a positive value. At this time, on the characteristics shown in FIG. 4 and FIG. 5, it operates in the area shown by _FA2 in FIG. 4 and the cosine value FT2 in FIG. 5 and at frequencies higher than the amplitude resonance frequency _{fr_Amp} and the phase difference resonance frequency _{fr_p-p.} Operate. The positive DC component is fed back to the adder 5 through the computing unit 14, and the carrier frequency fc is adjusted to the low frequency side.

振幅積が下記の（１２−１）式の関係を満たすと共に、位相差の余弦値が下記の（１２−２）式の関係を満たすときには、（９−１）式の第１項の負の値が第２項の正の値と同等の値となりＤＣ成分は概ね０となる。このとき、図４、図５に示す特性上では、図４の振幅積ＦＡ３、図５の余弦値ＦＴ３に示す領域で動作し、振幅共振周波数ｆ_{ｒ＿Ａｍｐ}、位相差共振周波数ｆ_{ｒ＿ｐ−ｐ}に近接又は同一周波数で動作する。このとき位相差θ_ｒは、位相共振周波数ｆ_{ｒ_Phase}に近い値ＦＨ３となり最大値となる。

When the amplitude product satisfies the relationship of the following equation (12-1) and the cosine value of the phase difference satisfies the relationship of the following equation (12-2), the negative of the first term of the equation (9-1) The value is equal to the positive value of the second term, and the DC component is approximately zero. At this time, on the characteristics shown in FIG. 4 and FIG. 5, it operates in the region shown by the amplitude product FA3 in FIG. 4 and the cosine value FT3 in FIG. 5 and approaches the amplitude resonance frequency _{fr_Amp} and the phase difference resonance frequency _{fr_p-p} . Or operate at the same frequency. At this time, the phase difference θ _r has a value FH3 close to the phase resonance frequency f _{r_Phase and} has a maximum value.

発振器６の初期発振周波数を、位相共振周波数ｆ_{ｒ_Phase}にロック可能に予め定められたロッキングレンジ内に設定することで、自動的にフィードバック制御して送信信号Ｖ_ＴＸと受信信号Ｖ_ＲＸの位相差θ_ｒを最大値とする位相共振周波数ｆ_{ｒ_Phase}に収束させることができる。

The initial oscillating frequency of the oscillator 6, by setting the phase resonance frequency f _{R_Phase} to lockably predetermined locking-range, automatically the phase difference of the feedback control to transmit the signal V _TX and the reception signal V _RX theta It can be made to converge to phase resonance frequency _{fr_Phase} which makes _r a maximum value.

At this time, the Q value can be increased by increasing the phase difference θ _r as much as possible, and the accuracy can be improved. On the other hand, in order to widen the locking range, it is desirable to reduce the Q value. Although these are in a trade-off relationship, when the circuit configuration of this embodiment is adopted, these can be adjusted as freely as possible, and the degree of freedom in design can be improved.

FIG. 7 is a view schematically showing frequency characteristics FX1 to FX3 of DC component output when the inventors simulate using a capacitive element as an auxiliary circuit. FIG. 7 shows a simulation result when the impedance Z _s of the SAW sensor 2 is inductive at a predetermined frequency (several hundreds of MHz).

  The inventors have confirmed that, by adjusting the capacitance value of the capacitive element forming the auxiliary circuit 16, the gradient change of the DC component output with respect to the carrier frequency fc can be made possible. That is, in order to widen the locking range, it is preferable to reduce the capacitance value (capacitance value C3) as shown in the characteristic FX1, and to increase the accuracy even if the locking range is narrowed, the capacitance value as shown in the characteristic FX3. It is desirable to increase (capacitance value C1).

<Description of Conventional Comparative Example>
Conventionally, it has been confirmed by the inventors that when configured as in Non-Patent Document 1, characteristic changes corresponding to frequency changes as shown in FIG. 8 can be obtained. That is, when the configuration shown in Non-Patent Document 1 is used, indices corresponding to the above-described amplitude product and phase difference will coincide at one resonance frequency f _r , and parameters for adjusting this resonance frequency f _r For example, even if the resistance value of the resistance between the transmitting node and the receiving node is adjusted, its Q value can only be adjusted, and the gradient (sensitivity) to the carrier frequency fc is set as shown in FIG. It has proved difficult to adjust.

<Summary of this embodiment>
According to this embodiment, the auxiliary circuit 16 is connected between the first node N1 and the second node N2. This auxiliary circuit 16, because it is provided to indicate the above-mentioned characteristics, the frequency can be accurately detected by converging the carrier frequency f _c to the phase resonance frequency _f r_phase. Moreover, the degree of freedom in design can be improved.

As shown in FIG. 9 corresponding to FIG. 8, the characteristics of the present embodiment are related to resonance because they have resonance relationships at an amplitude resonance frequency f _{r — Amp} and a phase resonance frequency f _{r —} phase with different amplitude products and phase differences. Accordingly, as shown in FIG. 7, the slope (sensitivity) with respect to the carrier frequency fc can be easily adjusted.

That is, when the auxiliary circuit 16 is, for example, a capacitive circuit, when the capacitance value is relatively small (capacitance value C3), the frequency difference between the amplitude resonance frequency _{fr_Amp} and the phase resonance frequency _{fr_phase} (| _{fr_phase −fr_Amp} It has been confirmed that |) becomes relatively large, and as shown by the characteristic FX1 in FIG. 7, the frequency change gradient of the output of the filter 13 can be made gentle, and a relatively wide locking range can be ensured. It becomes possible.

In addition, when the capacitance value of the capacitive circuit constituting the auxiliary circuit 16 is relatively large (capacitance value C1), the frequency difference (| _{fr_phase− fr_Amp} |) between the amplitude resonance frequency _{fr_Amp} and the phase resonance frequency _{fr_phase} is It has been confirmed that it becomes relatively small, and the frequency change gradient of the filter output can be made steep as shown by the characteristic FX3 in FIG. 7, and the detection accuracy of the resonance frequency can be enhanced. As a result, the accuracy in detection and the locking range can be adjusted in a trade-off relationship, and the degree of freedom in design can be increased. The auxiliary circuit 16 is illustrated as an example in FIG. 2 as a single element 16z (for example, a single capacitor element) as an example, but is not limited thereto.

In addition, the rocking range can be made wide by operating the phase resonance frequency _{fr_p-p} with different amplitude resonance frequency _{fr_Amp} .
The oscillator 6 of the transmitter 7 transmits the narrowband frequency-modulated signal to the carrier to the SAW sensor 2, and the receiver 15 transmits the transmission signal V _TX of the transmitter 7 and the received signal V _RX from the SAW sensor 2. The DC component (phase difference component between the output of the filter 11 and the output of the modulation signal generator 4 input to the phase comparator 12) according to the phase difference θ _r is fed back and superimposed on the modulation signal of the modulation frequency f _mod The oscillator 6 of the transmitter 7 oscillates a signal using a control signal _Vctrl as a signal in which a DC component (phase difference component) is superimposed on the modulation signal, and causes the carrier frequency f _c to converge on the resonance frequency f _r . Accordingly, the resonance frequency f _r of the SAW sensor 2 accurately be detected, it can provide a resonant frequency detection apparatus 3 of highly practical SAW sensor 2.

Second Embodiment
10 and 11 show an additional explanatory view of the second embodiment. In the present embodiment, an embodiment in which two SAW elements 119a and 119b (see FIG. 11) are provided as the SAW sensor 102 is shown. The SAW sensor 102 changes its propagation characteristics depending on the temperature change. In order to cancel this temperature dependent characteristic, it is desirable to use two or three or more SAW elements 119a, 119b as the SAW sensor 102 so that two physical quantities (strain, temperature) can be detected.

Here, in FIG. 10, when two SAW elements 119a and 119b are used, the SAW sensor 102 is described as “SAW1 + SAW2”. As shown in FIG. 11, the SAW sensor 102 is configured by connecting a plurality of SAW elements 119a and 119b in parallel. The SAW elements 119a and 119b are formed of resonant SAW elements having different resonance frequencies f _r1 and f _r2 , respectively. The SAW element 119a is a strip-like reflection formed on the piezoelectric substrate (without the reference numeral), with the interdigital electrode 20a and the same electrode film as the interdigital electrode 20a located on both sides of the interdigital electrode 20a. Device 21a are arranged. It propagates a transmission signal _{V TX} input through the antenna 17, 18 to the substrate of the SAW device 119a, and outputs to the side of the input node N102 of the amplifier 109a again through the antenna 18, 17. The SAW element 119b is also in the form of a strip formed on the piezoelectric substrate (without reference numeral), with the interdigital electrode 20b and the same electrode film as the interdigital electrode 20b located on both sides of the interdigital electrode 20b. The reflector 21b is arranged. It propagates a transmission signal _{V TX} input through the antenna 17, 18 to the substrate of the SAW device 119b, and outputs to the side of the input node N102 of the amplifier 109a again through the antenna 18, 17.

  The difference between the resonant frequency detection device 103 of the sensing system 101 of the second embodiment shown in FIG. 10 and the resonant frequency detection device 3 of the sensing system 1 of the first embodiment is the adder 5 and the oscillator 6 of the first embodiment. (VCO 6z), two amplifiers 8, 9, down-conversion unit 10 (mixer 10z), filter 11 (BPF 11z), phase comparator 12 (mixer 12z), filter 13 (LPF 13z), arithmetic unit 14 and respective SAW elements In a form suitable for 119a and 119b, two sets of two sets are provided. Hereinafter, a first transmitter 107a and a first receiver 115a configured to correspond to the first system SAW element 119a, and a second transmitter 107b and a second configured to correspond to the second system SAW element 119b. The receiver 115 b will be described separately.

The first transmitter 107a includes a modulation signal generator 4 that generates a modulation signal of the modulation frequency f _mod , an adder 105a, and an oscillator 106a. The second transmitter 107b includes the modulation signal generator 4, an adder 105b, and an oscillator 106b. The oscillators 106a and 106b are composed of, for example, a voltage controlled oscillator (VCO) 6z.

  The first receiver 115a is configured by connecting two amplifiers 108a and 109a, a down conversion unit 110a, a filter 111a, a phase comparator 112a, a filter 113a, and an arithmetic unit 14a. The second receiver 115b is also configured by connecting two amplifiers 108b and 109b, a down conversion unit 110b, a filter 111b, a phase comparator 112b, a filter 113b, and an arithmetic unit 14b. The down conversion units 110a and 110b are each configured of, for example, a mixer 10z. Each of the phase comparators 112a and 112b includes, for example, a mixer 12z. The filters 111a and 111b are each configured of, for example, a band pass filter (BPF) 11z. The filters 113a and 113b are each configured of, for example, a low pass filter (LPF) 13z.

The modulation signal of the modulation frequency f _mod generated by the modulation signal generator 4 is input to the oscillators 106a and 106b through the adders 105a and 105b, respectively. The adder 105a superimposes the modulation signal (for example, AC voltage) of the modulation signal generator 4 and the signal (for example, DC voltage) output from the filter 113a through the computing device 14a, and outputs it as the control signal _Vctrl1 to the oscillator 106a. . The oscillator 106a receives the signal input from the adder 105a as the control signal V _ctrl1 , controls the carrier (carrier frequency f _c1 ), and outputs the narrow band frequency-modulated signal after the control to the node N101 a. The node N101a is connected to the input of the amplifier 108a.

On the other hand, the adder 105b also superimposes the modulation signal (for example, AC voltage) of the modulation signal generator 4 and the signal (for example, DC voltage) output from the filter 113b through the computing unit 14b as a control signal _Vctrl2 on the oscillator 106b. Output. The oscillator 106 b receives the signal input from the adder 105 b as the control signal V _ctrl2 , controls the carrier (carrier frequency f _c2 ), and outputs the narrow band frequency-modulated signal after the control to the node N 101 b. The node N101b is connected to the input of the amplifier 108b.

In the present embodiment, an adder 122 is further provided to add the output signals of the oscillators 106a and 106b. The adder 122 adds the output signals of the two oscillators 106 a and 106 b and applies the result to the SAW sensor 102 through the auxiliary circuit 116. Auxiliary circuit 116 is connected between the input node N101a of amplifier 108a as the output node of the oscillator 106a, an input node N102 of the amplifier 109a of the received signal _{V RX.} The auxiliary circuit 116 is connected between an input node N101b of an amplifier 108b serving as an output node of the oscillator 106b and an input node N102 of the amplifier 109b.

Only one auxiliary circuit 116 is provided in two systems of the transmitting and receiving unit, and is configured to have the same characteristic as that of the first embodiment. The amplifiers 108a, 108b, 109a and 109b for amplification of the transmission signal V _TX and the reception signal V _RX have input impedances equal to the characteristic impedance (for example, 50 Ω) of the transmission line transmitting the output signals of the oscillators 106a and 106b, respectively. It is set to the level. These amplifiers 108a, 108b, 109a and 109b may not be provided as long as gain can be secured.

In the SAW sensor 102, the signals output from the two oscillators 106a and 106b are added by the adder 122 and input. _Therefore , the signal in which the carriers of the carrier frequencies f _c1 and f _c2 are narrow band frequency modulated is input. Therefore, in the reception signal V _RX that the receivers 115a and 115b input to the respective amplifiers 108a, 108b, 109a and 109b through the SAW sensor 102, the frequency components are f _c1 , f _c2 , f _c1 −f _mod and f _c1 + f _mod , There are approximately six types of f _c2 −f _mod and f _c2 + f _mod . Here, since the absolute value of f _c2 -f _c1 is much larger than f _mod , two modulated signals of the carrier frequencies f _c1 and f _c2 do not interfere with each other.

Further, the amplification of the amplifier 108a of the transmission signal _{V TX} and the reception signal _{V RX,} 108b, 109a, 109b is preset amplifiers in a predetermined ratio to a predetermined identical amplification degree or the amplification degree of each other. Amplifier 108a, is 108b amplifies the transmit signal _{V TX,} amplifiers 109a, 109b amplifies the received signal _{V RX} input through SAW sensor 102, and outputs the amplified signal downconversion unit 110a, to 110b.

Downconversion unit 110a, two amplifiers 108a, by mixing the amplified signal by 109a, down-conversion processing by a transmission signal _{V TX} a received signal _{V RX,} combining cancel the components of the carrier frequency _{f c1.} Similarly, the down conversion unit 110 b performs a down conversion process on the reception signal V _RX with the transmission signal V _TX by mixing the signals amplified by the two amplifiers 108 b and 109 b, and cancels the component of the carrier frequency _fc2. . As a result, the frequency components of the output signals of these down conversion units 110 a and 110 b are mainly composed of the component of the modulation frequency f _mod .

Filter 111a is a filter for passing the component of the modulation frequency _{f mod,} cuts unnecessary component (DC component, the modulation frequency _{f mod} components, twice the components such as the modulation frequency _{f mod).} Similarly, the filter 111b is a filter for passing the component of the modulation frequency _{f mod,} unnecessary components to cut (DC component, the modulation frequency _{f mod} components, twice the components such as the modulation frequency _{f mod).}

The phase comparator 112a compares the phases of the output signal of the modulation signal generator 4 and the output signal of the filter 111a, and outputs a phase comparison signal to the filter 113a. The phase comparator 112 a cancels components of the modulation frequency f _mod . The phase comparator 112b also compares the phases of the output signal of the modulation signal generator 4 and the output signal of the filter 111b, and outputs a phase comparison signal to the filter 113b. The phase comparator 112b cancels components of the modulation frequency f _mod .

The filter 113a passes the DC component, cuts unnecessary components (such as a component twice the modulation frequency f _mod ), outputs the cut component to the computing unit 14a, and feeds it back to the adder 105a. The filter 113b passes the DC component, cuts unnecessary components (such as a component twice the modulation frequency f _mod ), outputs the cut component to the computing unit 14b, and feeds it back to the adder 105b.

The adder 105 a adds the phase difference component corresponding to the phase difference to the output signal of the modulation signal generator 4 and outputs the _result as the control signal V _{ctrl 1} to the oscillator 106 a. Oscillator 106a adjusts the carrier frequency _{f c1} and outputs it to the amplifier 108a in accordance with the control signal _{V ctrl1.} The adder 105 b adds the phase difference component corresponding to the phase difference to the output signal of the modulation signal generator 4 and outputs the result as the control signal _{Vctrl 2} to the oscillator 106 b. Oscillator 106b adjusts the carrier frequency _{f c2} and outputs to the amplifier 108b in response to the control signal _{V ctrl2.}

These series of processes, with the carrier frequency _{f c1} of the carrier oscillator 106a outputs are repeated until it converges to the resonant frequency _{f r1} of the SAW device 119a constituting the SAW sensor 102, the carrier frequency of the carrier oscillator 106b outputs The process is repeated until f _c2 converges to the resonance frequency f _r2 of the SAW element 119 _b constituting the SAW sensor 102.

As a result, the receiver 115a of the first system can detect phase information according to the transmission / reception propagation characteristics of the SAW element 119a of the first system, and can adjust the carrier frequency _fc1 accordingly. Also, the receiver 115b of the second system can detect phase information according to the transmission / reception propagation characteristics of the SAW element 119b of the second system, and can adjust the carrier frequency _fc2 accordingly. Thus, the carrier frequencies f _c1 and f _c2 can be independently adjusted to the resonance frequencies f _r1 and f _r2 .

According to the present embodiment, even when two SAW elements 119a and 119b are used, the carrier frequencies f _c1 and f _c2 can be adjusted independently to the resonance frequencies f _r1 and f _r2 , respectively. As a result, distortion and temperature are obtained. The parameters of two physical quantities related to etc. can be derived, and two physical quantities (correlation, etc.) such as temperature dependency of strain can be determined. Also in the second embodiment, the auxiliary circuit 116 may be separately provided between the transmitter 107a and the receiver 115a, and between the transmitter 107b and the receiver 115b, as in the fourth embodiment described later.

Third Embodiment
12 and 13 show an additional explanatory view of the third embodiment. The third embodiment is characterized in that an envelope detection unit 210 is provided as the down conversion unit 10. About the part provided with the same function as description of 1st Embodiment, the same code is attached and description is abbreviate | omitted.

As shown in FIG. 12, the resonant frequency detection device 203 of the sensing system 201 includes a transmitter 7 and a receiver 215. The receiver 215 according to the present embodiment is configured by cascading an envelope detection unit 210, a filter 11, a phase comparison unit 12, a filter 13, and an arithmetic unit 14. The oscillator 6 of the transmitter 7 inputs the modulation signal generated by the modulation signal generator 4 through the adder 5, and inputs the signal input from the adder 5 as a control signal _Vctrl to control the carrier, The oscillation signal oscillated according to the narrow band frequency modulated signal which is a signal after control is output as the transmission signal V _TX to the node N1. The auxiliary circuit 16 is connected between the output node N1 of the oscillator 6 (the output node of the transmitter 7) and the input node N2 of the envelope detector 210 (the input node of the receiver 215).

The transmission signal V _TX is output to the SAW sensor 2 through the auxiliary circuit 16. The receiver 215 inputs the signal received through the SAW sensor 2 as a received signal V _RX . The envelope detection unit 210 performs envelope detection on the reception signal V _RX received from the SAW sensor 2. FIG. 13 shows an example of the electrical configuration of the envelope detection unit 210. As shown in FIG. The envelope detection unit 210 is configured using, for example, a diode D, a resistor R, and a capacitor C, and is configured by serially connecting the diode D and a parallel circuit of the resistor R and the capacitor C. The signal voltage input to the envelope detection unit 210 is rectified by the diode D and the resistor R, and can cut high frequency components according to the function of the capacitor C. The envelope can be detected directly by adjusting each parameter of each element D, R, C. This envelope signal (voltage) can be a signal whose main component is the modulation frequency f _mod .

The filter 11 shown in FIG. 12 passes the modulation frequency f _mod for the signal detected by the envelope detection unit 210 and cuts unnecessary components. The phase comparator 12 mixes (mixes) the signal after passing through the filter 11 and the modulation signal from the modulation signal generator 4 and outputs the signal after mixing (mixing) to the filter 13 as a phase difference component. The filter 13 passes the DC component (phase difference component) of the signal subjected to the phase comparison by the phase comparator 12 and feeds it back to the adder 5 through the calculator 14. The processing of the adder 5 is the same as the operation of the above-described embodiment, and hence the description thereof is omitted. Then, the carrier frequency f _c of the oscillator 6 can be controlled to the resonance frequency f _r . As shown in the present embodiment, even if the configuration in which the envelope is detected by the envelope detection unit 210 is adopted, the same effects as those of the above-described embodiment can be obtained.

Fourth Embodiment
FIG. 14 shows an additional explanatory view of the fourth embodiment. In the fourth embodiment, two SAW sensors 319a and 319b are provided. Further, transmitters 307a and 307b and receivers 315a and 315b are provided to correspond to the SAW sensors 319a and 319b. The SAW sensor 319a is a SAW sensor provided with a SAW element 119a, and the SAW sensor 319b is a SAW sensor provided with a SAW element 119b. The sensing system 301 includes a resonant frequency detection device 303. The resonant frequency detection device 303 includes transmitters 307a and 307b and receivers 315a and 315b.

  Two transmitters 307a, 307b and receivers 315a, 315b are provided. The transmitter 307a includes the modulation signal generator 4, the adder 305a, and the oscillator 306a, and the transmitter 308b includes the modulation signal generator 4, the adder 305b, and the oscillator 306b. The receiver 315a is configured by cascading an envelope detection unit 310a, a filter 311a, a phase comparator 312a, a filter 313a, and an operator 314a. The receiver 315b is configured by connecting an envelope detection unit 310b, a filter 311b, a phase comparator 312b, a filter 313b, and an arithmetic unit 314b in cascade. Each configuration of the transmitters 307a and 307b and the receivers 315a and 315b has the same function as the corresponding configuration described in the above embodiment (the configuration in which the tens and ones place the same reference numerals). , I omit the individual functional description.

  An auxiliary circuit 316a is provided between the output node N301a of the transmitter 307a and an input node N302a of the receiver 315a, and an auxiliary circuit 316b is provided between the output node N301b of the transmitter 307b and the input node N302b of the receiver 315b. Provided between the Also in this case, the phase comparators 312a and 312b (for example, mixers) of the receivers 315a and 315b are the output signals of the filters 311a and 311b and the modulation signal of the modulation signal generator 4 as in the contents described in the above embodiment. And to obtain a phase difference component and feed back it, the same operation and effect as those of the above-described embodiment can be obtained. Even in such a mode, the same effect as that of the above-described embodiment can be obtained.

Fifth Embodiment
FIG. 15 shows an additional explanatory view of the fifth embodiment. In the present embodiment, an oscillator (corresponding to a second oscillator) 423 is separately provided, and the oscillation frequency is changed based on the DC component after the oscillator 423 passes through the filter 13 and the calculator 14. 10 shows a form of acquiring the component of the modulation frequency f _mod by 10 (for example, the mixer 10z). FIG. 15 is a configuration diagram shown instead of FIG. 1, and parts different from FIG. 1 will be described.

  As shown in FIG. 15, the sensing system 401 includes a resonant frequency detection device 403, and the resonant frequency detection device 403 includes a transmitter 7 and a receiver 415. The receiver 415 is configured by connecting two amplifiers 8 and 9, the down conversion unit 10, the filter 11, the phase comparator 12, the filter 13, and the computing unit 14 in cascade, and further includes an oscillator 423.

The input node N401 of the amplifier 8 is not connected to the output node N1 of the oscillator 6. The output signals of the filter 13 and the calculator 14 of the receiver 415 are input to the adder 5 and to another oscillator 423. That is, the oscillator 423 inputs an oscillation signal of so-called one tone carrier frequency f _cr to the amplifier 8 based on the output signal (for example, DC voltage) of the filter 13 and the computing unit 14.

While the oscillator 423 is adjusting the carrier frequency f _cr as the second frequency, the adder 5 outputs the modulation signal of the modulation signal generator 4 (for example, the AC voltage of the modulation frequency f _mod ) and the output signal of the arithmetic unit 14 (For example, DC voltage) is added and output to the oscillator 6. The oscillator 6 receives the output signal (for example, DC voltage) of the filter 13 and the integrator 14 and oscillates. Oscillator 6 and 423 have the same control signal V _ctrl dependent frequency characteristics of the oscillation signal. Therefore, by down-conversion unit 10 mixes the output signal of the reception signal _{V RX} and the oscillator 423 from SAW sensor 2 (mixing), converging the carrier frequency _{f c} of the oscillation signal of the oscillator 6 and 423 _{f cr} The component of the modulation frequency f _mod can be obtained by the down conversion processing by the down conversion unit 10.

The filter 11 passes the component of the modulation frequency f _mod and cuts other unnecessary components, and the phase comparison unit 12 compares the phase of the output signal of the filter 11 with the modulation signal by the modulation signal generator 4, The DC component of the comparison result is fed back to the oscillator 6 from the filter 13 through the computing unit 14. Thus, it is possible to control the carrier frequency _{f c} of the signal oscillator 6 outputs the resonance frequency _{f r} of the SAW sensor 2. Also in such a configuration, the same effects as those of the above-described embodiment can be obtained.

Sixth Embodiment
FIG. 16 shows an additional explanatory view of the sixth embodiment. The sixth embodiment differs from the second embodiment in that, as shown in the fifth embodiment, a signal in which the oscillation frequency is changed based on an output signal (DC component) after passing through a filter and an arithmetic unit The component of the modulation frequency f _mod is acquired by the down conversion unit (for example, a mixer) using this. FIG. 16 is a block diagram instead of FIG. 10 of the second embodiment.

  As shown in FIG. 16, the sensing system 501 includes a resonant frequency detector 503, and the resonant frequency detector 503 includes transmitters 507a and 507b and receivers 515a and 515b. The transmitter 507a includes a modulation signal generator 4, an adder 505a, and an oscillator 506a, and the transmitter 507b includes the modulation signal generator 4, an adder 505b, and an oscillator 506b.

  The receiver 515a includes amplifiers 508a and 509a, a down conversion unit 510a, a filter 511a, a phase comparator 512a, a filter 513a, and an arithmetic unit 514a, and further includes an oscillator (corresponding to a second oscillator) 523a. The receiver 515b includes amplifiers 508b and 509b, a down conversion unit 510b, a filter 511b, a phase comparator 512b, a filter 513b, and an integrator 514b, and further includes an oscillator (corresponding to a second oscillator) 523b. The configuration of each of the transmitters 507a and 507b and the receivers 515a and 515b has the same function as the corresponding configuration described in the above embodiment (the configuration in which the tens and ones places the same symbol). , I omit the individual functional description. The configuration of the SAW sensor 502 is the same as that of the SAW sensor 102, and is configured by connecting a plurality of SAW elements 119a and 119b. An auxiliary circuit 516 is provided between the output node N501a of the transmitter 507a and the input node N502 of the receiver 515a, and between the output node N501b of the transmitter 507b and the input node N502 of the receiver 515b. ing. The auxiliary circuit 516 is shared between each transmitter 507 a and the receiver 515 a and between the transmitter 507 b and the receiver 515 b.

As shown in FIG. 16, the input node N501c of the amplifier 508a is not connected to the output node N501a of the oscillator 506a, and the input node N501d of the amplifier 508b is not connected to the output node N501b of the oscillator 506b. The output signal (for example, DC voltage) of the filter 513a of the receiver 515a and the integrator 514a is input to the adder 505a, and is also input to the separately provided oscillator 523a. Oscillator 523a controls the carrier frequency _{f c} on the basis of the signal output through the operation unit 514a from the filter 513a, the oscillation signal, which is inputted to the amplifier 508a through node N501c.

The output signals of the filter 513b of the receiver 515b and the computing unit 514b are input to the adder 505b and to an oscillator 523b provided separately. Oscillator 523b controls a carrier frequency _{f c} on the basis of the output signal of the filter 513b and calculator 514b, the oscillation signal is inputted to the amplifier 508b through the node N501d.

The oscillators 506a and 523a have, for example, the same control signal V _ctrl1 dependency on the frequency characteristics of their oscillation signals. Similarly, the oscillators 506b and 523b have the same control signal _Vctrl2 dependency, for example, on the frequency characteristics of the oscillation signal.

Therefore, the components of the modulation frequency f _mod can be acquired by mixing (mixing) the reception signal V _RX from the SAW sensor 502 with the output signals of the oscillators 523 a and 523 b by the down conversion units 510 a and 510 b. The filters 511a and 511b pass the component of the modulation frequency f _mod and cut other unnecessary components, and the phase comparators 512a and 512b output the output signals of these filters 511a and 511b and the modulation signal by the modulation signal generator 4 And DC components as the phase comparison signal from the filters 513a, 513b through the integrators 514a, 514b are fed back to the oscillators 506a, 506b, 523a, 523b. As a result, the carrier frequencies f _c1 and f _c2 of the signals output from the oscillators 506 a and 506 _b can be controlled to the resonance frequency f _r of each of the SAW elements 119 a and 119 _b of the SAW sensor 502. Also in such a configuration, the same effects as those of the above-described embodiment can be obtained.

Seventh Embodiment
FIG. 17 shows an additional explanatory view of the seventh embodiment. The sensing system 601 comprises a resonant frequency detector 603. The resonant frequency detection device 603 of the sensing system 601 includes a transmitter 7 including a modulation signal generator (corresponding to a modulation signal generation unit) 4, an adder 5, and an oscillator 6, amplifiers 8 and 9, a down conversion unit 10, and a filter 11. And a receiver 15 including a phase comparator (corresponding to a phase comparator) 12, a filter 13, and an operator.

  Also, the resonance frequency detection device 603 is configured by connecting an auxiliary circuit 616 between the transmitter 7 and the receiver 15. The auxiliary circuit 616 is configured by connecting in parallel a circuit in which a plurality of capacitors 16za, 16zb, 16zc and a plurality of switches Sa, Sb, Sc are respectively connected in series.

The resonance frequency detection device 603 further includes a monitor circuit 624 that monitors the output signal of the filter 13, and a control circuit 625. The control circuit 625 can perform on / off control of the switches Sa, Sb, and Sb, and performs on / off control of the plurality of switches Sa, Sb, and Sc of the auxiliary circuit 616 based on the monitoring result of the monitor circuit 624. Thus, the auxiliary circuit 616 can control the impedance. Further, the control circuit 625 is in a changeable carrier frequency f _c of the addition and subtraction control the control voltage Vctrl by outputting a control signal (e.g., DC voltage) to the adder 5 oscillator 6.

Sensing system 601 before starting the normal operation, the control circuit 625 switches Sa, Sb, changes and controls the carrier frequency f _c of the oscillator 6 with on-off controlled by changing the impedance of the auxiliary circuit 616 Sc. At this time, the monitor circuit 624 monitors the output value (corresponding to the phase difference component) of the filter 13. At this time, the control circuit 625 compares the specification of the predetermined locking range with the monitored actual locking range. The control circuit 625 compares the locking ranges by comparing the difference between the frequencies corresponding to the maximum value and the minimum value of the output value of the filter 13. Then, the control circuit 625 compares the specification of the locking range with the actual locking range to determine the impedance of the auxiliary circuit 616 in the closest condition, and determines the on / off state of the switches Sa, Sb, and Sc.

Then, the control circuit 625 operates the sensing system 601 normally with the impedance of the auxiliary circuit 616 fixed. Through such processing, the carrier frequency f _c can be converged to the resonance frequency f _r . As a result, the same effects as those of the above-described embodiment can be obtained.

  According to this embodiment, the optimum impedance of the auxiliary circuit 616 can be determined in accordance with the specification of the locking range. The configuration of the present embodiment is also applicable to, for example, a configuration using two sensor elements 102. Also, the contents of the above-described embodiments can be combined and applied.

In the drawing, 4 is a modulation signal generator (modulation signal generation unit), 6, 106a, 106b, 306a, 306b, 506a, 506b is an oscillator, 6z is a VCO (oscillator) 7, 107a, 107b, 307a, 307b, 507a. , 507b is a transmitter, 15, 115a, 115b, 215, 315a, 315b, 415, 515a, 515b is a receiver, 16, 116, 316a, 316b, 516, 616 is an auxiliary circuit, 16z, 16za, 16zb, 16zc is Auxiliary elements (auxiliary circuits) 19, 119a, 119b are SAW elements, 624 is a monitor circuit, 625 is a control circuit, N1, N101a, N101b, N301a, N301b, N501a, N501b are first nodes, N2, N102, N302a, N 302 b and N 502 are second nodes, f _c and f _c1 , F _c2 indicate the carrier frequency, and f _r , f _r1 and f _r2 indicate the resonant frequency of the SAW sensor.

Claims (10)

Modulation signal generator for outputting a modulated signal (4), and, the modulation signal control signal; oscillator narrowband frequency modulated signal to the carrier in response to an input to the control signal as _{(V ctrl V ctrl1, V ctrl2} ) 106a, 106b; 306a, 306b; 506a, 506b), and the narrow band frequency-modulated signal is provided in a resonant type SAW element (19; 119a, 119b). 102; 319a, 319b; 502) transmitters (7; 107a, 107b; 307a, 307b; 507a, 507b), and
A receiver (15; 115a, 115b; 215; 315a, 315b; 415; 515a, 515b) that receives the transmission signal transmitted to the SAW sensor by the transmitter via the SAW element of the SAW sensor;
The first node (N1; N101a, N101b; N301a, N301b; N501a, N501b) from which the transmitter transmits transmission signals, and the second node (N2; N102; N302a, N302b; N502) from which the receiver receives reception signals. The frequency characteristic of the amplitude product of the transmission signal and the reception signal when the transmission signal of the transmitter and the reception signal of the receiver are mixed is the minimum value at the amplitude resonance frequency ( _{fr_Amp} ), and the amplitude A characteristic that monotonically decreases in response to an increase in frequency in a frequency range lower than the resonance frequency and monotonously increases in response to an increase in frequency in a frequency range higher than the amplitude resonance frequency, and Maximum value at the phase difference resonance frequency ( _frp-p ) and lower than the phase difference resonance frequency An auxiliary circuit (16; 16z; 16za, 16zb,) configured to have characteristics of monotonously increasing in response to frequency increase and decreasing monotonically in response to frequency increase in a frequency region higher than the phase difference resonance frequency. 16zc; 116; 316a, 316b; 516; 616);
The receiver receives a transmission signal transmitted to the SAW sensor by the transmitter via the auxiliary circuit and the SAW sensor, and converts the reception signal from the SAW sensor to obtain the modulation signal. The phase difference component according to the phase difference between the frequency component and the modulation signal of the modulation signal generation unit is fed back to the transmitter and superimposed on the modulation signal of the modulation signal generation unit of the transmitter.
The oscillator of the transmitter oscillates a signal using the signal in which the phase difference component is superimposed on the modulated signal as the control signal to make the carrier frequency (f _c ; f _c1 , f _c2 ) the resonance frequency of the SAW sensor A resonant frequency detection device of a SAW sensor which converges to (f _r ; f _r1 , f _r2 ). In the resonance frequency detection device of the SAW sensor according to claim 1,
A resonant frequency detection device of a SAW sensor, wherein the auxiliary circuit is _{configured to operate the} amplitude resonant frequency ( _{fr_Amp} ) at a frequency different from the phase difference resonant frequency ( _{fr_p-p} ). In the resonance frequency detection device of the SAW sensor according to claim 1,
The auxiliary circuits (616, 16za, 16zb, 16zc) are configured to be able to change the impedance,
A control circuit (625) for changing and controlling the frequency of the carrier while changing the impedance of the auxiliary circuit before normal operation;
A monitor circuit (624) for monitoring a phase difference component by the receiver in a state of being change-controlled by the control circuit;
The resonance frequency detection device for a SAW sensor, wherein the control circuit is operated normally after the impedance of the auxiliary circuit is determined according to the monitor result by the monitor circuit. The resonance frequency detection apparatus for a SAW sensor according to any one of claims 1 to 3.
The receivers (15; 115a, 115b)
A down conversion unit (10; 10z; 110a, 110b) for down converting the reception signal from the SAW sensor by the transmission signal;
A filter (11: 11z; 111a, 111b) for passing frequency components of the modulated signal by filtering the signal downconverted by the downconversion unit;
A phase comparison unit (12; 12z; 112a, 112b) for detecting the phase difference component by phase comparison of the frequency component of the modulation signal that has passed through the filter and the modulation signal of the modulation signal generator of the transmitter; The resonance frequency detection apparatus of a SAW sensor provided. The resonance frequency detection apparatus for a SAW sensor according to any one of claims 1 to 3.
The receiver (215; 315a, 315b)
An envelope detection unit (210; 310a, 310b) for performing envelope detection on a received signal from the SAW sensor;
A filter (11; 311a, 311b) that passes frequency components of the modulated signal by filtering the signal whose envelope has been detected by the envelope detection unit;
A phase comparison unit (12; 312a, 312b) for detecting the phase difference component by phase comparison of the modulation frequency component of the modulation signal acquired by the envelope detection unit and the modulation signal by the modulation signal generator of the transmitter And a resonant frequency detector for a SAW sensor. The resonance frequency detection apparatus for a SAW sensor according to any one of claims 1 to 3.
The receiver (415; 515a, 515b) further comprises a second oscillator (423; 523a, 523b),
A down-conversion unit (10; 10z; 510a, 510b) for down-converting the reception signal from the SAW sensor by the second frequency of the second oscillator;
A filter (11: 11z; 511a, 511b) for passing the frequency component of the modulated signal by filtering the signal downconverted by the downconversion unit;
A phase comparison unit (12; 12z; 512a, 512b) for detecting the phase difference component by phase comparison of the frequency component of the modulation signal that has passed through the filter and the modulation signal by the modulation signal generator of the transmitter; Equipped
The phase difference component of the phase comparison unit is fed back to the transmitter and superimposed on the modulation signal of the modulation signal generation unit, and is also fed back to the control signal of the second oscillator to cause the second frequency to be the carrier of the transmission signal. A resonant frequency detection device of a SAW sensor which converges to a frequency (f _c ; f _c1 , f _c2 ). In the SAW sensor resonance frequency detection device according to claim 4 or 6,
The resonance frequency detection device of a SAW sensor, wherein the down conversion units (10; 10z; 110a, 110b; 510a, 510b) include a mixer (10z).   The resonance frequency detection device of a SAW sensor according to any one of claims 4 to 6, wherein the phase comparison unit (12; 12z; 112a, 112b; 312a, 312b; 512a, 512b) is a mixer (12z). The resonance frequency detection apparatus of a SAW sensor provided. 9. The resonant frequency detection device for a SAW sensor according to any one of claims 1 to 8.
The resonant frequency detection device of a SAW sensor, wherein the SAW sensor (102; 502) comprises a plurality of the SAW elements (119a, 119b). In the SAW sensor resonance frequency detection device according to claim 9,
The transmitters (107a, 107b; 507a, 507b) are provided corresponding to the plurality of SAW elements, and the receivers (115a, 115b; 515a, 515) are respectively corresponding to the plurality of SAW elements. Provided
The resonance frequency detection device for a SAW sensor, wherein the auxiliary circuit (116; 516) is shared between the plurality of transmitters and the plurality of receivers.

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