JP6374476B2 - Receiver - Google Patents

Description

  The present invention relates to a receiving apparatus.

  In wireless communication such as digital terrestrial television broadcasting, distortion may occur in a transmission path due to multipath interference, and reception accuracy may decrease. For this reason, in a receiving apparatus such as a digital television receiver, for example, an equalization process for correcting distortion generated in the transmission path is performed to prevent a decrease in reception accuracy (see, for example, Patent Document 1). .

JP 2010-268100 A

  However, the above-described conventional technology has room for further improvement in terms of improving the reception accuracy.

  The disclosed technique has been made in view of the above, and an object of the present invention is to provide a receiving apparatus capable of improving reception accuracy.

The receiving device disclosed in the present application includes a Fourier transform unit and a phase correction unit. The Fourier transform unit performs a fast Fourier transform process on the orthogonal frequency division multiplexed signal. The phase correction unit performs correction to match the phase of the orthogonal frequency division multiplexed signal after the fast Fourier transform process with a reference phase that is a phase when the fast Fourier transform process is performed at a reference position that arrives at a fixed period. .

  According to one aspect of the receiving device disclosed in the present application, it is possible to improve reception accuracy.

FIG. 1 is a block diagram illustrating the configuration of the receiving apparatus according to the present embodiment. FIG. 2 is a block diagram illustrating a configuration of the guard composition processing unit. FIG. 3 is an explanatory diagram of the guard composition process. FIG. 4A is a diagram illustrating an example in which impulse noise is included in the guard interval. FIG. 4B is an explanatory diagram of the first replacement process. FIG. 5A is a diagram illustrating an example in the case where impulse noise is included in the tail portion of the effective symbol. FIG. 5B is an explanatory diagram of the second replacement process. FIG. 6 is a diagram illustrating a circuit configuration example of the guard composition processing unit. FIG. 7 is an explanatory diagram of the enable signal. FIG. 8A is a diagram illustrating an example when the FFT window is set to an appropriate position. FIG. 8B is a diagram illustrating an example when the FFT window is set in front of an appropriate position. FIG. 9 is an explanatory diagram of the phase correction process. FIG. 10A is a diagram illustrating an example of a conventional delay profile. FIG. 10B is a diagram illustrating an example of a conventional delay profile. FIG. 11A is a diagram illustrating an example of a delay profile in the present embodiment. FIG. 11B is a diagram illustrating an example of a delay profile in the present embodiment. FIG. 12A is a diagram illustrating an example of a delay profile. FIG. 12B is a diagram illustrating conditions for setting the FFT window position. FIG. 13 is a diagram illustrating another example of the delay profile. FIG. 14 is a block diagram illustrating a configuration of the window position control unit. FIG. 15 is an explanatory diagram of the window position setting process. FIG. 16A is an explanatory diagram of window position adjustment processing. FIG. 16B is an explanatory diagram of the window position adjustment process. FIG. 16C is an explanatory diagram of the window position adjustment process. FIG. 16D is an explanatory diagram of the window position adjustment process. FIG. 17 is a flowchart showing the processing procedure of the window position setting process. FIG. 18 is a flowchart illustrating the processing procedure of the window position adjustment processing. FIG. 19A is an explanatory diagram of a conventional method of calculating a transmission line response. FIG. 19B is an explanatory diagram of a conventional method for calculating a transmission line response. FIG. 20 is a schematic explanatory diagram of a transmission path calculation unit according to the present embodiment. FIG. 21 is a block diagram illustrating a configuration of the transmission path calculation unit. FIG. 22A is a block diagram illustrating a configuration of the first noise removing unit. FIG. 22B is a block diagram illustrating a configuration of the second noise removing unit. FIG. 23A is an explanatory diagram of the time direction averaging process. FIG. 23B is a diagram illustrating a transmission path response estimation value after time direction averaging processing output from the first noise removal unit.

  Exemplary embodiments of a receiving device disclosed in the present application will be described below in detail with reference to the accompanying drawings. Hereinafter, an orthogonal frequency division multiplexed signal (hereinafter referred to as “broadcast signal”) that is a signal of an OFDM (Orthogonal Frequency Division Multiplexing) system that is broadcast using a plurality of carriers whose phases are orthogonal to each other (hereinafter referred to as “carrier”). Will be described.

  Hereinafter, an example in which the receiving apparatus is mounted on a moving body such as an automobile will be described. However, the receiving device does not necessarily need to be mounted on the moving body.

  First, the configuration of the receiving apparatus according to the present embodiment will be described with reference to FIG. FIG. 1 is a block diagram illustrating the configuration of the receiving apparatus according to the present embodiment. In FIG. 1, constituent elements necessary for explaining the characteristics of the receiving device are shown, and descriptions of general constituent elements are omitted as appropriate.

  As shown in FIG. 1, the receiving apparatus 1 according to the present embodiment includes a tuner unit 3, an impulse removing unit 4, a guard synthesis processing unit 5, an FFT unit 6, a phase correcting unit 7, and a delay profile calculating unit. 8 and a window position control unit 9. In addition, the receiving device 1 includes a transmission path calculation unit 10, an equalization processing unit 11, a hard decision unit 12, an error correction unit 13, and a demodulation unit 14.

  The tuner unit 3 is an acquisition unit that acquires a broadcast signal input from the antenna 2 via the antenna 2. The tuner unit 3 receives an OFDM broadcast signal, specifically, a broadcast signal in which a guard interval, which is a duplicate of the tail part of an effective symbol including broadcast contents, is added to the head of the effective symbol. The tuner unit 3 also detects and amplifies the received broadcast signal, converts the analog signal into a digital signal, and outputs the converted signal to the impulse removing unit 4.

  The impulse removing unit 4 is a processing unit that removes impulse noise included in the broadcast signal input from the tuner unit 3. Here, the impulse noise is instantaneous high-frequency noise generated by, for example, lightning or starting an electrical system.

  For example, the impulse removing unit 4 detects a section in which the signal level (hereinafter simply referred to as “level”) of the input broadcast signal is equal to or greater than a predetermined threshold as a section including impulse noise. Then, the impulse removing unit 4 replaces the data of the detected section with predetermined data (for example, 0). Thereby, impulse noise can be removed from the broadcast signal.

  The impulse removing unit 4 outputs the broadcast signal after removing the impulse noise and information indicating the section in which the impulse noise is detected (hereinafter referred to as “noise detection signal”) to the guard synthesis processing unit 5. Thus, the impulse removing unit 4 is an example of a noise detecting unit that detects impulse noise included in the broadcast signal.

  The guard synthesis processing unit 5 performs a guard synthesis process and a guard replacement process on the broadcast signal from which the impulse noise is input, which is input from the impulse removal unit 4. The configuration of the guard composition processing unit 5 will be described with reference to FIG. FIG. 2 is a block diagram showing a configuration of the guard composition processing unit 5.

  As shown in FIG. 2, the guard composition processing unit 5 includes a guard composition unit 51 and a guard replacement unit 52. The guard replacement unit 52 includes a first replacement processing unit 52a and a second replacement processing unit 52b.

  The guard combining unit 51 improves the reception accuracy of the broadcast signal by combining the guard interval and the tail part of the effective symbol. The contents of the guard composition processing performed by the guard composition unit 51 will be described with reference to FIG. FIG. 3 is an explanatory diagram of the guard composition process.

  As shown in FIG. 3, in the OFDM broadcast signal, a guard interval G, which is a duplicate of the tail portion C of the effective symbol D including the broadcast content, is added to the head of the effective symbol D in order to increase resistance to multipaths and the like. ing.

  When such a broadcast signal is input, the guard combining unit 51 first delays the broadcast signal by one symbol, that is, a time corresponding to the section length of the effective symbol D.

  Subsequently, the guard combining unit 51 combines the tail part C of the broadcast signal before the delay and the guard interval G of the broadcast signal after the delay. For example, the guard combining unit 51 generates an average value of the tail portion C of the broadcast signal before delay and the guard interval G of the broadcast signal after delay as the tail portion C ′ after combining.

  Thus, the guard combining unit 51 performs a guard combining process for combining the guard interval G of the broadcast signal received by the tuner unit 3 and the tail portion C of the broadcast signal. Thereby, even if an error has occurred in the tail portion C, for example, data can be complemented by the guard interval G, so that the reception accuracy can be improved.

  By the way, when impulse noise is included in the guard interval G or the tail portion C, the portion is replaced with predetermined data (for example, 0) by the impulse removing unit 4 in the previous stage. In such a case, the predetermined data that is not the original data replaced by the impulse removing unit 4 is combined with the original data, and there is a possibility that the reception accuracy may be lowered.

  Therefore, the guard replacement unit 52 replaces the data in which the impulse noise is included in the guard interval G and the tail portion C (that is, the data from which the impulse noise has been removed by the impulse removal unit 4) with the other data. It was decided to perform guard replacement processing.

  First, when the guard interval G is data from which impulse noise has been detected and removed by the impulse removing unit 4, the first replacement processing unit 52a tails the effective symbol D obtained by delaying the guard interval G by a predetermined time. A first replacement process for replacing part C is performed.

  The contents of the first replacement process will be described with reference to FIGS. 4A and 4B. FIG. 4A is a diagram illustrating an example in a case where impulse noise is included in the guard interval G. FIG. 4B is an explanatory diagram of the first replacement process.

  As shown in FIG. 4A, it is assumed that impulse noise is included in the guard interval G of the received broadcast signal. In such a case, the impulse removing unit 4 detects and removes the impulse noise, and outputs a noise detection signal indicating a section in which the impulse noise is detected to the first replacement processing unit 52a. The first replacement processing unit 52a refers to the noise detection signal to determine whether or not the guard interval G after the guard synthesis processing is data in which the impulse noise is detected and removed by the impulse removing unit 4, that is, the impulse. It is determined whether or not a section replaced with predetermined data by the removing unit 4 is included.

  When the first replacement processing unit 52a determines that the guard interval G after the guard combining process is data from which the impulse noise has been detected and removed by the impulse removing unit 4, as illustrated in FIG. 4B, The guard interval G of the subsequent broadcast signal is replaced with the tail portion C of the broadcast signal delayed by a predetermined time (for example, one symbol) and output.

  As a result, the guard interval G can be replaced with a tail portion C in which data is more reliable than the guard interval G.

  When the tail part C of the broadcast signal is data from which the impulse noise has been detected and removed by the impulse removing unit 4, the second replacement processing unit 52b performs the tail part of the broadcast signal after the guard combining process delayed by a predetermined time. A second replacement process is performed to replace C with the guard interval G of the broadcast signal.

  The contents of the second replacement process will be described with reference to FIGS. 5A and 5B. FIG. 5A is a diagram illustrating an example in the case where impulse noise is included in the tail portion C of the effective symbol D. FIG. 5B is an explanatory diagram of the second replacement process.

  As shown in FIG. 5A, it is assumed that impulse noise is included in the tail portion C of the received broadcast signal. In such a case, the impulse removing unit 4 detects and removes the impulse noise and outputs a noise detection signal indicating a section in which the impulse noise is detected to the second replacement processing unit 52b. The second replacement processing unit 52b refers to the noise detection signal to determine whether the tail portion C of the effective symbol D is data from which the impulse noise has been detected and removed by the impulse removing unit 4. In other words, the second replacement processing unit 52b is generated by using the tail part C including the section in which the tail part C of the broadcast signal after the guard combining process is replaced with predetermined data by the impulse removing unit 4. It is determined whether or not there is.

  When the second replacement processing unit 52b determines that the tail portion C of the effective symbol D is the data from which the impulse noise has been detected and removed by the impulse removing unit 4, as shown in FIG. The tail part C of the broadcast signal after the guard combining process delayed by one symbol) is replaced with the guard interval G of the broadcast signal not delayed and output.

  As a result, the tail portion C after the guard combining process can be replaced with a guard interval G in which data is more reliable than the tail portion C.

  Therefore, it is possible to avoid a situation in which the reception accuracy is lowered due to the synthesis of the predetermined data that is not the original data replaced by the impulse removing unit 4 with the original data.

  Next, a circuit configuration example of the above-described guard composition processing unit 5 will be described with reference to FIG. FIG. 6 is a diagram illustrating a circuit configuration example of the guard composition processing unit 5.

  As shown in FIG. 6, the guard synthesis processing unit 5 includes a data delay unit 511, a detection signal delay unit 512, an adder 513, a multiplier 514, a first selector 515, a second selector 516, 3 selector 517.

  Here, the guard composition unit 51 illustrated in FIG. 2 corresponds to the addition unit 513, the multiplication unit 514, and the first selector 515 illustrated in FIG. Further, the first replacement processing unit 52a and the second replacement processing unit 52b shown in FIG. 2 correspond to the second selector 516 and the third selector 517 shown in FIG. 6, respectively.

  The data delay unit 511 delays the input broadcast signal by a predetermined time (for example, one symbol) and outputs the delayed signal to the adder 513, the second selector 516, and the third selector 517, respectively.

  The detection signal delay unit 512 delays the input noise detection signal by a predetermined time (for example, one symbol) and outputs the delayed signal to the third selector 517. The detection signal delay unit 512 delays the noise detection signal by the same time as the data delay unit 511.

  Adder 513 adds the delayed broadcast signal input from data delay unit 511 and the pre-delay broadcast signal input from impulse remover 4, and outputs the result to multiplier 514. Multiplier 514 amplifies the addition result of adder 513 by a factor of 0.5 and outputs the result to first selector 515.

  The first selector 515 selects and outputs one of the broadcast signal after the guard combining process input from the multiplying unit 514 and the broadcast signal before the guard combining process input from the impulse removing unit 4 based on the enable signal.

  The enable signal is a signal indicating a section in which a broadcast signal after the guard combining process can be output. Here, the enable signal will be described with reference to FIG. FIG. 7 is an explanatory diagram of the enable signal.

  As shown in FIG. 7, in addition to the main wave having the highest signal level, the broadcast signal includes a preceding wave that arrives earlier in time than the main wave, a delayed wave that arrives later in time than the main wave, and the like. Are received in a mixed state. The enable signal is a signal indicating a section in which the guard intervals G of the main wave, the preceding wave, and the delayed wave overlap.

  The first selector 515 outputs the broadcast signal after the guard combining process only when the enable signal is High, that is, when the guard intervals G of the main wave, the preceding wave, and the delayed wave overlap. For the other sections, broadcast signals input from the impulse removing unit 4 are output. As a result, it is possible to prevent a reduction in reception accuracy by performing guard combining processing using components of other adjacent symbols, and it is possible to demodulate a broadcast signal with high reliability in the demodulator 14 described later. It becomes.

  The enable signal is generated by an enable signal generation unit (not shown) based on a delay profile calculated by, for example, a delay profile calculation unit 8 described later, and is output to the guard synthesis processing unit 5.

  Returning to FIG. 6, the second selector 516 will be described. The second selector 516 selects and outputs one of the delayed broadcast signal input from the data delay unit 511 and the broadcast signal input from the first selector 515 based on the enable signal and the noise detection signal.

  Specifically, the second selector 516 determines that when the enable signal and the noise detection signal are both high, that is, the data input from the impulse removing unit 4 is set to the guard intervals G of the main wave, the preceding wave, and the delayed wave. In the case of the data in the overlapping section and the section in which the impulse noise is detected, the data input from the data delay unit 511, that is, the tail portion C of the effective symbol D is output.

  On the other hand, when only one of the enable signal and the noise detection signal is High, or when both the enable signal and the noise detection signal are Low, the second selector 516 receives the data input from the first selector 515, that is, A guard interval G is selected and output. Thus, the second selector 516 corresponds to the first replacement processing unit 52a.

  The third selector 517 selects and outputs one of the broadcast signal input from the impulse removing unit 4 and the delayed broadcast signal input from the data delay unit 511 based on the enable signal and the delayed noise detection signal.

  Specifically, when both the enable signal and the delayed noise detection signal are High, that is, the data input from the data delay unit 511 is the tail portion C of the effective symbol D, and the impulse noise is detected. If it is a section, the data input from the impulse removing unit 4, that is, the guard interval G is selected and output.

  On the other hand, the third selector 517 receives an input from the data delay unit 511 when only one of the enable signal and the delayed noise detection signal is High or when both the enable signal and the delayed noise detection signal are Low. Selected data, that is, the tail portion C of the effective symbol D is selected and output. Thus, the third selector 517 corresponds to the second replacement processing unit 52b.

  Thus, the receiving apparatus 1 according to the present embodiment includes the guard replacement unit 52, so that when guard combining processing is performed, predetermined data that is not the original data replaced by the impulse removing unit 4 is originally It is possible to prevent the reception accuracy from being reduced by combining with the data.

  Returning to FIG. 1, the FFT unit 6 will be described. The FFT unit 6 performs FFT (Fast Fourier Transform) processing on the broadcast signal input from the guard synthesis processing unit 5, thereby converting the broadcast signal from a signal in the time domain to a signal in the frequency domain. Part. The FFT unit 6 outputs the broadcast signal after the FFT processing to the phase correction unit 7.

  The FFT unit 6 executes the FFT process for each predetermined processing section (hereinafter referred to as “FFT window”), and the FFT position is set by the window position control unit 9.

  Here, phase rotation occurs in the broadcast signal after the FFT processing depending on the position of the FFT window set by the window position control unit 9, that is, the FFT processing start position (hereinafter referred to as "FFT window position"). There is. This will be described with reference to FIGS. 8A and 8B. FIG. 8A is a diagram illustrating an example when the FFT window is set at an appropriate position, and FIG. 8B is a diagram illustrating an example when the FFT window is set ahead of the appropriate position.

  As shown in the left diagram of FIG. 8A, when the FFT window position is set to an appropriate position, specifically, at the beginning of the effective symbol D, as shown in the right diagram of FIG. No phase rotation occurs. On the other hand, when the FFT window position is set ahead of the effective symbol D as shown in the left diagram of FIG. 8B, phase rotation occurs in the broadcast signal after the FFT processing, as shown in the right diagram of FIG. 8B.

  As described above, the broadcast signal includes a main wave, a preceding wave, and a delayed wave. For this reason, for example, when the FFT window position is set in accordance with the leading position of the effective symbol D of the main wave, phase rotation does not occur in the main wave, but phase rotation occurs in the preceding wave and the delayed wave. Therefore, in a broadcast signal in which these main wave, preceding wave, and delay wave are mixed, phase rotation occurs by performing FFT processing. The position and number of the preceding wave and the delayed wave change every moment, and the FFT window position also changes every FFT processing. For this reason, broadcast signals having different phase rotation amounts are output for each FFT process.

  Therefore, the receiving apparatus 1 according to the present embodiment is provided with the phase correction unit 7 subsequent to the FFT unit 6. The contents of the phase correction processing performed by the phase correction unit 7 will be described with reference to FIG. FIG. 9 is an explanatory diagram of the phase correction process.

  As shown in FIG. 9, the FFT window is set to an optimum position for each FFT process by the window position control unit 9, and the FFT unit 6 outputs a broadcast signal having a different phase rotation amount for each FFT process. The phase correction unit 7 performs a process of correcting the phase of each broadcast signal after the FFT process so as to coincide with a predetermined reference phase.

  Specifically, the phase correction unit 7 performs FFT processing at a reference position where the phase rotation amount of each broadcast signal after FFT processing arrives at a constant period (hereinafter referred to as “FFT window reference position”). The phase of each broadcast signal after the FFT processing is rotated so that the phase rotation amount is as high as possible. Thereby, the apparent phase rotation amount of each broadcast signal after the FFT processing can be matched.

  Thus, in the receiving apparatus 1 according to the present embodiment, the phase correction unit 7 corrects the phase of the broadcast signal after the FFT processing so as to match the predetermined reference phase.

  Thereby, for example, even when an ICI canceller for removing an inter-carrier interference component (hereinafter referred to as “ICI: Inter Carrier Interference”) is provided in the receiving apparatus 1 from the broadcast signal after the FFT processing, No need to stop.

  That is, conventionally, broadcast signals after FFT processing are sequentially stored in a plurality of buffers, and a plurality of other (for example, three) broadcast signals so as to match the phase rotation amount of one of the broadcast signals. The phase was corrected and output. In this case, the phase rotation amount of the broadcast signal to be output changes depending on the phase rotation amount of the broadcast signal used as a reference.

  The ICI canceller estimates and removes whether an ICI component is included based on a change in the broadcast signal. For this reason, if the phase rotation amount of the broadcast signal after phase correction changes for each FFT process, the ICI canceller may not operate normally. Under such circumstances, conventionally, the ICI canceller is prevented from malfunctioning by stopping the ICI canceller during phase rotation correction.

  On the other hand, according to the receiving apparatus 1 according to the present embodiment, a broadcast signal having a constant phase rotation amount is output from the phase correction unit 7 even if the FFT window position changes for each FFT process. Thus, it is not necessary to stop the ICI canceller even during phase correction as in the prior art.

  In addition, as described above, a plurality of phase correction units are conventionally used. However, since only one phase correction unit 7 is used in the receiving apparatus 1 according to the present embodiment, the circuit scale may be reduced. it can.

  Furthermore, according to the receiving apparatus 1 according to the present embodiment, by providing the phase correction unit 7, the LMS unit 112 (see FIG. 20) provided in the delay profile calculation unit 8 (see FIG. 1) and the transmission path calculation unit 10 described later. ) Can be optimized, but this point will be described later.

  Returning to FIG. 1, the delay profile calculation unit 8 will be described. In FIG. 1, among the broadcast signals output from the phase correction unit 7, a scattered pilot signal that is a pilot signal (hereinafter referred to as “SP signal”) is denoted by “SP”, and a data signal is denoted by “data”. ". As shown in FIG. 1, the data signal is input to the equalization processing unit 11, and the SP signal is input to the delay profile calculation unit 8 and the transmission path calculation unit 10, respectively.

  The delay profile calculator 8 generates a delay profile based on the SP signal input from the phase corrector 7 and generates a delay profile that is information regarding the arrival time of the incoming wave such as the main wave, the preceding wave, and the delayed wave included in the broadcast signal. Output to the position controller 9.

  For example, the delay profile calculation unit 8 calculates a transfer function related to the transmission path of the broadcast signal based on the SP signal input from the phase correction unit 7. Subsequently, the delay profile calculation unit 8 calculates the arrival time of each incoming wave included in the broadcast signal by performing IFFT (Inverse Fast Fourier Transform) on the calculated transfer function. Then, the delay profile calculation unit 8 outputs information including the delay time of each incoming wave and the level of each incoming wave with respect to the arrival time of the main wave to the window position control unit 9 as a delay profile.

  Here, as described above, in the receiving apparatus 1 according to the present embodiment, the operation of the delay profile calculation unit 8 can be optimized by providing the phase correction unit 7 subsequent to the FFT unit 6. This point will be described with reference to FIGS. 10A, 10B, 11A, and 11B. 10A and 10B are diagrams illustrating an example of a conventional delay profile, and FIGS. 11A and 11B are diagrams illustrating an example of a delay profile in the present embodiment.

  Conventionally, when performing phase correction, phase correction is performed with reference to the FFT processing start position (FFT window position). For this reason, for example, the position of each incoming wave on the delay profile has changed between the case where no preceding wave exists as shown in FIG. 10A and the case where a preceding wave exists as shown in FIG. 10B.

  On the other hand, as described above, the phase correction unit 7 according to the present embodiment performs the post-FFT processing so that the phase rotation amount is as if the FFT processing was performed at the FFT window reference position that arrives at a fixed period. Rotate the phase of each broadcast signal. Specifically, the FFT window reference position is the center position of the delay profile.

  Thus, for example, even if the FFT window position changes between when there is no preceding wave as shown in FIG. 11A and when there is a preceding wave as shown in FIG. 11B, each incoming wave on the delay profile The position of does not change. Therefore, the operation of the delay profile calculation unit 8 can be optimized, for example, it is easy to estimate which incoming wave is the main wave and the preceding wave.

  The delay profile calculated by the delay profile calculation unit 8 is output to the window position control unit 9 (see FIG. 1). Here, before describing the configuration of the window position control unit 9, a conventional FFT window setting method will be described with reference to FIGS. 12A and 12B. FIG. 12A is a diagram illustrating an example of a delay profile, and FIG. 12B is a diagram illustrating conditions for setting the FFT window position.

  FIG. 12A shows an example in which one preceding wave (4) and three delayed waves (2), (3), (5) exist in addition to the main wave (1). In addition, the number attached | subjected to each incoming wave has shown the order with the high level.

  As shown in FIG. 12B, conventionally, in order to prevent intersymbol interference as much as possible, (a) “must capture the main wave”, (b) “capture the non-main wave from the incoming wave in order from the highest level”, The FFT window position is set according to setting conditions such as (c) “do not miss the preceding wave” and (d) “capture as much delayed wave as possible”. These setting conditions are assumed to have a higher priority in the order of (a) to (d).

  For example, according to the setting condition (a), the main wave (1) is selected, and according to the setting condition (b), the delay wave (2), (3), the preceding wave (4), and the delay wave (5) are selected in this order. The In accordance with the setting condition (c), the preceding wave (4) is selected, and in accordance with the setting condition (d), the delayed waves (2), (3), and (5) are selected in this order. Conventionally, these waves are comprehensively determined to select an incoming wave to be taken into a processing section (FFT window) of FFT processing. In the example shown in FIG. 12A, an FFT window for capturing the main wave (1), the delayed wave (2), and the preceding wave (4) is set.

  However, in the conventional setting method, the FFT window position may not be set appropriately. This point will be described with reference to FIG. FIG. 13 is a diagram illustrating another example of the delay profile.

  For example, in the case of the example shown in FIG. 13, conventionally, after selecting the main wave (1) and the delayed waves (2) and (3) having a high level, the preceding wave (4) and the delayed wave (5) are compared. In order to select the preceding wave (4) having a higher level, the FFT window is set at the position of the solid line shown in FIG.

  In this case, since the preceding wave (4) and the delayed waves (5) to (8) cannot be taken into the FFT window, there is a high possibility of causing intersymbol interference.

  For example, giving up the acquisition of the preceding wave (4) and receiving the delayed waves (5) to (8) may improve the receiving accuracy, but the receiving accuracy is not always improved depending on the situation. .

  Therefore, the window position control unit 9 according to the present embodiment adjusts the FFT window position in a step-track manner so that intersymbol interference does not occur more reliably.

  Here, the configuration of the window position control unit 9 according to the present embodiment will be described with reference to FIG. FIG. 14 is a block diagram illustrating a configuration of the window position control unit 9. As shown in FIG. 14, the window position control unit 9 includes a window position setting unit 91 and a window position adjustment unit 92.

  The window position setting unit 91 is a processing unit that sets the FFT window position until the delay profile is output from the delay profile calculation unit 8. Here, the contents of the window position setting process by the window position setting unit 91 will be described with reference to FIG. FIG. 15 is an explanatory diagram of the window position setting process.

  As shown in FIG. 15, the window position setting unit 91 acquires a broadcast signal output from, for example, the guard composition processing unit 5 (see FIG. 1). Subsequently, the window position setting unit 91 calculates a moving average value of the acquired broadcast signal and a signal obtained by delaying the broadcast signal by one symbol. Here, as described above, since the guard interval G is a copy of the tail portion C of the effective symbol D, the moving average value increases in the guard interval G and the tail portion C.

  Then, the window position setting unit 91 sets the peak position of the moving average value (hereinafter referred to as “guard correlation peak”), that is, the head position of the effective symbol D as the FFT window position.

  The window position adjustment unit 92 is a processing unit that sets the FFT window position after the delay profile is output from the delay profile calculation unit 8. The window position adjustment unit 92 performs a window position adjustment process for setting the current FFT window position by adjusting the previously set FFT window position based on the intensity of the incoming wave located outside the section of the FFT window. .

  Here, the contents of the window position adjustment process performed by the window position adjustment unit 92 will be described with reference to FIGS. 16A to 16D. 16A to 16D are explanatory diagrams of the window position adjustment process.

  In FIG. 16A, in addition to the main wave (1), there are one preceding wave (4) and four delayed waves (2), (3), (5), (6). In the example, the FFT window position is set so as to capture the main wave (1) and the delayed waves (2) and (3). The numbers given to the preceding wave (4) and the delayed waves (5) and (6) indicate the levels of these incoming waves.

  First, the window position adjustment unit 92 uses the level of the incoming wave located outside the previously set FFT window section, and moves the FFT window set last time to the right (delay side) and the left direction (preceding). Define the force to move to the side).

  Specifically, the force that moves the FFT window in the right direction is the total value of the incoming waves that protrude rightward with respect to the FFT window, and the force that moves the FFT window in the left direction is the FFT window. Is the total value of the levels of incoming waves that protrude in the left direction.

  In the case shown in FIG. 16A, the delayed waves (5) and (6) protrude to the right of the previously set FFT window, and these levels are “20” and “5”, respectively. The force to move in the right direction is “25”. Further, since the preceding wave (4) protrudes to the left of the FFT window set last time and the level of the preceding wave (4) is “15”, the force to move the FFT window to the left is “15”.

  Next, the window position adjustment unit 92 calculates the previously set FFT window moving direction and moving amount Δ based on the calculated left and right forces. Specifically, the window position adjustment unit 92 determines the larger one of the calculated left and right forces as the moving direction of the FFT window. In the case shown in FIG. 16A, since the force to move in the right direction is larger, the window position adjustment unit 92 determines the moving direction of the FFT window as “right direction”.

  Thus, the window position adjustment unit 92 depends on the magnitude relationship between the intensity of the incoming wave temporally preceding the section of the FFT window and the intensity of the incoming wave delayed in time than the section of the FFT window. The direction for moving the previously set FFT window position is determined.

  Further, the window position adjusting unit 92 increases the FFT window movement amount Δ as the calculated difference between the left and right forces increases. As described above, increasing the moving amount Δ is not limited to increasing the moving speed of the FFT window. For example, the moving amount Δ is a fixed amount, and the update interval is shortened as the difference between the left and right forces increases. The moving speed of the FFT window may be increased by.

  Subsequently, the window position adjustment unit 92 assumes that the previously set FFT window has been moved with the movement direction and movement amount Δ provisionally determined as described above (see FIG. 16B), and the FFT window is moved to the right. The force for moving to the left and the force for moving to the left are calculated again.

  In the case shown in FIG. 16B, there is no change in the left and right forces even after the temporary movement, and the force that moves the FFT window to the right is larger as before the movement. As described above, when the direction of the force does not change before and after the movement, the window position adjustment unit 92 sets the FFT window position after the temporary movement as the FFT window position of the current FFT processing.

  On the other hand, a case where the magnitude relationship between the left and right forces changes as a result of calculating the left and right forces again after temporarily determining the movement direction and the movement amount Δ will be described.

  In FIG. 16C, in addition to the incoming waves (1) to (6) shown in FIG. 16A, a preceding wave (7) further exists, and the main wave (1), delayed wave (2), An example in which the FFT window position is set so as to capture (3) and the preceding wave (7) is shown.

  Assuming that the previously set FFT window has been moved with the temporarily determined movement direction and movement amount Δ, the window position adjustment unit 92 again applies the force to move the FFT window to the right and the force to move to the left. calculate. As a result, the force for moving the FFT window after provisional movement to the right is “25”, the force for moving to the left is “39”, and the magnitude relationship between the left and right forces is reversed before and after the movement.

  In such a case, the window position adjustment unit 92 performs readjustment processing. For example, as shown in FIG. 16D, the window position adjustment unit 92 moves in a direction opposite to that during temporary movement (here, in the left direction) and is smaller than the movement amount Δ during temporary movement (for example, Δ In 2), the FFT window after the temporary movement is moved again. Then, the window position adjustment unit 92 sets the FFT window position after the re-movement as the FFT window position for the current FFT process.

  Note that the window position adjustment unit 92 may obtain the optimal solution by repeating the re-moving process until the magnitude relationship between the left and right forces does not change before and after the movement. Further, the window position adjustment unit 92 may set the previous FFT window position as it is as the current FFT window position, for example, as another readjustment process.

  Next, specific processing procedures of the above-described window position setting process and window position adjustment process will be described with reference to FIGS. 17 and 18, respectively. FIG. 17 is a flowchart showing the processing procedure of the window position setting process, and FIG. 18 is a flowchart showing the processing procedure of the window position adjustment process.

  Note that the window position setting process shown in FIG. 17 is executed until the delay profile is input from the delay profile calculation unit 8, and the window position adjustment process shown in FIG. 18 is executed after the delay profile is input. . Moreover, the window position setting part 91 and the window position adjustment part 92 perform the process sequence shown in FIG. 17 and FIG. 18 for every FFT process.

  First, the processing procedure of the window position setting process will be described with reference to FIG. As illustrated in FIG. 17, the window position setting unit 91 sets the FFT window position based on the guard correlation peak (step S101). That is, the window position setting unit 91 calculates a moving average value of the broadcast signal acquired from the guard composition processing unit 5 and a signal obtained by delaying the broadcast signal by one symbol, and is the peak position of the moving average value. The guard correlation peak is set as the FFT window position.

  Next, the processing procedure of the window position adjustment process will be described with reference to FIG. As illustrated in FIG. 18, the window position adjustment unit 92 acquires a delay profile from the delay profile calculation unit 8 (step S201).

  Subsequently, the window position adjustment unit 92 extracts path information before and after the environment of the FFT window position (step S202). That is, the window position adjustment unit 92 extracts information on the position and level of the incoming wave outside the previous FFT window section.

  Subsequently, the window position adjustment unit 92 calculates the left and right forces for moving the previous FFT window position (step S203), and temporarily calculates the moving direction and the movement amount Δ of the previous FFT window position based on the calculation result. Determine (step S204).

  Subsequently, the window position adjustment unit 92 calculates the left and right forces for moving the FFT window with respect to the FFT window position after the temporary movement with the temporarily determined movement direction and movement amount Δ (step S205).

  Subsequently, the window position adjustment unit 92 determines whether or not the direction of force (A) calculated in step S203 and the direction of force (B) calculated in step S205 are the same direction (step S206). And when it determines with (A) and (B) being the same direction (step S206, Yes), the window position adjustment part 92 sets the FFT window position after temporary movement to this FFT window position. (Step S207), the current window position adjustment processing ends. On the other hand, if (A) and (B) are not in the same direction in step S206 (step S206, No), the window position adjustment unit 92 performs readjustment processing (step S208) and performs the current window position adjustment processing. Finish.

  Thus, in the receiving apparatus 1 according to the present embodiment, the window position control unit 9 adjusts the previously set FFT window position based on the intensity of the incoming wave located outside the section of the FFT window. The FFT window position was set. As a result, the FFT window position can be set more appropriately, and the reception accuracy can be improved.

  In the above-described example, when the direction of the force does not change before and after the temporary movement, the FFT window position after the temporary movement is set as the FFT window position of the current FFT processing. In such a case, the adjustment unit 92 may move the FFT window slightly to the right. This is because it is usually difficult to think that the radio wave environment is constant, and basically the environment of main wave + delay wave is mainly, even if the direction of force does not change before and after temporary movement, By moving the FFT window in the right direction so that the main wave is always located on the left side of the FFT window, the FFT window can be set to a more optimal position.

  Further, the window position adjustment unit 92 may be configured to operate only on an incoming wave that has undergone intersymbol interference by simplifying the circuit. In such a case, the force to move the FFT window to the left and right using only the incoming wave existing in a predetermined range centered on the FFT window, for example, is not a target for processing all the incoming waves that protrude from the section of the FFT window. You may make it calculate. By doing in this way, even if the incoming wave that was within the range of the FFT window momentarily goes out of the range and deteriorates, it can immediately return to the state before deterioration.

  In addition, the window position adjustment unit 92 reduces the situation of missing a necessary incoming wave by calculating the force to move the FFT window to the left and right using only the incoming wave that exists in a predetermined range centered on the FFT window. can do.

  Note that the window position control unit 9 detects only an incoming wave having a level equal to or higher than a predetermined threshold from the delay profile input from the delay profile calculation unit 8 and sets it as a target of the window position control process. The number of incoming waves to be subjected to the window position control process is not particularly limited. The threshold value can be changed as appropriate. For example, the threshold value may be set to zero, and all incoming waves included in the delay profile may be processed.

  Further, in order to prevent erroneous determination, whether or not to move the FFT window position may be determined based on a plurality of control results instead of one control result.

  Next, the transmission path calculation unit 10 (see FIG. 1) will be described. The transmission path calculation unit 10 is a processing unit that calculates a transmission path response estimated value for each carrier of each broadcast signal. Here, a conventional method of calculating a transmission path response estimated value will be described with reference to FIGS. 19A and 19B. 19A and 19B are explanatory diagrams of a conventional method for calculating a transmission path response estimated value.

  As shown in FIG. 19A, conventionally, the SP signal is averaged using an averaging filter without a center tap to remove noise contained in the SP signal, and the SP signal after noise removal is used as a transmission channel response estimation value. Was output as.

  The averaging filter is, for example, a FIR (Finite Impulse Response) filter, and performs processing for averaging the SP signal using the tap coefficient input from the LMS unit. The LMS unit adaptively updates the tap coefficients of the averaging filter according to an optimization algorithm called an LMS (Least Mean Square) algorithm.

  For example, as shown in FIG. 19A, consider a case where the tap coefficient corresponding to the SP signal SP3 is updated. In such a case, first, the averaging filter uses four SP signals SP1, SP2, SP4, SP5 existing around the SP signal SP3 and tap coefficients Wn corresponding to the SP signals SP1, SP2, SP4, SP5. Calculate the weighted average value.

  Subsequently, the LMS unit updates the tap coefficient Wn based on the weighted average value calculated by the averaging filter (that is, the estimated value of the SP signal SP3) and the measured value of the SP signal.

  The SP signal is a signal having a known amplitude and phase, and the LMS unit obtains an error e (n) between the estimated value of the SP signal and the known value, and executes LMS so that the error e (n) is reduced. The tap coefficient Wn is updated in the section. The updated tap coefficient Wn + 1 is output to the averaging filter. The LMS unit updates the tap coefficient Wn corresponding to each SP signal by sequentially performing such processing along the time direction, for example.

  Then, as shown in FIG. 19B, the averaging filter performs the averaging process again using the updated tap coefficient Wn + 1. In this way, by performing the weighted average process using the peripheral SP signal, noise components such as AWGN (Additive White Gaussian Noise) having no correlation with the peripheral SP signal can be removed from the original SP signal. it can. Then, the averaging filter outputs the SP signal after the averaging process as a transmission path response estimated value.

  However, as described above, when an averaging filter without a center tap is used as the averaging filter, there is a possibility that the tap coefficient output from the LMS unit may not converge properly depending on the situation. For example, when the number of delay waves is large and the bandwidth has to be widened, the tap coefficient output from the LMS unit does not converge properly, and the characteristics of the averaging filter rise out of the band. There was a risk of distortion.

  This is because, in an environment where the number of delay waves is large and the bandwidth must be widened, the tap coefficient converges by the LMS unit in a form close to the impulse response (that is, a form in which only the center tap is large). In the method, since an averaging filter without a center tap is used, it can be considered that if a shape close to the impulse response is obtained, the out-of-band shape converges.

  Therefore, the transmission line calculation unit 10 according to the present embodiment uses an averaging filter having a center tap. This point will be described with reference to FIG. FIG. 20 is a schematic explanatory diagram of the transmission path calculation unit 10 according to the present embodiment.

  As illustrated in FIG. 20, the transmission path calculation unit 10 according to the present embodiment includes a time direction averaging filter 111 and an LMS unit 112. The LMS unit 112 is a coefficient updating unit including a subtraction unit 112a and an LMS execution unit 112b.

  Unlike the above-described conventional averaging filter, the time direction averaging filter 111 is an averaging filter such as an FIR filter having a center tap.

  For example, consider a case where the tap coefficient corresponding to the SP signal SP3 is updated as shown in FIG. In such a case, the time direction averaging filter 111 calculates the SP signal SP3 and the four SP signals SP1, SP2, SP4, SP5 existing around the SP signal SP3 and the tap coefficients Wn corresponding to the SP signals SP1 to SP5. Calculate the weighted average value used.

  Subsequently, the LMS unit 112 updates the tap coefficient Wn based on the weighted average value calculated by the time direction averaging filter 111 (that is, the estimated value of the SP signal SP3) and the measured value of the SP signal. In the LMS unit 112, the subtraction unit 112a subtracts the estimated value of the SP signal obtained by the time direction averaging filter 111 from the SP signal input from the phase correction unit 7 (see FIG. 1), thereby generating an error e (n). Is calculated. Then, the LMS execution unit 112b updates the tap coefficient Wn according to the LMS algorithm so that the error e (n) input from the subtraction unit 112a is reduced.

  Then, the time direction averaging filter 111 performs the above averaging process again using the updated tap coefficient Wn + 1, and outputs the SP signal after the averaging process to the equalization processing unit 11 as a transmission path response estimated value. .

  As described above, in the transmission path calculation unit 10 according to the present embodiment, the time direction averaging filter 111 performs temporal processing on the SP signal (here, SP signal SP3) to be averaged and the SP signal. The SP signals to be averaged are averaged using a plurality of SP signals before and after (here, SP signals SP1, SP2, SP4, SP5). Thereby, for example, even when the number of delay waves is large and the bandwidth needs to be widened, the tap coefficients output from the LMS unit 112 can be appropriately converged.

  By the way, when an averaging filter having a center tap such as the time direction averaging filter 111 according to the present embodiment is used, tap coefficients converge so that input = output. That is, convergence is performed so that the tap coefficient of the center tap is 1 and the other tap coefficients are 0.

  Therefore, the time direction averaging filter 111 according to the present embodiment multiplies the tap coefficient of the center tap output from the LMS unit 112 by a predetermined forgetting coefficient α.

  In this way, by using a value obtained by multiplying the tap coefficient of the SP signal to be averaged updated by the LMS unit 112 by the predetermined forgetting coefficient α as the tap coefficient corresponding to the SP signal, the input = It is possible to prevent the tap coefficients from converging so as to be output.

  The forgetting factor α is 0 <α <1. Here, it is assumed that the forgetting factor α is a fixed value, but the forgetting factor α may be a value that is automatically updated. For example, the transmission path calculation unit 10 may update the forgetting factor α to a value whose upper limit is tap coefficients other than the center tap × β (β> 1) for each tap coefficient updating process.

  Next, the configuration of the transmission path calculation unit 10 according to the present embodiment will be described with reference to FIG. FIG. 21 is a block diagram illustrating a configuration of the transmission path calculation unit 10. As illustrated in FIG. 21, the transmission path calculation unit 10 includes a first noise removal unit 101 and a second noise removal unit 102.

  The first noise removing unit 101 is a processing unit that removes noise included in the SP signal by averaging the SP signal in the time direction. The second noise removing unit 102 is a processing unit that removes noise included in the SP signal by averaging the SP signal in the carrier direction. The configurations of the first noise removing unit 101 and the second noise removing unit 102 will be described with reference to FIGS. 22A and 22B. 22A is a block diagram illustrating a configuration of the first noise removing unit 101, and FIG. 22B is a block diagram illustrating a configuration of the second noise removing unit 102.

  As shown in FIG. 22A, the first noise removing unit 101 includes a time direction averaging filter 111 and an LMS unit 112 shown in FIG. The transmission path response estimated value h1 that is the SP signal from which noise has been removed by the first noise removing unit 101 is input to the second noise removing unit 102.

  Here, the content of the time direction averaging process performed by the first noise removing unit 101 will be described with reference to FIGS. 23A and 23B. FIG. 23A is an explanatory diagram of the time direction averaging process, and FIG. 23B is a diagram illustrating the channel response estimation value after the time direction averaging process output from the first noise removing unit 101. Note that the black circles shown in FIG. 23A indicate SP signals. FIG. 23A shows an example in which noise is removed from the SP signal of a symbol surrounded by a broken line.

  The time direction averaging filter 111 is centered on the SP signal SP03 and removes the noise of the SP signal SP03 in the time period surrounded by a broken line, and the SP signal SP03 and the SP signals SP01, SP02, A weighted average value using SP04 and the corresponding tap coefficients is calculated (pattern 0).

  As shown in FIG. 23A, SP signals are inserted at a rate of one for every twelve carriers, for example. The time direction averaging filter 111 also performs an interpolation process for artificially increasing the SP signal (patterns 1 to 3).

  In the case of patterns 1 to 3, there is no SP signal at the center tap position. For this reason, for patterns 1 to 3, forgetting coefficient α multiplied by the tap coefficient of the center tap is set to 0, for example. Thereby, about the patterns 1-3, the same result as the case where the averaging filter without a center tap is used similarly to the past is obtained.

  Thus, the time direction averaging filter 111 can increase the SP signal in a pseudo manner as shown in FIG. 23B by performing the interpolation process. Therefore, according to the receiving apparatus 1 according to the present embodiment, noise removal processing can be performed using more SP signals than when interpolation processing is not performed, and therefore noise removal accuracy can be improved.

  The first noise removing unit 101 has four types of tap coefficient patterns corresponding to the patterns 1 to 4 described above, and updates the tap coefficients corresponding to the patterns 1 to 4 for each of the patterns 1 to 4. To do.

  The second noise removal unit 102 includes a carrier direction averaging filter 121 and an LMS unit 122, as shown in FIG. 22B. Similar to the time direction averaging filter 111, the carrier direction averaging filter 121 is an averaging filter such as an FIR filter having a center tap. The carrier direction averaging filter 121 averages the transmission path response estimation value h1 input from the first noise removing unit 101 in the carrier direction, in other words, in the frequency axis direction, thereby further reducing noise included in the SP signal. Remove. The LMS unit 122 is a coefficient updating unit having the same configuration as the LMS unit 112 included in the first noise removing unit 101. The transmission path response estimated value h2 that is the SP signal from which noise has been removed by the second noise removing unit 102 is input to the equalization processing unit 11.

  Returning to FIG. 1, the equalization processing unit 11 will be described. The equalization processing unit 11 is a processing unit that performs equalization processing for correcting the distortion of the data signal using the transmission path response estimated value output from the transmission path calculation unit 10. Specifically, the equalization processing unit 11 corrects the distortion of the data signal by dividing the data signal by the transmission path response estimated value. The data signal after the equalization processing is output to the hard decision unit 12.

  The hard decision unit 12 is a processing unit that performs a hard decision process on the data signal after the equalization process input from the equalization processing unit 11. Specifically, the hard decision unit 12 demaps each reception point of the data signal after the equalization processing onto a constellation represented by the in-phase component axis (I axis) and the quadrature component axis (Q axis). At this time, each reception point of the data signal is demapped to a position closer to the reference point of each mapping frame on the constellation as the influence of noise or reflection received on the transmission path is lower. The hard decision unit 12 outputs the data signal after the hard decision process to the error correction unit 13.

  The error correction unit 13 detects an error in the data signal input from the hard decision unit 12 using an error correction decoding method such as Viterbi decoding or Reed-Solomon decoding. To correct. The data signal after error correction is output to the demodulator 14.

  The demodulation unit 14 performs OFDM demodulation on the data signal input from the error correction unit 13 and outputs the demodulated signal to the output device 15. The output device 15 is, for example, a display device that displays digital television broadcast video, a speaker that outputs digital television broadcast audio, or the like.

  As described above, the receiving apparatus 1 according to the present embodiment includes the tuner unit 3, the impulse removing unit 4, the guard combining unit 51, and the guard replacing unit 52. The tuner unit 3 receives a broadcast signal in which a guard interval G is added to the head of the effective symbol D. The guard combining unit 51 performs a guard combining process for combining the guard interval G received by the tuner unit 3 and the tail portion C of the effective symbol D. The impulse removing unit 4 detects impulse noise included in the broadcast signal. Based on the noise detection signal from the impulse removing unit 4, the guard replacing unit 52 replaces the data in which the impulse noise is detected in the guard interval G and the tail portion C of the effective symbol D with the other data.

  The receiving device 1 according to the present embodiment includes an FFT unit 6 and a phase correction unit 7. The FFT unit 6 performs FFT processing on a broadcast signal that is an orthogonal frequency division multiplexed signal. The phase correction unit 7 corrects the phase of the broadcast signal after the FFT processing so as to match a predetermined reference phase.

  In addition, the receiving device 1 according to the present embodiment includes an FFT unit 6 and a window position control unit 9. The FFT unit 6 performs FFT processing on a broadcast signal that is an orthogonal frequency division multiplexed signal. The window position control unit 9 sets an FFT window position indicating the start position of the processing section of the FFT process for each FFT process. Further, the window position control unit 9 sets the current FFT window position by adjusting the previously set FFT window position based on the intensity of the incoming wave located outside the section of the FFT window.

  In addition, the receiving device 1 according to the present embodiment includes an FFT unit 6, a time direction averaging filter 111, a carrier direction averaging filter 121, and LMS units 112 and 122. The FFT unit 6 performs FFT processing on a broadcast signal that is an orthogonal frequency division multiplexed signal. The time direction averaging filter 111 and the carrier direction averaging filter 121 average the SP signal extracted from the broadcast signal after the FFT processing in the time direction and the carrier direction, respectively. The LMS units 112 and 122 update the tap coefficients of the time direction averaging filter 111 and the carrier direction averaging filter 121 according to the LMS algorithm. Then, the time direction averaging filter 111 and the carrier direction averaging filter 121 use the SP signal to be averaged and a plurality of SP signals that are temporally changed with respect to the SP signal. The target SP signal is averaged.

  Therefore, according to the receiver 1 according to the present embodiment, it is possible to improve the reception accuracy.

  Further effects and modifications can be easily derived by those skilled in the art. Thus, the broader aspects of the present invention are not limited to the specific details and representative embodiments shown and described above. Accordingly, various modifications can be made without departing from the spirit or scope of the general inventive concept as defined by the appended claims and their equivalents.

  As described above, the receiving device disclosed in the present application is effective when it is desired to improve the receiving accuracy, and in particular, application to an in-vehicle receiving device can be considered.

DESCRIPTION OF SYMBOLS 1 Receiver 2 Antenna 3 Tuner part 4 Impulse removal part 5 Guard composition process part 51 Guard composition part 52 Guard replacement part 52a First replacement process part 52b Second replacement process part 511 Data delay part 512 Detection signal delay part 513 Adder 514 Multiplier 515 First selector 516 Second selector 517 Third selector 6 FFT unit 7 Phase correction unit 8 Delay profile calculation unit 9 Window position control unit 91 Window position setting unit 92 Window position adjustment unit 10 Transmission path calculation unit 101 First noise Removal unit 111 Time direction averaging filter 112 LMS unit 102 Second noise removal unit 121 Carrier direction averaging filter 122 LMS unit 11 Equalization processing unit 12 Hard decision unit 13 Error correction unit 14 Demodulation unit 15 Output device

Claims (2)

A Fourier transform unit for performing a fast Fourier transform process on an orthogonal frequency division multiplexed signal;
And phase corrector phase Ru to match the criteria phase orthogonal frequency division multiplex signal after the fast Fourier transform,
An averaging filter that averages a pilot signal extracted from the orthogonal frequency division multiplexed signal whose phase is corrected by the phase correction unit;
A coefficient updating unit that updates the coefficient of the averaging filter according to a predetermined optimization algorithm ,
The averaging filter is
A receiving apparatus having a center tap, wherein only a tap coefficient of the center tap is multiplied by a predetermined forgetting factor . The averaging filter is
A time direction averaging filter that averages a pilot signal extracted from the orthogonal frequency division multiplexed signal whose phase is corrected by the phase correction unit in the time direction;
The receiving apparatus according to claim 1 , further comprising: a carrier direction averaging filter that averages a pilot signal extracted from the orthogonal frequency division multiplexed signal whose phase is corrected by the phase correction unit in a frequency direction.

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