JP6355251B2 - motor - Google Patents

Description

  In recent years, energy problems and the like have become apparent, and hybrid vehicles using motors have become widespread, and electric vehicles driven by motors have been developed. In addition, high efficiency regulations regarding industrial motors are attracting attention. The present invention relates to improvements in these motors, generators, and their drive technology. In other words, the present invention relates to higher efficiency and smaller size of the motor and lower cost of driving the motor.

  Permanent magnet type synchronous motors are often used as high-precision and high-response motors for main motors and industrial machines of hybrid vehicles. In these applications, so-called constant output characteristics based on field magnetic flux control are required to widen the range of the rotational speed of the motor. For example, as shown in FIG. 2, the motor is driven to a rotational speed four times the maximum rotational speed Nmx = 12000 rpm with respect to the base rotational speed Nba = 3000 rpm. In this case, the magnitude of the field magnetic flux at the maximum rotational speed Nmx is 1/4 with respect to the magnitude of the field magnetic flux at the base rotational speed Nba. Such a constant output characteristic is required, and a built-in magnet type synchronous motor IPMSM is often used as a permanent magnet type motor that requires control to reduce the field magnetic flux at the maximum rotation speed Nmx. Normally, the magnitude of the field magnetic flux is controlled by controlling the current supplied to the stator winding, and is realized by an advanced technology in which the motor characteristics and the motor control technology are closely related.

  Since this conventional permanent magnet type synchronous motor needs to have a small field magnetic flux at the maximum rotational speed Nmx, conversely, at a base rotational speed Nba = 3000 rpm and a low speed rotation below, with a torque T = 120 Nm, It is often necessary to energize a large field current component If1 to the stator windings. The field current component If1 is a reactive current and has a value that is so large that it cannot be ignored with respect to the torque current component It1. The current passed through the stator winding is a combined current obtained by combining the torque current component It1 of the active current and the field current component If1 of the reactive current, and the copper loss that is the Joule heat of the stator winding is a large value. Become. When it is assumed that a large torque operation at a base rotational speed Nba or less continues for a long time as in a steep climbing operation of an automobile, the size of the motor is roughly determined by a loss characteristic in the large torque operation.

  Examples of target motors will be described with reference to examples of specific numerical values with reference to a characteristic example of the rotational speed and torque in FIG. 2 and a vector diagram of each current in FIG. It is assumed that the maximum torque of this motor is 120 Nm, which is three times the continuous rated torque of 40 Nm. In FIG. 2, 24 is a point where continuous output is possible, and 23 is a short-time rated operating point. The operating point 24 is 3000 rpm and 40 Nm. For example, the output can be obtained by applying the torque current 261 and the field exciting current 262 shown in FIG. The combined current 263 is obtained as a vector sum of 261 and 262. At this time, the power factor is a cosine value of the phase 264. When 264 is 30 °, the power factor is cos (30 °) = 0.866.

  Next, each current and a power factor are shown in FIG.26 (b) about 23 operating points of short-time rating. Reference numeral 265 denotes a torque current that is three times larger than 261, 266 is a field excitation current, and 267 is a combined current. The operating point 23 is a point of maximum torque rated for a short time, and there is a demand for miniaturization and cost reduction in a normal motor design, so that a part of the motor may be magnetically saturated. In many cases, the power factor may drop to about 0.6. At that time, the phase 268 is 53.13 °, and the field excitation current 266 is larger than the torque current 265. Compared to an ideal motor with a power factor of 1, the copper loss increases to 1 / (0.6 × 0.6) = 2.777 times. The current capacity of the inverter also increases 1 / 0.6 = 1.666 times.

  Thus, when the reactive current field current component If1 is large, the power factor of the motor is low, and the copper loss is considerably larger than the copper loss equivalent value of the torque current component that is the effective current. There is a problem that the motor is increased in size and cost. At the same time, if the power factor is low, the power element such as an IGBT that supplies current to the stator windings becomes large, and there is a problem that the inverter becomes an expensive motor system.

  On the other hand, many induction motors are used for industrial purposes. When the induction motor is used with the characteristics as shown in FIG. 2 and the large torque operation at the base rotation speed Nba or less continues for a long time, the power factor in the low speed region of the motor becomes low, the motor becomes large, There is a problem that the motor becomes expensive. When a large torque operation is required in a region where the power factor is low, there is a problem that the inverter capacity is increased and the cost is increased.

  An example of a field winding type synchronous motor as another conventional motor is shown in FIG. 251 is a rotor shaft, 252 is a rotor, 253 is a field winding, 255 is a DC power source for field excitation, 256 and 257 are brushes, and 258 and 259 are slip rings. 25A is a stator and 25B is a stator winding. A motor case, a bearing, and a fixing member are not described.

  Since the field magnetic flux of the motor is generated by the field winding 253, it is reasonable, and the power supplied from the DC power supply 255 may be only the amount of resistance consumed by the field winding. There is no need to superimpose a field current component on the stator winding 25B in order to control the field magnetic flux. The current flowing through the stator is a torque current and is an effective current. There is no increase in copper loss due to the superposition of reactive currents, and there is no need to increase the current capacity of the inverter.

  However, brushes and slip rings may cause abnormal wear depending on the usage environment such as humidity and surrounding chemical substances, and have a long life, so there are maintenance burdens, service problems, and reliability problems. In addition, there is a problem of increasing the size due to installation space and maintenance space.

  Next, FIG. 27 shows a cross-sectional view of a rotor configuration having field windings connected in series with a diode. 279 is a soft magnetic body of the rotor. Field windings wound around 271 and 272 excite the rotor's N-pole magnetic pole 27F. 27G indicated by a bold line is a connection of the coil end portion of this field winding. Similarly, the field windings 273, 274, 275, 276, 277, and 278 are wound in series with their current directions aligned. An equivalent circuit of a field winding and a diode 27A are shown on the right side of the rotor 279. The winding 27B includes 271 and 272, the winding 27C includes 274 and 273, the winding 27D includes 271 and 272, the winding 27E includes 278 and 277, and the diode 27A is connected in series.

  Next, the operation of the rotor 279 will be described. First, when the field current Isf on the stator side starts to be energized and increases, each rotor magnetic pole is excited to produce a field magnetic flux φf, which increases in proportion to Isf. While the field magnetic flux φf is increasing, a negative voltage proportional to (−dφf / dt) and the number of turns is applied to the rotor field windings 27B, 27C, 27D, and 27E of FIG. 27 according to Neumann's formula. Vf is generated, and a reverse voltage is applied to the diode 27A. In this state, the rotor field current If does not flow. Next, when the field current Isf on the stator side decreases, the induced voltage of the rotor field windings 27B, 27C, 27D, and 27E is a positive value proportional to (−dφf / dt) and the number of turns according to Neumann's formula. Since the voltage Vf is reached, the field current If flows in the forward direction of the diode 27A. Even when the field current Isf on the stator side becomes zero, the field magnetic flux φf is held by the rotor field current If. If the stator-side field current Isf does not flow, the magnetic energy of the field magnetic flux φf is consumed due to the copper loss of the rotor field windings 27B, 27C, 27D, and 27E, and the field current Isf gradually increases. The field magnetic flux φf also gradually decreases in proportion to the decrease. Because of this operation, if the field magnetic flux φf is sometimes created by the field current Isf on the stator side, then the rotor field windings 27B, 27C, 27D, 27E and the diode 27A are connected to the field magnetic flux φf. Therefore, the field magnetic flux φf can be kept almost constant although it fluctuates to some extent and fluctuates.

  There is no mistake in the qualitative phenomenon and operation described above, but there are actually some practical problems. As described with reference to FIG. 26 (b), in usage and applications such as a main motor of an electric vehicle, a high torque is required for climbing up a steep slope, and the motor due to the magnitude of the burden in such operation. Size and motor cost are determined. That is, the motor efficiency, the linearity of the motor current and the motor torque, and the magnitude of the inverter burden in the large torque operation at the low speed and the medium speed determine the motor cost and the inverter cost. The generation of the field magnetic flux φf by the field current Isf on the stator side shown in FIG. 27 and the holding of the field magnetic flux φf by the rotor field windings 27B, 27C, 27D, 27E and the diode 27A are expected in the low torque region. Acts close to. However, in the large torque region, the loss in the field winding due to the field current If increases, the field magnetic flux itself operates magnetically in the nonlinear region, and the stator that excites the field magnetic flux φf. Side field current Isf increases, the torque current needs to be reduced during the energization time of the field current Isf in order to reduce the burden on the inverter, and therefore a large torque ripple occurs. The operating time width of the field current Isf also increases to generate vibration and noise due to torque ripple, etc., and when the duty that can be applied to the torque current decreases, it is necessary to increase the amplitude of the torque current accordingly. is there. Therefore, in the large torque region, the effect of the field magnetic flux holding function by the rotor field windings 27B, 27C, 27D, 27E and the diode 27A shown in FIG. 27 is reduced as a total evaluation. In other words, there is a limit to high efficiency, miniaturization, and cost reduction of the motor system.

  Next, the three-phase AC voltage and the three-phase AC current of the sine wave indicate that there is room for improvement in terms of the utilization factor of the components. The motor has been developed as a mainstream technology for three years, using a three-phase sine wave alternating current method. The three-phase AC method is the mainstream technology for both induction motors and synchronous motors. FIG. 6 shows a sinusoidal waveform of a three-phase AC voltage or current. 61 is the U phase, 62 is the V phase, and 63 is the W phase. If a three-phase sine wave voltage with an amplitude of 1 and a three-phase sine wave current are supplied, the U-phase power at that time is 64, the V-phase power is 65, and the W-phase power is 66. The average power value of each phase is ½, and the total power of the three phases is a constant value of 3/2. Considering that the output is 1 when the DC voltage is 1 and the DC current is 1, it can be seen that there is room for improvement since the sine wave is 1/2. If the ratio to the state that is most effectively utilized is referred to as a utilization factor, the utilization factor of the IGBT, which is a power element, is 50%, and it can be seen that the voltage and current characteristics are not fully utilized.

  Next, there is a difficulty in the winding method of the motor winding. The full-pitch winding of a three-phase motor is complicated because the windings of each phase intersect at the coil end, and various contrivances have been made by each manufacturer, but there is room for improvement. Examples thereof are shown in FIGS. 1 and 2 of Japanese Patent Application Laid-Open No. 2004-166476. Problem of increase in production cost due to low productivity, problem of increase in production cost due to expensive production equipment, problem of increase in size of motor due to decrease in winding space factor in slot, rotor shaft at coil end There are problems such as an increase in the size of the motor due to an increase in the direction length, and an increase in material costs accompanying an increase in the size of the motor.

JP-A-6-31786 (FIG. 1) Japanese Patent Laying-Open No. 2011-223713 (FIG. 1) JP 2004-166476 A (FIGS. 1 and 2)

The problem to be solved by claim 1 is a method for generating a field magnetic flux capable of reducing the size, cost and reliability of a motor.
The problem to be solved by claim 2 is the implementation of the field magnetic flux generation method of claim 1.
The problem to be solved by the third aspect is a rotor structure capable of more uniformly and more effectively generating the circumferential distribution of the field magnetic flux generated in the first aspect.
A problem to be solved by claim 4 is a rotor structure that can generate the field magnetic flux generated in claim 3 more effectively by using a permanent magnet.
The problem to be solved by the fifth aspect is a current energizing method that specifically links the field magnetic flux generated in the first aspect to the high efficiency of the three-phase motor, and a three-phase motor driving method.
The problem to be solved by the sixth aspect is a current energizing method that specifically links the field magnetic flux generated in the first aspect to the high efficiency of the motor, and a driving method for the five-phase motor.
The problem to be solved by claim 7 is to realize a simple five-phase motor as the specific motor configuration of claim 1.
The problem to be solved by claim 8 is to detect the magnitude of the field magnetic flux shown in claim 1.
The problem to be solved by claim 9 is a concrete motor structure that sends electric power necessary for the field winding to the rotor side in the embodiment of the power supply winding according to claim 1.
The problem to be solved by claim 10 is to realize a simple five-phase motor as the specific motor configuration of claim 1.
The problem to be solved by claim 11 is to realize a simple five-phase motor as the specific motor configuration of claim 1.
The problem to be solved by the twelfth aspect is to drive the motor at a high output without difficulty by the five-phase motor according to the first aspect.
The problem to be solved by claim 13 is a motor configuration and a sensorless position detection method for driving the motor of claim 1 by sensorless control even in a stationary state.

(Invention according to Claim 1)
The invention described in claim 1 relates to an AC motor having four or more PN poles and having three or more MN phase stators and rotors, and QN is an integer of 2 or more in the circumferential direction of the stator. A feed winding PSW that excites an AC magnetic flux component having a period of QN times 360 ° in electrical angle;
RN is an integer smaller than QN / 2, and the winding pitch in the circumferential direction is (360 ° × RN), and the first receiving winding PRW1 of the rotor that receives field power and the winding pitch in the circumferential direction Is (360 ° × RN), arranged at a circumferential position different from the first power receiving winding PRW1, and disposed on the rotor, the second power receiving winding PRW2 of the rotor for receiving the field power. , A rectifier REC1 that rectifies the output of the first power receiving winding PRW1, and a rectifier REC2 that is arranged in the rotor and rectifies the output of the second power receiving winding PRW2, and the N pole magnetic pole or S pole magnetic pole of the rotor or A field winding FM wound around both of them, and using the output of the rectifier REC1 and the output of the rectifier REC2 to pass a field current If to the field winding FM, the N pole Excitation of magnetic pole and magnetic pole It is the configuration of the chromatography data.
According to this configuration, the field excitation power can be supplied to the rotor side by using the stator winding by changing the connection method and the energization method of the stator winding, so that the size of the rotor field winding type motor can be reduced. Cost reduction can be realized.

(Invention according to claim 2)
The power supply winding PSW includes a plurality of U-phase windings that are one of the MN phases, and the winding U1M that is one of the U-phase windings. A U-phase winding U2M arranged at a circumferential position different from the winding U1M within a circumferential range of (360 ° × QN) in electrical angle, and a field current to the field winding FM The driving unit DRU1 for energizing the field excitation current component If2 by superimposing the U-phase current component on the winding U1M with If2 being the AC field excitation current component on the stator side necessary for energizing If, The motor configuration according to claim 1, further comprising: a drive unit DRU2 that applies a field excitation current component (−If2) to the winding U2M so as to be superimposed on the U-phase current component.
According to this configuration, it is possible to realize a specific field magnetic flux generation method.

(Invention according to claim 3)
According to a third aspect of the present invention, the N pole magnetic pole or the S pole magnetic pole of the rotor includes a plurality of parallel soft magnetic bodies and a gap, a resin, or a permanent magnet sandwiched between the soft magnetic bodies. The motor configuration according to claim 1.
According to this configuration, since the distribution state in the circumferential direction of the field magnetic flux generated in claim 1 can be generated more uniformly and more effectively, the induced voltage is trapezoidal closer to a rectangular wave from the conventional sine wave shape. As a result, the motor output can be improved, and as a result, the motor efficiency can be improved, the motor can be reduced in size and the motor can be reduced in cost. it can.

(Invention according to claim 4)
According to a fourth aspect of the present invention, there is provided the configuration of the motor according to the third aspect, further comprising a permanent magnet arranged in series in the direction of the magnetic flux and the magnetic flux.
According to this configuration, the electromagnetic burden for exciting the field magnetic flux according to the first aspect can be reduced by adding the permanent magnet.

(Invention according to claim 5)
According to a fifth aspect of the present invention, the stator includes each phase winding of a three-phase star connection, and the induced voltage of each phase winding is rectangular due to a magnetic flux component excited in the field winding of the rotor. A substantially trapezoidal induced voltage waveform close to a wave, and a substantially trapezoidal positive / negative current waveform has the above-mentioned substantially trapezoidal positive or negative shape with θrr being the width of the electrical angle at which the positive and negative voltages of the induced voltage are inverted. (Θrr + θst) ≦ 60 °, where θst is an electrical angle of a width in which the current of the current changes from zero to 80% of the peak value. According to this configuration, This is a current energization method that specifically combines the field magnetic flux generated in claim 1 with the high efficiency of the three-phase motor, and can drive the three-phase motor more efficiently and with less torque ripple.

(Invention according to claim 6)
According to a sixth aspect of the present invention, the stator includes each phase winding of a five-phase star connection, and the induced voltage of each phase winding is rectangular due to a magnetic flux component excited in the field winding of the rotor. In the trapezoidal induced voltage waveform close to a wave, the substantially trapezoidal positive and negative current waveforms have the substantially trapezoidal shape with θrr being the width of the electrical angle at which the positive and negative voltages of the induced voltage are inverted. 2. The motor structure according to claim 1, wherein (θrr + θsf) ≦ 36 °, where θsf is an electrical angle having a width in which a positive or negative current changes from zero to 60% of the peak value.
According to this configuration, the field magnetic flux generated in claim 1 is a current energization method that specifically links the high efficiency of the five-phase motor, and the five-phase motor can be driven more efficiently. The motor output can be improved by changing the induced voltage from a conventional sinusoidal shape to a trapezoidal shape that is closer to a rectangular wave, and the current waveform to a trapezoidal shape that is closer to a rectangular wave. At that time, although not sine wave theory, the torque ripple can be driven small.

(Invention according to claim 7)
The invention described in claim 7 is a PN pole motor having five phases and four or more poles, wherein the A phase windings are positioned at 0 ° and 180 ° in electrical angle on the circumference of the stator. The B-phase windings are arranged at 72 ° and 252 ° electrical angles on the stator circumference, and the C-phase windings are 144 ° and 324 ° electrical angles on the stator circumference. The D-phase windings are arranged at 216 ° and 36 ° electrical angles on the stator circumference, and the E-phase windings are 288 ° electrical angles on the stator circumference. The number of full-pitch windings of each phase arranged at a position of 216 ° and wound over two slots is set to (PN / 2-1) or less. It is the structure of the described motor.
According to this configuration, there is a problem that the intersection of each phase winding of the coil end portion of the five-phase winding is complicated, but this complexity is reduced by a technology that simplifies the intersection of each phase winding. It is possible to improve productivity and reduce the size of the coil end portion.

(Invention according to claim 8)
The invention according to claim 8 is a current sensor CT1 for detecting a current Iu1 energized to the winding U1M by the drive unit DRU1, and a current sensor for detecting a current Iu2 energized to the winding U2M by the drive unit DRU2. From CT2, Iu1s which is the output of the current sensor CT1, and Iu2s which is the output of the current sensor CT2, If2s is detected as a detected value of the AC field excitation current component on the stator side as (2 × If2s = Iu1s−Iu2s). The motor structure according to claim 2, further comprising an adding means for obtaining the motor.
According to this configuration, since the field current component can be detected with a little cost, the driving performance of the motor can be improved.

(Invention according to claim 9)
According to a ninth aspect of the present invention, the feed winding PSW is a field power supply winding for exciting an alternating magnetic flux component having a QN times period of 360 ° in electrical angle in the circumferential direction of the stator. The configuration of the motor according to claim 1.
According to this configuration, the field excitation power can be supplied to the rotor side using the stator winding by adding a small number of stator windings, so the rotor field winding type motor can be reduced in size and cost. can do.

(Invention according to claim 10)
The invention according to claim 10 is a five-phase motor having a number of poles that is an integral multiple of four poles, wherein the phase order of the five-phase alternating current is the order of A phase, B phase, C phase, D phase, and E phase. The circumferential position of each phase winding of the stator is described in the range from 0 ° to 720 ° in electrical angle, and the A-phase winding is wound around the slot having 0 ° electrical angle and 144 ° electrical angle. Similarly, a B-phase winding wound around a slot having an electrical angle of 72 ° and a slot having an air angle of 216 °, and similarly, winding on a slot having an electrical angle of 144 ° and a slot having an electrical angle of 288 °. A rotating C-phase winding, similarly, a D-phase winding wound around a slot with an electrical angle of 216 ° and an electrical angle of 360 °, and similarly with a slot with an electrical angle of 288 ° and an electrical angle The motor structure according to claim 1, further comprising an E-phase winding wound around a 72 ° slot.
According to this configuration, a problem that the crossing of the respective phase windings of the coil end portion of the five-phase winding becomes complicated occurs as an adverse effect of the motor efficiency improvement method according to the sixth aspect. The technology that simplifies the intersection can reduce this complication, improve productivity, and reduce the size of the coil end.

(Invention according to claim 11)
The invention according to claim 11 is a five-phase motor having a number of poles that is an integral multiple of eight poles, wherein the phase order of the five-phase alternating current is the order of A phase, B phase, C phase, D phase, and E phase. The circumferential position of each phase winding of the stator is described in the range from 0 ° to 1440 ° in electrical angle, and the concentrated winding A is wound around the slot having 0 ° electrical angle and 144 ° electrical angle. Similarly to the phase winding, similarly to the concentrated winding C-phase winding wound around the electrical angle 144 ° slot and the air angle 288 ° slot, similarly to the electrical angle 288 ° slot and electrical angle Concentrated E-phase winding wound around a 432 ° slot, as well as a concentrated B-phase winding wound around a slot of 432 ° in electrical angle and a slot of 576 ° in electrical angle, similarly A concentrated winding D-phase winding wound around a slot with an electrical angle of 576 ° and a slot with an electrical angle of 720 °, as well as an electrical angle And an A-phase winding, a C-phase winding, an E-phase winding, a B-phase winding, and a D-phase winding that are wound in the same manner in the slots of 720 ° to 1440 °. 1 is a configuration of the motor according to 1;
According to this configuration, the field excitation power can be supplied to the rotor side with a five-phase motor in which the winding of each phase of the coil end portion is small, and a motor with high productivity can be realized particularly in the stator winding. A rotor field winding motor can be reduced in size and cost, and a high-quality motor with small torque ripple can be realized.

(Invention according to Claim 12)
The invention according to claim 12 is a five-phase AC motor in which the phase order of the five-phase AC is the order of A-phase, B-phase, C-phase, D-phase, and E-phase, and the A-phase voltage is a sinusoidal voltage. The B phase voltage is a voltage similar to the A phase voltage having a phase difference of 72 ° with respect to the A phase voltage and the C phase voltage. The phase voltage is the same voltage as the A phase voltage having a phase difference of 72 ° with respect to the B phase voltage and the D phase voltage, and the D phase voltage has a phase difference of 72 ° with respect to the C phase voltage and the E phase voltage. The E-phase voltage is the same voltage as the A-phase voltage having a phase difference of 72 ° with respect to the D-phase voltage and the A-phase voltage, and the A-phase current is a sine wave. The B phase current is the same current as the A phase current having a phase difference of 72 ° with respect to the A phase current and the C phase current. Phase electricity The current is the same as the A-phase current having a phase difference of 72 ° with respect to the current and the D-phase current. The D-phase current is the same as the A-phase current having a phase difference of 72 ° with respect to the C-phase current and the E-phase current. 2. The motor structure according to claim 1, wherein the E-phase current is a current similar to the A-phase current having a phase difference of 72 ° with respect to the D-phase current and the A-phase current. .
According to this configuration, since the third harmonic voltage and the third harmonic current are superimposed on the voltage and current of each of the five phases in addition to the sine wave in the PWM control of the inverter, the sine wave voltage and the sine wave current are superimposed on each other. In comparison, the motor power output can be increased under the condition of the same peak value, and the motor can be made more efficient and smaller. And the torque ripple by the 3rd harmonic superposition does not generate | occur | produce theoretically.

(Invention according to Claim 13)
The invention described in claim 13 is such that the stator side AC field excitation current component necessary for energizing the field winding FM to the field winding FM is If2, and the energization to the power supply winding PSW is performed. By increasing and decreasing the field excitation current component If2, the field current If is increased or decreased, and a change in the voltage component induced by electromagnetic induction between the terminals of each phase winding of the stator is detected. The rotation position of the rotor is detected, and the motor according to claim 1.
According to this configuration, since the rotational position of the rotor can be detected from the windings of each phase, the sensorless operation of the motor is possible, and cost reduction and high reliability can be achieved. If the change in the circumferential magnetic impedance of the rotor is small, sensorless position detection may be difficult. In particular, in the case of starting from stationary or low-speed rotation, the case of large torque output, etc., this problem can be solved by the present invention.

  In the motor according to the present invention, the rotor field power is fed to the rotor side by utilizing the stator winding, so that the power factor of the three-phase current is high, and the motor can be reduced in size and cost. it can. The operation range is wide from starting from stationary, forward / reverse operation at low speed, and high speed rotation. Further, by optimizing the magnetic flux characteristics of the field and the accompanying torque current, it is possible to achieve high efficiency, miniaturization, and cost reduction of the motor. Further, by improving the winding structure to alleviate the problem of the complicated winding due to the above-mentioned torque current optimization measure, it is possible to improve the manufacturability of the winding and reduce the coil end portion.

3-phase 4-pole motor Speed-torque characteristics Stator winding layout Winding flux linkage Winding layout example Three-phase sine wave and power characteristics Rectangular voltage, current and power characteristics 4-pole rotor 3-phase current 5-phase 4-pole motor Stator winding layout 5-phase current 5-phase sine wave 5-phase 8-pole stator 5-phase 8-pole stator Teeth and slot shape Winding guide and stator 4-pole rotor 4-pole rotor 4-pole rotor Field current 5-phase vector 5-phase 8-pole motor 5 phase 12 pole motor Conventional motor with slip ring and brush Stator current 4-pole rotor Stator winding layout 3-phase 4-pole motor Stator winding layout Motor and drive circuit of the present invention Field power supply current If2 and field current If Rotor winding layout 3-phase 4-pole motor 3-phase 4-pole motor Stator winding layout 5-phase 4-pole motor Stator winding layout 5-phase 8-pole motor 4-pole rotor 5-phase motor voltage, current and power

  An object of the present invention is to achieve high efficiency of a motor, and to realize miniaturization and cost reduction of the motor. It is also necessary to reduce the complexity of the windings that appears as a negative effect of the high efficiency measures. The motor particularly targeted by the present invention is a synchronous motor that has a wide speed control range and performs so-called constant output control, and requires a large torque at a low speed. For example, it is a main motor of a hybrid vehicle, a main motor of an electric vehicle, and the like, and has a wide operating range from start-up from stationary, forward / reverse operation at low speed, and high speed rotation. Especially when climbing steep slopes, a large torque is required, which is important for equipment design. In many cases, the size of the motor is generally determined by loss, heat generation, and temperature rise at high torque at low speed rotation, and its characteristics are directly connected to the size of the motor and the cost of the motor. There are many similar needs for industrial motors.

  In addition, both high performance and low cost are necessary as the main motor needs of electric vehicles. For example, it is not just a motor configuration that can achieve high efficiency, but as will be described in detail later, it is possible that the rotor side field current can be detected from the stator side, and that the motor configuration can detect the sensorless position when stopped Is needed by. In addition, a stator optimization technique and a rotor optimization technique related to the present invention are also required.

  Embodiments of claims 1 and 2 will be described. FIG. 1 shows a cross-sectional view of a motor according to an embodiment of the present invention. The arrangement relationship of the stator windings at the coil end portion is added. This is a three-phase, four-pole field winding type synchronous motor. 1D is a stator core, and the U-phase winding is a U1-phase winding wound from 11 to 14 and a U2-phase winding wound from 17 to 1A. The V-phase winding is a winding wound from 13 to 16 and a winding wound from 19 to 1C. The W-phase winding is a winding wound from 15 to 18 and a winding wound from 1B to 12. Each of these windings has 12 teeth 1U of the stator on the circumference and is wound in each slot surrounded by these teeth. The stator windings in FIG. 1 are three-phase, four-pole, full-pitch winding, and concentrated winding. Each slot has a simple structure in which only one phase winding is arranged. Therefore, the productivity is excellent and the cost can be reduced. However, the torque ripple is likely to increase because of the simple structure.

  1E is a rotor, and 1F is a rotor shaft. 1V and 1X are N pole magnetic poles of the rotor, and 1W and 1Y are S pole magnetic poles of the rotor. The winding wound from 1H to 1P is a power receiving winding for receiving a magnetic field feeding magnetic flux supplied by the stator, the coil end connection is 27, and the winding has an electrical angle of 360 ° pitch. It has become. The winding wound from 1L to 1S is a power receiving winding for receiving the field feeding magnetic flux supplied by the stator, and the connection of the coil end portion is 28, and the winding has an electrical angle of 360 ° pitch. Thus, the receiving windings 1H and 1P have a phase difference of 180 ° in electrical angle. The windings wound around 1J and 1K are field windings for exciting the N pole magnetic pole 1V, and the connection of the coil end portions is 29. The winding wound around 1M and 1Z is a field winding for exciting the S pole magnetic pole 1W, and the connection of its coil end is 2A. The winding wound around 1Q and 1R is a field winding for exciting the N pole magnetic pole 1X, and the connection of the coil end portion is 2B. The winding wound around 1T and 1G is a field winding for exciting the S pole magnetic pole 1E, and the connection of its coil end is 2C.

  Next, FIG. 2 shows an example of characteristics of torque and power with respect to the rotational speed as an example of characteristics of the motor of FIG. The horizontal axis shows the rotational speed up to 12000 rpm, the left vertical axis shows the torque up to 120 Nm, and the right vertical axis shows the power up to 37.7 kW. 21 is a characteristic of the maximum torque of this motor, and the lower left side of 21 is an operable region. The operating point 24 is a point of a continuous rated torque of 40 Nm that can be continuously driven at a base rotational speed of 3000 rpm. Reference numeral 23 denotes a maximum torque point of 120 Nm, which is three times the continuous torque, and is a short-time rated torque. Reference numeral 22 denotes a maximum power characteristic indicating the power at each point of the maximum torque characteristic 21. For example, the operating point 23 is a torque point of 3000 rpm and 120 Nm, and the power at that time is 37.7 kW, which is the operating point 25 of power. The maximum power characteristic 22 is constant at 37.7 kW from 3000 rpm to 12000 rpm, and is also expressed as “constant output characteristic is 1: 4”.

The relationship among the motor torque T [Nm], the rotational angular velocity ω [rad], and the output power Po [W] is expressed by the following equation. It is a torque current [A] and Kt is a torque constant [Nm / A].
Po = ω × T (1)
= Ω × Kt × It (2)
On the other hand, when the electric power Pe [W] supplied to the motor is expressed as a DC motor and is represented by a motor voltage Vm and an induced voltage constant Kv [V / rad], the following equation is obtained.
Pe = Vm × It (3)
= Ω × Kv × It (4)
Since the electric power Pe and the output power Po are equal, it can be seen that the torque constant Kt and the induced voltage constant Kv are the same value when the equations (2) and (4) are compared. Here, the ideal state of the motor is assumed, and the winding resistance, iron loss, and mechanical loss are assumed to be zero.

Usually, in the range of 3000 to 12000 rpm, which is a constant output characteristic rotation speed range, the DC power supply voltage of the inverter is driven under constant conditions. Therefore, in order to keep the motor voltage Vm constant, the equations (3) and (4) Therefore, the induced voltage constant Kv is expressed by the following equation, and is a value inversely proportional to the rotational angular velocity.
Kv = Vm / ω (5)
As a result, the field current If that excites the field is proportional to the magnitude of the field magnetic flux and the induced voltage constant Kv, and thus has a value inversely proportional to the rotational angular velocity. Therefore, the field currents of the field windings 1J, 1K, 1M, 1Z, 1Q, 1R, 1T, and 1G shown in FIG. 1 are varied from the maximum value to 1/4 of the maximum value, and the rotation of the motor Along with the number, it is necessary to perform so-called field weakening control. In order to realize the output characteristics as shown in FIG. 2, the control of the field magnetic flux is an important technique. In the range of 3000 to 12000 rpm on the maximum power characteristic 22, the torque current It becomes a constant maximum current value from the equations (1) to (5). In reality, each equation is not uniform, but if the field current control can be freely performed at a base rotation speed of 3000 rpm or more, a region where the rotation control at the base rotation speed or more can be expanded. And a big merit occurs.

  Next, FIG. 3 shows a specific configuration and method in which field power is supplied from the stator side of the motor shown in FIG. 1, received at the rotor side, and supplied with field current. The winding 31 passes U1 phase current Iu1 through U1 phase windings 11 and 14 in FIG. 1, and the winding 32 passes U2 phase current Iu2 through U2 phase windings 17 and 1A in FIG. The winding 33 and the winding 34 are V-phase windings connected in series, and are the 13, 16 and 19, 1C windings of FIG. 1 and energize the V-phase current Iv. The winding 35 and the winding 36 are W-phase windings connected in series, and the W-phase current Iw is passed through the windings 15, 18 and 1B and 12 in FIG. Reference numeral 37 denotes a DC power supply voltage. Eight power control elements T1, T2, T3, T4, T5, T6, T7, and T8 are IGBTs and the like, and supply voltage and current to each of the phase windings. Each IGBT is provided with an antiparallel diode for reverse energization. A motor control unit 3X controls the motor, and controls the voltage and current of each IGBT. Also, field power supply control is performed. A field current is also detected from the detected values of the IGBT and the phase current. As the power control element, various elements such as MOSFET and SiC semiconductor can be used.

Here, the U1-phase winding 31 and the U2-phase winding 32 are connected in parallel, and the V-phase windings 33, 34 and the W-phase windings 35, 36 are connected in series. Is unbalanced. In order to eliminate this imbalance, the number of turns of the U1-phase winding 31 and the U2-phase winding 32 is set to be twice that of the other V-phase windings 33, 34 and W-phase windings 35, 36. . The U1 phase current Iu1 and the U2 phase current Iu2 are ½ current as shown in the following equation.
Iu = Iu1 + Iu2 (6)
By matching the U-phase current Iu in this way, the relationship can be made equivalent to the voltage and current of a normal three-phase star connection, and the relationship of the following equation can be maintained.
Iu + Iv + Iw = 0 (7)
When the number of turns of the U1-phase winding 31 and the U2-phase winding 32 is twice that of the other windings, the cross-sectional area of the winding can be halved. The total amount of phase copper wire is about the same. In FIG. 3, when the V-phase windings 33 and 34 and the W-phase windings 35 and 36 are respectively parallel windings, the U1-phase winding 31 and the U2-phase winding 32 have the same number of turns. The equation (7) can be maintained.

  In FIG. 3, the upper side from the alternate long and short dash line indicates the stator side, and the lower side from the alternate long and short dash line indicates the rotor side. The winding 38 is the receiving winding of 1H, 1P and 27 in FIG. 1, and the winding 3A is the receiving winding of 1L, 1S and 28 in FIG. These windings 38 and 3A are windings with a pitch of 360 ° and are arranged with a phase difference of 180 ° in electrical angle. FIG. 4 shows the magnetomotive force relationship acting on the windings 38 and 3A. The horizontal axis indicates the rotor circumferential direction in electrical angle. The vertical axis indicates the direction of magnetomotive force, and the upper side of the figure is the rotor direction on the inner diameter side. 4A shows a state in which a U-phase positive current is applied to the U-phase winding, the winding 41 is 11 in FIG. 1, the winding 43 is 14 in FIG. 1, and the winding 45 is 17 in FIG. Winding 47 is 1A in FIG. 1, and winding 49 is winding 41. If the current direction is the direction of the current symbol in the figure, a magnetomotive force in the direction of the arrow is generated. The electrical angle is an arrow in the opposite direction every 180 °, and in the range of 360 ° in the electrical angle, all 360 ° regions include one upward arrow and one downward arrow, and an electrical angle of 360 °. The total magnetomotive force of the range is zero. Therefore, the power receiving windings 38 and 3A are wound at an electrical angle of 360 ° and are not affected by the magnetomotive force generated by the current Iu of the U-phase winding shown in FIG. I understand. Similarly, the power receiving windings 38 and 3A are not affected by the V-phase current Iv and the W-phase current Iw.

  In the motor shown in FIG. 1, each winding pitch of the U, V, and W phase windings is an electrical angle of 180 °, and each winding cycle is an electrical angle of 360 °. When the U-phase winding is divided into a U1-phase winding and a U2-phase winding and used as a power supply winding, the winding period is 720 °. It can be said that the winding pitch of the power receiving windings 27 and 28 is an electrical angle of 360 °, and the winding period of these power receiving windings is an electrical angle of 720 °. However, each winding pitch of the U, V, and W phase windings of the three-phase concentrated winding motor of FIG. 35 is 120 ° in electrical angle, and the winding pitch of the feeding winding is the number of phases, poles, and windings of the motor. It depends on the format. The minimum condition of the winding period of the power supply winding is that the electrical angle is not 360 °, and there is no restriction on the winding pitch of the power supply winding. Further, if the winding pitch of the power receiving winding is an integer multiple of 360 ° (RN times) in electrical angle, it is at least unaffected by motor drive current such as three-phase current. The value RN of the winding pitch of the power receiving winding can be selected depending on the number of phases, the number of poles, the winding type of the power feeding winding of the target motor.

Next, field power feeding using the U1-phase winding 31 and U2-phase winding 32 in FIG. 3 and receiving by the receiving windings 38 and 3A will be described. Since the U1-phase current Iu1 and the U2-phase current Iu2 can be energized independently, it is possible to energize the feeding current If2, which is a field power component indicated by the 3G arrow. Specifically, the U1 phase current Iu1 and the U2 phase current Iu2 may be relatively increased or decreased with respect to the feeding current If2 which is a field power component, and the following equation is obtained from the relationship of the equation (6).
Iu1 = Iu / 2 + If2 (8)
Iu2 = Iu / 2-If2 (9)
The sum of the U1-phase current Iu1 and the U2-phase current Iu2 is not affected by the feeding current If2, and becomes the U-phase current Iu as shown in the following equation.
Iu1 + Iu2 = Iu / 2 + If2 + Iu / 2−If2 = Iu (10)
Further, the feeding current If2 of the field power component is expressed by the following equation as a difference between the U1-phase current Iu1 and the U2-phase current Iu2.
Iu1-Iu2 = 2 × If2

Note that the feeding current If2, which is a field power component, is an alternating current having a frequency Ff2 that enables electromagnetic coupling between the stator and the rotor like a transformer. On the other hand, the fundamental frequency Fu of the U-phase current Iu is expressed by the following equation using the number of revolutions Nr and the number of poles PN of the rotor.
Fu = PN / 2 × Nr / 60
Here, since there is a relationship between the AC frequency Ff2 of the current If2 of the field power component and the fundamental frequency Fu of the current Iu, it is necessary to consider. For example, Ff2 = (500 Hz) + Fu may be set. In addition, the waveform shape of the feeding current If2 that is an alternating current is reasonable if the rotor field current is a direct current and is a rectangular wave alternating current. However, there is no particular limitation, and since the circuit time constant of the field winding is large, the smoothing effect is great, and a sine wave, a triangular wave or the like is also possible.

  FIG. 4B shows only the feeding current If2 which is a field power component in the equations (8) and (9). The winding 301 is 11 in FIG. 1, the winding 302 is 14 in FIG. 1, the winding 303 is 17 in FIG. 1, the winding 304 is 1A in FIG. 1, and the winding 30C is the same as the winding 301. The directions of magnetomotive force due to the feeding current If2 which is a field power component are indicated by mesh arrows 30A and 30B. There is only one arrow in the range of electrical angle 360 °.

  FIG. 4C shows a magnetomotive force component that acts on the power receiving winding 38. The winding 305 is the winding 1H in FIG. 1, the winding 306 is 1P in FIG. 1, and the winding 307 is the same as the winding 305. The magnetomotive forces acting between the windings 305 and 306 are 30A, 42, and 44. Since the directions of 42 and 44 are opposite, they cancel each other, and only the magnetomotive force 30A remains. Only the power supply component of the field power component If2 shown in FIG. 4B is transmitted voltage and current by electromagnetic induction acting on the power receiving winding 38.

  Similarly, FIG. 4D shows a magnetomotive force component acting on the power receiving winding 3A. The winding 308 is the winding 1L in FIG. 1, and the winding 309 is 1S in FIG. The magnetomotive forces acting between the winding 308 and the winding 309 are 44, 30B, and 46. Since the directions of 44 and 46 are opposite, they cancel each other, and only the magnetomotive force 30B remains. Voltage and current are transmitted by electromagnetic coupling in which only the feeding component of the field power component If2 shown in FIG. 4B is applied to the power receiving winding 3A. Thus, by making the power receiving windings 38 and 3A into windings with an electrical angle of 360 ° pitch, that is, windings with an electrical angle of 720 ° cycle, the electrical angle is normally 180 ° pitch with a 360 ° cycle. The three-phase AC current component is prevented from being transmitted, and the electric field power can be transmitted from the stator side to the rotor side by creating a power supply configuration with an electrical angle of 360 ° pitch and a period of 720 °. The winding 31 and the winding 32 are both U-phase stator windings, but the U1-phase winding 31 arranged between an electrical angle of 0 ° and 360 ° and an electrical angle of 360 ° to 720 °. It is divided into U2-phase windings 32 arranged between each of them so that each current can be passed, and a current component having an electrical angle of 720 ° cycle can be passed. The U1-phase winding 31 and the U2-phase winding 32 are windings for passing a U-phase current component, and at the same time, are feeding windings as means for feeding field power by the feeding current If2.

Next, the voltage and current components transmitted to the power receiving winding 38 are connected to a full wave rectifier 39, and the power receiving winding 3A having a phase difference of 90 ° in mechanical angle and 180 ° in electrical angle is supplied to the full wave rectifier 3B. The output terminals of both full-wave rectifiers are connected in series. The outputs of both of these full-wave rectifiers are connected to the field windings 3C, 3D, 3E, and 3F that are connected in series with the current direction aligned, and energize the field current If. The winding 3C is the windings 1J and 1K in FIG. 1, the winding 3D is the windings 1Z and 1M in FIG. 1, the winding 3E is the windings 1Q and 1R in FIG. 1, and the winding 3F is the figure. 1 1G and 1T windings. The reason why two sets of the power receiving winding and the full wave rectifier are used is to allow field power to be supplied to the field winding at any rotational position. Further, at the moment when the output voltage of the full-wave rectifiers 39 and 3B decreases, the field current If is energized by the flywheel action via the four diodes of the full-wave rectifier. When it is desired to reduce the voltage drop of these four diodes, a diode 3G is added and connected as shown by a broken line to reduce the voltage drop of the diode part when the field current If is applied to the flywheel. It is possible to reduce the voltage drop of one diode.

  Since the field current If can be maintained for a certain period of time by a flywheel action via a diode, simplification can be made. Conversely, three power receiving windings can be used to form a three-phase rectifier. The form of the full-wave rectifiers 39 and 3B can be modified, and is not limited to a full-wave rectifier using four diodes. For example, FIG. 11 shown later shows an example in which the output terminals of the full-wave rectifiers 39 and 3B are connected in series. Use of other diodes such as series, parallel, or half-end rectification, depending on the voltage waveform, current waveform and electromagnetic coupling characteristics such as feeding winding and receiving winding to supply field power from the stator side, A rectification function can also be realized in combination. A rectifier using these diodes is not shown in FIG. 1, but is built in the rotor and fixed.

  Next, the three-phase AC voltage of the motor, the magnitude and frequency of the current, and the magnitude and frequency of the field power component If2 will be clarified, and the contents of the present invention will be made more concrete. As shown in FIG. 2, the motor shown in FIGS. 1 to 4 is a motor having a maximum output of 37.7 kW and a maximum torque of 120 Nm. Since the motor has four poles, the frequency Fu of the motor induced voltage is 400 Hz at 12000 rpm, which is a frequency proportional to the rotational speed. On the other hand, the output power of the field power component If2 is normally a resistance of the field winding of the rotor, which is about several hundred W, which is a small value compared to the motor output. As shown in FIG. 25, when power is supplied using a brush and slip ring, the space, cost, life, and reliability of the device become problems, but the original stator winding as shown in FIGS. If it is a method of utilizing, the burden of cost etc. is small. Further, the frequency Ff2 of the feeding current If2 that is the field power component can be freely selected, and there is no principle restriction. For example, as described above, Ff2 = (500 Hz) + Fu may be used. When the motor rotates at high speed, there is a problem of the frequency limit of the inverter. In such a case, conversely, the frequency Ff2 of If2 can be made lower than the motor frequency Fu. The frequency, voltage, and current of the feed current If2 can be designed for losses such as iron loss of the motor and the convenience of the control side.

  The method for feeding and receiving field power described with reference to FIG. 3 basically forms a transformer and supplies power from the stator side to the rotor side. The AC power supplied to the secondary side is rectified and the field current If is supplied to the field windings 3C, 3D, 3E, and 3F. Therefore, it is possible to efficiently supply only the electric power corresponding to the resistance loss of the field winding necessary for the rotor side. The feeding current If2 of the stator is sufficiently smaller than the respective phase currents of the motor.

  In the case of the conventional synchronous motor IPMSM with built-in magnet, the d-axis current, which is the field current component that excites the field magnetic flux, is superposed on the q-axis current, which is the torque current component. Since the d-axis current increases due to effects such as magnetic saturation, the combined current of the d-axis and q-axis increases, and the copper loss of the stator increases. It can be said that the power factor decreases and the reactive current for exciting the field magnetic flux increases. In this regard, in the configuration of FIG. 3, it is sufficient to supply a small amount of electric power corresponding to the copper loss of the field winding necessary for exciting the field magnetic flux, so that the stator current at the time of large torque output is greatly reduced. it can. As a result, motor efficiency can be improved, and downsizing and cost reduction are possible.

  In addition, the conventional synchronous motor IPMSM with built-in magnet performs field strengthening control and field weakening control in a high-speed rotation region higher than the base rotation speed. It is difficult to perform the increase / decrease control, and there is also a problem that the power factor is likely to decrease when a large torque is output. In this regard, in the configuration of FIG. 3, it is possible to supply the field power to the field winding of the rotor and functionally supply the field current through a path different from the control of the three-phase current. Therefore, the field magnetic flux can be controlled by increasing / decreasing accurately and without waste. As a result, motor field weakening control, field strengthening control, and motor constant output control can be realized accurately and efficiently. As specific motor characteristics, a variable speed range in a high-speed rotation region equal to or higher than the base rotation speed can be widened. That is, it is important to achieve both a large torque at low speed rotation and an expansion of the variable speed range. Also, in order to reduce copper loss in the rotor field winding, it is possible to use a permanent magnet in the rotor and to superimpose a field excitation current d-axis current on the stator winding. can do. Therefore, it is possible to design for optimum operation, high efficiency, and high performance using the permanent magnet, the d-axis current, and the field current If in combination, depending on the application and operating conditions.

  Further, in the motor of the present invention, the power factor can be greatly improved especially in the maximum torque region, and the reactive current can be greatly reduced, so that the current capacity of the inverter can be greatly reduced. Therefore, it is possible to reduce the size and cost of the inverter.

  In the conventional method shown in FIG. 27, the stator winding can supply field magnetic flux and magnetic flux energy, but cannot supply power to the rotor side like a transformer. The rotor field windings 27B, 27C, 27D, 27E and the diode 27A act to maintain the field magnetic flux, but the field current If itself cannot be increased directly. As a result, as described above, when the motor outputs a large torque, the burden of copper loss of the stator winding, the burden of the current of the inverter, the vibration due to the intermittent current, the problem of noise, and the like increase.

  The burden on the inverter shown in FIG. 3 will be described. The field power component If2 is differentially generated and supplied by the four IGBTs T1, T2, T3, and T4. It is a relationship of (8) and (9) Formula, and If2 is relatively small. Further, T1, T2, T3, and T4 of the four IGBTs also generate U-phase voltage and current, and the U-phase current Iu has the relationship of the expressions (6), (8), and (9). Therefore, the current capacities of T1, T2, T3, and T4 of the four IGBTs may be about ½ of the current capacities of T5 and T6 of the V-phase drive IGBT, and the number of U-phase drive IGBTs is four. It will increase, but the cost and space burden will not be doubled. In particular, in an inverter having a large output capacity to some extent, the IGBTs are often used in parallel connection. In such a case, the inverter configuration of FIG. 3 can be configured only by changing the connection relationship of the IGBTs. In that case, the cost burden is small.

The field power component If2 can be supplied by various modifications from the configuration shown in FIG. If a component having a period of 720 ° in terms of electrical angle is generated, not only the windings of the same phase, the power receiving windings 38 and 3A in FIG. 3 can receive the electric power. For example, the windings 31, 33, 35 are windings between 0 ° and 360 °, and the windings 32, 34, 36 are windings between 360 ° and 720 °, so these six windings. If the field power component can be superimposed on only one of the windings, a component having a period of 720 ° in electrical angle is created. It can also be superimposed on a plurality of windings. As another example, a field power component 53 can be added as shown in FIG. The winding 51 is the windings 11 and 14 in FIG. 1, and the winding 52 is the windings 17 and 1A in FIG. Current 54 is U-phase current Iu. These various modifications are also included in the present invention. Embodiments of the present invention different from those shown in FIGS. 1, 3, and 4 will be described later.

  Next, a specific configuration relating to the distribution of the field magnetic flux shown in FIG. First, the conventional general three-phase sine wave alternating current and the rectangular wave driving realized by the present invention will be described. FIG. 6 is a diagram showing the voltage and current of the three-phase sine wave alternating current and the power at each time point. The horizontal axis indicates an electrical angle of 0 ° to 360 °. The vertical axis in the upper diagram of FIG. 6 represents the magnitude of voltage and current. The lower diagram of FIG. 6 is synchronized with the upper diagram, and its vertical axis is power. Here, 61 is a U-phase induced voltage and applied current, 62 is a V-phase induced voltage and applied current, 63 is a W-phase induced voltage, and applied current. Assuming there is a case where the power factor is 1, the power of the U phase is 64, the power of the V phase is 65, the power of the W phase is 66, and the power of each phase is double frequency. Each average value is 0.5. The total power of the three phases is 67 to 1.5, which is always a constant value. These show the case where the voltage and current are three-phase sinusoidal alternating current.

  Now, if the rate of effective utilization of the electric wire is referred to as the electric wire utilization rate, since the waveform rate of the sine wave is 0.707, the power is 0.707 squared, and the electric wire utilization rate is 0.00. 5 and a small value in comparison with the utilization factor 1.0 of the DC voltage and DC current. Further, from the viewpoint of an IGBT, which is a semiconductor power element, it is possible to supply power most effectively in the case of a DC voltage and a DC current, and assuming that the drive utilization rate in that case is 1.0, as shown in FIG. The utilization factor of the IGBT in driving a sine wave voltage and sine wave current is 0.5. In order to realize high efficiency, miniaturization, and low cost of a motor system including driving, it is effective to improve the utilization rate of the IGBT.

  FIG. 7 is a diagram showing trapezoidal alternating current close to rectangular alternating current, and the horizontal axis indicates an electrical angle of 0 ° to 360 °. The vertical axis in the upper diagram of FIG. 7 represents the magnitude of voltage and current. The lower diagram of FIG. 7 is synchronized with the upper diagram, and its vertical axis is power. If 71 is an induced voltage and an applied current, the power is the square of 71 and 72 is. When the width of θrr which is the inclined portion of 71 becomes smaller, it approaches the rectangular wave alternating current, and the power average value of 72 approaches 1.0. Comparing 72 with the U-phase power 64 of the sine wave voltage and sine wave current of FIG. 6, it can be seen that the average value of 72 can be made larger. At present, motor drive theory premised on sine waves is the mainstream, and in control theory, addition and subtraction of sine wave vectors have been developed to a high degree in the d-axis and q-axis theories of rotational coordinates. It is used. On the other hand, the rectangular wave driving and trapezoidal wave driving described in the present invention are slightly different, and include many harmonic components in terms of frequency, and each of them needs to be individually considered. Further, even if the three-phase sinusoidal alternating current is a multi-phase of four or more phases, there are few principle merits, and three phases are the mainstream from the merits of simplification. The rectangular wave driving and trapezoidal wave driving of the present invention are not limited to three phases, and the degree of freedom increases in the case of four or more phases. In addition, motors that pursue miniaturization are often used in a magnetically nonlinear saturation region, and the motor shape, cost, etc. are often determined by the saturation characteristics in the large torque region of the motor. Therefore, unlike the linear theoretical viewpoint, it is also a problem of how to efficiently and effectively use a magnetic material having magnetic saturation characteristics.

  Next, FIG. 8 shows a rotor configuration capable of more effectively realizing the motor induced voltage of FIG. 7 and will be described. The magnetic poles 1V, 1W, 1X, and 1Y of the rotor 1E in FIG. 1 do not show the detailed rotor magnetic pole structure. When each field winding is wound around a structure in which electromagnetic steel sheets are laminated and a field current is applied, a field flux with a uniform magnetic flux distribution is generated on the surface of each rotor magnetic pole, and the induced voltage of each phase winding is It becomes a waveform like 71 of FIG. However, when a torque current is applied to each phase winding, the magnetic flux on each rotor magnetic pole surface is biased to one side due to the armature reaction, and the uniformity of the magnetic flux distribution is impaired. Therefore, the trapezoidal flat portion shown in 71 of FIG. 7 has an induced voltage of each phase winding is deformed to generate a portion having a large value and a portion having a small value. In order to reduce this problem, the rotor of FIG. 8 is provided with slit-like gaps 83 in each rotor magnetic pole. The circumferential magnetic flux component generated by the magnetomotive force acting in the circumferential direction of the rotor magnetic pole by the torque current is reduced by the slit-shaped gap 83. The magnetic flux component in the direction of the rotor magnetic pole acts on the stator side through an elongated magnetic path 84 sandwiched between the gaps 83.

  81 is an N pole magnetic pole of the rotor, and 82 is an S pole magnetic pole of the rotor. 88 and 8E, 8B and 8J are field power receiving windings, 87 and 8A, 8F and 8K are field windings that excite the N pole magnetic pole of the rotor, 8D and 8C, 89 and 8H are S of the rotor. It is a field winding for exciting the poles. Since the S pole and the N pole are connected in series, the field winding can excite the field magnetic flux even with the field winding on only one side. For example, the S pole field winding 8D, 8C, 89, and 8H can be removed. The slit-shaped gap 83 may be configured to be filled with a nonmagnetic material such as resin.

  Reference numeral 85 in FIG. 8 denotes a rotor outer periphery connecting portion necessary for obtaining the strength of the rotor, and also has an effect of suppressing the pulsation of the magnetic flux density on the rotor outer periphery in the arrangement period of the gap portion 83. However, if the width of the rotor outer periphery connecting portion is large, there is a problem that the circumferential magnetic flux passing therethrough increases and the rotor characteristics deteriorate. The dimensions are important values for motor characteristics. In order to obtain the strength against the centrifugal force of the rotor, the outer periphery of the rotor can be connected as shown by 185 in FIG. 18, and the outer periphery of the rotor can be circular. Further, in order to increase the strength of the rotor against centrifugal force, the outer periphery of the rotor may be covered with a reinforced plastic FRP such as carbon fiber. Further, reference numeral 86 in FIG. 8 denotes a rotor internal connecting portion necessary for obtaining the strength of the rotor. However, if the width of 86 is wide, the magnetic flux passing therethrough increases, and the effect of the slit-shaped gap 83 is reduced. Therefore, the width dimension of the rotor internal joint 86 is an important parameter for the rotor characteristics.

  Next, an embodiment of claim 4 will be described. In the rotor shown in FIG. 18, permanent magnets 182, 183, 186, 187 and the like are added to the four-pole rotor shown in FIG. 81 is an N pole magnetic pole of the rotor, and 82 is an S pole magnetic pole of the rotor. The magnetization direction of these permanent magnets is the magnetic pole direction of each rotor. For example, the permanent magnets 182 and 183 have an inner diameter direction of S pole and an outer diameter direction of N pole, and the permanent magnets 186 and 187 have an inner diameter direction of N pole and an outer diameter direction of S pole. Since these permanent magnets are excited in the direction of the magnetic poles of the rotor, there is an electromagnetic relationship in series with the excitation of field magnetic flux by field current such as field windings 87, 8A and 8C, 8D. Therefore, the permanent magnet assists the excitation of the field magnetic flux by the field current of the field winding.

  83 and the like are slit-like gaps, and 84 and the like are elongated magnetic paths sandwiched between the gaps. Reference numeral 86 denotes a rotor internal joint portion necessary for obtaining the strength of the rotor, but it is preferable that the width of the rotor internal joint portion is small in view of the electromagnetic characteristics of the rotor.

  189, 18E and the like are soft magnetic bodies between the elongated magnetic path 84 and the permanent magnet, and have an effect of widely dispersing the magnetic flux passing through the elongated magnetic path 84 in the circumferential direction. However, if the width of 189, 18E, etc. is too wide, the field magnetic flux moves in the circumferential direction due to the armature reaction or the like and changes, so it is necessary to make it an appropriate narrow width.

  The soft magnetic material around the permanent magnet serves as a magnetic path for a magnetic flux component in which a part of the magnetic flux generated by the permanent magnet is not supplied to the stator side. That is, when the field current is zero, part of the magnetic flux generated by the permanent magnet passes as leakage magnetic flux. Specifically, they are the circumferential magnetic paths 18C, 184, and 18D of the permanent magnet, the magnetic paths 18A and 18B on the outer diameter side of the permanent magnet, and the magnetic paths 181 and 185 on the outer diameter side of the slot. Some of these leakage fluxes are not necessarily harmful because they change the direction of the magnetic flux by the exciting current of the field winding and change to a magnetic flux that can be effectively used for the stator. These narrow magnetic paths of the soft magnetic material can be designed with a magnetic circuit considering the magnetic saturation characteristics of the soft magnetic material. That is, the width of each part can be designed in consideration of the necessity for the strength of the rotor and the magnetic characteristics.

  Further, the portions of the magnetic paths 181 and 185 on the outer diameter side of the slot in FIG. 18 work effectively in terms of rotor strength. However, there is also a problem that leakage flux increases. On the other hand, the rotor configuration in FIG. 8 shows an example in which the slots of the rotor are open. This slot opening can be used when winding the winding into the slot of the rotor. From the viewpoint of the characteristics of the rotor field, it is advantageous that the slot is opened. These configurations are also design choices. Further, although two permanent magnets are arranged for each magnetic pole of the rotor, it can be divided into one, three, four, or five.

  Next, FIG. 19 shows an example of another rotor using a permanent magnet. This is a structure in which permanent magnets 191, 192, 193, 194 are added to the distal end portion on the outer diameter side of the slit-shaped gap 83 of the rotor of FIG. 8. An enlarged view of the upper left part of FIG. 19 is shown in FIG. The magnetization direction of these permanent magnets is the magnetic pole direction of each rotor. For example, the permanent magnets 191, 192, 193, 194 have an inner diameter direction of S pole and an outer diameter direction of N pole. Since these permanent magnets are excited in the direction of the magnetic poles of the rotor, there is an electromagnetic relationship in series with the excitation of field magnetic flux by field current such as field windings 87, 8A and 8C, 8D. Therefore, the permanent magnet assists the excitation of the field magnetic flux by the field current of the field winding. Magnetic characteristics can be easily understood because the magnetic flux passing through the elongated magnetic paths 202, 203, and 204 sandwiched between the slit-shaped gaps 201 can be considered and designed for each magnetic path. For example, the magnetic flux passing through the elongated magnetic path 202 is sandwiched between the permanent magnets 191 and 192 and both permanent magnets, and passes through the narrowed soft magnetic body.

  The portion of the magnetic path 205 on the inner diameter side of the permanent magnets 191 and 192 has a shape in which the magnetic flux passing through the elongated magnetic path 202 sandwiched between the slit-shaped gap portions 201 is spread so as to easily pass through the permanent magnet 191. When the field current is zero, the soft magnetic body portion 207 having a narrow width between the permanent magnets 191 and 192 passes a part of the magnetic flux generated by the permanent magnet as a leakage magnetic flux and makes a round in the rotor. Magnetic flux. The broken lines with arrows in FIG. 20 indicate the passage of magnetic flux. At this time, the leakage magnetic flux is a magnetic flux component that does not contribute to the torque generation of the motor because it does not pass through the stator side. When the field current increases, the leakage magnetic flux gradually decreases, and the magnetic flux passing from the rotor side to the stator side increases. The width of the soft magnetic body portion 207 can be designed to take into account the magnetic saturation characteristics of the soft magnetic body. That is, when the field current is zero, the permanent magnet moderately assists the field current and field magnetic flux, and when a large field magnetic flux is required, the desired magnetic flux is generated by the large magnetic field current and the permanent magnet. Magnetic characteristics that can be generated in The characteristics of the permanent magnet, the shape of the permanent magnet, and the width of the soft magnetic body 207 are designed. The effects of these permanent magnets 191, 192, 193, 194 are a reduction in the field current capacity and a reduction in copper loss due to the field current. Since the field current can be reduced, the motor can be highly efficient and downsized.

  In addition, the permanent magnets shown in FIGS. 18, 19, and 20 produce field magnetic fluxes in an auxiliary manner. Therefore, since it can be used in a state where the coercive force is relatively small, the magnet characteristics can be improved and the field magnetic flux can be increased by magnetizing and magnetizing the permanent magnet. Conversely, at high speed rotation, by weakening the characteristics of the permanent magnet, the field magnetic flux can be weakened and the constant output characteristics can be easily realized. In particular, in the motor of the present invention, both field excitation using the field winding of the rotor and field excitation energizing the stator winding can be performed simultaneously, and the magnetizing ability is high. It is possible to vary the magnet characteristics using these currents even in the operating state of the motor.

  In the present invention, various rotors as shown in FIG. 40 can also be used. In FIG. 40, a semicircular shape of the rotor cross section is written, and in (b) and (c), the rotor shape is shown by a quadrant. In FIG. 40A, 370, 371, 372, and 373 are elongated slits, and the slits of 370, 372, and 373 are slits that are roughly connected between the rotor magnetic poles. It is a slit to the vicinity of the center. With this rotor structure, the rotor strength near the slit 371 can be secured.

  In the rotor of the left quadrant of FIG. 40B, the slits 374, 375, and 376 are structured from the rotor surface side toward the rotor center. Assuming that the d-axis is the direction of the rotor magnetic pole and the slit 375 is facing, there is an effect of shielding the magnetic flux in the q-axis direction in the vicinity of the rotor surface. There is no slit near the center of the rotor, and there is no magnetic flux shielding effect in the q-axis direction. The strength of the rotor can be sufficiently secured. In the left quadrant rotor of FIG. 40B, 37B permanent magnets are added in addition to the slits of 37C, 37D, and 37E. The magnetic flux magnet can be supplied evenly to the surface of the rotor magnetic pole.

  In the left quadrant rotor of FIG. 40C, 378 is a slit and 377 is a permanent magnet. There is a soft iron portion at the center of the rotor magnetic pole, and it is easy to pass the electromagnet magnetic flux due to the field current. The rotor of the right quadrant in (c) of FIG. 40 has a structure for supplying magnet magnetic flux in the direction of the magnetic poles of the rotor. In the rotor of FIG. 40 (d), ring-shaped or pipe-shaped permanent magnets are arranged on the outer periphery of the rotor. In particular, in a small motor, such a circular magnet in which a resin and a rare earth magnet are mixed and hardened is widely used. The strength itself is high, and the strength can be obtained by fixing to the rotor with an adhesive. The portion 378 of the permanent magnet is magnetized to the N pole, and the portion 379 is magnetized to the S pole. It is possible to vary the rotor field by increasing and decreasing the field current of the rotor by causing the permanent magnet and the magnetomotive force of the field winding to overlap each other. As in these examples, it is possible to realize a motor having a high overall capability by combining the capability of the permanent magnet with the capability of the field current.

  Next, an embodiment of claim 5 will be described with reference to FIG. The horizontal axis is the electrical angle of the rotation angle, and the vertical axis is the magnitude of the current. The induced voltage waveform of the three-phase motor of the present invention is a voltage waveform as shown in FIG. 7, and the U-phase, V-phase, and W-phase voltages Vu, Vv, and Vw have a phase difference of 120 ° in terms of electrical angle. ing. In the region of the width of θrr that is the slope of the induced voltage in FIG. 7, the induced voltage variation is large, and it is difficult to obtain the torque value as expected when a torque current is applied during this period. 91 in FIG. 9 is an example of a U-phase current waveform. The current waveform is a combination of a positive trapezoidal wave and a negative trapezoidal wave. While the U-phase current Iu is changing from a positive value to a negative value, the current value is zero or small, and the width of the section is θhz × 2. This θhz is set to a value slightly larger than ½ of θrr in FIG. 7, and the current is set to a small value close to zero in a section where the induced voltage change is large so that torque fluctuation does not increase. θst is a rotation angle at which the current changes from the value of the upper side of the trapezoid to zero. θht is a value that is 1/2 of θst, and is a rotation angle at which the current value changes in a trapezoidal shape by 1/2 as shown in FIG. Similarly, 92 is a V-phase current waveform, 93 is a W-phase current waveform, and the relative phase difference is 120 ° in electrical angle. Here, it is assumed that the trapezoidal current waveform of each phase is symmetrical. In addition, it is assumed that the current waveforms of the U, V, and W phases have the same shape and the phases are different by 120 °.

Next, the conditions under which the power of the three-phase motor is constant will be described. The motor output power Pm is represented by the following equation when idealized with the internal motor loss such as copper loss due to winding resistance being zero.
Pm = Vu × Iu + Vv × Iv + Vw × Iw (11)
First, the electrical angle in the section where the voltage fluctuation of the U-phase voltage Vu is large will be confirmed with reference to FIG. In the section from (180 ° −θhz) to (180 ° + θhz), since the U-phase current Iu is zero, it is necessary to output the full power of the motor in the V-phase and the W-phase. The V-phase voltage Vv is 1, the V-phase current Iv is 1, the W-phase voltage Vw is -1, the W-phase current Iw is -1, and the motor power Pm is 2 from the equation (11). Similarly, in the section from (180 ° + θhz) to (240 ° −θhz), the V-phase voltage Vv is 1, the V-phase current Iv is 1, and the U-phase voltage Vu and the W-phase voltage Vw are −1, so (Iu + Iw ) Is −1, the motor power Pm in equation (11) can be set to 2. In the interval from (240 ° −θhz) to (300 ° + θhz), the U-phase voltage Vu is −1, the U-phase current Iu is −1, and the V-phase voltage Vv and the W-phase voltage Vw are 1, so (Iv + Iw) Is 1, the motor power Pm in the equation (11) can be set to 2. In the interval from (300 ° + θhz) to (360 ° −θhz), the W-phase voltage Vw is 1, the W-phase current Iw is 1, and the U-phase voltage Vu and the V-phase voltage Vv are −1, so (Iu + Iv) is − If it is 1, the motor power Pm in the equation (11) can be set to 2. For the section from (360 ° −θhz) to 360 °, the U phase current Iu is zero, the V phase voltage Vv is −1, the V phase current Iv is −1, the W phase voltage Vw is 1, and the W phase current Iw. Is 1, and the motor power Pm is 2 from the equation (11). For the section from 0 ° to (0 ° + θhz), the U-phase current Iu is zero, the V-phase voltage Vv is −1, the V-phase current Iv is −1, the W-phase voltage Vw is 1, and the W-phase current Iw is 1 and the motor power Pm is 2 from the equation (11). In the interval from (0 ° + θhz) to (60 ° −θhz), the V-phase voltage Vv is −1, the V-phase current Iv is −1, and the U-phase voltage Vu and the W-phase voltage Vw are 1. Therefore, (Iu + Iw) is If it is 1, the motor power Pm in the equation (11) can be set to 2. In the interval from (60 ° −θhz) to (120 ° + θhz), the U-phase voltage Vu is 1 and the U-phase current Iu is 1, and the V-phase voltage Vv and the W-phase voltage Vw are −1, so (Iv + Iw) is If it is -1, the motor power Pm of the formula (11) can be set to 2. In the interval from (120 ° + θhz) to (180 ° −θhz), the W-phase voltage Vw is −1, the W-phase current Iw is −1, and the U-phase voltage Vu and the V-phase voltage Vv are 1, so (Iu + Iv) is If it is 1, the motor power Pm in the equation (11) can be set to 2. The above has described all the sections in FIG.

Here, let us consider the conditions under which the motor power Pm in equation (11) can be set to 2 in the current waveform of FIG. Considering the point from 96 in FIG. 9 to 180 °, it is the slope of the induced voltage in FIG. 7 that is the reference for the value of θht, that is, the value of θst / 2 and the value of θhz, which is the rotation angle of current increase / decrease. It is a value of 1/2 of the rotation angle of θrr. If the sum of these values is smaller than 30 °, each phase current can be controlled as shown in FIG. 9, and the motor power Pm in the equation (11) can be set to 2.
(Θrr / 2 + θst / 2) ≦ 30 ° (12)
(Θrr + θst) ≦ 60 ° (13)

  (Theta) rr of (13) is a value resulting from the shape of a motor induced voltage, and there exists a limit on motor design in making this value small. Since θst is a variable time of current and is determined by the controllability of current, for example, there is a time margin for low-speed rotation, so it can be set to a small value and close to zero. In the actual control, the U-phase current waveform 91 is a current waveform in which the trapezoid changes smoothly like the 97 current waveform. Therefore, there is an ambiguity in the evaluation criteria of equation (13). From the practical viewpoint of the motor, in the current waveform as shown in FIG. 9, the range of the current waveform having a value up to 80% of the current peak value is regarded as the upper side of the trapezoidal shape. From a practical viewpoint, this is a standard that partially allows 20% torque fluctuation. Due to the partial variation, the average torque decrease is 10% or less.

  Further, as described above, the angle width at which the current value of each phase takes the positive and negative peak values is 120 ° + (the width θhz × 2 of the section where the voltage fluctuation is large) in electrical angle. Then, for each phase (width θhz × 2 of a section where voltage fluctuation is large), the current value of that phase is set to zero or a small value that does not cause any harmful effects, and power is output in the other two phases. The other angle region is an angle region where the current values of the two phases change between zero and the peak value, and the absolute value of the sum of the current values of the two phases is the peak value of the phase current. Therefore, there is a degree of freedom in taking each value of the current values (Iu + Iv) = 1 of these two phases, and the shape shown in FIG. 9 can be modified. A trapezoidal shape with a smaller change rate and a smooth corner shape can also be used.

Next, an embodiment of claim 6 will be described. FIG. 10 shows a cross-sectional view of a 5-phase, 4-pole motor. Arrangement of the stator windings of the coil end portion, so complicated, to explain the example of 8-pole later shown in FIG. 14. A phase to the scan stator, B phase, C phase, D phase, by winding a full-pitch winding in the E-phase. The A phase windings are the A1 phase windings 101 and 106, and the A2 phase windings 10B and 10G, and are 0 ° and 180 ° at the electrical angle position. B-phase windings are 103 and 108, and 10D and 10J, and are 72 ° and 252 ° in electrical angle positions. The C-phase windings are 105 and 10A, and 10F and 10K, and are 144 ° and 324 ° in electrical angle positions. The D-phase windings are 107 and 10C and 10H and 102, and are 216 ° and 36 ° in electrical angle positions. The E-phase windings are 109 and 10E and 10K and 104, and are 288 ° and 108 ° in electrical angle positions. The rotor has the same configuration as the rotor shown in FIGS. Moreover, the rotor of the structure shown in FIG. 1, FIG. 8 is also applicable. 19 and 20 will be described in detail later.

  FIG. 11 is a diagram showing a method of connecting the windings of the motor of FIG. 10, a method of applying voltage and current, a method of supplying field power from the stator to the rotor, and a method of energizing the field current. Although the diagram of the three-phase motor shown in FIG. 3 has been edited for a five-phase motor, the concept of supplying field power from the stator side to the rotor side using the stator winding is the same, and the configuration is similar. ing. The winding 111 energizes the A1 phase current Ia1 with the A1 phase windings 101 and 106 in FIG. 10, and the winding 112 energizes the A2 phase current Ia2 with the 10B and 10G A2 phase windings in FIG. The winding 112 and the winding 11C are B-phase windings connected in series, which are both the windings 103, 108, 10D, and 10J in FIG. 10, and energize the B-phase current Ib. The winding 113 and the winding 11D are C-phase windings connected in series, and are both the windings 105, 10A, 10F, and 10L in FIG. 10 and energize the C-phase current Ic. The winding 114 and the winding 11E are D-phase windings connected in series, and are both windings 107, 10C, 10H, and 102 of FIG. 10, and energize the D-phase current Id. Winding 115 and winding 11F are E-phase windings connected in series, and are both 109, 10E, 10K, and 104 windings of FIG. 10, and energize E-phase current Ie. Reference numeral 37 denotes a DC power supply voltage, and 10 power control elements 116, 117, 118, 119, 11H, 11J, 11K, 11L, 11M, 11N, 11P, and 11Q are IGBTs, etc. Supply voltage and current. Each IGBT is provided with an antiparallel diode for reverse energization. 3Z is a motor control unit that controls the motor, and controls the voltage and current of each IGBT. Also, field power supply control is performed. A field current is also detected from the detected values of the IGBT and the phase current.

Here, the A1 phase winding 111 and the A2 phase winding 11B are connected in parallel, and the B phase windings 112 and 11C and the C phase windings 113 and 11D, the D phase winding 114, The 11E and E phase windings 115 and 11F are connected in series, and the A phase winding is unbalanced. In order to eliminate this imbalance, the number of turns of the A1-phase winding 31 and the A2-phase winding 32 is set to twice the number of turns of the windings of the other phases. Then, the A1 phase current Ia1 and the A2 phase current Ia2 are ½ of the current as shown in the following equation.
Ia = Ia1 + Ia2 (14)
By matching the A-phase current Ia in this way, the relationship can be made equivalent to the voltage and current of a normal five-phase star connection, and the relationship of the following equation can be maintained.
Ia + Ib + Ic + Id + Ie = 0 (15)
Note that when the number of turns of the A1-phase winding 111 and the A2-phase winding 11B is set to be twice that of the other windings, the cross-sectional area of the winding can be halved. The total amount of phase copper wire is about the same. Further, in FIG. 11, when the B phase windings 112 and 11C and the C phase windings 113 and 11D, the D phase windings 114 and 11E, and the E phase windings 115 and 11F are respectively parallel windings, Since the A1 phase winding 111 and the A2 phase winding 11B have the same arrangement relationship, the number of turns of the A1 phase winding 111 and the A2 phase winding 11B is the same as the number of turns of the other phase winding. ) Expression is maintained.

In FIG. 11, the upper side from the one-dot chain line indicates the stator side, and the lower side from the one-dot chain line indicates the rotor side. 11 is exactly the same as the lower side from the alternate long and short dash line in FIG. Further, the electromagnetic coupling of the magnetic flux components shown in FIG. 4 realizes the feeding and receiving of field power, and the relationship between them is the same. Next, the feeding of field power using the A1 phase winding 111 and the A2 phase winding 11B of FIG. 11 and the power reception of the power receiving windings 38 and 3A will be described. Since the A-phase currents Ia1 and Ia2 can be energized independently, the field power component If2 indicated by the arrow 11R can be energized. Specifically, the field power component If2 may be relatively increased or decreased relative to the A1 phase current Ia1 and the A2 phase current Ia2, and the following equation is obtained from the relationship of the equation (14).
Ia1 = Ia / 2 + If2 (16)
Ia2 = Ia / 2-If2 (17)
The sum of the A1 phase current Ia1 and the A2 phase current Ia2 is not affected by the field power component If2 and becomes the A phase current Ia as shown in the following equation.
Ia1 + Ia2 = Ia / 2 + If2 + Ia / 2−If2 = Ia (18)
Ia1-Ia2 = 2 × If2 (19)
The winding 111 and the winding 11B are both A-phase stator windings, but between the winding 111 arranged between 0 ° and 360 ° in electrical angle and between 360 ° and 720 ° in electrical angle The windings 11B are arranged so as to be capable of energizing each current, and a current component having an electrical angle of 720 ° cycle can be energized. The relationship between the voltage and current components transmitted to the power receiving windings 38 and 3A and the field current If supplied to the field windings 3C, 3D, 3E, and 3F is the same as in FIG. 3 and the description thereof. However, the example which connected the output terminal of full wave rectifier 39,3B in series is shown. Reference numerals 111 and 11B are A-phase windings, and at the same time, feed windings as a means for feeding field power.

  Next, FIG. 12 shows examples of voltage waveforms and current waveforms of the five-phase motor shown in FIG. For the three-phase case, as shown in FIGS. 6, 7, and 9, the utilization factor of each part of the motor is improved by approaching the rectangular wave rather than the sine wave, and it is possible to increase the efficiency in principle. . Similarly, for the five-phase motors of FIGS. 11 and 12, the trapezoidal wave-like voltage and current close to a rectangular wave are used to improve the motor efficiency, reduce the motor size, reduce the motor cost, and reduce the inverter size. Cost reduction is realized.

  In the case of the three-phase motor shown in FIGS. 3 and 9, the DC voltage source 37 can be energized to only one path of the three-phase windings of U, V, and W, and the utilization rate is 2/3. And 1/3 is a structure which cannot be used. On the other hand, in the configurations of FIGS. 11 and 12, the DC voltage source 37 can energize two paths among the five-phase windings A, B, C, D, and E. = 0.8 and the utilization rate can be improved. Since the induced voltage of each winding cannot be a rectangular wave, switching of phase current and variable time are necessary, it is appropriate that there is a redundancy of energization of 1/5 = 0.2.

An example of the A-phase voltage waveform Vqa is shown on the lower side of the sheet of FIG. The voltage amplitude is shown as 1.0, and the horizontal axis is the electrical angle of the rotation angle in the circumferential direction of the motor. It has a trapezoidal wave shape close to a rectangular wave whose voltage changes sharply in the vicinity of 0 ° and 180 °. The angular width at which the voltage changes is θsf. The B-phase voltage Vqa, C-phase voltage Vqc, D-phase voltage Vqd, and E-phase voltage Vqe are voltage waveforms having the same shape, which are delayed in phase by 72 ° in electrical angle, but are difficult to see because the waveforms overlap. Not done. In FIG. 12, Iqa is an A-phase current, Iqb is a B-phase current, Iqc is a C-phase current, Iqd is a D-phase current, and Iqe is an E-phase current. The motor output power Pm is represented by the following equation when idealized with the internal motor loss such as copper loss due to winding resistance being zero.
Pm = Vqa × Iqa + Vqb × Iqb + Vqc × Iqc
+ Vqd × Iqd + Vqe × Iqe (20)

The method of expressing the rotation angle in FIG. 12 is shown by a method similar to the three-phase case in FIG. It is assumed that the trapezoidal current waveform of each phase is symmetrical. Further, it is assumed that the current waveforms of the A, B, C, D, and E phases have the same shape and the phases are different by 72 °.
In the region of the width of θrr, which is the slope of the induced voltage, the induced voltage fluctuation is large, and it is difficult to obtain the torque value as expected when a torque current is applied during this period. The A-phase current Iqa in FIG. 12 has a current waveform that combines a positive trapezoidal wave and a negative trapezoidal wave. While the A-phase current Iqa is changing from a positive value to a negative value, the current value is zero or small, and the width of the section is θhy × 2. This θhy is set to a value slightly larger than ½ of θrr of the A-phase voltage waveform Vqa, and the current is set to a small value close to zero in a section where the induced voltage change is large so that torque fluctuation does not increase. θsf is a rotation angle at which the current changes from the value of the upper side of the trapezoid to zero. θhf is a value that is ½ of θsf, and is a rotation angle at which the current value changes in a trapezoidal shape by ½, as shown in FIG.

  Next, in the voltage characteristics and current characteristics shown in FIG. 12, the conditions under which a large motor output can be output and the motor output power is constant are described. Note that a constant motor output means that the torque ripple is zero. In FIG. 11, the A-phase winding is divided into an A1-phase winding 111 and an A2-phase winding 11B. However, for power considerations, it is assumed that there is one A-phase winding. It is a form connection. If the maximum voltage of each phase is 1.0 [unit voltage] and the maximum current is 1.0 [unit current], the maximum output is 2.0 [unit voltage × 2 for two paths of five windings. Unit current].

This condition is replaced with the angular width of FIG. First, during θrr when the voltage changes sharply, the torque also changes greatly, it is difficult to obtain a torque value that is stably controlled and expected, and the torque value is also small. Assume a value near zero. Therefore, in order to obtain 2.0 [unit voltage × unit current] that is the maximum output during the θrr, the voltage of the other four phases is 1.0 [unit voltage] or −1.0 [unit voltage]. The four-phase current must be 1.0 [unit current]. During θsf, the B-phase current Iqb is 1.0, and the D-phase current Iqd and the E-phase current Iqe are −1.0. Therefore, the sum (Iqa + Iqc) of the A-phase current 1qa and the C-phase current Iqc is 1.0. Find conditions that can be done. It is only necessary that the sum of θhy and θhf be 72 ° / 4 = 18 ° or less.
θrr / 2 + θsf / 2 ≦ 18 °
(Θrr + θsf) ≦ 36 ° (21)

(Theta) rr of (13) is a value resulting from the shape of a motor induced voltage, and there exists a limit on motor design to make this value small. θsf is a variable time of the current and is determined by the controllability of the current. For example, if it rotates at a low speed, there is a time margin, so it can be set to a small value and can be close to zero. Further, the A-phase current waveform Iqa is difficult to be trapezoidal when the rotational speed is increased in actual control, and becomes a current waveform in which the trapezoid changes smoothly. Therefore, there is an ambiguity in the evaluation criteria of equation (21). In particular, the five-phase motor current characteristics in FIG. 12 are more severe in angular conditions than in the case of the three-phase motor in FIG.
From a practical point of view of the motor, the allowable range is determined as 20% of the partial torque fluctuation in the current waveform of FIG. In the vicinity of 162 ° in FIG. 12, since the B-phase current Iqb is 1 and the D-phase current Iqd and the E-phase current Iqe are −1, when the sum of the A-phase current and the C-phase current (Iqa + Iqc) is 0.6, 1 + 0.6) /2=0.8, and 80% torque is output. Therefore, in a five-phase motor, the current waveform range up to 60% of the current peak value is regarded as the upper side of this trapezoidal shape. From a practical viewpoint, this is a standard that partially allows 20% torque fluctuation. Due to the partial variation, the average torque decrease is 10% or less. Further, there is a degree of freedom in taking each value of the current values (Iqa + Iqc) = 1 of these two phases, and the shape shown in FIG. 9 can be modified. A trapezoidal shape with a smaller change rate and a smooth corner shape can also be used.

  Note that the method of supplying the feeding current If2 that is the field power component can be variously modified from the configuration of FIG. If a component having a period of 720 ° in electrical angle is created, not only the windings of the same phase, the power receiving windings 38 and 3A in FIG. 11 can receive the power. For example, the configuration shown in FIG. 5 is also possible, and these various modifications are also included in the present invention.

  As shown in the explanation of FIG. 3, the motor configuration of FIG. 11 uses the stator windings to perform the functions of the brush and slip ring to supply power, so that the space, cost, life, and reliability of the device are increased. Is excellent in terms of. The features described in FIG. 3 have similar features and effects even in the motor configuration of FIG. 11, although the three phases are changed to five phases. In addition, the copper loss of the stator winding can be greatly reduced as compared with a motor of a type in which the field current is superimposed on the torque current. The reason for this is that the motor of FIG. 11 passes the current component that transmits the field power, but does not pass the field current component to the stator winding, so there is no decrease in the power factor as shown in FIG. Since the current amplitude does not increase, the copper loss of the stator can be greatly reduced. Note that the copper loss due to the stator current component for transmitting the field power from the stator to the rotor is small.

  Furthermore, by using a sine wave as shown in FIG. 6 and a trapezoidal wave-like voltage and current close to a rectangular wave as shown in FIG. The utilization factor 4/5 of each part of the motor is 0.8, which is a significant improvement compared to the utilization factor of sine wave voltage and current being 0.5. In the case of even-numbered phases such as four-phase and six-phase, the number of currents flowing in and out from the inverter coincides, so that there is a problem in that the output decreases when the alternating current is reversed and torque pulsation occurs. The 7-phase, 9-phase, etc. are preferable in that respect because one phase can perform the current switching operation as shown in FIG. 12, but there is a problem that the configuration of the motor and the inverter becomes complicated. Although the present invention does not exclude these, there are advantages and disadvantages, and it is necessary to use them properly depending on the application.

  Next, the burden on the inverter shown in FIG. 11 will be described. The feeding power If2, which is the field power component, is differentially generated and supplied by the four IGBTs 116, 117, 118, and 119. It is a relationship of (16), (17) Formula, and If2 is relatively small. The four IGBTs 116, 117, 118, and 119 also generate A-phase voltage and current, and the A-phase current Ia is expressed by the relationships of the equations (14), (16), and (17). Accordingly, the current capacities of 116, 117, 118, and 119 of the four IGBTs may be about ½ of the current capacities of 11H and 11J of the B-phase driving IGBT, and the number of the A-phase driving IGBTs is four. It will increase, but the cost and space burden will not be doubled. In particular, in an inverter having a large output capacity to some extent, the IGBTs are often connected in parallel, and in such a case, the inverter configuration of FIG. 11 can be configured only by changing the connection relationship of the transistors. In that case, the cost burden is small. From the standpoint of the utilization rate of the IGBT of the inverter, it can be said that the utilization rate is 0.8 because it can be used at all times in order to maximize the output of the 4 phases of the 5 phases. Since the maximum voltage and maximum current of the IGBT can be reduced by the increase in the utilization rate, the inverter can be reduced in size and cost.

  Next, 3 phase and 5 phase are compared. In the simple comparison, the utilization factor of each part of the three-phase motor explained in FIG. 3 etc. is 2/3, and the utilization factor of each part of the five-phase motor explained in FIG. 11 etc. is 4/5. The rate is high, and in principle, high efficiency and miniaturization are possible. As for the inverter, since the maximum voltage and maximum current are relatively clear, the five-phase motor is more advantageous in principle. Furthermore, when the field current component is energized to some extent with each current of the stator, the angular position resolution of each winding becomes important, and in that respect, the 5-phase motor has an angular resolution of 36 °, which is advantageous. Further, when several current modes are changed due to various control requirements, the five phases are characterized by a mathematically high canceling effect of low-order harmonics and extremely small torque ripple. Further, as will be described later, it can be driven by a five-phase sine wave current shown in FIG. Isa is an A-phase current, Isb is a B-phase current, Isc is a C-phase current, Isd is a D-phase current, and Ise is an E-phase current. The five-phase motor has an excellent electrical function, but has a problem that the number of windings is increased and complicated. As for the inverters, it is possible to reduce the total current capacity obtained by adding the energization capacities of the inverters, but there is a problem that the number of control elements increases and becomes complicated. To solve these problems, FIGS. 15 to 17 show the simplification of the coil end of the five-phase motor, FIGS. 23 and 24 show the simplification of the winding of the five-phase motor, and FIGS. For simplification, FIG. 41 illustrates one method of effective use of the five-phase inverter.

  Next, FIG. 14 will be described. The purpose is to simplify the winding of the coil end portion of the five-phase motor, shorten the length of the coil end portion in the rotor axial direction, and improve the productivity of the stator winding. Since the 5-phase motor has a large number of windings, the crossing of the windings at the coil end portion becomes complicated, and thus it is rarely used. In particular, in the case of a motor with an output of 5 KW or more, a practical 8-pole configuration, and full-pitch winding with a large winding coefficient, the winding arrangement of the coil end portion is as shown in FIG. There are many production problems. In order to take advantage of the above-mentioned features of the 5-phase motor, it is necessary to improve this problem at the same time.

  The number of slots in FIG. 14 is 40, the number of windings in the coil end portion is 20, and the rotor has 8 poles. The rotor can be various types of rotors. For example, the rotor shown in FIG. 19 can be arranged. A phase winding is 287, 28C, 28H, and 28N. B-phase windings are 288, 28D, 28J, and 28P. The C-phase windings are 289, 28E, 28K, and 28Q. The D-phase windings are 28A, 28F, 28L, and 28R. The E-phase windings are 28B, 28G, 28M, and 28S. Each coil end winding crosses the other four windings. For example, the U-phase winding 287 intersects four windings 28R, 28S, 288, and 289. There is a problem that the winding of each winding becomes complicated. In particular, it is difficult to cross the windings of two adjacent slots. Specifically, the intersection of the winding of the slot 141 and the winding of the slot 142, the intersection of the winding of the slot 141 and the winding of the slot 142, the intersection of the winding of the slot 143 and the winding of the slot 144, For example, the intersection of the winding of the slot 145 and the winding of the slot 146, the intersection of the winding of the slot 147 and the winding of the slot 148, and the like.

  FIG. 15 and FIG. 16 show a method for reducing the problem of winding crossings at the coil end portion. The shape of two adjacent slots at the coil end is modified to create a slot in which the center of the winding is moved to the outer diameter side and a slot in which the center of the winding is moved to the inner diameter side. By defining the winding order of the windings, the crossing of the windings can be improved more easily. The motor shown in FIG. 15 is the same as that shown in FIG. 14, has 40 slots, and the rotor has 8 poles. The rotor can be various types of rotors. For example, the rotor shown in FIG. 19 can be arranged. In FIG. 15, the A phase winding is 297, 29C, 29H, 29N, the B phase winding is 298, 29D, 29J, 29P, the C phase winding is 299, 29E, 29K, 29Q, and the D phase winding is 29A, 29F. 29L, 29R, and E-phase windings are 29B, 29G, 29M, and 29S.

The shape KSW of the slot 15A in FIG. 15 differs from the shape of the slot SP1 in FIG. 14 in that the outer diameter side of the slot is wide and the inner diameter side near the slot opening is narrow. On the other hand, the shape KSN of the adjacent slot 15B is wider on the inner diameter side of the slot and narrower on the inner diameter side of the slot. The purpose of these shapes KSW and KSN is related to the winding order of the windings, and the slot shape of the winding wound first with two windings arranged adjacent to each other in the circumferential direction is wide on the outer diameter side. KSW is used, and the winding that goes from the slot to the coil end side is guided to the outer diameter side as much as possible and wound.
Then, the slot shape of the winding wound after the two windings is a wide shape KSN on the inner diameter side, and the winding extending from the slot to the coil end side is wound so as to overlap the previously wound winding. The winding is performed so that the number of crossings of the two windings can be minimized. By this method, the winding intersection portion can be wound with less protrusion in the axial direction of the rotor, and the winding length of the coil end portion can be shortened.

Next, an example of a procedure for winding each winding around the stator core of FIG. 15 will be described. Initially, starting from a state where no winding is wound around the stator core, the A-phase winding 297 is wound around the slots 151 and 156. Next, the B phase winding 298 is wound around the slots 153 and 158. Next, the C-phase winding 299 is wound around the slots 155 and 15A. At this time, the winding 299 coming out of the slot 155 is wound around the A-phase winding 296 wound around the slot 156 and wound around the slot 15A. A double wire is added to the portion wound around the A-phase winding 296 to indicate the overlapping of the windings. Similarly, the windings of the other phases are shown with double lines added to the overlapping portions of the windings. Next, the D-phase winding 29A is wound around the slots 157 and 15C. Next, the E-phase winding 29B is wound around the slots 159 and 15E. Next, the A-phase winding 29C is wound around the slots 15B and 15G. Next, the B-phase winding 29D is wound around the slots 15D and 15J. Next, the C-phase winding 29E is wound around the slots 15F and 15L. Next, the D-phase winding 29F is wound around the slots 15H and 15N. Next, the E-phase winding 29G is wound around the slots 15K and 15Q. Next, the A-phase winding 29H is wound around the slots 15M and 15S. Next, the B phase winding 29J is wound around the slots 15P and 15U. Next, the C-phase winding 29K is wound around the slots 15R and 15W. Next, the D-phase winding 29L is wound around the slots 15T and 15Y.
Next, the E-phase winding 29M is wound around the slots 15V and 291. Next, the A-phase winding 29N is wound around the slots 15X and 293. Next, the B-phase winding 29P is wound around the slots 15Z and 295.
Next, the C-phase winding 29Q is wound around the slots 292 and 29T. Next, the D-phase winding 29R is wound around the slots 294 and 152. Next, the E-phase winding 29S is wound around the slots 296 and 154.

  Similar to the slot 15A, the slot with the wide outer diameter side KSW is 151, 153, 156, 158, 15C, 15E, 15G, 15J, 15L, 15N, 15Q, 15S, 15U, 15W, 15Y, 291, 293, 295. Similar to the slot 15B, the slots with the wide inner diameter side KSN are 152, 154, 157, 159, 15D, 15F, 15H, 15K, 15M, 15P, 15R, 15T, 15V, 15X, 15Z, 292, 294, 296. It is.

  Next, an example of the slot shape will be described with reference to FIG. FIG. 16 (a) is a cross-sectional shape of the orderly tooth of FIG. 14, and FIG. 16 (a) is a cross-sectional shape of a slot surrounded by the tooth of FIG. 14, that is, the tooth of FIG. 16 (a). is there. FIG. 16B is a shape in which the tooth shown in FIG. 16A is inclined with respect to the center line, and the position and shape of the tooth on the inner diameter side are not changed. The shape of the slot can be changed by the shape of FIG. At this time, since the slot shape is widened and narrowed, and there is a canceling effect, it can be considered that the cross-sectional area of the slot hardly changes. In principle, the slot cross-sectional area slightly decreases under the condition of a constant tooth width. In addition, since the tip shape of the teeth facing the rotor is not changed, the electromagnetic influence on the rotor is small. FIG. 16C shows a shape in which the teeth are inclined in the opposite direction to FIG.

By properly using the tooth shapes shown in FIGS. 16A, 16B, and 16C, various shapes as shown in FIGS. 16D, 16E, 16F, and 16G are used. A slot shape can be obtained.
The slot shape of FIG. 16 (e) is a slot shape in which the teeth of (b) are arranged on the left side and the teeth of (c) are arranged on the right side, and the outer diameter side is wide and the inner diameter side is narrow. The shape is a slot shape 15A or the like. The slot shape of (f) in FIG. 16 is a slot shape in which the teeth of (c) are arranged on the left side and the teeth of (b) are arranged on the right side, and the outer diameter side is narrow and the inner diameter side is wide. A shape such as a slot shape 15B. The slot shape of (g) in FIG. 16 is a slot shape in which the teeth of (c) are arranged on the left side and the teeth of (c) are arranged on the right side, and the slot shapes of slot shapes 155 and 29T in FIG. is there. In the vicinity of the start and end of winding on the stator core, a slightly special slot shape is formed due to the asymmetry.

  15 and 16, the complicated winding configuration of a 5-phase motor with a large number of poles, such as 8 poles, is improved by improving the three-dimensional intersection shape of two windings arranged next to each other. By reducing the number of wire crossings, it is possible to make winding easier, to reduce winding of the winding crossing in the rotor axial direction, and to reduce the winding length of the coil end. Can be shortened. The slot shape is not limited to the shape shown in FIG. 16, and various modifications can be made for the same purpose. Further, the winding order of the windings is not unambiguous, and various selections are possible, and these are also included in the present invention.

  Next, FIG. 17 is a diagram in which winding guides 171, 172, 173, 174, 175, 176 and the like that limit the arrangement positions of the windings are added to the configuration of FIG. 15. Although the example of winding of the coil end portion has been described with reference to FIG. 15, the position of the winding has a three-dimensional degree of freedom, and it is difficult to arrange the windings in an orderly manner at an expected position. If the winding position can be specified by any method, it is possible to wind the winding in a more orderly and simplified manner.

The winding guides 171, 172, 173, 174, 175, and 176 in FIG. 17 are rod-shaped winding guides that limit the arrangement position of the windings to some extent when winding the windings. For example, when the A-phase winding 297 that is wound first in FIG. 17 is wound, the winding guides 171, 172, 173 are wound so that the 297 does not protrude toward the slots 152, 153, 154, 155. The position can be limited. The winding position of the B-phase winding 298 to be wound next can be limited so that the winding guides 172, 173 and 174 do not protrude toward the slots 153, 154, 155 and 156. Since the A-phase winding 297 of the slot 156 has already been wound, the C-phase winding 299 to be wound next is wound around the slot 156 and its winding. The winding guides 174 and 175 can limit the winding positions so that the C-phase winding 299 does not protrude toward the slots 157, 158 and 159. Since the B-phase winding 298 of the slot 158 is already wound, the D-phase winding 29A to be wound next is wound around the slot 158 and its winding.
The winding guides 175 and 176 can limit the winding position so that the D-phase winding 29A does not protrude toward the slots 159, 15A, and 15B. Similarly to the C-phase winding 299 and the D-phase winding 29A, the E-phase winding 29B to be wound next is already wound with the C-phase winding 299 in the slot 15A. Wind over the winding. Similarly, the windings shown in FIG. 17 can be sequentially wound in the counterclockwise direction on the paper surface. When winding each winding around the stator core, the winding start and end winding conditions are different from most other windings, so the winding relationship is slightly different. It becomes.

  As shown in FIG. 17, by providing the winding guides 171, 172, 173, 174, 175, and the like, it is possible to wind each winding in a more orderly and simplified manner. In the example of FIG. 17, an example of a simple rod-shaped winding guide has been described. However, the number of winding guides is increased, and a winding guide having a three-dimensional shape in which the shape of the winding guide is refined with resin or the like is used. It is also possible to integrate the winding guides. As a result, it is possible to make the winding easier, to reduce winding of the winding intersection in the rotor axial direction, and to reduce the winding length of the coil end portion. .

  Furthermore, by using a winding guide that takes into consideration the winding order of the windings and the overlapping relationship between the windings, the winding can be directly wound by an automatic winding machine. Although the winding arrangement is a little complicated, the winding position is controlled while precisely controlling the winding position of the winding like an automatic winding machine of a concentrated winding motor, and a certain amount of winding tension is applied. can do. The winding guide can also be used as a terminal block for the winding. When a plurality of winding guides are integrated or integrated with resin or the like, a winding terminal block can be provided in the vicinity of the winding guide.

  12 and the description thereof, it has been explained that the utilization factor of each part of the motor can be improved by driving the motor with a trapezoidal wave-like voltage and current close to the rectangular wave of the five-phase motor. However, the five-phase current waveform in FIG. 12 includes a steep increase / decrease and there is no problem because the frequency is low at low speed rotation. However, it is not necessary to accurately energize the current of these steep changes at high speed rotation. Due to the inductance, it is difficult due to the limitation of the DC voltage source of the inverter. As a method for solving the problem at the time of high-speed rotation, a 5-phase sine wave A-phase current Isa, B-phase current Isb, C-phase current Issc, D-phase current Isd, and E-phase current Ise shown in FIG. To do. Since the rate of change of current can be greatly reduced, the voltage burden of current control is greatly reduced. As a motor output characteristic, a 5-phase motor is particularly effective in canceling low-order harmonics, so even if a 5-phase sine wave current is applied to a 5-phase trapezoidal wave motor, torque ripple is required in most applications. However, it can be driven with such a small value that does not cause a practical problem.

  In particular, as a characteristic of main motors such as electric vehicles, the number of rotations that require the maximum torque of the motor is a climbing operation on a steep slope, and therefore it is required that the maximum torque can be generated effectively by low-speed rotation of the motor. It is said that the maximum torque characteristics determine the size of the motor and the cost of the motor. There are many needs for large torque characteristics during low-speed rotation not only for automobiles but also for industrial machines.

  The current characteristics of the three-phase motor shown in FIG. 9 can also be made sinusoidal, and the same can be said. In addition, it is difficult to create a rectangular magnetic flux distribution and a rectangular induced voltage in the circumferential direction with a three-phase sinusoidal field current. It is difficult to create a rectangular magnetic flux distribution and a rectangular induced voltage in the circumferential direction with a five-phase sinusoidal field current. As shown in the present invention, a rectangular magnetic flux distribution can be created in the circumferential direction by the field current of the field winding wound around the rotor. If such a field magnetic flux can be varied by a field current, field magnetic flux control, constant output control, and the like are possible. As another example, in a rotor in which permanent magnets are arranged on the rotor surface, it is relatively easy to create a rectangular magnetic flux distribution in the circumferential direction, but it is difficult to achieve compatibility with field magnetic flux control.

  Next, in order to obtain the field magnetic flux distribution of the five-phase motor as shown in FIGS. 10 and 12 and the induced voltage characteristics close to the rectangular shape of the winding, a field current is applied to the field winding of the rotor of FIG. Can be energized. However, if the field excitation current component is energized to each phase winding of the stator and the increase and decrease of the field flux that forms a rectangular wave distribution in the circumferential direction can be controlled by the current of the stator winding, the response of the field flux Can be improved, and field weakening is also possible. As in the conventional concept of dq axis control, it is difficult to obtain an induced voltage characteristic close to a rectangle with a magnetomotive force having a sinusoidal distribution obtained as a vector sum of a plurality of phase currents in the stator winding.

  FIG. 21 shows each phase current of the stator that excites the field magnetic flux having a rectangular wave distribution in the circumferential direction. The horizontal axis indicates the rotation direction angle in electrical angle. Ida, Idb, Idc, Idd, and Ide are the A, B, C, D, and E phase field excitation current components. Between the width θm of 36 ° from 162 ° to 198 °, the A-phase field excitation current component Ida is set to 0.4 which is a positive peak value. The positive and negative vectors from the A phase to the E phase have a relationship as shown in FIG. The current Ida flowing into the A-phase winding flows out as the closest C-phase current Idc and d-phase current Idd in the reverse vector. Therefore, (Idc + Idd) = − 0.4. Also, the current (Ia1 + Ia2) in FIG. 11 flows into the U1-phase winding 111 and U2-phase winding 11B, and flows out through the C-phase windings 113 and 11D and the D-phase windings 11E and 114. To do. The A-phase field current component Ida in FIG. 21 is a current centered on 180 °, whereas the torque current component Iqa shown in FIG. The electrical angle is 90 ° in phase lag.

In the range of the width θn from 126 ° to 162 ° in FIG. 21, the C-phase field excitation current component Idc is set to −0.4, which is a negative peak value. In the meantime, similarly, (Ide + Ida) = 0.4. Between the width θp of 198 ° to 230 ° in FIG. 21, the D-phase field excitation current component Idd is set to −0.4 which is a negative peak value. During this time, the A-phase current Ida and the B-phase current Idb are similarly expressed by the following equations.
(Ida + Idb) = 0.4 (22)
Further, during this period, the A-phase current Ida and the B-phase current Idb change linearly, but it is sufficient if the expression (22) is satisfied and various values may be taken. For example, Ida and Idb may be constant values of 0.2 during this period. Further, equation (22) is a typical example, and as shown in equation (15), it is slightly modified in accordance with the spirit of the present invention so that the sum of all currents of the five-phase star connection becomes zero. You can also. In other angle ranges, the field current components have the same relationship.

  By supplying a five-phase field current component as shown in FIG. 21, each magnetic pole of the rotor as shown in FIG. 19 can be excited to increase or decrease the field magnetic flux. In addition, since a uniform magnetomotive force can be applied to almost the entire area of the rotor magnetic pole by these field current components, a uniform field magnetic flux can be distributed at each position of the rotor magnetic pole. As a result, a field magnetic flux having a rectangular wave distribution in the circumferential direction can be created, and an induced voltage characteristic close to the rectangular shape of the winding can be obtained. Since these five-phase field current components are driven by an inverter, they can be increased or decreased rapidly. Note that the effect of generating the field magnetic flux by the permanent magnet of FIG. 19 is constant and cannot be varied. Further, the generation of the field magnetic flux by the field current supplied to the field windings 87, 8A, 8C, 8D, 8F, 8K, 8H, 89 can increase the magnetic flux relatively quickly. However, when the field magnetic flux is reduced, the magnetic energy is held by the diode and the field winding. Therefore, there is a problem that the response speed is slow because there is no choice but to wait for the magnetic energy to be consumed by the resistance of the field winding. is there.

  Next, an embodiment of claim 7 will be described. FIG. 23 shows a cross-sectional view of the five-phase motor of the present invention. In the 8-pole motor, the arrangement relationship of the stator windings at the coil end portion is added. Windings are not disposed in the slot-shaped locations indicated by broken lines, but are removed. The five-phase, eight-pole motor shown in FIG. 14 has a problem in that the number of windings is large and the windings of the respective phases are complicated to cross each other, which makes the winding complicated. . In the configuration shown in FIG. 23, in order to alleviate this problem, half of the windings are regularly removed in a specific relationship, and the crossing relationship of the windings is greatly simplified. The motor of the present invention shown in FIG. 23 is a motor in which 10 full-pitch windings are arranged in 5 phases, 8 poles and 20 slots.

  The positional relationship between the windings 311, 312, 313, 314, 315, 316, 317, 318, 319, and 31A illustrated in FIG. 23 is the same as the positional relationship between the windings in FIG. An example of a procedure for winding the winding will be described. First, reference numeral 311 denotes an A-phase winding, which is wound from the slots 231 to 236. Reference numeral 312 denotes an E-phase winding, which is wound from the slot 239 to 23E. Reference numeral 313 denotes a D-phase winding wound around the slots 23H to 23N. 314 is a C-phase winding, which is wound from the slot 23R to 23W. Reference numeral 315 denotes a B-phase winding, which is wound from the slot 23Z to 23H. Since these windings 311, 312, 313, 314, 315 do not cross and interfere with other windings, they can be easily wound.

  Next, reference numeral 316 denotes a C-phase winding, which is wound from the slot 235 to 23A. At this time, windings are already wound around the slots 236 and 239. However, since other windings are not wound around the slots 236 and 239 later, they are wound so as to cover these windings. You can also. Similarly, the windings 317, 318, 319, 319, 31A are wound. These ten windings have few windings and can be easily wound as compared with the windings of the five-phase full-pitch winding stator of FIG. Further, even when compared with the winding of the three-phase full-pitch winding stator shown in FIG. 1, the winding crossing relationship is simple, so that winding can be easily performed.

  Next, the correction of the slot shape shown in FIG. 23 will be described. Slot shapes 233, 234, 237, 238, 23B, 23C, 23F, 23G, 23K, 23L, 23P, 23Q, 23T, 23U, 23X, 23Y, 31E, 31F, 31J, 31K indicated by broken lines are not necessary, and soft iron It can also be a part. However, as a matter of course, since it is desired to make the space for winding the winding as wide as possible, the cross-sectional area of the slot for winding the winding can be widened by the cross-sectional integral of the slot shape indicated by the broken line. Specifically, the slot cross-sectional area shown in 23A can be expanded to the location shown by the thick line 31B. Similarly, the slot cross-sectional area shown in 23D can be expanded to the location shown by the thick line 31C. This means that the cross-sectional area was nearly doubled. The other slots can be similarly enlarged. By increasing the cross-sectional area of each slot, the slot can be arranged almost uniformly on the circumference.

  As a result, windings equivalent to the winding amount in all the slots of the 5-phase, 8-pole, 40-slot all-pitch motor shown in FIG. 14 can be wound in each slot of FIG. Furthermore, since the winding configuration is simple compared to the winding of FIG. 14 and winding of the winding is easy, the winding space factor in the slot of FIG. 23 can be improved, and the total amount of insulator in the slot Can also be expected to decrease. Note that the shape of the teeth between the slots 23A and 23D needs to be a shape and a width that allow the magnetic flux to pass between the rotor and the stator without difficulty.

The winding configuration in FIG. 23 is a configuration in which two sets of motor configurations in which all-node windings are arranged in five phases, four poles, and ten slots with an electrical angle of 720 ° are arranged on the circumference, and include five phases, eight This is a motor in which all joint windings are arranged with poles and 20 slots. Since the electric angle of 720 ° is arranged symmetrically with respect to the center of the motor, the A-phase windings 311 and 318 are replaced with the A-phase windings 111 and 11B in FIG. To the rotor side. At this time, the winding pitch of the power receiving windings 31L and 31M of the rotor needs to be 720 ° in electrical angle. Similarly, the winding pitch of the power receiving windings 31N and 31P needs to be 720 ° in electrical angle. The field windings of the rotor are wound at a 180 ° pitch for each rotor magnetic pole and are the same as the other field windings shown in the figure. Further, the internal configuration of the rotor can be an 8-pole rotor in which the gap portion 201 and the magnets 201, 202, 203, and 204 as shown in FIGS. 19 and 20 are arranged as an 8-pole configuration. Can be improved.

Further, since the circumferential position of the opening of each slot in FIG. 23 determines the current vector of each winding, it is non-uniformly arranged every two in the circumferential direction, but the electrical angle is 0 °. , 72 °, 144 °, 216 °, 288 ° are not changed. Further, in FIG. 23, slot openings are provided at electrical positions in the circumferential direction at circumferential positions of 0 °, 36 °, 144 °, 180 °, 288 °, 324 °, 72 °, 108 °, 216 °, and 252 °. Are arranged, and two sets of them are arranged. In addition, by creating a small dent similar to the slot opening at or near the position of the slot opening indicated by the broken line, and providing a pseudo slot opening, the magnetic resistance pulsation caused by the slot opening is reduced. It is possible to reduce the torque ripple.

  The configuration of the five-phase motor shown in FIG. 23 is a configuration with an electrical angle of 720 °, and can also be configured as a motor with five phases, four poles, ten slots, and five full-pitch windings. In addition, five windings can be added to the configuration of the five-phase motor shown in FIG. 23, and the motor can be configured with five phase windings, eight poles, thirty slots and fifteen total windings. Add B-phase windings to 233 and 238, A-phase windings to 23B and 23G, E-phase windings to 23K and 23Q, D-phase windings to 23T and 23Y, and C-phase windings to 31E and 31K. . The torque balance as a 5-phase motor can be taken. Further, the method for facilitating the intersection of windings shown in FIGS. 15 and 16 can be applied. Further, winding of the winding can be facilitated by using a winding guide as shown in FIG.

  Next, FIG. 24 shows a cross-sectional view of another five-phase motor of the present invention. In a 12-pole motor, the arrangement relationship of the stator windings at the coil end portion is additionally shown. Windings are not disposed in the slot-shaped locations indicated by broken lines, but are removed. The five-phase, eight-pole motor shown in FIG. 14 has a problem in that the number of windings is large and the windings of the respective phases are complicated to cross each other, which makes the winding complicated. . In the configuration shown in FIG. 24, in order to alleviate this problem, 1/3 windings are regularly removed in a specific relationship, and the crossing relationship of the windings is greatly simplified. The motor of the present invention shown in FIG. 24 is a motor in which 20 full-pitch windings are arranged in 5 phases, 12 poles, 40 slots.

  An example of a procedure for winding the winding of FIG. 24 will be described. First, reference numeral 321 denotes an A-phase winding, which is wound from the slots 241 to 246. The slots of the A phase winding are 241, 246, 24B, 24C, 24D, 24E, 24F, 24G, 24H, 24J, and 24K. Next, similarly, D-phase winding 322, B-phase winding 323, E-phase winding 324, C-phase winding 325, A-phase winding 326, D-phase winding 327, B-phase winding 328 E-phase winding 329 And winding with the C-phase winding 32A. Since these windings do not cross or interfere with other windings, they can be easily wound. Next, in the windings whose winding order is reversed, -E phase winding 32B, -C phase winding 32C, -A phase winding 32D, -D phase winding 32E, -B phase winding 32F, -E phase winding 32G, -C phase winding 32H, -A phase winding 32J, -D phase winding 32K, -B phase winding 32L.

  These 20 windings have few intersecting windings and can be easily wound as compared with the windings of the five-phase full-pitch winding stator of FIG. Further, even when compared with the winding of the three-phase full-pitch winding stator shown in FIG. 1, the winding crossing relationship is simple, so that winding can be easily performed.

Next, the modification of the slot shape shown in FIG. 24 will be described. The slot shapes 242, 245, 248, 24B and the like indicated by broken lines are not necessary, and may be a soft iron part. However, as a matter of course, since it is desired to make the space for winding the winding as wide as possible, the cross-sectional area of the slot for winding the winding can be widened by the cross-sectional integral of the slot shape indicated by the broken line. Specifically, the slot cross-sectional area indicated by 24P can be expanded to the location indicated by the thick line 24Q. Similarly, the slot cross-sectional area indicated by 24S can be expanded to the location indicated by the thick line 24R. This means that the cross-sectional area can be enlarged to about 3/2 times. The other slots can be similarly enlarged. By increasing the cross-sectional area of each slot, the slot can be arranged almost uniformly on the circumference.

  As a result, compared with the amount of winding in the slot of the five-phase motor of the basic structure in which the winding is wound in all slots including the slot cross-sectional shape shown by the broken line, the equivalent winding is placed in each slot in FIG. It can be wound. Furthermore, since the winding configuration is simple compared to the winding of FIG. 14 and the winding of the winding is easy, the winding space factor in the slot of FIG. 24 can be improved, and the total amount of insulator in the slot Can also be expected to decrease. Note that the shape of the teeth between the slots 24P and 24S needs to be a shape and a width that allow a magnetic flux to pass between the rotor and the stator without difficulty.

  The winding configuration of FIG. 24 is a configuration in which two sets of motor configurations in which 10 full-pitch windings are arranged in 5 phases, 6 poles, 20 slots with an electrical angle of 360 ° × 3 = 1080 ° are arranged on the circumference. In this motor, 20 full-pitch windings are arranged in 5 phases, 12 poles, 40 slots. Since the configuration with an electrical angle of 1080 ° is symmetrical with respect to the center of the motor, the A-phase windings 321 and 326 are replaced with the A-phase windings 111 and 11B in FIG. To the rotor side. At this time, the winding pitch of the power receiving windings 24T and 24U of the rotor needs to be 1080 ° in electrical angle. The same winding pitch for the power receiving windings 24V and 24W is required to be 1080 ° in electrical angle. The field windings of the rotor are wound at a 180 ° pitch for each rotor magnetic pole and are the same as the other field windings shown in the figure. Further, the internal configuration of the rotor may be a 12-pole rotor in which gaps 201, 202, 203, 204 and magnets 191, 192, 193, 194 as shown in FIGS. 19 and 20 are arranged in an 8-pole configuration. This can improve the performance of the motor.

  In addition, since the circumferential position of the opening of each slot in FIG. 24 determines the current vector of each winding, it is unevenly arranged every two in the circumferential direction, but the electrical angle is 0 °. , 72 °, 144 °, 216 °, 288 ° are not changed. In addition, by creating a small dent similar to the slot opening at or near the position of the slot opening indicated by the broken line, and providing a pseudo slot opening, the magnetic resistance pulsation caused by the slot opening is reduced. It is possible to reduce the torque ripple.

  The configuration of the five-phase motor shown in FIG. 24 is a configuration with an electrical angle of 1080 °, and can also be configured as a motor with five phases, six poles, 20 slots, and ten full-pitch windings. In addition, the configuration of the 5-phase motor shown in FIG. 24 may be configured as a motor in which 15 windings are arranged in 5 phases, 12 poles, 30 slots, in which 5 windings are deleted. it can. For example, the windings 24B, 32D, 32F, 32G, and 32K are omitted. The torque balance as a 5-phase motor can be taken. Further, the method for facilitating the intersection of windings shown in FIGS. 15 and 16 can be applied. Further, winding of the winding can be facilitated by using a winding guide as shown in FIG.

  Next, in order to more clearly show the winding arrangement and interconnections of the motors of FIGS. 1, 3 and 4 shown in the first embodiment, FIG. 28 shows a winding arrangement diagram. FIG. 28 is a linearly developed view of the inner peripheral surface of the stator as viewed from the rotor side. In the horizontal direction of FIG. 28, the rotation angle in the circumferential direction is indicated by an electrical angle, and since it is a 4-pole motor, the entire circumference is 720 °. The thick line shows each tooth of the stator, and 12 teeth are arranged on the entire circumference. Between these teeth is a slot for inserting a winding, and slot numbers 1 to 12 are added. Since it is complicated and difficult to see when all the windings of the stator are described, two sets of the same stator teeth are shown in the upper and lower two stages of FIG. 28, and the U1 phase winding and U2 phase winding are shown in the upper stage. In the lower stage, the V-phase winding and the W-phase winding are shown. Actually, the upper winding and the lower winding overlap each other. In addition, windings indicated by substantially hexagonal shapes 686, 687, 688, 689, 68A, and 68B are so-called turtle shell type coils. This tortoiseshell type coil represents a substantially hexagonal shape with a single wire. For example, the tortoiseshell type coil is a coil in which a plurality of windings such as 50 turns are wound, and shows the entrance and exit of the coil.

  681 in FIG. 28 is a U1-phase winding terminal, and the turtle shell-shaped coil 686 is the windings 11 and 14 in FIG. The other end of the tortoiseshell type coil 686 is connected to 685 which is a neutral point of the star connection of the stator. Similarly, 682 is a U2-phase winding terminal, and the tortoiseshell type coil 687 is the windings 17 and 1A of FIG. The other end of the tortoiseshell type coil 687 is connected to 685 which is a neutral point of the star connection of the stator. The U1-phase current Iu1 is supplied to the winding terminal 681, the U2-phase current Iu2 is supplied to the winding terminal 682, and the relationship of the above equation (6) is established.

  683 in FIG. 28 is a V-phase winding terminal, the tortoiseshell type coil 688 is the windings 13 and 16 in FIG. 1, and the tortoiseshell type coil 68A is the windings 19 and 1C in FIG. The other end of the tortoiseshell type coil 68A is connected to 685 which is a neutral point of the star connection of the stator. 684 in FIG. 28 is a W-phase winding terminal, the tortoiseshell type coil 689 is the windings 15 and 18 in FIG. 1, and the tortoiseshell type coil 68B is the windings 1B and 12 in FIG. The other end of the tortoiseshell type coil 68B is connected to 685 which is a neutral point of the star connection of the stator. 68E indicated by a broken line represents the leftmost tooth, 68F indicated by the broken line represents the rightmost tooth, and the tortoiseshell type coil 68C indicated by the broken line is the same as the tortoiseshell type coil 68B. .

  Next, another embodiment of claim 1 will be described. 29 shows a motor having a configuration in which the number of coils inserted in one slot is two, three phases, four poles, all joint windings, concentrated windings. It can be said to be a modification of the motor of FIG. 1 which is the embodiment of claim 1 described above, and is equivalent in macro. The stator core is the same as the configuration of FIG. 1, and the configuration and arrangement of the windings are different. When only the U-phase current Iu, V-phase current Iv, and W-phase current Iw are applied to both motors, the current flowing in each slot of the motor in FIG. 1 and the total current flowing in the corresponding slot in the motor in FIG. 29 are exactly the same. Value. 29 has the same configuration as that of FIG.

690 is a stator core, and the U1-phase winding is a winding wound from 69D to 69G and a winding wound from 69N to 69M. The U2-phase winding is a winding wound from 69J to 69H and a winding wound from 69K to 69L. The V-phase winding is a winding wound from 69F to 69X, a winding wound from 69Z to 69Y, a winding wound from 501 to 502, and a winding wound from 69E to 503. The W-phase winding is a winding wound from 69R to 69S, a winding wound from 69U to 69T, a winding wound from 69V to 69W, and a winding wound from 69Q to 69P.
Compared with the number of windings in FIG. 1, the number of windings in FIG. 29 is doubled, but the total number of windings is the same.

  FIG. 30 is a diagram illustrating the arrangement and connection of the windings in FIG. Note that the layout diagram in FIG. 30 uses the same expression method as in FIG. 28, such as a tortoiseshell-shaped coil, teeth, slots, and a description of the winding layout divided into two upper and lower stages. 28, 30, 33, 35, 36, etc., the layout and connection diagrams of the windings show the slot numbers and the positions of the windings of the tortoiseshell type coils, and between the tortoiseshell type coils. Connection relationships are also shown. 701 in FIG. 30 is a U1-phase winding terminal, the tortoiseshell type coil 706 is the winding 69D and 69G in FIG. 29, and the tortoiseshell type coil 709 is the winding 69N and 69M in FIG. The other end of the tortoiseshell type coil 709 is connected to the neutral point 705 of the star connection of the stator. Similarly, reference numeral 702 denotes a U2-phase winding terminal, the tortoiseshell type coil 707 has windings 69J and 69H, and the tortoiseshell type coil 708 has windings 69K and 69L. The other end of the tortoiseshell type coil 708 is connected to the neutral point 705. The U1-phase current Iu1 is supplied to the winding terminal 701, the U2-phase current Iu2 is supplied to the winding terminal 702, and the relationship of the above equation (6) is established.

  Reference numeral 703 denotes a V-phase winding terminal, the tortoiseshell type coil 70A has windings 69F and 69X, the tortoiseshell type coil 70B has windings 69Z and 69Y, the tortoiseshell type coil 70C has windings 501 and 502, Tortoiseshell type coil 70D has windings 69E and 503. The other end of the turtle shell-shaped coil 70 </ b> D is connected to the neutral point 705. The V-phase current Iv is supplied to the winding terminal 703 and acts as one phase of the three-phase current, and the relationship of the above formula (6) is established.

  Reference numeral 704 denotes a W-phase winding terminal, the tortoiseshell type coil 70E has windings 69R and 69S, the tortoiseshell type coil 70F has windings 69U and 69T, and the tortoiseshell type coil 70G has windings 69V and 69W. The tortoiseshell type coil 70G is connected to the mechanical type coil 70H, and the tortoiseshell type coil 70H has windings 69Q and 69P. The other end of the tortoiseshell type coil 70H is connected to the neutral point 705. A W-phase current Iw is supplied to the winding terminal 704 to act as one phase of the three-phase current, and the relationship of the expression (6) is established.

Next, the electromagnetic action of the motor shown in FIGS. 29 and 30 will be described. Note that the slot into which the winding 69D of FIG. 29 is inserted is referred to as slot 1 and is named slot 1 to slot 12 in the counterclockwise order, and the slot numbers are added near the corresponding slot.
A specific example of energization of the U1 phase and the U2 phase will be described assuming that the number of turns of each turtle shell coil is one turn for the sake of simplicity. Now, the U-phase current component is supplied to the U1-phase terminal 701 and the U2-phase terminal 702 of FIG. However, since the purpose here is to consider the U-phase current component, the V-phase current and the W-phase current are ignored. The tortoiseshell type coil 706 of FIG. 30 is inserted into the slots 1 and 4, and the tortoiseshell type coil 709 is inserted into the slots 1 and 10, and a U1 phase current Iu1 = 5 A is applied.
The U2-phase current is Iu2 = 5 A, the tortoiseshell type coil 707 is inserted in the slot 4 and the slot 7, and the tortoiseshell type coil 708 is inserted in the slot 7 and the slot 10. 30A is supplied from the upper side to the lower side of the paper surface of FIG. 30, and the total current supplied to the slot 4 is supplied from 10A to the upper side of the paper surface of FIG. 30A is supplied from the upper side to the lower side of the paper surface of FIG. 30, and the total current supplied to the slot 10 is supplied from 10A to the upper side of the paper surface of FIG. As a result, a magnetomotive force acts between the slot 1 and the slot 4 in the direction from the back side to the front side on the paper surface of FIG. 30, and between the slot 4 and the slot 7, the magnetomotive force acts on the paper surface in FIG. A magnetomotive force acts between the slot 7 and the slot 10 from the back side to the front side on the paper surface of FIG. 30, and a magnetomotive force acts between the slot 10 and the slot 1 on the paper surface of FIG. Works. These magnetomotive forces are a state in which the U1 phase and the U2 phase are correctly acting as a U-phase current in a normal three-phase current.

  Next, the U1 phase terminal 701 and the U2 phase terminal 702 shown in FIG. 30 are supplied with a feeding current If2 that is a field power component for supplying field power instead of the U phase current component, so that the U1 phase terminal 701 is supplied. It is assumed that the U1 phase current Iu1 = 2A and the U2 phase current Iu2 = −2A are supplied to the U2 phase terminal 702. At this time, the current flowing through the V-phase winding and the W-phase winding is zero. The total current energized in the slot 1 is 4A energized from the top to the bottom of the page of FIG. 30, and the total current energized in the slot 4 is 0A because the U1 phase current Iu1 and the U2 phase current Iu2 cancel each other. The total current energized to 7 is energized by 4A from the bottom to the top of the page of FIG. 30, and the total current energized to the slot 10 is 0A because the U1 phase current Iu1 and the U2 phase current Iu2 cancel each other. As a result, a reverse current 4A is supplied to the slot 1 and the slot 7, and a magnetomotive force acts between the slot 1 and the slot 7 on the right side in FIG. A magnetomotive force acts between the slot 1 located at the right end of the slot from the front side to the back side on the paper surface of FIG. The circumferential width of these magnetomotive forces is 360 ° in electrical angle, and the circumferential period of the magnetomotive force is 720 °. Further, these magnetomotive forces are equivalent to excitation with an excitation winding having a circumferential winding pitch of 360 ° and a winding period of 720 °.

  In the explanation of FIG. 35, the receiving winding and the field winding will be described in detail later with reference to an arrangement diagram of the windings. If the winding pitch of the receiving winding is 360 °, the U phase and the V phase will be described. The winding period of the W-phase three-phase current is 360 °, and the total magnetomotive force in the circumferential direction 360 ° with respect to the three-phase current is zero, so that the magnetomotive forces of the three-phase currents Iu, Iv, Iw are the receiving windings. There is in principle no electromagnetic action on the wire. The power receiving winding receives the electromagnetic action of the feeding current If2 which is a field power component, and supplies power for energizing the field current to the field winding from the stator side to the rotor side by electromagnetic induction action. To do. In the above example, the U1 phase current Iu1 = 2A and a constant moment are considered. However, the feeding current If2 is an alternating current, and field power can be supplied by electromagnetic induction between the stator and the rotor without contact. it can.

  Next, another embodiment of claim 1 will be described. FIG. 34 shows an example of a motor configuration in which the motor configuration of FIG. 1 and the motor configuration of FIG. 29 are combined. The U1-phase winding and U2-phase winding wound in slot 1, slot 4, slot 7, and slot 10 in FIG. 34 are the same as those in FIG. 29, and the V-phase winding and W-phase wound in the other slots. The winding is the same as that shown in FIG. The aim of the motor of FIG. 34 is that the field power supply efficiency is slightly better in the configuration of FIG. 29, and the motor of FIG. 1 is better in terms of simplifying the windings. It is to make use of points. Thus, the motor of the present invention can be applied and modified in various forms such as distributed winding and short winding.

  Next, another embodiment of claim 1 will be described. FIG. 35 shows an example of the motor of the present invention having three phases, four poles, short winding and concentrated winding. The number of motor slots in FIG. 1 and FIG. 29 matched with twelve, whereas the number of motor slots in FIG. 35 is as small as six, and concentrated windings are excellent in productivity. The cost can be reduced. 342 is a U1-phase winding and 344 is a U2-phase winding. 34C and 34A are V-phase windings. Reference numerals 346 and 348 denote W-phase windings. The method shown in FIG. 3 can be applied to the energization method of the motor of FIG. 35, and field power can be supplied to the field winding of the rotor by the method shown in FIG. The rotor 1E is the same as other motors such as FIG.

  The three-phase concentrated winding motor is superior in winding manufacturability, shortening the coil end length of the winding, increasing the winding space factor, and enabling automatic winding / high speed winding. There is sex. For this reason, it is possible to realize high performance, miniaturization, and low cost of the motor, and it is widely used. However, since the performance of field weakening control by three-phase current is low, it is often used in applications that do not require field weakening characteristics so much. Therefore, by performing the field current control of the present invention, the field control performance is greatly improved, and it can be expected to be used in a region that could not be used until now.

  Next, an arrangement connection diagram of the power receiving winding and the field winding of the rotor will be described with reference to FIG. This rotor is a power receiving winding and a field winding of the rotor shown in FIGS. 1, 3, 18, 18, 29, 34, 35 and the like. FIG. 33 is a developed view in which the circumferential shape of the rotor surface is developed linearly. The horizontal axis in FIG. 33 is the electrical angle in the circumferential direction of the rotor, and since it is a 4-pole rotor, it indicates from 0 ° to 720 °. 33A and 33B shown in FIG. 33A are N pole magnetic poles of the rotor, and 339 and 33D are S pole magnetic poles. The tortoiseshell type coil 33F is a power receiving winding JD1 having a winding pitch of 360 ° in electrical angle, corresponds to the coil end 27 in FIG. 1, and corresponds to the power receiving winding 38 in FIG. On the other hand, the magnetomotive force having an electrical angle of 360 °, which is excited by the currents of the U-phase, V-phase, and W-phase windings on the stator side, is essentially zero. It is. Therefore, the output terminals 333 and 334 of the power receiving winding JD1 have a 360 ° cycle excited by the currents Iu, Iv, and Iw of the expression (7) of the U-phase, V-phase, and W-phase windings on the stator side. In principle, it is not affected by magnetomotive force. The tortoiseshell type coil 33G is also a power receiving winding JD2 having a winding pitch of 360 ° in electrical angle, corresponds to the coil end 28 in FIG. 1, and corresponds to the power receiving winding 3A in FIG. The power receiving winding JD2 is arranged so as to have a circumferential phase difference of 180 ° in electrical angle with respect to the power receiving winding JD1 of the turtle shell-shaped coil 33F. The output terminals of the power receiving winding JD2 are 335 and 336.

  It has been explained that if the winding pitch of the receiving winding is 360 ° in electrical angle, it is not affected by the three-phase current of the stator, but the winding pitch of the receiving winding is twice that of 360 ° in electrical angle, Even if it is an integer multiple such as three times, it is not affected by the three-phase current of the stator. Even in this case, the field power can be supplied to the power receiving winding on the rotor side by the feeding current on the stator side. Similar effects can be obtained if the winding pitch of the power receiving winding is an electrical angle of around 360 °. These are also included in the present invention.

  Next, voltages induced in the power receiving windings JD1 and JD2 will be described. FIG. 32 is a time chart to be described in detail later, and will be described using these waveforms. Now, let it be assumed that the feeding current component If2 flows from the terminal 701 in FIG. 30 to the terminal 702, and the current waveform is as shown in FIG. Then, when the slot 1 and the slot 7 shown in the layout diagram of the winding in FIG. 30 are at a position where the rotor rotates at a position facing the tortoiseshell type coil 33F of the rotor in FIG. And 334, the voltage Vrc1 shown in FIG. At that time, the power receiving winding JD2 is at the rotor rotation position where a phase difference of 180 ° in electrical angle is generated with respect to the phase of the magnetic flux having a period of 720 ° in electrical angle generated by the feeding current If2, so No voltage is generated at the output terminal, unlike Vrc2 in FIG.

  When the rotor is rotated from this rotor rotation position, the amplitude of Vrc1 decreases and the amplitude of Vrc2 increases in synchronization with the rotation position. At the position where the rotor rotation position is rotated by 180 ° in electrical angle, the voltage of Vrc1 becomes zero, and the voltage of Vrc2 that was zero becomes the maximum value as shown in FIG. Thus, depending on the rotor rotational position, a voltage due to the feeding current If2 is generated in either the power receiving winding JD1 or the power receiving winding JD2.

  The voltage Vrc1 is rectified to a DC voltage by the full-wave rectifier 39 in FIG. 3, and the voltage Vrc2 is rectified to a DC voltage by the full-wave rectifier 3B in FIG. The sum of the outputs is applied to the field windings 3C, 3D, 3E, and 3F, and the field current If is energized. The waveform of the field current If is a smoothed waveform such as the estimated value Ifs of the field current If shown in FIG. The time constant of the field circuit composed of the full wave rectifier and the field winding is a large time constant such as 100 msec, and the output of the full wave rectifier has a pulsating voltage component having a frequency twice as high as 500 Hz. Even if you have it, it is smooth enough. These field windings 3C, 3D, 3E, and 3F correspond to the tortoiseshell type coils 338, 33A, 33C, and 33E in FIG. The field current If is supplied from the terminals 331 and 332 in FIG.

  FIG. 33 (b) is a modified example of FIG. 33 (a), in which a receiving winding for connecting the tortoiseshell type coils 33M and 33N in series and performing flux linkage equivalent to the tortoiseshell type coil 33F is provided. Realized. Note that the power receiving winding in FIG. 33 (b) corresponding to the turtle shell coils 33G and 33J in FIG. 33 (a) is omitted because the arrangement and connection diagram is complicated and difficult to see. The feature of the configuration of FIG. 33 (b) is that the tortoiseshell type coil can be manufactured at the pitch of the rotor magnetic poles, so that the rotor winding is easily manufactured. As shown in FIG. 33 (b), the present invention also includes a configuration of the power receiving winding in which the winding pitch is an integral multiple of 360 °. Note that FIG. 33 (b) is disadvantageous because there are overlapping winding portions, but the number of turns of the power receiving winding is much smaller than the number of turns of the field winding. There is little problem with the winding space, and it is sufficient to evaluate which is better.

Next, an embodiment of claim 8 will be described. Although the U1-phase winding 31 and the U2-phase winding 32 are shown in FIG. 3, both windings also serve as a power supply winding for supplying field power from the stator side to the rotor side. It has been shown that when the AC voltage and AC current If2 indicated by the arrow 3G in FIG. 3 are supplied, the field current If is supplied to the field windings 3C, 3D, 3E, and 3F on the rotor side. The relationship between the U1 phase current Iu1 and the U2 phase current Iu2 and the feeding current If2 which is 3G in FIG. 3 is expressed by the following equation (23) from the equations (8) and (9).
If2 = (Iu1-Iu2) / 2 (23)
The U1-phase current Iu1 is driven and energized by the IGBTs T1 and T2 shown in FIG. 3, and the U2-phase current Iu2 is driven and supplied by the IGBTs T3 and T4 shown in FIG.

  Next, in order to explain a little more detailed operation, the block diagram of FIG. 31 is shown. FIG. 32 is a time chart showing signals and currents of the respective parts in FIG. 31 and will be described together. FIG. 31 is a block diagram for controlling the speed of the motor 739 shown in FIGS. 1 and 37, and the following three points are added to a general three-phase AC motor speed control device. First, as shown in FIG. 3, the number of IGBTs is increased from six to eight, and the number of current detectors is increased so that four currents can be driven. Second, a 74 L feeding current control unit indicated by a one-dot chain line for controlling the feeding current If2 which is a field power component is added. The third is an unbalance compensation unit that corrects an unbalance amount between the U-phase current component in the U1-phase current expressed by the equation (8) and the U-phase current component in the U2-phase current expressed by the equation (9). Has been added. The unbalance compensation unit is 74M indicated by a one-dot chain line.

  73A is a rotor position detector mechanically connected to the motor 739 of the present invention, and its output signal is connected to its interface 74E to output a speed signal 742 and a rotational position signal 74F. 738 is a four-phase inverter composed of eight IGBTs T1, T2, T3, T4, T5, T6, T7, and T8 in FIG. 3, and includes a U1-phase current Iu1, a U2-phase current Iu2, a V-phase current Iv, The W phase current Iw is output. 73B is a value Iu1s obtained by detecting the U1 phase current Iu1 driven by the IGBT T1 and T2 by the current detector 74W, and 73C is a current detector 74X that detects the U2 phase current Iu2 driven by the T3 and T4 of the IGBT. The values Iu2s and 73D are the values Iv1s and 73E obtained by detecting the V-phase current Iv driven by the IGBTs T5 and T6 by the current detector, and the values Iv1s and 73E are the W-phase current Iw driven by the IGBTs T7 and T8. This is the value Iw1s detected in (1). 73G is an adder that obtains the U-phase current detection value Ius from Iu1s and Iu2s from the equation (10). 74G is a converter for converting U, V, and W axis signals of fixed coordinates to d and q axes of rotation coordinates, 74A is a q axis current Iqs, and 74B is a d axis current Ids.

  741 is a motor speed command, and an adder 743 subtracts the speed signal 742 from the speed command 741 and feeds it back to the proportional integrator 744. Reference numeral 748 denotes field control information, which is information such as a field control mode. The field control unit 747 receives the speed signal 742 and outputs 74U that is the d-axis current command Idc and 74V that is the field current command Ifc. Reference numeral 745 denotes an adder that subtracts the q-axis current Iqs from the q-axis current command Iqc and performs feedback. The output is output to the proportional integrator 746, and the q-axis voltage command Vqc is output. 749 is an adder for subtracting the d-axis current Ids from the d-axis current command Idc and performing feedback. The output is output to the proportional integrator 749, and the d-axis voltage command Vdc is output. 74D is a converter for converting d and q axis signals of rotating coordinates into U, V and W axis signals of fixed coordinates. Its output 741 is a U-phase voltage command Vuc, output 732 is a V-phase voltage command Vvc, and output 733 is W. Phase voltage command Vwc. A pulse width modulator 737 outputs a U1-phase gate signal, a U2-phase gate signal, a V-phase gate signal, and a W-phase gate signal to the 4-phase inverter 738.

  Next, in the feeding current control unit 74L, the adder 73F sets a value If2s × 2 that is twice the feeding current detection value, which is the difference between the detection value Iu1s of the U1 phase current and the detection value Iu2s of the U2 phase current ( It stops based on Formula 8 and Formula (9). The computing unit 731 divides If2s × 2 by 2, and calculates the absolute value thereof to obtain the estimated field current value Ifs. When the output 74Z of the field fluctuation command unit 74T is zero, the output 74Y of the adder is the same as the field current command Ifc, and the adder 73M subtracts and feeds back the estimated field current Ifs, and the proportional integrator 73N Output to. 74S is a rectangular wave having a size of +1 and -1, and generates a 500 Hz rectangular wave, for example. A multiplier 73P multiplies the rectangular wave by the field voltage command Vfc output from the proportional integrator 73N, and outputs a power supply voltage command Vf2c. Adder 735 adds U-phase voltage command Vuc and field voltage command Vfc, and outputs U1-phase voltage command Vu1c. Adder 736 subtracts U-phase voltage command Vuc and field voltage command Vfc, and outputs U2-phase voltage command Vu2c. With the above operation, the voltage and current of the four phases are controlled, and the field current If is controlled.

  A signal relating to the field current is shown in the time chart of FIG. Now, it is assumed that U-phase voltage command Vuc is a certain value and (a) in FIG. Further, assuming that the field voltage command Vfc, which is the output of the proportional integrator 73N, is a certain value, the power supply voltage command Vf2c shown in FIG. 32B is output. Alternating current is 500 Hz. As described above, the frequency may be varied. The U1-phase voltage command Vu1c is the sum of the U-phase voltage command Vuc and the power supply voltage command Vf2c, and is (c) in FIG. The U2-phase voltage command Vu2c is the difference between the U-phase voltage command Vuc and the power supply voltage command Vf2c, and is (d) in FIG. When energizing according to these commands, the feeding current If2 is energized by the difference between them, and the feeding current detection value If2s which is ½ of the output of the adder 73G is as shown in FIG. The estimated field current value Ifs, which is the output of the arithmetic unit 73J, is (i) in FIG. The field current estimated value Ifs is subtracted from the field current command Ifc by the adder 73M and fed back.

  On the other hand, the operation on the rotor side is as follows. For example, the motor 739 is the motor shown in FIGS. 29 and 30, and the slot 1 and the slot 7 shown in the winding layout diagram of FIG. When the rotor rotational position is at a position facing the coil 33F, the voltage Vrc1 shown in FIG. 32 (f) is generated at the output terminals 333 and 334 of the turtle shell type coil 33F. At that time, the power receiving winding JD2 is at the rotor rotation position where a phase difference of 180 ° in electrical angle is generated with respect to the phase of the magnetic flux having a period of 720 ° in electrical angle generated by the feeding current If2, so No voltage is generated at the output terminal as in the case of Vrc2 in FIG. As described above, when the rotor is rotated from this rotor rotation position, the amplitude of Vrc1 decreases and the amplitude of Vrc2 increases in synchronization with the rotation position.

  The unbalance compensator 74M performs feedback control so that the U phase current components of the U1 phase current Iu1 and the U2 phase current Iu2 do not become unbalanced. 73H converts the rotational coordinate form into a fixed coordinate form and outputs a U-phase current command Iuc. The adder 74P adds Iuc / 2 and the feeding current command If2c to create a U1-phase current command Iu1c. The adder 74Q subtracts the U1-phase current detection value Iu1s from the U1-phase current command Iu1c and feeds it back to obtain an unbalance amount of the U-phase current component included in the U1-phase current, and the unbalance amount is used as the U1-phase voltage command. In addition, the unbalance amount is compensated and controlled by subtracting from the U2-phase voltage command. The unbalance compensation unit 74M is effective in establishing the expressions (8) and (9) with high accuracy.

  In this way, the feeding current If2 is detected, the estimated value Ifs of the field current If is calculated and obtained, and the field current If can be controlled. As a result, the motor can be driven with high torque, and the rotation region can be driven widely up to high-speed rotation with field weakening control. And since field power can be easily supplied to the rotor side in a non-contact manner, it is excellent in terms of high efficiency, miniaturization, and low cost of the motor. In FIG. 31, the example of the feedback configuration using the adder 73M and the like to supply the feeding current If2 has been described, but the feeding current If2 may be driven more simply. For example, an AC voltage may be simply superimposed on the U1-phase drive voltage and the U2-phase drive voltage. If the transfer function of the power supply path is relatively simple, the power supply current If2 can be supplied in proportion to the superimposed AC voltage.

  Next, an embodiment of claim 9 will be described. FIG. 36 is a winding arrangement diagram showing an example of a motor to which a power feeding winding PSWS for energizing a feeding current If2 is dedicated. FIG. 36 is shown by the same description method as the winding layout diagram of FIG. The tortoiseshell type coil 72C has a winding pitch of 360 ° in electrical angle. Tortoiseshell type coils 726, 727, 728, and 729 are U-phase windings. The V-phase winding and the W-phase winding shown on the lower side in the drawing of FIG. 36 are the same as the V-phase winding and the W-phase winding shown on the lower side of the drawing in FIG. The power supply winding 72C generates a magnetic flux PSWF having an electrical angle of 360 ° in the circumferential direction between the stator and the rotor by supplying an AC power supply current If2 to the input terminals 72A and 72B. Two power receiving windings having a rotor side electrical angle of 360 ° pitch and a phase difference of 180 ° electrical angle are configured to receive the magnetic flux PSWF by a combination of the two power receiving windings. . It can be said that the magnetic flux PSWF has an electrical angle of 360 ° in the circumferential direction and its cycle is an electrical angle of 720 °.

  Thus, by providing the power supply winding PSWS, the magnetic flux PSWF can be generated at a free place, so that field power can be supplied relatively efficiently depending on the motor structure. Further, since the feeding current and the phase current of the motor are separated, the current driving can be simplified. Since the dedicated power supply winding PSWS 72C is provided in this way and is configured independently of the normal three-phase winding, the number of windings increases and a drive circuit dedicated to the supply current If2 is required. In that respect, it gets complicated. On the other hand, however, compared to the configuration shown in FIG. 30, there is a feature that the burden on the three-phase winding can be reduced and the burden on the inverter shown in FIG. 3 can be reduced. Further, although the magnitude of the feeding current If2 depends on the motor size, it is 1/10 or less compared to the maximum phase current, and the space burden on the feeding winding PSWS is small. Further, the power supply winding PSWS can be devised to simplify the winding and the drive circuit, such as sharing a part of the winding with the phase winding of the motor.

  Next, an embodiment of claim 10 will be described. FIG. 37 shows a 5-phase motor with 5 phases, 4 poles, 10 slots, and a short pitch winding of 144 ° winding pitch. 37 in FIG. 37 is a stator core, and the number of slots is ten, which is half the number of full-pitch winding slots shown in FIG. Expressing the electrical angle from 0 ° to 720 °, 352 and 355 are A1 phase windings wound around the slots at the electrical angles of 0 ° and 144 °, and 354 and 357 are electrical angles of 72 ° and 216 °. B-phase winding wound around the slot of the position, 356 and 359 are C-phase winding wound around the slot at the electrical angle of 144 ° and 288 °, and 358 and 35B are at the electrical angle of 216 ° and 360 °. D-phase windings wound around slots, 35A and 35D are E-phase windings wound around slots at electrical angles of 288 ° and 72 °, 35C and 35F are slots at electrical angles of 360 ° and 504 ° A2 phase winding to be wound, 35E and 35H are B phase winding to be wound around slots at electrical angles of 432 ° and 576 °, and 35G and 35K are wound to slots at electrical angles of 504 ° and 648 ° C-phase windings, 35J and 351 have electrical angles of 576 ° and 7 D phase winding to be wound to 0 ° position slot, 35L and 353 are E-phase winding to be wound into position in the electrical angle 648 ° and 72 ° slot. The rotor 1E is similar to other motors such as FIG. 1, and various rotors shown in the present invention can be used.

  FIG. 38 shows an arrangement layout of windings of the five-phase motor shown in FIG. The 368 Tortoiseshell type coil is an A1 phase winding, and supplies an A1 phase current Ia1 from an A1 phase terminal 361. The 369 Tortoiseshell type coil is an A2 phase winding, and conducts an A2 phase current Ia2 from an A2 phase terminal 362. The turtle shell type coils 36A and 36E are B-phase windings, and a B-phase current Ib is supplied from a B-phase terminal 363. The tortoiseshell type coils 36B and 36F are C-phase windings, and a C-phase current Ic is supplied from a C-phase terminal 364. The tortoiseshell type coils 36C and 36G are D-phase windings, and conduct a D-phase current Id from a D-phase terminal 365. The 36D and 36H tortoiseshell type coils are E-phase windings, and an E-phase current Ie is supplied from an E-phase terminal 366. The phase order of the five-phase alternating current is the order of the A phase, the B phase, the C phase, the D phase, and the E phase, and the windings are arranged in the circumferential direction according to this phase order. The other end of each phase winding is connected to 367, the neutral point of the star connection. The current of each phase can be energized by the method shown in FIG. 11 to drive the five-phase motor.

  The characteristics of the five-phase motor shown in FIGS. 37 and 38 are that the winding pitch is a short-pitch winding with an electrical angle of 144 °, thereby reducing the crossing with the other-phase winding at the coil end portion. This is to simplify the winding structure. The number of slots in FIG. 37 is ten, which is half the number of full-pitch winding slots shown in FIG. 10, and windings of two phases are wound around each slot. The number of windings wound between the slots is ten, and the number of windings shown in FIG. 14 is the same. The number of times that the winding wound between the slots intersects with the winding of the other phase at the coil end portion is also halved. Compared with the motor of FIG. 10, the winding crossing of the coil end portion can be greatly simplified, so that the motor can be reduced in size and productivity can be reduced. Functionally, similarly to the full-pitch five-phase motor shown in FIG. 10, field weakening and strengthening control by d-axis current control is possible. Note that the power supply winding PSWS is wound around an appropriate slot in the five-phase motor of FIG. 37, and only the A-phase current is supplied to the A1-phase winding and the A2-phase winding.

Next, an embodiment of claim 11 will be described. FIG. 39 shows a 5-phase motor with 5 phases, 8 poles, 10 slots, and concentrated windings. The winding pitch is 144 °. Expressing the electrical angle from 0 ° to 1440 °, 395 is an A1 phase winding wound around the slots at electrical angles 0 ° and 144 °, 396 is to the slots at electrical angles 144 ° and 288 ° A winding C-phase winding 397 is wound around a slot at an electrical angle of 288 ° and 432 °, and an E-phase winding 398 is wound around a slot at an electrical angle of 432 ° and 576 °. 399 is a D-phase winding wound around slots at electrical angles 576 ° and 720 °, 39A is an A2-phase winding wound around slots at electrical angles 720 ° and 864 °, and 39B is an electrical angle 864. C-phase winding wound around the slots at the positions of 100 ° and 1008 °, 39C is the E-phase winding wound around the slots at the positions of the electrical angles of 1008 ° and 1152 °, and 39D is the positions of the electrical angles of 1152 ° and 1296 ° B-phase winding wound around the slot, 39E is electrical angle 12 This is a D-phase winding wound around slots at positions of 96 ° and 1440 °. The phase sequence of the five-phase alternating current is the sequence of the A phase, B phase, C phase, D phase, and E phase, and the windings of this motor are not arranged in the circumferential direction in accordance with the phase sequence. The rotor 1E is similar to other motors such as FIG. 1, and various rotors shown in the present invention can be used. The current of each phase can be energized by the method shown in FIG. 11 to drive the five-phase motor.

  The features of the five-phase motor of FIG. 39 are that the windings are concentrated windings, and that the winding pitch is a reasonably large electrical angle of 144 ° and the winding coefficient is as large as 0.95. Concentrated winding motors are excellent in winding manufacturability and have advantages in that the coil end length of the windings can be shortened, that the winding space factor can be increased, and that windings can be automatically wound / speeded up. . As a result, high performance, miniaturization, and cost reduction of the motor can be realized. On the other hand, the conventional concentrated winding has a problem that it is difficult to perform field weakening and strengthening control by the d-axis current control with the stator winding as in the full-pitch five-phase motor shown in FIG. However, in the present invention, since the field current can be controlled from the stator side, it is possible to realize a large torque at a low speed rotation and to drive with a wide rotation range up to a high speed rotation. Note that the power supply winding PSWS is wound around an appropriate slot in the five-phase motor of FIG. 37, and only the A-phase current is supplied to the A1-phase winding and the A2-phase winding.

Next, an embodiment of claim 12 will be described. FIG. 41 is a diagram illustrating the A-phase voltage, which is one phase in the five-phase motor, and the output power of the five-phase motor. Although the horizontal axis represents time, the time of one sine wave cycle is shown with 360 scales. The vertical axis shows both voltage and power. Reference numeral 402 denotes an A-phase sine wave voltage Va having an amplitude of 1. Now, it is assumed that the A-phase current Ia has an amplitude of 1 and the waveform shape is the same as Va. Similarly, the B phase is a B phase voltage Vb and a B phase current Ib whose phase is 72 ° behind the A phase, and the C phase is a C phase voltage Vc and a C phase current Ic whose phase is 72 ° behind the B phase. The phase is a D-phase voltage Vd and a B-phase current Id whose phases are delayed by 72 ° from the C-phase.
It is assumed that the E phase is an E phase voltage V and a B phase current Ie whose phase is delayed by 72 ° from the D phase. The shape is similar to that of the five-phase sine wave current shown in FIG. The five-phase power P5 at this time is 2.5 as shown in the following equation. It is 405 in FIG.
P5 = Va · Ia + Vb · Ib + Vc · Ic + Vd · Id + Ve · Ie = 2.5

  FIG. 12 shows a technique of rectangular wave and trapezoidal wave in order to increase the five-phase power P5 from 2.5 to 4.0. Here, a method of adding the third harmonic to the sine wave will be described. In FIG. 41, 401 is the third harmonic and its amplitude is 0.12. The value obtained by adding the sine wave 402 and the third harmonic is 403, the peak value is 0.89, and the vicinity of the peak value is flat and has a substantially constant value. Now, assuming that there is a rated voltage and a rated current of the IGBT that is the power element of the inverter, and that it is used with a certain allowable value, when controlling with the voltage and current of the waveform 403, the peak value is 0.88. Therefore, there is room and waste occurs. Therefore, both voltage and current can be set to 1 / 0.88 = 1.1364 times.

On the other hand, the characteristics of the five-phase voltage and current power P6 including the third harmonic will be described. Here, K3 = 0.12 and K4 = 1 / 0.88 = 1.1364.
Va = Ia = K4 {sin (wt) + K3 × sin (3 · wt)
Vb = Ib = K4 {sin (wt-72 °) + K3 × sin (3 · wt-72 °)
Vc = Ic = K4 {sin (wt-144 °) + K3 × sin (3 · wt-144 °)
Vd = Id = K4 {sin (wt−216 °) + K3 × sin (3 · wt−216 °)
Ve = Ie = K4 {sin (wt-288 °) + K3 × sin (3 · wt-288 °)
Then, the power P6 is expressed by the following equation.
P6 = Va * Ia + Vb * Ib + Vc * Ic + Vd * Id + Ve * Ie

  The five terms of the A, B, C, D, and E phases in this equation are 3 in the terms of the square of the fundamental wave, the product of the fundamental and the third harmonic, and the square of the third harmonic, respectively. This is the term expression. The sum of the five phases of the square term of the fundamental wave is 1.1364 × 1.1364 × 2.5 = 3.2285, and the sum of the five phases of the product term of the fundamental wave and the third harmonic is It becomes zero, and the sum of the five phases of the square term of the third harmonic is 0.004649. As a result, the five-phase power P6 is 406 in FIG. 41 and is 3.275. By driving with the voltage and current to which the third harmonic is added, the output power of the inverter and the motor can be increased by about 31%. Further, the waveform of 404 including the third harmonic does not have a steep change and has a smooth shape, so that the current of each phase can be controlled with relatively high accuracy without causing a large error.

  Here, another problem is that the induced voltage waveform of the five-phase motor needs to be designed to include a third-order harmonic as indicated by 404 in FIG. In this regard, the slightly trapezoidal induced voltage characteristic of 404 is far easier than the induced voltage characteristic of the sinusoidal shape of 402 from the structure of the motor and from design experience. In a three-phase motor, driving with a current including the third harmonic is impossible because the phases of the three-phase third harmonic are aligned. In the case of a five-phase motor, the sum of each third harmonic becomes zero, so there is no inconvenience in driving current.

  Next, an embodiment of claim 13 will be described. A main motor for an electric vehicle includes a start-up from a stationary state and a slope driving, and the motor needs to be able to detect a sensorless position in a low-speed and high-torque state. When the impedance difference between the d-axis and the q-axis is small, or when the impedance difference between the d-axis and the q-axis is small due to magnetic saturation of the motor at high torque, the motor characteristics may become a problem. In the motor shown in FIG. 1 and the like, the rotor surface is a soft magnetic material, the impedance difference between the d-axis and the q-axis is small, and sensorless position detection at low speed is difficult.

  The field current command Ifc of the output 74V of the field control unit 747 in FIG. 31 is a signal for commanding the magnitude of the field current If, and commands the magnitude of the field magnetic flux. On the other hand, the output 74Z of the field fluctuation command unit 74T is an AC signal having a frequency lower than that of the feed current command If2c. For example, if the field current command Ifc is a constant value of 1.0 and the AC signal 74Z is added by an adder, the field current command Ifpc with variation, which is the output 74Y, is as shown in FIG. It becomes a waveform. Thus, for example, a field current command is generated so that the field magnetic flux increases or decreases at about 50 Hz.

  If the field current of the rotor increases or decreases as shown in FIG. 32 (j), even when the rotor is stationary, a voltage corresponding to the increase or decrease of the field magnetic flux is induced in the winding of each phase. The magnetic pole position can be detected. When the magnitude of the field current command Ifpc with fluctuation of 74Y increases, it is determined whether the voltage generated in the phase winding is positive or negative, and whether the rotor magnetic pole is N pole or S pole Can be detected. The sensorless position detection of the rotor magnetic pole can be performed by the method as described above. Of course, sensor-less position detection by conventional motor-induced voltage detection is also possible as long as the rotation speed exceeds a certain level. Moreover, both methods can also be used together.

  Although the present invention has been described above, various modifications, applications, and combinations are possible. For example, the number of poles can be changed, for example, by transforming a 4-pole motor into 8 poles. A motor configuration in which the first motor is arranged on the outer diameter side, the second motor is arranged on the inner diameter side, and a total of two motors are arranged and integrated is possible. In the configuration in which the first rotor is arranged on the outermost diameter side, the second rotor is arranged on the innermost diameter side, and the first stator and the second stator are arranged in the middle, the winding of the first stator By wiring the windings of the second stator to each other, a toroidal winding can be obtained, and the coil end portion and the like can be greatly simplified. In the rotor of FIG. 8 and the like, a permanent magnet having a magnetization direction perpendicular to the magnetic flux direction of the rotor magnetic pole can be disposed in the slit-shaped gap 83. Armature reaction due to the torque current of the stator can be reduced. It is also possible to reinforce the gap 83 by filling it with a resin or the like that is a non-magnetic material, or to reduce vibration. As a strengthening measure against the centrifugal force of the rotor in FIG. 8, a part of the laminated electromagnetic steel plates can be replaced with a nonmagnetic stainless steel plate having no gap 83. In this case, the laminated electromagnetic steel plate and the stainless steel plate are connected by an adhesive or the like. Various rotors can be used in the motor of the present invention. Further, the motor of the present invention can be operated as a generator. Moreover, an aluminum wire etc. can also be used as a kind of winding, It is not limited. As the soft magnetic material, not only electromagnetic steel sheets but also various materials such as a pressure magnetic core can be used. Various permanent magnets can be used, and the strength of the magnet can be varied during use. It is possible to vary the magnet with the motor current, or to add a dedicated magnetizing, demagnetizing and demagnetizing device.

11, 14, 17, 1A U phase winding 13, 16, 19, 1C V phase winding 15, 18, 1B, 11 W phase winding 1D Stator 1E Rotor 1F Rotor shaft 1J, 1K Field winding 1Z, 1M Field winding 1Q, 1R Field winding 1G, 1T Field winding 1H, 1P Power receiving winding 1L, 1S Power receiving winding 1V, 1X N-pole magnetic pole 1W, 1Y S-pole magnetic pole

Claims (13)

For AC motors with 4 or more PN poles and 3 or more MN phase stators and rotors,
QN is an integer equal to or greater than 2, and a power supply winding PSW that excites an AC magnetic flux component with a period of QN times 360 ° in electrical angle in the circumferential direction of the stator;
RN is an integer smaller than QN / 2, and the circumferential winding pitch is (360 ° × RN), and the first receiving winding PRW1 of the rotor that receives field power,
A second power receiving winding PRW2 of a rotor for receiving field power, the winding pitch in the circumferential direction being (360 ° × RN), which is disposed at a circumferential position different from the first power receiving winding PRW1; ,
A rectifier REC1 disposed in the rotor and rectifying the output of the first power receiving winding PRW1,
A rectifier REC2 disposed in the rotor and rectifying the output of the second power receiving winding PRW2,
A field winding FM wound around the N-pole magnetic pole or S-pole magnetic pole of the rotor or both of them,
A motor that excites the N-pole magnetic pole and the S-pole magnetic pole by supplying a field current If to the field winding FM using the output of the rectifier REC1 and the output of the rectifier REC2. The feed winding PSW includes a plurality of U-phase windings that are one of the MN phases,
A winding U1M which is one of U-phase windings;
A U-phase winding U2M arranged at a circumferential position different from the winding U1M in a circumferential range of (360 ° × QN) in electrical angle;
The stator side AC field excitation current component necessary for applying the field current If to the field winding FM is set as If2, and the field excitation current is superimposed on the U phase current component Iu1 to the winding U1M. A drive unit DRU1 for energizing the component If2,
2. The motor according to claim 1, further comprising: a drive unit DRU <b> 2 that energizes the winding U <b> 2 </ b> M with a field excitation current component (−If2) superimposed on the U-phase current component Iu <b> 2. In the north pole or south pole of the rotor,
A plurality of soft magnetic materials in parallel;
The motor according to claim 1, further comprising a gap, a resin, or a permanent magnet sandwiched between the soft magnetic bodies. The motor according to claim 3, further comprising a permanent magnet arranged in series in the direction of the magnetic flux and the magnetic flux. The stator is equipped with a three-phase star connection for each phase winding,
In the induced voltage are approximately trapezoidal waveform of induced voltage closer to a square wave magnetic flux component which is excited in the field current If of the field winding FM rotor of each of the winding phase,
The substantially trapezoidal positive and negative alternating current waveforms have an electrical angle width at which the positive and negative voltages of the induced voltage are inverted as θrr,
2. The motor according to claim 1, wherein (θrr + θst) ≦ 60 °, where θst is an electrical angle having a width in which the substantially trapezoidal positive or negative current changes from zero to 80% of the peak value. .
Equipped with a 5-phase star connection each phase winding on the stator,
In the induced voltage trapezoidal induced voltage waveform closer to a square wave magnetic flux component which is excited in the field current If of the field winding FM rotor of each of the winding phase,
The substantially trapezoidal positive and negative alternating current waveforms have an electric angle width at which the positive and negative voltages of the induced voltage are reversed as θrr, and the substantially trapezoidal positive or negative current has a peak value from zero. The motor according to claim 1, wherein (θrr + θsf) ≦ 36 °, where θsf is an electrical angle having a width that changes to 60%. It is a PN pole motor with 5 phases and 4 poles,
A-phase windings are placed at 0 ° and 180 ° electrical angles on the circumference of the stator,
B-phase windings are placed at 72 ° and 252 ° electrical angles on the circumference of the stator,
C-phase windings are arranged at 144 ° and 324 ° in electrical angle on the circumference of the stator,
D-phase windings are placed at 216 ° and 36 ° electrical angles on the circumference of the stator,
The E phase windings are located at 288 ° and 108 ° electrical angles on the circumference of the stator,
The number of full-pitch windings of each phase wound over two slots is expressed as (PN / 2-1
The motor according to claim 1, wherein: A current sensor CT1 for detecting a current Iu1 energized to the winding U1M by the drive unit DRU1,
A current sensor CT2 for detecting a current Iu2 energized to the winding U2M by the drive unit DRU2,
Adding means for obtaining If2s as a detected value of the AC field excitation current component on the stator side as (2 × If2s = Iu1s−Iu2s) from Iu1s which is the output of the current sensor CT1 and Iu2s which is the output of the current sensor CT2. The motor according to claim 2, further comprising: 2. The power feeding winding PSW is a field power feeding winding PSWS for exciting an AC magnetic flux component having a period of QN times 360 ° in electrical angle in the circumferential direction of the stator. The motor described in. A 5-phase motor with a number of poles that is an integral multiple of 4 poles,
Describe the circumferential position of each phase winding of the stator in terms of electrical angle in the range of 0 ° to 720 °, with the phase order of five-phase alternating current as the order of A phase, B phase, C phase, D phase, E phase. A phase A winding wound around a slot with an electrical angle of 0 ° and a slot with an electrical angle of 144 °;
Similarly, a B-phase winding wound around a slot of 72 ° in electrical angle and a slot of 216 ° in air angle;
Similarly, a C-phase winding wound around a slot having an electrical angle of 144 ° and a slot having an electrical angle of 288 °,
Similarly, a D-phase winding wound around a slot with an electrical angle of 216 ° and a slot with an electrical angle of 360 °,
2. The motor according to claim 1, further comprising: a slot having an electrical angle of 288 ° and an E-phase winding wound around the slot having an electrical angle of 72 °. A 5-phase motor having a number of poles that is an integral multiple of 8 poles,
Describe the phase position of the 5-phase AC in the order of A phase, B phase, C phase, D phase, E phase, and the circumferential position of each phase winding of the stator in the range from 0 ° to 1440 ° in electrical angle. A concentrated winding A-phase winding wound around a slot with an electrical angle of 0 ° and a slot with an electrical angle of 144 °;
Similarly, a concentrated-winding C-phase winding wound around a slot having an electrical angle of 144 ° and a slot having an air angle of 288 °,
Similarly, a concentrated winding E-phase winding wound around a slot with an electrical angle of 288 ° and a slot with an electrical angle of 432 °,
Similarly, a concentrated B-phase winding wound around a slot of 432 ° in electrical angle and a slot of 576 ° in electrical angle;
Similarly, a concentrated D-phase winding wound around a slot with an electrical angle of 576 ° and a slot with an electrical angle of 720 °,
Similarly, it is provided with an A-phase winding, a C-phase winding, an E-phase winding, a B-phase winding, and a D-phase winding that are wound in the same manner as described above in each slot of 720 ° to 1440 ° in electrical angle. The motor according to claim 1. A five-phase AC motor in which the phase order of the five-phase alternating current is the order of A phase, B phase, C phase, D phase, E phase,
The A phase voltage is a voltage that includes a third harmonic voltage that is three times the sine wave frequency in the sine wave voltage,
The B phase voltage is the same voltage as the A phase voltage having a phase difference of 72 ° with respect to the A phase voltage and the C phase voltage.
The C phase voltage is the same voltage as the A phase voltage having a phase difference of 72 ° with respect to the B phase voltage and the D phase voltage.
The D phase voltage is a voltage similar to the A phase voltage having a phase difference of 72 ° with respect to the C phase voltage and the E phase voltage.
The E phase voltage is the same voltage as the A phase voltage with a phase difference of 72 ° with respect to the D phase voltage and the A phase voltage. The A phase current is a sine wave current that is three times higher than the sine wave frequency. Including current,
The B phase current is the same current as the A phase current having a phase difference of 72 ° with respect to the A phase current and the C phase current.
The C phase current is the same as the A phase current with a phase difference of 72 ° with respect to the B phase current and the D phase current.
The D phase current is the same current as the A phase current having a phase difference of 72 ° with respect to the C phase current and the E phase current.
2. The motor according to claim 1, wherein the E-phase current is a current similar to the A-phase current having a phase difference of 72 [deg.] With respect to the D-phase current and the A-phase current. The field excitation current component If2 energized to the power supply winding PSW is increased by setting If2 as the AC field excitation current component on the stator side necessary for energizing the field current If to the field winding FM, By increasing and decreasing the field current If, the field current If is increased or decreased,
2. The motor according to claim 1, wherein a rotational position of the rotor is detected by detecting a change in a voltage component induced by electromagnetic induction between terminals of each phase winding of the stator at that time.

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