JP6146288B2 - Electric motor control device - Google Patents

Description

  The present invention relates to a technique for controlling a rotary motor including an armature and a rotor that rotates relative to the armature.

  In particular, the present invention relates to a technique for controlling a synchronous motor based on a magnetic flux (hereinafter also referred to as “gap magnetic flux”) flowing through a gap between an armature and a rotor.

  Conventionally, various types of so-called primary magnetic flux control have been proposed for controlling a rotary motor based on a primary magnetic flux. In brief, the primary magnetic flux control is a technique for stably controlling the rotary motor by controlling the primary magnetic flux of the rotary motor according to the command value.

  The primary magnetic flux is a combination of the field magnetic flux generated by the rotor, which is a field, and the magnetic flux of the armature reaction generated by the armature current flowing in the armature winding, and appears as a gap magnetic flux.

  For example, the phase of the field magnetic flux Λ0 is adopted as the d-axis of the rotating coordinate system, and the phase of the primary magnetic flux [λ1] (indicating that the symbol [] (square bracket) is a vector. The phase difference between the δ axis and the d axis is considered as the load angle φ. However, here, the γ-axis is adopted at a phase advanced by 90 degrees with respect to the δ-axis. Further, δc axis and γc axis are defined as the control axes of the rotating coordinate system employed for controlling the primary magnetic flux. The δc axis and the γc axis correspond to the δ axis and the γ axis, respectively, and the phase difference of the δc axis with respect to the d axis is φc.

  In primary magnetic flux control, by making primary magnetic flux [λ1] equal to its command value (hereinafter referred to as “primary magnetic flux command value”), phase difference φc becomes equal to load angle φ, and δc axis coincides with δ axis. To do. Therefore, in the primary magnetic flux control, the γc-axis component of the primary magnetic flux command value is normally selected as zero.

  In such primary magnetic flux control, the torque of the rotary motor can be controlled in proportion to the γc-axis component of the armature current without depending on the rotational angular velocity.

  In Patent Document 2, Non-Patent Document 2, and Non-Patent Document 4, the names of the coordinate axes (δ axis, γ axis) of the rotating coordinate system are replaced with those of Patent Document 1, Non-Patent Document 1, and Non-Patent Document 3. Has been.

Japanese Patent No. 3672761 Japanese Patent No. 3860031

Kaku, Yamamura, Tsunehiro, “Position Sensorless Control Method for DC Brushless Motor”, IEEJ Transactions D, 1991, Vol. 111, No. 8, p. 639-644 Yabe, Sakabe Sakabe, “Sensorless driving of IPM motors using overmodulation PWM”, IEEJ Technical Committee Materials. RM, Rotating Machine Research Group 2001 (159), p. 7-12 Kamizato and three others, “Rotor position sensorless vector control method for reluctance motors”, Proceedings of the 1994 National Conference on Industrial Applications of the Institute of Electrical Engineers of Japan, p. 59-64 Ito, Toyosaki, Osawa, “High performance V / f control of permanent magnet synchronous motor”, IEEJ Transactions D, 2002, Vol. 122, No. 3, p. 253-259 Sakata, Fujimoto, “Complete Servomotor Tracking Control Method Based on Strict Model Considering Current Loop and PWM Hold”, IEEJ Transactions D, 2007, Vol. 127, No. 6, p. 587-593

  In order to drive the motor with high efficiency by primary magnetic flux control, it is necessary to change the primary magnetic flux command according to the load. However, the conventional technology does not perform control in consideration of the transient behavior of the primary magnetic flux with respect to fluctuations in the primary magnetic flux command value. Therefore, there is a problem that the primary magnetic flux does not sufficiently follow the fluctuation of the primary magnetic flux command value.

  In the primary magnetic flux control, a control shaft speed that stabilizes the rotational speed is generated from the speed command and the current. This presupposes that the primary magnetic flux is rotating in synchronization with the rotation of the control shaft. However, the conventional technology does not perform control in consideration of the transient behavior of the primary magnetic flux when changing the primary magnetic flux command value. Therefore, the rotation of the control shaft and the primary magnetic flux becomes asynchronous, and this speed stabilization may not work correctly.

  In view of such a problem, an object of the present invention is to improve the follow-up of the primary magnetic flux with respect to fluctuations in the primary magnetic flux command value.

  A first aspect of the motor control device according to the present invention is a rotary motor including an armature having an armature winding through which an armature current ([i]) flows, and a rotor that rotates relative to the armature. Is controlled in a rotating coordinate system that rotates with a first axis (δc) and a second axis (γc) advanced by 90 degrees relative to the first axis in the rotational direction of the rotor. . The apparatus controls a gap magnetic flux ([λ1]) that is a magnetic flux flowing through a gap between the armature and the rotor.

  The motor control device includes a calculation unit (102), a phase calculation unit (106), and a first coordinate conversion unit (104).

  The motor control device obtains a second angular velocity (ω1) at which the rotating coordinate system rotates based on a command value (ω *) of a first angular velocity (ω0) at which the rotor rotates and the armature current. You may further provide a speed calculation part (108,109,110).

  The calculation unit uses the magnetic flux command value ([Λ1 *]), which is a command value in the rotating coordinate system of the air gap magnetic flux, and a differential value thereof, and an angular velocity (ω1) at which the rotating coordinate system rotates. First voltage command values ([v *], [v * ′]), which are command values in the rotating coordinate system, of the voltage applied to the electric motor are obtained.

  The phase calculation unit integrates the angular velocity to obtain a phase (θ) of the rotating coordinate system.

  The first coordinate converter converts the first voltage command value with the phase (θ) and converts the voltage applied to the rotary motor to a second voltage command value ([V *])

  A second aspect of the motor control device according to the present invention is the first aspect, wherein the calculation unit (102) further calculates a product of a resistance value of the armature winding and the armature current. To determine the first voltage command value.

  A third aspect of the motor control device according to the present invention is the second aspect, wherein the armature current is coordinate-transformed by the phase (θ), and the first-axis component of the armature current is obtained. A second coordinate conversion unit (101) is further provided for determining a first current value (iδc) that is a second value and a second current value (iγc) that is a component of the second axis of the armature current.

  The calculation unit (102) differentiates a product of the resistance value (R) and the first current value and a first axis magnetic flux command value (λ1δc *) that is a component of the first axis of the magnetic flux command value. The product of the second magnetic flux command value (λ1γc *), which is a component of the second axis of the magnetic flux command value, and the angular velocity is subtracted from the sum of the first differential value (sλ1δc *) obtained in this way and the first differential value (sλ1δc *). The component (vδc *, vδc * ′) of the first axis of the voltage command value is obtained.

  The calculation unit further includes a product of the resistance value and the second current value (iγc), a second differential value (sλ1γc *) obtained by differentiating the second magnetic flux command value, and the first The product of the second axis of the first voltage command value (vγc *, vγc * ′) is obtained by adding the product of the magnetic flux command value and the angular velocity.

  A fourth aspect of the motor control device according to the present invention is any one of the first to third aspects, wherein a deviation of the air gap magnetic flux ([λ1]) from the magnetic flux command value ([Λ1 *]). Is further provided with a correction unit (103) for correcting the first voltage command value ([v * ′]) using the result of proportional-integral-derivative control.

  A fifth aspect of the electric motor control device according to the present invention is the fourth aspect, in which an estimated value ([λ1 ^]) of the air gap magnetic flux ([λ1]) is obtained and the magnetic flux is given to the correction unit. A part (105) is further provided.

  According to the first to third aspects of the electric motor control apparatus according to the present invention, when the primary magnetic flux command value fluctuates in the electric motor control in the rotating coordinate system, the followability of the primary magnetic flux can be improved as compared with the conventional case. .

  According to the fourth aspect of the motor control device of the present invention, feedback on the primary magnetic flux can be incorporated into the control.

  According to the fifth aspect of the motor control device of the present invention, it is not necessary to directly detect the primary magnetic flux.

It is a block diagram which shows the structure of the electric motor control apparatus concerning 1st Embodiment, and its peripheral device. It is a block diagram which illustrates the composition of the 1st calculation part. It is a block diagram which shows the structure of the electric motor control apparatus concerning 2nd Embodiment, and its peripheral device. It is a block diagram which illustrates the composition of the 2nd calculation part. It is a block diagram which shows the structure of the motor control apparatus concerning a 3rd form, and its peripheral device. It is a block diagram which illustrates the structure of a primary magnetic flux estimation part. It is a block diagram which shows the process of a discrete time system about the part corresponded to a 1st calculation part and a 2nd calculation part.

Basic idea of the present invention.
Prior to the detailed description of the embodiments, the basic idea of the present invention will be described. Of course, this basic idea is also included in the present invention.

  In the primary magnetic flux control, the dq coordinate system (the d axis is in phase with the field magnetic flux and the q axis is advanced by 90 degrees with respect to the d axis) relative to the phase of the field magnetic flux (that is, the rotary motor A δc-γc coordinate system is set which is advanced by phase difference φc (with respect to rotation of the rotor). The voltage applied to the rotary motor is adjusted so that the δc axis coincides with the δ axis in phase with the primary magnetic flux.

  Considering the relationship between these two rotating coordinate systems, the voltage equation in the δc-γc coordinate system can be obtained from the voltage equation in the dq coordinate system as follows.

  First, the voltage equation in the dq coordinate system includes the d-axis voltage vd as the d-axis component of the voltage applied to the rotary motor, the q-axis voltage vq as the q-axis component of the voltage applied to the rotary motor, and the electric machine included in the rotary motor. The d-axis inductance Ld as the d-axis component of the inductance of the slave winding, the q-axis inductance Lq as the q-axis component of the inductance of the armature winding, the resistance component R of the armature winding, and the d-axis as the d-axis component of the armature current Introducing the current id, the q-axis current iq that is the q-axis component of the armature current, the field magnetic flux Λ 0, the rotational angular velocity ω 0 of the rotary motor, and the operator s of time differentiation are expressed by the following equation (1).

Considering that the field magnetic flux Λ0 does not depend on time, the equation (1) can be transformed into the equation (2). Here, the vector [Λ0 0] ^t (the superscript “t” after the parentheses indicates transposition of the matrix: the same applies hereinafter) represents the field magnetic flux Λ0 in the dq coordinate system. The expression in the symbol {} (curly brace) in the second term on the right side of Expression (2) represents the primary magnetic flux [λ1] in the dq coordinate system.

  Now, since the δc-γc coordinate system is advanced with a phase difference φc with respect to the dq coordinate system, an expression obtained by advancing equation (2) with the phase difference φc is obtained. Specifically, rotation transformation is performed on the equation (2) using the rotation matrix C represented by the equation (3). Thereby, Formula (4) is obtained.

The vector [vδc vγc] ^t obtained from the δc-axis voltage vδc, which is the δc-axis component of the voltage applied to the rotary motor, and the γc-axis voltage vγc, which is the γc-axis component of the voltage applied to the rotary motor, is the left side of equation (4). Matches. Therefore, the vector is expressed by equation (5).

  Next, the mathematical expression in the symbol {} of the second term on the right side of Expression (5) is modified. This expression represents the primary magnetic flux [λ1] in the δc-γc coordinate system. By substituting equation (3) for this equation, equation (6) is obtained.

  The first term on the right side of Equation (6) is a magnetic flux (armature reaction) generated by the armature current [i] flowing, and the second term is a magnetic flux caused by the field magnetic flux Λ0.

  Furthermore, Formula (8) is obtained by introducing Formula (7).

  By substituting equation (8) into the equation in the symbol {} of the second term on the right side of equation (5), equation (9) is obtained.

  Since the mathematical expression in the symbol {} of the second term on the right side of Equation (9) represents the primary magnetic flux [λ1], the δc axis component λ1δc and the γc axis component λ1γc are introduced to obtain Equation (10).

  Equation (10) can be further transformed into Equation (12) via Equation (11).

  However, although equation (12) is an equation for the δc-γc coordinate system, it depends on the rotational angular velocity ω0 of the rotary motor. In order to control the primary magnetic flux [λ1] in the δc-γc coordinate system, it is necessary to control based on the rotational angular velocity ω1 of the δc-γc coordinate system. Immediately after the primary magnetic flux command value fluctuates, it must be considered that the rotational angular velocity ω0 does not coincide with the rotational angular velocity ω1 and the phase difference φc varies temporally.

Paying attention to this point, equation (12) is modified. First, with reference to equation (3), the rotation matrix C ^-1 is specifically shown to obtain equation (13).

  Then, the operator s is operated to obtain the expression (14). Considering that the phase difference φc also varies with time as described above, Equation (15) is obtained by specifically showing the rotation matrix C with reference to Equation (3). However, points (dots) added on the phase difference φc indicate time differentiation.

Equation (16) is obtained by calculating a matrix that is a coefficient of the primary magnetic flux [λ1δc λ1γc] ^t in Equation (15).

  The time differentiation of the phase difference φc is a relative angular velocity of the δc−γc coordinate system with respect to the dq coordinate system, and therefore can be grasped as an angular velocity difference (ω1−ω0). Thereby, Expression (16) is transformed into Expression (17).

  By combining the second term and the fourth term on the right side of Equation (17), Equation (18) is obtained. This led to a voltage equation for controlling the primary magnetic flux [λ1] in the δc-γc coordinate system. Equation (18) is a voltage equation that reflects the transient state.

In order to control the primary magnetic flux [λ1] in the δc-γc coordinate system, the δc-axis voltage vδc and the γc-axis voltage vγc are determined based on the command values λ1δc * and λ1γc * of the δc-axis component λ1δc and γc-axis component λ1γc, respectively. The respective command values vδc * and vγc * are obtained. Therefore, based on the equation (18), the vector [vδc * vγc *] ^t is obtained by the equation (19). This represents a command value (hereinafter referred to as “voltage command value”) [v *] of the voltage applied to the armature winding. The vector [λ1δc * λ1γc *] ^t represents a primary magnetic flux command value.

The first term and the second term on the right side of Equation (19) do not include time differentiation, and the third term includes time differentiation. In the conventional technique, only the steady operation is considered when obtaining the voltage command vector [vδc * vγc *] ^t . Therefore, the rotational angular velocity ω1 employed in the matrix that is the coefficient of the primary magnetic flux [λ1δc λ1γc] ^t in the second term on the right side is identified with the rotational angular velocity ω0, or the third term is ignored. The third term can be regarded as a transient term corresponding to the fluctuation of the primary magnetic flux command value.

  For example, in Patent Document 1, in the derivation of Equation (8) for setting the voltage command value, the primary magnetic flux control in a steady state is premised, and the transient behavior of the primary magnetic flux with respect to fluctuations in the primary magnetic flux command value can be taken into consideration. Absent.

  Moreover, in patent document 2, in a formula (8) which shows a voltage equation, the steady state is assumed.

  Further, in Non-Patent Document 1, in the derivation of the equation (8) for setting the voltage command value, although starting from the equation corresponding to the equation (18) of the present application, the temporal variation of the primary magnetic flux command value is accurate. Not reflected.

  Further, in Non-Patent Document 2, although the expression (13) for setting the voltage command value is derived from the expression (7) corresponding to the expression (18) of the present application, the expressions (12a), (12b) ), The primary magnetic flux command value is set to a constant value.

  Non-Patent Document 3 does not distinguish between the rotor angular velocity (ωr) and the rotational coordinate system (δc-γc coordinate system referred to in the present application) for performing control, and the transient state cannot be sufficiently considered.

  Further, Non-Patent Document 4 employs a voltage command generation formula that assumes control in a steady state, and does not consider a transient state.

  Unlike these conventional techniques, by using the voltage command value [v *] represented by the voltage command vector obtained based on the equation (19), the primary magnetic flux command value is used in the motor control in the δc-γc coordinate system. When fluctuates, the followability of the primary magnetic flux can be improved as compared with the conventional case.

  Of course, with respect to the voltage command value obtained based on the temporal fluctuation of the primary magnetic flux command value as in the present invention, the current feedback term referred to in Patent Literature 1, the correction term referred to in Patent Literature 2, and the non-patent literature. A term for feeding back the magnetic flux error described in 2 may be added as a feedback term.

The right side of Equation (19) is a known amount of resistance component R of the armature winding, an observable amount of armature current [i] (= [iδc iγc] ^t ), a rotational angular velocity ω1 of the δc-γc coordinate system, and This is grasped as a feedforward amount determined based on the input of the primary magnetic flux command value [Λ1 *] (= [λ1δc * λ1γc *] ^t ). Thus grasp the left side of the equation (19) as [vδc * _F vγc * ^_F] again feed forward term written as ^t, is expressed a feedback term and ^{[vδc * _B vγc * _B]} t, the time of the primary magnetic flux command value The voltage command value [vδc * vγc *] ^{t in} consideration of not only the fluctuation but also the feedback is expressed by the equation (20).

  As described above, feedback on the primary magnetic flux can be incorporated into the control.

  In the following embodiments, a description will be given by introducing a command value (hereinafter referred to as “rotational angular velocity command value”) ω * of the rotational angular velocity ω0 of the rotary motor.

First embodiment.
FIG. 1 is a block diagram showing a configuration of an electric motor control device 1 according to the present embodiment and its peripheral devices.

  The rotary motor 3 includes an armature (not shown) and a rotor that is a field. As technical common sense, an armature has an armature winding. The rotary motor 3 is, for example, an embedded magnet type three-phase motor, and a field magnet is embedded in the field. In the rotary motor 3, the rotor rotates relative to the armature.

The voltage supply source 2 includes, for example, a voltage control type inverter and its control unit, and applies a three-phase voltage to the rotary motor 3 based on a three-phase voltage command value [V *] = [Vu * Vv * Vw *] ^t. To do. As a result, a three-phase current [I] = [Iu Iv Iw] ^t flows in the rotary motor 3. However, components included in the voltage command value [V *] and the three-phase current [I] are described in the order of, for example, a U-phase component, a V-phase component, and a W-phase component.

  The motor control device 1 is a device that controls the rotational angular velocity ω0 of the rotary motor 3 based on the primary magnetic flux command value [Λ1 *] and the armature current [i].

  The electric motor control device 1 includes coordinate conversion units 101 and 104, a first calculation unit 102, an integrator 106, a constant multiplication unit 108, a subtractor 109, and a high-pass filter 110.

  The coordinate conversion unit 101 performs δc−γc on the three-phase current [I] for primary magnetic flux control (however, it is not limited to the case where the γc-axis component of the primary magnetic flux command value is set to zero as in normal primary magnetic flux control). Convert to armature current [i] in the rotating coordinate system. Since such rotation conversion to the rotating coordinate system is a known technique, a detailed description thereof will be omitted.

  The first calculator 102 obtains the voltage command value [v *] in the δc-γc rotating coordinate system according to the equation (19). A specific configuration will be described later.

The coordinate conversion unit 104 converts the voltage command value [v *] to a voltage command value [V *] in another coordinate system applied to the rotary motor 3. This “other coordinate system” may be, for example, a dq rotating coordinate system, an α-β fixed coordinate system (for example, the α axis is set in phase with the U phase), or UVW. It may be a fixed coordinate system or a polar coordinate system. Which coordinate system is adopted as the “other coordinate system” depends on what kind of control the voltage supply source 2 performs. For example, when the voltage command value [V *] is set in the α-β fixed coordinate system, [V *] = [Vα * Vβ *] ^t (where the component shown above is the α-axis component and will be shown later) Is the β-axis component).

  The integrator 106 calculates the phase θ of the δc axis with respect to the α axis based on the rotational angular velocity ω1. That is, the integrator 106 functions as a phase detector that detects the phase of the δc-γc rotating coordinate system, more precisely, the phase of the δc axis with respect to the α axis.

  Based on the phase θ, the coordinate conversion units 101 and 104 can perform the above-described coordinate conversion, respectively.

  The rotational angular velocity ω1 of the δc-γc coordinate system is set as the output of the subtractor 109. A value obtained by removing the DC component from the γc-axis component iγc of the armature current by the high-pass filter 110 and further multiplying by a predetermined gain Km by the constant multiplication unit 108 is subtracted from the command value ω * of the rotational angular velocity ω0 by the subtractor 109. A rotational angular velocity ω1 is obtained. That is, the constant multiplication unit 108, the subtractor 109, and the high-pass filter 110 can be grasped as a speed calculation unit that obtains the rotational angular velocity ω1 based on the command value ω * and the armature current [i].

  The primary magnetic flux control is ideally performed, and in the steady state, the γc axis component iγc is constant, and ω1 = ω0 = ω *.

  FIG. 2 is a block diagram illustrating the configuration of the first calculation unit 102. In the figure, “X” surrounded by a circle represents a multiplier, “+” surrounded by a circle represents an adder, and “Yen” added a circle represents a subtractor. Since the resistance component R of the armature winding is known, it can be set in the first calculation unit 102.

The first calculator 102 receives the primary magnetic flux command value [Λ1 *] (= [λ1δc * λ1γc *] ^t ). In normal primary magnetic flux control, λ1γc * = 0 is set, so that only the command value λ1δc * may be input and λ1γc * = 0 may be set in the first calculation unit 102.

The first calculation unit 102 also receives an armature current [i] = [iδc iγc] ^t , and multiplies the resistance component R by the multipliers to generate values R · iγc and R · iδc. The values R · iγc and R · iδc are given to adders 1022 and 1021, respectively.

  Values (−ω1 · λ1γc *) and ω1 · λ1δc * are obtained by the multipliers and are given to adders 1021 and 1022, respectively.

  The adder 1021 is further given a (differentiated) value sλ1δc *. The adder 1022 is further given a (differentiated) value sλ1γc *.

  As described above, the first calculation unit 102 obtains the voltage command value [v *] according to the equation (19).

  The calculation in the first calculation unit 102 uses the primary magnetic flux command value [Λ1 *] as a command value in the δc-γc rotational coordinate system of the primary magnetic flux [λ1], a differential value thereof, and the rotational angular velocity ω1. It is understood that the voltage command value [v *], which is the command value in the δc-γc rotating coordinate system, of the voltage applied to is obtained.

  If the resistance component R of the armature winding cannot be ignored, the product of this and the armature current [i] is also added to obtain the voltage command value [v *].

  It is desirable that the primary magnetic flux command value [Λ1 *] used for the calculation of the first calculation unit 102 has a high frequency component removed by a low-pass filter. The reason is as follows.

  When the fluctuation of the value of the primary magnetic flux command value [Λ1 *] is steep, the voltage command value [v *] and, in turn, the voltage command value [V *] increases transiently. However, the voltage supply source 2 (for example, a voltage control type inverter) has a limit in the voltage that can be output. Therefore, if the voltage command value [V *] becomes transiently large, the voltage actually applied to the rotary motor 3 may be saturated. In such a case, naturally, the followability of the primary magnetic flux [λ1] with respect to the primary magnetic flux command value [Λ1 *] also deteriorates.

  Therefore, in order to avoid such deterioration in followability, even if the value of the primary magnetic flux command value [Λ1 *] is steep, the voltage command value [v * ] Should not be excessive.

  On the other hand, if the value of the primary magnetic flux command value [Λ1 *] is steep, the followability of the primary magnetic flux [λ1] can be obtained by removing the high frequency component and using it for the calculation of the voltage command value [v *]. Will lower.

  Therefore, it is desirable to provide the low-pass filter from the viewpoint of appropriate design in consideration of the voltage range applied to the rotary motor 3 and the followability of the primary magnetic flux.

Second embodiment.
FIG. 3 is a block diagram showing the configuration of the motor control device 1 according to the present embodiment and its peripheral devices. In the configuration of the motor control device 1 according to the present embodiment, the second calculation unit 103 includes a first calculation unit 102, a coordinate conversion unit 104, and the configuration of the motor control device 1 according to the first embodiment. Have been added between.

  The second calculation unit 103 applies a feedback term to the calculation result from the first calculation unit 102 (this is adopted as the voltage command value [v *] in the first embodiment) [v * ′]. Add and output as voltage command value [v *].

Specifically, the second calculation unit 103 receives the primary magnetic flux [λ1] and the primary magnetic flux command value [Λ1 *], and obtains the magnetic flux deviation [ΔΛ] (= [λ1δc * −λ1δc λ1γc * −λ1γc] ^t . Then, proportional integral differential processing (PID control) is performed on the magnetic flux deviation [ΔΛ] to obtain the feedback term [B], and the feedback term [B] is added to the calculation result [v * ′], and the result Is output as a voltage command value [v *].

That is, the second calculation unit 103 treats the calculation result [v * ′] as the first term [vδc * _F vγc * _F] ^t of the equation (20), and treats the feedback term [B] as the second term of the equation (20). treated as ^{[vδc * _B vγc * _B]} t, it executes processing for calculating the left side [vδc * vγc ^{*] t} of formula (20). This is a process of correcting the voltage command value obtained in the first embodiment using a feedback term, and the second calculation unit 103 can be grasped as a correction unit that performs the process.

  However, in the present embodiment, the primary magnetic flux [λ1] is treated as an observable value or a value that has already been estimated.

  FIG. 4 is a block diagram illustrating the configuration of the second calculation unit 103. In the figure, + surrounded by a circle indicates an adder, and a circle with + − indicates a subtractor. The subtracter inputs the primary magnetic flux command value [Λ1 *] and the primary magnetic flux [λ1], and outputs a magnetic flux deviation [ΔΛ]. The adder inputs a feedback term [B] as a result of the proportional-integral-derivative process (PID control) and an operation result [v * ′], and outputs a voltage command value [v *].

  By adopting the feedback term in this way, it is possible to improve the followability to the primary magnetic flux command value [Λ1 *] of the primary magnetic flux [λ1] regardless of whether it is in a transient state or a steady state.

Third embodiment.
FIG. 5 is a block diagram showing the configuration of the motor control device 1 according to the present embodiment and its peripheral devices. In the configuration of the motor control device 1 according to the present embodiment, a primary magnetic flux estimation unit 105 is added to the configuration of the motor control device 1 according to the second embodiment.

  The primary magnetic flux estimation unit 105 estimates the primary magnetic flux [λ1] that has been treated as an observable value or an already estimated value in the second embodiment. Hereinafter, the estimated value of the primary magnetic flux [λ1] will be described as the primary magnetic flux estimated value [λ1 ^].

  As described above, the equations (6) and (8) indicate the primary magnetic flux [λ1]. In these equations, the field magnetic flux Λ0, the d-axis inductance Ld, and the q-axis inductance Lq are known quantities, and the δc-axis component (hereinafter referred to as “δc-axis current”) iδc and γc-axis components of the armature current [i]. iγc (hereinafter referred to as “γc-axis current”) is an observable quantity. Therefore, if the phase difference φc is obtained, the primary magnetic flux estimated value [λ1 ^] can be obtained from Equation (6) or Equations (7) and (8).

  However, since the phase difference φc is not an observable amount, it needs to be estimated as well. The phase difference φc has the relationship of equation (21) among the δc-axis voltage vδc, the γc-axis voltage vγc, the δc-axis current iδc, the γc-axis current iγc, the q-axis inductance Lq, and the resistance component R of the armature winding. .

  At this time, the δc-axis voltage vδc and γc-axis voltage vγc used may be obtained by converting the actually measured voltage by the coordinate conversion unit 101. However, FIG. 5 illustrates a case where the voltage command value [v *] that has already been obtained is adopted and used for the estimation of a new phase difference φc. Alternatively, the calculation result [v * ′] of the first calculation unit 102 may be used instead of the voltage command value [v *].

  FIG. 6 is a block diagram illustrating the structure of the primary magnetic flux estimation unit 105. The primary magnetic flux estimation unit 105 includes a delay unit 105a, a load angle estimation unit 105b, an armature reaction estimation unit 105c, a field magnetic flux vector generation unit 105d, and an adder 105e.

  The armature reaction estimation unit 105c receives the phase difference φc, the d-axis inductance Ld, the q-axis inductance Lq, the δc-axis current iδc, and the γc-axis current iγc, and calculates the first term on the right side of Equation (6).

  In FIG. 6, a matrix that is a coefficient of the armature current [i] of the first term on the right side of Expression (6) is indicated as {L}.

  The field magnetic flux vector generation unit 105d receives the field magnetic flux Λ0 and calculates the second term on the right side of Equation (6). In FIG. 6, the second term on the right side of Equation (6) is indicated by a vector [Λ0].

The adder 105e performs addition on the two components of the γc-axis component and the δc-axis component, thereby realizing the addition of the first term and the second term on the right side of the equation (6), and the primary magnetic flux estimated value [ λ1 ^] (= [λ1δc ^ λ1γc ^] ^t ) is output.

  In order to estimate the phase difference φc, the voltage command value [v *] obtained by the second calculation unit 103 at the previous control timing is used. In other words, the voltage command value [v *] obtained by the second calculation unit 103 is delayed by the delay unit 105a, and the phase difference φc according to the equation (21) by the load angle estimation unit 105b at the next control timing. Is calculated.

  Note that the voltage command value [v *] obtained at the present time may be used instead of the voltage command value [v *] obtained at the previous control timing. In this case, the delay unit 105a can be omitted.

  According to the present embodiment, direct detection of the primary magnetic flux [λ1] becomes unnecessary.

  In any of the above-described embodiments, the motor control device 1 includes a microcomputer and a storage device. The microcomputer executes each processing step (in other words, a procedure) described in the program. The storage device is composed of one or more of various storage devices such as a ROM (Read Only Memory), a RAM (Random Access Memory), a rewritable nonvolatile memory (EPROM (Erasable Programmable ROM), etc.), and a hard disk device, for example. Is possible. The storage device stores various information, data, and the like, stores a program executed by the microcomputer, and provides a work area for executing the program.

  It can be understood that the microcomputer functions as various means corresponding to each processing step described in the program, or can realize that various functions corresponding to each processing step are realized. In addition, the motor control device 1 is not limited to this, and various procedures executed by the motor control device 1 or various means or various functions implemented may be realized by hardware.

  For example, in a microcomputer, control is often processed in a discrete time system. Non-Patent Document 5 discloses a technique for controlling a motor in a discrete time system. As a conversion method for the following processing, the PTC (complete tracking control method) introduced in Non-Patent Document 5 may be adopted.

  In the following, a discrete time system process for obtaining the voltage command value [v *] in the second embodiment will be described.

  FIG. 7 is a block diagram illustrating discrete-time processing for portions corresponding to the first calculation unit 102 and the second calculation unit 103.

  A value appended with round brackets at the end of each element of the vector indicates a value that is updated every control cycle Ts. Specifically, the value with (n-1) is a value one minute before the control period Ts than the value with (n), and the value with (n + 1) is ( n) is a value one minute after the control cycle Ts from the value appended.

Processes S101 and S102 are both delay processes, and the primary magnetic flux command value [λ1δc * (n + 1) λ1γc * (n + 1)] ^t is delayed by one and two control periods Ts, respectively. Through these processes, the primary magnetic flux command value [λ1δc * (n) λ1γc * (n)] ^t and the primary magnetic flux command value [λ1δc * (n−1) λ1γc * (n−1)] ^t are obtained.

Process S103 is a subtraction process in which the primary magnetic flux command value [λ1δc * (n) λ1γc * (n)] ^t is subtracted from the primary magnetic flux command value [λ1δc * (n + 1) λ1γc * (n + 1)] ^t , and the control cycle Ts The difference in the primary magnetic flux command value for one minute is obtained. This difference is divided by the control cycle Ts in step S104, and a value corresponding to the differential value of the primary magnetic flux command value is obtained. This corresponds to the third term on the right side of Equation (19).

Process S105 is a matrix operation, and a matrix operation is performed on the primary magnetic flux command value [λ1δc * (n) λ1γc * (n)] ^t to obtain a value corresponding to the second term on the right side of Equation (19). .

The process S110 is a multiplication process, and the armature current [iδc (n−1) iγc (n−1)] ^t is multiplied by the resistance component R to obtain a product corresponding to the first term on the right side of the equation (19).

The process S111 is a subtraction process, and the primary magnetic flux [λ1δc (n−1) λ1γc (n−1)] ^t is subtracted from the primary magnetic flux command value [λ1δc * (n−1) λ1γc * (n−1)] ^t. Get the magnetic flux deviation. The process S109 performs PI control on the magnetic flux deviation and outputs a value corresponding to the feedback term.

Processes S106, S107, and S108 are addition processes, and the processing results of processes S104, S105, S109, and S110 are added. Thereby, a voltage command value [vδc * (n) vγc * (n)] ^t is obtained.

  It is obvious that such a process in a discrete time system can also be applied to the first embodiment. In addition, it is obvious that even the low-pass filter shown in the first embodiment can be processed in a discrete time system.

  As a conversion method for obtaining processing in a discrete time system, bilinear z conversion can be adopted in addition to standard z conversion.

Application to other technologies.
This case is not limited to the case where λ1γc * = 0 is selected as in the normal primary magnetic flux control. As can be seen from the derivation of the above equations and the handling of the primary magnetic flux command value [Λ1 *], it is apparent that the present invention can also be applied when the γc-axis component λ1γc * of the primary magnetic flux command value [λ1] is not zero.

  Non-Patent Document 4 introduces V / f control of a permanent magnet synchronous motor instead of primary magnetic flux control. However, in this document, the voltage command generation formula does not reflect the transient state, and it is considered that the efficiency in the transient state is lowered. Therefore, as described in the basic idea of the invention of the present application, by obtaining a voltage command reflecting a transient state, it can be expected to obtain the same effect as the present application.

  Moreover, although patent document 2 is not primary magnetic flux control, it can be anticipated that the same effect as this application will be acquired by reflecting a transient state in a voltage command generation type | formula.

DESCRIPTION OF SYMBOLS 1 Electric motor control apparatus 101,104 Coordinate conversion part 102 1st calculation part 103 2nd calculation part 105 Primary magnetic flux estimation part 106 Integrator

Claims (5)

A rotary electric motor including an armature having an armature winding through which an armature current ([i]) flows, and a rotor that rotates relative to the armature, a first shaft (δc) and the rotation of the rotor A device for controlling in a rotating coordinate system rotating with a second axis (γc) advanced by 90 degrees relative to the first axis in the direction,
A magnetic flux command value ([Λ1 *]) as a command value in the rotating coordinate system and a differential value of the air gap magnetic flux ([λ1]) as a magnetic flux flowing through the air gap between the armature and the rotor, and the rotational coordinates A calculation for obtaining a first voltage command value ([v *], [v * ′]) as a command value in the rotating coordinate system of the voltage applied to the rotary motor using the angular velocity (ω1) at which the system rotates. Part (102),
A phase calculation unit (106) for obtaining the phase (θ) of the rotating coordinate system by integrating the angular velocity;
First coordinates for converting the first voltage command value by the phase (θ) to obtain a second voltage command value ([V *]) as a command value in another coordinate system of the voltage applied to the rotary motor An electric motor control device comprising a conversion unit (104) and controlling the air gap magnetic flux.   The motor control apparatus according to claim 1, wherein the calculation unit (102) further obtains the first voltage command value using a product of a resistance value of the armature winding and the armature current. The armature current is coordinate-transformed by the phase (θ), and a first current value (iδc) that is a component of the first axis of the armature current and a second component of the armature current that is a component of the second axis. 2nd coordinate conversion part (101) which calculates | requires 2 electric current value (i (gamma) c)
Further comprising
The calculation unit (102)
A first differential value obtained by differentiating a product of the resistance value (R) and the first current value and a first axis magnetic flux command value (λ1δc *) as a component of the first axis of the magnetic flux command value ( sλ1δc *) is subtracted from the product of the second magnetic flux command value (λ1γc *), which is a component of the second axis of the magnetic flux command value, and the angular velocity, and the first axis of the first voltage command value Component (vδc *, vδc * ′)
The product of the resistance value and the second current value (iγc), the second differential value (sλ1γc *) obtained by differentiating the second magnetic flux command value, the first magnetic flux command value and the angular velocity The motor control device according to claim 2, wherein a product (vγc *, vγc * ′) of the second voltage of the first voltage command value is obtained by adding a product. A correction unit that corrects the first voltage command value ([v * ′]) using a result of proportional-integral-derivative control of a deviation of the air gap magnetic flux ([λ1]) from the magnetic flux command value ([Λ1 *]). 103)
The motor control device according to any one of claims 1 to 3, further comprising: A magnetic flux estimator (105) for obtaining an estimated value ([λ1 ^]) of the air gap magnetic flux ([λ1]) and giving it to the correction unit
The motor control device according to claim 4, further comprising:

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