JP6089608B2 - Control method of synchronous motor - Google Patents

Description

  The present invention relates to a system in which a synchronous motor (hereinafter referred to as PM motor) having a permanent magnet as a field source and not having a braking winding is controlled at a variable speed by an inverter or the like, and more particularly, to position sensorless control.

  The arrangement structure of the field magnet in the PM motor includes an SPM structure (SPMSM: Surface Permanent Synchronous Motor) that attaches a magnet to the surface of the field iron core and an IPM structure (IPMSM: Interior Permanent Magnet Magnet) that embeds the magnet inside the iron core. ) And can be broadly divided.

  Synchronous motors and synchronous generators having field windings are often defined as salient pole machines and non-salient pole machines depending on the iron core shape of the field pole. However, in the permanent magnet synchronous machine, since the relative permeability of the permanent magnet is as low as about 1, the SPM structure and the IPM structure having different arrangements of the iron core and the magnet have different magnetic characteristics. In general, an equivalent circuit of a permanent magnet synchronous machine expressed by a two-axis theory is a non-saliency in an SPM structure in which a field axis (N pole) d-axis inductance Ld and a q-axis inductance Lq that is an orthogonal axis thereof are substantially equal. The other IPM structure tends to have reverse salient pole characteristics in which the d-axis has a smaller inductance than the q-axis.

  Of course, if the shape of the field core is devised, it is possible to arbitrarily design the characteristics of non-salient poles and reverse salient poles regardless of the structure of SPM or IPM. Strictly, magnetic characteristics of salient poles and non-salient poles are used.

  Here, in the orthogonal coordinate system with the field pole (N pole, d axis) as a reference, the case where the inductance components of the Ld and Lq axes of the stator windings are substantially equal is “having magnetic non-saliency” It is defined as

  As a method of driving a motor without attaching a position sensor for detecting magnetic pole position information to the motor, a number of methods called “position sensorless control” have been developed.

  This “position sensorless control” uses a speed electromotive force, but when the speed becomes below 10% of the rated speed, the speed electromotive force is reduced by the field linkage flux of the PM motor. Therefore, the influence of a voltage error of a converter such as an inverter becomes relatively large with respect to the speed electromotive force, and there is a problem that the voltage error becomes a large disturbance and an accurate magnetic pole position cannot be estimated.

  As a method for solving the problem in the low speed region of the position sensorless control, a magnetic pole position estimation method in which a high frequency component is superimposed on a current or voltage has been developed for a motor having saliency (for example, Patent Documents 1 and 2).

JP 2003-153582 A JP 7-245981 A

Koji Tanaka and Ichiro Miki, “Position Sensorless Control of Embedded Magnet Synchronous Motor Using Extended Inductive Voltage”, IEEE Trans. IA, Vol 125, no. 9, pp 833-838, 2005

  FIG. 6 is a block diagram showing a position sensorless control method (Patent Document 1) for superimposing high-frequency components.

  This method of superimposing high-frequency components can estimate the salient pole direction of the motor (for example, the d-axis or q-axis direction), but it can determine the NS pole of the magnetic pole (for example, the positive and negative directions of the d-axis) Judgment) is not possible. For this reason, if the NS pole is determined incorrectly, the d-axis estimated phase has an error of 180 ° in electrical angle, and the polarity of the estimated q-axis is also reversed. For this reason, a torque in the direction opposite to the torque command is generated, causing a problem that the acceleration continues in the direction opposite to the direction in which the rotation is desired. Hereinafter, this abnormal phenomenon is referred to as “reverse runaway”.

  As described above, it is an object to provide a method for controlling a synchronous motor that can quickly return to normal operation even when reverse runaway occurs.

  The present invention has been devised in view of the above-described conventional problems, and one aspect thereof is a control method for a synchronous motor that controls a synchronous motor using a permanent magnet as a field source without a position / speed sensor, The first phase error component is estimated based on the high frequency voltage or high frequency current flowing in the synchronous motor by superimposing the high frequency on the voltage or current, and the speed electromotive force generated by the rotation of the synchronous motor and the estimated speed value The second phase error component is estimated based on the code, and speed control and phase calculation are performed based on a composite error component obtained by adding the first phase error component and the second phase error component.

  As one aspect thereof, the second phase error component is calculated by the following equation.

  As another aspect, the first phase error component is a value obtained by extracting a feature value from a high frequency component appearing in the q-axis component, and the second phase error component is a velocity electromotive force of the d-axis component, It is calculated based on the sign of the speed estimation value.

  Further, the composite error component is calculated by multiplying the first and second phase error components by a weight gain corresponding to the speed estimation value.

  According to the present invention, in the synchronous motor control method, even if reverse runaway occurs, it is possible to quickly return to normal operation.

FIG. 3 is a block diagram illustrating a method for controlling the synchronous motor according to the first embodiment. It is a block diagram which shows the control method of the synchronous motor in Embodiment 2. FIG. 10 is a block diagram illustrating a method for controlling a synchronous motor according to a third embodiment. It is a graph which shows an example of a weight gain. It is a time chart which shows the operation | movement of the synchronous motor in the prior art and Embodiment 3. It is a block diagram which shows an example of the control method of the conventional synchronous motor.

  Hereinafter, the control method of the synchronous motor in Embodiments 1-3 of this invention is demonstrated in detail based on drawing.

[Embodiment 1]
FIG. 1 is a block diagram illustrating position sensorless control according to the first embodiment. First, the high-frequency superposition method that is the basis of the present invention will be briefly described (more specifically, refer to Patent Document 1 and many other papers). In addition, this Embodiment 1 shows an example of the high frequency superposition method, The calculation of each part, etc. may be another method.

In the first embodiment, an example is shown in which control is performed on a speed command. The speed controller 11 compares the speed command ω ^* and the estimated speed value ω, and generates current commands id ^* and iq ^* such that the deviation becomes zero. The current control unit 12 compares the current commands id ^* and iq ^* with an average current detection value described later, and generates voltage commands Vd ^* and Vq ^* such that the deviation becomes zero. A high-frequency voltage Vdh, which will be described later, is superimposed on the d-axis voltage command Vd ^* and is output to the reverse rotation coordinate conversion unit 13 together with the q-axis voltage command Vq ^* .

The reverse rotation coordinate converter 13 converts the dq axis voltage commands Vd ^* and Vq ^* into U, V, and W phase voltage commands Vu, Vv, and Vw based on the estimated phase θ. The power converter 2 supplies a current to the PM motor 1 based on the voltage commands Vu, Vv, Vw of the U, V, and W phase components. The current at this time is detected by the current detector 3. The rotating coordinate conversion unit 4 converts the U, V, and W phase current detection values Iu, Iv, and Iw into dq axis current detection values Id and Iq based on the estimated phase θ.

The moving average unit 5 calculates the average current detection value of the current detection values Id and Iq on the dq axis. Here, one cycle of the high-frequency angular velocity ωh of the high-frequency voltage Vdh superimposed on the voltage commands Vd ^* and Vq ^* is defined as an average period. This average current detection value is fed back to the current control unit 12 and used for current control. Further, the high frequency current components Idh and Iqh are extracted by subtracting the current detection value from the current detection values Id and Iq. Since the current detection values Id and Iq include a harmonic component and a steady component, only the harmonic current components Idh and Iqh are extracted by an operation such as subtracting the average current detection value of one period from the band filter. Can do.

The high frequency generator 6 outputs a high frequency voltage Vdh superimposed on the d-axis voltage command Vd ^* . Specifically, the high frequency angular velocity command ωh is integrated by the integrator 6a and applied to the cos unit 6b, and the amplitude is adjusted to | Vh | by the amplitude adjusting unit 6c. The high-frequency voltage Vdh of the d axis whose amplitude is Vh and angular frequency is ωh is added to the voltage command Vd ^* , and the added value is output to the reverse rotation coordinate conversion unit 13. Further, the high frequency generator 6 integrates the high frequency angular velocity command ωh by the integrator 6 a, applies it to the sin unit 6 d, and outputs it to the feature amount calculator 7.

  The feature quantity calculation unit 7 extracts the feature quantities Idh_sin and Iqh_sin by multiplying the high-frequency current components Idh and Iqh by the outputs of the sin unit 6d and performing a moving average.

  The first phase error calculation unit 8 calculates a first phase error component (first phase error estimated value Δθ in the first embodiment) from the feature values Idh_sin and Iqh_sin by the following equation (1).

  The PI controller 9 generates a speed estimated value ω that makes the first phase error estimated value Δθ zero. The estimated speed value ω is output to the speed controller 11 and used for speed control. Further, the estimated speed value ω is converted into an estimated phase θ by integrating in the integrator 10, and this estimated phase θ is used for coordinate conversion of the rotating coordinate converting unit 4 and the reverse rotating coordinate converting unit 13.

The above is the outline of the high-frequency superposition method. The change from the high frequency superposition method in the first embodiment is that the detected current values id, iq and voltage commands Vd ^* , Vq ^* (or the first phase error estimated value Δθ obtained by superposing high frequencies are used. A speed electromotive force e based on the flux linkage of the permanent magnet is calculated from the voltage detection value), and a second phase error component (in the first embodiment, the second phase error estimated value obtained from the information on the speed electromotive force e) is calculated. Δθ ′) is added. A speed estimation value having a function that can be restored even if the runaway is reversed by applying control so that the composite phase error obtained by adding the first phase error estimation value Δθ and the second phase error estimation value Δθ ′ is zero. Get ω.

  Here, a method of estimating the second phase error estimated value Δθ ′ using the speed electromotive force e will be specifically described. In the first embodiment, compared to the conventional control block (FIG. 6), an induced voltage calculation unit 14 that calculates speed electromotive force ed, eq from voltage and current values, speed electromotive force ed, eq and speed estimated value ω The second phase error calculation unit 15 that calculates the second phase error estimated value Δθ ′ using the sign of, and an addition unit that adds the first phase error estimated value Δθ and the second phase error estimated value Δθ ′ 16 are added.

That is, in the induced voltage calculation unit 14, the speed electromotive force is calculated from the voltage commands Vd ^* and Vq ^* output from the current control unit 12 and the average current detection value obtained by removing the high frequency of the moving average unit 5 by the following equation (2). ed and eq are calculated. The calculation of the speed electromotive force e may be back-calculated from the voltage-current equation of the PM motor 1 as in the following equation (2), or there are many methods such as using an extended induced voltage observer (Non-patent Document 1). Here, an example using the voltage-current equation will be described.

  Next, the second phase error calculation unit 15 calculates the second phase error estimated value Δθ ′ by the following equation (3) using the speed electromotive forces ed and eq.

  Here, the sgn () function is a function that outputs 1 if the input is positive, -1 if the input is negative, and 0 if the input is zero.

  The adder 16 adds the second phase error estimated value Δθ ′ obtained by the speed electromotive force ed, eq to the first phase error estimated value Δθ due to the high frequency component, calculates a composite error component, and creates a new The phase error estimated value is output to the PI controller 9. Note that the estimated speed value ω used in the induced voltage calculator 14 and the second phase estimation calculator 15 is approximated by the value of the previous sample time in the PI calculator 9.

  According to the first embodiment, when the PM motor 1 is stopped, the speed electromotive force e becomes almost zero because the term of ω in the above equation (2) becomes zero, but in the high frequency superposition method, p · Ld · id and The component of the speed electromotive force of the high-frequency angular velocity command ωh appears in the term p · Ld · iq. By utilizing this, it is possible to extract the feature amount related to the first phase error estimated value Δθ and estimate the velocity and phase.

  Here, if the estimated phase is shifted from the NS pole by 180 °, when the engine is started as it is, a reverse runaway that accelerates in the reverse direction occurs as described above. Since the electromotive forces ed and eq are generated, the magnetic pole position is corrected using the second phase error estimated value Δθ ′ obtained from the speed electromotive forces ed and eq, so that the normal rotational direction can be restored. it can.

[Embodiment 2]
In the first embodiment, the tan ⁻¹ function which is a nonlinear function is used when calculating the first and second phase error estimated values Δθ and Δθ ′, but the sign of the value used for the tan ⁻¹ function is switched. In particular, if the denominator side changes, the calculation results (first and second phase error estimated values Δθ and Δθ ′) change greatly. If zero is input to the denominator side, an infinite value is output.

  For this reason, if a non-linear function is used in the control block for calculating the speed estimated value ω in this way, a small amount of input disturbance is abnormally amplified, so that the response gain of the PI control unit 9 in the next stage may be increased. It becomes difficult.

Therefore, in the second embodiment, an approximation of directly adding the respective phase error components without using tan ⁻¹ which is a nonlinear function is applied. As a result, the speed / phase estimation response (response gain of the PI control unit 9) can be increased by the amount that the disturbance amplification of the first and second phase error calculation units 8 and 15 can be suppressed.

  Since there is an error component to which the approximation is applied, there is a restriction that the second phase error estimated value Δθ ′ << π. However, in the convergence state of the system, Δθ′≈0 and the response is enhanced. There is no problem even if you apply. Conversely, during reverse runaway, the purpose is to generate a large component that inverts the phase error estimated value by 180 °, so long as there is only a component that can be corrected to 90 ° or more. Therefore, linearity is not necessary and there is no practical problem even if the approximation error is large.

  FIG. 2 is a block diagram illustrating a control apparatus for a synchronous motor according to the second embodiment. Here, only differences from the first embodiment will be described, and the same parts as those of the first embodiment will be denoted by the same reference numerals and the description thereof will be omitted. As shown in FIG. 2, the q-axis feature quantity computing unit 7 'extracts the feature quantity Iqh_sin from the high-frequency component Iqh that appears in the q-axis current detection value Iq. The d-axis feature quantity is omitted. The induced voltage calculation unit 14 'calculates the speed electromotive force ed of the d-axis component. In the second phase error calculation unit 15 ′, the speed electromotive force ed of the d-axis component is multiplied by the sign of the speed estimation value ω and −1 (that is, sgn (ω) · (−ed)), and the addition unit 16 Output to.

  The q-axis feature quantity Iqh_sin computed by the q-axis feature quantity computation unit 7 ′ is used as the first phase error component, and sgn (ω) · (−ed) computed by the second phase error computation unit 15 ′. Are directly added by the adder 16 as a second phase error component and output to the PI control unit 9. Others are the same as in the first embodiment.

As described above, according to the second embodiment, the output of the q-axis feature quantity calculation unit 7 ′ and the second phase error calculation unit 15 ′ are not used without using the non-linear function tan ⁻¹ (). By directly using the output, it is possible to reduce a sudden change due to the disturbance of the estimated phase error, and it is also possible to reduce the calculation amount. In addition, the same effects as those of the first embodiment are obtained.

[Embodiment 3]
The first phase error estimated value Δθ or the first phase error component iqh_sin of the high-frequency superposition method, and the second phase error estimated value Δθ ′ or the second phase error component sgn (ω) · ( -Ed), the gain of one of the phase errors obtained from the high-frequency superposition method and the speed electromotive force is multiplied to give a weight so that the influence of both components can be further adjusted. Become.

  In particular, the induced voltage calculators 14 and 14 'may generate a large value due to the reverse runaway prevention function, and a phase that is likely to occur particularly at an extremely low speed by multiplying the phase error component by a gain that is variable according to the frequency. The influence of the disturbance component of the error can be reduced.

FIG. 3 shows a control block of the synchronous motor in the third embodiment. By multiplying the output of the second phase error calculation unit 15 ′ in the second embodiment by the weight gain K _p by the multiplier 17, the first phase error component Iqh_sin and the speed electromotive force in the high frequency electromotive force method are multiplied. The balance with the phase error component sgn (ω) · (−ed) of 2 is adjusted. In the third embodiment, the first phase error component iqh_sin is fixed.

Figure 4 shows a graph of the weight gain K _p. As shown in FIG. 4, the weight gain K _p is 0.2 when the frequency of the estimated speed value ω is −20% or less and + 20% or more, and decreases as it approaches 0% from −20% and + 20%. It is set to be zero at -2% to + 2%. That is, the weight gain _Kp is set to be zero when the frequency of the speed estimation value ω is near zero.

  Next, the effects of the first to third embodiments will be described using data obtained by simulating the operation using a numerical calculator. FIG. 5 is a comparison chart showing simulation results when the conventional high-frequency superposition method (FIG. 6) and the third embodiment are applied. The PM motor 1 having saliency is shifted by 180 ° in the initial phase of the phase estimation output. It is a condition that it started in the state.

  In FIG. 5, (a) is a conventional high-frequency superposition method, and (b) is a control method in the third embodiment, and the upper stage shows three types of rotational speeds (pu): speed command, detected speed, and speed estimation. The lower part shows the magnetic pole positions of the detected phase and the estimated phase.

In the conventional high-frequency superposition method shown in FIG. 5A, the actual phase and the estimated phase remain 180 ° apart, and the actual speed accelerates in the direction opposite to the speed command, causing reverse runaway. On the other hand, when the third embodiment is applied, the estimated phase is shifted by 180 ° with respect to the actual phase at the start and the speed command and the actual speed are reversed, but there is almost no difference between the actual speed and the estimated speed. The actual speed is -0.1 p. Upon reaching u, it becomes effective weight gain K _p of FIG. 4, the later being able to return to the normal direction of rotation. Also, the estimated phase error can be corrected from 180 ° to around 0 °.

  In addition, when the load torque was changed from 0 to 100% at 0.75 s and the speed command was changed from 10% to -10% at 2.0 s, and the transient stability was confirmed, any condition was confirmed. However, the operation can be continued stably without step-out.

  Although the present invention has been described in detail only for the specific examples described above, it is obvious to those skilled in the art that various changes and modifications are possible within the scope of the technical idea of the present invention. Such variations and modifications are naturally within the scope of the claims.

  For example, in Embodiments 1 to 3, the high frequency voltage is superimposed on the voltage command, but the high frequency current may be superimposed on the current command.

1 ... PM motor (synchronous motor)
Vd ^* , Vq ^* ... voltage command id ^* , iq ^* ... current command idh, iqh ... high frequency current [Delta] [theta] ... first phase error estimated value (first phase error component)
Δθ ′: second phase error estimated value (second phase error component)
ed, eq ... speed electromotive force omega ... velocity estimates Idh_sin, Iqh_sin ... feature amount K _p ... weight gain

Claims (2)

A synchronous motor control method for controlling a synchronous motor using a permanent magnet as a field source without a position / speed sensor,
A first phase error component is estimated based on a high-frequency voltage or high-frequency current flowing in the synchronous motor by superimposing a high frequency on the voltage or current,
Estimating the second phase error component based on the speed electromotive force generated by the rotation of the synchronous motor and the sign of the speed estimation value,
There line speed control and phase calculated based on the composite error component obtained by adding a first phase error component and a second phase error components,
The first phase error component is:
The feature value is extracted from the high-frequency component that appears in the q-axis component.
The second phase error component is:
A method for controlling a synchronous motor, wherein the method is calculated based on a speed electromotive force of a d-axis component and a sign of a speed estimation value. 2. The method of controlling a synchronous motor according to claim 1 , wherein the composite error component is calculated by multiplying the second phase error component by a weight gain corresponding to the speed estimation value.

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