JP6055352B2 - Control device for power converter - Google Patents

Description

  The present invention relates to a control device for a power converter, and more particularly to control of a power converter connected between a plurality of DC power supplies and a common power line.

  Japanese Patent Application Laid-Open No. 2008-104271 (Patent Document 1) discloses that in a power conversion device for driving an AC motor, by switching each phase upper arm element of an inverter, a plurality of DC voltage sources are serially driven and driven in parallel. The structure which can select is described. In the configuration described in Patent Document 1, the DC voltage on the DC side of the inverter can be changed without providing a switch for series / parallel switching.

  JP 2012-70514 A (Patent Document 2) discloses an operation mode (series connection mode) in which DC / DC conversion is performed in a state where two DC power sources are connected in series under the control of a plurality of switching elements. There is described a configuration of a power converter capable of switching an operation mode (parallel connection mode) in which DC / DC conversion is performed in a state where two DC power supplies are used in parallel.

JP 2008-104271 A JP 2012-70514 A

  In the configuration described in Patent Document 1, the DC side voltage of the inverter includes a voltage in which two DC power sources are connected in series (during series driving) and a voltage of one DC power source (in parallel) according to the driving state. (During driving). For this reason, the DC side voltage cannot be controlled to an arbitrary voltage, and the degree of freedom in voltage is low.

  On the other hand, the power converter described in Patent Document 2 has a high degree of voltage freedom because the output voltage of the power line can be controlled according to the voltage command value with the switching of the operation mode.

  However, the power converter described in Patent Document 2 requires different control calculations in each of the serial connection mode and the parallel connection mode. For this reason, there is a possibility that the control accuracy is lowered due to a calculation delay corresponding to the switching of the operation mode.

  Furthermore, in the serial connection mode of the power converter of Patent Document 2, voltage control based on the output voltage and the voltage command value of the entire series DC power supply is executed, so the power distribution ratio between the DC power supplies cannot be controlled. For this reason, if you try to protect each DC power supply from overpower, the overall output power is limited by the performance of the DC power supply that was previously subjected to overpower restrictions, so it is not possible to take full advantage of the capabilities of both power supplies. There is sex. In addition, since the duty ratio in an equivalent step-up chopper circuit configured between a DC power source and a power line connected in series is directly calculated from the voltage deviation of the output voltage, there is a limit to increasing the control speed.

  In parallel connection mode, the output from one DC power supply is current controlled, while the output from the other DC power supply is voltage controlled, so that the voltage controlled DC power supply is sufficiently protected from overpower. Is difficult.

  The present invention has been made to solve such problems, and an object of the present invention is to provide a plurality of operation modes in a power converter connected between a plurality of DC power supplies and a common power line. It is to realize control that improves the controllability of the output voltage to the power line and the protection from overpower of each DC power supply by the control calculation applied in common.

  In one aspect of the present invention, in a control device for a power converter connected between a power line connected to a load and a plurality of DC power supplies, the power converter includes a plurality of switching elements, and It is configured to select and operate one operation mode among a plurality of operation modes having different modes of power conversion between the plurality of DC power supplies and the power lines. The control device includes means for calculating the total input / output power from the entire plurality of DC power sources to the power line based on the deviation between the voltage detection value of the power line and the voltage command value, and a plurality of DCs according to the change of the operation mode. Means for switching the power distribution ratio between the power supplies, means for setting each power command value of the plurality of DC power sources according to the total input / output power and the power distribution ratio, control means, and signal generation means Including. The control means calculates a duty ratio for controlling the output from the DC power supply based on the deviation of the current detection value with respect to the current command value obtained by dividing the power command value by the output voltage for each of the plurality of DC power supplies. . The signal generating means generates an on / off control signal for the plurality of switching elements based on the control pulse signal obtained according to the pulse width modulation by comparing the duty ratio calculated for each of the plurality of DC power supplies and the carrier wave. .

  Preferably, the control device further includes first protection means provided corresponding to a predetermined DC power source among the plurality of DC power sources. The first protection means limits the power command value of the predetermined DC power source set according to the power distribution ratio within a first power range set according to the state of the predetermined DC power source.

  More preferably, the control device further includes a second protection means. The second protection means limits the total input / output power within a second power range set according to the state of the plurality of DC power supplies. The first protection means is provided corresponding to each of the remaining DC power supplies except for one of the plurality of DC power supplies.

  Preferably, the plurality of DC power supplies are constituted by first and second DC power supplies. The plurality of switching elements include first to fourth switching elements. The first switching element is electrically connected between the first node and the power line. The second switching element is electrically connected between the second node and the first node. The third switching element is electrically connected between the third node and the second node that are electrically connected to the negative terminal of the second DC power supply. The fourth switching element is electrically connected between the negative terminal of the first DC power supply and the third node. The power converter further includes first and second reactors. The first reactor is electrically connected between the second node and the positive terminal of the first DC power supply. The second reactor is electrically connected between the first node and the positive terminal of the second DC power supply.

  More preferably, the plurality of operation modes have a first mode and a second mode. In the first mode, the first and second DC power supplies perform DC voltage conversion in parallel with the power line by ON / OFF control of the first to fourth switching elements. In the second mode, the third switching element is fixed on and the first, second, and fourth switching elements are on / off controlled, so that the power line is connected in series with the first and second DC power supplies. DC voltage conversion is performed between In the first mode, the power distribution ratio is variably set according to the states of the first and second DC power supplies. In the second mode, the power distribution ratio is set to a ratio based on the voltages of the first and second DC power supplies. Fixed.

  More preferably, in the first mode, the control means calculates the first and second duty ratios for the first and second DC power supplies by current feedback control based on a current command value. In the second mode, the control means executes current feedback control based on the current command value in one of the first and second DC power supplies, while the other of the first and second DC power supplies The first and second duty ratios are calculated by executing the feedforward control based on the output voltage of the DC power supply and the voltage command value without executing the feedback control. In each of the first and second modes, the signal generation means includes a first control pulse signal obtained by comparing the first duty ratio and the first carrier wave, the second duty ratio and the second carrier. On-off control signals for the first to fourth switching elements are generated based on the second control pulse signal obtained by the wave comparison. Further, in the second mode, the phase difference between the first carrier wave and the second carrier wave is determined by the pulse transition timing of the first control pulse signal and the pulse transition timing of the second control pulse signal. Control is variably performed in accordance with the calculated first and second duty ratios.

  More preferably, in both the first and second modes, the phase difference between the first carrier wave and the second carrier wave is the pulse transition timing of the first control pulse signal and the second control pulse signal. It is variably controlled according to the calculated first and second duty ratios so as to coincide with the transition timing of the first pulse.

  Preferably, the plurality of operation modes further include third to sixth modes. In the third mode, DC voltage conversion is performed between one DC power source of the first and second DC power sources and the power line by on / off control of the first to fourth switching elements, and The state where the other DC power supply of the second DC power supply is electrically disconnected from the power line is maintained. In the fourth mode, the first to fourth switching elements are fixed on and off, and one of the first and second DC power supplies is electrically connected to the power line, while the first and second DC power supplies are connected. The state where the other side of the power source is electrically disconnected from the power line is maintained. In the fifth mode, the first to fourth switching elements are fixed on and off, and the state where the first and second DC power supplies are connected in parallel to the power line is maintained. In the sixth mode, the first to fourth switching elements are fixed on and off, and the state where the first and second DC power supplies are connected in series to the power line is maintained. The power distribution ratio is set so that the entire input / output power is ensured only by the output from one of the DC power supplies in the third mode.

  Alternatively, preferably, the plurality of operation modes include a first mode and a third mode. In the first mode, the first and second DC power supplies perform DC voltage conversion in parallel with the power line by ON / OFF control of the first to fourth switching elements. In the third mode, DC voltage conversion is performed between one DC power source of the first and second DC power sources and the power line by on / off control of the first to fourth switching elements, and the first and second switching devices are also controlled. The state where the other DC power source of the two DC power sources is electrically disconnected from the power line is maintained. In the first mode, the power distribution ratio is variably set according to the states of the first and second DC power supplies, while in the third mode, the total input / output power is determined only by the output from one of the power supplies. It is set to be secured.

  More preferably, in the first mode, the control means calculates the first and second duty ratios for the first and second DC power supplies by current feedback control based on a current command value. In the third mode, the control means calculates one duty ratio of the first and second duty ratios for one DC power supply by current feedback control based on the current command value. In the first mode, the signal generating means compares the first control pulse signal obtained by comparing the first duty ratio and the first carrier wave with the second duty ratio and the second carrier wave. On-off control signals for the first to fourth switching elements are generated based on the obtained second control pulse signal. In the third mode, the signal generating means performs the first to second control pulse signals based on the first or second control pulse signal obtained by comparing the calculated duty ratio with the first or second carrier wave. On-off control signals for the first to fourth switching elements are generated so as to form two pairs of four switching elements that are commonly controlled to be turned on / off.

  Preferably, the control device further includes means for setting a circulating power value for charging / discharging between the first DC power source and the second DC power source. The power command value of the first DC power supply is set so as to be limited within the power range set according to the state of the first DC power supply in accordance with the overall input / output power, the power distribution ratio, and the circulating power value. The The power command value of the second DC power supply is set by subtracting the power command value of the first DC power supply from the total input / output power.

  Alternatively, preferably, the control unit performs current feedback control based on a current command value in one of the first and second DC power supplies in the second mode, while the first and second DC power supplies. On the other hand, current feedback control is not executed, and feedforward control based on the output voltage and voltage command value of the DC power supply is executed, whereby the first and second duties for the first and second DC power supplies, respectively. Calculate the ratio. The signal generation means generates on / off control signals for the first to fourth switching elements according to pulse width modulation based on the first and second duty ratios.

  According to the present invention, in a power converter connected between a plurality of DC power supplies and a common power line, the controllability of the output voltage to the power line can be controlled by the control calculation applied in common between the plurality of operation modes. In addition, it is possible to realize control that improves the protection against overpower of each DC power supply.

It is a circuit diagram which shows the structure of the power supply system containing the power converter according to Embodiment 1 of this invention. It is the schematic which shows the structural example of the load shown in FIG. It is a chart for demonstrating the several operation mode which the power converter shown in FIG. 1 has. It is a conceptual diagram which shows an example of the characteristic of both DC power supplies at the time of comprising the two DC power supplies shown in FIG. 1 with different types of power supplies. It is a circuit diagram explaining the 1st circuit operation in PB mode. It is a circuit diagram explaining the 2nd circuit operation in PB mode. It is a circuit diagram explaining DC / DC conversion (step-up operation) for the first DC power supply in the PB mode. It is a circuit diagram explaining DC / DC conversion (step-up operation) for the second DC power supply in the PB mode. It is a circuit diagram explaining the circuit operation in SB mode. It is a circuit diagram explaining DC / DC conversion (step-up operation) in the SB mode. It is a conceptual diagram explaining the basic concept of power converter control according to this Embodiment 1. FIG. It is a 1st block diagram for demonstrating the power converter control according to this Embodiment 1. FIG. FIG. 11 is a second block diagram for illustrating power converter control according to the first embodiment. It is a conceptual diagram for demonstrating the power flow in a PB mode within the power supply system by the power converter control according to Embodiment 1. FIG. It is a wave form diagram showing an example of control operation of a switching element according to PWM control in PB mode. It is a chart explaining the setting of the control signal and control data in the PB mode. It is a wave form diagram which shows the control operation example of PB mode in the case of using the carrier wave from which a phase differs. It is a wave form diagram explaining the electric current phase by the carrier phase control in PB mode. It is a circuit diagram explaining the electric current path | route in the predetermined period of FIG. FIG. 19 is a current waveform diagram of the switching element at the current phase shown in FIG. 18. It is a wave form diagram which shows the electric current phase when the phase difference between carrier waves = 0. FIG. 22 is a current waveform diagram of the switching element at the current phase shown in FIG. 21. It is a chart for explaining carrier phase control in the PB mode in each operation state of the DC power supply. It is a table explaining the setting of control signals and control data in each operation mode belonging to the boost mode. It is a conceptual diagram for demonstrating the power flow in the power supply system in the aB mode by the power converter control according to Embodiment 2. FIG. It is a conceptual diagram for demonstrating the power flow in the power supply system in bB mode by the power converter control according to Embodiment 2. FIG. It is a conceptual diagram for demonstrating the power flow in the power supply system in SB mode by power converter control according to Embodiment 2. FIG. It is a graph for demonstrating the setting of the control signal in SB mode. It is a figure explaining the state of two DC power supplies in SB mode. It is a wave form diagram which shows the control pulse signal in SB mode when carrier phase control is applied. It is a wave form diagram which shows the operation example of PB mode in the power converter control according to this Embodiment 2, and SB mode. It is a circuit diagram which shows the structural example of the power supply system containing the power converter according to Embodiment 3 of this invention. FIG. 33 is a chart for explaining a plurality of operation modes of the power converter shown in FIG. 32 and setting of control signals and control data in each operation mode.

  Embodiments of the present invention will be described below in detail with reference to the drawings. In the following, the same or corresponding parts in the drawings are denoted by the same reference numerals, and the description thereof will not be repeated in principle.

[Embodiment 1]
(Circuit configuration of power converter)
FIG. 1 is a circuit diagram showing a configuration of a power supply system including a power converter according to the first embodiment of the present invention.

  Referring to FIG. 1, power supply system 5 includes a plurality of DC power supplies 10 a and 10 b, a load 30, and a power converter 50.

  In the present embodiment, each of DC power supplies 10a and 10b is a secondary battery such as a lithium ion secondary battery or a nickel metal hydride battery, or a DC voltage excellent in output characteristics such as an electric double layer capacitor or a lithium ion capacitor. Consists of source elements. The DC power supply 10a and the DC power supply 10b correspond to a “first DC power supply” and a “second DC power supply”, respectively.

  The DC power supplies 10a and 10b can be configured by DC power supplies of the same type and the same capacity, or can be configured by DC power supplies having different characteristics and / or capacities. As described above, the DC power supplies 10a and 10b are concepts including large-capacity capacitors.

  Power converter 50 is connected between DC power supplies 10 a and 10 b and power line 20. Power converter 50 controls a DC voltage (hereinafter also referred to as output voltage VH) on power line 20 connected to load 30 in accordance with voltage command value VH *. That is, the power line 20 is provided in common for the DC power supplies 10a and 10b.

  The load 30 operates in response to the output voltage VH of the power converter 50. Voltage command value VH * is set to a voltage suitable for the operation of load 30. Voltage command value VH * may be variably set according to the state of load 30. Furthermore, the load 30 may be configured to be able to generate charging power for the DC power supplies 10a and 10b by regenerative power generation or the like.

Power converter 50 includes switching elements S1 to S4 and reactors L1 and L2. In the present embodiment, an IGBT (Insulated Gate Bipolar Transistor), a power MOS (Metal Oxide Semiconductor) transistor, a power bipolar transistor, or the like can be used as the switching element. Anti-parallel diodes D1 to D4 are arranged for switching elements S1 to S4. Further, the switching elements S1 to S4 can control on / off in response to the control signals SG1 to SG4, respectively. That is, the switching elements S1 to S4 are turned on when the control signals SG1 to SG4 are at a high level (hereinafter, H level), and are turned off when the control signals SG1 to SG4 are at a low level (hereinafter, L level).

  Switching element S1 is electrically connected between power line 20 and node N1. Reactor L2 is connected between node N1 and the positive terminal of DC power supply 10b. Switching element S2 is electrically connected between nodes N1 and N2. Reactor L1 is connected between node N2 and the positive terminal of DC power supply 10a.

  Switching element S3 is electrically connected between nodes N2 and N3. Node N3 is electrically connected to the negative terminal of DC power supply 10b. Switching element S4 is electrically connected between node N3 and ground line 21. The ground wiring 21 is electrically connected to the load 30 and the negative terminal of the DC power supply 10a.

  As can be understood from FIG. 1, the power converter 50 has a boost chopper circuit corresponding to each of the DC power supply 10a and the DC power supply 10b. That is, for DC power supply 10a, a current bidirectional first step-up chopper circuit having switching elements S1 and S2 as upper arm elements and switching elements S3 and S4 as lower arm elements is configured. Similarly, for the DC power supply 10b, a current bidirectional second step-up chopper circuit is configured with the switching elements S1 and S4 as upper arm elements and the switching elements S2 and S3 as lower arm elements. .

  A power conversion path formed between the DC power supply 10a and the power line 20 by the first boost chopper circuit and a power conversion path formed between the DC power supply 10b and the power line 20 by the second boost chopper circuit. Both include the switching elements S1 to S4.

  The control device 40 generates control signals SG1 to SG4 for controlling on / off of the switching elements S1 to S4 in order to control the output voltage VH to the load 30. Although not shown in FIG. 1, the voltage (hereinafter referred to as Va) and current (hereinafter referred to as Ia) of the DC power supply 10a and the voltage (hereinafter referred to as Vb) of the DC power supply 10b are omitted. And a detector (voltage sensor, current sensor) for the output voltage VH and the current (hereinafter referred to as Ib) and the output voltage VH are provided. Furthermore, it is also preferable to arrange detectors (temperature sensors) for the temperatures of the DC power supplies 10a and 10b (hereinafter referred to as Ta and Tb). The outputs of these detectors are provided to the controller 40.

  In the configuration of FIG. 1, switching elements S1 to S4 correspond to “first switching element” to “fourth switching element”, respectively, and reactors L1 and L2 correspond to “first reactor” and “second reactor”, respectively. Corresponds to "reactor" respectively.

FIG. 2 is a schematic diagram illustrating a configuration example of the load 30.
Referring to FIG. 2, load 30 is configured to include, for example, a traveling motor for an electric vehicle. Load 30 includes a smoothing capacitor CH, an inverter 32, a motor generator 35, a power transmission gear 36, and drive wheels 37.

  The motor generator 35 is a traveling motor for generating vehicle driving force, and is constituted by, for example, a multi-phase permanent magnet type synchronous motor. The output torque of the motor generator 35 is transmitted to the drive wheels 37 via a power transmission gear 36 constituted by a speed reducer and a power split mechanism. The electric vehicle travels with the torque transmitted to the drive wheels 37. Further, the motor generator 35 generates power by the rotational force of the drive wheels 37 during regenerative braking of the electric vehicle. This generated power is AC / DC converted by the inverter 32. This DC power can be used as charging power for DC power supplies 10 a and 10 b included in the power supply system 5.

  In a hybrid vehicle in which an engine (not shown) is mounted in addition to the motor generator, vehicle driving force required for the electric vehicle is generated by cooperatively operating the engine and the motor generator 35. At this time, it is also possible to charge the DC power supplies 10a and 10b using the power generated by the rotation of the engine.

As described above, the electric vehicle comprehensively represents a vehicle equipped with the electric motor for traveling, and includes a hybrid vehicle that generates vehicle driving force by the engine and the electric motor, and an electric vehicle and a fuel cell vehicle not equipped with the engine. It includes both.

(Power converter operation mode)
The power converter 50 has a plurality of operation modes in which DC power conversion modes between the DC power supplies 10a and 10b and the power line 20 are different.

FIG. 3 shows a plurality of operation modes that the power converter 50 has.
Referring to FIG. 3, the operation mode includes “boost mode (B)” in which the output voltage of DC power supplies 10a and / or 10b is boosted in accordance with periodic on / off control of switching elements S1 to S4, and switching element S1. Are roughly divided into “direct connection mode (D)” in which the DC power supplies 10 a and / or 10 b are electrically connected to the power line 20 with the on / off state of S 4 being fixed.

  In the boost mode, a “parallel boost mode (hereinafter referred to as PB mode)” in which DC / DC conversion is performed in parallel between DC power supplies 10 a and 10 b and power line 20, DC power supplies 10 a and 10 b and power line 20 connected in series are performed. And “series boost mode (hereinafter referred to as SB mode)” for performing DC / DC conversion with the. The PB mode corresponds to the “parallel connection mode” shown in Patent Document 2, and the SB mode corresponds to the “series connection mode” shown in Patent Document 2.

  Further, in the boost mode, a “single mode by DC power source 10a (hereinafter referred to as aB mode)” for performing DC / DC conversion with the power line 20 using only the DC power source 10a, and a power line using only the DC power source 10b. “Single mode by DC power supply 10b (hereinafter referred to as bB mode)” that performs DC / DC conversion with 20 is included. In the aB mode, as long as the output voltage VH is controlled to be higher than the voltage Vb of the DC power supply 10b, the DC power supply 10b is maintained in a state of being electrically disconnected from the power line 20 and is not used. Similarly, in the bB mode, as long as the output voltage VH is controlled to be higher than the voltage Va of the DC power supply 10a, the DC power supply 10a is maintained in a state of being electrically disconnected from the power line 20 and is not used. The

  In each of the PB mode, SB mode, aB mode, and bB mode included in the boost mode, output voltage VH of power line 20 is controlled in accordance with voltage command value VH *. Control of the switching elements S1 to S4 in each of these modes will be described later. As described above, by making voltage command value VH * a variable voltage suitable for the operation of load 30, power loss at load 30 can be suppressed in each mode included in the boost mode.

  In the direct connection mode, the “parallel direct connection mode (hereinafter referred to as PD mode)” in which the DC power supplies 10 a and 10 b are connected in parallel to the power line 20 and the DC power supplies 10 a and 10 b in series with the power line 20 are connected. “Series direct connection mode (hereinafter referred to as SD mode)” for maintaining the connected state is included.

  In the PD mode, the switching elements S1, S2, and S4 are fixed on, while the switching element S3 is fixed off. As a result, the output voltage VH becomes equal to the output voltages Va and Vb (strictly, the higher voltage of Va and Vb) of the DC power supplies 10a and 10. Since the voltage difference between Va and Vb causes a short circuit current in the DC power supplies 10a and 10b, the PD mode can be applied only when the voltage difference is small.

  In the SD mode, the switching elements S2 and S4 are fixed off, while the switching elements S1 and S3 are fixed on. As a result, the output voltage VH is equivalent to the sum of the output voltages Va and Vb of the DC power supplies 10a and 10 (VH = Va + Vb).

  Furthermore, in the direct connection mode, “direct connection mode of DC power supply 10a (hereinafter referred to as aD mode)” in which only DC power supply 10a is electrically connected to power line 20 and only DC power supply 10b is electrically connected to power line 20 “ "Direct connection mode (hereinafter referred to as bD mode) of DC power supply 10b".

  In the aD mode, the switching elements S1 and S2 are fixed on, while the switching elements S3 and S4 are fixed off. As a result, the DC power supply 10b is disconnected from the power line 20, and the output voltage VH is equal to the voltage Va of the DC power supply 10a (VH = Va). In the aD mode, the DC power supply 10b is not used because it is kept electrically disconnected from the power line 20. When the aD mode is applied in a state where Vb> Va, a short-circuit current is generated from the DC power supplies 10b to 10a via the switching element S2. For this reason, Va> Vb is a necessary condition for applying the aD mode.

  Similarly, in the bD mode, the switching elements S1 and S4 are fixed on, while the switching elements S2 and S3 are fixed off. As a result, the DC power supply 10a is disconnected from the power line 20, and the output voltage VH is equal to the voltage Vb of the DC power supply 10b (VH = Vb). In the bD mode, the DC power supply 10a is not used because it is kept disconnected from the power line 20. When the bD mode is applied in a state where Va> Vb, a short-circuit current is generated from the DC power supplies 10a to 10b via the diode D2. For this reason, Vb> Va is a necessary condition for applying the bD mode.

  In each of the PD mode, SD mode, aD mode, and bD mode included in the direct connection mode, the output voltage VH of the power line 20 is determined depending on the voltages Va and Vb of the DC power supplies 10a and 10b, and therefore must be directly controlled. Can not be. For this reason, in each mode included in the direct connection mode, the output voltage VH cannot be set to a voltage suitable for the operation of the load 30, which may increase the power loss in the load 30.

  On the other hand, in the direct connection mode, since the switching elements S1 to S4 are not turned on / off, the power loss of the power converter 50 is significantly suppressed. Therefore, depending on the operating state of the load 30, the application of the direct connection mode increases the power loss reduction amount in the power converter 50 more than the power loss increase amount of the load 30. There is a possibility that it can be suppressed.

  In FIG. 3, the PB mode corresponds to a “first mode”, the SB mode corresponds to a “second mode”, and the aB mode and the bB mode correspond to a “third mode”. The aD mode and the bD mode correspond to the “fourth mode”, the PD mode corresponds to the “fifth mode”, and the SD mode corresponds to the “sixth mode”.

  FIG. 4 is a conceptual diagram showing an example of characteristics of both DC power supplies when the DC power supplies 10a and 10b are configured with different types of power supplies. FIG. 4 shows a so-called Ragon plot in which energy is plotted on the horizontal axis and power is plotted on the vertical axis. In general, since the output power and stored energy of a DC power supply are in a trade-off relationship, it is difficult to obtain a high output with a high-capacity battery, and it is difficult to increase the stored energy with a high-power battery.

  Accordingly, one of the DC power supplies 10a and 10b is constituted by a so-called high-capacity type power supply with high stored energy, while the other is constituted by a so-called high-output type power supply with high output power. It is preferable. In this way, the energy stored in the high-capacity power supply is used for a long period of time, while the high-power power supply is used as a buffer to output the shortage due to the high-capacity power supply. Can do.

  In the example of FIG. 4, the DC power supply 10a is configured with a high-capacity power supply, while the DC power supply 10b is configured with a high-output power supply. Therefore, the operating range 110 of the DC power supply 10a has a narrower power output range than the operating range 120 of the DC power supply 10b. On the other hand, the energy range that can be stored in the operation region 120 is narrower than that in the operation region 110.

At the operating point 101 of the load 30, high power is required for a short time. For example, in an electric vehicle, the operating point 101 corresponds to a sudden acceleration due to a user's accelerator operation. On the contrary,
At the operating point 102 of the load 30, a relatively low power is required for a long time. For example, in an electric vehicle, the operating point 102 corresponds to continuous high speed steady running.

  The operating point 101 can be dealt with mainly by the output from the high-power DC power supply 10b. On the other hand, the operating point 102 can be dealt with mainly by the output from the high-capacity DC power supply 10a. As a result, in the electric vehicle, by using the energy stored in the high-capacity battery for a long time, the travel distance by the electric energy can be extended, and the acceleration performance corresponding to the accelerator operation of the user can be achieved. It can be secured promptly.

  In addition, when the DC power source is constituted by a battery, there is a possibility that output characteristics may be reduced at a low temperature, and charging / discharging may be limited to suppress the progress of deterioration at a high temperature. In particular, in an electric vehicle, a temperature difference may occur between the DC power supplies 10a and 10b due to a difference in mounting position. Therefore, in the power supply system 5, it is more efficient to use only one of the DC power supplies according to the state (particularly the temperature) of the DC power supplies 10a and 10b or according to the request of the load 30 as described above. There are cases that are appropriate. By providing a mode (aB mode, bB mode, aD mode, bD mode) that uses only one of the DC power supplies 10a, 10b as described above, these cases can be handled.

  That is, in power converter 50 according to the first embodiment, one of the plurality of operation modes shown in FIG. 3 is selected according to the state of DC power supplies 10a and 10b and / or load 30. Is done. The output voltage VH is controlled by operating the power converter 50 in the selected operation mode.

(Circuit operation in each operation mode)
Next, the circuit operation of the power converter 50 in each operation mode will be described. First, the circuit operation in the PB mode in which DC / DC conversion is performed in parallel between DC power supplies 10a and 10b and power line 20 will be described with reference to FIGS.

  As shown in FIGS. 5 and 6, DC power supplies 10 a and 10 b can be connected in parallel to power line 20 by turning on switching element S <b> 4 or S <b> 2. Here, in the parallel connection mode, the equivalent circuit differs depending on the level of the voltage Va of the DC power supply 10a and the voltage Vb of the DC power supply 10b.

  As shown in FIG. 5A, when Vb> Va, the DC power supplies 10a and 10b are connected in parallel via the switching elements S2 and S3 by turning on the switching element S4. An equivalent circuit at this time is shown in FIG.

  Referring to FIG. 5B, between the DC power supply 10a and the power line 20, the ON period and the OFF period of the lower arm element can be alternately formed by the ON / OFF control of the switching element S3. Similarly, between the DC power supply 10b and the power line 20, the ON and OFF periods of the lower arm element of the step-up chopper circuit can be alternately formed by controlling the switching elements S2 and S3 in common. The switching element S1 operates as a switch that controls regeneration from the load 30.

  On the other hand, as shown in FIG. 6A, when Va> Vb, the DC power supplies 10a and 10b are connected in parallel via the switching elements S3 and S4 by turning on the switching element S2. An equivalent circuit at this time is shown in FIG.

  Referring to FIG. 6B, between the DC power supply 10b and the power line 20, the ON period and the OFF period of the lower arm element can be alternately formed by the ON / OFF control of the switching element S3. Similarly, between the DC power supply 10a and the power line 20, the switching elements S3 and S4 are commonly controlled to be turned on / off, whereby the on period and the off period of the lower arm element of the boost chopper circuit can be alternately formed. The switching element S1 operates as a switch that controls regeneration from the load 30.

  Next, the step-up operation in the PB mode of power converter 50 will be described in detail with reference to FIGS.

  FIG. 7 shows DC / DC conversion (step-up operation) for the DC power supply 10a in the PB mode.

Referring to FIG. 7A, by turning on a pair of switching elements S3 and S4 and turning off a pair of switching elements S1 and S2, a current path 150 for storing energy in reactor L1 is formed. . Thereby, a state is formed in which the lower arm element of the boost chopper circuit is turned on.

  On the other hand, referring to FIG. 7B, by turning off the pair of switching elements S3 and S4 and turning on the pair of switching elements S1 and S2, the accumulated energy of reactor L1 is supplied to DC power supply 10a. A current path 151 is formed for output with energy. As a result, a state in which the upper arm element of the boost chopper circuit is turned on is formed.

  While the pair of switching elements S3 and S4 is turned on, the first period in which at least one of the switching elements S1 and S2 is turned off and the pair of switching elements S1 and S2 are turned on, while the switching element S3 , S4 and the second period in which at least one of them is turned off alternately, the current path 150 in FIG. 7A and the current path 151 in FIG. 7B are alternately formed.

  As a result, a boost chopper circuit having a pair of switching elements S1 and S2 equivalently as an upper arm element and a pair of switching elements S3 and S4 equivalently as a lower arm element is configured for the DC power supply 10a. In the DC / DC conversion operation shown in FIG. 7, since there is no current flow path to the DC power supply 10b, the DC power supplies 10a and 10b are non-interfering with each other. That is, it is possible to independently control power input / output to / from DC power supplies 10a and 10b.

  In such DC / DC conversion, the relationship expressed by the following equation (1) is established between the voltage Va of the DC power supply 10a and the output voltage VH of the power line 20. In the expression (1), the duty ratio during a period when the pair of switching elements S3 and S4 is turned on is Da.

VH = 1 / (1-Da) · Va (1)
FIG. 8 shows DC / DC conversion (step-up operation) for the DC power supply 10b in the PB mode.

  Referring to FIG. 8A, by turning on the pair of switching elements S2 and S3 and turning off the pair of switching elements S1 and S4, a current path 160 for storing energy in reactor L2 is formed. . Thereby, a state is formed in which the lower arm element of the boost chopper circuit is turned on.

  On the other hand, referring to FIG. 8 (b), by turning off the pair of switching elements S2 and S3 and turning on the pair of switching elements S1 and S4, the stored energy of reactor L2 is reduced to that of DC power supply 10b. A current path 161 is formed for output with energy. As a result, a state in which the upper arm element of the boost chopper circuit is turned on is formed.

  While the pair of switching elements S2 and S3 is turned on, the first period when at least one of the switching elements S1 and S4 is turned off, and the pair of switching elements S1 and S4 is turned on, while the switching element S2 , S3 are alternately turned off and the second period in which the current path 160 in FIG. 8A and the current path 161 in FIG. 8B are alternately formed.

  As a result, a step-up chopper circuit having a pair of switching elements S1 and S4 equivalently as an upper arm element and a pair of switching elements S2 and S3 equivalently as a lower arm element is configured for DC power supply 10b. In the DC / DC conversion operation shown in FIG. 8, since there is no current path including DC power supply 10a, DC power supplies 10a and 10b are non-interfering with each other. That is, it is possible to independently control power input / output to / from DC power supplies 10a and 10b.

  In such DC / DC conversion, the relationship expressed by the following equation (2) is established between the voltage Vb of the DC power supply 10b and the output voltage VH of the power line 20. In the equation (2), Db is a duty ratio during a period in which the pair of switching elements S2 and S3 is turned on.

VH = 1 / (1-Db) · Vb (2)
The circuit operation in the boost mode (aB mode, bB mode) using only one of the DC power supplies 10a, 10b is common to the circuit operations in FIGS. In the aB mode, the DC power supply 10b is not used by the switching operation shown in FIGS. 7A and 7B, while bidirectional DC / DC conversion is executed between the DC power supply 10a and the load 30. be able to.

  Similarly, in the bB mode, the DC power supply 10a is not used by the switching operation shown in FIGS. 8A and 8B, while bidirectional DC / DC conversion is performed between the DC power supply 10b and the load 30. Can be executed.

  Next, circuit operation in the SB mode in which DC / DC conversion is performed between DC power supplies 10a and 10b connected in series and power line 20 will be described with reference to FIGS. 9 and 10. FIG.

  As shown in FIG. 9A, the DC power supplies 10a and 10b can be connected in series to the power line 20 by fixing the switching element S3 to be on. An equivalent circuit at this time is shown in FIG.

  Referring to FIG. 9B, in the SB mode, the switching elements S2 and S4 are commonly controlled on and off between the DC power supplies 10a and 10b connected in series and the power line 20, thereby lowering the boost chopper circuit. The on period and the off period of the arm element can be alternately formed. The switching element S1 operates as a switch that controls regeneration from the load 30 by being turned on during the off period of the switching elements S2 and S4. In addition, the wiring 15 that connects the reactor L1 to the switching element S4 is equivalently formed by the switching element S3 that is fixed on.

Next, DC / DC conversion (step-up operation) in the SB mode will be described with reference to FIG.
Referring to FIG. 10 (a), switching element S3 is fixed on to connect DC power supplies 10a and 10b in series, while a pair of switching elements S2 and S4 is turned on and switching element S1 is turned off. . Thereby, current paths 170 and 171 for storing energy in reactors L1 and L2 are formed. As a result, a state in which the lower arm element of the boost chopper circuit is turned on is formed for the DC power supplies 10a and 10b connected in series.

  On the other hand, referring to FIG. 10B, the pair of switching elements S2 and S4 is turned off, and switching element S1 is turned on, contrary to FIG. Turned on. Thereby, a current path 172 is formed. The sum of the energy from DC power supplies 10a and 10b connected in series and the energy accumulated in reactors L1 and L2 is output to power line 20 through current path 172. As a result, a state in which the upper arm element of the boost chopper circuit is turned on is formed for the DC power supplies 10a and 10b connected in series.

A first period in which the pair of switching elements S2 and S4 is turned on while the switching element S1 is turned off while the switching element S3 is turned on, and the switching element S1 is turned on and the switching element S2 is turned on , S4 are alternately turned off and the second period is alternately repeated, whereby the current paths 170 and 171 in FIG. 10A and the current path 172 in FIG. 10B are alternately formed.

  In the DC / DC conversion in the SB mode, the relationship expressed by the following expression (3) is established among the voltage Va of the DC power supply 10a, the voltage Vb of the DC power supply 10b, and the output voltage VH of the power line 20. In the expression (3), the duty ratio in the first period when the pair of switching elements S2 and S4 is turned on is Dc.

Vo = 1 / (1-Dc) · (Va + Vb) (3)
However, when Va and Vb are different, or when the inductances of reactors L1 and L2 are different, the current values of reactors L1 and L2 at the end of the operation in FIG. Therefore, immediately after the transition to the operation of FIG. 10B, when the current of reactor L1 is larger, a difference current flows through current path 173. On the other hand, when the current of reactor L2 is larger, a difference current flows through current path 174.

  In the direct connection mode, it is understood that any of the PD mode, the SD mode, the aD mode, and the bD mode is realized by fixing the on / off states of the switching elements S1 to S4 according to FIG.

(Control operation of power converter)
Next, control of power converter 50 according to the first embodiment will be described. As will be apparent from the following description, one of the features of power converter control according to the present embodiment is that a common control operation is applied in each operation mode.

FIG. 11 is a diagram illustrating a basic concept of power converter control according to the first embodiment.
Referring to FIG. 11, the sum of output power Pa of DC power supply 10a and output power Pb of DC power supply 10b is defined as total power PH. In the following description, the power value during discharge of each DC power supply 10a, 10b and during powering operation of load 30 is expressed as a positive value, and the power value during charging of each DC power supply 10a, 10b and during regeneration operation of load 30 is expressed as a positive value. It shall be expressed as a negative value.

  The output voltage VH increases in a state where the total power PH is larger than the output power PL to the load 30 (PH> PL), but decreases in a state where PH <PL. Therefore, in the power converter control according to the first embodiment, the command value of total power PH is set according to voltage deviation ΔVH with respect to voltage command value VH * of output voltage VH. Further, by distributing the total power PH between the output powers Pa and Pb, the outputs of the DC power supplies 10a and 10b are subjected to power control (current control).

(Basic control operation in PB mode)
In the first embodiment, a description will be given of the control operation in the basic PB mode.

  12 and 13 are block diagrams for illustrating power converter control according to the first embodiment. FIG. 12 shows a configuration for a control calculation for setting the power command value of each DC power supply, and FIG. 13 shows a control calculation for controlling the output of each DC power supply according to the set power command value. The configuration of is shown.

  The functions of the functional blocks shown in FIGS. 12 and 13 are realized by software processing by a predetermined program executed by a control unit (ECU) 40 or hardware processing by a dedicated electronic circuit or the like.

  Referring to FIG. 12, control device 40 includes a voltage control unit 200 and a power management unit 290.

  The power management unit 290 determines the power upper limit value PHmax and the power lower limit value PHmin related to the total power PH, the power upper limit value Pamax and the power lower limit value Pamin of the DC power supply 10a based on the state of the DC power supplies 10a and 10b and / or the load 30. And a power distribution ratio k between the DC power supplies 10a and 10b.

  Each power upper limit value indicates the upper limit value of the discharge power, and is set to 0 or positive. When the power upper limit value is set to 0, it means that discharging from the DC power supply is prohibited.

  Similarly, each power lower limit value indicates the upper limit value of the charging power, and is set to 0 or negative. When the power lower limit value is set to 0, it means that charging of the DC power supply is prohibited.

  For example, power upper limit value Pamax and power lower limit value Pamin are set based on SOC (State of Charge) and / or temperature Ta of DC power supply 10a. Similarly, power upper limit value Pbmax and power lower limit value Pbmin can be set for DC power supply 10b as well, so that power upper limit value PHmax and power lower limit value PHmin for DC power supplies 10a and 10b as a whole can be set. For example, PHmax = Pamax + Pbmax and PHmin = Pamin + Pbmin. Also, the load power PL needs to be limited within the range of PHmax to PHmin. That is, the operation command value for the load 30 to perform the regenerative operation or the power running operation is generated by limiting the load power PL within a range where PHmin ≦ PL ≦ PHmax. For example, in the configuration example of FIG. 2, the load power PL by the motor generator 35 is determined by the product of the torque and the rotational speed, so that the torque command value is limited as necessary.

The power distribution ratio k can be determined based on, for example, the SOC balance between the DC power supplies 10a and 10b, the balance of the upper and lower limit power, or the like. The power distribution ratio k can be set to an arbitrary value of 0 ≦ k ≦ 1.0. As will be described in detail later, power distribution ratio k is switched according to the operation mode. Power management unit 290 further sets circulating power Pr for charging / discharging between DC power supplies 10a and 10b.

  The circulating power value Pr corresponds to the output power from the DC power supply 10a for charging the DC power supply 10b. For example, during powering operation, if k = 1 and Pr> 0 is set, the DC power supply 10b can be charged while the total power PH is supplied to the power line 20 by the output power of the DC power supply 10a. . On the other hand, when k = 0 and Pr <0 is set, the DC power supply 10a can be charged while the total power PH is supplied to the power line 20 by the output power of the DC power supply 10b.

  Further, during regenerative operation (PH <0), if k = 0 and Pr> 0 is set, the DC power supply 10b is generated by both the regenerative power from the load 30 and the output power from the DC power supply 10a. Can be charged. Conversely, if k = 1 and Pr <0 is set, the DC power supply 10a can be charged by both the regenerative power from the load 30 and the output power from the DC power supply 10b.

  When circulating power value Pr is not set (Pr = 0), charging / discharging between DC power supplies 10a and 10b is not executed. For example, when the SOCs of the DC power supplies 10a and 10b are unbalanced, the power management unit 290 sets the circulating power value Pr so as to promote the charging of the DC power supply on the low SOC side.

  Voltage control unit 200 sets power command values Pa * and Pb * of DC power supplies 10a and 10b based on the voltage deviation of output voltage VH. The voltage control unit 200 includes a deviation calculation unit 210, a control calculation unit 220, a limiter 230, a power distribution unit 240, a circulating power addition unit 250, a limiter 260, and a subtraction unit 270. In the configuration of FIG. 12, the limiter 260 corresponds to “first protection means”, and the limiter 230 corresponds to “second protection means”.

  Deviation calculation unit 210 calculates voltage deviation ΔVH (ΔVH = VH * −VH) in accordance with the difference between detected value of voltage command value VH * and output voltage VH. The control calculation unit 220 calculates the total power PHr required for voltage control based on the voltage deviation ΔVH. For example, the control calculation unit 220 sets PHr according to the following formula (4) by PI calculation.

PHr = Kp · ΔVH + Σ (Ki · ΔVH) (4)
In Expression (4), Kp is a proportional control gain, and Ki is an integral control gain. These control gains also reflect the capacitance value of the smoothing capacitor CH. By setting the total power PHr according to Equation (4), feedback control for reducing the voltage deviation ΔVH can be realized.

  Alternatively, when the output power PL to the load 30 can be predicted from the operating state of the load 30, the total power PHr required according to the equation (5) can be set by further reflecting the predicted value PL *. It is. In this way, the output voltage VH can be controlled in a form that feeds forward power consumption at the load 30.

PHr = Kp · ΔVH + Σ (Ki · ΔVH) + PL * (5)
Limiter 230 limits power command value PH * so as to be within the range of PHmax to PHmin set by power management unit 290. If PHr> PHmax, the limiter 230 sets PH * = PHmax. Similarly, when PHr <PHmim, limiter 230 sets PH * = PHmin. When PHmax ≧ PHr ≧ PHmin, PH * = PHr is set as it is. As a result, the total power command value PH * is determined.

  The power distribution unit 240 calculates the output power k · PH * to be shared by the DC power supply 10a based on the total power command value PH * and the power distribution ratio k. The circulating power adding unit 250 adds the power Par required by the DC power supply 10a by adding the k · Pa * calculated by the power distributing unit 240 and the circulating power value Pr set by the power managing unit 290. Calculate (Par = k · Pa * + Pr).

  Limiter 260 limits power command value Pa * of DC power supply 10a so as to be within the range of Pamax to Pamin set by power management unit 290. If Par> Pamax, the limiter 260 corrects Pa * = Pamax. Similarly, when PHa <Pamim, the limiter 260 corrects Pa * = Pamin. When Pamax> Par> Pamin, Pa * = Par as it is. Thereby, the power command value Pa * of the DC power supply 10a is determined.

  The subtraction unit 270 sets the power command value Pb * of the DC power supply 10b by subtracting the power command value Pa * from the total power command value PH * (Pb * = PH * −Pa *).

  FIG. 14 is a conceptual diagram for explaining the power flow in the power supply system according to the power command value set according to FIG.

  Referring to FIG. 14, total power command value PH * necessary for controlling output voltage VH to voltage command value VH * is distributed to power command values Pa * and Pb * according to power distribution ratio k. That is, basically, Pa * = k · PH and Pb * = (1−k) · PH * are set. As a result, the power according to the total power command value PH * for controlling the output voltage VH can be input / output to / from the power line 20 while controlling the power ratio between the DC power supplies 10a and 10b.

  Further, by setting the circulating power value Pr, the DC power supply 10b is charged by the output power from the DC power supply 10a (Pr> 0), or the DC power supply 10a is charged by the output power from the DC power supply 10b (Pr <0). )can do.

  Further, since power command value Pa * is reliably limited to a range between Pamax and Pamin by limiter 260, DC power supply 10a can be protected from overpower. That is, overcharge and overdischarge of the DC power supply 10a can be prevented.

  Further, the load power PL is limited within the range of PHmin to PHmax, and the total power command value PH * is reliably limited within the range of PHmax to PHmin by the limiter 230, so that the DC power supply 10b is also protected from overpower. Can protect. That is, overcharge and overdischarge of the DC power supply 10b can be prevented.

  Referring to FIG. 13, control device 40 controls current output from DC power supplies 10a, 10b according to power command values Pa *, Pb *, current control unit 300, current control unit 310, PWM (Pulse Width Modulation). ) Control unit 400 and carrier wave generation unit 410. The current control unit 300 controls the output of the DC power supply 10a by current control. The current control unit 310 controls the output of the DC power supply 10a by current control. In the configuration of FIG. 13, the current control units 300 and 310 correspond to “control means”, and the PWM control unit 400 corresponds to “signal generation means”.

  The current control unit 300 includes a current command generation unit 302, a deviation calculation unit 304, a control calculation unit 306, and an FF addition unit 308.

  The current command generator 302 sets the current command value Ia * of the DC power supply 10a based on the power command value Pa * and the detected value of the voltage Va (Ia * = Pa * / Va). Deviation calculation unit 304 calculates current deviation ΔIa (ΔIa = Ia * −Ia) according to the difference between current command value Ia * and the detected value of current Ia. The control calculation unit 306 calculates a control amount Dfba for current feedback control based on the current deviation ΔIa. For example, the control calculation unit 306 calculates the control amount Dfba according to the following formula (6) by PI calculation.

Dfba = Kp · ΔIa + Σ (Ki · ΔIa) (6)
In equation (6), Kp is a proportional control gain, and Ki is an integral control gain. These control gains are set separately from the equation (4).

  On the other hand, the FF control amount Dffa of the voltage feedforward control is set according to Equation (7) along Da = (VH−Va) / VH obtained by solving Equation (1) for Da (FIG. 16). reference).

Dffa = (VH * −Va) / VH * (7)
The FF adder 308 calculates the duty ratio Da related to the output control of the DC power supply 10a by adding the FB control amount Dfba and the FF control amount Dffa. The duty ratio Da is the lower arm element (switching element) of the step-up chopper circuit (FIG. 7) when DC / DC conversion is performed between the voltage Va of the DC power supply 10a and the output voltage VH, as in the expression (1). This corresponds to the duty ratio during a period in which S3, S4) are turned on.

  Similarly, the current control unit 310 corresponding to the DC power supply 10b includes a current command generation unit 312, a deviation calculation unit 314, a control calculation unit 316, and an FF addition unit 318.

  The current command generator 312 sets the current command value Ib * of the DC power supply 10b based on the power command value Pb * and the detected value of the voltage Vb (Ib * = Pb * / Vb). Deviation calculation unit 314 calculates current deviation ΔIb (ΔIb = Ib * −Ib) according to the difference between current command value Ib * and the detected value of current Ib. The control calculation unit 316 calculates a control amount Dfbb for current feedback control based on the current deviation ΔIb. For example, the control calculation unit 316 calculates the control amount Dfbb according to the following equation (8) by PI calculation.

Dfbb = Kp · ΔIb + Σ (Ki · ΔIb) (8)
In Expression (8), Kp is a proportional control gain, and Ki is an integral control gain. These control gains are set separately from the equations (4) and (6).

  On the other hand, the FF control amount Dffb of the voltage feedforward control is set according to the equation (9) along Db = (VH−Vb) / VH obtained by solving the equation (2) for Db (FIG. 16). reference). In equation (9), voltage command value VH * may be a detected value of output voltage VH.

Dffb = (VH * −Vb) / VH * (9)
The FF adder 318 calculates the duty ratio Db related to the output control of the DC power supply 10b by adding the FB control amount Dfbb and the FF control amount Dffb. The duty ratio Db corresponds to the duty ratio during the period when the lower arm elements (switching elements S2 and S3) of the boost chopper circuit (FIG. 8) are turned on, as in Expression (2).

  The PWM control unit 400 controls the switching elements S1 to S4 by pulse width modulation control based on the duty ratios Da and Db set by the current control units 300 and 310 and the carrier waves CWa and CWb from the carrier wave generation unit 410. Control signals SG1 to SG4 are generated.

  FIG. 15 is a waveform diagram showing an example of control operation of the switching element according to the PWM control in the PB mode. FIG. 15 shows an example of control operation in the PB mode using carrier waves CWa and CWb having the same phase.

  Referring to FIG. 15, carrier wave CWa used for PWM control of DC power supply 10a and carrier wave CWb used for PWM control of DC power supply 10b have the same frequency and the same phase.

  A control pulse signal SDa is generated based on a voltage comparison between the duty ratio Da for controlling the output of the DC power supply 10a and the carrier wave CWa. Similarly, control pulse signal SDb is generated based on a comparison between duty ratio Db for controlling the output of DC power supply 10b and carrier wave CWb. Control pulse signals / SDa and / SDb are inverted signals of control pulse signals SDa and SDb.

  FIG. 16 shows logical operation expressions and setting expressions for setting control signals SG1 to SG4 and FF control amounts Dffa and Dffb in the PB mode.

  As shown in FIG. 16, control signals SG1 to SG4 are set based on a logical operation of control pulse signals SDa (/ SDa) and SDb (/ SDb). Further, the FF control amounts Dffa and Dffb are set according to the above formulas (7) and (9).

  Switching element S1 forms an upper arm element in each of the step-up chopper circuits of FIGS. Therefore, control signal SG1 for controlling on / off of switching element S1 is generated by the logical sum of control pulse signals / SDa and / SDb. As a result, the switching element S1 realizes both functions of the upper arm element of the boost chopper circuit (DC power supply 10a) in FIG. 7 and the upper arm element of the boost chopper circuit (DC power supply 10b) in FIG. ON / OFF controlled.

  The switching element S2 forms an upper arm element in the boost chopper circuit of FIG. 7, and forms a lower arm element in the boost chopper circuit of FIG. Therefore, control signal SG2 for controlling on / off of switching element S2 is generated by the logical sum of control pulse signals / SDa and SDb. Thus, the switching element S2 realizes both functions of the upper arm element of the boost chopper circuit (DC power supply 10a) in FIG. 7 and the lower arm element of the boost chopper circuit (DC power supply 10b) in FIG. ON / OFF controlled.

  Similarly, the control signal SG3 of the switching element S3 is generated by the logical sum of the control pulse signals SDa and SDb. Thereby, the switching element S3 realizes both functions of the lower arm element of the boost chopper circuit (DC power supply 10a) in FIG. 7 and the lower arm element of the boost chopper circuit (DC power supply 10b) in FIG. ON / OFF controlled.

  Further, the control signal SG4 of the switching element S4 is generated by a logical sum of the control pulse signals SDa and / SDb. Thereby, the switching element S4 realizes both functions of the lower arm element of the boost chopper circuit (DC power supply 10a) of FIG. 7 and the upper arm element of the boost chopper circuit (DC power supply 10b) of FIG. ON / OFF controlled.

  In the PB mode, control signals SG2 and SG4 are set to complementary levels, so switching elements S2 and S4 are turned on and off in a complementary manner. Accordingly, the operation when Vb> Va shown in FIG. 5 and the operation when Va> Vb shown in FIG. 6 are naturally switched. Furthermore, DC power conversion according to the duty ratios Da and Db can be executed for the DC power supplies 10a and 10b by turning on and off the switching elements S1 and S3 in a complementary manner.

  Referring to FIG. 15 again, control signals SG1 to SG4 are generated based on control pulse signals SDa (/ SDa) and SDb (/ SDb) according to the logical operation expression shown in FIG. By turning on and off switching elements S1 to S4 based on control signals SG1 to SG4, current I (L1) flowing through reactor L1 and current I (L2) flowing through reactor L2 are controlled. The current I (L1) corresponds to the current Ia of the DC power supply 10a, and the current I (L2) corresponds to the current Ib of the DC power supply 10b.

  As described above, in the PB mode, the DC / DC conversion is performed in parallel between the DC power supplies 10a and 10b and the power line 20, and the output voltage VH can be controlled to the voltage command value VH *. In particular, by converting the voltage deviation of the output voltage VH into a power command value and performing current control on the outputs of the DC power supplies 10a and 10b, it is possible to reliably protect from overpower on an output power basis. Further, it is possible to easily control the power distribution ratio k and the circulating power value Pr between the DC power supplies 10a and 10b.

  In particular, the power command value can be directly limited to one of the DC power supplies 10a and 10b. In the configuration example of FIG. 12, the limiter 260 can reliably limit the power command value Pa * of the DC power supply 10 a within the range of Pamin ≦ Pa * ≦ Pamax. Thereby, the overpower of DC power supply 10a can be prevented strictly.

  The total power command value PH * is limited to within the range of PHmin to PHmax to set the power command value Pb * of the DC power supply 10b, and the load power PL is limited to within the range of PHmin to PHmax. The power supply 10b can also be indirectly protected from overpower. However, in the configuration example of FIG. 12, the DC power supply 10a in which the power command value Pa * is directly limited by the limiter 260 is more strictly protected from overpower than the DC power supply 10b. Therefore, it is preferable that the limiter 260 directly limit the power command value of the DC power source that requires more strict protection from overpower.

(Modification of PWM control in PB mode)
FIG. 17 shows an example of control operation in the PB mode when a phase difference is intentionally provided between the carrier waves CWa and CWb.

  Referring to FIG. 17, carrier wave CWa and carrier wave CWb have the same frequency, but a phase difference φ is provided between them. In the example of FIG. 17, the phase difference φ = 180 degrees.

  Similarly to FIG. 15, control pulse signal SDa is generated based on comparison between carrier wave CWa and duty ratio Da, and control pulse signal SDb is generated based on comparison between carrier wave CWb and duty ratio Db.

In FIG. 17, the duty ratios Da and Db have the same values as in FIG. Therefore, the control pulse signal SDa in FIG. 17 is the same in length as the H level period although the phase is different compared to the control pulse signal SDa in FIG. Similarly, the control pulse signal SDb of FIG. 17 has the same length of the H level period although the phase is different from that of the control pulse signal SDb of FIG.

  Therefore, by providing phase difference φ between carrier waves CWa and CWb, control signals SG1 to SG4 in FIG. 17 have waveforms different from control signals SG1 to SG4 in FIG. 15 and 17, it is understood that the phase relationship (current phase) between the current I (L1) and the current I (L2) is changed by changing the phase difference φ between the carrier waves CWa and CWb. Is done.

  On the other hand, it is understood that the average values of the currents I (L1) and I (L2) are equivalent between FIGS. 15 and 17 for the same duty ratios Da and Db. That is, the outputs of the DC power supplies 10a and 10b are controlled by the duty ratios Da and Db, and there is no effect even if the phase difference φ between the carrier waves CWa and CWb is changed.

  Therefore, in power converter 50 according to the embodiment of the present invention, the switching loss of switching elements S1 to S4 is controlled by carrier phase control that appropriately adjusts phase difference φ between carrier waves CWa and CWb in the PB mode. Reduction can be achieved.

  In the following, as a representative example, control in a state where both DC power sources 10a and 10b are in a power running state, that is, a state where current I (L1)> 0 and current I (L2)> 0 will be described.

  FIG. 18 is a waveform diagram for explaining a current phase by carrier phase control in the PB mode in the power converter 50.

  Referring to FIG. 18, since switching elements S2 to S4 are turned on until time Ta, the lower arm element of the boost chopper circuit is turned on for both DC power supplies 10a and 10b. Therefore, both currents I (L1) and I (L2) rise.

  At time Ta, switching element S2 is turned off, so that the lower arm element of the step-up chopper circuit is turned off with respect to DC power supply 10b, and current I (L2) starts to decrease. Instead of switching off the switching element S2, the switching element S1 is turned on.

  After the time Ta, the lower arm element of the boost chopper circuit is turned on for the DC power supply 10a, and the lower arm element of the boost chopper circuit is turned off for the DC power supply 10b. That is, the current I (L1) increases while the current I (L2) decreases. At this time, the current path in the power converter 50 is as shown in FIG.

  As understood from FIG. 19A, after the time Ta, the difference current between the currents I (L1) and I (L2) passes through the switching element S4. That is, the passing current of the switching element S4 becomes small.

  Referring to FIG. 18 again, when switching element S4 is turned off from the state after time Ta, the lower arm element of the step-up chopper circuit is turned off with respect to DC power supply 10a, so that current I (L1) is Start descent. Further, when the switching element S2 is turned on, the lower arm element of the step-up chopper circuit is turned on with respect to the DC power supply 10b, so that the current I (L2) starts to rise again. That is, the current path in the power converter 50 changes from the state shown in FIG. 19A to the state shown in FIG. In the state of FIG. 19B, since the difference current between the currents I (L1) and I (L2) passes through the switching element S2, the passing current of the switching element S2 becomes small.

  By turning off the switching element S4 in the state of FIG. 19A, the current at the time of turning off the switching element S4, that is, the switching loss can be reduced. Further, by turning on the switching element S2 in the state of FIG. 19B, the current when the switching element S2 is turned on, that is, the switching loss can be reduced.

  Therefore, the current phase, that is, the phase difference φ between the carrier waves CWa and CWb is set so that the falling start timing (maximum point) of the current I (L1) and the rising timing (minimum point) of the current I (L2) overlap. adjust. Accordingly, at time Tb in FIG. 18, the switching element S2 is turned on and the switching element S4 is turned off.

  Referring to FIG. 18 again, at time Tc, switching element S1 is turned off and switching element S4 is turned on. As a result, the lower arm element of the step-up chopper circuit is turned on for each of the DC power supplies 10a and 10b. As a result, the state before time Ta described above is reproduced, and both currents I (L1) and I (L2) rise.

  FIG. 20 shows current waveforms of the switching elements S2 and S4 in the current phase shown in FIG. FIG. 20A shows a waveform of the current I (S2) of the switching element S2, and FIG. 20B shows a waveform of the current I (S4) of the switching element S4.

  Referring to FIG. 20A, current I (S2) is I (S2) = I (L2) in the period up to time Ta and the period after time Tc. In the period from time Ta to Tb, since the switching element S2 is turned off, I (S2) = 0. In the period from time Tb to Tc, as shown in FIG. 19B, I (S2) = − (I (L1) −I (L2)).

  Referring to FIG. 20B, current I (S4) is I (S4) = I (L1) in the period up to time Ta and the period after time Tc. In the period from time Ta to Tb, as shown in FIG. 19A, I (S4) = − (I (L2) −I (L1)). In the period from time Tb to Tc, the switching element S4 is turned off, so that I (S4) = 0.

  FIG. 21 shows a current phase when the phase difference between carrier waves is set to φ = 0 under the same duty ratio as in FIG. 18 for comparison with FIG.

  Referring to FIG. 21, when the phase difference φ = 0 between carrier waves CWa and CWb, the timings (Tx, Ty, Tz, Tw) at which currents I (L1) and I (L2) rise / fall are different. It will be a thing.

  Specifically, in a state where switching element S1 is turned off and switching elements S2 to S4 are turned on before time Tx, both currents I (L1) and I (L2) rise. Then, the switching element S4 is turned off at time Tx, whereby the current I (L1) starts to decrease. The switching element S1 is turned on instead of the switching element S4 being turned off.

  At time Ty, the switching element S3 is turned off, and the current I (L2) starts to decrease. The switching element S4 is turned on instead of the switching element S3 being turned off. As a result, both currents I (L1) and I (L2) drop.

At time Tz, the switching element S2 is turned off and the switching element S2 is turned off.
3 turns on. As a result, the lower arm element of the step-up chopper circuit is turned on with respect to the DC power supply 10a, so that the current I (L1) rises again. Furthermore, at time Tw, the switching element S1 is turned off and the switching element S2 is turned on. Thereby, since the state before time Tx is reproduced, both currents I (L1) and I (L2) rise.

  FIG. 22 shows current waveforms of switching elements S2 and S4 in the current phase shown in FIG. FIG. 22A shows the waveform of the current I (S2) of the switching element S2, and FIG. 22B shows the waveform of the current I (S4) of the switching element S4.

  Referring to FIG. 22A, current I (S2) is I (S2) = I (L2) in the period up to time Tx and the period after time Tw. In the period from time Tx to Ty, a current path similar to that in FIG. 19B is formed, and therefore I (S2) = − (I (L1) −I (L2)). Then, during the period from time Ty to Tz, it operates as an upper arm element for the DC power supply 10a, so that I (S2) = − I (L1). In the period from time Ty to Tz when both of the currents I (L1) and I (L2) drop, the switching element S2 operates as an upper arm element with respect to the DC power supply 10a, so I (S2) = − I (L1 ) In the period from time Tz to Tw, the switching element S2 is turned off, so that I (S2) = 0.

  Referring to FIG. 22B, current I (S4) is I (S4) = I (L1) in the period up to time Tx and the period after time Tw. In the period from time Tx to Ty, the switching element S4 is turned off, so that I (S4) = 0. Since the switching element S4 operates as an upper arm element with respect to the DC power supply 10b in the period from the time Ty to Tz when both of the currents I (L1) and I (L2) drop, I (S4) = − I (L2) Become. Between time Tz and Tw, a current path similar to that in FIG. 19A is formed, and therefore I (S2) = − (I (L2) −I (L1)).

  From the comparison between the current I (S2) generated at time Tb in FIG. 20A and the current I (S2) generated at time Tw in FIG. 22A, the phase difference φ is set so as to be the current phase in FIG. It is understood that by adjusting, the turn-on current of the switching element S2, that is, the switching loss at the time of turn-on is reduced. Further, from the comparison of the current I (S2) at times Tb to Tc in FIG. 20A and the current I (S2) at times Ty to Tz in FIG. Is also reduced.

Similarly, from the comparison between the current I (S4) at time Tb in FIG. 20B and the current I (S4) at time Tx in FIG. 22B, the current phase in FIG. It is understood that by adjusting the phase difference φ, the turn-off current of the switching element S4, that is, the switching loss at the time of turn-off is reduced. Furthermore, from the comparison between the current I (S4) at time Ta to Tb in FIG. 20B and the current I (S4) at time Ty to Tz in FIG. Thus, the loss in the switching elements S1 to S4 can be reduced by providing the phase difference φ between the carrier waves CWa and CWb. As shown in FIG. 18, in the state where both DC power supplies 10a and 20 are powered, the current I (L1) starts to decrease (maximum point) and the current I (L2) increases (timing point). By setting the phase difference φ so that they overlap, that is, the turn-on timing of the switching element S2 and the turn-off timing of the switching element S4 coincide with each other, loss in the switching elements S1 to S4 is suppressed.

  As a result, DC power conversion between the DC power supplies 10a and 20 and the power line 20 (load 30) can be executed with high efficiency. With such a phase difference φ, the falling timing (or rising timing) of the control pulse signal SDa and the rising timing (or falling timing) of the control pulse signal SDb overlap. In other words, it is necessary to adjust the phase difference φ so that the pulse transition timing of the control pulse signal SDa matches the pulse transition timing of the control pulse signal SDb. The transition timing indicates the timing at which the H level / L level of the pulse is switched.

  As can be understood from FIGS. 15 and 17, the control pulse signals SDa and SDb vary depending on the duty ratios Da and Db. Therefore, it can be understood that the phase difference φ at which the current phase as shown in FIG. For this reason, the relationship between the duty ratios Da and Db and the phase difference φ by the carrier phase control is obtained in advance, and the corresponding relationship is preliminarily mapped (hereinafter also referred to as “phase difference map”) or a function equation (hereinafter referred to as “level” It can be stored in the control device 40 as a “phase difference calculation formula”.

  In the PWM control for current control in the DC power supplies 10a and 10b in the PB mode described in FIGS. 12 and 13, the phase difference for carrier phase control is based on the calculated duty ratios Da and Db. φ can be calculated. Then, the carrier wave generation unit 410 (FIG. 13) generates carrier waves CWa and CWb so as to have the calculated phase difference φ, thereby suppressing the loss in the switching elements S1 to S4 with high efficiency DC / DC conversion can be realized.

  Although FIGS. 18-22 demonstrated the state where both DC power supply 10a and 20 were power running, the same carrier phase control can be performed also in another state.

FIG. 23 is a chart for illustrating carrier phase control according to the first embodiment of the present invention in each operation state of the DC power supply. Referring to FIG. 23, in state A, the above-described DC power supplies 10a and 10b Both are in a power running state. As shown in FIG. 18, the current phase is such that the fall timing (maximum point) of current I (L1) and the rise timing (minimum point) of current I (L2) overlap at Tb in the figure. The phase difference φ of the carrier wave is adjusted. Thereby, the turn-on loss of switching element S2 and the turn-off loss of switching element S4 in Tb can be reduced. Furthermore, as described above, the conduction loss of the switching element S4 in the period of Ta to Tb and the conduction loss of the switching element S2 in the period of Tb to Tc can be reduced.

  In state B, both DC power supplies 10a and 10b are in a regenerative state. In this state, the carrier wave has a current phase so that the rising timing (minimum point) of the current I (L1) and the falling timing (maximum point) of the current I (L2) overlap at Tb in the figure. Adjust the phase difference φ. Thereby, the turn-on loss of switching element S4 and the turn-off loss of switching element S2 in Tb can be reduced. Furthermore, as described above, the conduction loss of the switching element S2 during the period of Ta to Tb and the conduction loss of the switching element S4 during the period of Tb to Tc can be reduced.

  In state C, DC power supply 10a is in a regenerative state, while DC power supply 10b is in a powering state. In this state, the carrier wave has a current phase such that the fall timing (maximum point) of the current I (L1) and the fall timing (minimum point) of the current I (L2) overlap with Ta in the figure. Adjust the phase difference φ. Thereby, the turn-on loss of switching element S3 and the turn-off loss of switching element S1 in Ta can be reduced. Furthermore, as described above, the conduction loss of the switching element S1 during the period of Ta to Tb and the conduction loss of the switching element S3 during the period of Tc to Ta can be reduced.

  Furthermore, in the state D, the DC power supply 10a is in a power running state, while the DC power supply 10b is in a regenerative state. In this state, the phase difference φ of the carrier wave is adjusted so that the rising timing of the current I (L1) and the rising timing of the current I (L2) have a current phase overlapping at Tc in the drawing. Thereby, the turn-on loss of the switching element S1 and the turn-off loss of the switching element S3 at Tc can be reduced. Furthermore, as described above, the conduction loss of the switching element S1 during the period of Tb to Tc and the conduction loss of the switching element S3 during the period of Tc to Ta can be reduced.

  Thus, the phase difference φ for reducing the loss in the switching elements S1 to S4 differs depending on the combination of the power running / regenerative state of the DC power supplies 10a and 10b. Therefore, it is preferable to set the above-described phase difference map or phase difference calculation formula for each power running / regenerative state combination (states A to D in FIG. 23).

  Thus, in power converter 50 according to the present embodiment, DC / DC conversion in the PB mode for controlling output voltage VH to voltage command value VH * according to the control calculation shown in FIGS. 12 and 13. The carrier phase control by the carrier wave generator 410 (FIG. 13) can be combined. Thereby, when the voltage deviation of the output voltage VH is converted into a power command value and the output of each DC power supply 10a, 10b is current-controlled, the DC / DC conversion with high efficiency in which the loss of the switching elements S1 to S4 is reduced. Can be executed.

[Embodiment 2]
In the first embodiment, the control operation in the PB mode according to the control configuration of FIGS. 12 and 13 has been described. As shown in FIG. 3, in power converter 50 according to the first embodiment, a boost mode in which output voltage VH is controlled to voltage command value VH * is set to aB mode, bB mode in addition to PB mode. And SB mode exists. In Embodiment 2, power converter control according to the present embodiment in these operation modes will be described.

  Also in the aB mode, bB mode, and SB mode, the output voltage VH is controlled to the current command value VH * by sharing the control configuration according to FIGS. 12 and 13 described in the first embodiment.

  FIG. 24 is a table for explaining setting of control signals and control data in each operation mode belonging to the boost mode.

  Referring to FIG. 24, the control configurations shown in FIGS. 12 and 13 are shared in each operation mode in the boost mode. Then, by changing the power distribution ratio k, the DC power source to be subjected to the current feedback control, and the arithmetic logic of the control signals SG1 to SG4, the operation mode is dealt with.

  In the PB mode, as described in the first embodiment, the power distribution ratio k can be arbitrarily set within the range of 0 ≦ k ≦ 1.0, and the circulating power value Pr is also arbitrary in terms of control. The value can be set. As described above, in the PB mode, both currents Ia and Ib of DC power supplies 10a and 10b are controlled according to current command values Ia * and Ib * set based on a power command value for controlling output voltage VH. Is done.

(Control operation in aB mode)
In the aB mode, the DC power supply 10b is not used by the boost chopper circuit formed by the switching elements S1 to S4 by the switching operation shown in FIGS. Bidirectional DC / DC conversion is performed between the loads 30). Therefore, in the aB mode, switching elements S1 to S4 are controlled in accordance with control pulse signal SDa based on duty ratio Da for controlling the output of DC power supply 10a. Specifically, switching elements S3 and S4 constituting the lower arm element of the step-up chopper circuit shown in FIGS. 7A and 7B are commonly turned on / off in accordance with control pulse signal SDa. Similarly, switching elements S1 and S2 constituting the upper arm element of the step-up chopper circuit are commonly turned on / off in accordance with control pulse signal / SDa.

  24 and 12, in the aB mode, as in the PB mode, the total power command value is calculated based on the voltage deviation ΔVH of the output voltage VH by the deviation calculation unit 210, the control calculation unit 220, and the limiter 230. PH * is set. Since DC power supply 10b is not used, power upper limit PHmax and power lower limit PHmin given to limiter 230 can be set equal to power upper limit Pamax and power lower limit Pamin of DC power supply 10a. . Correspondingly, in the aB mode, the operation command value of the load 30 is generated while being limited to a range satisfying Pamin ≦ PL ≦ Pamax.

  In the aB mode, the DC power supply 10b is not used (charging / discharging avoidance), and thus the circulating power Pr = 0 is fixed. Further, by fixing the power distribution ratio k = 1.0, the power command value Pa * = PH * is set, while the power command value Pb * = 0 is set. At this time, the limiter 260 can also protect the electric power command value Pa * so that it does not deviate from the range of Pamax to Pamin, that is, prevent overpower from being generated in the DC power supply 10a. Therefore, in the aB mode, one of limiters 230 and 260 can be deactivated.

  Furthermore, in the configuration of FIG. 13, the current feedback control is executed only for the DC power supply 10a. That is, as in the PB mode, the current control unit 300 performs the feedback control shown in Expression (6) based on the current deviation between the current command value Ia * set according to the power command value Pa * and the detected value of the current Ia. Then, the duty ratio Da is calculated (Da = Dfba + Dfba) by feedforward control based on the voltage ratio shown in Expression (7).

  In contrast, in the aB mode, the control pulse signal SDb is unnecessary as described above, and thus the operation of the current control unit 310 can be stopped. That is, the calculation of the duty ratio Db is stopped.

  FIG. 25 is a conceptual diagram for explaining the power flow in the power supply system in the aB mode.

  Referring to FIG. 25, in the aB mode, power command value PH * for controlling output voltage VH to voltage command value VH * is all distributed to DC power supply 10a. That is, power PL input / output to / from load 30 is covered only by DC power supply 10a. Further, since the circulating power value Pr is fixed at 0, charging / discharging between the DC power supplies 10a and 10b does not occur.

  Even in the aB mode, the load power PL and the power command value Pa * are reliably limited within the range of Pamax to Pamin by the limiters 260 and / or 290. For this reason, the direct-current power supply 10a used alone can be protected from overpower. Also, in the aB mode, by calculating the duty ratio Da by feedback control of the current Ia of the DC power supply 10a, the voltage deviation ΔVH can be quickly compared with control in which the duty ratio Da is calculated only by feedback control of the output power VH. Can be resolved.

(Control operation in bB mode)
In the bB mode, the DC power supply 10a is not used by the boost chopper circuit formed by the switching elements S1 to S4 by the switching operation shown in FIGS. 8A and 8B, while the DC power supply 10b and the load 30 are not used. Bidirectional DC / DC conversion is performed between them. Therefore, in the bB mode, switching elements S1 to S4 are controlled according to control pulse signal SDb based on duty ratio Db for controlling the output of DC power supply 10b. Specifically, switching elements S2 and S3 constituting the lower arm element of the step-up chopper circuit shown in FIGS. 8A and 8B are commonly turned on / off in accordance with control pulse signal SDb. Similarly, switching elements S1 and S4 constituting the upper arm element of the boost chopper circuit are commonly controlled to be turned on / off according to control pulse signal / SDb. Referring to FIG. 24 and FIG. Similar to the aB mode, the total power command value PH * is set based on the voltage deviation ΔVH of the output voltage VH. Since the DC power supply 10a is not used in the bB mode, the power upper limit PHmax and the power lower limit PHmin given to the limiter 230 are set to be equal to the power upper limit Pbmax and the power lower limit Pbmin of the DC power supply 10b. Similarly, the circulating power Pr = 0 is fixed.

  Further, by fixing the power distribution ratio k = 0, the power command value Pb * = PH * is set while the power command value Pa * = 0. In this case, the limiter 260 does not need to be restricted. That is, in the bB mode, the limiter 230 can directly protect the DC power supply 10b from overpower.

  Further, in the configuration of FIG. 13, the current feedback control is executed only for the DC power supply 10b. That is, as in the PB mode, the current control unit 310 performs the feedback control shown in Expression (8) based on the current deviation between the current command value Ib * set according to the power command value Pb * and the detected value of the current Ib. Then, the duty ratio Db is calculated (Db = Dfbb + Dfbb) by feedforward control based on the voltage ratio shown in Expression (9).

  On the other hand, in the bB mode, the control pulse signal SDa is unnecessary as described above, and thus the operation of the current control unit 300 can be stopped. That is, the calculation of the duty ratio Da is stopped.

  FIG. 26 is a conceptual diagram for explaining the power flow in the power supply system in the bB mode.

  Referring to FIG. 26, in the bB mode, power command value PH * necessary for controlling output voltage VH to voltage command value VH * is all distributed to DC power supply 10b. That is, the power PL input / output to / from the load 30 is covered only by the DC power supply 10b. Further, since the circulating power value Pr is fixed at 0, charging / discharging between the DC power supplies 10a and 10b does not occur.

  Also in the bB mode, the power upper limit value PHmax and the power lower limit value PHmin given to the limiter 230 can be set equal to the power upper limit value Pbmax and the power lower limit value Pbmin of the DC power supply 10b. As a result, power command value Pb * is reliably limited within the range of Pbmax to Pbmin. Further, in the bB mode, the operation command value of the load 30 is generated while being limited within the range of Pbmin ≦ PL ≦ Pbmax. As a result, the DC power supply 10b that is used alone can be protected from overpower. Further, in the bB mode, by performing feedback control of the current Ib of the DC power supply 10b, the generated voltage deviation ΔVH can be quickly eliminated as compared with the control in which the DC voltage VH is canceled by direct feedback control.

(SB mode control operation)
Next, the control operation in the SB mode will be described.

  FIG. 27 is a conceptual diagram for explaining the power flow in the power supply system in the SB mode.

  Referring to FIG. 27, in the SB mode, bidirectional DC / DC conversion is performed with power line 20 (load 30) with DC power supplies 10a and 10b connected in series. Therefore, the currents flowing through the DC power supply 10a and the DC power supply 10b are common (Ia = Ib). For this reason, the output power Pa of the DC power supply 10a and the output power Pb of the DC power supply 10b cannot be directly controlled.

  The ratio between the electric power Pa and Pb is automatically determined according to the following equation (10) depending on the ratio between the voltages Va and Vb.

Pa: Pb = Va: Vb (10)
Further, the power PH input / output to / from the power line 20 is obtained by the sum (Pa + Pb) of the output power from both DC power supplies, as in the PB mode (PH = Pa + Pb).

  Basically, in the SB mode, DC / DC conversion between the DC voltage (Va + Vb) and the output voltage VH is executed by the boost chopper circuit shown in FIG. For this reason, as shown in FIG. 28, according to the duty ratio Dc shown in the above equation (3), the period during which the pair of switching elements S2 and S4 constituting the lower arm element is turned on, and the upper arm element A period in which the switching element S1 is turned on is complementarily provided. That is, the control pulse signal SDc generated by comparing the duty ratio Dc and the carrier wave is set as the control signals SG2 and SG4. Similarly, the control signal SG1 is obtained by a control pulse signal / SDc obtained by inverting the control pulse signal SDc. Control signal SG3 is fixed at H level in order to keep switching element S3 on.

  In the power converter control according to the second embodiment, the output voltage VH is controlled to the current command value VH * by sharing the control configuration according to FIGS. 12 and 13 described in the first embodiment even in the SB mode. Further, by combining the carrier phase difference control described in the first embodiment, control signals SG1 to SG4 in the SB mode are generated using control pulse signals SDa (/ SDa) and SDb (/ SDb) in FIG. The

  Referring to FIGS. 24 and 12 again, in the SB mode, the power distribution ratio k is the current value of the voltages Va and Vb of the DC power supplies 10a and 10b according to the equation (11) obtained along the equation (10). Detection value).

k = Va / (Va + Vb) (11)
In the SB mode, charging / discharging between the DC power supplies 10a and 10b cannot be performed, so the circulating power value Pr = 0 is set.

  Thus, in the configuration of FIG. 12, the total power command value PH * is set based on the voltage deviation ΔVH of the output voltage VH, as in the SB mode. Total power command value PH * can be set within a range of PHmax to PHmin by limiter 230. Further, in accordance with the power distribution ratio k based on the current voltages Va and Vb between the DC power supplies 10a and 10b connected in series according to the equation (11), the total power command value PH * becomes the power command value Pa * and Distributed to Pb *. At this time, the limiter 260 limits the power command value Pa * within the range of Pamax to Pamin.

  As shown in FIG. 24, in the SB mode, since Ia = Ib, the current feedback control is executed by only one of the DC power supplies 10a and 10b. For example, it is possible to directly limit the power command value, that is, current feedback control is performed on the DC power supply 10a that is strictly protected from overpower.

  Referring to FIG. 13 again, as in the PB mode, current control unit 300 uses equation (6) based on the current deviation between current command value Ia * set according to power command value Pa * and the detected value of current Ia. The duty ratio Da is calculated (Da = Dfba + Dfba) by the feedback control shown in FIG. 5 and the feedforward control based on the voltage ratio shown in the equation (7).

  On the other hand, in the current control unit 310, the current feedback control is not executed by setting the control gain in the control calculation unit 316, specifically, Kp and Ki in Expression (8) to zero. Therefore, the current control unit 310 calculates the duty ratio Db only by feedforward control based on the voltage Vb (Db = Dffb). The FF control amount Dffb can be set according to Equation (9).

  The PWM control unit 400 controls the switching elements S1 to S4 by pulse width modulation control based on the duty ratios Da and Db set by the current control units 300 and 310 and the carrier waves CWa and CWb from the carrier wave generation unit 410. Control signals SG1 to SG4 are generated.

  In the SB mode, carrier phase difference control as described below is executed in order to generate control signals SG1 to SG4 using control pulse signals SDa (/ SDa) and SDb (/ SDb).

  As shown in FIG. 29, since DC power supplies 10a and 10b are connected in series in the SB mode, both DC power supplies 10a and 10b are in a power running state (state A in FIG. 23) and DC power supplies 10a and 10b. There is only one of the states (state B in FIG. 23) in which both of them are regenerated.

  Therefore, in the control operation in the SB mode, the phase difference φ between the carrier waves is switched so that the turn-on of the switching element S2 and the turn-off of the switching element S4 overlap as shown in the states A and B of FIG. It is set so that the turn-on of the element S4 and the turn-off of the switching element S2 overlap.

  That is, the positions of the carrier waves CWa and CWb are such that the falling timing of the control pulse signal SDa and the rising timing of the control pulse signal SDb, or the rising timing of the control pulse signal SDa and the falling timing of the control pulse signal SDb overlap. By setting the phase difference φ, the current phases shown in the states A and B in FIG. 23 are realized.

  Consider the duty ratios Da and Db at this time. By transforming equation (1), the following equation (12) is obtained for Da.

Da = (VH−Va) / VH (12)
Similarly, the following equation (13) is obtained for Db by modifying equation (2).

Db = (VH−Vb) / VH (13)
As shown in FIG. 28, control signal SG3 in the PB mode is generated based on the logical sum of control pulse signals SDa and SDb. Therefore, when the phase difference φ is set so that the falling (or rising) timing of the control pulse signal SDa and the rising (or falling) timing of the control pulse signal SDb overlap, when VH> (Va + Vb) is satisfied, It is understood that the ratio of the H level period of the control signal SG3 in the PB mode exceeds 1.0. That is, when VH> (Va + Vb), the control signal SG3 is fixed to the H level also by PWM control common to the PB mode with the duty ratios Da and Db.

  FIG. 30 is a waveform diagram showing a control pulse signal in the SB mode when carrier phase control is applied.

  As shown in FIG. 28, control signal SG1 in the PB mode is generated based on the logical sum of control pulse signals / SDa and / SDb.

  Referring to FIG. 30, when phase difference φ is set as described above, the rising timing of control pulse signal / SDa and the rising timing of control pulse signal / SDb overlap. Therefore, the duty ratio DSG1 of the control signal SG1 = (1−Da) + (1−Db). That is, DSG1 is expressed by the following equation (14).

DSG1 = (Va + Vb) / VH (14)
On the other hand, the duty ratio Dc is expressed by the following equation (15) by modifying the equation (3).

Dc = 1− (Va + Vb) / VH (15)
Therefore, if SG1 = / SGc according to the logical operation in the SB mode of FIG. 28, the duty ratio DSG1 of the control signal SG1 is expressed by the following equation (16).

DSG1 = 1−Dc = (Va + Vb) / VH (16)
As described above, when the phase difference φ is set according to the carrier phase control described above, the logical operation based on the control pulse signals SDa and SDb based on the duty ratios Da and Db, specifically, the logical sum of / SDa and / SDb. Thus, a signal having the same duty ratio as the control pulse signal / SDc based on the duty ratio Dc can be generated. That is, the control signal SG1 in the SB mode can be generated based on the control pulse signals SDa and SDb.

  As shown in FIG. 28, the control signals SG2 and SG4 in the SB mode are inverted signals of the control signal SG1. The logical operation result of not (/ SDb or / SDa) is the logical product (SDb and SDa) of SDa and SDb. Therefore, control signals SG2 and SG4 to be set according to control pulse signal SDc can also be generated based on the logical operation of control pulse signals SDa and SDb.

  Thus, in the SB mode, the carrier phase control is applied, and the phase difference φ is set so that the pulse transition timing is matched between the control pulse signal SDa (/ SDa) and the control pulse signal SDb (/ SDb). Is done. The carrier wave generator 410 generates the carrier waves CWa and CWb so as to have such a phase difference φ, and as shown in FIG. 28, control to be set based on the duty ratio Dc in the SB mode. Signals SG1 to SG4 can be generated from control pulse signals SDa and SDb based on duty ratios Da and Db.

  Specifically, as described above, control signal SG3 is a signal fixed at the H level by the logical sum of control pulse signals SDa and SDb. Control signal SG1 can be generated to have a duty equivalent to that of PWM control based on duty ratio Dc by the logical sum of control pulse signals / SDa and / SDb. In the SB mode, the control signals SG2 and SG4 set complementarily to the control signal SG1 can also be generated by the logical product of the control pulse signals SDa and SDb.

  Note that the phase difference φ in the SB mode also follows a preset phase difference map or phase difference calculation formula that stores the relationship between the duty ratios Da and Db and the phase difference φ, as in the carrier phase control in the PB mode. , And can be calculated based on the duty ratios Da and Db (FIG. 13) calculated in the SB mode.

  FIG. 31 is a waveform diagram showing an operation example of the PB mode and the SB mode in the power converter control according to the second embodiment.

  Referring to FIG. 31, a command for switching from PB mode to SB is issued at the peak of carrier wave CWa. Before the switching command is generated, control signals SG1 to SG4 are generated based on the duty ratios Da and Db calculated by the current control of each of the DC power supplies 10a and 10b.

  When the switching command is issued, the control signal in the SB mode is immediately calculated based on the control pulse signals SDa and SDb at that time without newly calculating the duty ratio Dc according to the logical operation expression shown in FIG. SG1 to SG4 can be generated.

  Therefore, the control signals SG1 to SG4 in the SB mode can be generated using the duty ratios Da and Db in common with other operation modes belonging to the boost mode including the PB mode. In particular, as described with reference to FIG. 31, the switching process between the PB mode and the SB mode can be executed without causing a control delay when the operation mode is switched.

  As described above, with the power converter control according to the first and second embodiments, each operation belonging to the boost mode for controlling the output voltage VH to the voltage command value VH * with respect to the control operation of the power converter 50 shown in FIG. The control configuration shown in FIGS. 12 and 13 can be shared between the modes.

  Specifically, the common control calculation according to FIGS. 12 and 13 is applied between the operation modes by switching the power distribution ratio k and the control gain of the current control units 300 and 310 between the operation modes. Is possible. For this reason, it is possible to reduce the control calculation load in control of the power converter 50 which selectively applies a plurality of operation modes.

  Furthermore, since the duty ratio Da can be calculated by the current feedback control of the DC power supply 10a, the voltage deviation ΔVH in the SB mode can be quickly eliminated as compared with the control for calculating the duty ratio (Dc) by the feedback control of the output voltage VH. can do.

  In general, in the region of VH> (Va + Vb), it is preferable to select the PB mode because reducing the current level by applying the SB mode improves the efficiency of the power supply system. However, in the SB mode, since power distribution between the DC power supplies 10a and 10b cannot be controlled, when an SOC imbalance or the like occurs or the output of any DC power supply is limited (Par) When> Pamax or Par <Pamin, etc.), it is preferable to switch the operation mode to the PB mode. As described above, by sharing the control calculation among the operation modes, it becomes possible to smoothly switch the operation modes, and thus the controllability can be further improved.

[Embodiment 3]
In the third embodiment, application of power converter control according to the present embodiment to a power converter having a configuration different from that in FIG. 1 will be described.

  FIG. 32 is a circuit diagram showing a configuration example of a power supply system including a power converter according to the third embodiment of the present invention.

  Referring to FIG. 32, power converter 50 # according to the third embodiment includes boost chopper circuits 6 and 7. The step-up chopper circuit 6 performs bidirectional DC / DC conversion between the DC power supply 10 a and the power line 20 connected to the load 30. Boost chopper circuit 6 includes switching elements S5 and S6 and a reactor L1.

  The step-up chopper circuit 7 performs bidirectional DC / DC conversion between the DC power supply 10b and the DC power supply 10a and the common power line 20. Boost chopper circuit 7 includes switching elements S7 and S8 and a reactor L2.

  Anti-parallel diodes D5 to D8 are arranged for switching elements S5 to S8. Further, the switching elements S5 to S8 can control on / off in response to control signals SG5 to SG8 from the control device 40, respectively.

  Thus, power converter 50 # differs from power converter 50 according to the present embodiment in that boost chopper circuits 6 and 7 are provided independently for each of DC power supplies 10a and 10b. Yes. Boost chopper circuits 6 and 7 can be controlled independently.

  The control device 40 generates control signals SG5 to SG8 for controlling on / off of the switching elements S5 to S8 in order to control the output voltage VH.

FIG. 33 shows a plurality of operation modes possessed by power converter 50 # shown in FIG.
Referring to FIG. 33, power converter 50 # can select other boost mode and direct connection mode except SB mode and SD mode in power converter 50. That is, power converter 50 # has an operation mode of PB mode, aB mode and bB mode belonging to the boost mode, and PD mode, aD mode and bD mode belonging to direct connection mode.

  In the PB mode, by controlling the boost chopper circuits 6 and 7 independently, the control can be performed similarly to the PB mode in the first embodiment. That is, according to the configuration of FIGS. 12 and 13, the power distribution ratio k (0 ≦ k ≦ 1.0) and the circulating power value Pr can be set according to the state of the DC power supplies 10a and 10b. Thus, with the configuration of FIG. 12, the DC power sources 10a and 10b reflect the power distribution ratio k and the circulating power value Pr from the total power command value PH * for controlling the output voltage VH to the voltage command value VH *. Power command values Pa * and Pb * can be set.

  Furthermore, according to the configuration of FIG. 13, both currents Ia and Ib of DC power supplies 10a and 10b are controlled according to current command values Ia * and Ib * set based on the power command value for controlling output voltage VH. As described above, the duty ratios Da and Db can be calculated. Since boost chopper circuits 6 and 7 are independently controlled, control signals SG5 and SG6 of switching elements S5 and S6 of boost chopper circuit 6 are generated based on control pulse signal SDa. Specifically, the control signal SG6 = / SGa of the switching element S6 constituting the lower arm element, and the control signal SG5 = SGa of the switching element S5 constituting the upper arm element.

  Similarly, the control signals SG7 and SG8 of the switching elements S7 and S8 of the boost chopper circuit 7 are generated based on the control pulse signal SDb. Specifically, the control signal SG8 = / SGb of the switching element S8 constituting the lower arm element, and the control signal SG7 = SGb of the switching element S7 constituting the upper arm element.

  Also in PB mode of power converter 50 #, voltage command values Pa * and Pb * are set by converting the voltage deviation of output voltage VH into a power command value in accordance with the control configuration shown in FIGS. Thus, the current of the outputs of the DC power supplies 10a and 10b can be controlled. Furthermore, the limiters 230 and 260 enable reliable protection from overpower on an output power basis. Further, the power distribution ratio k and the circulating power value Pr between the DC power supplies 10a and 10b can be easily controlled.

  In the PB mode of power converter 50 #, the current paths of switching elements S5 and S6 and the current paths of switching elements S7 and S8 do not overlap. Therefore, even if carrier phase control is applied, switching elements S5 to S8 Power loss cannot be reduced. Therefore, it is not necessary to apply carrier phase control, and the phase difference φ can be fixed (typically, φ = 0).

  In the aB mode, the DC power supply 10b is not used by operating only the boost chopper circuit 6, while the output voltage VH is set to the voltage by bidirectional DC / DC conversion between the DC power supply 10a and the power line 20. Control to command value VH * is possible. That is, similarly to the aB mode (Embodiment 1) of power converter 50, by setting power distribution ratio k = 1.0 and circulating power value Pr = 0, limiter 230 or 260 causes Pamin ≦ Pa * ≦ Pamax. The power command value Pa * of the DC power supply 10a to be used can be set (Pa * = PH *).

  Further, in the configuration of FIG. 13, the current control unit 300 corresponding to the DC power supply 10a operates in the same manner as the PB mode of the power converter 50, and is based on the current feedback control (current command value Ia *) and the voltage ratio. The duty ratio Da is calculated by feedforward control (Da = Dfba + Dfba). On the other hand, in the aB mode, since the calculation of the control pulse signal SDb is unnecessary, the operation of the current control unit 310 can be stopped.

  In the aB mode, the switching elements S7 and S8 constituting the boost chopper circuit 7 are kept off. On the other hand, switching elements S5 and S6 constituting boosting chopper circuit 6 are turned on / off according to control pulse signal SDa (/ SDa) generated by pulse width modulation control based on duty ratio Da.

  In the bB mode, the DC power supply 10a is not used by operating only the step-up chopper circuit 7, while the output voltage VH is set to the voltage by bidirectional DC / DC conversion between the DC power supply 10b and the power line 20. Control to command value VH * is possible. Thus, similarly to the bB mode (Embodiment 1) of power converter 50, by setting power distribution ratio k = 0 and circulating power value Pr = 0, limiter 230 enables power protection of Pbmin ≦ Pb * ≦ Pbmax. , The power command value Pb * of the DC power supply 10b to be used can be set (Pb * = PH *).

  In the configuration of FIG. 13, the current control unit 310 corresponding to the DC power supply 10b operates in the same manner as in the PB mode of the power converter 50, and feeds forward based on current feedback control (current command value Ib *) and voltage ratio. According to the control, the duty ratio Db is calculated (Db = Dfbb + Dfbb). On the other hand, in the bB mode, since the calculation of the control pulse signal SDa is not necessary, the operation of the current control unit 300 can be stopped.

  In the bB mode, the switching elements S5 and S6 constituting the boost chopper circuit 6 are kept off. On the other hand, the switching elements S7 and S8 constituting the boost chopper circuit 7 are turned on / off according to the control pulse signal SDb (/ SDb) generated by the pulse width modulation control based on the duty ratio Db.

  In the PD mode, the switching elements S5 and S7 are fixed on, while the switching elements S6 and S8 are fixed off. As a result, similarly to the SD mode in the power converter 50, the output voltage VH is equivalent to the output voltages Va and Vb (strictly, the higher voltage of Va and Vb) of the DC power supplies 10a and 10. . Similar to the power converter 50, the voltage difference between Va and Vb causes a short-circuit current in the DC power supplies 10a and 10b. Therefore, the PD mode can be applied only when the voltage difference is small.

  In the aD mode, the switching element S5 is fixed on, while the switching elements S6 to S8 are fixed off. Thereby, like the aD mode in the power converter 50, the DC power supply 10b is disconnected from the power line 20, and the output voltage VH is equivalent to the voltage Va of the DC power supply 10a (VH = Va). As described above, Va> Vb is a necessary condition for applying the aD mode.

  In the bD mode, the switching element S7 is fixed on, while the switching elements S5, S6, and S8 are fixed off. As a result, as in the bD mode in the power converter 50, the DC power supply 10a is disconnected from the power line 20, and the output voltage VH is equivalent to the voltage Vb of the DC power supply 10b (VH = Vb). As described above, Vb> Va is a necessary condition for applying the bD mode.

  Thus, power converter 50 # according to the third embodiment also belongs to the boost mode in which output voltage VH is controlled to voltage command value VH *, similarly to power converter 50 described in the first and second embodiments. The control configuration shown in FIGS. 12 and 13 can be shared among a plurality of operation modes (PB mode, aB mode, and bB mode). Similarly to the power converter 50, the PD mode, the aD mode, and the bD mode can be realized as the direct connection mode.

  Specifically, by switching the power distribution ratio k or the like between the operation modes, the common control calculation according to FIGS. 12 and 13 can be applied between the operation modes. Therefore, it is possible to reduce the control calculation load in the control of power converter 50 # that selectively applies a plurality of operation modes. Furthermore, since the duty ratios Da and Db of the boost chopper circuits 6 and 7 can be calculated by feedback control of the currents Ia and Ib, the generated voltage deviation ΔVH can be quickly compared with the control calculated by feedback control of the output voltage VH. Can be resolved.

  In the present embodiment, power converters 50 and 50 # that perform DC / DC conversion between two DC power supplies 10a and 10b and common power line 20 are illustrated, but three or more DC power supplies are illustrated. The control configuration shown in FIGS. 12 and 13 can be applied to a configuration in which a power supply is provided. For example, power converter 50 # can be extended to provide a boost chopper circuit in parallel corresponding to each of n (n ≧ 3) DC power supplies. In this case, the power distribution ratio among the n DC power sources is set for n (n ≧ 3) DC power sources, and the limiter 260 is set for (n−1) DC power sources. The power command value limitation equivalent to that according to FIG. 13 can be executed. At this time, the power protection for the remaining one DC power supply is indirectly secured by the limitation of the total power command value PH * by the limiter 230 (FIG. 13). Regarding the configuration of power converter 50 #, not only the illustrated boost chopper but also a buck-boost converter can be applied to at least one DC power supply instead of the boost chopper for the converters arranged in parallel. is there.

  Furthermore, the load 30 will be described in terms of confirmation that it can be configured by any device as long as it is a device that operates with the DC voltage VH. That is, in the present embodiment, the example in which the load 30 is configured to include the electric motor for traveling of the electric vehicle has been described, but the application of the present invention is not limited to such a case.

  The embodiment disclosed this time should be considered as illustrative in all points and not restrictive. The scope of the present invention is defined by the terms of the claims, rather than the description above, and is intended to include any modifications within the scope and meaning equivalent to the terms of the claims.

  5, power supply system, 6, 7 step-up chopper circuit, 10a, 10b DC power supply, 15 wiring, 20 power line, 21 ground wiring, 30 load, 32 inverter, 35 motor generator, 36 power transmission gear, 37 drive wheel, 40 control device, 50, 50 # power converter, 150, 151, 160, 161, 170-174 current path, 200 voltage control unit, 210, 304, 314 deviation calculation unit, 220, 306, 316 control calculation unit, 230, 260 limiter, 240 power distribution unit, 250 circulating power addition unit, 270 subtraction unit, 290 power management unit, 300, 310 current control unit, 302, 312 current command generation unit, 308, 318 addition unit, 400 PWM control unit, 410 carrier wave generation Part, CH smoothing capacitor, CWa, CWb carrier wave, D 1 to D8 Reverse parallel diode, Da, Db, Dc Duty ratio, Dfba, Dfbb Control amount (feedback control), Dffa, Dffb control amount (feed forward control), Ia, Ib Current (DC power supply), Ia *, Ib * Current command value, k Power distribution ratio, L1, L2 reactor, N1, N2, N3 nodes, PH total power, PH * Total power command value, PH total power, PHmax, Pamax, Pbmax Power upper limit value, PHmin, Pamin, Pbmin power lower limit value, Pr circulating power value, S1 to S8 power semiconductor switching element, SDa, / SDa, SDb, / SDb, SDc, / SDc control pulse signal, SG1 to SG8 control signal, VH output voltage, VH * voltage Command value, Va, Vb voltage (DC power supply).

Claims (12)

A control device for a power converter connected between a power line connected to a load and the first and second DC power supplies, and including a plurality of switching elements,
The power converter performs parallel DC voltage conversion between the first and second DC power supplies and the power line, and the first and second connected in series by ON / OFF control of the plurality of switching elements. A plurality of operations that are configured to be able to switch between DC voltage conversion between the DC power source and the power line, and that have different modes of power conversion between the first and second DC power sources and the power line. Configured to operate by selecting one of the modes of operation;
The controller is
Means for calculating total input / output power from the entire first and second DC power sources to the power line based on a deviation between a voltage detection value of the power line and a voltage command value;
Means for switching a power distribution ratio between the first and second DC power sources in response to a change in the operation mode;
Means for setting respective power command values of the first and second DC power sources according to the overall input / output power and the power distribution ratio;
For each of the first and second DC power supplies, a duty ratio for controlling output from the DC power supply is determined based on a deviation of a current detection value with respect to a current command value obtained by dividing the power command value by an output voltage. Control means for computing;
Based on the logical operation of the first and second control pulse signals obtained according to the pulse width modulation by comparing the duty ratio calculated for each of the first and second DC power supplies and a carrier wave, Signal generating means for generating on / off control signals of a plurality of switching elements,
The control apparatus for a power converter, wherein the logical operation is different between the plurality of operation modes. A control device for a power converter connected between a power line connected to a load and the first and second DC power supplies, and including a plurality of switching elements,
The power converter is configured to operate by selecting one operation mode from among a plurality of operation modes having different modes of power conversion between the first and second DC power supplies and the power line. ,
The controller is
Means for calculating total input / output power from the entire first and second DC power sources to the power line based on a deviation between a voltage detection value of the power line and a voltage command value;
Means for switching a power distribution ratio between the first and second DC power sources in response to a change in the operation mode;
Means for setting respective power command values of the first and second DC power sources according to the overall input / output power and the power distribution ratio;
For each of the first and second DC power supplies, a duty ratio for controlling output from the DC power supply is determined based on a deviation of a current detection value with respect to a current command value obtained by dividing the power command value by an output voltage. Control means for computing;
The plurality of switchings based on the first and second control pulse signals obtained in accordance with the pulse width modulation by comparing the duty ratio calculated for each of the first and second DC power sources and a carrier wave. Signal generating means for generating an on / off control signal of the element,
The plurality of switching elements are:
A first switching element electrically connected between a first node and the power line;
A second switching element electrically connected between a second node and the first node;
A third node electrically connected to the negative terminal of the second DC power supply and a third switching element electrically connected between the second node;
A fourth switching element electrically connected between the negative terminal of the first DC power source and the third node;
The power converter is
A first reactor electrically connected between the second node and a positive terminal of the first DC power supply;
A control device for a power converter, further comprising: a second reactor electrically connected between the first node and a positive terminal of the second DC power supply. The plurality of operation modes are:
A first mode in which the first and second DC power supplies perform DC voltage conversion in parallel with the power line by ON / OFF control of the first to fourth switching elements;
The third switching element is fixed on and the first, second, and fourth switching elements are turned on / off, so that the first and second DC power supplies are connected in series with the power line. A second mode for performing DC voltage conversion between
The power distribution ratio is variably set in accordance with the state of the first and second DC power supplies in the first mode, while the first and second DC power supplies in the second mode. It is fixed to a ratio based on the voltage, the control device for a power converter according to claim 2 Symbol placement. The control means calculates first and second duty ratios by current feedback control based on the current command value for the first and second DC power supplies in the first mode,
The control means performs current feedback control based on the current command value in one of the first and second DC power supplies in the second mode, while the first and second DC power supplies. On the other side of the power supply, the first and second duty ratios are calculated by executing feedforward control based on the output voltage of the DC power supply and the voltage command value without executing the current feedback control.
In each of the first and second modes, the signal generating means includes a first control pulse signal obtained by comparing the first duty ratio and the first carrier wave, the second duty ratio, and Generating the on / off control signals of the first to fourth switching elements based on a second control pulse signal obtained by comparing the second carrier wave;
In the second mode, the phase difference between the first carrier wave and the second carrier wave is the pulse transition timing of the first control pulse signal and the pulse transition of the second control pulse signal. The power converter control device according to claim 3 , wherein the controller is variably controlled in accordance with the calculated first and second duty ratios so as to match the timing. In both the first and second modes, the phase difference between the first carrier wave and the second carrier wave is the pulse transition timing of the first control pulse signal and the second control pulse. 5. The control device for a power converter according to claim 4 , wherein the control device is variably controlled according to the calculated first and second duty ratios so as to match the transition timing of the pulse of the signal. The plurality of operation modes are:
The on-off control of the first to fourth switching elements performs DC voltage conversion between one of the first and second DC power supplies and the power line, and the first and second switching elements. A third mode in which the other DC power source of the DC power source is kept electrically disconnected from the power line;
On and off of the first to fourth switching elements are fixed, and one of the first and second DC power supplies is electrically connected to the power line, while the first and second DC power supplies A fourth mode in which the other is maintained electrically disconnected from the power line;
A fifth mode in which the first to fourth switching elements are fixed on and off, and the first and second DC power supplies are connected in parallel to the power line;
A sixth mode in which the first to fourth switching elements are fixed on and off and the first and second DC power supplies are connected in series to the power line; and
The power distribution ratio, in the third mode, the set as one the entire output power only by the output from the DC power source is secured, according to any one of claims 3-5 Power converter control device. The plurality of operation modes are:
A first mode in which the first and second DC power supplies perform DC voltage conversion in parallel with the power line by ON / OFF control of the first to fourth switching elements;
The on-off control of the first to fourth switching elements performs DC voltage conversion between one of the first and second DC power supplies and the power line, and the first and second switching elements. A third mode in which the other DC power source of the DC power source is kept electrically disconnected from the power line, and
The power distribution ratio is variably set in accordance with the state of the first and second DC power supplies in the first mode, whereas in the third mode, only by the output from the one power supply. The control device for a power converter according to claim 2 , wherein the control device is set so that the entire input / output power is secured. The control means calculates first and second duty ratios by current feedback control based on the current command value for the first and second DC power supplies in the first mode,
In the third mode, the control means calculates one duty ratio of the first and second duty ratios by current feedback control based on the current command value for the one DC power supply,
In the first mode, the signal generating means includes a first control pulse signal obtained by comparing the first duty ratio and the first carrier wave, the second duty ratio and the second carrier wave. Generating the on / off control signals of the first to fourth switching elements based on a second control pulse signal obtained by comparison with
The signal generating means is based on the first or second control pulse signal obtained by comparing the calculated one duty ratio with the first or second carrier wave in the third mode. so as to constitute the two portions of the pair of the first to fourth switching elements are commonly off control, to generate the on-off control signal of the fourth switching element from the first, according to claim 7, wherein Power converter control device. Means for setting a circulating power value for charging / discharging between the first DC power source and the second DC power source;
The power command value of the first DC power supply is limited within a power range set according to the state of the first DC power supply according to the total input / output power, the power distribution ratio, and the circulating power value. Is set to
Wherein the power command value of the second DC power supply, the set by the overall input power subtracting the power command value of the first DC power source, control of the power converter of claim 2 Symbol placement apparatus. The control means performs current feedback control based on the current command value in one of the first and second DC power supplies in the second mode, while the first and second DC power supplies. On the other side of the power supply, the current feedback control is not executed, and feedforward control based on the output voltage of the DC power supply and the voltage command value is executed, whereby the first and second DC power supplies for each of the first and second DC power supplies. And calculating the second duty ratio,
4. The control device for a power converter according to claim 3 , wherein the signal generation unit generates an on / off control signal for the first to fourth switching elements according to the pulse width modulation based on the first and second duty ratios. 5. . The power command value of the predetermined DC power supply provided corresponding to a predetermined DC power supply of the first and second DC power supplies and set according to the power distribution ratio is set to the state of the predetermined DC power supply. The control apparatus of the power converter of any one of Claims 1-10 further provided with the 1st protection means for restrict | limiting within the 1st electric power range set according to. A second protection means for limiting the overall input / output power within a second power range set according to a state of the first and second DC power supplies;
12. The control device for a power converter according to claim 11 , wherein the first protection means is provided corresponding to each of the remaining DC power supplies excluding one of the first and second DC power supplies.

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