JP6043774B2 - Grid-connected inverter device and distributed power supply system including the same - Google Patents

Description

  The present invention relates to a grid-connected inverter device, and more particularly to a technique for reducing loss.

  In recent years, in order to prevent global warming, distributed power systems using fuel cells, solar power, wind power, and the like as energy sources have become widespread. Generally, the system is provided with a grid-connected inverter device so that the power obtained from the above-mentioned energy source can be supplied to a commercial system. In order to promote effective use of energy and downsizing of the system, a grid-connected inverter device with low loss is desired.

  On the other hand, Patent Document 1 discloses a conventional technique for reducing the loss of an inverter device.

Japanese Patent Publication No. 7-8145

  In Patent Document 1, when the load current is detected and the polarity of the load voltage and the polarity of the load current are positive or negative, the two switch elements (Q1 and Q2 or A technique is described in which the on / off operation of Q3 and Q4) is prevented so as to reduce the loss due to the switching element on / off. However, Patent Document 1 does not consider the influence of ripples included in the load current, that is, the output current of the single-phase bridge inverter circuit 1.

  Here, an overview of the grid-connected inverter device is touched. The grid-connected inverter device outputs pulsed power (hereinafter referred to as pulse power) from the bridge circuit, smoothes it with a filter circuit, and outputs it to the commercial system. For this reason, generally, the output current of the bridge circuit includes a ripple as shown in FIG.

  Next, consider a case where the technique of Patent Document 1 described above is applied to a grid-connected inverter device. In general, the power factor of the grid-connected inverter device is 0.95 to 1.0, and there is almost no phase difference between the grid voltage and grid current. Therefore, for example, in a small current region near the zero cross of the system voltage, the output current of the bridge circuit is small, and the polarity of the output current may repeat a positive polarity and a negative polarity every switching cycle due to the ripple described above. That is, in a small current region near the zero cross, the polarity determination of the output current becomes unstable, and the switching operation of the bridge circuit may become unstable. In this case, for example, the system current may be distorted, or the loss due to ON / OFF of the switch element (hereinafter referred to as switching loss) may not be sufficiently reduced.

  Accordingly, an object of the present invention is to provide a system interconnection inverter device that realizes a stable switching operation in a small current region near the zero crossing of the system voltage and has a small loss.

In order to achieve the above object, the present invention has a bridge circuit, a filter circuit, and a control means for controlling the bridge circuit, and the power obtained from a DC power supply is supplied through the bridge circuit and the filter circuit in this order. In the grid-connected inverter device for supplying to the power supply, the bridge circuit includes a plurality of switches, and the control unit controls the switches so that distortion of a current flowing through the commercial system is suppressed; An operation mode switching means for switching between the mode 1 and the mode 2, a switching signal generating means for driving the plurality of switches, and the bridge. Current detection means for detecting the output current of the circuit and the amplitude value of the ripple of the output current of the bridge circuit; To comprise a ripple detecting unit, the amplitude value of the average output current Ii of the bridge circuit (ave) | Ii (ave) | is compared with the ripple amplitude Ir of the output current Ii before Kivu ridge circuit, Ir is When it is larger than | Ii (ave) |, the grid interconnection inverter device which is set as the mode 1 is used.

  According to the present invention, the grid interconnection has a bridge circuit, a filter circuit, and a control means for controlling the bridge circuit, and supplies power obtained from a DC power source to a commercial system through the bridge circuit and the filter circuit in this order. In the inverter device, the bridge circuit is composed of a plurality of switches, and the control means suppresses distortion of the current flowing through the commercial system from the period before the predetermined period until the predetermined period elapses with the zero cross point of the commercial system voltage as a base point. Mode 1 for controlling the switch as described above and mode 2 for controlling the switch so as to reduce the loss of the switch in a period from the elapse of a predetermined period to the next predetermined period. A stable switching operation can be realized in a small current region, and a grid-connected inverter device with little loss can be provided.

  Alternatively, according to the present invention, a system that has a bridge circuit, a filter circuit, and a control unit that controls the bridge circuit, and supplies power obtained from a DC power supply to a commercial system through the bridge circuit and the filter circuit in this order. In the interconnected inverter device, the bridge circuit includes a first switch, a second switch, a third switch, and a fourth switch, and both ends of the DC power supply are connected in series to the first switch and the second switch. Are connected to both ends of the first switch, the third switch and the fourth switch connected in series, the connection point of the first switch and the second switch, and the third switch and the fourth switch. The connection point is connected to the filter circuit, the filter circuit is connected to the commercial system, and the control means has a period from a predetermined period before the predetermined period elapses after the zero cross point of the commercial system voltage. When the commercial system voltage is positive, the first switch is turned on, the second switch is turned off, and the third and fourth switches are complementary PWM. When the commercial system voltage is negative, In mode 1 in which the third switch is turned on and the fourth switch is turned off and the first and second switches are complementary PWM, the voltage of the commercial system is positive during the period from the elapse of a predetermined period to the next predetermined period. When the first switch is turned on, the second switch and the third switch are turned off and the fourth switch is PWMed. When the commercial system voltage is negative, the third switch is turned on. By having the fourth switch and the mode 2 in which the first switch is turned off and the second switch is PWM, a stable switching operation is realized in a small current region near the zero cross of the system voltage, It is possible to provide a loss less system interconnection inverter device.

  Further, according to the present invention, there is provided a bridge circuit in which both ends of the first switch and the second switch connected in series, and both ends of the third switch and the fourth switch connected in series are connected to both ends of the DC power supply. The first and third switches are composed of IGBTs and diodes to reduce conduction loss, and the second and fourth switches are composed of MOSFETs and diodes to enable high-speed switching and reduce switching loss. Can be made.

BRIEF DESCRIPTION OF THE DRAWINGS The block diagram of the single phase system connection inverter apparatus by the 1st Example of this invention. FIG. 1 is a voltage / current waveform diagram when the operation mode switching means 120 is not used (operation waveform based on a PWM signal in mode 2). The enlarged view of the period (B) and (F) of FIG. The enlarged view of the period (C) and (E) of FIG. FIG. 1 is a voltage-current waveform diagram when the operation mode switching means 120 is not used (operation waveform by a PWM signal in mode 1). FIG. 2 is a voltage / current waveform diagram when the operation mode switching means 120 is used in FIG. 1. The control means of the single phase system connection inverter apparatus by the 1st example of the present invention. FIG. 1 is a voltage / current waveform diagram when the operation mode switching means 120 is used (when the output current is larger than that in FIG. 7). The block diagram of the single phase system connection inverter apparatus by the 2nd Example of this invention. The voltage-current waveform figure in the 2nd Example of this invention. FIG. 11 is a detailed diagram of the switching operation of FIG. 10. The block diagram of the distributed power supply system by the 3rd Example of this invention. The block diagram of a grid connection inverter apparatus.

  Hereinafter, the present invention will be described with reference to examples.

  FIG. 1 is a circuit configuration diagram of a single-phase grid-connected inverter device 1 according to a first embodiment of the present invention.

  A main circuit of the single-phase grid-connected inverter device 1 will be described. Connected to both ends of the DC power supply 10 are a bypass capacitor 60, a series connection body of the switching elements 20 and 21, and a series connection body of the switching elements 22 and 23. Free wheel diodes 40 to 43 are connected in parallel to the switching elements 20 to 23, respectively. A series connection body of an inductor 70, a capacitor 61, and an inductor 71 is connected between the connection point of the switching elements 20 and 21 and the connection point of the switching elements 22 and 23. Both ends of the capacitor 61 are connected to the commercial system 80. In the present embodiment, IGBTs are used for the switching elements 20 and 22. Further, MOSFETs are used for the switching elements 21 and 23.

  The switching element 20 and the diode 40, the switching element 21 and the diode 41, the switching element 22 and the diode 42, the switching element 23 and the diode 43 described above are respectively the first switch, the second switch, the third switch, and the fourth switch. Configure the switch. The bypass capacitor 60 and the first to fourth switches constitute a single-phase bridge circuit that converts DC power output from the DC power supply 10 into AC pulsed power.

  The output terminal of the single-phase bridge circuit is a connection point between the first switch and the second switch, and a connection point between the third switch and the fourth switch. The inductor 70, the capacitor 61, and the inductor 71 constitute a filter circuit that smoothes the AC pulse power output from the single-phase bridge circuit and outputs it to the system. The input ends of the filter circuit are both ends of a series connection body composed of an inductor 70, a capacitor 61, and an inductor 71, and the output ends are both ends of the capacitor 61.

  The single-phase grid-connected inverter device 1 is provided with a control means 100 for controlling the main circuit described above. The control means 100 will be described. The voltage detection means 102 is connected to both ends of the bypass capacitor 60 in order to detect the voltage on the input side of the single-phase grid-connected inverter device 1, and the voltage detection means 104 is connected to both ends of the capacitor 61 in order to detect the voltage on the output side. Is connected. The voltage detection means 102 and 104 are connected to the grid connection control means 108. Further, a current sensor 90 is inserted on the positive electrode side of the DC power supply 10 to detect the current on the input side, and the switching elements 20 and 21 on the output side of the single-phase bridge circuit to detect the current on the output side. A current sensor 91 is inserted between the connection point and the inductor 70. The current sensor 90 and the current sensor 91 are connected to the current detection unit 101 and the current detection unit 103, respectively. The current detection unit 101 and the current detection unit 103 are connected to the grid connection control unit 108. PWM generation means 106 and 107 are connected to the grid interconnection control means 108. The PWM generation unit 106 is connected to the switching unit 109 and the ripple detection unit 111. The PWM generation unit 107 is connected to the switching unit 109. The switching unit 109 is connected to the drive unit 105. The drive means 105 is connected to the gates of the switching elements 20-23. Further, the voltage detection means 102 and 104 are connected to the ripple detection means 111. The current detection unit 103 and the ripple detection unit 111 are connected to the comparison unit 110. The comparison unit 110 is connected to the switching unit 109.

  The control means 100 performs the following control so that the single-phase system interconnection inverter device 1 outputs desired power to the commercial system 80. The grid connection control means 108 detects the voltage and current on the input side and output side from the voltage detection means 102 and 104 and the current detection means 101 and 103, and flows through the system voltage Vac of the commercial system 80 and the commercial system 80. The modulation factor is calculated so that the current Iac has the same phase (power factor 0.95 to 1.0), and is output to the PWM generators 106 and 107. The PWM generation means 106 and 107 generate a PWM signal for switching the switching elements 20 to 23 by comparing the modulation factor output from the grid connection control means 108 with the carrier signal, and via the switching means 109. Output to the drive means 105. The drive means 105 amplifies the PWM signal output from the PWM generation means 106 or 107, and drives the gates of the switching elements 20-23.

  Here, the switching unit 109 and the comparison unit 110 described above constitute an operation mode switching unit 120. The operation mode switching unit 120 has a function of transmitting either the PWM generation unit 106 or the PWM signal generated by the PWM generation unit 107 to the drive unit 105 based on a method described later. The operation mode switching means 120 and the ripple detection means 111 are means related to the present invention.

  The operation of this embodiment will be described below. In order to clarify the operation and effect of the present invention, the operation mode switching means 120 and the ripple detection means 111 according to the present invention will be described separately when they are used and when they are used.

(Operation when the operation mode switching means 120 and the ripple detection means 111 are not used)
First, FIG. 2 shows operation waveforms when the switching elements 20 to 23 are switched by the PWM signal generated by the PWM generation means 106 (hereinafter referred to as mode 1) without using the operation mode switching means 120 in FIG. . Vac is the voltage of the commercial system 80 (hereinafter referred to as system voltage). Ii is an output current of the single-phase bridge circuit (Ii is a waveform in which a sine wave is superimposed on the second triangular wave from the top in FIG. 2). Ii (ave) is an average current of Ii, which is substantially equal to the system current Iac, which is a current smoothed by the filter circuit. Vg (20) to Vg (23) are gate drive voltages of the switching elements 20 to 23. Hereinafter, a description will be given with reference to FIG.

  When Vac is a positive half cycle, Vg (20) is continuously on and Vg (21) is continuously off. Vg (22) and Vg (23) are alternately turned on / off (hereinafter referred to as complementary PWM). When Vac is a negative half cycle, Vg (20) and Vg (21) perform complementary PWM. Vg (22) is continuously turned on, and Vg (23) is continuously turned off.

  In addition, between Vg (20) and Vg (21) and between Vg (22) and Vg so that switching elements 20 and 21 and switching elements 22 and 23 connected in series are not turned on. A dead time is provided between (23) and (23). When the above-described dead time is not provided, there is a possibility that both the switching elements 20 and 21 or the switching elements 22 and 23 are turned on due to the difference between the turn-on time and the turn-off time in the switching element. In this case, a large short-circuit current flows through the loop of the DC power supply 10 → the switching element 20 → the switching element 21 or the DC power supply 10 → the switching element 22 → the switching element 23, and the switching element may be destroyed.

  Hereinafter, the operation will be described divided into periods (A) to (F) in the figure. The period (A) is a period in which the system voltage Vac is positive and the output current Ii of the single-phase bridge circuit is positive. Periods (B) and (F) are periods in which Vac is positive and the upper limit of Ii is positive and the lower limit of Ii is negative due to ripple. The difference between the periods (B) and (F) is only the difference in whether the average current Ii (ave) of Ii increases or decreases over time, and there is no difference related to the switching operation of the single-phase bridge circuit. . Based on the same idea, the period (D) is a period in which Vac is negative and Ii is negative. Periods (C) and (E) are periods in which Vac is negative and the upper limit of Ii is positive and the lower limit of Ii is negative due to ripple. The difference between the periods (C) and (E) is only the difference in whether the average current Ii (ave) of Ii increases or decreases over time, and there is no difference related to the switching operation of the single-phase bridge circuit. .

  First, the periods (A), (B), and (F) in which the system voltage Vac is positive will be described.

  In the period (A), when the switching element 22 is off and the switching element 23 is on, a current flows through the path of the DC power source 10 → the switching element 20 → the inductor 70 → the capacitor 61 → the inductor 71 → the switching element 23. Next, when the switching element 22 is on and the switching element 23 is off, a current flows through the path of the switching element 20 → the inductor 70 → the capacitor 61 → the inductor 71 → the diode 42. As described above, no current flows through the switching element 22 in the period (A), and there is no problem even if the switching element 22 is not turned on.

  In order to explain the operations in the periods (B) and (F), FIG. 3 shows operation waveforms in which the periods are enlarged. This will be described with reference to FIG. In FIG. 3, t0 is the time when Ii = 0 when the output current Ii becomes negative to positive, t1 is the time when the output current Ii> 0 and Ii starts to transition from rising to falling, and t2 Is the time when Ii = 0 when the output current Ii becomes negative from positive, t3 is the time when the output current Ii <0 and Ii starts to transition from falling to rising, and t4 is the time when the output current Ii is This is the time when Ii = 0 from negative to positive.

  First, the operation from time t0 to t2 when the output current Ii of the bridge inverter circuit is positive will be described. At time t0-t1, when the switching element 22 is off and the switching element 23 is on, a current flows through the path of the DC power source 10 → the switching element 20 → the inductor 70 → the capacitor 61 → the inductor 71 → the switching element 23. Next, at time t1-t2, when the switching element 23 is off and the switching element 22 is on, a current flows through the path of the switching element 20 → the inductor 70 → the capacitor 61 → the inductor 71 → the diode 42. That is, the operation in the period (A) is the same until the time t0-t2 when Ii is positive.

  Next, the operation from time t2 to t4 when the output current Ii of the single-phase bridge circuit becomes negative will be described. At t2-t3, when the switching element 22 is on and the switching element 23 is off, a current flows through the path of the capacitor 61 → the inductor 70 → the diode 40 → the switching element 22 → the inductor 71. Next, at time t3 to t4, when the switching element 22 is turned off and the switching element 23 is turned on, a current flows through the path of the capacitor 61 → the inductor 70 → the diode 40 → the DC power supply 10 → the switching element 23 and the diode 43. That is, there is a period during which current flows through the switching element 22 between time t2 and t4 when Ii becomes negative.

  Next, returning to FIG. 2, the periods (C) to (E) in which the system voltage Vac is negative will be described.

  In the period (D), when the switching element 20 is off and the switching element 21 is on, a current flows through the path of the DC power source 10 → the switching element 22 → the inductor 71 → the capacitor 61 → the inductor 70 → the switching element 21. Next, when the switching element 20 is on and the switching element 21 is off, a current flows through the path of the switching element 22 → the inductor 71 → the capacitor 61 → the inductor 70 → the diode 40. As described above, no current flows through the switching element 20 during the period (D), and there is no problem even if the switching element 20 is not turned on.

  In order to explain the operations in the periods (C) and (E), FIG. 4 shows operation waveforms in which the periods are enlarged. This will be described with reference to FIG. In FIG. 4, t0 ′ is the time when Ii = 0 when the output current Ii becomes negative from positive, and t1 ′ is the time when the output current Ii <0 and Ii starts to transition from falling to rising. , T2 ′ is the time when Ii = 0 when the output current Ii becomes negative to positive, t3 ′ is the time when the output current Ii> 0 and Ii starts to transition from rising to falling, and t4 ′. Is the time when Ii = 0 when the output current Ii changes from positive to negative.

  First, the operation from time t0 ′ to t2 ′ when the output current Ii of the single-phase bridge circuit is negative will be described. At time t0′-t1 ′, when the switching element 20 is off and the switching element 21 is on, current flows through the path of the DC power source 10 → the switching element 22 → the inductor 71 → the capacitor 61 → the inductor 70 → the switching element 21. Flowing. Next, at time t1′-t2 ′, when the switching element 21 is off and the switching element 20 is on, a current flows through the path of the switching element 22 → the inductor 71 → the capacitor 61 → the inductor 70 → the diode 40. That is, the operation is similar to that in the period (D) until time t0′-t2 ′ where Ii is negative.

  Next, the operation from time t2 ′ to t4 ′ when the output current Ii of the single-phase bridge circuit becomes positive will be described. When the switching element 20 is on and the switching element 21 is off at t2′-t3 ′, a current flows through the path of the capacitor 61 → the inductor 71 → the diode 42 → the switching element 20 → the inductor 70. Next, at time t3′-t4 ′, when switching element 20 is turned off and switching element 21 is turned on, a current flows through the path of capacitor 61 → inductor 71 → diode 42 → DC power supply 10 → switching element 21 and diode 41. That is, there is a period during which a current flows through the switching element 20 between time t2 ′ and t4 ′ when Ii becomes positive.

  As described above, when the switching elements 20 to 23 are switched by the mode 1 PWM signal without using the operation mode switching means 120 and the ripple detection means 111 in FIG. 1, the switching elements (period ( Since the switching element 22 in A) and the switching element 20) in the period (D) are switched, an extra switching loss occurs.

  On the other hand, the operation waveform when the switching elements 20 to 23 are switched by the PWM signal generated by the PWM generation means 107 (hereinafter referred to as mode 2) without using the operation mode switching means 120 and the ripple detection means 111 in FIG. Is shown in FIG. Hereinafter, a description will be given with reference to FIG.

  When Vac is a positive half cycle, Vg (20) is continuously turned on and Vg (21) and Vg (22) are continuously turned off. Then, Vg (23) is turned on / off (hereinafter referred to as PWM). When Vac is a negative half cycle, Vg (22) is continuously turned on, and Vg (20) and Vg (23) are continuously turned off. Vg (21) is PWMed. In this way, FIG. 5 does not perform complementary PWM performed between Vg (20) and Vg (21) and between Vg (22) and Vg (23) as compared with FIG. 2, but Vg (21) or Vg ( 23) is PWMed.

  Since no current flows through the switching element 22 during the period (A) and through the switching element 20 during the period (D), the output current Ii of the single-phase bridge circuit is as shown in FIG. It is the same. However, in the periods (B), (C), (E), and (F), distortion as shown in FIG. 5 may occur in Ii. This is because the current on the negative side of Ii shown in FIG. 2 cannot flow through the switching element 22 during the periods (B) and (F), and Ii is on the positive side. The cause is an increase. Similarly, in the periods (C) and (E), since the switching element 20 is turned off, the current on the positive side of Ii shown in FIG. 2 cannot flow to the switching element 20, and Ii is on the negative side. The cause is an increase. On the other hand, however, FIG. 5 has the effect that the switching frequency of the switching elements 20 and 22 is small and the switching loss is small compared to FIG.

(Operation when the operation mode switching means 120 and the ripple detection means 111 are used)
FIG. 6 shows operation waveforms when the operation mode switching means 120 and the ripple detection means 111 are used in FIG. Hereinafter, description will be made with reference to FIG. 1 and FIG. The operation mode switching means 120 discriminates the above-mentioned periods (A) to (F), and the PWM generation means 107 and the switching means so as to transmit the mode 2 PWM signal to the drive means 105 in the periods (A) and (D). 109 and the drive means 105 are connected, and in the periods (B), (C), (E), and (F), the PWM generation means 106, the switching means 109, and the drive are transmitted so as to transmit the mode 1 PWM signal to the drive means 105. The means 105 is connected.

  Next, a method in which the operation mode switching unit 120 determines the periods (A) to (F) will be described.

  First, the determination conditions are as follows. The operation mode switching means 120 compares the amplitude value | Ii (ave) | of the average output current Ii (ave) of the single-phase bridge circuit with the amplitude value Ir of the ripple of Ii, and Ir is | Ii (ave) | If larger, it is determined as periods (B), (C), (E), and (F) during which the polarity of the current is switched during the switching operation of the single-phase bridge circuit. On the other hand, when Ir is smaller than | Ii (ave) |, the operation mode switching unit 120 determines the periods (A) and (D).

  Here, a method for detecting the Ir by the ripple detecting means 111 will be described. For example, when the system voltage Vac is positive and the switching element 20 is turned on and the switching element 23 transitions from OFF to ON, the voltage Vpn of the DC power supply 10 is output to the output terminal of the single-phase bridge circuit, The voltage is applied to the inductor 70, the capacitor 61, and the inductor 71 that constitute the filter circuit. Accordingly, a voltage obtained by subtracting Vac from Vpn is applied to the inductors 70 and 71, and the current of the inductors 70 and 71, that is, the output current Ii of the single-phase bridge circuit increases. Half of the increased current value corresponds to the ripple amplitude Ir. Thus, Ir can be obtained from the calculation based on Vpn, the ON time of the switching element 23 (the ON time of the switching element 21 when Vac is negative), Vac, and the combined inductance of the inductors 70 and 71. . Therefore, the ripple detection means 111 receives Vpn from the voltage detection means 102, the ON time of the switching elements 21 and 23 from the PWM generation means 106, and the system voltage Vac from the voltage detection means 104, and those values and stored inductors. Ir is calculated based on the combined inductance of 70 and 71. This calculation is performed each time the switching element 21 or 23 is switched, and Ir is updated. The ripple detection unit 111 holds and outputs Ir before being updated while Ir is updated.

  On the other hand, the ripple amplitude value Ir can be detected by other methods. An example is shown in FIG. In FIG. 7, a ripple detection unit 112 is provided instead of the ripple detection unit 111 in FIG. The current detection means 103 is connected to the input of the ripple detection means 112, and the comparison means 110 is connected to the output. The ripple detection means 112 includes a high-pass filter that cuts off the frequency of a commercial system (50 Hz or 60 Hz) and a gun-line detection circuit. In general, the arc detection circuit is a circuit that extracts only the arc line when there is target information in the arc line in the time series of the electrical signal. When the value of the output current Ii of the single-phase bridge circuit is input from the current detection unit 103 to the ripple detection unit 112, the frequency of the commercial system is first cut by the above-described high-pass filter, so that only a ripple component is obtained. When this signal is input to the arcuate line detection circuit, only the arcuate line of the ripple component is extracted, that is, only the Ir amplitude value as shown in FIG. 6 is output. Further, Ir can be obtained by other methods and is not limited to the present embodiment.

  As shown in FIG. 6, the amplitude value Ir of the ripple thus obtained is not a constant value but takes a maximum value in the middle of the system voltage Vac from zero to peak or from peak to zero.

  From here, it returns to description of operation | movement of an operation mode switching means. The comparison unit 110 receives the amplitude value | Ii (ave) | of the average output current Ii (ave) of the single-phase bridge circuit from the current detection unit 103 and the ripple amplitude value Ir of Ii from the ripple detection unit 111. . Comparing means 110 compares | Ii (ave) | and Ir, and outputs Low when | Ii (ave) | is greater than Ir, and outputs High when vice versa. The switching unit 109 connects the PWM generation unit 107, the switching unit 109, and the drive unit 105 so that the mode 2 PWM signal is transmitted to the drive unit 105 when the output of the comparison unit 110 is Low, and the output of the comparison unit 110 is High. In this case, the PWM generation means 106, the switching means 109, and the drive means 105 are connected so as to transmit the PWM signal of mode 1 to the drive means 105.

  By these operations, in the small current region (periods (B), (C), (E), (F)) near the zero cross of the system voltage Vac, the switching elements 20 to 23 are switched by the complementary PWM signal of mode 1. Thus, distortion of the system current Iac can be suppressed. In the large current region (periods (A) and (D)), the switching elements 20 to 23 can be switched by the PWM signal of mode 2 having a smaller number of switching times than that of mode 1, thereby reducing the switching loss.

  FIG. 8 shows an operation waveform when the system current Iac is increased as compared with FIG. As described above, the ripple amplitude value Ir of the single-phase bridge circuit is determined from Vpn, the ON time of the switching elements 21 and 23, Vac, and the combined inductance of the inductors 70 and 71, and is independent of the system current Iac. Therefore, the amplitude value Ir of the ripple is the same value as in FIG. On the other hand, since Iac is increased and the average output current Ii (ave) of the single-phase bridge circuit is increased, time periods (A) and (D) in which | Ii (ave) | On the contrary, the periods (B), (C), (E), and (F) in which | Ii (ave) | is smaller than Ir are narrowed. As a result, the period for operating in mode 2 is widened, while the period for operating in mode 1 is conversely narrowed. That is, in this embodiment, in the period in which the amplitude value of the ripple is larger than the average output current amplitude value of the single-phase bridge circuit, centering on the zero cross of the system voltage, the mode operates within the period, and the outside of the period Then, it operates in mode 2. The period changes in inverse proportion to the magnitude of the output current.

  As described above, in this embodiment, by using the operation mode switching means 120 according to the present invention, the distortion of the system current near the zero cross of the system voltage is suppressed, and the switching frequency of the switching element is reduced to reduce the switching loss. Can be reduced. As a result, it is possible to obtain the effect of providing a single-phase grid-connected inverter device with little loss.

  In order to reduce the loss of the switching element, it is generally desirable to use an IGBT with a relatively low switching speed but a small conduction loss for the switching elements 20 and 22 having a small number of times of switching. For the switching elements 21 and 23 having a large number of times of switching, it is generally desirable to use MOSFETs that can switch at high speed and have low switching loss. Considering this, in this embodiment, the switching elements 20 and 22 are IGBTs, and the switching elements 21 and 23 are MOSFETs. However, IGBTs or MOSFETs may be used for all switching elements, and the present invention is not limited to this embodiment.

  In the present embodiment, the single-phase grid-connected inverter has been described. The above control is also performed in a three-phase grid-connected inverter equipped with a single-phase bridge circuit as a three-phase bridge circuit, a sensor that detects the voltage and current of the three-phase system, and a PWM generation means / ripple detection means that supports three phases. The same effect can be expected.

  FIG. 9 is a configuration diagram of the single-phase grid-connected inverter device 2 according to the second embodiment of the present invention. In FIG. 9, the same components as those in FIG.

  Differences between FIG. 9 and FIG. 1 will be described. Switching elements 30 to 33 are connected in parallel to the switching elements 20 to 23, respectively. That is, in this embodiment, the switching element 20, the diode 40 and the switching element 30, the switching element 21 and the diode 41 and switching element 31, the switching element 22 and the diode 42 and switching element 32, the switching element 23 and the diode 43 and the switching element 33. Respectively constitute a first switch, a second switch, a third switch, and a fourth switch. The gates of the switching elements 30 to 33 are connected to the control means 200. The control unit 200 has all the functions of the control unit 100 described in the first embodiment, and also has a function of driving the gates of the switching elements 30 to 33. MOSFETs are used for the switching elements 30 and 32. The switching elements 31 and 33 are IGBTs. In the present embodiment, the loss can be further reduced as compared with the first embodiment by including the switching elements 30 to 33.

  Next, the operation of this embodiment will be described. The operation waveform of each part is shown in FIG. Vg (30) and Vg (32) are gate drive voltages of the switching elements 30 and 32, respectively. The switching element 30 switches during the periods (C) and (E). Further, the switching element 32 switches during the periods (B) and (F). That is, the switching elements 30 and 32 are switched when the single-phase system interconnection inverter device 2 is in the mode 1. Vg (31) and Vg (33) are gate drive voltages of the switching elements 31 and 33, respectively. The switching elements 31 and 33 are driven by substantially the same signals as the switching elements 21 and 23, respectively, but there is a slight difference in the turn-off time.

  First, the effect of adding the switching elements 31 and 33 will be described. Details of the switching operation of the switching elements 21 and 31 are shown in FIG. Ic (31) is a current between the collector and emitter of the switching element 31. Id (21) is the current between the drain and source of the switching element 21. When the control means 200 turns on the switching elements 21 and 31 at time t0 ″ at the same time, a current flows through each element. Generally, when comparing IGBT with MOSFET, IGBT has lower resistance, so Ic (31) is more In other words, by adding the switching element 31, the conduction loss of the switching element 21 can be reduced, and the loss of the second switch constituted by the switching elements 21 and 31 and the diode 41 is reduced. it can.

However, in general, the IGBT is turned off slower than the MOSFET, and the IGBT has a larger turn-off loss than the MOSFET. Therefore, the control unit 200 turns off the gate of the switching element 31 before the switching element 21. After time t1 ″ when the gate of the switching element 31 is turned off, Ic (31) decreases and Id (21) increases. At time t2 ″, Ic (31) becomes zero. Id (21) is a current obtained by adding Ic (31) before time t1 ″. Since the switching element 21 is on during this time t1 ″ to t2 ″, the collector-emitter of the switching element 31 In the meantime, the ON voltage of the switching element 21 is applied. Generally, this ON voltage is about several volts, so that almost no switching loss occurs when the switching element 31 is OFF. Next, at time t3 ″, the switching element 21 21 gate is turned off.

  Between time t3 ″ and t4 ″, Id (21) decreases to zero, and switching loss occurs in the switching element 21. Since the switching element 21 in the present embodiment uses a MOSFET as in the first embodiment, the switching loss of the switching element 21 occurring between the times t3 ″ and t4 ″ is the same as in the first embodiment. is there.

  As described above, in this embodiment, the addition of the switching element 31 can reduce the conduction loss of the switching element 21, and the loss of the second switch composed of the switching elements 21 and 31 and the diode 41 can be reduced to the embodiment. 1 can be reduced.

  The time from the time t1 ″ at which the gate of the switching element 31 is turned off to the time t3 ″ at which the gate of the switching element 21 is turned off is desirably a sufficient time for the switching element 31 to be turned off. For example, the control unit 200 may be provided with an oscillator that emits a pulse having a time of about 10 nsec to 1 usec, and the time of the pulse may be used to set the time from t1 ″ to t3 ″.

  Further, the time necessary and sufficient for the switching element 31 to turn off is proportional to the current value that has flowed through the switching element 31 before time t1 ″, so that the current that has flowed through the switching element 31, that is, the output of the single-phase bridge circuit. It is desirable to change the pulse width emitted by the oscillator described above in proportion to the current Ii.

  Alternatively, the control means 200 is provided with a current detection means for detecting the current between the collector and the emitter of the switching element 31, and this current detection means detects the time when the current of the switching element 31 becomes zero and turns off the switching element 21. However, the same effect can be obtained.

  Furthermore, in the present embodiment, the switching elements 21 and 31 are turned on at the same time. In general, however, there is no significant difference in the turn-on speed between the IGBT and the MOSFET, so whichever is turned on first. Also good.

  Note that the switching operation of the switching elements 23 and 33 is performed by changing Vg (21), Vg (31), Ic (21), and Id (31) in FIG. 11 to Vg (23), Vg (33), and Ic ( 23) and Id (33), and the description thereof is omitted. By adding the switching element 33, the conduction loss of the switching element 23 can be reduced, and the loss of the fourth switch composed of the switching elements 23 and 33 and the diode 43 can be reduced.

  Next, the effect of adding the switching elements 30 and 32 will be described. As shown in FIG. 10, the switching element 30 switches only in the periods (C) and (E), and the switching element 32 switches only in the periods (B) and (F).

  The operation in the periods (B) and (F) when the switching element 32 is added will be described with reference to FIG. Here, Vg (22) in FIG. 3 is read as Vg (32).

  A current flows through the switching element 32 between time t1 and time t3 when Vg (32) is on. First, at time t1-t2, when the switching elements 23 and 33 are off and the switching element 32 is on, current flows through the path of the switching element 20 → the inductor 70 → the capacitor 61 → the inductor 71 → the diode 42 and the switching element 32. Flows. That is, since the switching element 32 is a MOSFET, a current can flow from the source to the drain. Thereby, the current flowing through the diode 42 is reduced, and the loss of the second switch constituted by the switching elements 22 and 32 and the diode 42 can be reduced.

  At time t2-t3, when the switching element 32 is on and the switching elements 23 and 33 are off, current flows through the path of the capacitor 61 → the inductor 70 → the diode 40 → the switching elements 22 and 32 → the inductor 71. That is, current is shunted by the switching elements 22 and 32, and the loss of the second switch constituted by the switching elements 22 and 32 and the diode 42 can be reduced.

  Next, operations in the periods (C) and (E) when the switching element 30 is added will be described with reference to FIG. 4 described above. Here, it is assumed that Vg (20) in FIG. 4 is read as Vg (30).

  A current flows through the switching element 30 from time t1 ′ to time t3 ′ when Vg (30) is on. First, at time t1′-t2 ′, when switching elements 21 and 31 are off and switching element 30 is on, switching element 22 → inductor 71 → capacitor 61 → inductor 70 → path of diode 40 and switching element 30 Current flows. That is, since the switching element 30 is a MOSFET, a current can flow from the source to the drain. Thereby, the current flowing through the diode 40 is reduced, and the loss of the first switch constituted by the switching elements 20 and 30 and the diode 40 can be reduced.

  At time t2'-t3 ', when switching element 30 is on and switching elements 21 and 31 are off, current flows through the path of capacitor 61 → inductor 71 → diode 42 → switching elements 20 and 30 → inductor 70. . That is, the current is shunted by the switching elements 20 and 30, and the loss of the first switch constituted by the switching elements 20 and 30 and the diode 40 can be reduced.

  As described above, this embodiment can further reduce the loss as compared with the first embodiment by including the switching elements 30 to 33. In particular, the switching elements 31 and 33 reduce the loss of the switching elements 21 and 23 that switch in a large current region (periods (A) and (D) in FIG. 10), and the effect of reducing the loss is great. As a result, also in the present embodiment, it is possible to obtain an effect of providing a single-phase grid-connected inverter device with little loss.

  In the present embodiment, the single-phase grid-connected inverter has been described. The above control is also performed in a three-phase grid-connected inverter equipped with a single-phase bridge circuit as a three-phase bridge circuit, a sensor that detects the voltage and current of the three-phase system, and a PWM generation means / ripple detection means that supports three phases. The same effect can be expected.

  FIG. 12 is a block diagram of a distributed power supply system according to a third embodiment of the present invention. 300 is the single-phase grid-connected inverter device 2 shown in the second embodiment. 11 is a solar cell panel.

  The solar cell panel 11 corresponds to the DC power source 10 in the first and second embodiments. 11 is not limited to a solar cell panel, but may be any DC voltage source. For example, the AC output of a fuel cell or wind power generator may be rectified to DC.

  Since the distributed power system 300 includes the single-phase grid-connected inverter device 1 described in the first embodiment or the single-phase grid-connected inverter device 2 described in the second embodiment, the power loss is higher than that of the conventional distributed power system. There is little heat generation. For this reason, a heatsink and a cooling fan can be made smaller than before, and a compact and low-loss distributed power supply system can be obtained.

  In addition, this invention is not limited to the Example mentioned above, Various modifications are included. For example, the above-described embodiments have been described in detail for easy understanding of the present invention, and are not necessarily limited to those having all the configurations described. Further, it is possible to add, delete, and replace other configurations for a part of the configuration of each embodiment.

  Further, each of the above-described configurations, functions, and the like may be realized by hardware by designing a part or all of them with, for example, an integrated circuit. Further, each of the above-described configurations, functions, and the like may be realized by software by a processor interpreting and executing a program that realizes each function. Information on programs that implement each function can be stored in a recording device such as a memory, a hard disk, or an SSD (Solid State Drive), or a recording medium such as an IC card, an SD card, or a DVD.

  In addition, the control lines and information lines are those that are considered necessary for the explanation, and not all the control lines and information lines on the product are necessarily shown. In practice, it can be considered that almost all the components are connected to each other.

1, 2, 301 Single-phase grid-connected inverter device 10 DC power supply 11 Solar cell panels 20-23, 30-33 Switching elements 40-43 Diode 60 Bypass capacitor 61 Capacitor 70, 71 Inductor 80 Commercial system 90, 91 Current sensor 100 , 200 Control means 101, 103 Current detection means 102, 104 Voltage detection means 105 Drive means 106, 107 PWM generation means 108 System interconnection control means 109 Switching means 110 Comparison means 111, 112 Ripple detection means 120 Operation mode switching means 300 Dispersion Power system

Claims (10)

In a grid-connected inverter device having a bridge circuit, a filter circuit, and a control means for controlling the bridge circuit, and supplying power obtained from a DC power source to a commercial system through the bridge circuit and the filter circuit in this order ,
The bridge circuit comprises a plurality of switches,
The control means includes
A mode 1 for controlling the switch so as to suppress distortion of a current flowing through the commercial system; and a mode 2 for controlling the switch so as to reduce the loss of the switch. The mode 1 and the mode 2 Operation mode switching means for switching between,
Switching signal generating means for driving the plurality of switches;
Current detecting means for detecting an output current of the bridge circuit;
Ripple detection means for detecting the amplitude value of the ripple of the output current of the bridge circuit,
Comparing the ripple amplitude Ir of the output current Ii before Kivu ridge circuit, Ir is | | the amplitude value of the average output current Ii of the bridge circuit (ave) | Ii (ave) Ii (ave) | than A grid-connected inverter device characterized in that the mode 1 is set to a large case. In the grid interconnection inverter device according to claim 1,
The bridge circuit is configured by connecting both ends of a first switch and a second switch connected in series, and both ends of a third switch and a fourth switch connected in series to both ends of a DC power supply,
In the mode 1, when the voltage of the commercial system is positive, the control means turns on the first switch, turns off the second switch, and complements the third and fourth switches. When the voltage of the commercial system is negative, the third switch is turned on and the fourth switch is turned off, and the first and second switches are complementary PWMed,
In the mode 2, when the voltage of the commercial system is positive, the control means turns on the first switch, turns off the second switch and the third switch, and turns off the fourth switch. PWM is performed, and when the voltage of the commercial system is negative, the third switch is turned on, the fourth switch and the first switch are turned off, and the second switch is PWMed. Grid-connected inverter device. In a grid-connected inverter device having a bridge circuit, a filter circuit, and a control means for controlling the bridge circuit, and supplying power obtained from a DC power source to a commercial system through the bridge circuit and the filter circuit in this order ,
The bridge circuit includes a first switch, a second switch, a third switch, and a fourth switch,
Both ends of the DC power supply are connected to both ends of a first switch and a second switch connected in series, and both ends of a third switch and a fourth switch connected in series,
The connection point of the first switch and the second switch, the connection point of the third switch and the fourth switch are connected to the filter circuit,
The filter circuit is connected to the commercial system,
The control means includes
When the voltage of the commercial system is positive, the first switch is turned on, the second switch is turned off, and the third and fourth switches are complementary PWM, and the commercial system voltage is negative. And mode 1 for turning on the third switch and turning off the fourth switch and performing complementary PWM on the first and second switches;
When the voltage of the commercial system is positive, the first switch is turned on, the second switch and the third switch are turned off and the fourth switch is PWMed, and the voltage of the commercial system is negative. At the time of having the third switch on, the fourth switch and the mode 2 for turning off the first switch and PWMing the second switch,
An operation mode switching means for switching between the mode 1 and the mode 2;
Switching signal generating means for driving the plurality of switches;
Current detecting means for detecting an output current of the bridge circuit;
Ripple detection means for detecting the amplitude value of the ripple of the output current of the bridge circuit,
Comparing the ripple amplitude Ir of the output current Ii before Kivu ridge circuit, Ir is | | the amplitude value of the average output current Ii of the bridge circuit (ave) | Ii (ave) Ii (ave) | than A grid-connected inverter device characterized in that the mode 1 is set to a large case. In the grid connection inverter apparatus of Claim 2 or 3,
The ripple detection means calculates the amplitude value of the ripple of the output current in the bridge circuit based on the voltage of the DC power supply, the voltage of the commercial system, and the ON time of the second switch or the fourth switch. A grid-connected inverter device having a function of calculating. In the grid connection inverter apparatus in any one of Claims 1 thru | or 3,
The ripple detection means comprises a high-pass filter that cuts off the frequency of the commercial system and an arc detection circuit. In the grid connection inverter apparatus of Claim 2 or 3,
The grid-connected inverter device, wherein the first and third switches are composed of IGBTs and diodes, and the second and fourth switches are composed of MOSFETs and diodes. In the grid connection inverter apparatus of Claim 2 or 3,
The first switch and the third switch are composed of IGBT, MOSFET and diode, and in the mode 1, the control means causes the MOSFET of the first switch to perform complementary PWM with the second switch, and A grid-connected inverter device, wherein a MOSFET of a third switch is complementary PWM with the fourth switch. In the grid connection inverter apparatus of Claim 2 or 3,
The second switch and the fourth switch are composed of an IGBT, a MOSFET, and a diode, and the control unit is configured to make the IGBT of the second and fourth switches more than the MOSFET of the second and fourth switches. A grid-connected inverter device that is turned off 10 n to 1 u seconds ahead.   A distributed power supply system comprising the grid-connected inverter device according to any one of claims 1 to 8.   The distributed power supply system according to claim 9, wherein the DC power supply of the grid-connected inverter device is a solar battery panel.