JP6038282B2 - Voltage conversion circuit - Google Patents

Description

  The present invention relates to a voltage conversion circuit.

  As a basic configuration of the voltage conversion circuit, for example, there is a configuration described in Patent Document 1 below. This voltage converter circuit is a booster circuit (boost chopper) that releases magnetic energy stored in the reactor by arbitrarily adjusting and switching the switch element to obtain boosted voltage by obtaining an induced voltage. . In the above boost chopper, all the magnetic energy for boosting is stored in a single reactor, so that there is a problem that the reactor becomes very large in order to prevent magnetic saturation of the core.

  Therefore, in order to solve the above problem, for example, a technique described in Patent Document 2 below has been proposed. In the technique described in Patent Document 2, a transformer magnetically coupled to a magnetic component constituting a voltage conversion circuit is used, and magnetic saturation of the core is prevented while canceling out DC magnetization caused by DC current. The size is reduced.

Japanese Unexamined Patent Publication No. 2003-111390 (5th page, FIG. 2) Japanese Patent No. 4098299 (page 20, FIG. 2)

  According to the voltage conversion circuit according to Patent Document 2 described above, since the current directions of the primary winding and the secondary winding of the transformer are opposite, the DC magnetization in the core is canceled and the core is magnetically saturated. Since it becomes difficult, a small winding (coil) can be employed, and there is an advantage that the voltage conversion circuit can be miniaturized.

  However, in this conventional voltage conversion circuit, focusing on the smoothing capacitor, the current ripple of the smoothing capacitor (hereinafter simply referred to as the capacitor current ripple) is equal to the current ripple of the reactor. The reactor current ripple is uniquely determined, and therefore the capacitor current ripple is also determined.

  Since the capacitor current ripple affects the design of the smoothing capacitor, the inductance of the reactor is determined due to restrictions on the capacitor current ripple, and there is a problem that the design of the reactor is restricted.

  The present invention has been made to solve the above-described problems, and while reducing the magnetic saturation of the reactor, it is possible to increase the degree of freedom in designing the reactor and the transformer and to reduce the size of the device. An object is to provide a voltage conversion circuit.

The voltage conversion circuit according to the present invention has a low voltage terminal, a high voltage terminal, and a reference terminal, a DC voltage between the low voltage terminal and the reference terminal, and between the high voltage terminal and the reference terminal. A magnetic cancellation type transformer in which a primary winding and a secondary winding are magnetically coupled via a core and are reversely wound at a turns ratio of 1: 1. a first reactor for energy storage, which is provided independently of the connected to the primary winding of the transformer the primary winding is connected to the secondary winding of the transformer such two A second reactor for energy storage provided independently of the secondary winding; a first switching element for controlling energization from the primary winding to the reference terminal; and the high voltage from the primary winding. A second switch element for controlling energization to the terminal and the secondary winding to A third switch element for controlling the energization of the quasi-terminal, and a fourth switch element for controlling the energization of the the high voltage terminal from said secondary winding, said first reactor and said second Either one of the reactors is provided on the proximity side of the high-voltage terminal of the transformer, and inductances of the primary winding and the secondary winding of the transformer are the first reactor and the second reactor, respectively. It is set in the range of 1 to 10 times the inductance of the reactor.

According to the voltage conversion circuit of the present invention, the reactor current ripple is determined by the combination of the inductance of the reactor and the transformer, and the inductance of the reactor is selected without being restricted by the reactor current ripple. can do. As a result, the magnetic saturation of the reactor can be alleviated and the design freedom of the reactor and the transformer can be expanded, and each inductance is selected and designed so that the volume of the reactor and the transformer in the device is minimized. This makes it possible to reduce the size and cost of the device.

It is a circuit diagram which shows the structure of the voltage conversion circuit in Embodiment 1 of this invention. It is explanatory drawing which shows the 1st operation state for demonstrating the pressure | voltage rise operation of the voltage converter circuit in Embodiment 1 of this invention. It is explanatory drawing which shows the 2nd operation state for demonstrating the pressure | voltage rise operation of the voltage converter circuit in Embodiment 1 of this invention. It is a graph which shows the relationship between the ratio of the reactor of the voltage converter circuit in Embodiment 1 of this invention, and the inductance of a transformer, and the current ripple of a smoothing capacitor. It is a graph which shows the correlation with the ratio of the inductance of the reactor of a voltage conversion circuit and transformer inductance in Embodiment 1 of this invention, and the volume of a reactor or a transformer. It is a circuit diagram which shows the modification of the voltage conversion circuit in Embodiment 1 of this invention. It is a circuit diagram which shows the structure of the voltage converter circuit in Embodiment 2 of this invention. It is a top view which shows the structural example of the transformer of the voltage conversion circuit in Embodiment 2 of this invention. It is a circuit diagram which shows the structure of the voltage converter circuit used as a reference example. It is explanatory drawing which shows the 1st operation state for demonstrating the pressure | voltage rise operation of the voltage converter circuit of a reference example. It is a time chart which shows the waveform in the 1st operation state in FIG. It is explanatory drawing which shows the 2nd operation state for demonstrating the pressure | voltage rise operation of the voltage converter circuit of a reference example. It is a time chart which shows the waveform in the 2nd operation state in FIG.

  First, in order to facilitate understanding of the contents of the present invention, a voltage conversion circuit as a reference example will be described first based on FIGS. 9 to 13. Next, the contents of the embodiment of the present invention will be described.

FIG. 9 is a circuit diagram showing a configuration of a voltage conversion circuit as a reference example of the present invention.
This voltage conversion circuit includes smoothing capacitors 5 and 6, a reactor 7, a transformer 8, and four switch elements 12, 14, 16 and 18. A smoothing capacitor 5 is connected between the low voltage terminal 2 and the reference terminal (GND) 4, and a smoothing capacitor 6 is connected between the high voltage terminal 3 and the reference terminal 4.

  A first end of a reactor 7 is connected to the low voltage terminal 2, and a pair of third and fourth T-type circuits connected in parallel to each other are connected between the second end of the reactor 7 and the high voltage terminal 3. Is provided. The third T-type circuit includes the primary winding 9 of the transformer 8 and the two switch elements 12 and 14. The fourth T-type circuit includes the secondary winding 10 of the transformer 8 and the two switch elements 16 and 18.

  In this case, the reactor 7 is for accumulating magnetic energy for boosting. The first ends of the primary winding 9 and the secondary winding 10 of the transformer 8 are connected to the reactor 7 in common. The transformer 8 is of a magnetic canceling type in which the primary winding 9 and the secondary winding 10 are magnetically coupled via the core 11 and are reversely wound with each other at a turn ratio of 1: 1.

  In the third T-type circuit, the collector and the emitter of the switch element 12 are connected between the primary winding 9 and the reference terminal 4, and the switch element 14 is connected between the primary winding 9 and the high voltage terminal 3. The emitter and collector are connected. Similarly, in the fourth T-type circuit, the collector and the emitter of the switch element 16 are connected between the secondary winding 10 and the reference terminal 4, and between the secondary winding 10 and the high voltage terminal 3. The emitter and collector of the switch element 18 are connected to each other.

  Further, diodes 13, 15, 17, and 19 are individually connected in parallel to the switch elements 12, 14, 16, and 18, respectively. A gate signal for on / off control is applied to the gate of each switch element 12, 14, 16, 18 from a control device (not shown).

  Next, the operation of the voltage conversion circuit of the above reference example, particularly the boosting operation here, will be described with reference to FIGS.

  FIG. 10 shows the flow of current in each part of the voltage conversion circuit when only the switch element 12 of the third T-type circuit is turned on and the primary winding 9 of the transformer 8 is energized. FIG. 12 shows the current flow in each part of the voltage conversion circuit when only the switch element 16 of the fourth T-type circuit is turned on and the secondary winding 10 of the transformer 8 is energized.

  In the state shown in FIG. 10, a low-level gate signal is input to each of the gates of the switch elements 14 and 18 and the switch element 16 and is turned off. An on / off gate signal G12 is input to the gate of the switch element 12.

  Here, since the DC voltage Vi from the DC power source (not shown) is input to the low voltage terminal 2, when the high level gate signal G12 is input to the gate of the switch element 12, the transformer is turned on. Excitation current I1 flows through the primary winding 9 of the eight. The exciting current I1 flows through the low voltage terminal 2, the reactor 7, the primary winding 9, the switch element 12, and the reference terminal 4. While the gate signal G12 of the switch element 12 is at the high level, the exciting current I1 gradually increases. When the gate signal G12 is inverted to a low level and the switch element 12 is turned off, the exciting current I1 '(indicated by a broken line in FIG. 10) decreases and finally becomes zero. This exciting current I 1 ′ flows to the high voltage terminal 3 through the primary winding 9 and the diode 15.

  Thus, when the excitation currents I1 and I1 'described above flow through the primary winding 9 of the transformer 8, the excitation current I2 is generated in the secondary winding 10 based on the mutual induction action. This excitation current I2 flows to the high voltage terminal 3 via the diode 19. Here, when the circuit constants of the third and fourth T-type circuits are substantially equal, the excitation current I2 of the secondary winding 10 is substantially the same shape as the excitation currents I1 and I1 ′ as shown in FIG. And have substantially the same value based on the turns ratio (1: 1). The smoothing capacitor 6 is charged by the excitation current I2 and the excitation current I1 '(broken line). As a result, the boosted DC voltage Vo is output to the high voltage terminal 3.

  Next, in the state shown in FIG. 12, a low-level gate signal is input to each of the gates of the switch elements 14 and 18 and the switch element 12 and is turned off. An on / off gate signal G16 is input to the gate of the switch element 16.

  Here, since the DC voltage Vi from the DC power source (not shown) is input to the low voltage terminal 2, when the high level gate signal G16 is input to the gate of the switch element 16, the transformer is turned on. Excitation current I3 flows through 8 secondary windings 10. This exciting current I3 flows via the low voltage terminal 2, the reactor 7, the secondary winding 10, the switch element 16, and the reference terminal 4. While the gate signal G16 of the switch element 16 is at the high level, the exciting current I3 gradually increases. When the gate signal G16 is inverted to a low level and the switch element 16 is turned off, the exciting current I3 '(indicated by a broken line in FIG. 12) decreases and finally becomes zero. This exciting current I 3 ′ flows to the high voltage terminal 3 through the secondary winding 10 and the diode 19.

  Thus, when the excitation currents I3 and I3 'described above flow through the secondary winding 10 of the transformer 8, the excitation current I4 is generated in the primary winding 9 based on the mutual induction action. This excitation current I4 flows to the high voltage terminal 3 via the diode 15. Here, when the circuit constants of the third and fourth T-type circuits are substantially equal, as shown in FIG. 13, the excitation current I4 of the primary winding 9 is substantially the same shape as the excitation currents I3 and I3 ′. It has changing characteristics and occurs at substantially the same value based on the turns ratio (1: 1). Then, the smoothing capacitor 6 is charged by the excitation current I4 and the excitation current I3 '(broken line), and as a result, the boosted DC voltage Vo is output to the high voltage terminal 3.

  As described above, the switching elements 12 and 16 are alternately turned on and off in a time-sharing manner by the gate signals G12 and G16 given from the control device (not shown), so that the high voltage terminal 3 is boosted, A stabilized DC voltage Vo is output.

  Here, the respective currents I1, I1 ′, I4 flowing through the primary winding 9 of the transformer 8 are collectively referred to as ic1, and the respective I2, I3, I3 ′ flowing through the secondary winding 10 are also described. Is collectively represented as ic2, the reactor current ib flowing through the reactor 7 is the sum of the primary winding current ic1 and the secondary winding current ic2. Since the reactor current ib changes in the period Ton during which the switch elements 12 and 16 are on, assuming that the reactor current ripple generated in the reactor 7 is Ib, Ib is given by the following formula (1).

  Further, when the circuit equation is established from FIG. 10, the following equation (2) is derived from the voltage equation of the path through which the primary winding current ic1 (I1) flows and the voltage equation of the path through which the secondary winding current ic2 (I2) flows. Is obtained. However, the magnetic coupling degree of the transformer 8 is 1, the inductance of the primary winding 9 and the inductance of the secondary winding 10 are both equal, and the inductance of the reactor 7 is Lb.

  If equation (2) is modified, the following equation (3) is obtained.

  From the above equations (1) and (3), the reactor current ripple Ib is given by the following equation (4).

  Here, the reactor current ripple Ib represented by the equation (4) is equal to the capacitor current ripple of the smoothing capacitor 5.

  As described above, in the voltage conversion circuit shown in FIG. 9 given as a reference example, the current directions of the primary winding 9 and the secondary winding 10 of the transformer 8 are opposite to each other. Since the core 11 is less likely to be magnetically saturated, a small winding (coil) can be employed, and the voltage conversion circuit can be downsized.

  However, in the voltage conversion circuit of this reference example, focusing on the smoothing capacitor 5, since the smoothing capacitor current ripple is equal to the reactor current ripple Ib, the inductance of the reactor 7 is determined when the step-up ratio is determined as shown in Equation (4). The reactor current ripple Ib is uniquely determined by Lb, and therefore the capacitor current ripple is also determined.

  Since the capacitor current ripple affects the design of the smoothing capacitor 5, there is a problem that the inductance Lb of the reactor 7 is determined due to restrictions on the capacitor current ripple and the design of the reactor 7 is restricted.

Embodiment 1 FIG.
Next, a voltage conversion circuit according to Embodiment 1 of the present invention will be described.
FIG. 1 is a circuit diagram showing a configuration of a voltage conversion circuit according to Embodiment 1 of the present invention. Components corresponding to or corresponding to those of the voltage conversion circuit shown as the reference example in FIG. 9 are denoted by the same reference numerals.

  The voltage conversion circuit of the first embodiment includes smoothing capacitors 5 and 6, two first and second reactors 21 and 22, a transformer 8, and four switch elements 12, 14, 16, and 18. A smoothing capacitor 5 is connected between the low voltage terminal 2 and the reference terminal 4 (GND), and a smoothing capacitor 6 is connected between the high voltage terminal 3 and the reference terminal 4.

  A pair of first and second T-type circuits connected in parallel with each other are provided between the low voltage terminal 2 and the high voltage terminal 3. The first T-type circuit includes a first reactor 21, a primary winding 9 of the transformer 8, and two switch elements 12 and 14. The second T-type circuit includes a second reactor 22, a secondary winding 10 of the transformer 8, and two switch elements 16 and 18.

  In this case, the first and second reactors 21 and 22 are for accumulating magnetic energy for boosting, and the first ends of the first and second reactors 21 and 22 are common to the low voltage terminal 2. It is connected. The transformer 8 is of a magnetic canceling type in which the primary winding 9 and the secondary winding 10 are magnetically coupled via the core 11 and are reversely wound with each other at a turn ratio of 1: 1. In addition, each of the switch elements 12, 14, 16, and 18 is an IGBT (Insulated Gate Bipolar Transistor) here.

  In the first T-type circuit, the second end of the first reactor 21 is connected to the connection terminal 28 on the first end side of the primary winding 9 of the transformer 8. Between the connection terminal 32 on the second end side of the primary winding 9 and the reference terminal 4, the collector and the emitter of the switch element 12 that controls energization from the primary winding 9 to the reference terminal 4 are connected. . Between the connection terminal 32 on the second end side of the primary winding 9 and the high voltage terminal 3, the emitter-collector of the switch element 14 that controls energization from the primary winding 9 to the high voltage terminal 3 is connected. ing.

  Similarly, in the second T-type circuit, the second end of the second reactor 22 is connected to the connection terminal 29 on the first end side of the secondary winding 10 of the transformer 8. Between the connection terminal 33 on the second end side of the secondary winding 10 and the reference terminal 4, the collector-emitter of the switch element 16 that controls energization from the secondary winding 10 to the reference terminal 4 is connected. ing. Between the connection terminal 33 on the second end side of the secondary winding 10 and the high voltage terminal 3, there is a gap between the emitter and the collector of the switch element 18 that controls energization from the secondary winding 10 to the high voltage terminal 3. It is connected.

  Further, the diodes 13, 15, 17, and 19 are individually connected in parallel to the switch elements 12, 14, 16, and 18, respectively. A gate signal for on / off control is given to a gate of each switch element 12, 14, 16, 18 from a control device (not shown). Reference numeral 41 denotes a common connection point of the switch elements 12 and 14 of the first T-type circuit, and reference numeral 42 denotes a common connection point of the switch elements 16 and 18 of the second T-type circuit.

  Thus, in the first embodiment, the first reactor 21 is connected via the connection terminal 28 of the primary winding 9 of the transformer 8, and the second reactor 22 is connected to the secondary winding of the transformer 8. 10 connection terminals 29 are connected. Here, each of the reactors 21 and 22 is not shared as a part of the primary winding 9 or the secondary winding 10 of the transformer 8, and is physically connected to each of the windings 9 and 10 of the transformer 8. In addition, it is divided into an independent structure.

  In the claims, the first switch element is the switch element 12, the second switch element is the switch element 14, the third switch element is the switch element 16, and the fourth switch element is the switch element 18. , Respectively.

  The operation of the voltage conversion circuit having such a configuration, particularly the boosting operation here will be described with reference to FIGS.

  FIG. 2 shows the flow of current in each part of the voltage conversion circuit when only the switch element 12 of the first T-type circuit is turned on and the primary winding 9 of the transformer 8 is energized. FIG. 3 shows a current flow in each part of the voltage conversion circuit when only the switch element 16 of the second T-type circuit is turned on and the secondary winding 10 of the transformer 8 is energized.

  In the state shown in FIG. 2, a low-level gate signal is input to each of the gates of the switch elements 14 and 18 and the switch element 16 and is turned off. An on / off gate signal is input to the gate of the switch element 12.

  Here, since the DC voltage Vi from the DC power source (not shown) is input to the low voltage terminal 2, when a high level gate signal is input to the gate of the switch element 12, the transformer 8 is turned on. An exciting current I5 flows through the primary winding 9 of the first coil. This exciting current I5 flows through the low voltage terminal 2, the first reactor 21, the primary winding 9, the switch element 12, and the reference terminal 4. While the gate signal of the switch element 12 is at the high level, the exciting current I5 gradually increases. When the gate signal of the switch element 12 is inverted to a low level, the excitation current I5 '(indicated by a broken line in FIG. 2) decreases and finally becomes zero. This exciting current I <b> 5 ′ flows to the high voltage terminal 3 through the first reactor 21, the primary winding 9, and the diode 15.

  When the excitation currents I5 and I5 'flow through the primary winding 9 of the transformer 8, the excitation current I6 is generated in the secondary winding 10 based on the mutual induction action. This excitation current I6 flows to the high voltage terminal 3 via the diode 19. Here, when the circuit constants of the first and second T-type circuits are substantially equal, the excitation current I6 of the secondary winding 10 is substantially equal to the excitation currents I5 and I5 ′ as in the case shown in FIG. And have substantially the same value based on the turn ratio (1: 1). The smoothing capacitor 6 is charged by the excitation current I6 and the excitation current I5 '. As a result, the boosted DC voltage Vo is output to the high voltage terminal 3.

  Next, in the state shown in FIG. 3, a low-level gate signal is input to each of the gates of the switch elements 14 and 18 and the switch element 12 and is turned off. An on / off gate signal is input to the gate of the switch element 16.

  Here, since the DC voltage Vi from the DC power source (not shown) is input to the low voltage terminal 2, when a high level gate signal is input to the gate of the switch element 16, the transformer 8 is turned on. Excitation current I7 flows through the secondary winding 10. That is, the exciting current I7 flows through the low voltage terminal 2, the second reactor 22, the secondary winding 10, the switch element 16, and the reference terminal 4. While the gate signal of the switch element 16 is at the high level, the exciting current I7 gradually increases. When the gate signal of the switch element 16 is inverted to a low level and the switch element 16 is turned off, the exciting current I7 '(indicated by a broken line) decreases and finally becomes zero. This exciting current I 7 ′ flows to the high voltage terminal 3 through the second reactor 22, the secondary winding 10, and the diode 19.

  When the excitation currents I7 and I7 'flow through the secondary winding 10 of the transformer 8, the excitation current I8 is generated in the primary winding 9 based on the mutual induction action. This excitation current I8 flows to the high voltage terminal 3 via the diode 15. Here, when the circuit constants of the first and second T-type circuits are substantially equal, the excitation current I8 of the primary winding 9 is substantially equal to the excitation currents I7 and I7 ′ as in the case shown in FIG. And have substantially the same value based on the turns ratio (1: 1). The smoothing capacitor 6 is charged by the excitation current I8 and the excitation current I7 ', and as a result, the boosted DC voltage Vo is output to the high voltage terminal 3.

  In this way, the switch elements 12 and 16 are alternately turned on and off in a time-sharing manner by the gate signal supplied from the control device (not shown), so that the high voltage terminal 3 is boosted and stabilized. DC voltage Vo is output.

  Here, the currents I5, I5 ′, I8 flowing through the primary winding 9 of the transformer 8 are collectively referred to as ic3, and the currents I6, I7, I7 ′ flowing through the secondary winding 10 are also described. Are collectively referred to as ic4, the current flowing through the first reactor 21 is equal to ic3, and the current flowing through the second reactor 22 is equal to ic4.

  Now, if the sum of the currents flowing through the first and second reactors 21 and 22 is expressed as an input current ie, the input current ie is divided into a current ic3 flowing through the primary winding 9 and a current ic4 flowing through the secondary winding 10. Is the sum of Since the input current ie changes in the period Ton during which the switch elements 12 and 16 are on, assuming that the ripple of the input current ie (hereinafter simply referred to as input current ripple) is Ie, Ie is expressed by the following equation (5). Given.

  Further, when the circuit equation is established from FIG. 2, the following equation (6) is derived from the voltage equation of the path through which the primary winding current ic3 (I5) flows and the voltage equation of the path through which the secondary winding current ic4 (I6) flows. Is obtained. However, here, the magnetic coupling degree of the transformer 8 is 1, the inductance of the primary winding 9 and the inductance of the secondary winding 10 are both equal to Lc, and the inductances of both reactors 21 and 22 are both equal to Ld.

  When the equation (6) is solved for (dic3 / dt) and (dic4 / dt), the following equation (7) is obtained.

  From the above equations (5) and (7), the input current ripple Ie is given by the following equation (8).

Here, the input current ripple Ie expressed by Equation (8) is equal to the capacitor current ripple of the smoothing capacitor 5.

Equation (8) shows that the input current ripple Ie according to the first embodiment, that is, the capacitor current ripple Ie, is the inductance Ld of the first and second reactors 21 and 22, the primary winding 9 and the secondary winding of the transformer 8. It is determined by the inductance Lc of the line 10.

  Here, paying attention to the relationship between the inductance Ld of each of the reactors 21 and 22 and the inductance Lc of the primary winding 9 and the secondary winding 10 of the transformer 8, the ratio of the two Lc and Ld is expressed as an inductance ratio P. Then, the following equation (9) is defined.

From the above equations (8) and (9), the capacitor current ripple Ie is given by the following equation.

  Here, from equation (10), the capacitor current ripple Ie when the inductance Lc of the primary winding 9 and the secondary winding 10 of the transformer 8 is zero is normalized as 1, and the inductance ratio P (= Lc) described above is normalized. When the correlation between / Ld) and the capacitor current ripple Ie is obtained, the graph shown in FIG. 4 is obtained. As can be seen from FIG. 4, as the inductance ratio P (= Lc / Ld) increases, the capacitor current ripple Ie tends to decrease.

  From the relationship shown in FIG. 4, if the transformer 8 is further provided for the first and second reactors 21 and 22 and the inductance ratio P is made larger than the case of only the first and second reactors 21 and 22. Although the advantage that the capacitor current ripple Ie can be reduced is obtained, the external shapes of the reactors 21 and 22 and the transformer 8 are not always the same.

  Therefore, when the capacitor current ripple Ie is constant, the reactors 21 and 22 and the transformer 8 are actually designed, the transformer 8 is omitted, and only the first and second reactors 21 and 22 are provided. FIG. 5 shows the result of examining how the volume of the entire device changes when the transformer 8 having the predetermined inductance Lc is provided from the state with reference to the volume.

  In FIG. 5, the total volume of the first and second reactors 21 and 22 (hereinafter referred to as the reactor volume) is Vd, the volume of the transformer 8 (hereinafter referred to as the transformer volume) is Vc, and the reactor volume Vd and the transformer The total volume of the vessel volume Vc (hereinafter referred to as the device volume) is Vl (= Vd + Vc), and when the transformer 8 is not provided (therefore, when Lc is zero, that is, when the reactor ratio P = 0) The relationship of each volume of Vd, Vc, and Vl when the reactor volume Vd at the time is normalized as 10 is shown.

  In order to reduce the voltage conversion circuit while minimizing the generation of the capacitor current ripple Ie, the reactor volume Vd when the transformer 8 is not provided (that is, when the inductance ratio P = 0) is used as a reference. It is desirable that the device volume Vl when the transformer 8 is provided is small.

  Here, as can be seen from FIG. 5, if a transformer 8 is newly provided for the first and second reactors 21 and 22 and the inductance ratio P is made larger than zero, each winding 9, 10 of the transformer 8. As a minimum structural volume is required to obtain the predetermined inductance Lc, the device volume Vl increases.

  Subsequently, when the inductance ratio P is increased, the transformer volume Vc gradually increases along with this, but each reactor volume Vd is remarkably reduced. When the inductance ratio P = 1, the device volume Vl is the initial value. It became equal to 10.

  Furthermore, when the inductance ratio P is increased, the device volume Vl takes a minimum value and then increases. This is due to the fact that the decrease in the reactor volume Vd is slowed and the increase in the transformer volume Vc is increased. The inductance ratio P was 10 when the device volume Vl reached the initial value of 10 again.

  From the above, in order to reduce the size of the voltage conversion circuit while suppressing the generation of the capacitor current ripple Ie as small as possible, the selection range of the inductance ratio P (= Lc / Ld) is 1 to 10, and therefore each of the reactors 21 and 22 is selected. It is desirable to select the inductance Lc of the primary winding 9 and the secondary winding 10 of the transformer 8 in a range of 1 to 10 times the inductance Ld of the transformer. In particular, by setting the selection range of the inductance ratio P to 3 to 6, the device volume Vl can be reduced to a minimum level of about 6 with respect to the initial value of 10, and thus is a more preferable selection range.

  As described above, FIG. 4 shows the relationship between the required capacitor current ripple Ie of the smoothing capacitor 5 and the inductance ratio P. Further, FIG. 5 shows the relationship between the device volume Vl and the inductance ratio P. Therefore, the inductance Ld of the first and second reactors 21 and 22 that provides the required capacitor current ripple Ie of the smoothing capacitor 5, the primary winding 9 of the transformer 8, while minimizing the device volume Vl as much as possible. The value of the inductance Lc of the secondary winding 10 can be specifically determined. Thereby, since it becomes possible to perform optimal structure design with respect to the 1st, 2nd reactors 21 and 22 and the transformer 8, the effect that size reduction and cost reduction of a power converter device can be achieved compared with the past. Is obtained.

  As described above, in the voltage conversion circuit according to the first embodiment, the first and second reactors 21 and 22 divided into two and the transformer 8 are arranged in combination, and the primary winding of the transformer 8 is arranged. By selecting the inductance Lc of the line 9 and the secondary winding 10 within a range of 1 to 10 times the inductance Ld of each reactor 21, 22, each reactor 21 and 22 is relaxed while alleviating the magnetic saturation. , 22 and the design of the transformer 8 can be expanded, and the apparatus can be reduced in size and cost.

  The present invention is not limited to the circuit configuration of the first embodiment shown in FIG. 1, and for example, as shown in FIG. 6, the primary winding 9 and both switch elements 12, It is good also as a structure which has arrange | positioned the 1st reactor 21 between the 14 common connection points 41. FIG. Although not shown, the same effect as described above can be obtained even if the second reactor 22 is arranged between the secondary winding 10 of the transformer 8 and the common connection point 42 of both switch elements 16 and 18. It is done.

Embodiment 2. FIG.
FIG. 7 is a circuit diagram showing a configuration of the voltage conversion circuit according to the second embodiment, and components corresponding to or corresponding to those in the circuit diagram of the first embodiment shown in FIG. 1 are denoted by the same reference numerals.

  When manufacturing equipment, the ease of manufacture and assembly is an important factor. This is directly linked to component costs and production efficiency, and improvements are always required. Therefore, in the second embodiment, consideration should be given to the arrangement relationship between the first and second reactors 21 and 22 and the transformer 8 and the winding structure of the primary winding 9 and the secondary winding 10 of the transformer 8. Then, the structure which improved the ease of manufacture or an assembly is demonstrated.

  The features of the second embodiment are as follows. That is, the first ends of the primary winding 9 and the secondary winding 10 of the transformer 8 are made common and connected to the low voltage terminal 2 as one connection terminal 31, and the second end of the primary winding 9 is connected. The first reactor 21 is disposed and connected between the end connection terminal 32 and the common connection point 41 of both switch elements 12, 14, the second end connection terminal 33 of the secondary winding 10 and both switches The second reactor 22 is arranged and connected between the common connection points 42 of the elements 16 and 18. Other configurations are the same as those of the first embodiment shown in FIG.

  The structure of the transformer 8 in such a configuration is shown in FIG. In the transformer 8 of FIG. 8, a winding 34 is wound around a toroidal core 11, and a first end of a wire 35 is connected to the middle of the winding 34 via a tap 30. Two ends serve as connection terminals 31 for connection to the low voltage terminal 2. Further, both ends of the winding 34 are a connection terminal 32 on the primary winding 9 side and a connection terminal 33 on the secondary winding 10 side, respectively.

  Next, correspondence between the transformer of FIG. 8 and the circuit diagram of FIG. 7 will be described. The portion of the winding 34 from one connection terminal 32 to the intermediate tap 30 in FIG. 8 is the primary winding 9 in FIG. 7, and the portion of the winding 34 from the other connection terminal 33 to the intermediate tap 30 is It becomes the secondary winding 10 of FIG. The connection terminal 31 is connected to the low voltage terminal 2 of FIG. 7, the connection terminal 32 on the primary winding 9 side is connected to the first end of the first reactor 21, and the connection terminal on the secondary winding 10 side. 33 is connected to the first end of the second reactor 22.

According to the configuration of the second embodiment, the primary winding 9 and the secondary winding 10 can be integrated and configured simply by winding the winding 34 around the core 11 of the transformer 8. Therefore, wiring connection points can be reduced, and the effect of simplifying the assembly process and improving production efficiency can be obtained. Moreover, since the primary winding 9 and the secondary winding 10 of the transformer 8 can be formed in one step, an effect of reducing the manufacturing cost can be obtained.
The operation as the voltage conversion circuit and the accompanying effects are all the same as in the first embodiment.

  In addition, the shape of the core 11 of the transformer 8 in this Embodiment 2 is not limited to the toroidal, and it is needless to say that it can be applied to any shape as long as it has the same function in operation. Yes.

  In the first and second embodiments described above, the IGBT is described as the switching elements 12, 14, 16, and 18 of the circuit configuration. However, the present invention is not limited to this, and switching elements such as MOSFETs are used. It goes without saying that it can be done.

  In the first and second embodiments, the coupling degree of the transformer 8 is described as 1. However, the present invention is not limited to this, and the same effect can be obtained even in a circuit configuration having a coupling degree of 1 or less. Needless to say.

  Further, in each of the first and second embodiments described above, the case where the DC power source is connected between the low voltage terminal 2 and the reference terminal 4 to perform the step-up operation has been described as an example, but the step-down operation can be performed. . For the step-down operation, a DC power source is connected between the high voltage terminal 3 and the reference terminal 4, and the switch elements 12 and 16 are both turned off by a gate signal given from a control device (not shown). The switch elements 14 and 18 are alternately turned on and off in a time division manner, whereby a stabilized DC voltage Vi stepped down to the low voltage terminal 2 is obtained.

Embodiment 3 FIG.
In the first and second embodiments, the switch elements 12, 14, 16, 18 and the diodes 13, 15, 17, 19 of the circuit configuration are mainly composed of silicon (Si) such as IGBT and MOSFET. did.

  In recent years, in addition to those formed of silicon as such switching elements and diodes, those formed of wide band gap semiconductors having a larger band gap than silicon have been put into practical use. In this case, the material of the wide band gap semiconductor can be composed of, for example, silicon carbide (SiC), a gallium nitride-based material, or diamond.

  Since switch elements and diode elements formed of such wide band gap semiconductors have high voltage resistance and high allowable current density, the switch elements and diodes can be miniaturized. By using a diode or a diode, a semiconductor module incorporating these elements can be miniaturized.

Therefore, in the third embodiment, silicon carbide (SiC) is applied as a material for the switch elements 12, 14, 16, 18 in particular.
Other configurations are the same as those of the first embodiment shown in FIG.

  Generally, in a high voltage circuit exceeding 600 V, for example, a switching element made of silicon carbide can suppress a switching loss smaller than a switching element made of silicon such as IGBT. The switching frequency of the element can be increased.

  For example, in the voltage conversion circuit of the first embodiment (FIG. 1), when the capacitor current ripple Ie is kept constant, if the switching frequency is increased, the inductance Ld of each of the reactors 21 and 22 and the transformer are proportionally increased. 8, the inductance Lc of the primary windings 9 and 10 can be reduced. This is a very effective method for downsizing the voltage conversion circuit. That is, high frequency driving of the switch element is required.

  However, the iron cores used in the reactors 21 and 22 and the transformer 8 generate a loss (iron loss) due to the ripple of current flowing through the windings, and the iron loss is higher as the ripple frequency is higher even with the same ripple amount. growing.

  Here, the frequency characteristics of the iron core differ depending on the material, and the iron loss generated differs even when the same ripple is applied. Furthermore, since the iron loss has a dependency relationship with the inductance, the relationship between the reactors 21 and 22, the iron core material of the transformer 8, and the inductances Lc and Ld is a factor that greatly affects the miniaturization of the voltage conversion circuit.

  That is, when the switch elements 12, 14, 16, and 18 made of silicon carbide are applied to the voltage conversion circuit, for example, when the reactors 21 and 22 and the transformer 8 are driven at a high frequency to reduce the size, for example, a transformer 8 is made of a material such as ferrite that has low loss even at high frequencies, and the inductance ratio P (= Lc / Ld) is increased. That is, the inductance Lc of each of the windings 9 and 10 of the transformer 8 is changed to the reactor 21, By configuring so as to be larger than the inductance Ld of 22, the distribution of the iron loss in the reactors 21 and 22 and the transformer 8 can be changed, and the overall iron loss can be suppressed small.

As described above, in the third embodiment, the first and second reactors 21 and 22 divided into two and the transformer 8 are arranged in combination, and the primary winding 9 and two of the transformer 8 are arranged. By selecting the inductance Lc of the secondary winding 10 in the range of 1 to 10 times the inductance Ld of each of the reactors 21 and 22, in addition to being able to realize downsizing and cost reduction, silicon carbide By switching the switch elements 12, 14, 16, and 18 configured as described above at a high frequency, heat generation can be suppressed and the cooler can be made smaller. Therefore, the voltage conversion circuit can be further downsized than the case of the first embodiment. Cost reduction is possible.
Since other operations and effects are the same as those of the first embodiment, detailed description thereof is omitted here.

  Note that the present invention is not limited to the configurations of the first to third embodiments described above, and the first to third embodiments can be freely combined or implemented within the scope not departing from the gist of the present invention. The configurations of the first to third embodiments can be appropriately changed or omitted.

Claims (7)

It has a low voltage terminal, a high voltage terminal, and a reference terminal, and performs voltage conversion between a DC voltage between the low voltage terminal and the reference terminal and a DC voltage between the high voltage terminal and the reference terminal. A voltage conversion circuit to perform,
A magnetic canceling transformer in which a primary winding and a secondary winding are magnetically coupled via a core and reversely wound at a turns ratio of 1: 1;
A first reactor for independent energy storage provided in the said primary winding is connected to the primary winding of the transformer,
A second reactor for independent energy storage provided in the said secondary windings being connected to the secondary winding of the transformer,
A first switch element for controlling energization from the primary winding to the reference terminal;
A second switch element for controlling energization from the primary winding to the high voltage terminal;
A third switch element for controlling energization from the secondary winding to the reference terminal;
A fourth switch element for controlling energization from the secondary winding to the high voltage terminal;
With
One of the first reactor and the second reactor is provided on the proximity side of the high voltage terminal of the transformer,
The voltage conversion circuit in which the inductance of the primary winding and the secondary winding of the transformer is set in a range of 1 to 10 times the inductance of the first reactor and the second reactor. The first reactor has a first end connected to the primary winding and a second end connected to a common connection point between the first switch element and the second switch element. Voltage conversion circuit. The first reactor has a first end connected to the secondary winding, and a second end connected to a common connection point of the third switch element and the fourth switch element. The voltage conversion circuit described. The first reactor has a first end connected to the second end of the primary winding, a second end connected to a common connection point with the first and second switch elements, and the second reactor. The reactor has a first end connected to the low voltage terminal, a second end connected to the first end of the secondary winding, and a first end of the primary winding connected to the low voltage terminal. The voltage conversion circuit according to claim 1, wherein a second end of the secondary winding is connected to a common connection point with the third and fourth switch elements. The first reactor has a first end connected to the low voltage terminal, a second end connected to the first end of the primary winding, and the second reactor has a first end connected to the second end. The second end of the primary winding is connected to a common connection point with the third and fourth switch elements, and the second end of the primary winding is connected to the first and second switches. 2. The voltage conversion circuit according to claim 1, wherein the voltage conversion circuit is connected to a common connection point with an element, and the first end of the secondary winding is connected to the low voltage terminal. 6. The voltage conversion circuit according to claim 1, wherein each of the switch elements is formed of a wide band gap semiconductor. The voltage conversion circuit according to claim 6 , wherein the wide band gap semiconductor is silicon carbide, a gallium nitride-based material, or diamond.

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