JP5993690B2 - Power converter and control method - Google Patents

Description

The present invention relates to a power conversion device that converts DC power into AC power.

  As background art of this technical field, there is JP-A-11-289769 (Patent Document 1). This publication provides “a power converter that uses the basic configuration of a three-phase AC power converter and can easily cope with a high-frequency single-phase load. U-X phase and V-Y Inverter circuit 3 having switching element arms for the three phases of the WZ phase and the WZ phase is used, and the outputs R and S of the switching element arms for the two U-X and VY phases are shared. In the half cycle period of the single-phase AC output, the upper arm is connected to the other AC output terminal B. The switching elements X and Y for two circuits of the lower arm are controlled to be turned on, and the switching elements U and V for two circuits of the upper arm and the switching elements U and V of the lower arm are controlled in the half cycle period following the cycle period. Each switching element Z By controlling the emissions, AC output terminal A, is described as obtaining the single-phase AC "to B (see Abstract).

JP 11-289769 A

The above prior art has the following problems. A power converter that uses a configuration of a three-phase AC power converter and can handle a high-frequency single-phase load is applied to a system that requires a high-frequency power source with a frequency of 10 kHz or more, such as a non-contact supply system. When used, the switching frequency of each switching element in one cycle is also 10 kHz or more.
When switching is performed at such a high frequency, the temperature rise due to generation loss of general-purpose switching elements such as IGBTs frequently used in power converters in recent years is considerably high.
In addition, in the conventional power converter, two positive and negative voltage pulses are generated in one cycle. When the load connected to the output of the power converter is an inductive load, either positive or negative Since the generation loss of the second pulse is larger, the temperature rise of the switching element that generates the second pulse is particularly larger than that of other switching elements. On the other hand, when the load connected to the output of the conventional power converter is a capacitive load, the first pulse generates more loss in both plus and minus, so the temperature of the switching element that generates the first pulse The increase is particularly large compared to other switching elements.
In both inductive load and capacitive load, if the switching element becomes hot due to the generated loss and the junction temperature of the semiconductor that constitutes the switching element exceeds the allowable value, normal operation may not be possible. There is sex. Therefore, it is necessary to sufficiently cool the switching element that generates a pulse with a large generation loss.
However, in order to enhance the cooling of the switching element, it may be necessary to increase the size of the cooling fan, the cooling fin, or the like, which causes a problem of increasing the size of the power conversion device main body and increasing the cost.
The present invention has been made in order to solve the above-described problems in a power conversion device that uses a configuration of a power conversion device for three-phase alternating current and can cope with a high-frequency single-phase load. An object of the present invention is to obtain a power conversion device having an increased cooling capacity of switching elements.

In order to solve the above problems, for example, the configuration described in the claims is adopted.
The present application includes a plurality of means for solving the above-described problems. For example, the present application includes an inverse conversion unit that converts DC power having a three-phase switching element arm into AC power, and the inverse conversion unit includes The outputs of the switching element arms of any two circuits of the three-phase switching element arms are connected in common to be one AC output terminal, and the output of the remaining one circuit switching element arm is the other AC output terminal. In the power converter configured to control conduction of one and the other switching elements independently in each of the two circuits and generate a single-phase alternating current between the one and the other AC output terminals, Of the pulses generated by the upper and lower arms of the two circuits during one cycle, the switching element arm that generates the pulse with the smaller generation loss is the Wherein the cooling among the location of the switching element arms are arranged in the most difficult places.

  ADVANTAGE OF THE INVENTION According to this invention, the local overheating of a switching element can be prevented and the cooling efficiency of a switching element can be improved.

It is an example of the block diagram of the non-contact supply system using the power converter device in Example 1. FIG. It is an example of the block diagram of the inverter circuit part in Example 1 and Example 2. FIG. It is the figure which showed the timing of the control action at the time of connecting an inductive load to the power converter device in Example 1, and an output waveform. It is the figure which showed the timing and output waveform of control operation at the time of connecting a capacitive load to the power converter device in Example 1. FIG. It is an example of the block diagram of the non-contact supply system using the power converter device in Example 2. FIG. It is the figure which showed the timing and output waveform of control operation of the power converter device in Example 2. FIG. It is the flowchart which showed control until the control apparatus in Example 1 and Example 2 supplies an ON control signal to each switching element of each phase.

  Hereinafter, examples will be described with reference to the drawings.

In this embodiment, an example of a non-contact supply system that supplies power to a high-frequency single-phase load in a non-contact manner using a power conversion device will be described.
FIG. 1 is an example of a configuration diagram of a non-contact supply system including an AC power supply 100, a power conversion device 110, and a high-frequency single-phase load 120. The AC power supply 100 is a commercial three-phase AC power supply or the like normally supplied from an electric power company or the like, and is connected to the converter circuit 111. The converter circuit 111 is configured by a three-phase bridge type rectifier circuit using a diode, for example, and functions to convert AC power supplied from the AC power supply 100 into DC. The electric power converted into the direct current is smoothed by passing through the smoothing capacitor 112 and input to the inverter circuit 113. The inverter circuit 113 operates in accordance with a switching signal input from the control device 114, converts the supplied DC power into a predetermined high frequency, for example, a single-phase AC power of 10 kHz, for example, with a predetermined current, and a conductive path for a feeder line. To 121.

  Therefore, the inverter circuit 113 is a main circuit of the inverse conversion unit, and includes three switching elements U, V, W connected to the positive electrode side P of the DC power smoothed by the smoothing capacitor 112, and the negative electrode side. It is composed of a total of six switching elements, three switching elements X, Y, and Z connected to N. In this example, an IGBT (insulated gate bipolar transistor) is used as each switching element, and flywheel diodes are connected in antiparallel to each of them.

  By the way, in such a main circuit of the inverter, the DC positive-side switching element is called an upper arm, and the negative-side switching element is called a lower arm. The switching element U of the upper arm and the switching element X of the lower arm become a switching element arm for one phase of the UX phase, and the switching element V of the upper arm and the switching element Y of the lower arm become one phase of the VY phase. The switching element arm of the upper arm and the switching element Z of the lower arm become the switching element arm for one phase of the WZ phase, and the UX phase, the VY phase, and the W- A switching element arm for three phases of the Z phase is formed.

  In the power converter for three-phase AC, a three-phase AC voltage is obtained by the output of the switching element arm for three phases. In the inverter circuit 113 of the power converter 110 according to the present embodiment, first, for three phases. The outputs R and S of the switching element arm for two phases of the switching element arms, that is, the UX phase switching element arm and the VY phase switching element arm are connected in common, and this is connected to the inverter circuit 113. One AC output terminal A is used. And the output T of the remaining switching element arm WZ phase is taken out as the other AC output terminal B, whereby a single-phase output power conversion device is configured.

  The high-frequency single-phase load 120 includes a conductive line 121 for a power supply line that allows a high-frequency single-phase current supplied from the output terminal A and the output terminal B, a reactance adjustment capacitor 122, and one or more mobile carts, for example. 123, 124,...

  Here, these movable carriages 123, 124,... Are object transport vehicles used in, for example, a clean room, are supplied with electric power in a non-contact manner from the conductive line 121, and move by driving a traveling motor. It is supposed to be. The conductive line 121 is two insulated lines that reciprocate along the movement path of the movable carriages 123, 124,. This is combined with a transformer that has only the secondary windings mounted on the movable carriages 123, 124,..., Except for one part of the iron core, and is open-circuited, and the conductive line 121 is moved as the primary winding. The carriages 123, 124,... Can receive electric power without contact.

  And in such non-contact power transmission, the higher the frequency of the power supply, the smaller the size of the magnetic flux intervening in the power transmission, and the smaller the device. A single-phase power supply is required.

  By the way, the reactance of the conductive line 121 is represented by 2πfL where L is the inductance and f is the frequency of the current flowing through the electric wire, and the magnitude is proportional to the inductance L and the frequency f. The inductance L of the conductive line 121 is proportional to the length of the conductive line, but is very small, for example, about several tens of μH. Therefore, the frequency f of the flowing current is several Hz to several hundred Hz as in a general three-phase induction motor. In this case, the reactance becomes small, and its length is not particularly problematic.

  However, since the frequency f of the current output from the power converter 110 of this embodiment to the conductive line 121 is as high as 10 kHz, for example, the reactance of the conductive line 122 is very important depending on the length thereof. That is, the longer the conductive line 121, the greater the reactance, making it difficult to flow a predetermined current at a high frequency. In this embodiment, therefore, a phase advance capacitor 122 is inserted in series with the conductive line 121 in order to reduce the reactance of the conductive line 121.

  When the capacitance of the phase advance capacitor 122 is C, the reactance is represented by 1 / 2πfC. Here, the reactance of the entire conductive line 121 is expressed by (2πfL) − (1 / 2πfC). When the result becomes positive, the reactance becomes inductive reactance, and the power factor of the load 120 becomes the delay power factor. On the other hand, when (2πfL) − (1 / 2πfC) is negative, it becomes capacitive reactance, and the power factor of the load 120 becomes the leading power factor.

  Next, the control device 114 in the power conversion device 110 in the present embodiment will be described. The control device 114 is composed of a circuit board on which a microcomputer or the like is mounted, and is turned on for the switching elements of the U-phase and V-phase of the upper arm of the inverter circuit 113 and the switching elements of the X-phase and Y-phase of the lower arm. It serves to supply a signal. The supply of the ON control signal to the remaining W-phase and Z-phase switching elements is not shown.

  Specifically, the current i flowing through the load 120 is detected by the current detector CT, and from the detection result, the control device 114 controls the U phase so that the output current i matches the output current command value given from the outside. The ON times T1, T3, T5, and T7 of the V-phase, X-phase, and Y-phase switching elements are controlled independently. As a result, two pulses of voltage are generated between the output terminals AB on the plus side and the minus side in one cycle, and the current flowing through the conductive line 121 can be controlled.

  For example, when the actual number of moving carriages 123, 124,... Is small and the output current command value is set to a small value, control is performed to shorten the pulse widths T1, T3, T5, and T7. At this time, depending on the output current command value, when it becomes smaller, one of these pulse widths, for example, the second pulse width T3, T7, becomes zero, and the other pulse width T1, There may be a situation where only T5 is controlled. In this state, the loss of the switching element that generates the second pulse is almost zero, and the generated loss in all the six switching elements is the smallest.

  On the other hand, for example, when the actual number of moving carriages 123, 124... Is large and the output current command value is set to a large value, the first pulse width T1, T3, and the second pulse width T5, Control is performed to increase T7. When the first pulse widths T1 and T3 and the second pulse widths T5 and T7 are controlled to be the longest, all of the pulse widths T1, T3, T5, and T7 are equal and are antiparallel to the respective switching elements of the inverter circuit 113. The return periods T2 and T6 in which current flows back to the connected flywheel diode and the return periods T4 and T8 in which energy is fed back to the smoothing capacitor 112 are the shortest or zero. At this time, the output current i is in the most growing state, and the loss generated in the entire six switching elements is also maximized.

Next, the structural arrangement of the inverter circuit 113 of the power conversion device of this embodiment will be described with reference to FIG.
The inverter circuit 113 includes a cooling fin 200, a U-phase switching element unit 211, a V-phase switching element unit 212, a W-phase switching element unit 213, and an X-phase switching element unit 214 arranged on the cooling fin 200. And a Y-phase switching element portion 215 and a Z-phase switching element portion 216.

Each switching element section uses IGBT (insulated gate bipolar transistor) as each switching element, and has a configuration in which flywheel diodes are connected in reverse parallel to each other.
.. 216 are connected to each other by a copper bar, an electric wire or the like so as to constitute the inverter circuit 113, and further, a smoothing capacitor CB is constituted so as to constitute the power converter 110. The AC output terminals A and B are also connected by a copper bar, an electric wire or the like, but are omitted in FIG.

  At this time, the V-Y phase switching element portions 212 and 215 receive heat from the UX phase switching element portions 211 and 214 and the WZ phase switching element portions 213 and 216 arranged on the left and right sides. Cooling becomes the most difficult. Therefore, consideration for cooling is required.

Next, the control operation of the inverter circuit 113 performed by the signal output from the control device 114 and the generation loss of the switching element that generates the first pulse and the second pulse when the load 120 becomes an inductive load are shown in FIG. This will be described with reference to a timing chart.
As already described, the control device 114 controls the ON time of the U-phase, V-phase, X-phase, and Y-phase in accordance with the actual operation number of the moving carriages 123, 124. Although the width of the pulse and the second pulse is determined, the second pulse may not be generated when the actual number of operations is small or when the output current command value is set small.
The generated loss of the entire switching element at that time is smaller than that when both the first pulse and the second pulse are generated. Therefore, regarding the generation loss, a state where both the first pulse and the second pulse, which are in a large state, are generated will be described.

  In FIG. 3, T represents a period of one cycle of single-phase alternating current to be generated between the alternating current output terminals A and B. Therefore, if the frequency of the output single-phase AC voltage is f, T = 1 / f. First, during the cycle period T / 2, which is half of this one cycle period T, the Z-phase switching element of the lower arm is turned on (conduction), and the V-phase and U-phase switching elements are controlled in the upper arm. The first pulse and the second pulse on the plus side are generated at the output of the power conversion device by alternately performing on-control in turn.

  Next, during the remaining half cycle period T / 2, the W-phase switching element of the upper arm is similarly turned on, and the Y-phase and X-phase switching elements are alternately turned on sequentially in the lower arm. Thus, a negative first pulse and a second pulse are generated at the output of the power converter. Thus, control is performed so that a single-phase AC voltage having a frequency f is generated between the AC output terminals A and B. In general, in order to control the inverter main circuit, a dead time is generally provided for the operation of the switching element.

At this time, the output current i has a waveform close to a sine wave as shown in FIG. 3, and when the load 120 is an inductive load as described above, it has a delayed power factor.
When the switching elements U, V, W, X, Y, and Z repeat switching in the switching pattern shown in FIG. 3, each switching element generates a loss and the temperature rises.

  Incidentally, the loss of the switching element is divided into a steady loss and a switching loss, and the switching loss is further divided into a turn-on loss and a turn-off loss. Both of these losses depend on the output voltage pulse and the output current i in FIG. Since both the first pulse and the second pulse are output from the smoothing capacitor 112 and have the same peak value, it is necessary to compare the loss generated by each pulse with the output current i.

In FIG. 3, i31 indicates the instantaneous value of the output current when the positive first pulse rises, and i32 indicates the instantaneous value of the output current when the positive first pulse falls. In FIG. 3, i33 indicates the instantaneous value of the output current when the positive first pulse rises, and i34 indicates the instantaneous value of the output current when the positive first pulse falls. Also, V _PN in FIG. 3 is a peak value of the first pulse and a second pulse generated 1 output terminal A when the arm U-phase and V-phase of inverter circuit 113 is turned on, the B, -V _PN is the peak values of the first pulse and the second pulse generated at the output terminals A and B in FIG. 1 when the lower arm X phase and Y phase of the inverter circuit 113 are turned on.

  Here, since the load 120 is an inductive load and has a delay power factor, when comparing the instantaneous values i31 and i33 of the current at the rising edge of the first and second pulses, i33 becomes larger, 1. When the instantaneous values i32 and i34 of the current at the fall of the first and second pulses are compared, i34 becomes larger. As described above, since the turn-on loss and the turn-off loss of the switching element depend on the current value, both losses are larger in the second pulse.

  The steady loss in the first pulse depends on the integrated value of the output current i in the period T1, and the steady loss in the second pulse depends on the integrated value of the output current i in the period T3. The steady loss in the pulse is larger.

  However, when the actual number of moving carriages 123, 124,... Is small and the output current command value is set to a small value, the period T3 may be extremely shorter than the period T1. The steady loss in the second pulse is smaller than the steady loss in the first pulse.

  When the total loss of the switching element is compared here, the loss generated by the second pulse is larger in all of the turn-on loss, the turn-off loss, and the steady loss, except when the period T3 is extremely shorter than the period T1. Therefore, the total loss is larger in the switching element that generates the second pulse than in the switching element that generates the first pulse.

Even when the period T3 is extremely shorter than the period T1, the ratio of the turn-on loss and the turn-off loss is higher than the steady loss in the apparatus that requires high-frequency power such as 10 kHz as in this embodiment. Is much larger, the switching element that generates the second pulse has a greater loss than the switching element that generates the first pulse.
These losses are the same for the negative first pulse and the second pulse only by reversing the polarity.

  Therefore, when the load 120 is an inductive load, the second pulse is generated in a state where two pulses are generated on each of the positive side and the negative side in one cycle regardless of the lengths of the periods T1, T3, T5, and T7. The switching element that generates the loss has a larger loss. Here, in the first embodiment, the switching elements 211 to 216 are arranged as shown in FIG. 2, and the VY phase switching elements 212 and 215 receive heat from the adjacent switching elements and rise in temperature. At the same time, the element is the most difficult to cool.

Therefore, among the first pulse and the second pulse, the first pulse with relatively small generation loss is generated in the switching element VY phase, and the second pulse with large generation loss is structurally easy to cool the switching element U- X. Control to generate in phase.

  By doing in this way, local overheating of a switching element can be prevented and the cooling efficiency of a switching element can be improved.

Next, the loss of the switching element that generates the first pulse and the second pulse when the load 120 becomes a capacitive load will be described with reference to FIG.
In FIG. 4, i41 indicates the instantaneous value of the output current when the positive first pulse rises, and i42 indicates the instantaneous value of the output current when the positive first pulse falls. Further, i43 in FIG. 4 indicates an instantaneous value of the output current when the positive first pulse rises, and i44 indicates an instantaneous value of the output current when the positive first pulse falls.

  Here, since the load 120 is a capacitive load and has a leading power factor, when comparing the instantaneous values i41 and i43 of the current at the first and second pulse rising, i41 becomes larger, 1. When comparing the instantaneous values i42 and i44 of the current at the fall of the first and second pulses, i42 becomes larger. As described above, since the turn-on loss and the turn-off loss of the switching element depend on the current value, both losses are larger in the first pulse.

  The steady loss in the first pulse depends on the integrated value of the output current i in the period T1, and the steady loss in the second pulse depends on the integrated value of the current i in the period T3. In the non-contact supply system of the present embodiment, when the period T1 and the period T3 are compared, the control is performed such that T1 is longer than T3 or equal to T1 and T3. The steady loss of becomes larger than the steady loss of T3.

  Here, when the total loss of the switching element is compared, the loss generated by the first pulse is larger in all of the turn-on loss, the turn-off loss, and the steady loss, and therefore the total loss is also the direction of the switching element that generates the first pulse. Becomes larger than the switching element that generates the second pulse. These losses are the same for the negative first pulse and the second pulse only by reversing the polarity.

  Therefore, when the load 120 is a capacitive load, the first pulse is generated in a state where two pulses are generated on each of the positive side and the negative side in one cycle regardless of the lengths of the periods T1, T3, T5, and T7. The generated switching element has a particularly large loss. In the first embodiment, the switching elements 211 to 216 are arranged as shown in FIG. 2, and the VY phase switching elements 212 and 215 receive heat from the adjacent switching elements and rise in temperature. The element is located in the place where cooling is most difficult.

  Therefore, of the first pulse and the second pulse, the switching element VY generates a second pulse having a relatively small generation loss, and the switching element U-V is structurally easy to cool the first pulse having a large generation loss. Control to generate.

  By doing in this way, local overheating of a switching element can be prevented and the cooling efficiency of a switching element can be improved.

  As described above, in the non-contact power feeding system according to the present embodiment, when the load 120 is an inductive load, the first pulse is generated by the switching element VY as shown in FIG. When the load 120 is a capacitive load, the first pulse is generated by the switching element U-X and the second pulse is generated by the switching element V- as shown in FIG. Control is performed so that Y is generated.

FIG. 7 is a flowchart showing the control until the control device 114 supplies the ON control signal to the switching elements of the U phase, the V phase, the X phase, and the Y phase.
First, the control device 114 determines whether or not the high-frequency single-phase load 120 connected to the power conversion device 113 is inductive or capacitive (S701). In this power converter 113, whether the property of the high-frequency single-phase load to be connected is inductive or capacitive is measured in advance and manually input to this power converter 113 for determination. It is also possible to perform automatic determination by providing a circuit for determining the nature of the load in the power conversion device.

  For example, immediately after starting the power converter, a low voltage is generated for the load for a short time, and at this time, it is detected whether the phase of the current flowing through the load is delayed or advanced with respect to the generated low voltage. For example, there may be a method of providing a circuit for determining that a delay power factor becomes an inductive load when it is, and a lead power factor when it is advanced, a capacitive load.

  Next, the pulse generated by the switching element of each phase is determined according to the result of the load property determination. That is, when the load property determination result is an inductive load, the positive first pulse is the V phase, the positive second pulse is the U phase, the negative first pulse is the Y phase, and the negative second pulse is the X phase. (S702). On the other hand, if the load property determination result is capacitive load, the positive first pulse is the U phase, the positive second pulse is the V phase, the negative first It is determined that the pulse is generated in the X phase and the negative second pulse is generated in the Y phase (S703).

  Finally, an ON control signal is supplied to each of the U-phase, V-phase, X-phase, and Y-phase switching elements (S704). At this time, as described above, the lengths of the ON times T1, T3, T5, and T7 of the switching elements of the U-phase, V-phase, X-phase, and Y-phase depend on the high-frequency single-phase load 120. Is controlled to become the output current command value.

  Thereby, local overheating of a switching element can be prevented and the cooling efficiency of a switching element can be improved.

  In this embodiment, in a non-contact supply system that supplies power to a high-frequency single-phase load in a non-contact manner using a power converter, no capacitor for reactance adjustment is inserted in a conductive line connected to the power converter. An example of the case will be described.

  FIG. 5 is an example of a configuration diagram of a non-contact supply system including a commercial power source 500, a power conversion device 510, and a high-frequency single-phase load 520. The reactance adjustment capacitor is not connected to the load 520, The portion is the same as that shown in FIG. 1, and the structural arrangement of the inverter control circuit 520 is the same as that shown in FIG.

  As already described in the first embodiment, the reactance of the load 520 depends on the length of the conductive line 522. Therefore, when the conductive line is long, the reactance adjustment is performed to supply desired power to the mobile carriages 523, 524. It is necessary to insert a capacitor for reducing the reactance.

  However, when the conductive line 522 is very short and the power consumption of the mobile carriages 523, 524... Receiving power from the conductive line 522 in a non-contact manner is very small, or when the output current command value is set to be very small. As shown in FIG. 5, it is possible to realize a non-contact power feeding system without inserting a reactance adjustment capacitor.

  At this time, the reactance of the entire high-frequency single-phase load 520 is positive because there is no capacitance C of the reactance adjustment capacitor, and the high-frequency single-phase load 520 is an inductive load.

Next, the control operation of the inverter circuit 513 performed by the signal output from the control device 514 and the generated loss of the switching element will be described with reference to the timing chart of FIG.
The control operation of the switching elements U, V, W, X, Y, and Z in the inverter circuit 513 is the same as that already described in the first embodiment, and the first pulse and the first pulse on the plus side in the period of one cycle. Two pulses and negative first and second pulses are generated, and pulse widths T1, T3, and T5, T7 are independently controlled in accordance with the actual number of moving carriages 523, 524, etc. and the output current command value. Operate. The output current i at this time is not close to a sine wave but has a linear waveform as shown in FIG.

  Further, as described in the first embodiment, the loss of each switching element is divided into a steady loss and a switching loss, and the switching loss is further divided into a turn-on loss and a turn-off loss. Both of these losses depend on the output voltage pulse and the output current i in FIG. Since both the first pulse and the second pulse are output from the smoothing capacitor 512 and have the same peak value, it is necessary to compare the loss generated by each pulse with the output current i.

  In FIG. 6, i61 indicates the instantaneous value of the output current when the positive first pulse rises, and i62 indicates the instantaneous value of the output current when the positive first pulse falls. In FIG. 6, i63 indicates the instantaneous value of the output current when the positive first pulse rises, and i64 indicates the instantaneous value of the output current when the positive first pulse falls.

Also, V _PN in FIG. 6 is a peak value of the first pulse and the second pulse generated when the arms U and V phases of inverter circuit 513 is turned on Figure 5 the output terminal A, to B, -V _PN is the peak value of the first pulse and the second pulse generated at the output terminals A and B in FIG. 1 when the lower arm X phase and Y phase of the inverter circuit 513 are turned on.

Here, the loss of the switching element when the first pulse and the second pulse are generated in the state where the generation loss in the entire six switching elements is maximized will be described.
The current value i63 grows from the current value i61 during the period T1 during which the first pulse is generated, and decreases during the reflux period T2. The rate of decrease in current during the reflux period T2 depends on the length of the conductive path 521 and the number of mobile carriages 523, 524... As described above, the pulse widths T1, T3, T5, and T7 are controlled to be equally longest. On the other hand, the return periods T2 and T4 and the feedback periods T4 and T6 are controlled to be shortest.

  Thereby, the current growth rate in the period T1 does not fall below the current decrease rate in the reflux period T2. Therefore, the current value i63 is larger than the current value i61, and the turn-on loss in the second pulse is larger than the turn-on loss in the first pulse.

  Further, the current value i64 decreases from the current value i62 during T2, which is the reflux period, and grows during the period of T3 during which the second pulse is generated. In a state where the total generated loss is maximized, the current growth rate in the period T3 does not fall below the current decrease rate in the period T2. Therefore, the current value i64 is larger than the current value i62, and the turn-off loss in the second pulse is larger than the turn-off loss in the first pulse.

  Next, the steady loss of the positive first pulse depends on the integrated value of the output current i in the supply period T1. On the other hand, the steady loss of the positive second pulse depends on the integrated value of the output current i in the supply period T2. Here, since the control is performed so that the periods T1 and T3 are equal in the state where the generated loss in all the six switching elements is also the maximum, the integrated value of the output current i is T2 compared with the period T1. The period is higher.

  Therefore, the loss of the switching element when the first pulse and the second pulse are generated is particularly large in the switching element that generates the second pulse in all of the steady loss, the turn-on loss, and the turn-off loss. The same applies to the negative first pulse and the second pulse only by reversing the polarity.

  On the other hand, as described in FIG. 2, the switching elements V and Y are subjected to the heat generated by the left and right switching elements U, X and W, Z, so that cooling is most difficult. Therefore, in the power conversion apparatus 500 in the present embodiment, the first pulse and the second pulse of the first pulse having a relatively small generation loss are generated by the switching elements V and Y, and the second pulse having a large generation loss is structured. Furthermore, the switching elements U and V that are relatively easy to cool are controlled so as to be generated.

  Further, the control until the control device 514 supplies the ON control signal to the switching elements of the U phase, the V phase, the X phase, and the Y phase at this time is the same as that in the first embodiment as shown in the flowchart of FIG. It becomes street.

  First, the control device 514 determines whether or not the high-frequency single-phase load 520 connected to the power converter 513 is inductive or capacitive. Since no reactance adjustment capacitor is inserted in the high-frequency single-phase load 520, the determination result is an inductive load (S701).

  Next, the pulse generated by the switching element of each phase is determined according to the result of the load property determination. Since the load property judgment result is an inductive load, the positive first pulse is generated in the V phase, the positive second pulse is generated in the U phase, the negative first pulse is generated in the Y phase, and the negative second pulse is generated in the X phase. (S702).

  Finally, an ON control signal is supplied to each of the U-phase, V-phase, X-phase, and Y-phase switching elements (S704). At this time, as described above, the lengths of the ON times T1, T3, T5, and T7 of the switching elements of the U-phase, V-phase, X-phase, and Y-phase depend on the high-frequency single-phase load 520. Is controlled to become the output current command value.

  Thereby, local overheating of the switching element can be prevented, and the cooling efficiency of the switching element can be increased.

DESCRIPTION OF SYMBOLS 100 ... AC power supply 110 ... Power converter 111 ... Converter circuit 112 ... Smoothing capacitor 113 ... Inverter circuit 114 ... Control circuit 120 ... High frequency single phase load 121 ... Conductive path 122 ... Reactance adjustment capacitor 123 ... Moving cart 124 ... Moving cart 200 ... cooling fin 211 ... U phase switching element part 212 ... V phase switching element part 213 ... W phase switching element part 214 ... X phase switching element part 215 ... Y phase switching element part 216 ... Z phase switching Element unit 500 ... AC power supply 510 ... Power converter 511 ... Converter circuit 512 ... Smoothing capacitor 513 ... Inverter circuit 514 ... Control circuit 520 ... High-frequency single-phase load 521 ... Conductive path 523 ... Moving carriage 524 ... Moving carriage

Claims (6)

An inverse conversion unit for converting DC power having a three-phase switching element arm into AC power;
The outputs of any two switching element arms of the three-phase switching element arms of the inverse conversion unit are connected in common to form one AC output terminal, and the output of the remaining one circuit switching element arm is the other. on which the AC output terminal, said second circuit each a, conducts controlled independently of each switching element of one and the other, and configured to generate single-phase alternating current to between the one and the other AC output terminal In the power converter,
Of the pulses generated by the upper and lower arms of the two circuits during one cycle, the switching element arm that generates the pulse with the smaller generation loss is cooled in the installation place of the switching element arms of the two circuits. Is a power converter characterized by being arranged in the most difficult place. The power conversion device according to claim 1,
When the load connected to the output terminal is an inductive load, the switching element arm for generating the first pulse having the earlier time among the pulses generated by the upper arm and the lower arm of the two circuits during one cycle. Is arranged at a place where the cooling is most difficult among the places where the switching element arms of the two circuits are installed. The power conversion device according to claim 1,
When the load connected to the output terminal is a capacitive load, the switching element arm for generating the second pulse having the later time among the pulses generated by the upper arm and the lower arm of the two circuits in one cycle. Is arranged at a place where the cooling is most difficult among the places where the switching element arms of the two circuits are installed. An inverse conversion unit for converting DC power having a three-phase switching element arm into AC power;
The outputs of any two switching element arms of the three-phase switching element arms of the inverse conversion unit are connected in common to form one AC output terminal, and the output of the remaining one circuit switching element arm is the other. on which the AC output terminal, said second circuit each a, conducts controlled independently of each switching element of one and the other, and configured to generate single-phase alternating current to between the one and the other AC output terminal A method for controlling a power converter,
Of the pulses generated by the upper and lower arms of the two circuits during one cycle, the pulse generation with the smaller generation loss is the most difficult place for cooling among the installation positions of the switching element arms of the two circuits. A control method for a power converter, wherein the switching element arm is controlled so as to be generated by the switching element arm. It is a control method of the power converter device according to claim 4,
Determine whether the load connected to the output terminal is an inductive load or a capacitive load,
In the case where the determination is an inductive load, the switching element arm arranged at the place where the cooling is most difficult among the installation places of the switching element arms of the two circuits, and the upper arm and the lower arm of the two circuits are Among the pulses generated during one period, the first pulse having the earlier time is generated, and the other switching element arm generates the first pulse generated by the upper and lower arms of the two circuits during one period. , Generate the second pulse with the later time ,
Wherein when the determination is a capacitive load, to generate the second pulse in the switching element arm cooling is arranged in the most difficult places among the location of the switching element arm of the two circuits, the other switching A control method for a power converter, wherein the element arm is controlled to generate the first pulse. It is a control method of the power converter device according to claim 5,
To determine whether the load is an inductive load or a capacitive load, a low voltage is generated for the load for a short time immediately after the power converter is started, and the phase of the current flowing through the load at that time is the generated low voltage. A method for controlling a power conversion apparatus, comprising: detecting whether the vehicle is late or advanced, determining that the vehicle is delayed as an inductive load, and determining the vehicle being advanced as a capacitive load.

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