JP5975797B2 - Power converter - Google Patents

Description

  The present invention relates to a power conversion device.

  The power conversion device includes a plurality of switching elements, and can convert DC power into AC power and can convert AC current into DC power by a switching operation of the switching elements. The switching element is generally controlled based on a pulse width modulation method (hereinafter referred to as a PWM method) using a carrier wave that changes at a constant frequency.

  In an electric vehicle such as an electric vehicle or a hybrid vehicle, a DC voltage from a DC power source is converted into an AC voltage to control a motor generator for driving the vehicle, or AC power induced in the motor generator is converted into DC power. A power conversion device is used for the purpose. A smoothing capacitor is provided in parallel with a DC power source (for example, a battery) on the DC side of such a power converter. This smoothing capacitor prevents the power converter from becoming unstable due to a steep current change that occurs during power transfer between the battery and the inverter.

  Generally, in an electric vehicle, the main relay is turned off when an ignition switch that commands operation stop is turned off. At this time, from the viewpoint of safety, it is preferable to remove the electric charge accumulated in the smoothing capacitor. As a method of discharging the remaining charge of the smoothing capacitor, for example, there is a method of connecting a discharge resistor in parallel with the smoothing capacitor.

  However, in this method, a discharge path is always formed even during vehicle operation due to the discharge resistance. Therefore, it is necessary to increase the discharge resistance value in terms of efficiency. For this reason, it is difficult to discharge the smoothing capacitor at high speed when the vehicle is stopped. There is also a method of adding a discharge circuit device in which a switch is provided in series with the discharge resistance so that the discharge resistance value is kept low while the vehicle is running, but the cost increases due to the addition of the switch and the switch control circuit. End up.

  In view of this, there has been proposed a method of quickly discharging the remaining charge of the smoothing capacitor by controlling the power converter without adding a discharge circuit device (for example, Patent Document 1).

  Patent Document 1 discloses that the power converter is controlled so that positive and negative high-frequency currents are supplied in a short cycle that does not contribute to rotation considering the inertia of the motor generator. According to this control, the remaining charge of the smoothing capacitor can be discharged in a state where the motor generator does not substantially generate rotational torque. The high-frequency current mainly becomes a conduction loss when flowing through the stator winding of the motor generator, etc., and it is thought that the magnetic loss (eddy current loss, hysteresis loss, etc.) of the motor generator also occurs at the same time. Is converted into thermal energy.

  However, in such a conventional discharge control method, rotational torque is not substantially generated when the motor generator is stopped, but rotational torque is generated when the motor generator is rotating (that is, the vehicle is accelerated / decelerated). There was a possibility.

Japanese Patent No. 4241163

  Then, an object of this invention is to provide the power converter device which does not generate | occur | produce a rotational torque, when discharging a smoothing capacitor.

In order to solve the above problems, for example, the configuration described in the claims is adopted. The present application includes a plurality of means for solving the above-described problems. For example, the present application includes a plurality of series circuits in which a switching element for an upper arm and a switching element for a lower arm are connected in series. a power switching circuit for generating an AC power by receiving power by cutting the main relay if the rotational speed of a state in which the power supply from the battery to the inverter is shut off motor is equal to or less than a predetermined value, at least one in Rukoto changing the cycle of the control pulses of the phase twice equivalent carrier cycle, the alternating current having a period corresponding to twice the carrier period is supplied to the electric motor, discharging the accumulated charge in the smoothing capacitor by the alternating current A power conversion device having a function to be configured is configured.

  ADVANTAGE OF THE INVENTION According to this invention, the power converter device which does not generate | occur | produce a rotational torque when discharging a smoothing capacitor can be provided.

The system diagram which shows the system of a hybrid vehicle. The circuit diagram which shows the structure of the electric circuit shown in FIG. An explanatory view showing PWM control. 1 is a diagram illustrating a pulse modulator according to Embodiment 1. FIG. The figure which shows the example of normal PWM pulse phase control. FIG. 3 is a diagram illustrating PWM pulse phase control according to the first embodiment. The figure which shows the update timing of on / off of the PWM pulse of Example 1. FIG. The figure which shows the relationship between an input waveform and a carrier. The experimental result of the pulse shift by Example 1. FIG. The experimental result of the pulse shift by Example 1. FIG. FIG. 3 is an explanatory diagram of PWM pulse phase control according to the first embodiment. FIG. 3 is an explanatory diagram of PWM pulse phase control according to the first embodiment. The figure which shows the line voltage at the time of the normal PWM control and the pulse shift by Example 1. FIG. 6 is a diagram illustrating PWM pulse phase control according to the second embodiment. The figure which shows the update timing of ON of the PWM pulse of Example 2, and OFF. The figure which shows the relationship between an input waveform and a carrier. Explanatory drawing of the PWM pulse phase control of Example 2. FIG. Explanatory drawing of the PWM pulse phase control of Example 2. FIG. FIG. 6 is a diagram illustrating a line voltage during pulse shift according to the second embodiment.

  Embodiments of the present invention will be described with reference to the drawings.

Note that the AC power output from the power converter is supplied to an inductance circuit composed of a rotating electrical machine or the like, and an AC current flows based on the action of the inductance. In the following embodiments, a rotating electrical machine that acts as an electric motor or a generator as an inductance circuit will be described as an example.
The use of the present invention to generate AC power for driving the rotating electrical machine is optimal from the viewpoint of effect, but it can also be used as a power conversion device that supplies AC power to an inductance circuit other than the rotating electrical machine.
[Example 1]
A power conversion device according to an embodiment of the present invention is a power conversion device that generates AC power for driving a rotating electrical machine of a hybrid vehicle (hereinafter referred to as HEV) or a pure electric vehicle (hereinafter referred to as EV). This is an applied example. The power converter for HEV and the power converter for EV are common in basic configuration and control, and as a representative example, a control configuration when the power converter according to the embodiment of the present invention is applied to a hybrid vehicle. The circuit configuration of the power converter will be described with reference to FIGS.

  FIG. 1 is a diagram showing a control block of a hybrid vehicle (hereinafter referred to as “HEV”). Engine EGN and motor generator MG1 generate vehicle running torque. Motor generator MG1 not only generates rotational torque but also has a function of converting mechanical energy applied from the outside to motor generator MG1 into electric power.

  Motor generator MG1 may be a synchronous machine or an induction machine, and operates as an electric motor or a generator depending on the operation method as described above. When motor generator MG1 is mounted on an automobile, it is desirable to obtain a small size and high output, and a permanent magnet type synchronous motor using a magnet such as neodymium is suitable. The permanent magnet type synchronous motor generates less heat from the rotor than the induction motor, and is excellent for automobiles from this viewpoint.

  The output torque on the output side of the engine EGN is transmitted to the motor generator MG1 via the power distribution mechanism TSM, and the rotation torque from the power distribution mechanism TSM or the rotation torque generated by the motor generator MG1 is transmitted via the transmission TM and the differential gear DEF. Transmitted to the wheels. On the other hand, during regenerative braking operation, rotational torque is transmitted from the wheels to motor generator MG1, and AC power is generated based on the supplied rotational torque. The generated AC power is converted to DC power by the power conversion device 200 as described later, and the high-voltage battery 136 is charged, and the charged power is used again as travel energy.

  Next, the power conversion device 200 will be described. The inverter circuit 140 is electrically connected to the battery 136 via the DC connector 138, and power is exchanged between the battery 136 and the inverter circuit 140. When motor generator MG1 is operated as an electric motor, inverter circuit 140 generates AC power based on DC power supplied from battery 136 via DC connector 138, and supplies it to motor generator MG1 via AC connector 188. . The configuration comprising motor generator MG1 and inverter circuit 140 operates as a first motor generator unit.

  In the present embodiment, the first motor generator unit is operated as an electric unit by the electric power of the battery 136, so that the vehicle can be driven only by the power of the motor generator MG1. Furthermore, in the present embodiment, the battery 136 can be charged by operating the first motor generator unit as a power generation unit by the power of the engine 120 or the power from the wheels to generate power.

  Although omitted in FIG. 1, the battery 136 is also used as a power source for driving an auxiliary electric motor. The auxiliary motor is, for example, an electric motor that drives a compressor of an air conditioner or an electric motor that drives a control hydraulic pump. DC power is supplied from the battery 136 to the auxiliary power module, and the auxiliary power module generates AC power and supplies it to the auxiliary electric motor. The auxiliary power module has basically the same circuit configuration and function as the inverter circuit 140, and controls the phase, frequency, and power of alternating current supplied to the auxiliary electric motor.

  The power conversion device 200 includes a capacitor module 500 for smoothing the DC power supplied to the inverter circuit 140.

  The power conversion device 200 includes a communication connector 21 for receiving a command from a host control device or transmitting data representing a state to the host control device. Power conversion device 200 calculates a control amount of motor generator MG1 by control circuit 172 based on a command input from connector 21, further calculates whether to operate as an electric motor or a generator, and based on the calculation result. The control pulse is generated, and the control pulse is supplied to the driver circuit 174. The driver circuit 174 generates a driving pulse for controlling the inverter circuit 140 based on the supplied control pulse.

  Next, the configuration of the electric circuit of the inverter circuit 140 will be described with reference to FIG. In the following description, an insulated gate bipolar transistor is used as a switching element, and hereinafter abbreviated as IGBT. The IGBT 328 and the diode 156 that operate as the upper arm, and the IGBT 330 and the diode 166 that operate as the lower arm constitute the series circuit 150 of the upper and lower arms. The inverter circuit 140 includes the series circuit 150 corresponding to three phases of the U phase, the V phase, and the W phase of the AC power to be output.

  These three phases correspond to the three phase windings of the armature winding of the motor generator MG1 in this embodiment. The series circuit 150 of the upper and lower arms of each of the three phases outputs an alternating current from a connection point 169 (intermediate electrode) that is a midpoint portion of the series circuit. The connection point 169 is connected to the motor generator MG1 through an AC bus bar 802 (described later) that connects the AC terminal 159 and the AC connector 188.

  The collector electrode 153 of the IGBT 328 of the upper arm is electrically connected to the capacitor terminal 506 on the positive electrode side of the capacitor module 500 via the positive electrode terminal 157. The emitter electrode of the IGBT 330 of the lower arm is electrically connected to the capacitor terminal 504 on the negative electrode side of the capacitor module 500 via the negative electrode terminal 158.

  As described above, the control circuit 172 receives a control command from the host control device via the connector 21, and based on this, the IGBT 328 that configures the upper arm or the lower arm of each phase series circuit 150 that constitutes the inverter circuit 140. And a control pulse that is a control signal for controlling the IGBT 330 is generated and supplied to the driver circuit 174.

  Based on the control pulse, the driver circuit 174 supplies a drive pulse for controlling the IGBT 328 and IGBT 330 constituting the upper arm or the lower arm of each phase series circuit 150 to the IGBT 328 and IGBT 330 of each phase. IGBT 328 and IGBT 330 perform conduction or cutoff operation based on the drive pulse from driver circuit 174, convert DC power supplied from battery 136 into three-phase AC power, and supply the converted power to motor generator MG1. Is done.

  The IGBT 328 includes a collector electrode 153, a signal emitter electrode 155, and a gate electrode 154. The IGBT 330 includes a collector electrode 163, a signal emitter electrode 165, and a gate electrode 164. A diode 156 is electrically connected between the collector electrode 153 and the emitter electrode 155. A diode 166 is electrically connected between the collector electrode 163 and the emitter electrode 165.

  As the switching power semiconductor element, a metal oxide semiconductor field effect transistor (hereinafter abbreviated as MOSFET) may be used. In this case, the diode 156 and the diode 166 are unnecessary. As a power semiconductor element for switching, IGBT is suitable when the DC voltage is relatively high, and MOSFET is suitable when the DC voltage is relatively low.

  The capacitor module 500 includes a capacitor terminal 506 on the positive electrode side, a capacitor terminal 504 on the negative electrode side, a power supply terminal 509 on the positive electrode side, and a power supply terminal 508 on the negative electrode side. The high-voltage DC power from the battery 136 is supplied to the positive-side power terminal 509 and the negative-side power terminal 508 via the DC connector 138, and the positive-side capacitor terminal 506 and the negative-side capacitor of the capacitor module 500. The voltage is supplied from the terminal 504 to the inverter circuit 140.

  On the other hand, the DC power converted from the AC power by the inverter circuit 140 is supplied to the capacitor module 500 from the positive capacitor terminal 506 and the negative capacitor terminal 504, and is connected to the positive power terminal 509 and the negative power terminal 508. Is supplied to the battery 136 via the DC connector 138 and accumulated in the battery 136.

  The control circuit 172 includes a microcomputer (hereinafter referred to as “microcomputer”) for calculating the switching timing of the IGBT 328 and the IGBT 330. The input information to the microcomputer includes a target torque value required for the motor generator MG1, a current value supplied from the series circuit 150 to the motor generator MG1, and a magnetic pole position of the rotor of the motor generator MG1.

  The target torque value is based on a command signal output from a host controller (not shown). The current value is detected based on a detection signal from the current sensor 180. The magnetic pole position is detected based on a detection signal output from a rotating magnetic pole sensor (not shown) such as a resolver provided in the motor generator MG1. In this embodiment, the current sensor 180 detects the current value of three phases, but the current value for two phases may be detected and the current for three phases may be obtained by calculation. .

  The microcomputer in the control circuit 172 calculates the d and q axis current command values of the motor generator 192 based on the input target torque value, and detects the calculated d and q axis current command values. Based on the difference between the d and q axis current values, the d and q axis voltage command values are calculated, and a pulsed drive signal is generated from the d and q axis voltage command values. The control circuit 172 has a function of generating a drive signal of a method according to an embodiment of the present invention described later.

  Here, the d-axis is an axis that is the direction of the main magnetic flux by the rotor magnet, and the q-axis is an axis that is electrically orthogonal to the d-axis.

  This method is a modulation method for controlling the switching operation of the IGBTs 328 and 330 that are switching elements based on the ripple of the AC waveform to be output and the magnetic pole position signal of the motor.

  When driving the lower arm, the driver circuit 174 amplifies a pulse-like modulated wave signal and outputs it as a drive signal to the gate electrode of the corresponding IGBT 330 of the lower arm. When driving the upper arm, the reference potential level of the pulsed modulated wave signal is shifted to the reference potential level of the upper arm, and then the pulsed modulated wave signal is amplified and used as a drive signal. , Output to the gate electrode of the IGBT 328 of the corresponding upper arm. As a result, each IGBT 328, 330 performs a switching operation based on the input drive signal. In this way, by the switching operation of the IGBTs 328 and 330 performed in accordance with the drive signal (drive signal) from the control unit 174, the inverter circuit 140 converts the voltage supplied from the battery 136, which is a DC power supply, to 2π / 3 rad in electrical angle. The output voltage is converted into U-phase, V-phase, and W-phase output voltages that are shifted every time and supplied to a motor generator 192 that is a three-phase AC motor. The electrical angle corresponds to the rotation state of the motor generator 192, specifically the position of the rotor, and periodically changes between 0 and 2π. By using this electrical angle as a parameter, the switching states of the IGBTs 328 and 330, that is, the output voltages of the U phase, the V phase, and the W phase can be determined according to the rotation state of the motor generator 192.

  In addition, the microcomputer in the control circuit 172 detects abnormality (overcurrent, overvoltage, overtemperature, etc.) and protects the series circuit 150. For this reason, sensing information is input to the control circuit 172. For example, information on the current flowing through the emitter electrodes of the IGBTs 328 and IGBTs 330 is input to the corresponding drive units (ICs) from the signal emitter electrode 155 and the signal emitter electrode 165 of each arm. Thereby, each drive part (IC) detects overcurrent, and when overcurrent is detected, it stops the switching operation of corresponding IGBT328 and IGBT330, and protects corresponding IGBT328 and IGBT330 from overcurrent.

  Information on the temperature of the series circuit 150 is input to the microcomputer from a temperature sensor (not shown) provided in the series circuit 150. In addition, voltage information on the DC positive side of the series circuit 150 is input to the microcomputer. The microcomputer performs overtemperature detection and overvoltage detection based on such information, and stops switching operations of all the IGBTs 328 and IGBTs 330 when an overtemperature or overvoltage is detected.

  FIG. 3 shows a motor control system by the control circuit 172 of this embodiment. A torque command T * as a target torque value is inputted to the control circuit 172 from a host control device. The torque command / current command converter 210 is based on the input torque command T * and the electrical angular velocity ωre calculated by the angular velocity calculator 260 based on the magnetic pole position signal θre detected by the rotating magnetic pole sensor 193. The d-axis current command signal Id * and the q-axis current command signal Iq * are obtained using the torque-rotation speed map data stored in advance. The d-axis current command signal Id * and the q-axis current command signal Iq * obtained by the torque command / current command converter 210 are output to the current controller (ACR) 220.

  The current controller (ACR) 220 is a phase of the d-axis current command signal Id * and the q-axis current command signal Iq * output from the torque command / current command converter 210 and the phase of the motor generator 192 detected by the current sensor 180. The current detection signals lu, lv, and lw are based on the Id and Iq current signals converted on the d and q axes by the magnetic pole position signal θre from the rotating magnetic pole sensor in a three-phase two-phase converter (not shown) on the control circuit 172. Thus, the d-axis voltage command signal Vd * and the q-axis voltage command signal Vq * are respectively calculated so that the current flowing through the motor generator 192 follows the d-axis current command signal Id * and the q-axis current command signal Iq *. The d-axis voltage command signal Vd * and the q-axis voltage command signal Vq * obtained by the current controller (ACR) 220 are output to the pulse modulator 230.

  The pulse modulator 230 has six types corresponding to the U-phase, V-phase, and W-phase upper and lower arms, respectively, based on the d-axis voltage command signal Vd *, the q-axis voltage command signal Vq *, and the magnetic pole position signal θre from the rotating magnetic pole sensor. The pulse signal is generated. Then, the generated pulse signal is output to the driver circuit 174, and a drive signal is output to each switching element.

  As described above, a pulse signal is output as a modulated wave from the control circuit 172 to the driver circuit 174. In response to this modulated wave, a drive signal is output from the driver circuit 174 to the IGBTs 328 and 330 of the inverter circuit 140.

  FIG. 4 shows details of the pulse modulator 230. The two-phase three-phase converter converts the d-axis voltage command signal Vd * and the q-axis voltage command signal Vq * to each phase based on the d-axis voltage command signal Vd *, the q-axis voltage command signal Vq *, and the magnetic pole position signal θre. Are converted to voltage command values Vu *, Vv *, Vw * and sent to the pulse width converter. The pulse width converter generates a pulse signal based on the voltage command values Vu *, Vv *, and Vw * of each phase that are phase-controlled by the PWM pulse phase control means.

  In the motor control system shown in FIG. 3, for example, a control cycle of about several hundred μs is determined in advance as a control cycle for the motor generator 192 in response to a request from the system performance. The pulse modulator 230 repeatedly calculates the states of the IGBTs 328 and 330 serving as switching elements for each control period. Depending on the calculation result, a pulse signal in the next control cycle is generated and output to the driver circuit 174.

  The PWM pulse phase control of this embodiment will be described with reference to FIGS.

  FIG. 5 shows an example of normal PWM pulse phase control. The carrier Tc is a triangular wave, and the pulses Vu, Vv, Vw of each phase are symmetrical with respect to the carrier peak Tc_peak.

  FIG. 6 is a diagram showing PWM pulse phase control of this embodiment. This figure shows a case where the pulse shift is performed on the V phase. Twice the carrier period is set to one period, and the phase interval between the center of the on period and the center of the off period of the pulse width modulation signal Vv is made unequal. Thereby, positive and negative high-frequency voltages are created. At this time, the average torque becomes zero.

  In other words, in this embodiment, the three phases are switched and controlled so that the average torque becomes zero. Therefore, since control is performed so that the voltage that cancels the induced voltage generated during rotation of the rotating electrical machine is generated from the inverter, the average current of the rotating electrical machine is 0, no brake torque is generated, and no torque is generated during rotation. Can be controlled.

  In this way, the voltage of the smoothing capacitor can be discharged by changing at least one PWM pulse phase in a PWM cycle and energizing the motor current positively or negatively.

  FIG. 7 shows the update timing of ON / OFF of the pulse. (A) is a pulse waveform at the time of normal PWM, (b) is an example of updating at the valley of the carrier Tc, and (c) is an example of updating at the peak of the carrier Tc.

  FIG. 8 shows the relationship between the input waveform and the carrier Tc. (A) is when the maximum value of the input waveform is equal to or greater than the maximum value of the carrier and the minimum value of the input waveform is equal to or greater than the minimum value of the carrier. When the value is equal to or less than the minimum value, (c) shows an example in which the maximum value of the input waveform is equal to or less than the carrier maximum value and the minimum value of the input waveform is equal to or less than the carrier minimum value. In these three cases, the discharge rate is fastest when the maximum value of the input waveform shown in (b) is greater than or equal to the carrier maximum value and the minimum value of the input waveform is less than or equal to the carrier minimum value.

  FIG. 9 shows the experimental results when the pulse shift according to the present embodiment is performed. An enlarged view of the area surrounded by the square in the upper diagram of FIG. 9 is the lower diagram of FIG. 9. In the lower diagram of FIG. 9, each phase current is pulsated by a pulse generated by the line voltage of the motor, which causes motor loss and decreases the capacitor voltage. At the same time, since the average torque is controlled to be zero, the motor remains stopped even during the discharging period.

  FIG. 10 shows the experimental results when the pulse shift according to the present embodiment is performed during the rotation of the rotating electrical machine. An enlarged view of the area enclosed by the square in the upper diagram of FIG. 10 is the lower diagram of FIG. This lower diagram shows that the phase shifted six times during one rotation of the motor is switched, and positive and negative pulses are generated in the vicinity of the switching. The inverter generates a voltage pulse so as to cancel the induced voltage component, and is pulse-shifted so that the capacitor is discharged. Since the pulse shift is controlled so that the average torque becomes zero, no brake torque is generated in the motor.

  11 and 12 show details of the pulse shift. As shown in FIG. 11, in the U phase and the W phase (PWM pulses Vu, Vw), the interval between the center of the ON pulse and the center of the OFF pulse is controlled to be equal. On the other hand, the PWM pulse Vv is pulse-shifted, and the interval between the center of the ON pulse and the center of the OFF pulse becomes unequal. This control is performed when the rotational speed of the electric motor is not more than a predetermined value. In FIG. 11, the pulse shift is performed in the V phase, but the effect of this embodiment can be obtained by performing the pulse shift in at least one phase.

That is, it is only necessary to change the pulse period of at least one phase to be equivalent to twice the carrier period. Specifically, in one carrier period, the phase of the pulse is displaced so as to be away from the peak of the carrier. In the carrier period, the pulse period may be changed to be equivalent to twice the carrier period by displacing the pulse phase so as to be concentrated on the peak of the carrier. As a result, the motor current can be passed positively and negatively, and the accumulated charge in the smoothing capacitor can be discharged.

  Further, when the rotation speed of the electric motor is equal to or less than a predetermined value, the pulse width of at least one phase may be changed for each carrier cycle. Specifically, by changing the pulse so as to increase the width of the pulse in one carrier period and changing the pulse so as to decrease the width of the pulse in the next carrier period, the width of the pulse is changed for each carrier period. Can be changed.

  FIG. 12 shows the U-V phase voltage during the pulse shift, and it can be seen that positive and negative line voltages are generated by the pulse shift. One cycle of the line voltage waveform is twice the carrier cycle.

FIG. 13 is a diagram showing the line voltage during normal PWM control and during pulse shift according to this embodiment. (D) is an example of a line voltage during normal PWM control. One cycle of the line voltage waveform is equal to the carrier cycle, and the line voltage changes at equal intervals within one cycle. (A) is a special case of the line voltage shown in FIG. 12, and the line voltage in the latter half of the cycle is continuously generated. Further, the waveform of the line voltage may be as shown in (b). In this case, the negative line voltage waveform in FIG. 12 appears as a positive line voltage waveform. (C) is a special case of the line voltage shown in (b), and the line voltage in the latter half of the cycle is continuously generated. One period of the line voltage waveform in (a)-(c) is twice the carrier period.
[Example 2]
Another embodiment of the present invention will be described below.

  In the PWM pulse phase control of the present embodiment shown in FIG. 14, the V-phase pulse width modulation signal Vv is shifted to the right with respect to the carrier peak Tc_peak, and positive and negative high-frequency voltages are generated. At this time, the average torque becomes zero.

  In other words, in this embodiment, the three phases are switched and controlled so that the average torque becomes zero. Therefore, since control is performed so that the voltage that cancels the induced voltage generated during rotation of the rotating electrical machine is generated from the inverter, the average current of the rotating electrical machine is 0, no brake torque is generated, and no torque is generated during rotation. Can be controlled.

  In this way, the voltage of the smoothing capacitor can be discharged by changing at least one PWM pulse phase in a PWM cycle and energizing the motor current positively or negatively.

  FIG. 15 shows the update timing of the pulse on / off. (A) is a pulse waveform at the time of normal PWM, (b) is an example in the case of updating in the peaks and valleys of the carrier Tc, and (c) is an example in which the on / off timing of (b) is reversed.

  FIG. 16 shows the relationship between the input waveform and the carrier Tc. (A) is when the maximum value of the input waveform is equal to or greater than the maximum value of the carrier and the minimum value of the input waveform is equal to or greater than the minimum value of the carrier. When the value is equal to or less than the minimum value, (c) shows an example in which the maximum value of the input waveform is equal to or less than the carrier maximum value and the minimum value of the input waveform is equal to or less than the carrier minimum value. In these three cases, the discharge rate is fastest when the maximum value of the input waveform shown in (b) is greater than or equal to the carrier maximum value and the minimum value of the input waveform is less than or equal to the carrier minimum value.

  17 and 18 show details of the pulse shift of this embodiment. As shown in FIG. 17, in the U-phase and W-phase (PWM pulses Vu, Vw), control is performed so that the positions of the center of the ON pulse and the center of the OFF pulse coincide. On the other hand, the PWM pulse Vv is pulse-shifted and controlled so that the center of the ON pulse and the center of the OFF pulse do not coincide with the U phase and the V phase. This control is performed when the rotational speed of the electric motor is not more than a predetermined value. In FIG. 12, the pulse shift is performed in the V phase, but the effect of this embodiment can be obtained by performing the pulse shift in at least one phase.

  That is, when the rotation speed of the electric motor is equal to or lower than a predetermined value, control may be performed so that the phase interval between the center of the on period and the center of the off period of the upper arm pulse of at least one phase is unequal. As a result, the motor current can be passed positively and negatively, and the accumulated charge in the smoothing capacitor can be discharged.

  FIG. 12 shows the U-V phase voltage during the pulse shift, and it can be seen that positive and negative line voltages are generated by the pulse shift. One cycle of the line voltage waveform is twice the carrier cycle.

  FIG. 19 is an example of the line voltage during pulse shift and the line voltage during sine wave PWM control. (A), (b) has shown the example of the line voltage at the time of pulse shift, (c) has shown the example of the line voltage at the time of sine wave PWM control. In (a)-(c), one period of the waveform of the line voltage is equal to the carrier period. In this example, the pulse is turned on and off twice in one cycle, and two pulses appear. Within one period, in (a), the two pulses have the same polarity, but the pulse widths do not match. In (b), the polarities of the two pulses do not match, but the pulse widths match. In (c), the polarity and pulse width of the two pulses are identical. Thus, by performing the pulse shift, the pulse polarity or the pulse width can be made unequal.

  The above-described embodiments and operational effects are merely examples, and the present invention is not limited to the configurations of the above-described embodiments.

136 Battery 138 DC connector 140 Inverter circuit 144 Power switching circuit 150 Series circuit of upper and lower arms 153 and 163 Collector electrode 154 and 164 Gate electrode 155 Emitter electrode 156 and 166 Diode 157 Positive terminal (P terminal)
158 Negative terminal (N terminal)
159 AC terminal 164 Gate electrode 165 Emitter electrode 169 Connection point 170 Control unit 172 Control circuit 174 Driver circuit 180 Current sensor 186 AC power line 188 AC connector 192, 194 Motor generator 193 Rotating magnetic pole sensor 200 Power converter 210 Torque command / current command conversion 220 Current Controller (ACR)
230 Pulse modulator 260 Angular velocity calculator 328, 330 IGBT
500 Capacitor module

Claims (5)

It has a plurality of series circuits in which a switching element for the upper arm and a switching element for the lower arm are connected in series, and includes a power switching circuit that generates AC power by receiving DC power,
When the rotational speed of a state in which the power supply from the battery by cutting the main relay to the inverter is interrupted electric motor is below a predetermined value, changes the cycle of the control pulses of at least one phase twice equivalent carrier period in Rukoto is, the alternating current having a period corresponding to twice the carrier period is supplied to the electric motor, the power converter having a function of discharging charges accumulated in the smoothing capacitor by the alternating current. The power conversion device according to claim 1,
In one carrier period displaces the control pulses of the phase so away from the mountain of the carrier, by the next carrier period to displace the control pulses of the phase so as to concentrate the mountain of the carrier, the period of the control pulse a power conversion device for varying the equivalent to twice the carrier period. It has a plurality of series circuits in which a switching element for the upper arm and a switching element for the lower arm are connected in series, and includes a power switching circuit that generates AC power by receiving DC power,
When the rotational speed of a state in which the power supply from the battery by cutting the main relay to the inverter is interrupted electric motor is below a predetermined value, by changing the width of the control pulses of at least one phase for each carrier period Rukoto A power converter having a function of passing an alternating current having a period corresponding to twice the carrier period to the motor and discharging the accumulated charge of the smoothing capacitor by the alternating current . The power conversion device according to claim 3,
In one carrier period alters the control pulse to increase the width of the pulses, by the next carrier period changing the control pulse so as to reduce the width of the control pulses, the width of the control pulse power converter that changes for each of the carrier cycle. It has a plurality of series circuits in which a switching element for the upper arm and a switching element for the lower arm are connected in series, and includes a power switching circuit that generates AC power by receiving DC power,
When the rotational speed of the motor is below a predetermined value, by performing a pulse shift control in so that a phase interval unequal of centers of OFF period of the ON period of the control pulse of the upper arm of at least one phase A power conversion device having a function of passing an alternating current having a period corresponding to twice the carrier period to the motor and discharging the accumulated charge of the smoothing capacitor by the alternating current .