JP5821026B2 - Charger - Google Patents

Description

  The present invention relates to a charging device for charging a secondary battery.

  In general, a charging device for charging a secondary battery is required to change an output voltage and an output current over a wide range. That is, since the terminal voltage of the secondary battery changes greatly according to the state of charge, the charging device is required to change the output voltage over a wide range from a low voltage to a rated voltage according to the terminal voltage of the secondary battery. . In addition, when the secondary battery is rapidly charged, the charging current becomes large, and when the secondary battery is auxiliary charged, the charging current becomes weak. Therefore, the charging device has a wide range of output currents according to the charging current of the secondary battery. It is required to be changed over time.

  This type of charging device mainly employs a configuration using a series control type constant voltage power supply and a configuration using a switching power supply. In particular, a charging device using a switching power supply is widely used because it has a higher power conversion efficiency and can be reduced in size compared to a charging device using a series-controlled constant voltage power supply.

  By the way, since the resonance type switching power supply has good circuit efficiency and suppresses noise, it is considered that when the resonance type switching power supply is used for the charging device, a small and low cost charging device can be provided.

  As a charging device using a resonance type switching power supply, for example, a configuration shown in FIG. 13 has been proposed. The illustrated charging device includes two switching elements Q11 and Q12, resonance inductors L11, L12, L13, and L14, and resonance capacitors C11 and C12. Inductors L11 and L12 are magnetically coupled to inductors L15 and L16. The electric power extracted from the inductors L15 and L16 is rectified and smoothed by the diodes D11 and D12, the inductor L17, and the capacitor C13, and used for charging the secondary battery B1 (see Patent Document 1). The charging device described in Patent Document 1 is configured using a push-pull type switching power supply, and the switching elements Q11 and Q12 are controlled to be turned on and off by the control unit CN11.

  The control unit CN11 adjusts the output to the secondary battery B1 by controlling the dead-off time for turning off the two switching elements Q11 and Q12 and the drive frequency for turning on and off the switching elements Q11 and Q12. . That is, the control unit CN11 adjusts the dead-off time according to the magnitude of the charging current, operates in the resonance mode when the charging current is large, and operates in the non-resonance mode when the charging current is small as in trickle charging. . Further, the controller CN11 changes the drive frequency while keeping the dead-off time constant when changing the output of the charging device, and increases the drive frequency when reducing the output.

  When the charging current is reduced, the controller CN11 lengthens the dead-off time when both the switching elements Q11 and Q12 are turned off, and performs switching in a region where the resonance voltage applied to the switching elements Q11 and Q12 is minimized. In this operation, a pseudo resonance operation is performed. Further, constant voltage charging is performed near the full charge of the secondary battery B1, and a plurality of non-resonant modes having different dead-off times are set during constant voltage charging, and any non-resonant mode is set according to the charging current. By selecting the resonance mode, an increase in switching loss is suppressed.

  As a charging device using a resonant switching power supply, for example, a configuration shown in FIG. 14 has also been proposed. This charging apparatus employs a half-bridge type switching power supply having a series circuit of two switching elements Q21 and Q22. A series circuit of a capacitor C21, an inductor L21, and a primary winding N12 of the transformer T2 is connected in parallel to the switching element Q22. Further, a capacitor C22 is connected in parallel to the primary winding N12 of the transformer T2. The electric power extracted from the secondary winding N22 of the transformer T2 is rectified and smoothed by the diodes D21 and D22 and the capacitor C23 and used for charging the secondary battery B2 (see Patent Document 2). The capacitors C21 and C22, the inductor L21, and the transformer T2 constitute a resonance circuit. Switching elements Q21 and Q22 are alternately turned on and off by control circuit CN21.

  Control circuit CN21 decreases the driving frequency of switching elements Q21 and Q22 in a region where the charging current is large, and increases the driving frequency of switching elements Q21 and Q22 in a region where the charging current is small. The capacitor C21 and the inductor L21 are set so as to correspond to a low driving frequency in a region where the charging current is large, and the capacitor C22 is set so as to correspond to a high driving frequency in a region where the charging current is small. This configuration adjusts the charging current by changing the driving frequency. Further, by having a plurality of resonance points, it is possible to maintain a resonance state over a wide range of charging currents.

  In addition to the above-described configuration, the resonance type switching power supply includes a composite resonance type switching power supply in which current resonance and voltage resonance by partial resonance are combined as shown in FIG. The illustrated example is a half bridge type having a series circuit of two switching elements Q31 and Q32, and the configuration of the resonance circuit is different from the configuration shown in FIG. That is, the capacitor C31 is connected in parallel to the low-side switching element Q32, and a series circuit of the inductor L31, the primary winding N31 of the transformer T3, and the capacitor C32 is connected in parallel to the capacitor C31. In this configuration, the capacitors C31 and C32, the inductor L31, and the transformer T3 form a resonance circuit.

  Further, the electric power extracted from the secondary winding N32 of the transformer T3 is rectified and smoothed by the diodes D31 and D32 and the capacitor C33. Unlike Patent Documents 1 and 2, the configuration example shown in FIG. 15 shows a configuration in which DC power is supplied to the load LD instead of the secondary battery. In the figure, a control circuit for controlling on / off of the switching elements Q31 and Q32 is omitted.

  An operation example of the configuration shown in FIG. 15 is shown in FIG. 16A and 16B show the gate-source voltage Vgs of the switching elements Q31 and Q32. FIGS. 16C and 16D show the drain-source voltage Vds (solid line) and the drain of the switching elements Q31 and Q32. Current Id (broken line) is shown. Further, FIG. 16E shows the voltage across the capacitor C32, and FIG. 16F shows the current flowing through the diodes D31 and D32. In this operation, a dead-off time in which the two switching elements Q31 and Q32 are both turned off is provided.

  The driving frequency of the switching elements Q31 and Q32 is set lower than the resonance frequency of the series resonance circuit constituted by the capacitor C32, the inductor L31, and the transformer T3. With this configuration, the operation is a composite resonance in which current resonance and voltage resonance are combined, and can be operated with high efficiency and low noise.

JP 2006-230104 A JP 2010-263683 A

  The charging device disclosed in Patent Document 1 performs constant voltage charging in which a charging current is a constant current in a region where the charging current is large, and an output voltage of the charging device is a constant voltage in a region where the charging current is small. The configuration described in Patent Document 1 operates from a resonance operation, a pseudo resonance operation, and a plurality of resonance modes in accordance with the state of charging so as to achieve high efficiency and low noise in all areas of charging. There is a problem that the circuit configuration becomes complicated. In addition, since a push-pull type switching power supply that is a voltage type converter is used, the transformer T1 is large when a large charging current is required as in the case where the secondary battery B1 having a large capacity is rapidly charged. There is a possibility of becoming.

  Since the charging device shown in Patent Document 2 is a half-bridge type, the transformer T2 can be made smaller than when a push-pull type configuration is adopted. However, when performing rapid charging of the secondary battery B2 having a large capacity, it is difficult to maintain the resonance operation only by controlling the driving frequency over a wide range of charging conditions. Further, the turn ratio between the primary winding N21 and the secondary winding N22 of the transformer T2 is designed on the assumption that the battery voltage is the highest, so that an excessive charging current flows when the battery voltage is low. There is a fear.

  Further, as in the configuration described in Patent Document 2, the half-bridge type switching power supply using series resonance has a drive frequency higher than the resonance frequency of the resonance circuit in order to avoid stress of the switching elements Q21 and Q22. It is necessary to operate with. Therefore, if a predetermined charging current is made to flow in an area where the remaining capacity of the secondary battery B2 is low and the battery voltage is low, the operating frequency becomes very high, resulting in a decrease in efficiency due to an increase in switching loss. Noise may increase.

  In the half-bridge type switching power supply using the complex resonance shown in FIG. 15, the input voltage Vin and the output voltage Vout have a relationship of Equation 1 where the capacitance of the capacitor C32 is Ci and the inductance of the inductor L31 is Lr. It is known.

ただし、ｎはトランスの巻数比、ＬｐはトランスＴ３の励磁インダクタンス、τは図１６に示した電力非伝達期間である。

However, n is the turns ratio of the transformer, Lp is the exciting inductance of the transformer T3, and τ is the power non-transmission period shown in FIG.

  As can be seen from Equation 1, in order to change the output voltage Vout, the power non-transmission period τ may be changed. That is, the output voltage Vout changes by changing the drive frequency of the switching elements Q31 and Q32. However, since the charging device is required to change the output voltage and output current over a wide range as described above, it is not easy to maintain the optimum resonance condition in all regions of the output voltage and output current. Absent.

  The present invention is a charging device that enables quick charging of a secondary battery, and uses a resonance operation for a wide range of charging condition changes while using a relatively simple circuit configuration, thereby achieving high efficiency and low noise. It is another object of the present invention to provide a charging device that can reduce the size of the transformer.

In order to achieve the above object, a charging device according to the present invention includes a series circuit of two switching elements connected between both ends of a DC power source that is an input power source, a charge control circuit that controls on / off of the switching elements, Rectifying and smoothing the output of the resonance circuit for complex resonance operation, which is connected in parallel to one of the switching elements and includes a series circuit of a primary winding and a capacitor of the inductor, the transformer, and the secondary winding of the transformer. A rectifying / smoothing circuit that supplies a charging current to the secondary battery, and a state detection circuit that detects a battery voltage that is a terminal voltage of the secondary battery and a charging current that flows to the secondary battery, and the charging control circuit includes an inductor and a transformer While controlling the on / off of the two switching elements while forming a power non-transmission period in a frequency range higher than the resonance frequency determined by the primary winding and the capacitor, Switching is performed so that the charging current is constant until the battery voltage detected by the state detection circuit reaches the specified target voltage after charging starts, and charging is performed at a constant voltage when the battery voltage reaches the target voltage. The switching voltage according to the battery voltage is provided in a period in which the on / off of the element is controlled and the charging current is a constant current, and two switching elements are provided in the first control region in which the battery voltage is lower than the switching voltage. of by adjusting the duty ratio of the on-period, varied as the battery voltage is reduced low enough time ratio, in the second control region the battery voltage is the switching voltage above the two switching elements It is characterized in that the driving frequency of the switching element is lowered as the battery voltage is increased while keeping the time ratio constant so that the ON periods are equal.

In this charging device, the charging control circuit, Te first control region odor, two one ON period of one switching element of the switching element kept constant, the ON period of the remaining one of the switching elements It is preferable to control the duty ratio by adjusting .

In this charging device, the charging control circuit, in the first control region, in the region the battery voltage state detecting circuit detects is prescribed threshold voltage or more, two of the first switching element of the switching element maintaining the on-time constant, and adjusting the on-period of the second switching element, in the region the battery voltage is less than the threshold voltage, the coercive Chi the oN period of the second switching element to be constant, and the it is preferable to adjust the on-period of the first switching element.

  In this charging device, the charging device is suitable for a plurality of types of secondary batteries having different battery voltages at the time of full charge, and the charging control circuit is configured to perform charging determined for each specification of the secondary battery in the second control region. It is preferable to change the drive frequency of the switching element so that a current flows.

  In this charging apparatus, after the battery voltage reaches the target voltage, the charging control circuit reduces the charging current by increasing the driving frequency with the lapse of time while the ON periods of the two switching elements are equal, From the time when the power non-transmission period disappears, it is preferable to keep one on-period of the switching element constant and to shorten or lengthen the other on-period of the switching element as the charging current decreases.

  In this charging apparatus, after the battery voltage reaches the target voltage, the charging control circuit reduces the charging current by increasing the driving frequency with the lapse of time while the ON periods of the two switching elements are equal, From the time when the power non-transfer period disappears, one switching element's on-period is kept constant, and the smaller the charging current, the shorter or longer the other switching element's on-period, and the charging current is less than the specified threshold current. In a certain region, it is preferable to keep the other ON period of the switching element constant and to lengthen or shorten the one ON period of the switching element.

  In this charging apparatus, after the battery voltage reaches the target voltage, the charging control circuit intermittently performs on / off operation of the switching element to provide a period for flowing a charging current and a period for stopping the charging, and charging with time. It is preferable to reduce the charging current by extending the period for stopping the current.

  In this charging device, the switch inserted in the path for rectifying and smoothing the output from both ends of the secondary winding of the transformer, and turning off the switch until the battery voltage exceeds the specified voltage, the secondary winding of the transformer It is preferable to include a switch control unit that rectifies and smoothes the output from the provided intermediate tap and turns on the switch when the battery voltage exceeds a specified voltage to rectify and smooth the output from both ends of the secondary winding of the transformer. .

  In this charging apparatus, the transformer is preferably a leakage transformer, and the inductor preferably uses the leakage inductance of the leakage transformer.

  In this charging apparatus, the secondary battery is preferably a lithium ion battery.

  According to the configuration of the present invention, in a charging device that enables quick charging of a secondary battery, the resonance operation is utilized for a wide range of charging condition changes while using a relatively simple circuit configuration. There is an effect that the efficiency and low noise can be achieved and the transformer can be miniaturized. In particular, using a half-bridge type switching power supply that uses complex resonance, it is possible to perform resonance operation in a region where charging power is high and to perform partial resonance operation even in a region where charging power is low. It is possible to provide a charging device. Moreover, since it is a half-bridge type, the transformer is small and contributes to cost reduction.

It is a circuit diagram showing an example of a circuit of an embodiment. It is a circuit diagram which shows the other circuit example of embodiment. FIG. 3 is an operation explanatory diagram illustrating the first embodiment. It is operation | movement explanatory drawing which shows control same as the above. It is operation | movement explanatory drawing which shows the voltage waveform and current waveform of the principal part in the same as the above. FIG. 9 is an operation explanatory diagram illustrating control of Embodiment 2. It is operation | movement explanatory drawing which shows the voltage waveform and current waveform of the principal part in the same as the above. FIG. 10 is an operation explanatory diagram illustrating the third embodiment. FIG. 11 is an operation explanatory diagram illustrating a voltage waveform and a current waveform of main parts in the fourth embodiment. It is operation | movement explanatory drawing which shows the other operation example same as the above. (A) and (b) are the circuit diagrams of the principal part which shows Embodiment 5. FIG. It is operation | movement explanatory drawing which shows the voltage waveform and current waveform of the principal part in the same as the above. It is a circuit diagram which shows a conventional structure. It is a circuit diagram which shows another conventional structure. It is a circuit diagram which shows the example of a half-bridge type switching power supply. FIG. 16 is an operation explanatory diagram of the circuit shown in FIG. 15.

(Embodiment 1)
The charging device described below as an embodiment includes a half-bridge converter as a main circuit, as shown in FIG. 1 or FIG.

  The charging device shown in FIG. 1 includes a diode bridge DB and a smoothing capacitor C2, and obtains an input power supply by full-wave rectifying an AC power supply AC such as a commercial power supply with the diode bridge DB and smoothing with the smoothing capacitor C2. .

  A series circuit of two switching elements Q1, Q2 is connected in parallel to the smoothing capacitor C2. For example, a power MOSFET is used for switching elements Q1, Q2. A series circuit of an inductor Lr, a primary winding N1 of a transformer T, and a capacitor Cr is connected in parallel to both ends (drain-source) of one switching element Q2. Inductor Lr, primary winding N1 and capacitor Cr constitute a resonant circuit. Further, the drain-source capacitance Cv of the switching element Q2 is also used as a part of the resonance circuit. In addition to the drain-source capacitance Cv, a resonance capacitor (not shown) may be connected in parallel to the drain-source to form a resonance circuit.

  The secondary winding N2 of the transformer T has a center tap, and the output of the secondary winding N2 is rectified and smoothed by a rectifying and smoothing circuit using two diodes D1, D2 for rectification, a smoothing capacitor C4, and an inductor L1. And supplied to the secondary battery B. Although the secondary battery B is assumed to be a lithium ion battery, it may be another secondary battery such as a nickel metal hydride battery or a nickel cadmium battery. Here, the turn ratio between the primary winding N1 and the secondary winding N2 of the transformer T is set so that a sufficiently large charging current is obtained in the vicinity of the rated voltage of the secondary battery B.

  In a state where the secondary battery B is connected to the charging device, a series circuit of the inductor L1, the secondary battery B, and the current detection resistor R1 is formed at both ends of the smoothing capacitor C4, and the charge flowing through the secondary battery B A current is detected as a voltage across the resistor R1. In addition, a voltage detection voltage dividing circuit composed of a series circuit of two resistors R2 and R3 is connected in parallel with the secondary battery B, and the terminal voltage of the secondary battery B (hereinafter referred to as “battery voltage”). ) Is detected as the voltage across the resistor R3. That is, the state detection circuit is configured by the resistors R1, R2, and R3.

  The illustrated charging device includes a capacitor C5 connected in parallel with the smoothing capacitor C2, and further includes a capacitor C6 connected between a connection point between the switching element Q2 and the capacitor Cr and a center tap of the transformer T. . An AC power supply AC is connected to the AC input terminal of the diode bridge DB via a fuse F, a noise prevention capacitor C1, a line filter FT, and the like.

  Information on the charging current detected as the voltage across the resistor R1 and information on the battery voltage detected as the voltage across the resistor R3 are input to the charging control circuit CN. The charging control circuit CN determines the timing for turning on / off the switching elements Q1, Q2 based on the charging current and the battery voltage, and outputs a timing signal for turning on / off the switching elements Q1, Q2. The timing signal is input to the drive circuit DR via the photocouplers PC1 and PC2 for insulation, and the drive signal applied to the switching elements Q1 and Q2 is generated in the drive circuit DR. The drive signal controls on / off of the switching elements Q1, Q2 according to an instruction by the timing signal.

  The charging device shown in FIG. 2 has a configuration in which the transformer T is replaced with a leakage transformer T1 with respect to the main circuit of FIG. 1, and the inductor L1 is omitted by using a leakage inductance for resonance. Other configurations are the same as those of the charging apparatus shown in FIG.

  Below, operation | movement of the charging device shown in FIG. 1, FIG. 2 is demonstrated. As shown in FIG. 3, the present embodiment employs a charging method corresponding to constant voltage / constant current charging. That is, the charging control circuit CN sets the charging current to a constant current when the secondary battery B is at a low voltage due to discharging, and performs supplementary charging with a weak current when the battery voltage reaches the specified target voltage Vt. Further, on / off of the switching elements Q1, Q2 is controlled. The target voltage Vt is set to the rated voltage of the secondary battery B, for example. In the operation example shown in FIG. 3, a constant charging current is applied to the secondary battery B that has been discharged and the terminal voltage has dropped considerably, and after reaching the target voltage Vt, the charging current is gradually reduced. Show.

  Incidentally, the charging device shown in FIGS. 1 and 2 uses a resonance type switching power source that performs soft switching using a resonance circuit. More specifically, it has a composite resonance type configuration in which a plurality of resonance elements are combined with a half-bridge type switching power supply. The switching power supply having this configuration can be operated with high efficiency by appropriately adjusting the on / off timing of the switching elements Q1, Q2. However, as shown in FIG. 3, since the battery voltage changes greatly in the region where the charging current is a constant current, on / off of the switching elements Q1, Q2 is controlled so that soft switching is performed in the entire range of the battery voltage. It is difficult to do.

  Therefore, in the present embodiment, the region where the charging current is a constant current is divided into a plurality of control regions A1 and A2 (2 in the illustrated example) as shown in FIG. The on-duty is changed according to the control areas A1 and A2. The control areas A1 and A2 are divided by the magnitude of the battery voltage and the specified switching voltage Vs. For example, the switching voltage Vs is defined such that the power non-transmission period τ is substantially 0 in the relationship shown in Equation 1, and is set to be slightly larger than Vin / 2n. Hereinafter, the control region A1 in which the battery voltage is less than the switching voltage Vs is referred to as a first control region, and the control region A2 in which the battery voltage is equal to or higher than the switching voltage Vs is referred to as a second control region.

  Since the horizontal axis of FIG. 4 is the battery voltage and the charging current is a constant current, the charging power increases in proportion to the battery voltage. In FIG. 4, characteristic a indicates the driving frequency of the switching elements Q1 and Q2, characteristic b indicates the time ratio of the ON periods of the switching elements Q1 and Q2, and characteristic c indicates the efficiency of the charging device. The ratio of the ON periods of the switching elements Q1 and Q2 is the ratio of the ON period of the high-side switching element Q1 to the total of the ON periods of the switching elements Q1 and Q2, and is expressed as a percentage. Therefore, when the ON periods of both switching elements Q1 and Q2 are equal to 1: 1, the duty ratio is 50%.

  In the first control region A1, the charge control circuit CN sets the time ratio of the ON period to less than 50% and increases the time ratio as the battery voltage is higher. In the second control region A2, the charge control circuit CN increases the time ratio. First, keep the time ratio of the on period constant at 50%. In addition, the charge control circuit CN sets the drive frequency lower as the battery voltage is higher in the second control region A2. In FIG. 4, also in the first control region A1, the driving frequency decreases as the battery voltage increases, but this is because the driving frequency decreases as a result of increasing the time ratio and approaching 50%. There is no adjustment of the drive frequency in the first control region A1.

  In the above-described operation, changing only the drive frequency in accordance with the battery voltage in the second control region A2 corresponds to changing the power non-transfer period τ in Equation 1. That is, when the drive frequency is lowered, the power non-transmission period τ is increased, and it can be seen from Equation 1 that the output voltage rises.

  On the other hand, in the second control region A2, when the battery voltage is in the vicinity of the switching voltage Vs, the adjustment range of the power non-transmission period τ is reduced and the drive frequency only is maintained if the on-period time ratio is maintained at 50%. Even if is adjusted, it is difficult to adjust the output voltage. That is, when the battery voltage is close to the switching voltage Vs, it is difficult to maintain complex resonance.

  Therefore, in the first control region A1, the charging control circuit CN maintains the charging current at a constant current by changing the on-period of the high-side switching element Q1 to change the on-period time ratio. That is, in the first control region A1, the charging control circuit CN shortens the ON period of the switching element Q1 as the battery voltage is lower. With this operation, in the first control region A1, the drive frequency increases as the battery voltage decreases. In the first control region A1, the case where the on-period of the high-side switching element Q1 is changed while the on-period of the low-side switching element Q2 is kept constant has been described. However, the on-period of the switching element Q1 is constant. Even if the ON period of the switching element Q2 is changed while maintaining the same, the same result can be obtained.

  FIG. 5 shows the drain-source voltage Vds (solid line) and the drain current Id (broken line) of the switching elements Q1, Q2 in the first control region A1 and the second control region A2. Therefore, in each switching element Q1, Q2, a period in which voltage Vds is not 0 corresponds to an off period. In FIG. 5, the dead-off time as a period during which both switching elements Q1, Q2 are turned off is omitted. The dead-off time does not need to be changed in the first control area A1 and the second control area A2. The operations shown in (1) to (5) in FIG. 5 correspond to the battery voltage regions shown in (1) to (5) in FIG.

  In the first control region A1 in which the battery voltage is lower than the switching voltage Vs, the on period of the switching element Q1 is shorter than the on period of the switching element Q2, as shown in FIGS. The time ratio becomes smaller. Therefore, in the first control region A1, the drain current Id of the high-side switching element Q1 has a triangular wave shape, and the peak is relatively large. However, the charging control circuit CN sets the on period relatively long to the extent that the power non-transmission period τ occurs for the low-side switching element Q2, and allows the resonance current to flow through the drain current Id.

  As apparent from FIGS. 5 (1) and (2), in the first control region A1, the ON period of the low-side switching element Q2 is kept substantially constant, and the switching voltage Q1 of the switching element Q1 increases as the battery voltage approaches the switching voltage Vs. The on period is long. By this operation, the charging current from the charging device to the secondary battery B is maintained at a constant current. Further, as the battery voltage increases, the slope of the drain current Id of the switching element Q1 becomes gentler and the peak is reduced.

  As described above, in the first control region A1, since the resonance current cannot flow in the drain current Id of one switching element Q1, the efficiency is lowered, but the first control region A1 has a relatively small charging power. Therefore, there is little loss and the effect of the decrease in efficiency is relatively small.

  In the second control region A2 in which the battery voltage is equal to or higher than the switching voltage Vs, as shown in FIGS. 5 (3), (4), and (5), the ON periods of the switching elements Q1 and Q2 are equal and the time ratio is kept constant. The drive frequency changes. The ON period of the switching elements Q1, Q2 is set so that a resonance current flows through the drain current Id of the switching elements Q1, Q2, and the drive frequency is set lower as the battery voltage increases. That is, the ON period of switching elements Q1, Q2 becomes longer as the battery voltage is higher.

  In the second control region A2, as the battery voltage increases, the drive frequency decreases and the on-periods of the switching elements Q1, Q2 become longer. That is, as the battery voltage increases, the power non-transmission period τ in Equation 1 is extended, and the output voltage increases. The power non-transmission period τ corresponds to the period of the shoulder portion after the resonance current flows in the drain current Id shown in FIGS. In the second control region A2, the charging device is operated in the composite resonance mode, and a relatively high efficiency is maintained.

  As described above, in the configuration of the present embodiment, when the secondary battery B is charged with a constant current, in the first control region A1 where the battery voltage is less than the switching voltage Vs, the switching element Q1 and the switching element Q2 One ON period is controlled. In the second control region A2 where the battery voltage is equal to or higher than the switching voltage Vs, the drive frequency is controlled while maintaining the duty ratio at 50%. By such an operation, the secondary battery B is charged with a constant current while maintaining the charging current at a desired current value, and the resonance current due to the composite resonance is continuously supplied for at least a part of the period regardless of the battery voltage. And maintain a relatively high efficiency.

  As shown in FIG. 4, in the first control region A1, it is considered that the efficiency decreases as the battery voltage decreases because of the following causes. That is, it can be considered that the peak value of the drain current Id of the switching element Q1 increases and the loss at the switching elements Q1, Q2, transformer T, diodes D1, D2, etc. increases due to the increase in driving frequency. . However, the efficiency is reduced in the first control region A1 where the charging power is relatively small, and high efficiency can be maintained in the second control region A2 where the charging power is large. Therefore, the ratio of the period in which the power loss is large in the process of charging the secondary battery B is small, and as a result, the power loss in the charging process is suppressed.

  In the above-described embodiment, the ON period of the switching element Q1 is changed in the first control region A1, but the same operation is performed even if the ON period of the switching element Q2 is changed.

(Embodiment 2)
In the first embodiment, two regions of a first control region A1 and a second control region A2 are set based on the battery voltage. However, as shown in FIG. The first control area A1 in FIG. 3 is further divided into two control areas A11 and A12, which is different from the first embodiment. The first control region A1 is the same as the first embodiment in that the time ratio of the ON period of the switching elements Q1, Q2 is changed. However, in the first embodiment, the ON period is changed only for one of the two switching elements Q1 and Q2. However, in the present embodiment, different switching elements Q1 and Q2 are used in the control regions A11 and A12. Vary the on period.

  Control areas A11 and A12 are divided based on the battery voltage. The control region A11 is a region where the battery voltage is less than a prescribed threshold voltage Vu (<Vs), and the control region A12 is a region where the battery voltage is equal to or higher than the threshold voltage Vu. The threshold voltage Vu is appropriately determined based on the efficiency of the charging device.

  In the control region A12 where the battery voltage is equal to or higher than the threshold voltage Vu, the charging control circuit CN is turned on of one of the switching elements Q1, Q2 according to the battery voltage, as in the first control region A1 of the first embodiment. The duty ratio of the on period is changed by changing the period. For example, the lower the battery voltage, the shorter the on-period of the switching element Q1, thereby reducing the duty ratio. As a result, the lower the battery voltage, the higher the driving frequency. On the other hand, in the control region A11 where the battery voltage is less than the threshold voltage Vu, the charging control circuit CN changes the other ON period of the switching elements Q1 and Q2 in accordance with the battery voltage and changes the time ratio of the ON period. Let That is, if the ON period of the switching element Q1 changes in the control region A12, the ON period of the switching element Q2 changes in the control region A11.

  Here, in the control region A12, the lower the battery voltage, the shorter the on-period of the switching element Q1, thereby reducing the time ratio. However, in the control region A11, the lower the battery voltage, the longer the on-period of the switching element Q2. The duty ratio of the switching element Q1 becomes relatively small.

  An operation example of this embodiment is shown in FIG. FIG. 7 shows the drain-source voltage Vds (solid line) of the switching elements Q1 and Q2 and the drain current Id of the switching elements Q1 and Q2, as in FIG. Also, (1), (2), and (3) in FIG. 7 correspond to (1), (2), and (3) in FIG.

  In the control region A12, as shown in FIGS. 7 (2) and (3), the on period of the switching element Q2 is not changed, but the on period of the switching element Q1 is changed, and the on period is shortened as the battery voltage is lower. ing. Accordingly, as shown in FIG. 6, the lower the battery voltage, the smaller the on-period time ratio. In the control region A12, the on-period of the switching element Q1 is shortened, so that the drive frequency increases as the battery voltage decreases.

  On the other hand, in the control region A11, as shown in FIGS. 7A and 7B, the ON period of the switching element Q1 is changed without changing the ON period of the switching element Q1, and the ON period is increased as the battery voltage is lower. It is long. Therefore, also in the control region A11, the lower the battery voltage, the smaller the time ratio of the switching element Q1, and the longer the ON period of the switching element Q2, the lower the battery voltage, the lower the drive frequency. ing.

  As described in the first embodiment, one factor that the efficiency decreases as the battery voltage decreases in the first control region A1 is an increase in loss due to an increase in the driving frequency of the switching elements Q1 and Q2. . Therefore, the loss is suppressed by controlling the drive frequency to be lowered in the region where the battery voltage is low as in the present embodiment, and as a result, the decrease in efficiency is suppressed. In addition, the time ratio of the on-period changes in the first control region A1 by the operation of the present embodiment, so that the secondary battery B can be charged with a constant current regardless of the battery voltage.

  Similarly to the first embodiment, the switching element Q1 and the switching element Q2 may be replaced with each other in the operation of the present embodiment. Other configurations and operations are the same as those in the first embodiment, and thus description thereof is omitted.

(Embodiment 3)
Although the above-described embodiment assumes that only one type of secondary battery B is supported, the charging device may be required to charge a plurality of types of secondary batteries having different characteristics. For example, a single charging device may have to deal with a plurality of types of battery packs having different battery voltages when fully charged.

  This embodiment demonstrates the structure for respond | corresponding to the multiple types of secondary battery from which the characteristic at the time of charge respond | corresponds with one charger. Now, assuming three types of battery packs (secondary batteries) having different battery voltages when fully charged, the target voltages for charging each battery pack are Va, Vb, and Vc, respectively (Va> Vb> Vc). . In addition, the charging currents Ia, Ib, and Ic of each battery pack are controlled as indicated by broken lines in FIG. In the illustrated example, when the charging currents Ia, Ib, and Ic are constant currents, the relationship is Ia> Ib> Ic. When the charging currents Ia, Ib, and Ic change as shown in FIG. 8 as the charging time elapses, the battery voltage changes as shown by a solid line in FIG. In the illustrated example, the charging currents Ia, Ib, and Ic at a constant current increase as the target voltages Va, Vb, and Vc increase.

  In these three types of battery packs, the charging current Ia for the battery pack having the highest battery voltage at the time of full charge is the largest. Therefore, the design conditions of the charging device are determined so that the output current of the charging current Ia can be sufficiently obtained. . In addition, the design condition is determined so that the resonance current of the composite resonance flows even for the battery pack having the lowest battery voltage when fully charged. Design conditions for obtaining an appropriate output current are determined by the winding design of the transformer T (see FIGS. 1 and 2). In order to operate in the same manner as in the above-described embodiment, the switching voltage Vs is determined to be lower than the lowest target voltage Vt.

  Similar to the first embodiment, the present embodiment is divided into a first control region A1 and a second control region A2 depending on the magnitude of the battery voltage and the switching voltage Vs. Similarly to the first embodiment, in the second control region A2, the on-time time ratio is made constant, and the drive frequency is lowered as the battery voltage increases. However, it is necessary to change the drive frequency according to the target voltages Va, Vb, and Vc for each battery pack. Therefore, it is necessary to determine the type of the battery pack, grasp the battery voltage when fully charged, and set the drive frequency according to the battery pack.

  In order to determine the type of battery pack, the battery pack voltage is detected by connecting a resistance to the battery pack to detect the resistance value that is mounted to indicate the number of secondary battery cells in the battery pack. Any one of the techniques for measuring Further, the charge control circuit CN includes a data table for determining the drive frequency according to the battery voltage at the time of full charge detected by the number of cells of the battery pack. If the data table is set so as to correspond to the battery pack of 2 to 8 cells, it is possible to cope with not only the current battery pack but also future battery pack development.

  As in the present embodiment, the winding of the transformer T is appropriately designed, the switching voltage Vs is appropriately set, and the drive frequency is changed according to the specifications of the battery pack, so that the battery voltage is at least the target voltage Va. , Vb, and Vc can be operated in a complex resonance range. That is, it becomes possible to deal with a plurality of types of battery packs having different specifications with one charging device. Other configurations and operations are the same as those in the first embodiment, and the technique described in the second embodiment can also be adopted in this embodiment.

(Embodiment 4)
In the first to third embodiments, the control in the region where the charging current is a constant current has been described. However, in the present embodiment, the battery voltage is set to the target voltage by charging with a constant current in the configuration shown in FIGS. The control when performing supplementary charging (constant voltage charging) after reaching Vt will be described.

  During the period in which the secondary battery B is charged with the charging current as a constant current, the on-period time ratio is constant and the drive frequency is changed in the second control region A2 where the battery voltage is equal to or higher than the switching voltage Vs. . On the other hand, during the period of supplementary charging, the charging current changes greatly. For example, it is necessary to change the charging current to about 1/50 compared with charging at a constant current. I can't. In the present embodiment, the charging current is changed by changing the time ratio of the ON period in the period of supplementary charging.

  The operations shown in (6) to (8) in FIG. 9 correspond to the battery voltage regions shown in (6) to (8) in FIG. In FIG. 9, the solid line indicates the drain-source voltage Vds of the switching elements Q1 and Q2, and the broken line indicates the drain current Id. As is clear from FIG. 9, in the region where the charging current is large as shown in (6), high efficiency is obtained by using the complex resonance while maintaining the duty ratio at 50%. In the charging current shown in (7), both the ON periods of the switching elements Q1 and Q2 are shortened to increase the driving frequency, and the power non-transmission period τ is shortened to reduce the charging current. In the region where the charging current is small as shown in (8), only the ON period of the switching element Q1 is shortened to reduce the duty ratio, thereby reducing the charging current.

  In the operation of the present embodiment, during the period of performing supplementary charging, in the region where the charging current is large, the driving frequency is controlled while maintaining the time ratio of both switching elements Q1 and Q2 at approximately 50%, and the region where the charging current is small. Then, the ON period of one switching element Q1 is shortened. By this operation, supplementary charging can be performed at a constant voltage, and in a region where the charging current is large, operation is performed with high efficiency by utilizing composite resonance. Further, although loss occurs in a region where the charging current is small, since the charging power is small, the ratio of loss in the charging process is small, and high efficiency can be expected as a whole.

  By the way, as explained in the second embodiment, the duty ratio of the on period is reduced by shortening one on period of the switching elements Q1 and Q2 and lengthening the other on period. Therefore, in the region (8) in FIG. 3, as described above, the duty ratio is reduced by shortening the ON period of the switching element Q1, and in the region (9) in FIG. 3, the ON period of the switching element Q2 is reduced. It is desirable to perform control to make the duty ratio of the switching element Q1 relatively small by increasing the length. The transition from the operation of shortening the ON period of the switching element Q1 to the operation of extending the ON period of the switching element Q2 defines the threshold current with respect to the charging current, and the ON period of the switching element Q2 is lengthened below the threshold current. That's fine. This operation prevents the drive frequency from becoming too high during the period in which the charging current is weak, resulting in a reduction in loss.

  In order to substantially reduce the charging current during the period of performing supplementary charging, the charging current may be intermittently supplied. FIG. 10 shows an operation example in which the charging current is passed intermittently. (6) to (8) in FIG. 10 correspond to (6) to (8) in FIG. 3. In FIG. 10, the period T1 is a period during which the charging current is flowing, and the period T2 is the period in which the charging current is stopped. Represents the period. The vertical axis in FIG. 10 corresponds to the on / off of the switching elements Q1, Q2. As shown in (6) of FIG. 3, during the period when the charging current is large, the charging current is continuously supplied as shown in (6) of FIG. 10, but as shown in (8) of FIG. In a period where the current is small, the charging current is intermittently passed as shown in FIG. Moreover, the substantial charging current becomes smaller as the period T1 during which the charging current flows and the period T2 during which the charging current does not flow become longer.

  By adopting the operation shown in FIG. 10, even if the charging current is a weak current, the operation shown in (6) of FIG. Can achieve high efficiency and suppress the generation of noise. That is, high-efficiency and low-noise operation is possible in the entire area where auxiliary charging is performed.

  The auxiliary charging operation described in the present embodiment can be used in combination with the techniques of the first to third embodiments. Further, the on period of the switching element Q2 may be used instead of the on period of the switching element Q1.

(Embodiment 5)
In the above-described embodiment, in order to cope with the battery voltage or the charging current, the driving frequency of the switching elements Q1 and Q2 and the time ratio of the on period are controlled. In the present embodiment, in order to cope with a large change in the charging voltage, a configuration in which an intermediate tap is provided in the secondary winding N2 of the transformer T is added as shown in FIG. In the figure, only the secondary side of the transformer T is shown. The configuration of the primary side of the charging device is the same as that of the charging device shown in FIGS.

  In the configuration shown in FIG. 11A, the secondary winding N2 of the transformer T includes two intermediate taps in addition to the center tap. The two intermediate taps are provided so that the number of turns between the center taps is equal. The anode of the diode D1 is connected to one end of the secondary winding N2, and the anode of the diode D2 is connected to the other end of the secondary winding N2. The cathodes of both diodes D1, D2 are connected in common. The anode of the diode D3 is connected to one intermediate tap of the secondary winding N2, and the anode of the diode D4 is connected to the other intermediate tap. The cathodes of both diodes D3 and D4 are connected in common.

  Further, a switch SW is inserted between the cathodes of the diodes D1 and D2 and the cathodes of the diodes D3 and D4, and a smoothing capacitor C4 is connected between the cathodes of the diodes D3 and D4 and the center tap of the secondary winding N2. The Similar to the charging device shown in FIGS. 1 and 2, an inductor L <b> 1 is inserted between the smoothing capacitor C <b> 4 and the secondary battery B.

  As shown in FIG. 11B, a switch element Qs such as a power MOSFET is used so that the switch SW is automatically opened and closed according to the specifications of the secondary battery B, and the switch element Qs is opened and closed. It is controlled according to the battery voltage. That is, the battery voltage is detected as the voltage at the connection point between the resistors R2 and R3, and the voltage at the connection point between the resistors R2 and R3 is compared with the reference voltage Vref using the comparator CP. The comparator CP is provided with hysteresis, and the input voltage for shifting the output from the H level to the low level is set lower than the input voltage for shifting the output from the L level to the H level.

  The switch element Qs is connected to the output of the comparator CP via the drive circuit DR1 so that the switch element Qs is turned on when the output of the comparator CP is at H level and turned off when the output is at L level. That is, the switches R2 and R3, the comparator CP, and the drive circuit DR1 constitute a switch control unit SC that controls on / off of the switch element Qs.

  The switch control unit SC turns on the switch element Qs when the voltage at the connection point of the resistors R2 and R3 representing the battery voltage is equal to or higher than the reference voltage Vref, and smoothes it through the diodes D1 and D2 from both ends of the secondary winding N2 of the transformer T. The capacitor C4 is charged. Further, the switch control unit SC turns off the switch element Qs when the voltage at the connection point of the resistors R2 and R3 representing the battery voltage is equal to or lower than the predetermined voltage with respect to the reference voltage Vref, and the switch control unit SC The smoothing capacitor C4 is charged from the tap through the diodes D3 and D4. That is, the voltage applied to the smoothing capacitor C4 is switched according to the battery voltage.

  By the above-described operation, in the region where the charging current is kept constant, as shown in FIG. 12, the ON / OFF of the switch element Qs and the driving frequency (characteristic a) are maintained while keeping the on-period time ratio (characteristic b) constant. It becomes possible to cope with a change in battery voltage. As described above, since the voltage taken out from the secondary winding N2 of the transformer T is switched by turning on and off the switch element Qs, the range in which the change in the battery voltage can be handled only by the change in the drive frequency is widened, and the drive is performed over a wide range of battery voltages. Constant current control is possible only by changing the frequency. As a result, it is possible to make the switching voltage Vs lower than that of the configuration of the first embodiment, and it is possible to continue the composite resonance operation over a wide range of the battery voltage and to flow the charging current. That is, the efficiency can be increased as compared with the configuration of the first embodiment, and the generation of noise is suppressed.

  The voltage for switching on and off the switch element Qs is set by the reference voltage Vref. Since the battery voltage rises with time during charging, the drive frequency changes with the voltage Vx corresponding to the reference voltage Vref as a boundary as shown in FIG. However, since hysteresis is given to the comparator CP, even if the battery voltage fluctuates in the vicinity of the reference voltage Vx, the drive frequency does not fluctuate (chatter). Further, as shown as a characteristic c in FIG. 12, the efficiency does not change greatly in the region where the battery voltage is around the voltage Vx, and the efficiency decreases only in the first control region A1 where the battery voltage is less than the switching voltage Vs. To do.

  When the drive frequency is adjusted according to the battery voltage in order to make the charging current constant, as described in the first embodiment, the drive frequency is decreased to increase the power non-transmission period τ as the battery voltage increases. Set. Therefore, the voltage Vx described above is slightly higher than the battery voltage when the power non-transmission period τ is almost zero in a state where the smoothing capacitor C4 is charged from both ends of the secondary winding N2 of the transformer T through the diodes D1 and D2. Set to a higher voltage. In a region where the battery voltage is lower than the voltage Vx, the switch element Qs is turned off, and the smoothing capacitor C4 is charged through the diodes D3 and D4 from the intermediate tap of the secondary winding N2 of the transformer T. Therefore, it becomes possible to change the drive frequency in a range almost the same as the region where the battery voltage is equal to or higher than the voltage Vx.

  Through the above-described operation, in the first control region A1, as in each of the above-described embodiments, the on period of only one switching element Q1 is shortened and composite resonance does not occur, but the on period of the other switching element Q2 is The length is set so as to cause a complex resonance. Further, in the second control region A2 where the battery voltage is equal to or higher than the switching voltage Vs, the on-period time ratio is kept constant and the drive frequency is relatively lower than the first control region A1, so the battery voltage is Regardless, the operation of complex resonance is performed, and high-efficiency and low-noise operation becomes possible.

  Other configurations and operations are the same as those of the above-described embodiments, and the above-described characteristic configuration of the present embodiment can be used in combination with other embodiments. In the above configuration example, two intermediate taps are provided in the secondary winding N2, but any number of intermediate taps can be provided as long as the number is an even number.

  In addition, since the charging device of each embodiment mentioned above employ | adopts a half-bridge type | mold switching power supply, compared with a push pull type | mold structure, the voltage applied to the primary winding of the trans | transformer T becomes substantially half. That is, the number of turns of the primary winding N1 is smaller than that of the push-pull type switching power supply. If the step-down ratio of the transformer T is constant between the half-bridge type and the push-pull type, the number of turns of the secondary winding N2 decreases as the number of turns of the primary winding N1 decreases. As a result, loss due to the transformer T is reduced, which contributes to miniaturization of the transformer T. As a result, it is possible to provide a highly efficient and compact charging device.

  In the above configuration example, the input power source of the switching power source is a DC power source obtained by rectifying the AC power source AC with the diode bridge DB and smoothing with the smoothing capacitor C2, but the input power source is a DC power source with another configuration. Also good. For example, another DC power source such as a solar cell may be used.

B Secondary battery C2 Smoothing capacitor (DC power supply)
C4 smoothing capacitor (rectifying smoothing circuit)
CN charge control circuit Cr capacitor (resonance circuit)
Cv drain-source capacitance (resonance circuit)
D1, D2 diode (rectifier smoothing circuit)
DB Diode bridge (DC power supply)
L1 inductor (rectifier smoothing circuit)
Lr inductor (resonant circuit)
Q1, Q2 switching element R1, R2, R3 resistance (state detection circuit)
SC switch controller T transformer (resonance circuit)
T1 leakage transformer

Claims (10)

A series circuit of two switching elements connected between both ends of a DC power source as an input power supply, a charge control circuit for controlling on / off of the switching elements, and an inductor connected in parallel to one of the switching elements; A resonance circuit for composite resonance operation comprising a series circuit of a primary winding of a transformer and a capacitor; a rectification smoothing circuit that rectifies and smoothes the output of the secondary winding of the transformer and supplies a charging current to the secondary battery; A state detection circuit that detects a battery voltage that is a terminal voltage of the secondary battery and a charging current that flows to the secondary battery, and the charge control circuit includes the inductor, the primary winding of the transformer, and the capacitor While controlling the on / off of the two switching elements while forming a power non-transmission period in a frequency range higher than the resonance frequency determined by Until the battery voltage detected by the state detection circuit reaches a specified target voltage from the start of charging, the charging current becomes a constant current, and when the battery voltage reaches the target voltage, charging is performed at a constant voltage. controlling on and off of the switching element, further, in a period in which the charging current and the constant current, provided the switching voltage corresponding to the battery voltage, in the first control region the battery voltage is less than the switching voltage, before Symbol 2 by adjusting the duty ratio of the on-period of pieces of the switching elements, varied as the battery voltage is reduced low enough time ratio, in the second control region the battery voltage is the switching voltage or the The driving frequency of the switching element is lowered as the battery voltage is increased while keeping the time ratio constant so that the ON periods of the two switching elements are equal. Charging apparatus according to symptoms. The charge control circuit, the Te first control region odors, keeping one ON period of one switching element of said two of said switching element constant, adjust the ON period of the remaining one of the switching elements charging apparatus according to claim 1, characterized by controlling the time ratio by. The charge control circuit comprises in a first control region, in a region the battery voltage where the state detecting circuit detects is prescribed threshold voltage or more, the first switching element of said two of said switching elements maintaining the on-time constant, and adjusting the on-period of the second switching element, in the region the battery voltage is less than the threshold voltage, the coercive Chi the oN period of said second switching element to be constant, and charging apparatus according to claim 1, wherein adjusting the on-period of the first switching element.   A charging device adapted to a plurality of types of secondary batteries having different battery voltages at full charge, wherein the charging control circuit has a charging current determined for each specification of the secondary battery in the second control region. The charging device according to claim 1, wherein a driving frequency of the switching element is changed so as to flow.   The charge control circuit reduces the charge current by increasing the drive frequency over time in a state where the ON periods of the two switching elements are equal after the battery voltage reaches the target voltage, and at least power 5. The on-period of one of the switching elements is kept constant from the time when the non-transmission period disappears, and the other on-period of the switching element is shortened or lengthened as the charging current decreases. The charging device according to any one of the above.   The charge control circuit reduces the charge current by increasing the drive frequency over time in a state where the ON periods of the two switching elements are equal after the battery voltage reaches the target voltage, and at least power From the time when the non-transmission period disappears, one on period of the switching element is kept constant, and as the charging current decreases, the other on period of the switching element is shortened or lengthened, and the charging current is less than a specified threshold current. 5. The region according to claim 1, wherein the other ON period of the switching element is kept constant and the one ON period of the switching element is lengthened or shortened. The charging device described.   The charging control circuit intermittently turns on and off the switching element after the battery voltage reaches the target voltage to provide a period for flowing a charging current and a period for stopping the charging current. The charging device according to any one of claims 1 to 4, wherein the charging current is reduced by lengthening the period of stopping.   A switch inserted in a path for rectifying and smoothing the output from both ends of the secondary winding of the transformer, and turning off the switch until the battery voltage exceeds a specified voltage, to the secondary winding of the transformer A switch controller for rectifying and smoothing the output from the provided intermediate tap and turning on the switch when the battery voltage exceeds the specified voltage to rectify and smooth the output from both ends of the secondary winding of the transformer; The charging device according to any one of claims 1 to 7, further comprising:   The charging device according to claim 1, wherein the transformer is a leakage transformer, and the inductor uses a leakage inductance of the leakage transformer.   The charging device according to claim 1, wherein the secondary battery is a lithium ion battery.

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