JP5707028B2 - Power converter - Google Patents

Description

  The present invention relates to a power conversion device.

  Conventionally, a technique for efficiently driving a load using a plurality of power supplies has been disclosed. For example, Patent Document 1 discloses a power conversion device that is connected to a plurality of power supplies and generates a drive voltage for driving a motor from output voltages of the plurality of power supplies. Specifically, from each of a plurality of power supplies, one cycle of the motor electrical angle is divided into 2N (N is a natural number) times the number of motor phases according to the electrical angle, and a plurality of rectangles are divided in this divided section. Output wave voltage pulses and distribute power from multiple power sources. Thereby, when outputting electric power from a plurality of power supplies, a stable rectangular wave state can be maintained and fluctuations in the output voltage can be suppressed.

JP 2009-38891 A

  However, according to Patent Document 1, although the above-described effects can be achieved, since a pulse is output by dividing a plurality of electrical angle periods, the number of times of switching per electrical angle period increases, resulting in a switching loss. It can grow.

  The present invention has been made in view of such circumstances, and an object thereof is to reduce switching loss while suppressing fluctuations in output voltage.

In order to solve such a problem, the present invention includes first switch means corresponding to a low-potential-side power supply and second switch means corresponding to a high-potential-side power supply, As control for the switch means, the first switch means is turned on only once, the first switch means is turned on only once , and then the second switch means is turned on only once during the conduction period of the first switch means. It shall be the non-conductive mixture was allowed to. As a result, only one output voltage pulse for each phase corresponding to one cycle of electrical angle is generated.

  According to the present invention, since the change in the output voltage within one electrical angle cycle can be suppressed to a small level, the fluctuation of the output voltage can be suppressed, and the short circuit can be suppressed. Further, by generating one output voltage pulse corresponding to one electrical angle cycle, it is possible to reduce the number of times of switching in one electrical angle cycle, so that it is possible to reduce switching loss.

Explanatory drawing which shows typically the whole structure of the control system containing the power converter device concerning 1st Embodiment. Explanatory drawing which shows the system configuration centering on the power converter 30 typically Illustration of output voltage pulse Explanatory drawing of the on / off state of each switch 31 (31a, 31b), 34, 37 regarding the U phase Explanatory drawing which shows typically the whole structure of the control system containing the power converter device concerning 2nd Embodiment. Explanatory drawing which shows typically the system configuration centering on the power converter 30 concerning 2nd Embodiment. Explanatory diagram showing the waveform of the U-phase output voltage pulse Explanatory drawing of the ON / OFF state of each switch 61 (61a, 61b), 64 (64a, 64b), 57 regarding the U phase Explanatory drawing of the ON / OFF state of each switch 61 (61a, 61b), 64 (64a, 64b), 57 regarding the U phase Explanatory drawing which shows pulse width p2 with respect to pulse width p1 and amplitude V2 of the electrical angle second harmonic component in the case of the prescribed phase voltage amplitude command V * and the respective power supply voltages Vdc_a and Vdc_b Explanatory drawing which shows typically the whole structure of the control system containing the power converter device concerning 4th Embodiment. The block diagram which shows the structure of the pulse command production | generation part 52 of 4th Embodiment. Explanatory drawing which shows typically the whole structure of the control system containing the power converter device concerning 5th Embodiment. Explanatory drawing of switch temperature Tsw_ab, Tsw_cd and distribution correction coefficient rto_th Explanatory drawing which shows typically the whole structure of the control system containing the power converter device concerning 5th Embodiment. Explanatory drawing which shows the waveform of the output voltage pulse and the U-phase output current when the power distribution ratio rto_pa is changed for the same phase voltage amplitude command V *

(First embodiment)
FIG. 1 is an explanatory diagram schematically showing an overall configuration of a control system including a power conversion device according to the first embodiment of the present invention. In the present embodiment, a control system applied to a drive motor for an electric vehicle will be described. This control system is mainly configured by the motor 10, the power converter 30, and the control unit 50, and the power converter 30 and the control unit 50 constitute the power conversion device according to the present embodiment.

  The motor 10 includes, for example, a plurality of phase windings star-connected around a neutral point (in this embodiment, three phase windings including a U-phase winding, a V-phase winding, and a W-phase winding) Is a three-phase AC synchronous motor. The motor 10 is driven by an interaction between a magnetic field generated by supplying three-phase AC power to each phase winding from a power converter 30 described later and a magnetic field generated by a permanent magnet of the rotor. The rotor of the motor 10 is connected to the input shaft of the automatic transmission.

  FIG. 2 is an explanatory diagram schematically showing a system configuration centered on the power converter 30. The power converter 30 is connected to a plurality of power sources connected in series (first and second power sources 20 and 21 in this embodiment). The power converter 30 synthesizes and generates output voltage pulses from the output voltages of the power supplies 20 and 21 corresponding to the conduction states of the switches 31 to 39 described later (power conversion means). The output voltage pulse is generated corresponding to each phase, and the power converter 30 outputs the output voltage pulse of each phase to the three-phase AC synchronous motor 10 that is a load, thereby driving the motor 10.

  Here, the first and second power supplies 20 and 21 are independent DC power supplies, and are connected in series by connecting the positive electrode of the first power supply 20 and the negative electrode of the second power supply 21. ing. As each power supply 20, 21, for example, a battery such as a nickel metal hydride battery or a lithium ion battery can be used. A common bus 41 is connected to the positive electrode of the first power supply 20 and the negative electrode of the second power supply. Further, a positive electrode bus 42 is connected to the positive electrode of the second power source 21, and a negative electrode bus 43 is connected to the negative electrode of the first power source 20. A smoothing capacitor 44 is provided between the common bus 41 and the negative electrode bus 43, and a smoothing capacitor 45 is provided between the common bus 41 and the positive electrode bus 42.

In the power converter 30, bidirectional switches (switch means) 31 to 33 capable of controlling bidirectional conduction are connected between the common bus 41 and the output terminals corresponding to the three phases. The individual bidirectional switches 31 to 33 are connected in parallel with a pair of unidirectional switches 31a / 31b to 33a / 33b, each of which can control conduction in one direction, with their conduction directions being opposite to each other. It is configured. Each of the switches 31a / 31b to 33a / 33b is mainly composed of a semiconductor switch (for example, a switching element such as a transistor such as an IGBT), and each semiconductor switch has a diode (not shown) to have a withstand voltage. ) are series connected.

  In the power converter 30 of this configuration, the potential of the positive bus 42 is not reversed, and the potential of the positive bus 42 is higher than the potential of the common bus 41. For this reason, unidirectional conduction is established between the positive bus 42 connected to the positive electrode of the second power supply 21 and the output terminals corresponding to the three phases, like the upper arm of a general three-phase inverter. Are connected to the unidirectional switches 34 to 36, respectively. Further, between the negative bus 43 connected to the negative electrode of the first power supply 20 and the output terminals corresponding to the three phases, the unidirectional switches 37 to 39 are connected to each other. Each of the unidirectional switches 34 to 39 is mainly composed of a semiconductor switch (for example, a switching element such as a transistor such as an IGBT), and each of the semiconductor switches has a reflux diode connected in reverse parallel.

The on / off state of each of the switches 31 to 39 , that is, switching between conduction and interruption (switching operation) is controlled through a gate drive signal output from the control unit 50. The individual switches 31 to 39 are turned on by being turned on by the control unit 50, and are turned off (cut off) by being turned off.

  Referring again to FIG. 1, the control unit 50 is a control unit that controls the power converter 30, and controls the output torque of the motor 10 that is a load through the power converter 30 (control unit). As the control unit 50, a microcomputer mainly composed of a CPU, a ROM, a RAM, and an I / O interface can be used. The control unit 50 performs a calculation for controlling the power converter 30 according to a control program stored in the ROM. Then, the control unit 50 outputs a control signal (gate drive signal) calculated by this calculation to the power converter 30.

  The control unit 50 has a torque control unit 51, a pulse command generation unit 52, and a gate drive signal generation unit 53, when this is viewed functionally, and a torque command for the motor 10 supplied from the outside. Based on the value Te *, the conduction state of each of the switches 31 to 39 provided in the power converter 30 is controlled by rectangular wave driving. Here, the rectangular wave drive controls the power converter 30 by making the phase of the output voltage pulse output from the power converter 30 variable with respect to one cycle of the electrical angle that is the electrical rotation phase of the motor 10. This is a driving method.

  The torque control unit 51 outputs a phase voltage amplitude command V * and a phase voltage phase command δ * of the motor 10 based on the torque command value Te * and the motor rotation speed ω. Here, the phase voltage amplitude command V * is an amplitude related to a three-phase output voltage command output from the power converter 30 in accordance with the torque command value Te * and the motor rotation speed ω. The phase voltage phase command δ * is a command value for manipulating the phase of the three-phase output voltage command output from the power converter 30.

  When the motor 10 is a synchronous motor, the motor torque can be expressed by the relationship between the amplitude of the voltage applied to the motor and the phase of the voltage. Therefore, when various parameters relating to the motor 10 are known, the phase voltage amplitude command V * and the phase voltage phase command δ * can be obtained from the relational expression based on the torque command value T *. In the present embodiment, a correspondence relationship between the torque command value Te * and the motor rotational speed ω, the phase voltage amplitude command V * and the phase voltage phase command δ * is acquired in advance through experiments and simulations, and a map defining the correspondence relationship. Is held by the torque control unit 51. Then, the torque control unit 51 calculates the phase voltage amplitude command V * and the phase voltage phase command δ * from the torque command value Te * and the motor rotation speed ω with reference to the map. The calculated phase voltage amplitude command V * is output to the pulse command generation unit 52, and the calculated phase voltage phase command δ * is output to the gate drive signal generation unit 53.

  The pulse command generator 52 includes a phase voltage amplitude command V *, a voltage of the first power supply 20 (hereinafter referred to as “first power supply voltage”) Vdc_a, and a voltage of the second power supply 21 (hereinafter referred to as “second power supply”). Vdc_b) and power distribution ratio rto_pa are input. The first and second power supply voltages Vdc_a and Vdc_b are detected by a voltage sensor (not shown). The power distribution ratio rto_pa is supplied from an external system (not shown). Here, the power distribution ratio rto_pa is a parameter indicating the output ratio of the first power supply 20. When power is output only from the first power supply 20, “1” is input as the power distribution ratio rto_pa, and when power is output only from the second power supply 21, “0” is input as the power distribution ratio rto_pa. . Further, when power is evenly distributed between the first power supply 20 and the second power supply 21, “0.5” is input as the power distribution ratio rto_pa.

The pulse command generator 52 calculates the first and second distribution voltage commands Va * and Vb * based on the phase voltage amplitude command V * and the power distribution ratio rto_pa (see Formula 1).

  Here, the first distribution voltage command Va * indicates the amplitude of the output voltage command output from the first power supply 20 according to the power distribution ratio rto_pa, and the phase voltage amplitude command V * and the power distribution ratio rto_pa. Is calculated as an integrated value. The second distribution voltage command Vb * indicates the amplitude of the output voltage command output from the second power supply 21 according to the power distribution ratio rto_pa, and the first distribution voltage command Va * is changed from the phase voltage amplitude command V *. Is calculated as a subtraction value obtained by subtracting. Since the output current does not change abruptly due to the inductance component of the load, the power ratio supplied from each power source 20, 21 can be manipulated by manipulating the amplitude ratio of the output voltage from each power source 20, 21. it can.

  FIG. 3 is an explanatory diagram of the output voltage pulse. Based on the first and second distribution voltage commands Va * and Vb * and the first and second power supply voltages Vdc_a and Vdc_b, the pulse command generation unit 52 supplies the first and second power supplies 20 and 21 to each other. Corresponding pulse widths p1 and p2 are calculated. This pulse width is a parameter for setting the pulse width of the output voltage of each of the power supplies 20 and 21 constituting the output voltage pulse.

  FIG. 4A is an explanatory diagram showing, for example, the waveform of an output voltage pulse output from the U-phase output terminal for one electrical angle cycle. Here, the potential of the negative electrode bus 43 is set as a reference. The output voltage pulse generated by the output voltage of the first power supply 20 has a symmetrical waveform centered on “π / 2” with “0” to “π” as a range. Specifically, in the electrical angle, the output voltage falls from the first power supply voltage Vdc_a to 0 at the timing of “p1”, and then the output voltage changes from 0 to the first power supply voltage Vdc_a at the timing of “π−p1”. Get up to. That is, the pulse width p1 of the first power supply 20 sets the width of the pulse relating to the output voltage of the first power supply 20 constituting the output voltage pulse. By operating this pulse width p1, the first The voltage amplitude from the power source 20 can be controlled.

  The second power supply 21 outputs the positive voltage on the condition that it is the positive voltage output period of the first power supply 20. The output voltage pulse generated by the output voltage of the second power supply 20 has a symmetrical waveform centered on “3π / 2” with “π” to “2π” as a range. Specifically, the output voltage rises from the first power supply voltage Vdc_a to a voltage corresponding to the sum of the first and second power supply voltages Vdc_a and Vdc_b at the timing of “π + p2” in the electrical angle, and thereafter “2π−p2 The output voltage corresponding to the first power supply voltage Vdc_a rises at the timing "." That is, the pulse width p2 of the second power supply 20 corresponds to the pulse width related to the output voltage of the second power supply 20 constituting the output voltage pulse (specifically, it corresponds to the non-output period of the second power supply 20). The pulse amplitude) is set, and the voltage amplitude from the second power source 21 can be controlled by manipulating the pulse width p2 of the second power source 21.

These pulse widths p1 and p2 and the amplitude V1 of the fundamental electrical wave component of the output voltage pulse generated from the power supply voltages Vdc_a and Vdc_b satisfy the relationship shown in the following equation.

In the formula, the first term on the right side is represented by the voltage amplitude (first distributed voltage command Va *) of the first power source 20 and the first pulse width p1, and the second term on the right side is the second power source. This is expressed by a voltage amplitude of 21 (second distributed voltage command Vb *) and a second pulse width p2. Using these relationships and the above-described power distribution ratio rto_pa, the first distribution voltage command Va * and the second distribution voltage command Vb * are expressed by the following equations.

The pulse command generator 52 can calculate the pulse widths p1 and p2 of the power supplies 20 and 21 based on the following equation.

  In order to reduce the load on the control unit 50, for the calculation of the inverse cosine, only the calculation part of the inverse cosine is tabulated, the table is referred to based on the input numerical values, and each pulse width p1, p2 is calculated. May be calculated. The calculated pulse widths p1 and p2 are output to the gate drive signal generator 53.

  FIG. 3B is an explanatory diagram showing a waveform of a U-phase output voltage pulse. The gate drive signal generation unit 53 drives the switches 31 to 39 of the power converter 30 to obtain the output voltage pulse of each phase based on the pulse widths p1 and p2 and the phase voltage phase command δ *. A gate drive signal is generated.

  Based on the phase voltage phase command δ *, the gate drive signal generation unit 53 operates the phase of the output voltage pulse generated with the pulse widths p1 and p2 based on the electrical angle. With this operation, the output voltage pulses generated with the pulse widths p1 and p2 are offset in a direction in which the phase is delayed by the phase voltage phase command δ * based on the basic waveform of FIG. (Refer FIG.3 (b)). Thereby, the voltage phase of the electrical angle fundamental wave component of the output voltage pulse is in a state corresponding to the voltage phase of the electrical angle fundamental wave component of the output voltage command.

The waveform of the V-phase output voltage pulse is a waveform whose phase is delayed by 2π / 3 with respect to the waveform of the U-phase output voltage pulse based on the output voltage pulse generated with each pulse width p1, p2. It becomes. The waveform of the W-phase output voltage pulse is a waveform whose phase is delayed by 2π / 3 with respect to the waveform of the V-phase output voltage pulse based on the output voltage pulse generated with each pulse width p1, p2. It becomes.

  As described above, the gate drive signal generation unit 53 determines the rising phase (ON phase) and the falling phase (OFF) for the output voltage pulse of each phase from the pulse widths p1 and p2 and the phase voltage phase command δ *. Phase). The gate drive signal generation unit 53 generates gate drive signals for the switches 31 to 39 from the on / off phase.

FIG. 4 is an explanatory diagram of on / off states of the switches 31 (31a, 31b), 34, and 37 related to the U phase. Hereinafter, the description will be given focusing on the U phase, but the same applies to the other phases. In FIG. 4, when the gate drive signals S31a, S31b, S34, and S37b related to the switches 31a, 31b, 34, and 37 are at a high level, the switches 31a, 31b, 34, and 37 are turned on. The gate drive signal generator 53 compares the electrical angle θ of the motor 10 with the phases θ31a, θ37, θ31b, and θ34 shown in the following equations. Here, the phase θ31a is the off phase of the switch 31a, the phase θ37 is the off phase of the switch 37, the phase θ31b is the off phase of the switch 31b, and the phase θ34 is the off phase of the switch 34. .

  When the condition that the electrical angle θ is larger than each of the off phases θ31a, θ37, θ31b, and θ34 is satisfied, the gate drive signal generation unit 53 falls the pulse, that is, the corresponding switches 31a, 31b, 34, and 37. Gate drive signals S31a, S31b, S34, and S37b are generated so as to turn off. For example, as a condition that the electrical angle θ is larger than the pulse phase θ31a, the gate drive signal S31a is generated so as to turn off the switch 31a. Here, the gate drive signal generation unit 53 generates the gate drive signals S31a and S37 as an inverted signal by setting the switch 31a and the switch 37 as a set, and sets the switch 31b and the switch 34 as a set and sets the gate as the inverted signal. Drive signals S31b and S34 are generated. For each set, gate drive signals S31a to S37b are generated by adding drive signal times that are simultaneously turned off. That is, the phase θ31a is also the on phase of the switch 37, the phase θ37 is also the on phase of the switch 31a, the phase θ31b is also the on phase of the switch 34, and the phase θ34 is also the on phase of the switch 31b.

  Here, the relationship between the off phase θ37 of the switch 37 and the off phase θ31b of the switch 31b is such that the off phase θ31b of the switch 31b is larger than the off phase θ37 of the switch 37, so that the output voltage pulse of each phase is It has a convex waveform. When the pulse widths p1 and p2 are zero, that is, when the off phase θ37 of the switch 37 and the off phase θ31 of the switch 31b coincide, the output voltage pulse of each phase is a simple rectangular wave. It becomes. Therefore, in this specification, “during the conduction period of the switch means corresponding to the power source on the low potential side” includes the entire range from the start point to the end point of the period.

  As described above, in the present embodiment, the control unit 50 controls the low-potential-side power supply (mains) as control for each switch (for example, each U-phase switch 31 (31a, 31b), 34, 37) in one electrical angle cycle. In the embodiment, the switch 31a corresponding to the first power supply 20) is turned on, and the high-potential-side power supply (in this embodiment, the second power supply is turned on during the conduction period of the switch 31a corresponding to the low-potential-side power supply 20). The switch 34 corresponding to the power source 21) is turned on. Thereby, the control unit 50 generates one output voltage pulse corresponding to one cycle of the electrical angle through the power converter 30.

  In this case, the control unit 50, based on the output voltage command and the voltages Vdc_a and Vdc_b of the power supplies 20 and 21, respectively, the pulse p1 and p2 width of the output voltage of the power supplies 20 and 21 constituting the output voltage pulse, The pulse phase δ * for manipulating the phase of the output voltage pulse is calculated. Then, the control unit 50 generates one output voltage pulse corresponding to one electrical angle cycle based on the calculation result.

  According to such a configuration, the individual power sources 20 and 21 output one pulse within one electrical angle cycle, and the output voltage pulse is synthesized. The change in the output voltage of each phase in one electrical angle cycle is as follows. Only up and down for 21 series voltages. Thereby, the change of the output voltage within one cycle of the electrical angle is minimized, and the fluctuation of the output voltage can be suppressed. In addition, since one output voltage pulse corresponding to one electrical angle cycle can reduce the number of times the switches 31 to 39 are switched in one electrical angle cycle, it is possible to reduce switching loss. In addition, since the order of conduction of the switches 31 to 39 corresponding to the low-potential side and high-potential side power supplies is regulated, occurrence of a short circuit can be suppressed. As a result, the power of the two power sources 20 and 21 can be distributed to control the power source power. Further, by calculating the pulse widths p1 and p2 from the power supply voltages Vdc_a and Vdc_b and the output voltage command, the output voltage command of the power converter 30 can be changed even when the power supply voltages Vdc_a and Vdc_b change. This can be realized by manipulating the pulse widths p1 and p2 and the pulse phase δ *.

In the present embodiment, the control unit 50 further sets the pulse widths p1 and p2 corresponding to the power sources 20 and 21 based on the power distribution ratio rto_pa that defines the distribution ratio of the power from the power sources 20 and 21, respectively. Calculate. According to such a configuration,
By considering the power distribution ratio rto_pa, the power of each of the power supplies 20 and 21 can be distributed. Thus, power can be distributed with a small voltage change and a small loss by manipulating the pulse width of the voltage to be synthesized without providing a converter for adjusting a plurality of power supply powers.

  Specifically, the control unit 50 outputs the output voltage command for the power supply corresponding to the power distribution ratio rto_pa for each power supply based on the phase voltage amplitude command V * which is the amplitude of the output voltage command and the power distribution ratio rto_pa. Is calculated as distributed voltage commands Va * and Vb *, and the power sources 20 and 21 are assigned to the power sources 20 and 21 based on the voltage Vdc_a and Vdc_b of each power source and the distributed voltage commands Va * and Vb * of the power sources 20 and 21, respectively. Corresponding pulse widths p1 and p2 are respectively calculated. According to this configuration, the distribution of the power supply is realized by the distribution of the phase voltage amplitude command V *, and the output voltage pulse corresponding to the output voltage command is output by obtaining the pulse widths p1 and p2 corresponding to the amplitude. be able to. In the present embodiment, the control unit 50 controls the power supplies 20 and 21 so that the voltage phase of the electrical angle fundamental wave component of the output voltage pulse corresponds to the voltage phase of the electrical angle fundamental wave component of the output voltage command. The phase of the output voltage pulse generated according to the pulse widths p1 and p2 corresponding to is controlled.

According to this configuration, after calculating the pulse widths p1 and p2, the phase of the output voltage pulse corresponding to the pulse widths p1 and p2 is manipulated. As a result, the amplitude of the output voltage command can be realized by the pulse widths p1 and p2, and the phase of the output voltage pulse can be output as the phase of the output voltage command. As a result, the voltage can be manipulated at an arbitrary phase, and voltage phase control becomes possible.

  In the present embodiment, the plurality of power supplies 20 and 21 are connected in series. Since the plurality of power supplies 20 and 21 are connected in series, the magnitude relationship between the input potentials is uniquely determined, so that it is easy to select a power supply (potential) such as a low voltage or a high voltage. Thereby, simplification of a calculation required for control can be achieved. As a result, the cost of the apparatus can be reduced and the operational reliability can be improved. Note that the number of power supplies may be three or more.

  It is preferable that the voltage change of the output voltage pulse within one cycle of the electrical angle is not more than twice the voltage value from the minimum potential to the maximum potential. Thereby, the voltage change within one cycle can be minimized, and the loss caused by the voltage change can be reduced. Therefore, the efficiency of the power converter 30 and its load can be improved.

  In this specification, the term “detection” is used not only to directly detect a corresponding parameter using a sensor or the like, but also to estimate the parameter from factors that affect the corresponding parameter. In addition, the term includes indirect detection methods.

(Second Embodiment)
Hereinafter, a control system according to the second embodiment of the present invention will be described. The control system according to the second embodiment is different from that of the first embodiment in the configuration of the power converter 30. The description of the configuration common to the first embodiment will be omitted, and the following description will focus on the differences.

  FIG. 5 is an explanatory diagram schematically showing the overall configuration of a control system including the power conversion device according to the second embodiment, and FIG. 6 focuses on the power converter 30 according to the second embodiment. It is explanatory drawing which shows a system structure typically. The power converter 30 is connected to a plurality of power supplies (first and second power supplies 20 and 21 in this embodiment) connected in parallel to each other. The power converter 30 synthesizes and generates output voltage pulses from the output voltages of the power supplies 20 and 21 corresponding to the conduction states of switches 61 to 69 described later. The output voltage pulse is generated corresponding to each phase, and the power converter 30 outputs the output voltage pulse of each phase to the three-phase AC synchronous motor 10 that is a load, thereby driving the motor 10.

Here, the first and second power sources 20 and 21 are independent DC power sources. As each power supply 20, 21, for example, a battery such as a nickel metal hydride battery or a lithium ion battery can be used. A negative electrode bus (common bus) 43 is connected to each of the negative electrodes of the first and second power supplies 20 and 21. The positive electrode of the first power source 20 is connected to the first positive electrode bus 42a, and the positive electrode of the second power source 21 is connected to the second positive electrode bus 42b. A smoothing capacitor 44 is provided between the negative electrode bus 43 and the first positive electrode bus 42a, and a smoothing capacitor 45 is provided between the negative electrode bus 43 and the second positive electrode bus 42b.
In the power converter 30 of this embodiment, bidirectional switches (switch means) 61 to 63 capable of controlling bidirectional conduction between the first positive electrode bus 42a and the output terminals corresponding to the three phases. Are connected to each other. Similarly, bidirectional switches (switch means) 64 to 66 are connected between the second positive electrode bus 42b and the output terminals corresponding to the three phases. The individual bidirectional switches 61 to 66 are connected in parallel with a pair of unidirectional switches 61a / 61b to 66a / 66b, each of which can control conduction in one direction, with their conduction directions being opposite to each other. It is configured. Further, a unidirectional switch (switch means) 67 capable of controlling unidirectional conduction is provided between the negative electrode bus 43 and the output terminals corresponding to the three phases, similarly to the lower arm of a general three-phase inverter. To 69 are connected to each other.

  Due to such a configuration of the power converter 30, the control unit 50 drives the power converter 30 in the following form. In the present embodiment, the first power supply voltage Vdc_a and the second power supply voltage Vdc_b are input to the gate drive signal generation unit 53, respectively.

  First, the pulse command generator 52 of the control unit 50 compares the power supply voltages Vdc_a and Vdc_b to set the high voltage VH and the low voltage VL. Specifically, when the first power supply voltage Vdc_a is larger than the second power supply voltage Vdc_b, the first power supply voltage Vdc_a is set as the high voltage VH, and the second power supply voltage Vdc_b is set as the low voltage VL. Is set. In contrast, when the first power supply voltage Vdc_a is equal to or lower than the second power supply voltage Vdc_b, the second power supply voltage Vdc_b is set as the high voltage VH, and the first power supply voltage Vdc_a is set as the low voltage VL. Is done.

Further, the pulse command generation unit 52 compares the power supply voltages Vdc_a and Vdc_b, and based on the comparison result and the power distribution ratio rto_pa, the distribution voltage command VH * related to the high voltage side power supplies 20 and 21; The distributed voltage command VL * related to the low-voltage power sources 20 and 21 is calculated. Specifically, when the first power supply voltage Vdc_a is larger than the second power supply voltage Vdc_b, the distribution voltage commands VH * and VL * are calculated as shown in Equation 6, and the first power supply voltage Vdc_a is calculated. Is less than or equal to the second power supply voltage Vdc_b, the distribution voltage commands VH * and VL * are calculated as shown in Equation 7.

Then, the pulse command generator 52 calculates the pulse widths p1 and p2 related to the high voltage side and low voltage side power supplies 20 and 21 based on the following equations.

FIG. 7 is an explanatory diagram showing a waveform of a U-phase output voltage pulse. Based on the pulse widths p1 and p2 calculated in the pulse command generation unit 52 and the phase voltage phase command δ * calculated in the torque control unit 51, the gate drive signal generation unit 53 outputs the output voltage pulse of each phase. Therefore, a gate drive signal for driving each of the switches 61 to 69 of the power converter 30 is generated.

  Based on the phase voltage phase command δ *, the gate drive signal generation unit 53 operates the phase of the output voltage pulse generated with the pulse widths p1 and p2 based on the electrical angle. By this operation, the output voltage pulses generated with the respective pulse widths p1 and p2 are offset in a direction in which the phase is delayed by the phase voltage phase command δ * based on the basic waveform of FIG. (Refer FIG.7 (b)).

    The waveform of the V-phase output voltage pulse is a waveform whose phase is sent by 2π / 3 with respect to the waveform of the U-phase output voltage pulse based on the output voltage pulse generated with each pulse width p1, p2. It becomes. The waveform of the W-phase output voltage pulse is a waveform whose phase is sent by 2π / 3 with respect to the waveform of the V-phase output voltage pulse based on the output voltage pulse generated with each pulse width p1, p2. It becomes.

Hereinafter, the description will be given focusing on the U phase, but the same applies to the other phases. First, the gate drive signal generation unit 53 calculates basic on / off phases θa, θb, θc, and θd based on the pulse widths p1 and p2 and the phase voltage phase command δ *. The basic on / off phases θa to θd satisfy the following relationship.

  The gate drive signal generation unit 53 compares the power supply voltages Vdc_a and Vdc_b and assigns the basic on / off phases θa to θd as the off phases of the switches based on the comparison result.

  FIG. 8 is an explanatory diagram of the on / off states of the switches 61 (61a, 61b), 64 (64a, 64b), 67 related to the U phase. When the first power supply voltage Vdc_a is larger than the second power supply voltage Vdc_b (Vdc_a> Vdc_a), that is, when the high voltage VH is the first power supply voltage Vdc_a, each basic on / off phase is set as follows. The

θa: OFF phase of switch 64a (ON phase of switch 67)
θb: OFF phase of switch 67 (ON phase of switch 64a)
θc: OFF phase of switch 64b (ON phase of switch 61a)
θd: OFF phase of switch 61a (ON phase of switch 64b)
When the condition that the electrical angle θ is larger than each of the off phases θa to θd is satisfied, the gate drive signal generation unit 53 causes the pulse to fall, that is, turns off the corresponding switches 64a, 67, 64b, 61a. The gate drive signals S64a, S67, S64b and S61a are generated. In addition, the gate drive signal generation unit 53 sets the switch 64a and the switch 67, the switch 61a and the switch 64b as a set, and generates an inverted signal as the gate drive signals S54a, S57, S54b and S51a in each set. Note that the switch 61b generates a gate drive signal that is always on in one electrical angle cycle.

  FIG. 9 is an explanatory diagram of the on / off states of the switches 61 (61a, 61b), 64 (64a, 64b), 67 related to the U phase. When the first power supply voltage Vdc_a is equal to or lower than the second power supply voltage Vdc_b (Vdc_a ≦ Vdc_a), that is, when the high voltage VH is the second power supply voltage Vdc_b, each basic on / off phase is set as follows: Is done.

θa: OFF phase of switch 61a (ON phase of switch 67)
θb: OFF phase of switch 67 (ON phase of switch 61a)
θc: OFF phase of switch 61b (ON phase of switch 64a)
θd: OFF phase of switch 64a (ON phase of switch 61b)
When the condition that the electrical angle θ is larger than each of the off phases θa to θd is satisfied, the gate drive signal generation unit 53 causes the pulse to fall, that is, turns off the corresponding switches 61a, 67, 61b, and 64a. The gate drive signals S61a, S67, S61b, and S64a are generated. In addition, the gate drive signal generation unit 53 generates the inverted signals as the gate drive signals S61a, S67, S61b, and S64a for each set of the switch 61a and the switch 67, the switch 61b, and the switch 64a. The switch 64b generates a gate drive signal that is always on in one electrical angle cycle.

  As described above, according to the present embodiment, the power supplies 20 and 21 are configured in parallel, but the change in the output voltage in one cycle of the electrical angle is only the voltage difference between the power supplies 20 and 21. Further, the power of the two power sources 20 and 21 can be distributed and the power source power can be controlled while reducing the number of times the switches 61 to 59 are switched.

  In the second embodiment, a step of determining the magnitude of the voltage and changing the off-phase assignment of the switches 61 to 69 is necessary, so that more complicated processing than that of the first embodiment is required. However, the power supplies 20 and 21 can be connected in parallel.

(Third embodiment)
Hereinafter, a control system according to a third embodiment of the present invention will be described. The control system according to the third embodiment is different from that of the first embodiment in the arithmetic processing by the pulse command generation unit 52 of the control unit 50. The pulse command generation unit 52 according to the present embodiment does not generate the pulse widths p1 and p2 using the power distribution ratio rto_pa, but from the power supply voltages Vdc_a and Vdc_b and the phase voltage amplitude command V *. Pulse widths p1 and p2 are generated. The description of the configuration common to the first embodiment will be omitted, and the following description will focus on the differences.

  As described in the first embodiment, the amplitude of the fundamental electric wave component of the output voltage pulse generated from the pulse widths p1 and p2 and the power supply voltages Vdc_a and Vdc_b is expressed by Equation 2. In Equation 2, the first term on the right side is represented by the voltage amplitude (first distribution voltage command Va *) of the first power source 20 and the first pulse width p1, and the second term on the right side is the second power source. This is expressed by a voltage amplitude of 21 (second distributed voltage command Vb *) and a second pulse width p2. In the present embodiment, the pulse widths p1 and p2 are calculated using these relationships and the distribution voltage commands Va * and Vb *.

  First, the phase voltage amplitude command V * input to the pulse command generation unit 52 is given as the amplitude V1 of the fundamental wave component of the output voltage pulse.

On the other hand, the amplitude V2 of the electrical angle second harmonic component included in the output voltage pulse is expressed by the following equation.

Here, when a value within the pulse width p1 of the first power supply 20 is selected, the second power supply 21 is used in order to realize the phase voltage amplitude command V * with the amplitude V1 of the fundamental wave component of the output voltage pulse. The pulse width p2 of the above has the following relationship.

  When the amplitude V2 of the electrical angle second harmonic component corresponding to the pulse widths p1 and p2 calculated in this way is evaluated, there is a set of pulse widths p1 and p2 having the minimum value. Therefore, a set of pulse widths p1 and p2 corresponding to the phase voltage amplitude command V * is selected in advance so as to minimize the amplitude V2 of the electrical angle second-order harmonic component, and is mapped. The pulse command generation unit 52 inputs the phase voltage amplitude command V *, refers to the map, and specifies each pulse width p1, p2 corresponding to the phase voltage amplitude command V *.

  When the power supply voltages Vdc_a and Vdc_b fluctuate, these values are also used as inputs, and a set of the pulse widths p1 and p2 corresponding to the phase voltage amplitude command V * and the power supply voltages Vdc_a and Vdc_b It is only necessary to select in advance so as to minimize the amplitude V2 of the angular second-order harmonic component and map this. In this case, by normalizing the phase voltage amplitude command V * and the first power supply voltage Vdc_a with the second power supply voltage Vdc_b (V * / Vdc_b, Vdc_a / Vdc_b), the two-dimensional map is not used. The pulse widths p1 and p2 may be specified from the dimension map. The identified pulse widths p1 and p2 are output to the gate drive signal generator 53.

  FIG. 10 is an explanatory diagram showing the pulse width p2 with respect to the pulse width p1 and the amplitude V2 of the electrical second-order harmonic component in the case of the prescribed phase voltage amplitude command V * and the power supply voltages Vdc_a and Vdc_b. As shown in the figure, when the pulse width p1 of the first power supply 20 is “p1x”, the second harmonic included in the output voltage pulse is set when the pulse width p2 of the second power supply 21 is set to “p2x”. The component amplitude V2 can be suppressed to zero.

  As described above, in the present embodiment, the control unit 50 includes the second harmonic included in the output voltage pulse among the combinations of the pulse widths p1 and p2 that realize the phase voltage amplitude command V * with the amplitude of the fundamental wave component of the output voltage pulse. A pair whose wave component amplitude is closest to zero is generated as a pulse width corresponding to each of the power supplies 20 and 21. According to such a configuration, distortion of the output voltage can be reduced and the harmonic current can be reduced. Thereby, not only the noise from the load to the outside world can be reduced, but also the loss due to the harmonic current can be reduced. If the load is a motor, the generation of torque ripple can be suppressed, and the control performance can be improved.

(Fourth embodiment)
Hereinafter, a control system according to a fourth embodiment of the present invention will be described. The control system according to the fourth embodiment is different from that of the third embodiment in that the current output from the power converter 30 to the motor 10 is detected, thereby detecting the harmonic current. is there. The description of the configuration common to the third embodiment will be omitted, and the following description will be focused on the differences.

  FIG. 11 is an explanatory diagram schematically illustrating an overall configuration of a control system including the power conversion device according to the fourth embodiment. A U-phase current Iu and a W-phase current Iw detected by the current sensor are input to the pulse command generation unit 52 of the control unit 50 according to the present embodiment. As shown in FIG. 12, the pulse command generation unit 52 of the present embodiment includes a second harmonic detection unit 52a, a PI control unit 52b, and a p2 calculation unit 52c.

  The second harmonic detection unit 52a detects the second harmonic current based on the U-phase and W-phase currents Iu and Iw, and outputs the amplitude of the second harmonic current (harmonic detection means). ). As a method of detecting a harmonic current of a specific order, for example, a method of performing coordinate conversion on rotating coordinates synchronized with the harmonic and inputting the current value after coordinate conversion to a low-pass filter can be mentioned. The details of such a harmonic current detection method are disclosed in Japanese Patent No. 3809978, so refer to them if necessary. In addition to this method, the second harmonic current may be detected using a band pass filter.

  The PI control unit 52b sets the pulse width p1 corresponding to the first power supply 20 so that the amplitude of the detected second harmonic current follows the command value 0 by PI control. The PI control unit 52b outputs the set pulse width p1 to the p2 calculation unit 52c. The p2 calculation unit 52c calculates the pulse width p2 using Formula 11 shown in the third embodiment based on the pulse width p1 from the PI control unit 52b. The calculated pulse widths p1 and p2 are output to the gate drive signal generator 53, respectively.

  Thus, in the present embodiment, the control unit 50 calculates the pulse width corresponding to each of the power supplies 20 and 21 based on the amplitude of the fundamental wave component of the output voltage command and the amplitude of the harmonic current. According to this configuration, it is possible to manipulate the harmonic current value while outputting the amplitude of the fundamental wave component of the output voltage command. Thereby, the efficiency of the load of the power converter 30 can be improved, and if the load is the motor 10, the torque ripple can be reduced by controlling the harmonic current.

  Further, according to the present embodiment, unlike the third embodiment, it is not necessary to have data obtained by calculating the pulse widths p1 and p2, and the memory of the control unit 50 can be reduced in size.

  Note that the harmonic current of the output current can be estimated by calculating the fundamental voltage component and the harmonic voltage component included in the output voltage pulse width.

(Fifth embodiment)
Hereinafter, a control system according to a fifth embodiment of the present invention will be described. The control system according to the fifth embodiment is different from that of the first embodiment in that the temperature of the switches 31 to 39 is detected and thereby the respective pulse widths p1 and p2 are calculated. The description of the configuration common to the first embodiment will be omitted, and the following description will focus on the differences.

  FIG. 13 is an explanatory diagram schematically illustrating an overall configuration of a control system including the power conversion device according to the fifth embodiment. In the pulse command generation unit 52 of the control unit 50 according to the present embodiment, information regarding the temperature of each of the switches 31 to 39 detected by the temperature sensor that is the temperature detection unit is input as temperature information Tsw. Hereinafter, the calculation processing of the pulse command generation unit 52 will be described by focusing on the U phase, but the same applies to the other phases.

  FIG. 14 is an explanatory diagram of the switch temperatures Tsw_ab and Tsw_cd and the distribution correction coefficient rto_th. First, the pulse command generation unit 52, based on the temperature information Tsw, the temperature of the switch 31a that controls conduction from the positive electrode of the first power supply 20 to the output terminal, and from the output terminal to the negative electrode of the first power supply 20. The higher one of the temperatures of the switch 37 that controls the conduction of the power is set as the first power supply temperature Tsw_ab. Further, the pulse command generation unit 52 determines the temperature of the switch 31b that controls conduction from the output terminal to the first power supply 20 side based on the temperature information Tsw, and the positive electrode of the second power supply 21 to the output terminal. The higher one of the temperatures of the switch 34 that controls conduction is set as the second power supply temperature Tsw_cd.

  As shown in FIG. 14, when one of the power supply temperatures Tsw_ab and Tsw_cd exceeds a preset determination temperature Tsw_th, the pulse command generator 52 distributes a distribution correction coefficient according to the degree of excess. The distribution correction coefficient rto_th is set so that rto_th approaches 1 to 0. The obtained distribution correction coefficient rto_th is multiplied by the first or second distribution voltage command Va *, Vb * distributed using the power distribution ratio rto_pa.

  In the present embodiment, it is assumed that the first power supply temperature Tsw_ab exceeds the determination temperature (temperature set based on the operation limit) Tsw_th. In this case, the pulse command generation unit 52 multiplies the first distribution voltage command Va * by the distribution correction coefficient rto_th, and updates the first distribution voltage command Va * with this multiplication value.

On the other hand, the pulse command generator 52 calculates a difference ΔVa * as shown in the following equation based on the first distribution voltage command Va * and the distribution correction coefficient rto_th.

  The pulse command generator 52 adds the difference ΔVa * to the second distribution voltage command Vb *, and updates the second distribution voltage command Vb * with this added value.

  As described above, the pulse command generation unit 52 calculates the pulse widths p1 and p2 using the calculated first and second distribution voltage commands Va * and Vb *.

  Although only the case where the power supply a is multiplied by the distribution correction coefficient is shown, the first and second distribution voltage commands Va * and Vb * are calculated (updated) from the same viewpoint also when the power supply b is corrected. .

Thus, in this embodiment, the control unit 50 corresponds to each power source 20, 21 based on the temperature of each of the switches 31-39, the voltages Vdc_a, Vdc_b of the power sources 20, 21, and the output voltage command. The pulse widths p1 and p2 to be calculated are calculated. According to such a configuration,
When there is a possibility that the temperature of any of the switches 31 to 39 rises and exceeds the operation limit, the on-time of the switches 31 to 39, that is, the pulse width is shortened by performing distribution correction. To do. Further, in order to realize the output voltage command, the pulse width from the other power sources 20 and 21, that is, the ON time of the switches 31 to 39 is increased. Thereby, since the temperature rise of the specific switches 31 to 39 can be suppressed in advance, the stable operation of the power converter 30 can be provided.

(Sixth embodiment)
Hereinafter, a control system according to a sixth embodiment of the present invention will be described. The control system according to the sixth embodiment is different from that of the first embodiment in that each pulse width p1, p2 is calculated from the viewpoint of reducing the switching loss. The description of the configuration common to the first embodiment will be omitted, and the following description will focus on the differences.

  FIG. 15 is an explanatory diagram schematically illustrating an overall configuration of a control system including the power conversion device according to the fifth embodiment. Hereinafter, the calculation processing of the pulse command generation unit 52 will be described by focusing on the U phase, but the same applies to the other phases.

  FIG. 16 is an explanatory diagram showing waveforms of the output voltage pulse and the U-phase output current when the power distribution ratio rto_pa is changed with respect to the same phase voltage amplitude command V *. In the drawing, when the first power supply voltage Vdc_a is “1”, the second power supply voltage Vdc_b has a relationship of “5”.

Since the switching loss Wsw is proportional to the integrated value of the voltage and current, the switching loss can be obtained from the voltage change during switching and the U-phase output current at that time. The switching loss Wsw_a shown in FIG. 16A is expressed by the following equation.

  Here, A is a constant for obtaining the switching loss. When I0 is “0” and I1 is “1”, the switching loss Wsw_a is “5 × A”.

  FIG. 6B shows an output waveform in which the fundamental component of the output voltage pulse is the same as that in FIG. 6A, but the switching loss Wsw_b is different from the karate shown in FIG. Here, I0 is “2”, I1 is “0.5”, and the switching loss Wsw_b is “4.5 × A”. Thus, it can be seen that the magnitudes of the switching losses are different even when the output voltage pulse has the same fundamental wave component.

  Therefore, in the present embodiment, the switching loss of each switch 31 to 39 is estimated by performing the above-described calculation, and the loss of each switch 31 to 39 is estimated. Then, the relationship between each pulse width p1, p2 and the output current and output voltage is calculated in advance so as to minimize the estimated value of loss, and this is stored in the pulse command generation unit 52 as a database. In addition to the torque command value Te * and the rotational speed ω (output frequency), the torque control unit 51 outputs an estimated output current value I in addition to the phase voltage amplitude command V * and the phase voltage phase command δ *. To 52. Thereby, the pulse command generation unit 52 calculates the pulse widths p1 and p2 based on the estimated value I of the output current and the phase voltage amplitude command V * after referring to the database.

  Thus, in this embodiment, the control unit 50 corresponds to each power source 20, 21 based on the loss of each of the switches 31-39, the voltages Vdc_a, Vdc_b of the power sources 20, 21, and the output voltage command. The pulse widths p1 and p2 to be generated are respectively generated. According to such a configuration, concentration of loss in specific switches 31 to 39, that is, increase in heat generation and temperature rise can be suppressed in advance. In addition, the output voltage for driving the motor 10 can be realized while minimizing the loss of the power converter 30, and the efficiency of the power converter 30 can be improved.

  Moreover, the loss of each switch 31-39 is performed based on the output current I and the voltage change of an output voltage pulse. According to this configuration, it is possible to estimate the transient loss due to the switching of the switches 31 to 39. Then, by operating the pulse widths p1 and p2 from among the pulse widths p1 and p2 that realize the output voltage command, it is possible to drive with the pulse widths p1 and p2 that reduce the switching loss. Thereby, highly efficient power conversion can be realized.

  In the present embodiment, the pulse command generation unit 52 of the control unit 50 functions as loss estimation means for estimating the losses of the switches 31 to 39, and the torque control unit 51 of the control unit 50 It functions as current detection means for detecting the output current.

  As mentioned above, although embodiment of this invention was described, this invention is not limited to each embodiment mentioned above, A suitable change is possible within the scope of the present invention. For example, the determination methods of the pulse widths p1 and p2 shown in the above-described embodiments may be prepared, and the determination methods may be switched.

DESCRIPTION OF SYMBOLS 10 ... Motor 20 ... 1st power supply 21 ... 2nd power supply 30 ... Power converter 31-33 ... Bidirectional switch 34-39 ... Unidirectional switch 50 ... Control unit 51 ... Torque control part 52 ... Pulse command generation part 53 ... Gate drive signal generator

Claims (10)

Connected in series two power or each anode was connected to the two power source commonly connected to generate a driving voltage of the motor which is the load by the output voltage pulses generated synthesized from the two power A power converter,
Switch means for connecting between the positive electrode of the power source and the motor is provided corresponding to each power source, and the negative electrode having the lowest potential of the two power sources or the commonly connected negative electrode and the motor Power conversion means for generating an output voltage pulse from each output voltage of the power source, according to the conduction state of each of the negative electrode side switch means and the switch means, comprising a negative electrode side switch means to be connected,
Control means for controlling the conduction state of each of the negative electrode side switch means and the switch means,
The switch means includes first switch means corresponding to a low-potential side power supply, and second switch means corresponding to a high-potential side power supply,
As the control for each switch means in one cycle of the electrical angle of the motor, the control means conducts the first switch means only once , and after conducting the first switch means once, the first switch means Only one output voltage pulse for each phase corresponding to one cycle of the electrical angle is generated by conducting the second switch means once during the conduction period of the switch means and then turning it off. A power conversion device.   The control means, based on the output voltage command and the voltage of each power supply, a pulse width for manipulating the pulse width of the output voltage of each power supply constituting the output voltage pulse, and the phase of the output voltage pulse, The power conversion device according to claim 1, wherein the output voltage pulse is generated based on the calculation result. 3. The power conversion device according to claim 2, wherein the control unit further calculates a pulse width corresponding to each power source based on a power distribution ratio that defines a distribution ratio of power from each power source. 4. . The control means, for each power supply, based on the amplitude of the output voltage command and the power distribution ratio, calculates the output voltage command amplitude for the power supply according to the power distribution ratio as a distribution voltage command, The power converter according to claim 3 , wherein a pulse width corresponding to each power source is calculated based on a voltage of each power source and a distributed voltage command of each power source. The control means includes a phase of the output voltage pulse generated according to a pulse width corresponding to each power supply, a voltage phase of an electrical angle fundamental wave component of the output voltage pulse, and an electrical angle fundamental wave of the output voltage command. 5. The power conversion device according to claim 3 , wherein the power conversion device is operated based on the calculated pulse phase so that the voltage phase of the component corresponds. 6. The control means includes
Based on the relationship between the voltage of the first power supply, the voltage of the second power supply, and the pulse width corresponding to each power supply for realizing the amplitude of the output voltage command with the fundamental wave component of the output voltage pulse, Storage means for storing a result of calculating in advance a set of pulse widths of each power source having the smallest second harmonic component contained in the output voltage pulse corresponding to the command;
Of the set of pulse widths corresponding to each power source that realizes the amplitude of the output voltage command with the amplitude of the fundamental wave component of the output voltage pulse, the amplitude of the second harmonic component contained in the output voltage pulse is the smallest. 3. The power conversion device according to claim 2 , wherein a set of near pulse widths is generated as a pulse width corresponding to each power source by reading from the storage means. Further comprising harmonic detection means for detecting a harmonic current of the output current from the power conversion means,
The control means calculates a pulse width corresponding to each power source so that the harmonic current is minimized based on the amplitude of the fundamental component of the output voltage command and the amplitude of the harmonic current. The power conversion device according to claim 2 characterized by things. Temperature detecting means for detecting the temperature of each of the switch means;
The control means is configured such that the detected temperature of the first switch means and the second switch means exceeds the determination temperature based on the temperature of each of the switch means, the voltage of each power source, and the output voltage command. The power distribution ratio according to claim 2, wherein the power distribution ratio is changed so that the power distribution ratio through the switch means having the larger degree becomes smaller, and the pulse width corresponding to each power source is calculated. Conversion device. Loss estimation means for estimating the respective losses of the switch means;
Based on the respective losses of the switch means, the voltages of the respective power supplies, and the output voltage command, the control means is configured so that the total loss of the first switch means and the second switch means is minimized. The power conversion apparatus according to claim 2, wherein a pulse width corresponding to each power source is generated by changing a power distribution ratio from the power supply. Current detection means for detecting an output current from the power conversion means;
The power converter according to claim 9 , wherein the loss estimation unit estimates each loss of the switch unit based on a product of the output current and a voltage change of the output voltage pulse. .

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