JP5658342B2 - Power converter - Google Patents

Description

  The present invention relates to a power conversion device that drives an AC motor. In particular, the power is composed of a power converter (forward converter) that converts AC power into DC power, and a power converter (reverse converter) that drives the AC motor by converting DC power into AC power. The present invention relates to a conversion device.

  For example, in an electric railway using a single-phase AC overhead line as a power source, the power conversion device includes a power converter (forward converter, converter) that converts single-phase AC power into DC power, and an electric motor that converts DC power into AC power. It is comprised from the power converter (inverse converter, inverter) which drives. As long as the single-phase AC power is rectified, the DC power supplied by the forward converter in principle pulsates at twice the power supply frequency. By increasing the capacity of the smoothing capacitor placed between the forward converter and the inverse converter, the pulsation of the capacitor voltage can be reduced. However, the capacity of the smoothing capacitor can be reduced in terms of installation volume, mass, and cost. Since voltage is limited, voltage pulsation cannot be completely eliminated.

変調率実効値Ｙｍ、インバータ角周波数ω１とし、インバータＵ相変調率ｙｍｕを次のようにおく。

実際のインバータＵ相出力電圧ｖｍｕは、Ｕ相変調率ｙｍｕとコンデンサ電圧Ｅｃｆ’の積であるから、式１と式２の積を求めると、

となる。式３より、インバータ角周波数ω１成分（基本波成分）のほかに、インバータ角周波数ω１と、電圧脈動角周波数ω０の和ω１＋ω０と差ω１−ω０の成分が生じていることが分かる。モータは印加する周波数が高くなるにつれてインピーダンスが大きくなる性質を持つ。逆に低周波成分についてはインピーダンスが小さくなる。このためインバータ角周波数ω１が電圧脈動角周波数ω０に接近すると両者の差ω１−ω０が小さくなり、振幅ΔＥｃｆ・Ｙｍが微小であっても無視できない電流が流れてしまう。このようなインバータ角周波数が電圧脈動角周波数に接近したときに発生する低周波脈動をビートノイズと呼ぶ。 This shows the adverse effects when the capacitor voltage is pulsating. The capacitor voltage Ecf ′ is assumed to pulsate around the DC component Ecf with an effective value amplitude ΔEcf and an angular frequency ω0, and is determined as follows.

The modulation factor effective value Ym and the inverter angular frequency ω1 are set, and the inverter U-phase modulation factor ymu is set as follows.

Since the actual inverter U-phase output voltage vmu is the product of the U-phase modulation factor ymu and the capacitor voltage Ecf ′, when the product of Equation 1 and Equation 2 is obtained,

It becomes. From equation 3, it can be seen that in addition to the inverter angular frequency ω1 component (fundamental wave component), there are components of the inverter angular frequency ω1 and the sum ω1 + ω0 and the difference ω1−ω0 of the voltage pulsation angular frequency ω0. The motor has the property that the impedance increases as the applied frequency increases. On the other hand, the impedance is low for low frequency components. For this reason, when the inverter angular frequency ω1 approaches the voltage pulsation angular frequency ω0, the difference ω1−ω0 between the two becomes small, and even if the amplitude ΔEcf · Ym is very small, a non-negligible current flows. Such low frequency pulsation that occurs when the inverter angular frequency approaches the voltage pulsation angular frequency is called beat noise.

となる。式５より、出力電圧ｖｍｕ’に含まれる周波数成分は基本波成分（インバータ角周波数ω１成分）のみとなり、ビートノイズの発生を完全に抑制できることが分かる。 Such a beat noise phenomenon has been known for a long time, and there are some countermeasure techniques. For example, as in the invention described in Patent Document 1, there is known a technique capable of eliminating beat noise by detecting a capacitor voltage and correcting a modulation rate (pulse width) so as to cancel the pulsation of the capacitor voltage. In the invention described in Patent Document 1, the corrected modulation factor ymu ′ is obtained by multiplying the modulation factor effective value Ym by the DC component Ecf of the capacitor voltage and dividing by the capacitor voltage Ecf ′ as shown in Equation 4.

It becomes. From Equation 5, it can be seen that the frequency component included in the output voltage vmu ′ is only the fundamental wave component (inverter angular frequency ω1 component), and the occurrence of beat noise can be completely suppressed.

式６を時間積分して位相の操作量に変換し、インバータＵ相変調率ｙｍｕ’を書き直すと次のようになる。

ここでコンデンサ電圧の脈動振幅ΔＥｃｆは直流成分Ｅｃｆに対して十分小さい（ΔＥｃｆ≪Ｅｃｆ）とすると、次式のように近似できる。

式１と式４の積より、インバータＵ相出力電圧ｖｍｕ’を求めると、

となる。なお微小量ΔＥｃｆの二乗項は無視できるものとして近似した。式９より、インバータ角周波数ω１と電圧脈動角周波数ω０の差の成分ω１−ω０は消え、ビートノイズの発生を抑制できることが分かる。ただし、式９と式３を比較すると和の成分ω１＋ω０の振幅が２倍に増加していることが分かる。しかし、モータは印加する周波数が高くなるにつれてインピーダンスが大きくなるため、和の成分のノイズ電流は無視できる。 However, in the case of using up to a modulation rate of 100% as in a power converter for driving an electric motor used in an electric railway, the modulation rate cannot be operated any more, so the technique described in Patent Document 1 cannot be applied. For such applications, for example, as in the invention described in Patent Document 2, a technique is known in which beat noise can be eliminated by detecting a pulsating component of a capacitor voltage and operating an inverter angular frequency. In the invention described in Patent Document 2, the pulsation component of the capacitor voltage is detected from Equation 1 and multiplied by ω0 / Ecf is set as the operation amount Δω1 of the inverter angular frequency.

Equation 6 is time-integrated and converted into a phase manipulated variable, and the inverter U-phase modulation factor ymu ′ is rewritten as follows.

Here, assuming that the pulsation amplitude ΔEcf of the capacitor voltage is sufficiently small (ΔEcf << Ecf) with respect to the DC component Ecf, it can be approximated by the following equation.

From the product of Equation 1 and Equation 4, the inverter U-phase output voltage vmu ′ is obtained.

It becomes. The square term of the minute amount ΔEcf was approximated as negligible. From Equation 9, it can be seen that the component ω1-ω0 of the difference between the inverter angular frequency ω1 and the voltage pulsation angular frequency ω0 disappears and the occurrence of beat noise can be suppressed. However, comparing Equation 9 with Equation 3, it can be seen that the amplitude of the sum component ω1 + ω0 has doubled. However, since the impedance of the motor increases as the applied frequency increases, the noise current of the sum component can be ignored.

  As described above, the inventions described in Patent Document 1 and Patent Document 2 use the detected capacitor voltage to operate the inverter modulation rate (pulse width) or inverter angular frequency, thereby generating low-frequency pulsation generated in the motor current. This is a technology for suppressing (beat noise). Even if the low frequency pulsation of the motor current is suppressed, the original pulsation of the capacitor voltage cannot be eliminated. These two technologies are feed-forward control. If an error or delay occurs in the detection of the capacitor voltage or the inverter control output, the low-frequency pulsation of the motor current may not be suppressed and the pulsation may remain or the operation amount may be reduced. On the contrary, it may become unstable.

式１１に示すように、モータ三相交流電流に対し、インバータ出力電圧を位相基準として座標変換を行うと、モータ電流の実効値振幅を得ることができる。

ここでモータ三相交流電圧にインバータ角周波数ω１と電圧脈動角周波数ω０の差の成分ω１−ω０の脈動が重畳されている場合を考える。基本波成分（インバータ角周波数ω１成分）の実効値振幅をＶｍとし、差の成分の実効値振幅をΔＶｍとおく。

インバータ角周波数ω１と電圧脈動角周波数ω０の差の成分ω１−ω０の電圧脈動に対して、電流は遅れ位相で流れるので、δ≧０とすると

となる。式１３に対し、インバータ出力電圧を位相基準として座標変換を行うと次のようになる。

式１４より、座標変換を行うと脈動成分の周波数がもともとの電圧脈動角周波数ω０にシフトされることが分かる。このモータ電流の脈動振幅ΔＩｍは、もともとのコンデンサ電圧の脈動と異なり、ビートノイズ抑制制御を適切に行うことにより減少することが期待できる。このように特許文献３記載の発明は、座標変換後のモータ電流より脈動成分を抽出し、これに応じてインバータ角周波数を操作することで、ビートノイズ抑制をフィードバック制御で構成する技術である。フィードバック制御のため、フィードバック制御ゲインの設定にはある程度の余裕があり、所望の応答時間で収束していくように設計できるという利点を持つ。加えて、モータ三相交流の段階では脈動周波数がインバータ角周波数ω１に応じて移動していたのに対して、座標変換後の段階では脈動周波数がω０一定となるため、脈動成分を抽出するためのフィルタが設計しやすい等の利点も持つ。 On the other hand, the invention described in Patent Document 3 detects the motor current at which the low-frequency pulsation actually occurs, and operates the inverter angular frequency according to the extracted pulsation component, thereby configuring beat noise suppression by feedback control. Technology. The effective phase amplitude Im, the inverter angular frequency ω1, the current phase when the inverter output voltage is used as a phase reference is θ, and the motor three-phase AC current is set as follows.

As shown in Equation 11, when coordinate conversion is performed on the motor three-phase AC current using the inverter output voltage as a phase reference, the effective amplitude of the motor current can be obtained.

Here, consider a case where the pulsation of the difference component ω1-ω0 of the difference between the inverter angular frequency ω1 and the voltage pulsation angular frequency ω0 is superimposed on the motor three-phase AC voltage. The effective value amplitude of the fundamental wave component (inverter angular frequency ω1 component) is Vm, and the effective value amplitude of the difference component is ΔVm.

Since the current flows in a delayed phase with respect to the voltage pulsation of the difference component ω1-ω0 of the difference between the inverter angular frequency ω1 and the voltage pulsation angular frequency ω0, if δ ≧ 0

It becomes. When coordinate conversion is performed with respect to Equation 13 using the inverter output voltage as a phase reference, the following is obtained.

From Equation 14, it can be seen that when coordinate conversion is performed, the frequency of the pulsation component is shifted to the original voltage pulsation angular frequency ω0. Unlike the original capacitor voltage pulsation, the motor current pulsation amplitude ΔIm can be expected to decrease by appropriately performing beat noise suppression control. As described above, the invention described in Patent Document 3 is a technique in which beat noise suppression is configured by feedback control by extracting a pulsation component from the motor current after coordinate conversion and manipulating the inverter angular frequency accordingly. For feedback control, there is a certain margin in the setting of the feedback control gain, and there is an advantage that it can be designed to converge with a desired response time. In addition, the pulsation frequency moved according to the inverter angular frequency ω1 at the motor three-phase AC stage, whereas the pulsation frequency is constant at ω0 at the stage after coordinate conversion, so that the pulsation component is extracted. The filter has the advantage of being easy to design.

Japanese Examined Patent Publication No. 61-48356 Japanese Patent Laid-Open No. 2-119573 JP 2003-111500 A

  As a technique for the beat noise phenomenon, the invention described in Patent Document 1 cannot be used in a field where the modulation factor is up to 100%, such as a power converter for driving a motor used in an electric railway. The inventions described in Patent Documents 1 and 2 are feedforward control. If an error or delay occurs in the detection of the capacitor voltage or the inverter control output, the low frequency pulsation of the motor current may not be suppressed and the pulsation may remain. Or, there is a possibility that the operation amount is excessive and the operation becomes unstable.

  The invention described in Patent Document 3 has two problems. First, when the inverter angular frequency ω1 is close to the voltage pulsation angular frequency ω0 in Expression 12, the difference component ω1−ω0 is almost DC, so the current delay phase δ in Expression 13 is almost zero. As ω1 moves away from the voltage pulsation angular frequency ω0, the phase δ increases. That is, in Equation 14, the phase δ of the pulsating component after coordinate conversion varies depending on the inverter angular frequency ω1. From Equation 1 and Equation 6, the operation amount of the inverter angular frequency is ideally in phase with the pulsating component of the capacitor voltage. For this reason, when detecting a pulsation component from the motor current after the coordinate conversion and obtaining the operation amount of the inverter angular frequency using the pulsation component, it is necessary to correct the phase δ. Since the phase δ varies depending on the inverter frequency ω1, it is difficult to design the control system. When the phase δ varies greatly, it is difficult to ensure the stability of the feedback control in the first place.

式１５に対し、インバータ出力電圧を位相基準として座標変換を行うと次のようになる。

式１６より、座標変換を行うと、インバータ角周波数ω１と電圧脈動角周波数ω０の和の成分ω１＋ω０の脈動成分も、もともとの電圧脈動角周波数ω０にシフトされることが分かる。すなわち、座標変換後の段階では、差の成分ω１−ω０と和の成分ω１＋ω０の区別がつかなくなってしまうことを意味する。式８より、インバータ角周波数ω１のみを操作する限り、差の成分ω１−ω０と和の成分ω１＋ω０を同時にゼロにすることはできず、差の成分ω１−ω０をゼロにしようとすると和の成分ω１＋ω０を２倍に増幅してしまう。この結果、座標変換後のモータ電流より抽出した角周波数ω０の脈動成分を極小にすることは可能であるが、決してゼロにすることはできない。このため、角周波数ω０の脈動成分をゼロにするようなフィードバック制御を構成しようとすると、条件によっては発散してしまう場合がある。 Further, as shown in Expression 15, it is assumed that a current flows with a delay phase ε with respect to the voltage pulsation of the sum component ω1 + ω0 of the inverter angular frequency ω1 and the voltage pulsation angular frequency ω0.

When coordinate conversion is performed with respect to Equation 15 using the inverter output voltage as a phase reference, the result is as follows.

From Equation 16, it can be seen that when coordinate conversion is performed, the pulsation component of the sum ω1 + ω0 of the inverter angular frequency ω1 and the voltage pulsation angular frequency ω0 is also shifted to the original voltage pulsation angular frequency ω0. That is, at the stage after the coordinate conversion, it means that the difference component ω1−ω0 and the sum component ω1 + ω0 cannot be distinguished. From Equation 8, as long as only the inverter angular frequency ω1 is operated, the difference component ω1−ω0 and the sum component ω1 + ω0 cannot be made zero at the same time. ω1 + ω0 is amplified twice. As a result, it is possible to minimize the pulsation component of the angular frequency ω0 extracted from the motor current after coordinate conversion, but it cannot be zero. For this reason, if it is going to comprise feedback control which makes the pulsation component of angular frequency (omega) 0 zero, it may diverge depending on conditions.

  In order to solve the above-described problems, the present invention provides a DC power supply, a smoothing capacitor that stabilizes DC power supplied from the DC power supply, and a power converter that converts DC power supplied from the DC power supply into AC power. AC motor driven by AC power supplied by the power converter, AC current detection means for detecting current flowing into the AC motor, and AC voltage frequency or phase applied to the AC motor as a reference, Coordinate conversion means for converting an AC current detection value detected by the AC current detection means into a DC amount, and an AC voltage output by the power converter so that an output value of the coordinate conversion means matches a command value In the power conversion device constituted by the first current control means, means for detecting a low frequency component of the AC current detection value detected by the AC current detection means, Low-frequency component of the AC current detection value is the most important feature in that it comprises a second current control means for operating the output AC voltage of the power converter to be zero.

  The power conversion device according to the present invention includes an AC current detection unit that detects current flowing into the AC motor, and an AC current detection value detected by the AC current detection unit based on an AC voltage frequency or phase applied to the AC motor. In addition to the coordinate conversion means for converting to a direct current amount, and the first current control means for operating the AC voltage output from the power converter so that the output value of the coordinate conversion means matches the command value, the AC current detection Means for detecting a low frequency component of the detected AC current value detected by the means, and second current control means for operating the AC voltage output from the power converter so that the low frequency component of the detected AC current value becomes zero. , The low frequency pulsation of the alternating current can be suppressed at the same time as making the fundamental wave amplitude (alternating current amplitude) of the alternating current coincide with the command value. That is, suppression of the beat noise phenomenon can be realized by feedback control.

  Moreover, regardless of the modulation rate of the power converter, if the pulsation frequency of the alternating current is sufficiently lower than the alternating voltage frequency applied to the alternating current motor, the pulsation can be suppressed by the second current control means.

It is a block diagram of the 1st example of the present invention. It is a block diagram of the 2nd example of the present invention. It is an example of the direct current | flow component detection means in the 1st and 2nd Example of this invention. It is an example of the phase calculating means in the 2nd Example of this invention.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings.

A first embodiment of the present invention is shown in FIG. In FIG. 1, a power converter of the present invention includes a single-phase AC voltage source 1 and a power converter 2 (forward converter, converter) that converts single-phase AC power supplied from the single-phase AC voltage source 1 into DC power. A smoothing capacitor 3 that stabilizes the DC power supplied from the power converter 2, a power converter 4 (inverse converter, inverter) that converts the DC power supplied from the power converter 1 into AC power, Three-phase AC motor 5 driven by AC power supplied from converter 4 and three-phase AC current detecting means for detecting three-phase AC currents Iu, v, and w flowing from power converter 4 to three-phase AC motor 5 6 and DC voltage detection means 7 for detecting the applied voltage Ecf of the smoothing capacitor 3.
Further, based on the reference phases θu, v, w of the three-phase AC voltages Vu, v, w applied to the three-phase AC motor 5 by the three-phase AC currents Iu, v, w detected by the three-phase AC current detection means 6. The coordinate conversion means 10 that performs coordinate conversion and converts it into two-phase DC currents Id and q, and the deviation between the two-phase DC current command values Id and q * and the two-phase DC currents Id and q output from the coordinate conversion means 10 The subtractor 11 to be obtained, the current control means 12 for outputting the DC voltage manipulated variable ΔVd, q so that the current deviation output from the subtractor 11 becomes zero, and the two-phase DC current command values Id, q * are input, The DC voltage generating means 13 for obtaining the two-phase DC voltages Vd and q applied to the AC motor 5 based on the AC voltage frequency applied to the AC motor 5 and the motor constant of the AC motor 5, and the two output from the DC voltage generating means 13 Phase DC voltage Vd, q and current control The adder 14 for obtaining the sum of the two-phase DC voltage manipulated variables ΔVd and q output from the stage 12 is input, and the output of the adder 14 is input, and the voltage amplitude Vr and the voltage phase angle δ are obtained by converting from orthogonal coordinates to polar coordinates. The coordinate conversion means 15 to be output and the addition for obtaining the sum of the reference phase θu, v, w of the three-phase AC voltage Vu, v, w applied to the three-phase AC motor 5 and the voltage phase angle δ output from the coordinate conversion means 15 The sine wave function generator 17 that outputs a three-phase alternating current sine wave based on the phase output from the adder 16 and the adder 16, the voltage amplitude Vr that is output from the coordinate conversion means 15, and the output from the sine wave function generator 17. The product of three-phase alternating current sine waves is obtained, and the three-phase alternating currents Iu, v, and w detected by the multiplier 18 that outputs the three-phase alternating current voltages Vu, v, and w and the three-phase alternating current detecting means 6 are Coordinate conversion means 1 for converting into phase alternating current Ia, b And DC component detection means 20 for detecting low frequency components ΔIa and b, and orthogonal two-phase AC current output by DC component detection means 20. Current controller 21 that outputs a quadrature two-phase AC voltage manipulated variable ΔVa, b such that the low frequency component ΔIa, b of the current is zero, and a quadrature two-phase AC voltage manipulated variable ΔVa, b output by the current controller 21 The coordinate conversion means 22 that inputs and outputs the three-phase AC voltage manipulated variables ΔVu, v, and w, the three-phase AC voltages Vu, v, and w that the multiplier 18 outputs, and the three-phase AC that the coordinate conversion means 22 outputs. The adder 23 for calculating the sum of the voltage manipulated variables ΔVu, v, w, the three-phase AC voltage output from the adder 23, and the smoothed capacitor voltage Ecf output from the DC voltage detection means 7 are input, and the power converter 4 Pulse signals Su, v, w for driving It comprises a PWM generator 24 that outputs.

  A second embodiment of the present invention is shown in FIG. FIG. 2 shows an embodiment of the present invention when the modulation rate is 100%, that is, the AC voltage amplitude output from the power converter is not operable and only the AC voltage phase is operable. In FIG. 2, the power converter of the present invention includes a single-phase AC voltage source 1 and a power converter 2 (forward converter, converter) that converts single-phase AC power supplied from the single-phase AC voltage source 1 into DC power. A smoothing capacitor 3 that stabilizes the DC power supplied from the power converter 2, a power converter 4 (inverse converter, inverter) that converts the DC power supplied from the power converter 1 into AC power, Three-phase AC motor 5 driven by AC power supplied from converter 4 and three-phase AC current detecting means for detecting three-phase AC currents Iu, v, and w flowing from power converter 4 to three-phase AC motor 5 6 and DC voltage detection means 7 for detecting the applied voltage Ecf of the smoothing capacitor 3.

  Further, based on the reference phases θu, v, w of the three-phase AC voltages Vu, v, w applied to the three-phase AC motor 5 by the three-phase AC currents Iu, v, w detected by the three-phase AC current detection means 6. The coordinate conversion means 10 that performs coordinate conversion and converts it into two-phase DC currents Id and q, and the deviation between the two-phase DC current command values Id and q * and the two-phase DC currents Id and q output from the coordinate conversion means 10 The subtractor 11 to be obtained, the current control means 12 for outputting the DC voltage manipulated variable ΔVd, q so that the current deviation output from the subtractor 11 becomes zero, and the two-phase DC current command values Id, q * are input, The DC voltage generating means 13 for obtaining the two-phase DC voltages Vd and q applied to the AC motor 5 based on the AC voltage frequency applied to the AC motor 5 and the motor constant of the AC motor 5, and the two output from the DC voltage generating means 13 Phase DC voltage Vd, q and current control The adder 14 for obtaining the sum of the two-phase DC voltage manipulated variables ΔVd and q output from the stage 12 is input, and the output of the adder 14 is input, and the voltage amplitude Vr and the voltage phase angle δ are obtained by converting from orthogonal coordinates to polar coordinates. The coordinate conversion means 15 to be output and the addition for obtaining the sum of the reference phase θu, v, w of the three-phase AC voltage Vu, v, w applied to the three-phase AC motor 5 and the voltage phase angle δ output from the coordinate conversion means 15 16, a coordinate conversion means 19 that converts the three-phase alternating currents Iu, v, and w detected by the three-phase alternating current detection means 6 into orthogonal two-phase alternating currents Ia and b, and an orthogonal two that is output from the coordinate conversion means 19. The DC component detection means 20 for detecting the low frequency components ΔIa and b by inputting the phase AC currents Ia and b, and the low frequency components ΔIa and b of the orthogonal two-phase AC current output by the DC component detection means 20 become zero. Such quadrature two-phase AC voltage manipulated variable Δ Current controller 21 for outputting a and b, and coordinate conversion means for inputting quadrature two-phase AC voltage manipulated variable ΔVa and b output from current controller 21 and outputting three-phase AC voltage manipulated variable ΔVu, v and w 22, the three-phase AC voltage phase output from the adder 16, the AC voltage angular frequency ω, and the three-phase AC voltage operation amount ΔVu, v, w output from the coordinate conversion means 22 are input, and the three-phase AC voltage phase Sum of phase calculation means 30 for outputting manipulated variables Δθu, v, w, three-phase AC AC phase output from adder 16, and three-phase AC voltage phase manipulated variables Δθu, v, w output from phase calculator 30 An adder 31 for obtaining θ′u, v, w, a three-phase AC phase output from the adder 31, a sine wave function generator 32 for outputting a three-phase AC sine wave, and a sine wave function generator 32 Input three-phase AC sine wave, 1 if input is positive, 0 if negative The code discriminator 33 is configured to output the pulse signal.

  FIG. 3 is an example showing details of the DC component detection means 20 in FIGS. 1 and 2. In FIG. 3, the DC component detection means 20 receives the two-phase DC current command values Id, q *, and the reference phases θu, v, w of the three-phase AC voltages Vu, v, w applied to the three-phase AC motor 5. Coordinate conversion means 40 that outputs the orthogonal two-phase alternating current command values Ia and b *, the orthogonal two-phase alternating current command values Ia and b * output from the coordinate conversion means 40, and the coordinate conversion It comprises a subtractor 41 for obtaining a difference ΔIa, b between the orthogonal two-phase alternating currents Ia, b output from the means 19.

  FIG. 4 is an example showing details of the phase calculation means 30 in FIG. In FIG. 4, the phase calculation means 30 normalizes the three-phase AC voltage operation amounts ΔVu, v, w output from the coordinate conversion means 22 with the smoothing capacitor voltage Ecf output from the DC voltage detection means 7, and operates the modulation rate. A divider 50 for obtaining the quantity, a gain 51 for multiplying the modulation rate manipulated variable output by the divider 50 by 2π to convert it into a phase manipulated variable, and a three-phase AC voltage phase θ′u, v, w are input. A cosine wave function generator 52 for outputting a phase AC cosine wave, and a sign discriminator for inputting a three-phase AC cosine wave output by the cosine wave function generator 52 and outputting 1 if the input is positive and 0 if negative. 53, a multiplier 54 for obtaining the product of the phase manipulated variable output by the gain 51 and the output of the sign discriminator 53, and the AC voltage angular frequency ω are input. If the frequency exceeds a predetermined frequency, 1 is continuously output. Hysteresis that continues to output 0 if the frequency falls below A scan comparator 55, based on the output of the hysteresis comparator 55, and the input of the multiplier 54 from the switch (switch) 56 as output, or blocking.

1 Single-phase AC voltage source 2 Power converter (forward converter, converter)
3 Smoothing capacitor 4 Power converter (inverse converter, inverter)
5 Three-phase AC motor 6 AC current detection means 7 DC voltage detection means 10 Coordinate conversion means (three-phase AC → two-phase DC)
DESCRIPTION OF SYMBOLS 11 Subtractor 12 Current control means 13 DC voltage generation means 14 Adder 15 Coordinate conversion means (orthogonal coordinates → polar coordinates)
16 Adder 17 Sine wave function generator 18 Multiplier 19 Coordinate conversion means (three-phase AC → two-phase AC)
20 DC component detection means 21 Current controller 22 Coordinate conversion means (two-phase AC → three-phase AC)
23 Adder 24 PWM generator 30 Phase calculation means 31 Adder 32 Sine wave function generator 33 Sign discriminator 40 Coordinate conversion means (two-phase DC → two-phase AC)
41 Subtractor 50 Divider 51 Gain 52 Cosine Wave Function Generator 53 Sign Discriminator 54 Multiplier 55 Hysteresis Comparator 56 Switch (Switch)
Ecf Smoothing capacitor voltage Ia, b * Quadrature two-phase alternating current command value Ia, b Quadrature two-phase alternating current ΔIa, b Low frequency component of quadrature two-phase alternating current Id, q * Two-phase alternating current command value Id, q Phase DC current Iu, v, w Three-phase AC current Su, v, w Three-phase pulse signal Vu, v, w Three-phase AC voltage ΔVu, v, w Three-phase AC voltage manipulated variable ΔVa, b Quadrature two-phase AC voltage operation Amount Vd, q Two-phase DC voltage ΔVd, q Two-phase DC voltage manipulated variable Vr Voltage amplitude δ Voltage phase angle θu, v, w Three-phase AC voltage reference phase θ'u, v, w Three-phase AC voltage phase Δθu, v, w Three-phase AC voltage phase manipulated variable ω AC voltage angular frequency

Claims (1)

DC power supply, smoothing capacitor for stabilizing DC power supplied from the DC power supply, first power converter for converting DC power supplied from the DC power supply into AC power, and the first power converter An AC motor driven by the AC power supplied by the power supply, AC current detection means for detecting a current flowing from the first power converter to the AC motor, and an AC voltage frequency or phase applied to the AC motor as a reference First coordinate conversion means for converting the AC current detection value detected by the AC current detection means into a DC amount, and the first coordinate conversion means so that the output value of the first coordinate conversion means matches the DC current command value. In the power converter configured by the first current control means for operating the AC voltage output from the one power converter,
Means for obtaining a deviation between the quadrature two-phase alternating current command value and the quadrature two-phase alternating current detection value, and an operation amount of the quadrature two-phase alternating voltage value according to the deviation, the was converted to the same coordinate system as the AC voltage manipulated variable output by the first current control means, before Symbol said first power converter by adding an AC voltage manipulated variable output by the first current control means A second current control means for operating the AC voltage output from
Said means for determining the deviation of the orthogonal two-phase alternating current command value and the quadrature two-phase alternating current detected values, said referenced to the AC voltage frequency or phase applied to the AC motor, the DC current command using the first current control means Second coordinate conversion means for converting the value into a quadrature two-phase alternating current command value, and third coordinate conversion means for converting the alternating current detection value detected by the alternating current detection means into a quadrature two-phase alternating current detection value When configured by the orthogonal two-phase alternating current command value and the third coordinate transformation means of the orthogonal two-phase alternating current detected value means you calculating the deviation of the output by the output by the second coordinate transformation means The power converter characterized by the above-mentioned.

Images