JP5634819B2 - Ultrasonic diagnostic equipment - Google Patents

Description

  The present invention relates to an ultrasonic diagnostic apparatus, and more particularly to a technique for reducing unnecessary signal components.

  In order to improve the image quality of an ultrasonic image, it is desired to reduce unnecessary signals (unnecessary components) such as side lobes, grating lobes, and noise in received signal processing.

  In the receive beamformer, a plurality of element reception signals from a plurality of vibration elements are subjected to phasing processing (delay processing) and then added for focusing. After the addition, beam data as an RF signal is obtained. The beam data is used for forming an ultrasonic image. In the case of the reflected wave from the focus point, since the phases are uniform among the plurality of element reception signals after the delay processing, an RF signal having a large amplitude after addition is obtained. On the other hand, in the case of a reflected wave from a place other than the focus point, since the phases do not match among the plurality of element reception signals, only an RF signal with low amplitude can be obtained.

  As a conventional technology to reduce unwanted signal components, PCF (Phase Coherence Factor) that focuses on the standard deviation of the phase of the received signal of each channel is used as an evaluation index for phase disturbance, reducing unwanted signal components in the received signal (Non-patent Document 1). However, in such a conventional technique, it may be determined that the phase is disturbed even if the phases are actually aligned. That is, when the image is weighted to the amplitude information using PCF, artifacts due to side lobes, grating lobes, and the like are reduced, but local black spots may occur in the substantial part. This will be specifically described below.

PCF is an evaluation index that focuses on the standard deviation of the phase of the signal of each channel. The following formula (1) shows the calculation formula of PCF.

The variable sf (k) in the above equation (1) is defined as follows.

Here, i is a channel number (i = 1, 2,... N), k is a depth, Si (k) is an analysis signal for an RF reception signal after delay processing (the reception signal is a complex signal) SIi (k) is the real component of the analytic signal, SQi (k) is the imaginary component of the analytic signal, φi (k) is the instantaneous value of the phase determined from the analytic signal, and σ (φi (k)) is the standard deviation of the instantaneous phase value. From the viewpoint of standard deviation, the degree of phase disturbance in the signal arrangement direction is evaluated. γ is a parameter that determines the sensitivity of PCF (where γ ≧ 0). phi ^A (k) is a variable that has been introduced in order to reduce the effects of aliasing that occurs when determining the arctangent tan ^-1, if .phi.i (k) is less than zero π to .phi.i (k) The added value is taken, and when φi (k) is zero or more, a value obtained by subtracting π from φi (k) is taken. In this case, an operation corresponding to the baseline shift is performed on the premise of folding. However, in equation (2), the smaller one of σ (φi (k)) and σ (φi ^A (k)) is selected for each depth. If PCF is negative, PCF (k) = 0.

  When all the channels are perfectly aligned, the standard deviation of the phase is 0, and the PCF is 1. On the other hand, when the phase is disturbed, the standard deviation of the phase increases, so the PCF becomes smaller than 1. By weighting the amplitude information in the subsequent stage with this PCF, it is possible to weaken the influence of a signal having a disordered phase that includes many unnecessary signals.

However, in the process using PCF, the degree of phase disturbance may or may not be properly evaluated. FIG. 5 shows an example of a phase distribution in which the degree of phase disturbance is properly evaluated. FIG. 6 shows an example of a phase distribution in which the degree of phase disturbance is not properly evaluated. As shown in FIG. 5, when the phases of all the channels are distributed with a spread not exceeding π, the degree of phase disturbance can be evaluated without the influence of aliasing by introducing φ ^A (k). More specifically, FIG. 5 shows both φi (k) and φi ^A (k). The former is composed of portions 100a and 100b before the return and a return portion 100d, and the return portion 100d is observed as a return phenomenon of the true portion 100c. Φi ^A (k) is generated as a result of performing the calculation of the above equation (5) on such a phase distribution (see each arrow in the figure for the shift calculation). It is distributed near the baseline, and its standard deviation σ (φi ^A (k)) is a small value (by the way, σ (φi (k)) is a large value, which is not adopted from equation (2)).

On the other hand, as shown in FIG. 6, when the phase is distributed with a spread exceeding π, depending on the distribution position of the phase, there is a case where the influence of phase folding cannot be avoided even if φi ^A (k) is introduced. Specifically, φi (k) is composed of portions 102a, 102b, 102c, and 102d before the return and a return portion 102f. The folded portion 102f is observed as a folding phenomenon of the true portion 102e. When φi ^A (k) is generated by applying the above equation (5) to φi (k), it is a discontinuous waveform consisting of both ends and an intermediate part, and overestimates the degree of phase disturbance. Will end up. Originally, the degree of phase disturbance should be evaluated as a waveform as shown in FIG. 7, but the aliasing correction by φi ^A (k) functions when the phase is distributed in a range exceeding π. It will disappear. For this reason, when the amplitude information is weighted using PCF, there is a portion where the amplitude is extremely weakened in the substantial part, which may appear as a black spot in the local image.

J. Camacho, et al, "Phase Coherence Imaging", IEEE trans. UFFC, vol. 56, No. 5, 2009.

  An object of the present invention is to make it possible to more accurately evaluate the state of phase disturbance for a plurality of element reception signals when calculating coefficients used for reducing unnecessary signal components.

  An ultrasonic diagnostic apparatus according to the present invention includes: a plurality of vibration elements that transmit and receive ultrasonic waves and output a plurality of element reception signals; a delay processing unit that performs a delay process on the plurality of element reception signals; A complex conversion processing unit that generates a plurality of complex signals by applying complex conversion processing to a plurality of element reception signals after delay processing, and performs a complex operation between adjacent signals in a signal sequence composed of the plurality of complex signals A phase difference calculation unit that obtains a plurality of phase differences by executing, a coefficient calculation unit that obtains a coefficient for suppressing unnecessary signal components based on the plurality of phase differences, and a plurality of element reception signals based on the coefficients And a gain variable unit that allows the gain of the received signal to be varied.

  According to the above configuration, a plurality of complex signals are generated from the plurality of element reception signals, and the phase difference is obtained by complex operation between the complex signals. That is, a plurality of phase differences are calculated in the arrangement direction of the plurality of element reception signals. These indicate the degree of phase disturbance in the same direction as a whole. A plurality of phase differences are quantified as an integrated value or an average value of the plurality of phase differences, and then a coefficient for gain adjustment, that is, an unnecessary signal component suppression is calculated. Preferably, the phase difference between the signals is calculated from the arc tangent to the result value of the complex operation between the two complex signals. Since the phase difference is used, the problem of aliasing hardly occurs.

  The complex conversion process is executed after the delay process, but the complex conversion process may be executed before the delay process. An addition process is performed to generate a reception signal (beam data) after the delay process. In this case, a plurality of element reception signals after the delay process may be added, or after the delay process and complex conversion process. A plurality of later complex signals may be added. The variable gain can be realized by other methods by multiplying the coefficient or by subtracting a signal corresponding to the coefficient. The phase difference may be calculated between adjacent elements in the vibration element row, or the phase difference may be calculated between representative vibration elements set discretely.

  Preferably, means for calculating an absolute value for each of the plurality of phase differences is provided, and the coefficient calculation unit calculates the coefficient based on an integrated value or an average value of the plurality of absolute values for the plurality of phase differences. To do. In the case where the phase difference has a positive or negative sign, if the absolute value thereof is taken, the degree of phase disturbance can be accurately quantified as an integrated value or an average value thereof.

  Preferably, the coefficient calculation unit generates the coefficient so that the gain of the received signal is small when the integrated value or the average value is large, and the received signal when the integrated value or the average value is small. The coefficient is generated so as to increase the gain. Both the integrated value and the average value are homogeneous as a result of quantification. According to the average value, even if the number of channels (the number of reference signals) varies, there is an advantage that it is not affected by it.

  Preferably, the coefficient is calculated at each depth along each beam direction, and a one-dimensional smoothing process is performed on a plurality of coefficients arranged in the beam direction, or arranged in the beam direction and the beam arrangement direction. Means for performing two-dimensional smoothing on a plurality of coefficients is provided, and the gain of the received signal is varied using the smoothed coefficients. According to this configuration, it is possible to alleviate steep fluctuations in image quality.

  According to the present invention, the state of phase disturbance can be more accurately evaluated for a plurality of element reception signals. Therefore, it is possible to prevent or reduce image quality deterioration such as blackout by appropriately suppressing unnecessary signal components.

1 is a block diagram showing a preferred embodiment of an ultrasonic diagnostic apparatus according to the present invention. It is a block diagram which shows the structural example of the coefficient calculating part shown in FIG. It is a block diagram which shows other embodiment. It is a block diagram which shows the structural example of the coefficient calculating part shown in FIG. It is a figure which shows the phase distribution in which the aliasing has occurred, and the phase distribution after the aliasing countermeasure, and shows the case where the degree of phase disturbance can be accurately evaluated. It is a figure which shows the phase distribution in which the aliasing has occurred, and the phase distribution after the aliasing countermeasure, and shows the case where the degree of phase disturbance cannot be accurately evaluated. It is a figure which shows the phase distribution as a comparative example.

  DESCRIPTION OF EXEMPLARY EMBODIMENTS Hereinafter, preferred embodiments of the invention will be described with reference to the drawings.

  FIG. 1 shows a preferred embodiment of an ultrasonic diagnostic apparatus according to the present invention, and FIG. 1 is a block diagram showing the overall configuration thereof. This ultrasonic diagnostic apparatus is used in the medical field, and is an apparatus that forms an ultrasonic image based on a reception signal obtained by transmitting and receiving ultrasonic waves to and from a living body. In this embodiment, a B-mode tomographic image is formed as an ultrasound image, but a Doppler image or the like may be formed as a matter of course.

  In FIG. 1, an array transducer 10 is arranged in an ultrasonic probe and is composed of a plurality of vibration elements 12. The plurality of vibration elements 12 are arranged in a straight line. Of course, they may be arranged in an arc shape. An ultrasonic beam (transmission beam, reception beam) is formed using the plurality of vibration elements 12 and is electronically scanned. As electronic scanning methods, electronic sector scanning, electronic linear scanning, and the like are known. It is also possible to use a 2D array transducer instead of the 1D array transducer.

  The transmission unit 14 is a transmission beam former. That is, the transmission unit 14 supplies a plurality of transmission signals having a predetermined delay relationship to the plurality of vibration elements 12 during transmission. As a result, a transmission beam is formed. At the time of reception, the reflected wave from each point in the living body is received by the array transducer 10. As a result, a plurality of reception signals (element reception signals) are generated and output to the amplification unit 16. The amplification unit 16, the delay unit 20, and the signal addition unit 24 constitute a reception unit, and the reception unit is a reception beamformer.

  The amplification unit 16 includes a plurality of amplifiers 18. A delay unit 20 is provided at the subsequent stage, and the delay unit 20 includes a plurality of delay units 22. Delay processing (phasing processing) is executed by these delay devices 22. The delay time given to each delay unit 22, that is, delay data is supplied from the control unit 42. An apodization processing unit may be provided after the delay unit 20. The A / D converter is not shown. A plurality of reception signals (a plurality of element reception signals) after the delay processing are input to the signal addition unit 24. Therefore, the plurality of reception signals are added to form a reception beam electronically. The received signal after the phasing addition processing is output to the detection unit 26. The detection unit 26 is a known circuit that performs detection processing. An input unit 44 is connected to the control unit 42.

  In the present embodiment, an unnecessary signal component suppression unit 28 is provided to suppress unnecessary signal components. Specifically, it includes a coefficient calculation unit 30 and a multiplier 32. The coefficient calculation unit 30 is a circuit that calculates a coefficient (evaluation value) for suppressing unnecessary signal components based on a plurality of element reception signals after delay processing and before addition processing. The multiplier 32 executes the component suppression processing. In the multiplier 32, the received signal after detection is multiplied by a coefficient, and the received signal is suppressed (reduced) according to the magnitude of the unnecessary signal component. The method for obtaining the coefficient will be described in detail later.

  The received signal after unnecessary signal suppression processing is sent to the signal processing unit 34. The signal processing unit 34 executes various signal processing such as logarithmic conversion, and the processed signal is sent to the image forming unit 36. The image forming unit 36 is configured by a digital scan converter (DSC) in this embodiment. Thereby, a B-mode tomographic image is constructed. The image data is sent to the display 40 via the display processing unit 38. In this embodiment, a B-mode image is formed, but a two-dimensional blood flow image or the like may be formed. In the above description, the multiplication of the coefficient is performed in the process of suppressing the unnecessary signal component. However, the unnecessary signal component may be suppressed or reduced by using other methods such as signal subtraction.

  Smoothing processing may be performed on a one-dimensional coefficient sequence made up of a plurality of coefficients arranged in the depth direction, and the smoothed coefficients may be given to the multiplier. Alternatively, the two-dimensional smoothing process may be performed on the two-dimensional coefficient sequence aligned in the depth direction and the beam scanning direction, and the smoothed coefficient may be given to the multiplier. This can alleviate a sharp change in image quality.

In the present embodiment, when evaluating the degree of phase disturbance, a phase difference of received signals is obtained between adjacent channels, and an evaluation index configured by an average value of absolute values of a plurality of phase differences is used. Specifically, an example of an evaluation index is shown in the following formulas (6) and (7). Equations (6) and (7) can be used selectively. In them, the evaluation index is expressed as PTF (k) (PTF: Phase Transit Factor).

The variable Δφi (k) in the above equation (8) is defined as follows.

  Here, i is a channel number (i = 1, 2,... N−1), k indicates the depth, and N indicates the number of channels. Si (k) is the analysis signal for the RF reception signal after delay processing (the reception signal is expressed as a complex signal), SIi (k) is the real component of the analysis signal, and SQi (k) is the analysis The imaginary component of the signal, φi (k) is the instantaneous value of the phase obtained from the analysis signal, σ (φi (k)) is the standard deviation of the instantaneous value, and p is a parameter that determines the sensitivity of the PCF (However, p ≧ 0). If the evaluation value is negative, PTF (k) = 0. In addition, as long as 0 ≦ PTF (k) ≦ 1 is satisfied and a monotone decreasing function with respect to A (k) is satisfied, the mathematical formula for obtaining the evaluation index is not limited to the formulas (6) and (7).

  The above equation (9) is for obtaining the phase difference Δφi (k) between adjacent channels. The above equation (8) obtains A (k) as an average value of the phase differences Δφi (k) across all adjacent channels. Actually, an average value for the absolute value of the phase difference is calculated. The reason why the absolute value is used is to prevent the phase difference from being positive and negative, and canceling out both. Both the above formulas (6) and (7) are formulas that decrease PTF (k) as A (k) increases. In the above calculation, the phase difference between channels is not captured, but the phase difference between channels is captured. It becomes possible to evaluate.

  That is, when the phases are perfectly aligned, A (k) = 0, and the evaluation index PTF (k) is 1. When the phase disturbance increases, A (k) takes a large value, and the evaluation value PTF (k) decreases. By using such an evaluation index, it is possible to evaluate the phase distribution shown in FIG. 6 and the phase distribution shown in FIG. 7 as the same degree of phase disturbance. The present invention is not limited to this example, and when the phase is distributed with a spread exceeding π in the signal string direction, it is possible to reduce the possibility of overestimating the phase disturbance due to the influence of aliasing. By using this evaluation index as a weighting coefficient and thereby weighting the amplitude information, the case where the amplitude is extremely weakened in the substantial part is improved, and local blackout is suppressed.

FIG. 2 shows a configuration example of the coefficient calculation unit 30 shown in FIG. s ₁ (k) to s _N (k) indicate received signals after delay processing. These received signals are input to a plurality of Hilbert transformers 50 and converted into complex signals, respectively. Instead of the Hilbert converter 50, a complex signal converter such as a quadrature detector may be used. A plurality of complex signals are generated as outputs of the plurality of Hilbert transformers 50. Each complex signal consists of a real part and an imaginary part. A phase difference calculator 52 is provided between adjacent channels or adjacent received signals, and each phase difference calculator 52 calculates a phase difference Δφi (k) for two complex signals based on the above equation (9). Each absolute value calculator 54 performs the absolute value calculation in the equation (8), and the average processor 56 executes the average calculation in the equation (8). Thereby, the average value A (k) is obtained. This indicates the average value of the phase difference and indicates the degree of phase disturbance. The coefficient calculator 58 calculates the evaluation index PTF (k) from the average value A (k) based on the equation (6) or the equation (7). This is used as a weighting coefficient. A coefficient is calculated for each depth and is multiplied by the received signal after detection. As a result, unnecessary signal components are suppressed. When the degree of phase disturbance is large, the evaluation index PTF (k) as a coefficient multiplied by the received signal is small. That is, the degree of suppression of the received signal is increased. When the degree of disturbance is small, the evaluation index PTF (k) coefficient is large. That is, the degree of suppression of the received signal is reduced. It is desirable to provide a one-dimensional or two-dimensional smoothing circuit after the coefficient calculator 58.

  3 and 4 show another embodiment. In this embodiment, a plurality of Hilbert transformers 50 are provided between the plurality of delay units 22 and the signal adding unit 24A. That is, in the embodiment shown in FIGS. 1 and 2, the complex signal conversion is performed inside the coefficient calculation unit. However, in the embodiment shown in FIGS. 3 and 4, the complex signal conversion is performed inside the reception unit. Has been implemented. In FIG. 3, the complex signal from each Hilbert transformer 50 is output to the signal adder 24A and the coefficient calculator 30A. A complex signal, which is the addition result from the signal adder 24A, is input to the detector 26A, where detection is performed. As a result, a received signal (scalar signal) after detection is obtained, and a multiplier 32 multiplies the received signal by a gain adjustment coefficient. FIG. 4 shows a specific configuration example of the coefficient calculation unit 30A shown in FIG. A plurality of complex signals are input to a plurality of phase difference calculators 52, and a phase difference is calculated for each adjacent channel. The subsequent configuration is the same as that shown in FIG.

  In the above embodiment, the phase difference is calculated between adjacent elements, but the phase difference may be calculated between a plurality of elements discretely selected from a plurality of elements.

10 array transducer, 20 delay unit, 24 signal addition unit, 28 unnecessary signal component suppression unit, 30 coefficient calculation unit, 32 multiplier.

Claims (4)

A plurality of vibration elements that transmit and receive ultrasonic waves and output a plurality of element reception signals;
Delay processing means for performing delay processing on the plurality of element reception signals;
A complex transformation processing unit that generates a plurality of complex signals by applying complex transformation processing to the plurality of element reception signals after the delay processing;
A phase difference calculation unit for obtaining a plurality of phase differences by executing complex calculation between adjacent signals in a signal sequence composed of the plurality of complex signals;
A coefficient calculation unit for obtaining a coefficient for suppressing unnecessary signal components based on the plurality of phase differences;
A gain variable unit configured to vary a gain of a reception signal based on the plurality of element reception signals based on the coefficient;
Including
The phase difference calculation unit calculates a phase difference as a difference between a phase obtained from the i-th complex signal and a phase obtained from the i + 1-th complex signal for each of the adjacent signals.
An ultrasonic diagnostic apparatus. The apparatus of claim 1.
Means for calculating an absolute value for each of the plurality of phase differences is provided;
The ultrasonic diagnostic apparatus, wherein the coefficient calculation unit calculates the coefficient based on an integrated value or an average value of a plurality of absolute values for the plurality of phase differences. The apparatus of claim 2.
The coefficient calculation unit generates the coefficient so that the gain of the received signal is small when the integrated value or the average value is large, and the gain of the received signal is when the integrated value or the average value is small. An ultrasonic diagnostic apparatus characterized by generating the coefficient so as to increase. The apparatus of claim 3.
The coefficients are calculated at each depth along each beam direction,
A means for performing a one-dimensional smoothing process on a plurality of coefficients arranged in the beam direction, or a means for performing a two-dimensional smoothing process on a plurality of coefficients arranged in the beam direction and the beam arrangement direction is provided.
An ultrasonic diagnostic apparatus, wherein the gain of the received signal is varied using a smoothed coefficient.

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