JP5527187B2 - Semiconductor device - Google Patents

Description

  The present application relates to a semiconductor device, in particular, a semiconductor device that functions as a rectifying element, and a semiconductor device that functions as a switching element in which rectifying elements are connected in parallel.

  A diode in which a semiconductor P-type and an N-type are joined is widely used as a rectifying element. Since the diode generally requires a forward voltage of about 0.7 V, there is a problem that the power loss is large.

  Regarding this problem, a technique is known in which the power consumption of the diode is reduced by detecting the voltage between the terminals of the diode and short-circuiting the terminals with a switch during forward conduction.

JP 7-297699 A

  The diode is required to have a high breakdown voltage characteristic with a reverse bias. However, with the configuration described in Patent Document 1, it is difficult to achieve high breakdown voltage characteristics required in the reverse direction.

  The present invention has been proposed in view of the above-described problems, and an object thereof is to provide a power semiconductor device that constitutes a diode and a switch having a high breakdown voltage and a low resistance.

  A semiconductor device disclosed in the present application is a semiconductor device functioning as a rectifying element, a source having a first transistor that serves as an anode of the rectifying element, a higher breakdown voltage than the first transistor, and a source having the first transistor A depletion type second transistor having a gate connected to the drain of the first transistor, a gate connected to the source of the first transistor, and a drain serving as a cathode of the rectifying element; and a higher breakdown voltage than the first transistor; A depletion-type third transistor having a drain connected to the drain of the second transistor, the source voltage of the first transistor and the source voltage of the third transistor are compared, and the first transistor A comparator for controlling a gate voltage of the transistor.

  According to the disclosed semiconductor device, a high breakdown voltage and low resistance diode and switch can be realized by combining a high breakdown voltage transistor and a low breakdown voltage transistor.

It is a circuit diagram of a 1st embodiment. It is a characteristic view of transistors Q1, Q2, and Q3. It is a circuit diagram of a 2nd embodiment. It is a circuit diagram of a 3rd embodiment. It is a circuit diagram which shows a diode bridge. It is a circuit diagram of a 4th embodiment. It is a circuit diagram which shows the drive circuit of a three-phase motor. It is a circuit diagram which shows the parallel connection of IGBT and a diode. It is a circuit diagram of a 5th embodiment. It is a circuit diagram which shows the power supply circuit for describing 6th Embodiment. It is a figure explaining operation | movement of the power supply circuit of FIG. FIG. 11 is a circuit diagram of a power supply circuit in which a diode provided in the power supply circuit of FIG. 10 is replaced with the semiconductor device described in FIG. 3. It is a figure explaining the reason a reverse current flows. It is a circuit diagram of a 6th embodiment. FIG. 15 is a circuit diagram of a power supply circuit in which a diode provided in the power supply circuit of FIG. 10 is replaced with the semiconductor device described in FIG. 14. FIG. 16 is a diagram for explaining the operation of the power supply circuit of FIG. 15.

  FIG. 1 shows a circuit diagram of the first embodiment. The power semiconductor device of this embodiment includes transistors Q1, Q2, and Q3, a comparator 10, a diode D1, and a capacitor C1. Here, as shown in the characteristic diagram of FIG. 2, the transistor Q1 is a normally-off enhancement type transistor, and the transistors Q2 and Q3 are normally-on depletion type transistors. Further, the transistors Q2 and Q3 are transistors having a higher breakdown voltage (for example, several hundreds to 1000V) than the transistor Q1. For example, in this embodiment, the transistor Q1 is an n-channel silicon MOSFET, and the transistors Q2 and Q3 are high electron mobility transistors (HEMTs) made of a wide band gap semiconductor such as gallium nitride (GaN). The comparator 10 has a non-inverting input terminal and an inverting input terminal. When the voltage applied to the inverting input terminal is higher than the voltage applied to the non-inverting input terminal, the comparator 10 outputs a low logic level signal and is low. In this case, the comparator outputs a high logic level signal.

  The transistor Q2 has a source connected to the drain of the transistor Q1, a gate connected to the source of the transistor Q1, and a drain serving as the cathode K. The transistor Q3 has a drain and a gate connected to the drain and gate of the transistor Q2, respectively, and a source connected to the anode of the diode D1 and the inverting input terminal of the comparator 10. The cathode of the diode D1 is connected to the power supply terminal of the comparator 10 and the capacitor C1. The source of the transistor Q1 is connected to the back gate, the power supply terminal of the comparator 10, the non-inverting input terminal of the comparator 10, and the capacitor C1, and becomes the anode A. The output terminal of the comparator 10 is connected to the gate of the transistor Q1.

  The power semiconductor device of the present embodiment configured as described above prevents current from flowing when a high voltage is applied to the cathode K side, and prevents a current from flowing when the high voltage is applied to the anode A side. Conducts current. This rectifying operation will be described. Here, it is assumed that the anode A is the ground. When a positive high voltage is applied to the cathode K, the source voltages of the depletion type transistors Q2 and Q3 rise, and the transistors Q2 and Q3 operate in a direction in which the drain current ID does not flow. As shown in FIG. 2, the transistors Q2 and Q3 have a negative threshold voltage Vth1, and when the drain current ID = 0, the gate voltage viewed from the sources of the transistors Q2 and Q3 is the absolute value of the threshold voltage Vth1 | Vth1 | is lower. Since the gates of the transistors Q2 and Q3 are connected to the anode A and are at the ground level, as a result, a voltage increased to the absolute value | Vth1 | of the threshold voltage Vth1 is generated at the sources of the transistors Q2 and Q3. The capacitor C1 is charged through the diode D1. Further, since a positive voltage is applied to the inverting input terminal of the comparator 10, the output of the comparator 10 is at the ground level, and the transistor Q1 is turned off.

  When a negative voltage is applied to the cathode K, the source voltages of the depletion type transistors Q2 and Q3 become negative, and the transistors Q2 and Q3 operate in a direction in which the drain current ID flows more. Since a negative voltage is applied to the inverting input terminal of the comparator 10, the output of the comparator 10 becomes positive and the transistor Q1 is turned on. At this time, the charge necessary for turning on the transistor Q1 is supplied from the capacitor C1, and backflow is prevented by the diode D1.

  As described above, the power semiconductor device of this embodiment functions as a diode that allows current to flow only when a negative voltage is applied to the cathode K side. A high electron mobility transistor (HEMT) having a high breakdown voltage characteristic can be used for the transistors Q2 and Q3, so that a high breakdown voltage characteristic with a reverse bias can be realized. In the reverse bias in which a positive high voltage is applied to the cathode K, the voltage generated at the sources of the transistors Q2 and Q3 is a voltage (for example, 2 to 3 V) increased to the absolute value | Vth1 | of the threshold voltage Vth1. . Therefore, the transistor Q1 may be a low breakdown voltage silicon MOSFET, and the on-resistance can be reduced. As a result, a low resistance with a forward bias can be realized, and power loss can be suppressed. In addition, unlike ordinary diodes, since minority carriers are not used, high-speed operation is possible and high-frequency rectification is possible.

  It is also conceivable to use the source of the transistor Q2 as the input of the comparator 10 without providing the transistor Q3. However, in FIG. 1, when the source of the transistor Q2 instead of the transistor Q3 is connected to the comparator 10, if the on-resistance of the transistor Q1 is small, the level of the voltage input to each input terminal of the comparator 10 approaches. The comparison operation of 10 may become unstable. In the present embodiment, by providing the monitoring transistor Q3, the operation of the comparator 10 is ensured and the stability is ensured. In addition, a large current (for example, 100 A) may flow through the transistor Q2, whereas the current flowing through the transistor Q3 is very small compared to the transistor Q2. Therefore, the transistor Q3 is very small as compared with the transistor Q2, and is equivalent to no chip size loss.

  Further, when a positive high voltage is applied to the cathode K, the capacitor C1 is charged, and when a negative voltage is applied to the cathode K, a reverse flow is prevented by the diode D1 and a charge is supplied from the capacitor C1 to the comparator 10. Like to do. Therefore, if the input to the anode A and the cathode K is an alternating input that alternately repeats positive and negative, it is not necessary to separately supply power to the comparator 10, and a self-supporting operation is possible.

  FIG. 3 shows a circuit diagram of the second embodiment. Compared with the first embodiment, the second embodiment has a configuration in which the diode D1 is replaced with a transistor Q4. The transistor Q4 is, for example, a p-channel silicon MOSFET, the drain is connected to the source of the transistor Q3, the source is connected to the power supply terminal of the comparator 10 and the capacitor C1, and the gate is connected to the output terminal of the comparator 10. .

  Assuming that the anode A is ground, when a positive high voltage is applied to the cathode K, the output of the comparator 10 is at the ground level, so that the transistor Q4 is turned on. When a negative voltage is applied to the cathode K, the output of the comparator 10 becomes positive, so that the transistor Q4 is turned off. Therefore, the transistor Q4 in the second embodiment performs the same function as the diode D1 in the first embodiment. In the second embodiment, the forward voltage drop of the diode D1 can be eliminated by using the transistor Q4 instead of the diode D1. Therefore, the voltage generated at the source of the transistor Q3 can be used for the power source of the comparator 10 without waste.

  FIG. 4 shows a circuit diagram of the third embodiment. The third embodiment is different from the first embodiment in that the gate of the transistor Q3 is connected to the source of the transistor Q2. When the gate of the transistor Q3 is connected to the source of the transistor Q2, when the positive voltage is applied to the cathode K to charge the capacitor C1, the absolute value of the threshold voltage Vth1 with respect to the anode A is applied to the source of the transistor Q3. A voltage that rises up to twice | Vth1 | is generated. Therefore, the third embodiment is effective when the voltage required by the comparator 10 cannot be sufficiently ensured, and the drive capability of the transistor Q1 can be improved.

  FIG. 5 is a circuit diagram showing a general diode bridge rectifier circuit in which a diode is incorporated in a bridge circuit and full-wave rectification of an AC voltage is performed. In FIG. 5, when the output of the AC power supply AC is a positive half cycle, the diodes D30 and D20 are turned on, and the current flows through a path through the diode D30, the load R10, and the diode D20. In the half cycle in which the output of the AC power supply AC is negative, the diodes D10 and D40 are turned on, and current flows through a path through the diode D10, the load R10, and the diode D40. Further, the voltage is smoothed by the capacitor C10. In this way, direct current supplied from the alternating current power supply AC to the load R10 is obtained.

  Here, as described above, the power semiconductor devices according to the first to third embodiments function as diodes and do not need to supply power separately, and can operate independently. Therefore, it can be considered that the power semiconductor devices of the first to third embodiments are applied to the diodes D10 to D40 in FIG. In this case, depending on the offset voltage of the comparator 10, the diode D10 and the diode D20 in FIG. 5 or the diode D30 and the diode D40 are turned on at the same time. As a result, a reverse current flows and a large power loss may occur.

  The fourth embodiment shown in FIG. 6 is an embodiment corresponding to the above problem. The negative terminal of the voltage source that supplies the voltage V1 is connected to the source of the transistor Q1. The resistors R1 and R2 are connected in series between the positive terminal of the voltage source that supplies the voltage V1 and the source of the transistor Q3. In the present embodiment, the non-inverting input terminal of the comparator 10 is connected to the source of the transistor Q1, and the inverting input terminal is connected to the voltage dividing points of the resistors R1 and R2. Since the other configuration is the same as that of the second embodiment described with reference to FIG. 3, in FIG. 6, the same reference numerals are given to the respective parts corresponding to those of FIG.

  The operation of this embodiment will be described assuming that the anode A is the ground. When the cathode K has a positive voltage with respect to the anode A, a positive voltage is applied to the inverting input terminal of the comparator 10, and thus the comparator 10 turns off the transistor Q1. On the other hand, when a negative voltage is applied to the cathode K with respect to the anode A, the comparator 10 operates so that the inverting input terminal is at the same ground level as the non-inverting input terminal. Assume that the voltage of the cathode K is Vk, and the current from the voltage source that supplies the voltage V1 to the source of the transistor Q3 through the resistors R1 and R2. Since no current flows into the inverting input terminal of the comparator 10, the current (V1-0) / R1 flowing through the resistor R1 is equal to the current (0-Vk) / R2 flowing through the resistor R2. From (V1-0) / R1 = (0-Vk) / R2, the voltage of the cathode K becomes Vk = −V1 × R2 / R1. Therefore, when a negative voltage is applied to the cathode K with respect to the anode A, the comparator 10 turns on the transistor Q1 so that the cathode K voltage is −V1 × R2 / R1, and the cathode K voltage is When larger than −V1 × R2 / R1, the transistor Q1 is turned off.

  As described above, the power semiconductor device according to the fourth embodiment is not turned on unless the voltage of the cathode K is surely negative with respect to the anode A. As a result, even in the diode bridge as shown in FIG. 5, it is possible to prevent the diode D10 and the diode D20 or the diode D30 and the diode D40 from being turned on simultaneously.

  FIG. 7 is a circuit diagram showing a drive circuit for a general three-phase motor. An IGBT (Insulated Gate Bipolar Transistor) is connected to the motor M in order to apply a high voltage V10 or a ground voltage to the U phase, V phase, and W phase. In this configuration, voltages V11 to V16 are applied to the control terminals of the IGBT1 to IGBT6 by PWM (Pulse Width Modulation) control to control the rotation speed of the motor M.

  In FIG. 7, when the IGBT is in the OFF state, the rotating motor M operates as a generator, and thus it is necessary for the current to flow in a direction opposite to that when the IGBT is in the ON state. However, current cannot flow from the emitter E to the collector C in the off state. Therefore, as shown in FIG. 8, a diode D having the same current capacity as that of the IGBT is required in parallel with each IGBT.

  The fifth embodiment shown in FIG. 9 functions as a parallel connection of an IGBT and a diode as shown in FIG. Each power supply terminal of the comparator 20 is connected in common with each power supply terminal of the comparator 10. The negative terminal of the voltage source that supplies the voltage V2 is connected to the source of the transistor Q1. The inverting input terminal of the comparator 20 is connected to the positive terminal of the voltage source that supplies the voltage V2, and the non-inverting input terminal is a control terminal Cont to which a control signal is input. The output terminal of the comparator 20 is connected to the non-inverting input terminal of the comparator 10. Since the other configuration is the same as that of the fourth embodiment described with reference to FIG. 6, the same reference numerals are given to the components corresponding to FIG.

  The operation of this embodiment will be described. When the control signal input to the control terminal Cont is a low logic level, the output of the comparator 20 is the voltage level of the emitter E. Therefore, the power semiconductor device functions as a diode as in the fourth embodiment of FIG. On the other hand, when the control signal input to the control terminal Cont is at a high logic level, the output of the comparator 20 becomes H level and the H level is applied to the non-inverting input terminal of the comparator 10. Therefore, the comparator 10 turns on the transistor Q1. Therefore, the power semiconductor device conducts the emitter E from the collector C.

  When this power semiconductor device is applied to the drive circuit of the three-phase motor shown in FIG. 7, the following configuration is obtained. That is, three series circuits in which two power semiconductor devices are connected in series are connected in a bridge shape between the power supplies of the high voltage V10. Then, a coil of each phase (U phase, V phase, W phase) of the three-phase motor is connected to each connection point between the two power semiconductor devices in the three series circuits. In addition, the drive circuit outputs a control signal (external control signal) to a control terminal Cont connected to the non-inverting input terminal of the comparator 20 of these power semiconductor devices. This power semiconductor device can be applied not only to a three-phase motor but also to various multi-phase motors. In this case, a series circuit in which two power semiconductor devices are connected in series is connected to a plurality of power supplies corresponding to the polyphase in a bridge shape. And the coil of each phase of a multiphase motor is connected to each connection point between two power semiconductor devices in this series circuit. The multiphase motor drive circuit also outputs a control signal to the comparator 20 of these power semiconductor devices.

  As described above, the power semiconductor device according to the fifth embodiment functions as a diode when the control signal input to the control terminal Cont is at a low logic level, and conducts when the control signal is at a high logic level. A function as a parallel connection of such an IGBT and a diode can be realized. Here, in FIG. 8, the current flows through the IGBT or the diode D depending on the flowing direction. Therefore, both the IGBT and the diode D need to have an element size corresponding to the allowable current. In FIG. 8, the same breakdown voltage is required for the IGBT and the diode D with respect to the maximum applied voltage in the off state of the IGBT. As a result, both the IGBT and the diode D can pass a large current and must have a high breakdown voltage, so that the overall element size becomes large. On the other hand, in the fifth embodiment of FIG. 9, the high voltage is applied only to the transistors Q2 and Q3, which are high electron mobility transistors (HEMT) having a high withstand voltage characteristic. Only a low voltage is applied. Further, the current flows through the transistors Q1 and Q2 regardless of the flowing direction. Therefore, the overall element size can be reduced. Further, the fifth embodiment of FIG. 9 has a lower resistance than that of FIG. 8 and can suppress heat dissipation. For example, when applied to a motor drive of a hybrid vehicle, the heat dissipation system is simplified. This is considered to be possible, and is effective in terms of weight and space.

  FIG. 10 is a circuit diagram showing an example of a power supply circuit 1 for explaining the sixth embodiment. The power supply circuit 1 includes a step-up switching regulator 30, a capacitor (capacitor) C <b> 2 that accumulates part of the current output from the switching regulator 30 as charges, and terminals T <b> 1 and T <b> 2 connected to a load 40.

  The switching regulator 30 includes a coil (inductor) L1 connected to a voltage source that supplies a voltage V20, a transistor Q5 provided between a reference power supply (reference voltage), eg, ground GND, and the coil L1, and the coil L1 and transistor Q5. And a diode D17 having an anode connected to the connection point N1, and a pulse generation circuit 31 that generates a pulse for driving the transistor Q5.

  The transistor Q5 is an NMOS transistor, for example, and in the following description, the transistor Q5 is described as an NMOS transistor. The transistor Q5 has a drain connected to the other end of the coil L1, and a source connected to the ground GND. The pulse generation circuit 31 alternately outputs a high-level conduction pulse that makes the transistor Q5 conductive and a low-level non-conduction pulse that makes the transistor Q5 non-conductive. The coil L1 receives magnetic current from a voltage source that supplies the voltage V20 when the transistor Q5 is conductive, accumulates magnetic energy, and releases the accumulated magnetic energy as current when the transistor Q5 is non-conductive. The diode D17 is a rectifying diode, and has an anode connected to the connection point N1 and a cathode connected to the connection point N2.

  FIG. 11 is a diagram for explaining the operation of the power supply circuit 1 in FIG. 10. In order from the top, the drive pulse Dp output from the pulse generation circuit 31, the on / off state of the transistor Q 5, and the voltage Vc at the connection point N 1 The on / off state of the diode D17, the current Ic flowing through the coil L1, the current Id flowing through the diode D17, and the current Iq flowing through the transistor Q5 are shown.

  The operation of the power supply circuit 1 in FIG. 10 will be described below with reference to FIG. First, the operation of the power supply circuit 1 from time X0 to time X1 in FIG. 11 will be described. When the pulse generation circuit 31 outputs a high level drive pulse Dp to the transistor Q5, the transistor Q5 is turned on (conductive state) by this high level drive pulse Dp. At this time, the voltage Vc at the connection point N1 becomes the ground GND. Further, the output voltage Vout at the terminal T1 becomes higher than the voltage at the connection point N1 (ground GND) due to the charge accumulated in the capacitor C2. As a result, the diode D17 is reverse-biased and the diode D17 is turned off. When the transistor Q5 is turned on, a current flows from the voltage source that supplies the voltage V20 to the coil L1, and the coil L1 accumulates magnetic energy. At this time, the current Ic of the coil L1 increases, the current Id becomes 0A, and the current Iq increases. Note that while the transistor Q5 is in an on state, that is, while the current Id is 0 A, the load 40 is operated by the electric charge already stored in the capacitor C2.

  Next, the operation between time X1 and time X2 in FIG. 11 will be described. When the pulse generation circuit 31 outputs a low-level drive pulse Dp to the transistor Q5, the transistor Q5 is turned off (non-conductive) by the low-level drive pulse Dp. Then, a counter electromotive force E ′ is generated at both ends of the coil L1. At this time, the voltage Vc at the connection point N1 rapidly increases (rises). The voltage Vc is “E + E ′” where E is the voltage of the voltage V20 and E ′ is the back electromotive force of the coil L1.

  As the voltage Vc rises, the diode D17 is turned on, and current starts to flow from the anode to the cathode of the diode D17. As a result, the coil current Ic flows as the current Id, and the current Iq becomes 0A. Thereafter, the coil current Ic generated by the counter electromotive force E ′ of the coil L1 gradually decreases, and the current Id also decreases. At this time, the current Id is accumulated as a charge in the capacitor C2 and supplied to the load 40 via the terminal T1. The output voltage Vout is “E + E′−Vf”, where E is the voltage V20, E ′ is the back electromotive force of the coil L1, and Vf is the forward voltage drop of the diode D17.

  In this way, the pulse generation circuit 31 outputs a high-level conduction pulse and a low-level non-conduction pulse to the transistor Q5, so that the conduction and non-conduction of the transistor Q5 are repeated and the magnetism stored in the coil L1 is accumulated. The current Id is supplied to the output terminal T1 side by energy, and the output voltage Vout is set to a boosted voltage of the voltage V20.

  12 shows a circuit diagram of the power supply circuit 2 in which the diode D17 provided in the power supply circuit 1 of FIG. 10 is replaced with the semiconductor device described in FIG. The reason for this replacement is to prevent power loss due to the forward voltage drop of the diode D17. In the switching regulator 30 'of FIG. 12, the anode A of the semiconductor device described in FIG. 3 is connected to the connection point N1, and the cathode K is connected to the connection point N2. Since the other configuration is the same as that of the power supply circuit 1 described with reference to FIG. 10, in FIG. 12, the same reference numerals are given to the respective parts corresponding to those in FIG.

  When the diode D17 is replaced with the semiconductor device shown in FIG. 3, a reverse current flows from the cathode K to the anode A when the semiconductor device changes from the on state to the off state. Then, this reverse current is absorbed by the ground GND, and the power supplied to the capacitor C2 and the load 40 is lost.

  FIG. 13 is a diagram for explaining the reason why the reverse current flows. From the top, the drive pulse Dp output from the pulse generation circuit 31, the on / off state of the transistor Q5, the voltage Vc at the connection point N1, and the comparator 10 shows the voltage Vcmpin applied to the inverting input terminal 10, the comparison signal CmpOut output from the comparator 10, the on / off state of the transistor Q1, and the current Id.

  First, the operation of the power supply circuit 2 between time X0 and time X1 in FIG. 13 will be described. When the pulse generation circuit 31 of FIG. 12 outputs a high level drive pulse Dp to the transistor Q5, the transistor Q5 is turned on by this high level drive pulse Dp, and the voltage Vc at the node N1 is applied to the ground GND. Become. Further, the voltage at the connection point N2 (output voltage Vout) becomes higher than the ground GND due to the electric charge accumulated in the capacitor C2. That is, a ground voltage is applied to the anode A connected to the connection point N1, and a voltage higher than the ground GND is applied to the cathode K connected to the connection point N2.

  Then, as described in FIG. 1, the voltage Vcmpin corresponding to the threshold voltage Vth1 of the transistors Q2 and Q3 is applied to the inverting input terminal of the comparator 10 in FIG. The comparator 10 outputs a low level comparison signal CmpOut because the voltage (ground GND) applied to the non-inverting input terminal is lower than the voltage Vcmpin applied to the inverting input terminal.

  However, the timing at which the low level comparison signal CmpOut is output is not the same as the timing at which the voltage Vcmpin is applied to the inverting input terminal of the comparator 10, and is slightly delayed as indicated by the symbol DLY. This delay is a delay of about several tens of ns or more in time. This delay is inevitably caused by the characteristics of the comparator which is an analog circuit, and it is extremely difficult to shorten this delay.

  During this delay DLY, the comparator 10 continues to output the high level comparison signal CmpOut, as shown in FIG. 13, and the transistor Q1 remains in the ON state. During this time, a reverse current flows from the cathode K to the anode A. This reverse current is absorbed by the ground GND, and as a result, the power supplied to the capacitor C2 and the load 40 is lost.

  Although the reverse current flows for only several tens of ns, the power loss due to the reverse current cannot be ignored for the following reason. That is, at present, there is an increasing need for downsizing the power supply circuit mounted on the electric device, and in order to realize downsizing of the power supply circuit, it is necessary to downsize the coil L1. In order to reduce the size of the coil L1 and boost the input voltage to a predetermined output voltage, it is necessary to increase the frequency of the drive pulse Dp generated by the pulse generation circuit 31 and shorten the cycle of the drive pulse Dp. Therefore, the frequency of the drive pulse Dp is increased from several hundred KHz to several MHz, and the cycle of the drive pulse Dp is shortened. For example, when the frequency of the drive pulse Dp is 1 MHz, the cycle of the drive pulse Dp is as short as 1 μs.

  When the frequency of the drive pulse Dp is increased and the cycle is shortened in this way, even when the transistor Q1 is switched from the on state to the off state in one cycle of the drive pulse Dp, even in several tens of ns in one cycle of the drive pulse Dp. When a reverse current flows and a power loss occurs, the total power loss due to the reverse current during the operation of the power supply circuit becomes so large that it cannot be ignored. Therefore, in the sixth embodiment, technical means for reliably preventing this reverse current is added to the semiconductor device of FIG.

  FIG. 14 shows a circuit diagram of the sixth embodiment. In the sixth embodiment, in order to prevent the reverse current from flowing, the first level of the comparison signal (control signal) CmpOut output from the comparator 10 is changed to the control signal Ctrl input from the external circuit to the control terminal Cont. A logic circuit that changes to the second level in response, such as an AND circuit 50, is provided between the comparator 10 and the gate of the transistor Q1. The first level is a high level that makes the transistor Q1 conductive, and the second level is a low level that makes the transistor Q1 nonconductive.

  The AND circuit 50 receives the comparison signal CmpOut from the comparator 10 and the control signal Ctrl from the control terminal Cont, takes a logical product of the voltage of the comparison signal CmpOut and the voltage of the control signal Ctrl, and outputs the result as an output signal. Output to the gate of the transistor Q1 as AndOut. The power necessary for the AND circuit 50 to operate is supplied from the capacitor C1 as described with reference to FIG. Since other structures are the same as those of the semiconductor device described with reference to FIG. 3, in FIG. 14, the same reference numerals are given to the respective parts corresponding to those of FIG. Note that the above logic circuit may be added not only to the semiconductor device in FIG. 3 but also to the semiconductor devices in FIGS.

  15 is a circuit diagram of the power supply circuit 5 showing a configuration in which the semiconductor device (high voltage diode) of FIG. 3 in the power supply circuit 2 of FIG. 12 is replaced with the semiconductor device of FIG. In the switching regulator 60 of the power supply circuit 5, the anode A of the semiconductor device of FIG. 14 is connected to the connection point N1, the cathode K is connected to the connection point N2, and the control terminal Cont is an inversion of the drive pulse Dp of the pulse generation circuit 31. A pulse is input via the level converter 33.

  In addition to the configuration of the power supply circuit 2 in FIG. 12, the power supply circuit 5 in FIG. 15 further includes an inverter 32 that outputs to the level converter 33 an inversion pulse Inv obtained by inverting the drive pulse Dp generated by the pulse generation circuit 31; A level converter 33 is provided that converts (increases) the voltage level of the inversion pulse Inv and outputs it as a control signal Ctrl to the control terminal Cont. That is, the external control signal Ctrl based on the drive pulse Dp generated by the pulse generation circuit 31 is input to the AND circuit 50 (logic circuit) of the semiconductor device in FIG.

  The reason why the level converter 33 is provided will be described below. The comparator 10 operates by the electric charge stored in the capacitor C1 and outputs a comparison signal CmpOut. The voltage level of the comparison signal CmpOut is higher than the voltage level of the drive pulse Dp generated by the pulse generation circuit 31. Therefore, the level converter 33 increases the voltage level of the inversion pulse Inv of the drive pulse Dp and outputs it to the AND circuit 50 as the control signal Ctrl. By doing so, the AND circuit 50 can take a logical product of the voltage of the comparison signal CmpOut and the voltage of the control signal Ctrl. The inverter 32 is supplied with a voltage from a voltage source that supplies the voltage V20, and the level converter 33 is supplied with a voltage from a voltage source (not shown), for example. Since other configurations are the same as those of the power supply circuit 1 described with reference to FIG. 10, in FIG. 15, the same reference numerals are given to the respective parts corresponding to FIG. 10, and description thereof is omitted.

  FIG. 16 is a diagram for explaining the operation of the power supply circuit 5 of FIG. 15, in order from the top, the drive pulse Dp output from the pulse generation circuit 31, the on / off state of the transistor Q5, and the voltage Vc at the connection point N1. The voltage Vcmpin applied to the inverting input terminal of the comparator 10, the comparison signal CmpOut output from the comparator 10, the control signal Ctrl, the output signal AndOut of the AND circuit 50, and the ON / OFF state of the transistor Q1 are shown.

  The operation of the power supply circuit 5 in FIG. 15 will be described below with reference to FIG. First, the operation of the power supply circuit 5 between time X0 and time X1 in FIG. 16 will be described. As described with reference to FIG. 13, when the pulse generation circuit 31 outputs the high-level drive pulse Dp to the transistor Q5, the timing at which the voltage Vcmpin corresponding to the threshold voltage Vth1 of the transistor Q3 is applied to the inverting input terminal of the comparator 10. The comparator 10 outputs a low-level comparison signal CmpOut to the AND circuit 50 after the delay DLY. However, a low level external control signal Ctrl, which is an inverted pulse of the high level drive pulse Dp, is input to the AND circuit 50 at a timing when the pulse generation circuit 31 outputs the high level drive pulse Dp to the transistor Q5.

  The AND circuit 50 calculates the logical product of the voltage of the comparison signal CmpOut and the voltage (low level) of the control signal Ctrl, and outputs a low level output signal AndOut to the gate of the transistor Q1. As a result, since the transistor Q1 is turned off in the delay time DLY, it is possible to prevent a reverse current from flowing from the cathode K to the anode A.

  As described above, the transistor Q1 is forcibly turned off based on the control signal from the external circuit, so that it is possible to prevent a reverse current flowing when the semiconductor device functioning as a rectifying element is switched from the on state to the off state. it can.

  Next, the operation between time X1 and time X2 in FIG. 16 will be described. Also in this case, as described with reference to FIG. 11, when the pulse generation circuit 31 outputs the low-level drive pulse Dp to the transistor Q5, the transistor Q5 is turned off (non-conductive state) by the low-level drive pulse Dp. And the voltage Vc rises. Then, the voltage of the cathode K becomes lower than the anode A of the semiconductor device of FIG. As a result, as described in FIG. 1, the negative voltage Vcmpin is applied to the inverting input terminal of the comparator 10 in FIG. 15, and the comparator 10 outputs the high level comparison signal CmpOut to the AND circuit 50. Further, at the timing when the pulse generation circuit 31 outputs the low-level drive pulse Dp to the transistor Q5, a high-level external control signal Ctrl that is an inversion pulse of the low-level drive pulse Dp is input to the AND circuit 50.

  The AND circuit 50 then outputs a high level output signal AndOut to the gate of the transistor Q1. Then, the transistor Q1 is turned on, and the current Id flows through the load 40. Note that the on-state time of the transistor Q1 is shortened by the delay time DLY. However, since this shortening time is as short as about several tens of ns as described above, the power supplied to the load 40 can be prevented from becoming unstable.

  Here, the correspondence with the claims is as follows.

  The transistors Q1, Q2, Q3, Q4, and Q5 are examples of the first, second, third, fourth, and fifth transistors recited in the claims, respectively. The comparator 10 is an example of a comparator described in claims, and the comparator 20 is an example of a comparator described in claims.

  The capacitor C1 is an example of a capacitor described in the claims, and the diode D1 is an example of a diode described in the claims. The voltage source that supplies the voltage V1 is an example of a first voltage source, and the voltage source that supplies the voltage V2 is an example of a second voltage source. The voltage source that supplies the voltage V20 is an example of a third voltage source. The resistors R1 and R2 are examples of voltage dividing resistors. The coil L1 is an example of a coil described in the claims.

  As described above in detail, according to each of the embodiments, a high breakdown voltage and low resistance diode and switch can be realized by combining the high breakdown voltage transistor Q2 and the low breakdown voltage transistor Q1 in a cascode configuration. it can.

  Note that the present invention is not limited to the above-described embodiments, and various modifications and changes can be made without departing from the spirit of the present invention.

  For example, in each of the above embodiments, the transistors Q2 and Q3 are high electron mobility transistors (HEMTs) made of a wide band gap semiconductor such as gallium nitride (GaN). However, the present invention is not limited to this. Other depletion type transistors having high breakdown voltage characteristics may be used.

  The comparator 10 may be an operational amplifier (op amp). For example, the comparator 10 of the fourth and fifth embodiments may be an operational amplifier.

  In the third embodiment of FIG. 4, the transistor Q3 may be connected in multiple stages to adjust the source voltage of the monitoring transistor Q3.

  In addition, it goes without saying that the embodiments may be appropriately combined and used.

  The above embodiment is summarized as follows.

(Appendix 1)
A semiconductor device that functions as a rectifying element,
A first transistor whose source is the anode of the rectifying element;
A depletion type second having a higher breakdown voltage than the first transistor, a source connected to the drain of the first transistor, a gate connected to the source of the first transistor, and a drain serving as the cathode of the rectifier element. Transistors
A depletion-type third transistor having a higher breakdown voltage than the first transistor and having a drain connected to the drain of the second transistor;
A comparator that compares the source voltage of the first transistor with the source voltage of the third transistor and controls the gate voltage of the first transistor;
A semiconductor device comprising:

(Appendix 2)
One power supply terminal of the comparator is connected to a source of the first transistor;
A fourth transistor connected between the source of the third transistor and the other power supply terminal of the comparator, the gate voltage of which is controlled by the comparator;
A capacitor connected between both power supply terminals of the comparator;
The semiconductor device according to appendix 1, wherein:

(Appendix 3)
One power supply terminal of the comparator is connected to a source of the first transistor;
A diode connected between the source of the third transistor and the other power supply terminal of the comparator;
A capacitor connected between both power supply terminals of the comparator;
The semiconductor device according to appendix 1, wherein:

(Appendix 4)
4. The semiconductor device according to any one of appendices 1 to 3, wherein a gate of the third transistor is connected to a source of the second transistor.

(Appendix 5)
The semiconductor device according to any one of appendices 1 to 3, wherein a gate of the third transistor is connected to a gate of the second transistor.

(Appendix 6)
A first voltage source having a negative terminal connected to the source of the first transistor and outputting a voltage obtained by adding the first voltage to the source voltage of the first transistor to the positive terminal;
A voltage dividing resistor connected in series between the source of the third transistor and the positive terminal of the first voltage source;
With
The semiconductor device according to any one of appendices 1 to 6, wherein one input terminal of the comparator is connected to a voltage dividing point of the voltage dividing resistor.

(Appendix 7)
The source of the first transistor is connected to a non-inverting input terminal of the comparator, and the source of the third transistor is connected to an inverting input terminal of the comparator. A semiconductor device according to claim 1.

(Appendix 8)
A semiconductor device that functions as a switching element in which rectifying elements are connected in parallel,
A first transistor whose source is one terminal of the switching element;
A depletion type having a higher breakdown voltage than the first transistor, a source connected to the drain of the first transistor, a gate connected to the source of the first transistor, and a drain serving as the other terminal of the switching element. A second transistor;
A depletion-type third transistor having a higher breakdown voltage than the first transistor and having a drain connected to the drain of the second transistor;
A first voltage source having a negative terminal connected to the source of the first transistor and outputting a voltage obtained by adding the first voltage to the source voltage of the first transistor to the positive terminal;
A voltage dividing resistor connected in series between the source of the third transistor and the positive terminal of the first voltage source;
A second voltage source having a negative terminal connected to the source of the first transistor and outputting a voltage obtained by adding a second voltage to the voltage of the source of the first transistor to the positive terminal;
A comparator for comparing the voltage of the control signal of the switching element and the output voltage of the second voltage source;
A comparator that compares an output voltage of the comparator with a divided voltage of the voltage dividing resistor and controls a gate voltage of the first transistor;
A semiconductor device comprising:

(Appendix 9)
Further, a first level for conducting the first transistor of the control signal output from the comparator between the gate of the comparator and the first transistor in response to the external control signal is set to the first transistor. A logic circuit for making the second level non-conductive,
The semiconductor device according to any one of appendices 1 to 5, characterized in that:

(Appendix 10)
Additional notes 1 to 9, wherein the second transistor and the third transistor are high electron mobility transistors (HEMTs), and the first transistor is a silicon MOSFET. The semiconductor device according to any one of the above.

(Appendix 11)
A bridge rectifier circuit that full-wave rectifies an AC voltage by incorporating the semiconductor device according to appendix 6 into a bridge circuit.

(Appendix 12)
A plurality of series circuits in which the two semiconductor devices according to appendix 8 are connected in series are connected in a bridge shape between the power supplies, and coils of each phase of the multiphase motor are connected to each connection point between the two semiconductor devices in the plurality of series circuits. A drive circuit for a multi-phase motor that connects and outputs a control signal to a comparator of these semiconductor devices.

(Appendix 13)
A coil connected to a third voltage source;
A fifth transistor provided between a reference power source and the coil;
The semiconductor device according to appendix 9, wherein an anode is connected to a connection point between the coil and the fifth transistor;
A pulse generation circuit for generating a pulse for driving the fifth transistor;
With
The switching regulator, wherein the external control signal based on the drive pulse generated by the pulse generation circuit is input to a logic circuit of the semiconductor device.

10 Comparator
20 Comparator
1, 2, 5 Power supply circuit 30, 30 ', 60 Switching regulator 31 Pulse generation circuit 32 Inverter,
33 level converter,
40 load,
50 AND circuit,
C1 Capacitor D1 Diode Q1-Q5 Transistor R1, R2 Resistance L1 Coil

Claims (10)

A semiconductor device that functions as a rectifying element,
A first transistor whose source is the anode of the rectifying element;
A depletion type second having a higher breakdown voltage than the first transistor, a source connected to the drain of the first transistor, a gate connected to the source of the first transistor, and a drain serving as the cathode of the rectifier element. Transistors
A depletion-type third transistor having a higher breakdown voltage than the first transistor and having a drain connected to the drain of the second transistor;
A comparator that compares the source voltage of the first transistor with the source voltage of the third transistor and controls the gate voltage of the first transistor;
A semiconductor device comprising: One power supply terminal of the comparator is connected to a source of the first transistor;
A fourth transistor connected between the source of the third transistor and the other power supply terminal of the comparator, the gate voltage of which is controlled by the comparator;
A capacitor connected between both power supply terminals of the comparator;
The semiconductor device according to claim 1, comprising: One power supply terminal of the comparator is connected to a source of the first transistor;
A diode connected between the source of the third transistor and the other power supply terminal of the comparator;
A capacitor connected between both power supply terminals of the comparator;
The semiconductor device according to claim 1, comprising: 4. The semiconductor device according to claim 1, wherein a gate of the third transistor is connected to a source of the second transistor. 5. The semiconductor device according to claim 1, wherein a gate of the third transistor is connected to a gate of the second transistor. A first voltage source having a negative terminal connected to the source of the first transistor and outputting a voltage obtained by adding the first voltage to the source voltage of the first transistor to the positive terminal;
A voltage dividing resistor connected in series between the source of the third transistor and the positive terminal of the first voltage source;
With
The semiconductor device according to any one of claims 1 to 5, characterized in that one input terminal of the comparator is connected to the dividing point of the voltage dividing resistors. The source of the first transistor is connected to a non-inverting input terminal of the comparator, and the source of the third transistor is connected to an inverting input terminal of the comparator. The semiconductor device according to any one of the above. A semiconductor device that functions as a switching element in which rectifying elements are connected in parallel,
A first transistor whose source is one terminal of the switching element;
A depletion type having a higher breakdown voltage than the first transistor, a source connected to the drain of the first transistor, a gate connected to the source of the first transistor, and a drain serving as the other terminal of the switching element. A second transistor;
A depletion-type third transistor having a higher breakdown voltage than the first transistor and having a drain connected to the drain of the second transistor;
A first voltage source having a negative terminal connected to the source of the first transistor and outputting a voltage obtained by adding the first voltage to the source voltage of the first transistor to the positive terminal;
A voltage dividing resistor connected in series between the source of the third transistor and the positive terminal of the first voltage source;
A second voltage source having a negative terminal connected to the source of the first transistor and outputting a voltage obtained by adding a second voltage to the voltage of the source of the first transistor to the positive terminal;
A comparator for comparing the voltage of the control signal of the switching element and the output voltage of the second voltage source;
A comparator that compares an output voltage of the comparator with a divided voltage of the voltage dividing resistor and controls a gate voltage of the first transistor;
A semiconductor device comprising: Further, a first level for conducting the first transistor of the control signal output from the comparator between the gate of the comparator and the first transistor in response to the external control signal is set to the first transistor. A logic circuit for making the second level non-conductive,
The semiconductor device according to claim 1, wherein: 10. The second transistor and the third transistor are high electron mobility transistors (HEMT), and the first transistor is a silicon MOSFET, and the first transistor is a silicon MOSFET. The semiconductor device according to any one of the above.

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