JP5470697B2 - Electric power steering device - Google Patents

Description

The present invention relates to electric power steering apparatus.

  Conventionally, many motor control devices used in an electric power steering device (EPS) or the like have any phase of the motor (any of U, V, and W) due to disconnection of a power supply line or a contact failure of a drive circuit. An abnormality detection means is provided that can detect the occurrence of the abnormality when a current-carrying failure occurs. And when generation | occurrence | production of the said abnormality is detected, the structure which stops motor control rapidly and aims at fail safe is common.

However, in EPS, as the motor control stops, the steering characteristics change greatly. That is, in order for the driver to perform an appropriate steering operation, a larger steering force is required. Based on this point, conventionally, there is a motor control device that continues motor control using two phases other than the current-carrying failure phase as a current-carrying phase even when the occurrence of a current-carrying failure phase is detected as described above (for example, Patent Document 1). As a result, the application of assist force to the steering system can be continued, and an increase in the driver's burden associated with fail-safe can be avoided.
JP 200326020 A

  However, when the motor control is continued using the two phases other than the current supply failure occurrence phase as the current supply phase when the current supply failure phase is generated as in the above-described conventional example, for each current supply phase as shown in FIG. In the configuration in which sine wave energization is performed (the example shown in the figure is when the U phase is abnormal and the V and W phases are energized), the steering feeling is inevitably deteriorated due to the occurrence of torque ripple.

  That is, as shown in FIG. 24, when the transition of the motor current during the conventional two-phase drive is expressed in the d / q coordinate system, the q-axis current command value that is the control target value of the motor torque is constant. Regardless, the actual q-axis current value changes sinusoidally. In other words, there is a problem that the assist force fluctuates greatly when the motor drive is continued in a state where the original output performance cannot be obtained because the motor current corresponding to the required torque is not generated. The room was left.

The present invention was made to solve the above problems, and its object is to enable you to ensure a smooth to high output performance of the motor rotation during two phase drive due to the generation of the current-carrying failure and to provide a conductive power steering apparatus.

In order to solve the above problems, the invention according to claim 1 is a motor control signal output means for outputting a motor control signal, and a drive circuit for supplying three-phase drive power to the motor based on the motor control signal. And an abnormality detecting means capable of detecting an energization failure occurring in each phase of the motor, wherein the motor control signal output means generates the motor control signal by executing current control based on the rotation angle of the motor. In addition, when the energization failure occurs, in the electric power steering apparatus that executes the output of the motor control signal using the two phases other than the energization failure occurrence phase as the energization phase, when the energization failure occurs, the motor control signal output means Passes through each energized phase a phase current that changes in a secant curve or a cosecant curve with an asymptotic line of a predetermined rotation angle corresponding to the energization failure occurrence phase. Ku executes the current control, and performs a current limiting for limiting the phase current within a predetermined range, to correct the rotation angle to the steering direction and the same direction, and the gist.

According to the above configuration, the required torque can be obtained except for the predetermined rotation angle corresponding to the asymptote (the current limit range in the vicinity of the predetermined rotation angle when there is a limit to the phase current value energized in each phase). A corresponding motor current can be generated. As a result, even when a poorly energized phase is generated, the motor drive can be continued while ensuring high output performance without causing large torque ripple. Also, smooth motor rotation without so-called catching can be realized. That is, in an electric power steering apparatus using a motor as a drive source, the influence of the output performance of the motor and the smoothness of the rotation on the steering feeling is extremely large. In this respect, according to the above-described configuration, it is possible to realize a good steering feeling that is excellent in followability to the steering operation and has no catching feeling even during two-phase driving.

  The gist of the invention described in claim 2 is that the motor control signal output means corrects the rotation angle in the delay direction when the steering direction and the rotation direction of the motor are different.

  According to a third aspect of the present invention, the motor control signal output means corrects the rotation angle in the advance direction when the steering direction and the rotation direction of the motor match.
Is the gist.

  In the current control, the phase of the actual current supplied to the motor due to the presence of a time delay element such as a mechanical factor such as the motor “rat”, a delay in calculation time, or a phase delay in current control, There is a tendency to lag behind the phase of the current command value in the current control. Therefore, the rotation of the motor can be smoothed by compensating for the phase delay of the actual current value with respect to the current command value by proceeding with the rotation in the advance direction in consideration of the time delay element during the two-phase driving.

The gist of the invention described in claim 4 is that the motor control signal output means corrects the rotational angle in the advance direction as the rotational angular velocity of the motor increases.
  That is, the phase delay of the actual current value with respect to the current command value caused by the time delay element becomes more remarkable as the rotational angular velocity of the motor is faster. Therefore, according to the above configuration, the phase delay of the actual current value with respect to the current command value can be compensated more accurately.

  The gist of the invention described in claim 5 is that the motor control signal output means is that the current control when the energization failure occurs is current feedback control.
  In other words, one of the most influential time delay elements is a phase delay in current feedback control. Therefore, as described above, by applying the current feedback control to the one that outputs the motor control signal (supply of driving power) when the energization failure occurs, a more remarkable effect can be obtained. .

  According to a sixth aspect of the present invention, as the current control, the motor control signal output means executes current feedback control in a d / q coordinate system when the energization failure does not occur, and phase current when the energization failure occurs. The gist is to execute feedback control.
  That is, phase current feedback control tends to have a larger phase delay than current feedback control in the d / q coordinate system. Therefore, a more prominent effect can be obtained by applying it to a motor that outputs a motor control signal by executing phase current feedback control during two-phase driving as in the above configuration.

According to the present invention, it is possible to provide a photoelectric power steering apparatus which can ensure a smooth to high output performance of the motor rotation during two phase drive due to the generation of the current-carrying failure.

Hereinafter, an embodiment in which the present invention is embodied in an electric power steering apparatus (EPS) will be described with reference to the drawings.
FIG. 1 is a schematic configuration diagram of the EPS 1 of the present embodiment. As shown in the figure, a steering shaft 3 to which a steering wheel (steering) 2 is fixed is connected to a rack 5 via a rack and pinion mechanism 4. It is converted into a reciprocating linear motion of the rack 5 by the and pinion mechanism 4. The rudder angle of the steered wheels 6 is changed by the reciprocating linear motion of the rack 5.

  Further, the EPS 1 includes an EPS actuator 10 as a steering force assisting device that applies an assist force for assisting a steering operation to the steering system, and an ECU 11 as a control unit that controls the operation of the EPS actuator 10. .

  The EPS actuator 10 of the present embodiment is a so-called rack-type EPS actuator in which a motor 12 that is a driving source thereof is arranged coaxially with the rack 5, and an assist torque generated by the motor 12 is a ball screw mechanism (not shown). Is transmitted to the rack 5 via. In addition, the motor 12 of this embodiment is a brushless motor, and rotates by receiving supply of three-phase (U, V, W) driving power from the ECU 11. And ECU11 as a motor control apparatus controls the assist force given to a steering system by controlling the assist torque which this motor 12 generate | occur | produces (power assist control).

  In the present embodiment, a torque sensor 14 and a vehicle speed sensor 15 are connected to the ECU 11. Then, the ECU 11 executes the operation of the EPS actuator 10, that is, power assist control, based on the steering torque τ and the vehicle speed V detected by the torque sensor 14 and the vehicle speed sensor 15, respectively.

Next, the electrical configuration of the EPS of this embodiment will be described.
FIG. 2 is a control block diagram of the EPS of this embodiment. As shown in the figure, the ECU 11 includes a microcomputer 17 as motor control signal output means for outputting a motor control signal, and a drive circuit 18 for supplying three-phase drive power to the motor 12 based on the motor control signal. ing.

  The drive circuit 18 of this embodiment is a known PWM inverter in which three arms corresponding to each phase are connected in parallel with a pair of switching elements connected in series as a basic unit (arm). The output of the motor control signal defines the on-duty ratio of each switching element constituting the drive circuit 18. Then, a motor control signal is applied to the gate terminal of each switching element, and each switching element is turned on / off in response to the motor control signal, so that the DC voltage of the in-vehicle power supply (not shown) becomes three-phase (U, V, W) is converted into drive power and supplied to the motor 12.

  In the present embodiment, the ECU 11 detects the current sensors 21u, 21v, 21w for detecting the phase current values Iu, Iv, Iw energized to the motor 12, and the rotation angle (electrical angle) θ of the motor 12. A rotation angle sensor 22 is connected for this purpose. Then, the microcomputer 17 sends a motor to the drive circuit 18 based on the phase current values Iu, Iv, Iw and the rotation angle θ of the motor 12 detected based on the output signals of these sensors, and the steering torque τ and the vehicle speed V. Output a control signal.

  More specifically, the microcomputer 17 according to the present embodiment determines an assist force (target assist force) to be applied to the steering system based on the steering torque τ and the vehicle speed V, and causes the motor 12 to generate the assist force. The motor control signal is generated by executing current control based on the detected phase current values Iu, Iv, Iw and the rotation angle θ.

  Specifically, the microcomputer 17 includes a current command value calculation unit 23 as a current command value calculation unit that calculates a current command value as an assist force to be applied to the steering system, that is, a motor torque control target value, and a current command value calculation. And a motor control signal generation unit 24 as a motor control signal generation unit that generates a motor control signal based on the current command value calculated by the unit 23.

  Based on the steering torque τ and the vehicle speed V detected by the torque sensor 14 and the vehicle speed sensor 15, the current command value calculation unit 23 sets q in the d / q coordinate system as a current command value corresponding to the control target value of the motor torque. The shaft current command value Iq * is calculated and output to the motor control signal generator 24. On the other hand, the motor control signal generator 24 includes the q-axis current command value Iq * output from the current command value calculator 23 and the phase current values Iu, Iv, Iw detected by the current sensors 21u, 21v, 21w. , And the rotation angle θ detected by the rotation angle sensor 22 is input. Then, the motor control signal generation unit 24 executes current feedback control in the d / q coordinate system based on the phase current values Iu, Iv, Iw and the rotation angle θ (electrical angle), thereby performing the motor control signal. Is generated.

  More specifically, the motor control signal generation unit 24 of the present embodiment performs three-phase phase voltage command values Vu * and Vv by executing current feedback control (d / q axis current F / B) in the d / q coordinate system. A first current control unit 24a for calculating * and Vw * is provided. At normal times, a motor control signal is generated based on the phase voltage command values Vu *, Vv *, and Vw * calculated by the first current control unit 24a.

  As shown in FIG. 3, each phase current value Iu, Iv, Iw input to the first current control unit 24a is input to the three-phase / two-phase conversion unit 25 together with the rotation angle θ, and the three-phase / 2-phase The converter 25 converts the d / q coordinate system into a d-axis current value Id and a q-axis current value Iq. The q-axis current value Iq is input to the subtractor 26q together with the q-axis current command value Iq * input from the current command value calculation unit 23, and the d-axis current value Id is the d-axis current command value Id * (Id). * = 0) and input to the subtractor 26d.

  The d-axis current deviation ΔId and q-axis current deviation ΔIq calculated in the subtracters 26d and 26q are input to the corresponding F / B control units 27d and 27q, respectively. In each of the F / B control units 27d and 27q, the d-axis current command value Id * and the q-axis current command value Iq * output from the current command value calculation unit 23 are the d-axis current value Id and the actual current value. Feedback control is performed to follow the q-axis current value Iq.

  That is, the F / B control units 27d and 27q multiply the inputted d-axis current deviation ΔId and q-axis current deviation ΔIq by a predetermined F / B gain (PI gain), thereby obtaining a d-axis voltage command value Vd * and The q-axis voltage command value Vq * is calculated. The calculated d-axis voltage command value Vd * and q-axis voltage command value Vq * are input to the two-phase / three-phase conversion unit 28 together with the rotation angle θ, and the two-phase / three-phase conversion unit 28 outputs three-phase signals. Converted to phase voltage command values Vu *, Vv *, Vw *. Then, the first current control unit 24 a outputs the phase voltage command values Vu *, Vv *, and Vw * to the PWM conversion unit 30.

  The PWM converter 30 generates duty command values αu, αv, αw based on the input phase voltage command values Vu *, Vv *, Vw *, and further indicates the duty command values αu, αv, αw. A motor control signal having an on-duty ratio is generated. Then, as shown in FIG. 2, the microcomputer 17 outputs the motor control signal generated by the motor control signal generator 24 to each switching element (the gate terminal thereof) constituting the drive circuit 18, thereby The operation of the drive circuit 18, that is, the supply of drive power to the motor 12 is controlled.

[Control mode when an abnormality occurs]
As shown in FIG. 2, in the ECU 11 of the present embodiment, the microcomputer 17 is provided with an abnormality determination unit 31 for identifying the abnormality mode when any abnormality occurs in the EPS 1. Then, the ECU 11 (microcomputer 17) changes the control mode of the motor 12 according to the abnormality mode specified (determined) by the abnormality determination unit 31.

  More specifically, an abnormality signal S_tr for detecting an abnormality in the mechanical system of the EPS actuator 10 is input to the abnormality determination unit 31, and the abnormality determination unit 31 receives the input abnormality signal. Based on S_tr, an abnormality in the mechanical system in EPS 1 is detected. The abnormality determination unit 31 includes the detected phase current values Iu, Iv, Iw, the rotational angular velocity ω, and the q-axis current deviation calculated in the motor control signal generation unit 24 (first current control unit 24a). ΔIq and duty command values αu, αv, αw, etc. for each phase are input. And the abnormality determination part 31 detects generation | occurrence | production of abnormality in a control system based on each of these state quantities.

  Specifically, the abnormality determination unit 31 of the present embodiment monitors the q-axis current deviation ΔIq in order to detect the occurrence of an abnormality related to the entire control system, such as a failure of the torque sensor 14 or a failure of the drive circuit 18. That is, the q-axis current deviation ΔIq is compared with a predetermined threshold value, and if the q-axis current deviation ΔIq is equal to or greater than the threshold value (continuous for a predetermined time), it is determined that an abnormality has occurred in the control system. To do.

  Further, the abnormality determination unit 31 disconnects or drives a power line (including a motor coil) based on the phase current values Iu, Iv, Iw, the rotational angular velocity ω, and the duty command values αu, αv, αw of each phase. The occurrence of an energization failure phase caused by a contact failure of the circuit 18 is detected. The detection of the occurrence of a poorly energized phase is performed by detecting the phase current value Ix of the X phase (X = U, V, W) below the predetermined value Ith (| Ix | ≦ Ith) and the rotational angular velocity ω within the target range for disconnection determination ( When | ω | ≦ ω0), the state in which the duty command value αx corresponding to the phase is not in the predetermined range (αLo ≦ αx ≦ αHi) corresponding to the predetermined value Ith and the threshold value ω0 defining the determination target range continues. Depending on whether or not.

  In this case, the predetermined value Ith serving as the threshold value of the phase current value Ix is set to a value near “0”, and the threshold value ω0 of the rotational angular velocity ω is set to a value corresponding to the maximum rotational speed of the motor. . The threshold values (αLo, αHi) relating to the duty command value αx are set to a value smaller than a lower limit value that can be taken by the duty command value αx and a value larger than the upper limit value in normal control.

  That is, as shown in the flowchart of FIG. 4, the abnormality determination unit 31 determines whether or not the detected phase current value Ix (absolute value thereof) is equal to or less than a predetermined value Ith (step 101), and the predetermined value Ith If it is below (| Ix | ≦ Ith, Step 101: YES), it is subsequently determined whether or not the rotational angular velocity ω (the absolute value thereof) is not more than a predetermined threshold value ω0 (Step 102). When the rotational angular velocity ω is equal to or smaller than the predetermined threshold ω0 (| ω | ≦ ω0, step 102), it is determined whether the duty command value αx is within the predetermined range (αLo ≦ αx ≦ αHi). If it is determined (step 103) and it is not within the predetermined range (step 103: NO), it is determined that an energization failure has occurred in the X phase (step 104).

  When the phase current value Ix is larger than the predetermined value Ith (| Ix |> Ith, step 101: NO), when the rotational angular velocity ω is larger than the threshold ω0 (| ω |> ω0, step 102: NO), Alternatively, if the duty command value αx is within the predetermined range (αLo ≦ αx ≦ αHi, step 103: YES), it is determined that no energization failure has occurred in the X phase (normal X phase, step 105).

  That is, when an energization failure (disconnection) occurs in the X phase (any one of the U, V, and W phases), the phase current value Ix of the phase is “0”. Here, in the case where the X-phase phase current value Ix is “0” or “a value close to 0”, there may be the following two cases in addition to the occurrence of such disconnection.

− When the rotational angular velocity of the motor reaches the maximum number of rotations − When the current command itself is substantially “0” Based on this point, in the present embodiment, first, the X-phase phase current value Ix to be determined is set to a predetermined value. By comparing with the value Ith, it is determined whether or not the phase current value Ix is “0”. Then, it is determined whether or not the two cases where the phase current value Ix is “0” or “a value close to 0” except when the wire is disconnected. It is determined that a disconnection has occurred.

  That is, if the extreme duty command value αx is output even though the rotational angular velocity ω is not such that the phase current value Ix is equal to or less than the predetermined value Ith in the vicinity of “0”, the energization failure of the X phase is poor. Can be determined to have occurred. And in this embodiment, it has the structure which specifies the phase in which the conduction failure generate | occur | produced by performing the said determination sequentially about each phase of U, V, and W. FIG.

  Although not shown in the flowchart of FIG. 4 for convenience of explanation, the above determination is made on the assumption that the power supply voltage is equal to or higher than a specified voltage necessary for driving the motor 12. Then, the final abnormality detection determination is made based on whether or not the state in which it is determined that the energization failure has occurred in the predetermined step 104 has continued for a predetermined time or more.

  In the present embodiment, the ECU 11 (microcomputer 17) switches the control mode of the motor 12 based on the result of the abnormality determination in the abnormality determination unit 31. Specifically, the abnormality determination unit 31 outputs the result of abnormality determination including the above-described failure of energization as an abnormality detection signal S_tm, and the current command value calculation unit 23 and the motor control signal generation unit 24 receive the input. The calculation of the current command value according to the abnormality detection signal S_tm to be performed and the generation of the motor control signal are executed. As a result, the control mode of the motor 12 in the microcomputer 17 can be switched.

  More specifically, the ECU 11 of the present embodiment is a “normal control mode” that is a normal control mode, and a “assist stop mode” that is a control mode when an abnormality that should stop driving the motor 12 occurs. , And “two-phase drive mode” which is a control mode in the case where energization failure occurs in any of the phases of the motor 12, and the above three broad control modes. When the abnormality detection signal S_tm output from the abnormality determination unit 31 corresponds to the “normal control mode”, the current command value calculation unit 23 and the motor control signal generation unit 24 respectively Calculation of the current command value and generation of the motor control signal are executed.

  On the other hand, when the abnormality detection signal S_tm output from the abnormality determination unit 31 is in the “assist stop mode”, the current command value calculation unit 23 and the motor control signal generation unit 24 respectively stop driving the motor 12. Calculation of the current command value and generation of the motor control signal are executed. The “assist stop mode” is selected not only when an abnormality occurs in the mechanical system or the torque sensor 14, but when an abnormality occurs in the power supply system, an overcurrent may occur. Can be mentioned. In addition, in the “assist stop mode”, there is a case where the output of the motor 12 is gradually reduced, that is, the assist force is gradually reduced and then stopped after the drive of the motor 12 is stopped immediately. The motor control signal generator 24 gradually reduces the value (absolute value) of the q-axis current command value Iq * output as the current command value. Then, after the motor 12 is stopped, the microcomputer 17 is configured to open each switching element constituting the drive circuit 18 and open a power relay (not shown).

  Further, the abnormality detection signal S_tm corresponding to the “two-phase drive mode” includes information for specifying the energization failure occurrence phase. When the abnormality detection signal S_tm output from the abnormality determination unit 31 corresponds to this “two-phase drive mode”, the motor control signal generation unit 24 sets two phases other than the current-carrying failure occurrence phase as current-carrying phases. In order to continue motor driving, the motor control signal is generated.

  More specifically, as shown in FIG. 2, the motor control signal generator 24 of this embodiment obtains the phase voltage command values Vu *, Vv *, Vw * by executing current feedback control in the d / q coordinate system. In addition to the first current control unit 24a that calculates, a second current control unit 24b that calculates the phase voltage command values Vu **, Vv **, and Vw ** by executing phase current feedback control is provided. When the abnormality detection signal S_tm input from the abnormality determination unit 31 corresponds to the “two-phase drive mode”, each phase voltage command value Vu * calculated by the second current control unit 24b. *, Vv ** and Vw ** are used to output motor control signals.

  More specifically, as shown in FIG. 3, the second current control unit 24 b of the present embodiment selects a control phase that selects one of the remaining two phases other than the detected conduction failure phase as a control phase. And a phase current command value calculation unit 33 for calculating a phase current command value Ix * (X = U, V, or W) for the phase selected as the control phase. Then, by executing the phase current feedback control based on the deviation between the phase current value Ix selected as the control phase and the phase current command value Ix * (Ix **), two phases other than the conduction failure occurrence phase are transferred to the conduction phase. Each phase voltage command value Vu **, Vv **, Vw ** is calculated to execute the motor drive.

  Specifically, the phase current command value Ix * output from the phase current command value calculation unit 33 is input to the guard processing unit 34. The phase current command value Ix ** after the guard process is input to the subtractor 35 together with the phase current value Ix of the phase selected as the control phase by the control phase selection unit 32. The subtractor 35 calculates the phase current deviation ΔIx by subtracting the phase current value Ix from the phase current command value Ix *, and outputs the calculated phase current deviation ΔIx to the F / B control unit 36. The F / B control unit 36 calculates the phase voltage command value Vx * for the control phase by multiplying the input phase current deviation ΔIx by a predetermined F / B gain (PI gain).

  The phase voltage command value Vx * calculated by the F / B control unit 36 is input to the phase voltage command value calculation unit 37. And the phase voltage command value calculating part 37 calculates each phase voltage command value Vu **, Vv **, Vw ** based on the phase voltage command value Vx * about the control phase.

  That is, the energization failure occurrence phase cannot be energized, and the phase of each energized phase during two-phase driving is shifted by π / 2 (180 °). Accordingly, the phase voltage command value of the phase where the power failure has occurred is “0”, and the phase voltage command value of the other current phase can be calculated by inverting the sign of the phase voltage command value Vx * related to the control phase. The second current control unit 24b of the present embodiment is configured to output the phase voltage command values Vu **, Vv **, and Vw ** calculated in this way to the PWM conversion unit 30. ing.

  Here, the phase current command value calculation unit 33 according to the present embodiment, when performing two-phase driving, excludes a predetermined rotation angle corresponding to the phase where the energization failure occurs, and the control target value (q-axis) of the required torque, that is, the motor torque. A phase current command value Ix * is calculated such that a motor current (q-axis current value Iq) corresponding to the current command value Iq *) is generated.

  Specifically, the phase current command value calculation unit 33 is a phase current command value of one of the remaining two phases based on the following formulas (1) to (3) according to the energization failure occurrence phase. Calculate Ix *.

That is, according to the above formulas (1) to (3), the secant curve (reciprocal of cos θ (secant: sec θ)) or cosecant curve with predetermined rotation angles θA and θB corresponding to the energization failure occurrence phase as asymptotic lines. A phase current command value Ix * that changes in the form (reciprocal of sin θ (cosecant: cosec θ)) is calculated (see FIG. 5). Then, by executing the phase current feedback control based on the phase current command value Ix * that changes into such a secant curve or a cosecant curve, theoretically, predetermined rotation angles θA, θB corresponding to the asymptote Except for the motor current (q-axis current value Iq) corresponding to the required torque (q-axis current command value Iq *) can be generated (see FIG. 6).

  5 and 6 are examples in which the U phase is a current-carrying phase and the two phases V and W are current-carrying phases. Of the two rotation angles corresponding to the asymptotic lines, In the electrical angle range of 0 ° to 360 °, when the smaller value is the rotation angle θA and the larger value is the rotation angle θB, the rotation angles θA and θB are “90 °” and “270 °, respectively. " The predetermined rotation angles θA and θB when the V phase is a current-carrying failure occurrence phase are “30 °” and “210 °”, respectively, and the predetermined rotation angles θA and θA when the W-phase is a current conduction failure occurrence phase. θB is “150 °” and “330 °”, respectively (not shown).

  In practice, since there is an upper limit to the current (absolute value) that can be passed through the motor coils 12u, 12v, 12w of each phase, in the present embodiment, the guard processing unit 34 calculates the phase current command value. A guard process for limiting the phase current command value Ix * output from the unit 33 to within a predetermined range (−Ix_max ≦ Ix * ≦ Ix_max) is executed. Note that “Ix_max” is the maximum value of current that can be passed through the X phase (U, V, W phase), and this maximum value is defined by the rated current of each switching element constituting the drive circuit 18. The Therefore, in the range where the guard process is performed (current limiting range: θ1 <θ <θ2, θ3 <θ <θ4), the phase current command value Ix ** after the guard process is the upper limit value ( Ix_max) or the lower limit (−Ix_max).

  That is, the microcomputer 17 of the present embodiment performs the phase current feedback control so as to energize each energized phase with a secant curve or a cosecant curve during two-phase driving, Except for current limiting ranges (θ1 <θ <θ2, θ3 <θ <θ4) set in the vicinity of predetermined rotation angles θA and θB corresponding to asymptotes, a motor current corresponding to the required torque is generated. As a result, even when an energization failure phase occurs, the assist force is continuously applied while maintaining a good steering feeling without causing a large torque ripple.

Next, the abnormality determination and control mode switching by the microcomputer and the motor control signal generation processing procedure during two-phase driving will be described.
As shown in the flowchart of FIG. 7, the microcomputer 17 first determines whether or not any abnormality has occurred (step 201). If it is determined that an abnormality has occurred (step 201: YES), then It is determined whether the abnormality is a control system abnormality (step 202). Next, when it is determined in step 202 that a control abnormality has occurred (step 202: YES), it is determined whether or not the current control mode is the two-phase drive mode (step 203), and the two-phase drive mode is determined. If not (step 203: NO), it is determined whether or not the abnormality of the control system is the occurrence of a poorly energized phase (step 204). If it is determined that a current-carrying failure phase has occurred (step 204: YES), a motor control signal is output that uses the remaining two phases other than the current-carrying failure phase as a current-carrying phase (two-phase drive mode, step). 205).

  As described above, the output of the motor control signal in the two-phase drive mode is a phase current command that changes into a secant curve or a remainder curve with predetermined rotation angles θA and θB corresponding to the energization failure occurrence phase as asymptotic lines. This is performed by calculating a value and executing phase current feedback control based on the phase current command value.

  That is, as shown in the flowchart of FIG. 8, the microcomputer 17 first determines whether or not the current-carrying failure occurrence phase is the U phase (step 301), and if it is the U phase (step 301: YES). In step S302, the phase current command value Iv * for the V phase is calculated based on the equation (1). Next, the microcomputer 17 executes a guard process calculation for the phase current command value Iv *, and limits the phase current command value Iv ** after the guard process within a predetermined range (step 303). Then, by executing phase current feedback control based on the phase current command value Iv ** after the guard processing, a phase voltage command value Vv * for the V phase is calculated (step 304), and based on the phase voltage command value Vv *. Thus, phase voltage command values Vu **, Vv **, and Vw ** for each phase are calculated (Vu ** = 0, Vv ** = Vv *, Vw ** =-Vv *, step 305).

  On the other hand, if it is determined in step 301 that the energization failure occurrence phase is not the U phase (step 301: NO), the microcomputer 17 determines whether the energization failure occurrence phase is the V phase (step 306). When the defect occurrence phase is the V phase (step 306: YES), the phase current command value Iu * for the U phase is calculated based on the above equation (2) (step 307). Next, the microcomputer 17 executes a guard process calculation for the phase current command value Iu *, and limits the phase current command value Iu ** after the guard process within a predetermined range (step 308). Then, phase current feedback control based on the phase current command value Iv ** after the guard processing is executed (step 309), and each phase voltage command value Vu * calculated by the execution of the phase current feedback control is The phase voltage command values Vu **, Vv **, and Vw ** of the phase are calculated (Vu ** = Vu *, Vv ** = 0, Vw ** = − Vu *, step 310).

  If it is determined in step 306 that the energization failure occurrence phase is not the V phase (step 306: NO), the microcomputer 17 determines the phase current command value Iv * for the V phase based on the above equation (3). Is calculated (step 311), and then the guard process calculation is executed to limit the phase current command value Iv ** after the guard process within a predetermined range (step 312). Then, the phase current feedback control based on the phase current command value Iv ** after the guard processing is executed (step 313), and the remaining based on the phase voltage command value Vv * calculated by the execution of the phase current feedback control. Two-phase (V, W phase) phase voltage command values Vu ** and Vw ** are calculated (Vu ** = − Vv *, Vv ** = Vv *, Vw ** = 0, step 314).

  The microcomputer 17 generates a motor control signal based on the phase voltage command values Vu **, Vv **, and Vw ** calculated in step 305, step 310, or step 314, and outputs the motor control signal to the drive circuit 18. (Step 315).

  If it is determined in step 201 that there is no abnormality (step 201: NO), as described above, the microcomputer 17 executes the current feedback control in the d / q coordinate system to execute the motor control signal. The output is executed (normal control mode, step 206). Further, when it is determined in step 202 that an abnormality other than the control system has occurred (step 202: NO), when it is determined in step 203 that the two-phase drive mode has already been established (step 203: YES), or the above If it is determined in step 203 that an abnormality other than the occurrence of a poorly energized phase has occurred (step 203: NO), the microcomputer 17 shifts to the assist stop mode (step 207). And the output of the motor control signal for stopping the drive of the motor 12, opening of the power relay, etc. are executed.

[Rotation angle correction control]
Next, a mode of rotation angle correction (offset) control during two-phase driving in this embodiment will be described.

  As shown in FIG. 3, in the present embodiment, the second current control unit 24b corresponding to the two-phase drive at the time of occurrence of a poorly energized phase has a phase current command value Ix * in the phase current feedback control at the time of the two-phase drive. Rotation angle correction control unit 40 for correcting the rotation angle (electrical angle) θ of the motor 12 input to the second current control unit 24b is provided to compensate for a phase shift between the phase current value Ix and the phase current value Ix. ing. Then, by executing the phase current feedback control based on the rotation angle θ ′ corrected by the rotation angle correction control unit 40, the motor rotation during the two-phase driving is smoothed, and there is no feeling of catching. The structure is designed to achieve a good steering feeling.

  That is, in a configuration in which current control is performed so as to energize a phase current that changes in a secant curve or a cosecant curve for each energized phase during two-phase driving as in the present embodiment, this is the energized phase. The sign of each phase current value is reversed across the rotation angles θA and θB corresponding to the asymptote (see FIG. 5). Therefore, when there is a phase shift between the phase current command value Ix * as the current command value and the phase current value Ix that is the actual current (see FIG. 9), the rotation angle θ of the motor 12 is asymptotically as described above. When passing through the predetermined rotation angles θA and θB corresponding to the line, there is a region in the vicinity of the predetermined rotation angles θA and θB where the sign of the current command value in the current control and the sign of the actual current value do not match. Arise. That is, there is a region where a current is generated that rotates the motor 12 in the reverse direction, and the smooth rotation of the motor 12 is prevented by the generation of the current in the reverse direction, so that the motor follows the steering operation. As a result, there is a risk that the driver may feel so-called catching.

  In consideration of this point, the microcomputer 17 (second current control unit 24b) of the present embodiment, as described above, between the phase current command value Ix * and the phase current value Ix in the phase current feedback control at the time of two-phase driving. In order to compensate for a phase shift between them, the rotation angle θ of the motor 12 which is the basis of the phase current feedback control is corrected (offset).

  Specifically, in the present embodiment, the rotation angle correction control unit 40 provided in the second current control unit 24 b corrects the rotation angle θ in the advance direction according to the rotation direction of the motor 12. Then, the phase current command value calculation unit 33 calculates the phase current command value Ix * based on the corrected rotation angle θ ′.

  In other words, in current control, the phase of the actual current supplied to the motor due to mechanical factors such as the motor's “rat”, delay in calculation time, or the presence of time delay elements such as phase delay in current control. However, there is a tendency to lag behind the phase of the current command value in the current control. Therefore, in consideration of the time delay element in the two-phase drive, the phase angle of the phase current value Ix with respect to the phase current command value Ix * is compensated by advancing the rotation angle θ in the advance direction according to the rotation direction. Is possible. In this embodiment, when the rotation angle θ of the motor 12 passes through the predetermined rotation angles θA and θB corresponding to the asymptote, the current command value generated in the vicinity of the predetermined rotation angles θA and θB. The region in which the sign and the sign of the actual current value do not match is reduced, so that the motor rotation is smoothed and the followability to the steering operation is improved.

  More specifically, as shown in FIG. 10, the rotation angle correction control unit 40 of this embodiment calculates a basic correction amount ε1 for correcting the rotation angle θ in the advance direction according to the rotation direction of the motor 12. And a rotational angular velocity correction amount computing unit 42 that computes a rotational angular velocity correction amount ε2 for correcting the rotational angle θ according to the rotational angular velocity ω of the motor 12.

  A steering torque τ is input to the basic correction amount calculation unit 41 of the present embodiment, and the basic correction amount calculation unit 41 uses a motor based on the input steering torque τ (sign thereof). 12 rotation directions are determined. Based on the determination result, a basic correction amount ε1 for correcting the rotation angle θ input to the rotation angle correction control unit 40 in the advance direction is calculated. In the present embodiment, the determination of the rotation direction of the motor 12 and the calculation of the basic correction amount ε1 in the basic correction amount calculation unit 41 are performed using a map 41a in which the steering torque τ and the basic correction amount ε1 are associated. Performed by calculation. In the present embodiment, in the map 41a, the basic correction amount ε1 is set to be a constant value corresponding to the sign of the steering torque τ.

  Further, the rotational angular velocity correction amount calculation unit 42 calculates a rotational angular velocity correction amount ε2 that corrects the rotational angle θ in the advance direction more as the detected rotational angular velocity ω of the motor 12 is faster (the absolute value is larger). To do. In this embodiment, the rotational angular velocity correction amount calculation unit 42 also has a map 42a in which the rotational angular velocity ω and the rotational angular velocity correction amount ε2 are associated. In the map 42a, the rotational angular velocity correction amount ε2 is The absolute value of the rotational angular velocity ω is set so as to increase as the absolute value thereof increases. Then, the rotational angular velocity correction amount calculation unit 42 calculates the rotational angular velocity correction amount ε2 by referring to the input rotational angular velocity ω in the map 42a.

  The basic correction amount ε1 calculated by the basic correction amount calculation unit 41 and the rotation angular velocity correction amount ε2 calculated by the rotation angular velocity correction amount calculation unit 42 are input to the adder 43 together with the rotation angle θ. Is added to the rotation angle θ input to the rotation angle correction control unit 40. That is, the rotation angle correction control unit 40 of the present embodiment corrects the input rotation angle θ in a predetermined angle advance direction according to the rotation direction of the motor 12, and the higher the rotation angular velocity ω of the motor 12, the more Largely correct in the direction of advance. Then, the corrected rotation angle θ ′ is output to the phase current command value calculation unit 33.

(Verification)
Next, the utility of the rotation angle correction control will be verified.
FIGS. 11 and 12 are both graphs showing the relationship between the steering torque τ and the rotational angular velocity ω of the motor during two-phase driving, that is, the followability of the motor to the steering operation. FIG. FIG. 12 is a graph when the rotation angle correction control is performed. In each of these drawings, the waveform L indicated by the broken line indicates the transition of the steering torque τ, and the waveform M indicated by the solid line indicates the transition of the rotational angular velocity ω of the motor.

  As shown in FIG. 11, when the rotation angle correction control is not performed, when the steering operation is performed relatively slowly to the left and right (section t2), and when the steering operation is performed quickly to the left and right (section t3), Both followability of the rotational angular velocity ω of the motor with respect to the transition of the steering torque τ is low. In particular, when the steering operation is performed slowly (section t1), only a very small rotational angular velocity ω is output even though the value of the steering torque τ has reached its detection limit (| τ0 |). Not. That is, the motor hardly rotates, and a so-called catch is generated.

  On the other hand, when the rotation angle correction control is performed, as shown in FIG. 12, when the steering operation is performed relatively slowly to the left and right (sections t5 and t7), the steering operation is quickly performed to the left and right. In both cases (sections t6 and t8), the followability of the rotational angular velocity ω of the motor with respect to the transition of the steering torque τ is high. In particular, even when the steering operation is performed slowly (section t4), it can be seen that the rotational angular velocity ω of the motor follows the steering operation.

  That is, when the rotation angle θ of the motor 12 passes through the predetermined rotation angles θA and θB corresponding to the asymptote by executing the rotation angle correction control, the current command value generated in the vicinity of the predetermined rotation angles θA and θB. The area where the sign of the current and the sign of the actual current value do not match can be reduced to suppress the generation of current that causes the motor 12 to rotate in the reverse direction. As a result, it is possible to infer that excellent followability to the steering operation as described above is realized by ensuring smooth motor rotation. Further, the graph of FIG. 12 does not reach the detection limit (| τ0 |) of the steering torque τ without increasing the rotational angular velocity ω (absolute value). Therefore, also from this point, it can be said that a good steering feeling without a feeling of being caught is realized.

As described above, according to the present embodiment, the following operations and effects can be obtained.
(1) When an energization failure occurs in any phase of the motor 12, the microcomputer 17 sets two energies other than the energization failure occurrence phase as energization phases to the energization failure occurrence phases. The current control is executed so as to generate a phase current that changes into a secant curve or a cosecant curve with the corresponding predetermined rotation angle as an asymptotic line, thereby continuing the output of the motor control signal. The microcomputer 17 includes a rotation angle correction control unit 40 that can correct (offset) the input rotation angle θ of the motor 12. During two-phase driving in which two phases other than the above-described energization failure phase are energized phases, between the phase current command value Ix * as the current command value in the current control and the phase current value Ix that is the actual current flow In order to compensate for the phase shift, the rotation angle θ which is the basis of the current control is corrected.

  According to the above configuration, the motor current corresponding to the required torque (q-axis current command value Iq *) excluding the predetermined rotation angles θA and θB corresponding to the asymptote (and the current limiting range set in the vicinity thereof). (Q-axis current value Iq) can be generated. As a result, even when a poorly energized phase occurs, it is possible to continue motor driving, that is, application of assist force, while ensuring high output performance without causing large torque ripple. Further, by compensating for the phase shift between the current command value and the phase current value during the two-phase drive, when the rotation angle θ passes through the predetermined rotation angles θA and θB corresponding to the asymptote, By reducing the region where the sign of the current command value and the sign of the actual current value generated in the vicinity of the predetermined rotation angles θA and θB do not match, it is possible to suppress the generation of current that causes the motor 12 to rotate in the reverse direction. it can. As a result, it is possible to achieve a good steering feeling that ensures smooth motor rotation, has excellent followability to steering operation, and has no feeling of catching.

(2) The rotation angle correction control unit 40 corrects the rotation angle θ in the advance direction according to the rotation direction of the motor 12.
In other words, in current control, the phase of the actual current supplied to the motor due to mechanical factors such as the motor's “rat”, delay in calculation time, or the presence of time delay elements such as phase delay in current control. However, there is a tendency to lag behind the phase of the current command value in the current control. Therefore, the phase delay of the actual current value with respect to the current command value can be compensated by considering the time delay element during the two-phase drive and by advancing the rotation angle θ in the advance direction according to the rotation direction.

(3) The rotation angle correction control unit 40 corrects the rotation angle θ in a more advanced direction as the rotation angular velocity ω of the motor 12 increases.
That is, the phase delay of the actual current value with respect to the current command value caused by the time delay element becomes more remarkable as the rotational angular velocity ω of the motor 12 is faster. Therefore, according to the above configuration, the phase delay of the actual current value with respect to the current command value can be compensated more accurately.

  (4) The microcomputer 17 includes a first current control unit 24a that calculates three-phase phase voltage command values Vu *, Vv *, and Vw * by executing current feedback control in the d / q coordinate system, and phase current feedback control. And a second current control unit 24b that calculates the phase voltage command values Vu **, Vv **, and Vw ** by execution. During normal operation, a motor control signal is output based on the phase voltage command values Vu *, Vv *, and Vw * calculated by the first current control unit 24a. The motor control signal is output based on the phase voltage command values Vu **, Vv **, and Vw ** calculated by the second current control unit 24b.

  In other words, one of the most influential time delay elements is a phase delay in current feedback control. In particular, phase current feedback control tends to have a larger phase delay than current feedback control in the d / q coordinate system. Therefore, when the two-phase driving is performed and the motor control signal is output by executing the phase current feedback control during the two-phase driving, the above-described configurations (1) to (3) can be applied to achieve more remarkable effects. Can be obtained.

(Utility during low-speed steering)
Next, the utility at the time of low speed steering will be described in detail with respect to the rotation angle correction control according to the configuration of the present embodiment.

  In FIG. 12, attention should be paid to the transition of the waveforms M and L in the section t4 corresponding to the case where the steering operation is performed slowly during the rotation angle correction control, that is, at the time of low speed steering. In this section t4, it can be seen that the waveforms M and L corresponding to changes in the steering torque τ and the rotational angular velocity ω vibrate little by little. However, this vibration is not caused by the steering operation of the tester (driver).

  That is, when the rotation angle correction according to the configuration of the present embodiment is not performed (see FIG. 11, section t1), the rotation angular velocity is obtained by performing the rotation angle correction even though the steering operation is performed. The waveform M indicating the change of oscillates around “0”. That is, a phenomenon occurs in which the rotation direction of the motor 12 is changed in small increments by executing the rotation angle correction control. At the time of low speed steering, the reversing operation in which the rotation direction is changed in small increments is a main mechanism for suppressing the occurrence of the catch.

  More specifically, as shown in FIG. 13, during the two-phase drive, by limiting the current in the vicinity of the predetermined rotation angles θA and θB, the torque in the steering direction near the predetermined rotation angles θA and θB. A rotation angle range in which (the sum of steering torque and assist torque) falls below the reaction torque (axial force) in the return direction, that is, a section where the steering speed is decelerated (deceleration section: θa <θ <θa ′, θb < θ <θb ′). And at the time of low speed steering, the existence of this deceleration section becomes a cause of the above-mentioned catch.

  That is, when the speed of entry into the deceleration zone is “ωin”, the escape speed is “ωout”, the motor inertia is “Jm”, and the deceleration energy in the deceleration zone is “−En”, Equation (4) is established.

Note that the “rush speed” in this case is, for example, the value of the rotational angular speed ω at the rotational angle θa when the steering direction is “left to right” in FIG. 13, and the “escape speed” is the same as the rotation speed. This is the value of the rotational angular velocity ω at the angle θa ′.

  Therefore, in order for the escape speed ωout to be greater than “0”, that is, to pass through the deceleration zone without stopping, the rush speed ωin must be faster than the critical speed ωcr expressed by the following equation (5). .

That is, as shown in FIG. 14, at the time of low speed steering (ω ≦ ωcr) where the rotational angular velocity ω is equal to or lower than the critical velocity ωcr, the deceleration zone cannot be passed. The rotational angular velocity ω is “0” at a rotational angle θp that is closer to the rotational angle θa ′ that is the escape position (the rotational angle θa side that is the entry angle). Here, in this deceleration zone, the torque in the return direction (reaction torque) is larger than the torque in the steering direction (sum of the steering torque and the assist torque) (“steering torque” + “assist torque” <“anti-reverse” Force torque (axial force) "). For this reason, the motor once stops at the same rotation angle θp and then rotates in the reverse direction. Finally, the motor stops at a rotation angle θa in which the torque in the steering direction and the torque in the return direction are balanced (when there is no special change in the steering torque), and as a result, it is shown in a section t1 in FIG. Thus, a so-called catching occurs in which the rotation of the motor does not follow the steering operation occurs.

  On the other hand, as shown in the section t4 in FIG. 12, when the rotation angle correction control is executed, the occurrence of the catch as shown in the section t1 in FIG. 11 is not observed. Instead, the reversing operation as described above in which the rotation direction of the motor is changed in small increments is recognized.

  FIG. 15 is a waveform diagram showing changes in motor rotation angle (electrical angle) and motor rotation angular velocity during rotation angle correction control. As shown in the figure, such a reversing operation of the motor occurs in the vicinity of the predetermined rotation angles θA and θB where the signs of the respective phase current values are reversed. It can be seen that the rotational angular velocity ω of the motor is increased by repeating the reversing operation.

  That is, by executing the rotation angle correction control according to the configuration of the present embodiment, at the time of low-speed steering (ω ≦ ωcr) where the rotation angular velocity ω is equal to or lower than the critical velocity ωcr, the motor performs the predetermined rotation angle θA, The inversion operation is repeated near θB. Then, by repeating the reversing operation, the rotational angular velocity ω is increased, and a rush velocity ωin faster than the critical velocity ωcr is obtained, so that the vehicle passes through the deceleration section that occurs in the vicinity of the predetermined rotational angles θA and θB, that is, is caught. It is the structure which suppresses generation | occurrence | production of.

  More specifically, as described above, the rotation angle correction control unit 40 of the present embodiment has a predetermined basic correction amount ε1 according to the steering torque τ (sign), that is, the direction of steering operation (steering direction), and Two correction amounts of the rotation angular velocity correction amount ε2 corresponding to the rotation angular velocity ω of the motor 12 are calculated (see FIG. 10). 16A and 16B show a case where the basic correction amount ε1 and the rotational angular velocity correction amount ε2 are superimposed on each other, that is, the relationship between the total of these respective correction amounts and the rotational angular velocity ω for each steering direction. It is shown in.

  As shown in FIGS. 16 (a) and 16 (b), the rotation angle correction control unit 40 of the present embodiment is in the advance direction during normal rotation of the motor 12, that is, in the normal state where the motor 12 rotates in the steering direction. However, in some cases, a correction amount that corrects the rotation angle θ in the delay direction may be calculated.

  Specifically, the value of the basic correction amount ε1 calculated by the basic correction amount calculation unit 41 is constant according to the steering direction (sign of the steering torque τ), but is calculated by the rotational angular velocity correction amount calculation unit 42. The value of the rotational angular velocity correction amount ε2 to be increased increases as the rotational angular velocity ω (absolute value) increases. Accordingly, in the region where the steering direction and the rotation direction of the motor 12 are opposite and the rotation angular velocity ω is small, the sign of the correction amount is different from the sign of the rotation angle θ, that is, the rotation angle θ is corrected in the delay direction. Such a correction amount is calculated. In this embodiment, the rotation angle correction in the delay direction realizes the repetition of the reversing operation in the deceleration zone in the vicinity of the predetermined rotation angles θA and θB and the increase in the rotation angular velocity ω.

  That is, by correcting the rotation angle θ in the delay direction, an actual current is generated in the vicinity of the predetermined rotation angles θA and θB so as to rotate the motor 12 in the direction opposite to the steering direction (see FIG. 9). ). In the present embodiment, when the rotation direction of the motor is reversed in the deceleration section in the vicinity of the predetermined rotation angles θA and θB, the assist force (in the opposite direction to the steering direction that assists the reverse rotation) ( (Reverse assist) is provided.

  Note that FIG. 17 shows the transition of the q-axis current command value Iq * as the current command value for rotating the motor in the steering direction and the q-axis current value Iq as the actual current value at the rotation angle θ (electrical angle). Although this is a waveform diagram shown for each, it can be confirmed in FIG. 17 that the reverse assist occurs in the vicinity of the predetermined rotation angles θA and θB.

  Then, while repeating the reversing operation as described above, the energy given as the reverse assist is converted into the rotational angular velocity ω, and the rush velocity ωin exceeding the critical velocity ωcr is obtained, thereby passing through the deceleration section. ing.

  That is, as described above, when the rush speed ωin (for example, the rotational angular speed ω at the rotational angle θa) to the deceleration section is equal to or lower than the critical speed ωcr, it is impossible to pass through the deceleration section. Then, after stopping at an intermediate rotation angle θp1, the motor rotates backward in the return direction (see FIG. 14).

  However, as shown in FIG. 18, by assisting the reverse rotation by applying the reverse assist force as described above, at the time of the reverse rotation, the entry position to the deceleration zone at the rotational angular speed ω faster than the entry speed ωin ( The rotation angle θa (θp0)) is passed. Then, after returning to the rotation angle θp2 positioned in the return direction larger than the entry position, it rotates again in the steering direction. At this time, the steering system is twisted due to the reverse rotation in the return direction, and the absolute value of the steering torque τ detected by the torque sensor 14 becomes large. Then, by accelerating again in the steering direction by a large assist force calculated based on the steering torque τ, the rush speed ωin is faster than the previous entry, that is, the rotation angle θp1 that is the maximum arrival point at the previous entry. The rush speed ωin that can reach the rotation angle θp3 located in the advance direction is acquired.

  When the re-entry speed ωin is less than or equal to the critical speed ωc, that is, when the rotation angle θp3 that is the maximum arrival point is within the deceleration zone, the reversing operation and acceleration by applying the reverse assist are repeated. Thus, the rush speed ωin faster than the critical speed ωcr is obtained, and the vehicle passes through the deceleration section.

In addition, you may change this embodiment as follows.
In the present embodiment, the present invention is embodied in an electric power steering device (EPS), but may be embodied in a motor control device used for applications other than EPS.

  In the present embodiment, the ECU 11 as the motor control device is roughly divided into three control modes: “normal control mode”, “assist stop mode”, and “two-phase drive mode”. However, the mode of motor control when an abnormality occurs is not limited to these modes. In other words, any configuration may be applied as long as the motor control is executed by using two phases other than the energization failure phase as the energization phase when the energization failure phase occurs. Also, the method of abnormality detection (determination) is not limited to the configuration of the present embodiment.

  In the present embodiment, the current command value calculation unit 23 outputs a phase current command value for one of the two phases other than the energization failure occurrence phase during two-phase driving, and the motor control signal generation unit 24 After calculating the phase voltage command value for the phase, the phase voltage command value for the other phase is calculated based on the calculated value. However, the present invention is not limited to this, and the current command value calculation unit 23 may output phase current command values for both two phases other than the energization failure occurrence phase.

  In this embodiment, based on the above equations (1) to (3), when the U phase or W phase is abnormal, the V phase phase current command value Iv * is calculated, and when the V phase is abnormal, U The phase current command value Iu * of the phase is calculated. However, not limited to this, the W-phase current command value (Iw *) is calculated when the U-phase or V-phase is abnormal, and the U-phase current command value (Iu *) is calculated when the W-phase is abnormal. It is good also as a structure of carrying out. In addition, each phase current command value in this case can be calculated by reversing the signs of the above formulas (1) to (3).

  Furthermore, the phase current command value at the time of occurrence of an energization failure does not necessarily have to be completely the same as that calculated by the above equations (1) to (3). That is, even if a phase current command value that changes to a nearly secant curve or a substantially cosecant curve with an asymptotic line as a predetermined rotation angle, or changes in approximation to this, an effect close to that of the present embodiment is obtained. Can be obtained. However, when the phase current command value is calculated based on the above equations (1) to (3), it is possible to generate a motor current closest to the required torque, and the phase current command calculated based on each equation Needless to say, a method that calculates a value close to a value provides a more remarkable effect.

  In the present embodiment, the basic correction amount ε1 based on the steering torque τ (sign), that is, the direction of steering operation (steering direction) is the case where the steering direction and the rotation direction of the motor 12 match, and the case where they do not match. In both cases, it was fixed (see FIGS. 10 and 16 (a) and 16 (b)). However, the configuration is not limited to this, and the basic correction amount ε1 may be changed depending on whether the steering direction and the motor rotation direction coincide with each other. For example, as shown in FIGS. 19A and 19B, when the steering direction and the motor rotation direction are different, the basic correction amount ε1 may be increased as compared with the case where they are the same.

  Further, in this embodiment, the rush speed ωin that can pass through the deceleration section in the vicinity of the predetermined rotation angles θA and θB in which the direction of each phase current switches, that is, the rotation angular speed ω is faster than the critical speed ωcr, When the motor rotation direction coincides, the rotation angle θ is corrected in the advance direction, thereby suppressing the occurrence of reverse assist and ensuring smooth rotation. When the rotational angular velocity ω is equal to or lower than the critical velocity ωcr and the steering direction and the motor rotational direction are different, the rotational angle θ is corrected in the delay direction, and the resulting reverse assist force is converted into the rotational angular velocity ω. Thus, the rush speed ωin is increased to a speed higher than the critical speed ωcr to suppress the occurrence of catching. However, the present invention is not limited to this, and as shown in FIGS. 20A and 20B, only the rotation angle correction in the advance direction when the steering direction coincides with the motor rotation direction is performed, or FIG. As shown in b), only the rotation angle correction in the delay direction when the steering direction and the motor rotation direction are different may be performed.

  In addition, as a form of current control, the phase current feedback control in the three-phase AC coordinates (U, V, W) as in the present embodiment is not necessarily required. For example, according to the following equations (6) to (8), a d-axis current command value Id * that changes in a tangent curve (tangent) shape with predetermined rotation angles θA and θB corresponding to the energization failure occurrence phase as asymptotic lines. Is calculated. Further, the present invention may be applied to a configuration in which a motor control signal is generated by executing a current feedback control in a d / q coordinate system based on the d-axis current command value Id * (see FIG. 22, FIG. Example). In addition, the present invention is not limited to feedback control, and may be applied to a device that outputs a motor control signal by executing open control.

  In the present embodiment, the rotation angle correction control unit 40 includes a basic correction amount calculation unit 41 that calculates a basic correction amount ε1 for correcting the rotation angle θ in the advance direction according to the rotation direction of the motor 12, and a motor. And a rotational angular velocity correction amount calculating unit 42 for calculating a rotational angular velocity correction amount ε2 for correcting the rotational angle θ according to twelve rotational angular velocities ω. However, the present invention is not limited to this, and a configuration in which only correction of the rotation angle θ according to the rotation direction of the motor 12 or correction of the rotation angle θ according to the rotation angular velocity ω may be implemented.

The schematic block diagram of an electric power steering device (EPS). The block diagram which shows the electric constitution of EPS. The control block diagram of a microcomputer (motor control signal generation part). The flowchart which shows the process sequence of an electricity supply failure phase detection. Explanatory drawing which shows transition of each phase electric current at the time of two-phase drive (at the time of U-phase electricity failure). Explanatory drawing which shows transition of the q-axis current at the time of two-phase drive (at the time of U-phase conduction failure). The flowchart which shows the process sequence of abnormality determination and control mode switching. The flowchart which shows the process sequence of the motor control signal generation at the time of two-phase drive. Explanatory drawing which shows the phase delay of the actual electric current value with respect to the electric current command value in electric current control. The control block diagram of a rotation angle correction control part. The graph which shows the relationship between the steering torque at the time of two-phase drive (without rotation angle correction | amendment), and the rotation angular velocity of a motor. The graph which shows the relationship between the steering torque at the time of two-phase drive (with rotation angle correction | amendment) and the rotation angular velocity of a motor. Explanatory drawing which shows the deceleration area of the predetermined rotation angle vicinity where the direction of each phase current switches. Explanatory drawing which shows the generation | occurrence | production mechanism of the catch at the time of low speed steering. The wave form diagram which shows transition of the motor rotation angle (electrical angle) and motor rotation angular velocity at the time of rotation angle correction control. (A) (b) The figure which showed the relationship between correction amount and rotation angular velocity for every steering direction. The wave form diagram which shows transition of the q-axis current command value as a current command value and the q-axis current value as an actual current value for each rotation angle (electrical angle). Explanatory drawing which shows the mechanism of the rotation angular velocity rise by repetition of reversing operation. (A) (b) The figure which showed the relationship between the correction amount and rotational angular velocity of another example for every steering direction. (A) (b) The figure which showed the relationship between the correction amount and rotational angular velocity of another example for every steering direction. (A) (b) The figure which showed the relationship between the correction amount and rotational angular velocity of another example for every steering direction. Explanatory drawing which shows transition of the d-axis current and q-axis current at the time of the two-phase drive (at the time of U-phase conduction failure) of another example. Explanatory drawing which shows the aspect of the two-phase drive which uses the two phases other than the conventional energization failure generation phase as an energization phase. Explanatory drawing which shows transition of the d-axis current and q-axis current at the time of the conventional two-phase drive.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 1 ... Electric power steering apparatus (EPS), 10 ... EPS actuator, 11 ... ECU, 12 ... Motor, 12u, 12v, 12w ... Motor coil, 17 ... Microcomputer, 18 ... Drive circuit, 23 ... Current command value calculating part, 24 ... motor control signal generation unit, 24a ... first current control unit, 24b ... second current control unit, 31 ... abnormality determination unit, 33 ... phase current command value calculation unit, 36 ... F / B control unit, 37 ... phase voltage Command value calculation unit, 40 ... rotation angle correction control unit, 41 ... basic correction amount calculation unit, 42 ... rotation angular velocity correction amount calculation unit, Ix, Iu, Iv, Iw ... phase current value, Ix *, Iu *, Iv * , Iw * ... phase current command value, Ix_max ... maximum value, Vx *, Vu *, Vv *, Vw *, Vu **, Vv **, Vw ** ... phase voltage command value, Id ... d-axis current value, Iq ... q-axis current value, Id * ... d-axis current command value, Iq * ... q-axis current command value, θ, θ ', θA, θB ... times Corner, ω ... rotation angular velocity.

Claims (6)

Motor control signal output means for outputting a motor control signal, a drive circuit for supplying three-phase drive power to the motor based on the motor control signal, and an abnormality detection capable of detecting an energization failure occurring in each phase of the motor The motor control signal output means generates the motor control signal by executing current control based on the rotation angle of the motor, and when the energization failure occurs, two phases other than the energization failure occurrence phase are generated. In the electric power steering apparatus that executes the output of the motor control signal as the energization phase,
At the time of occurrence of the energization failure, the motor control signal output means, for each of the energization phases, changes to a secant curve or a remainder curve with a predetermined rotation angle corresponding to the energization failure occurrence phase as an asymptotic line. An electric power steering characterized in that the current control is performed so as to energize current, the current is limited to limit the phase current within a predetermined range, and the rotation angle is corrected in the same direction as the steering direction. apparatus.   The electric power steering apparatus according to claim 1, wherein
  The motor control signal output means corrects the rotation angle in a delay direction when the steering direction and the rotation direction of the motor are different;
  An electric power steering device.   In the electric power steering according to claim 1 or 2,
  The motor control signal output means corrects the rotation angle in the advance direction when the steering direction and the rotation direction of the motor coincide;
  An electric power steering device. The electric power steering according to any one of claims 1 to 3,
The electric power steering apparatus according to claim 1, wherein the motor control signal output means corrects the rotation angle in a more advanced direction as the rotation angular velocity of the motor is faster. The electric power steering according to any one of claims 1 to 4,
The motor control signal output means, wherein said current control during the current-carrying failure occurrence, it is a current feedback control, the electric power steering apparatus according to claim. In the electric power steering device according to claim 5 ,
The motor control signal output means executes, as the current control, current feedback control in a d / q coordinate system when the energization failure does not occur, and phase current feedback control when the energization failure occurs. Electric power steering device.