JP5246028B2 - Microwave amplifier - Google Patents

Description

  The present invention relates to a microwave amplifier used in a microwave band including a millimeter wave band.

  FIG. 10 is an example of the microwave amplifier disclosed in Non-Patent Document 1. In FIG. 10, 1 is an FET (Field Effect Transistor), 2 is an input matching circuit, 3 is an output matching circuit, 4 and 21 are inductors, and 5 and 22 are capacitors. The inductor 4 and the capacitor 5 constitute an impedance transformation circuit 9.

Yasuyuki Ito and Naoki Takagi, “Basics and Applications of MMIC Technology,” Realize, May 31, 1996, p. 142

If the input side equivalent circuit of the FET is a circuit composed of LCR in order to simplify the discussion of the conventional configuration, the circuit on the input side in FIG. 10 can be represented by a circuit as shown in FIG. The FET input side equivalent circuit is composed of a gate-source capacitance Cgs, an internal resistance Rin, and a gate inductor Lg. When the input impedance of FET and Z _FET, Z _FET is converted into the impedance Z ₁ by the impedance transformer circuit 9 consisting of the inductor 4 and a capacitor 5. Next, Z ₁ is converted to Z ₂ by the inductor 21 and the capacitor 22 in the same manner as described above. Non-Patent Document 1 shows that wide-band characteristics can be obtained by changing impedance in stages using a multistage impedance transformation circuit.

  However, when the conventional configuration is to be realized on, for example, an MMIC (Monolithic Microwave Integrated Circuit), the following problem occurs. MIM (Metal-Insulator-Metal) capacitors are often used when creating capacitors on the MMIC. It is known that the capacitance values of MIM capacitors vary due to process variations in dielectric thickness.

FIG. 12 shows the calculation results of the reflection characteristics of the conventional circuit when the capacitance values of the capacitors 5 and 22 in FIG. 11 vary by ± 20%. The element values used for the calculation are shown in FIG.
As can be seen from FIG. 12, the reflection characteristics greatly change due to variations in the capacitance values of the capacitors (hereinafter referred to as C variations). For example, the fluctuation due to the C variation of the lower limit frequency at which the reflection characteristic becomes −10 dB is about 3.7 GHz in this calculation range, and the frequency is greatly changed. In addition, the design operation band of this conventional microwave amplifier is 30 to 40 GHz, and it is desirable that the reflection characteristic be about -10 dB or less in this frequency range. However, when C variation occurs in the illustrated range, a frequency at which the reflection characteristic becomes as large as −5 dB or more is generated. As described above, the conventional configuration has a simple configuration including an inductor and a capacitor, and can obtain a wide band characteristic.

  The present invention has been made in order to solve the above-described problem, and an object of the present invention is to reduce the frequency fluctuation of the circuit with respect to the C variation, improve the yield of the microwave amplifier with respect to the process variation, and reduce the cost.

A microwave amplifier according to the present invention includes:
An amplifying element and a matching circuit connected to the amplifying element, and performing an amplifying operation in a predetermined operating band corresponding to the matching circuit,
The matching circuit includes:
A first impedance transformation circuit including a capacitor connected to the amplifying element;
It is connected in series to the first impedance transformation circuit, and has an electrical length of (−90 ° + 180 ° × n) to (180 ° × n) (n is a positive integer) at the frequency of the predetermined operating band. A second impedance transformation circuit including a transmission line and a series resonance circuit including a capacitor connected in parallel to the other end of the transmission line and having a resonance frequency lower than the frequency of the predetermined operating band;
It is characterized by comprising.

  According to the present invention, there is an effect that it is possible to suppress deterioration of characteristics due to frequency fluctuations even when the capacitance value of the capacitor varies.

1 is a circuit configuration diagram showing a microwave amplifier according to a first embodiment of the present invention. 1 is a circuit diagram of an input side circuit of a microwave amplifier according to a first embodiment of the present invention. The figure which shows the input reflection characteristic with respect to C dispersion | variation of the microwave amplifier by Embodiment 1 of this invention Table showing element values used for characteristic calculation of microwave amplifier according to embodiment 1 of the present invention Configuration diagram showing a microwave amplifier according to a second embodiment of the present invention Configuration diagram showing a microwave amplifier according to a third embodiment of the present invention The figure which shows the input reflection characteristic with respect to C dispersion | variation of the microwave amplifier by Embodiment 3 of this invention Configuration diagram showing a microwave amplifier according to a fourth embodiment of the present invention Configuration diagram showing a microwave amplifier according to a fifth embodiment of the present invention Circuit diagram showing a conventional microwave amplifier Conventional microwave amplifier input side circuit configuration diagram The figure which shows the input reflection characteristic with respect to C dispersion | variation of the conventional microwave amplifier A table showing the element values used for calculating the characteristics of conventional microwave amplifiers

Embodiment 1 FIG.
1 is a circuit configuration diagram showing a microwave amplifier according to a first embodiment of the present invention. In FIG. 1, 1 is an FET (Field Effect Transistor) which is an amplifying element, 2 is an input matching circuit which is a matching circuit, 3 is an output matching circuit, 4 and 7 are inductors, 5 and 8 are capacitors, 6 Is a transmission line. A first impedance transformation circuit 9 is configured by connecting the inductor 4 and the capacitor 5 in an inverted L shape. A series resonant circuit 10 is constituted by the inductor 7 and the capacitor 8 and is connected between the transmission line 6 and the ground. The transmission line 6 and the series resonant circuit 10 constitute a second impedance transformation circuit. In this embodiment, the predetermined operating band is set from 30 GHz to 40 GHz, and the center frequency is 35 GHz. In FIG. 1, an FET 1 and an input matching circuit 2 are integrated as an MMIC (Monolithic Microwave Integrated Circuit). The capacitors 5 and 8 are made of MIM (Metal-Insulator-Metal) capacitors.

Next, the operation will be described. In order to simplify the discussion, it is assumed that the input side equivalent circuit of the FET 1 is a circuit made of LCR. The transmission line 6 is set so that the electrical length at the band center frequency is 90 ° or more and 180 ° or less. At this time, the transmission line 6 can be approximately divided as an equivalent circuit into a transmission line having an electrical length of 180 ° and a negative inductor. Therefore, FIG. 1 can be represented by the circuit shown in FIG. When the input impedance of the FET1 and Z _FET, Z _FET is converted by the impedance transformer circuit 9 to an impedance _{Z 2 (= 1 / Y 2} ), Z 2 is the impedance Z ₄ by transmission line 6 and the series resonance circuit 10 (= 1 / Y ₄ ) The series resonant circuit 10 has a resonance frequency set to a value lower than the band center frequency, and operates as a shunt inductor at the band center frequency. The transmission line 6 and the series resonant circuit 10 also operate as an impedance transformation circuit. As described above, the present embodiment uses a multi-stage impedance transformation circuit, and can obtain wideband characteristics as in the conventional configuration.

Next, frequency variation with respect to variation in the capacitance value of the capacitor (hereinafter referred to as C variation) will be described. As a representative value of the frequency in the predetermined operating band, the center frequency of the operating band is considered here, and the value of this angular frequency is ω ₀ . At this time, the admittance Y ₂ is expressed as follows using the admittance Y ₁ and the capacitor C ₁ .
Y ₂ (ω ₀ ) = Y ₁ (ω ₀ ) + jω ₀ C ₁ (1)
Y ₁ (ω ₀ ) = G (ω ₀ ) + jB (ω ₀ )
Y ₂ = G + j (B + ω ₀ C ₁ ) (2)
It becomes. Here, Y ₂ = Y ₂ (ω ₀ ), G = G (ω ₀ ), and B = B (ω ₀ ).

Next, the impedance Z ₃ (= 1 / Y ₃ ) is obtained. The transmission line 6 is set so that the electrical length at the band center frequency is 90 ° or more and 180 ° or less. This electrical length is a value when n is 1 in a value of (−90 ° + 180 ° × n) or more and (180 ° × n) or less (n is a positive integer). For this reason, the transmission line 6 can be equivalently divided into a transmission line having an electrical length of 180 ° and a virtual transmission line having an electrical length of −90 ° or more and 0 ° or less.

Even if a transmission line whose electrical length is n times 180 ° at ω ₀ is connected, the impedance remains unchanged, and this transmission line can be ignored. Further, a virtual transmission line having an electrical length of −90 ° or more and 0 ° or less can be considered by replacing it with a virtual negative inductor −L _Zc as shown in FIG.
L ≒ Z tanθ / ω ₀ (3)
Here, L is the equivalent inductance, Z is the characteristic impedance of the transmission line, and θ is the electrical length of the transmission line.
The reason why the virtual negative inductor can be derived in this way is that the electrical length of the transmission line 6 is set to be 90 ° or more and 180 ° or less.

となる。式（４）に式（２）を代入し、Y₃を求めると、

となる。 From the above, the required impedance Z ₃ is

It becomes. Substituting equation (2) into equation (4) to find Y ₃ ,

It becomes.

ここで、Y₃' はＣばらつきがないときのアドミッタンスである。また、Δ_Cbaraの変化に対するアドミッタンスの実数部の変化は小さいので、式（６）では省略している。 The C variation of C _1, represented by replacing C ₁ C ₁ and (1 + Δ _Cbara), into Equation (5), when then transforming this expression as an expression indicating a change to Δ _Cbara, Y ₃ is It can be approximately expressed by the following formula.

Here, Y ₃ ′ is an admittance when there is no C variation. Moreover, _since the change of the real part of the admittance with respect to the change of Δ _Cbara is small, it is omitted in the equation (6).

直列共振回路１０がシャントのインダクタとして動作するためにはY_Sの虚数部が、Im[Y_S]＜０である必要があるから、直列共振回路は以下の式を満たす必要がある。

すなわち、帯域中心周波数より直列共振回路１０の共振周波数が低いことが条件となる。ここでは、直列共振回路１０の共振周波数を帯域中心周波数より低い値に設定しているので、式（８）は満たされている。 Next, the admittance Y _S of the series resonant circuit 10 is considered as follows.

For series resonance circuit 10 operates as a shunt inductor imaginary part of Y _S is, Im [Y _S] it is necessary <0, the series resonant circuit should satisfy the following equation.

That is, the condition is that the resonance frequency of the series resonance circuit 10 is lower than the band center frequency. Here, since the resonance frequency of the series resonance circuit 10 is set to a value lower than the band center frequency, the equation (8) is satisfied.

ここで、Y_S' はＣばらつきがないときのアドミッタンスである。 Then the C dispersion of C ₂ in the same manner as described above, is substituted into equation (7) as _{C 2 = C 2 (1 +} Δ Cbara) Y S is approximately expressed by the following equation.

Here, Y _S 'is an admittance when there is no C variation.

よって、ＣばらつきΔ_Cbara に対するアドミッタンスの変動ΔY₄は、

である。 Since C ₁ and C ₂ are manufactured by the same process, the C variation ratio Δ _Cbara in the equations (6) and (9) can be set to the same value. Therefore, the admittance Y ₄ is expressed as follows from the equations (6) and (9).

Therefore, the admittance variation ΔY _{4 with} respect to the C variation Δ _Cbara is

It is.

Consider the values of the first and second terms in square brackets in equation (11).
Since the admittance Y ₂ has a value close to a real number by the impedance transformation circuit 9, the absolute value of the imaginary part B + ω ₀ C ₁ of Y ₂ is smaller than the real part G. In addition, in the first term in the square brackets in the formula (11), the constant 1 in the braces of the molecule has a small influence and can be ignored. Therefore, considering the leading negative sign, the first term in the square brackets in equation (11) is a negative value as a whole.

In addition, the second term in the square brackets in equation (11) is clearly a positive value. Therefore, in the equation (11), it can be seen that the first term and the second term in the brackets cancel each other, and the coefficient of Δ _Cbara becomes small. For this reason, in the present embodiment, the fluctuation of the admittance Y ₄ due to the C variation can be suppressed small.

  In the above description, the operation has been described with respect to the center frequency of the operation band. However, the same operation and effect can be obtained even when an arbitrary frequency within the operation band is considered.

In order to minimize the influence of frequency fluctuation due to C variation, it is ideal to satisfy the following conditions.

  Hereinafter, a specific calculation example using the matching circuit will be shown to clarify the effect of the present embodiment. FIG. 3 shows the calculation result of the input reflection characteristic when the capacitance value of the capacitor varies ± 20%. The element values used for the calculation are shown in FIG. In FIG. 3, for example, the fluctuation due to the C variation in the lower limit frequency at which the reflection characteristic becomes −10 dB is suppressed to about 0.3 GHz in this calculation range. This is a very small value with respect to the value of about 3.7 GHz in FIG.

  In FIG. 3, even if there is C variation in the illustrated range, the reflection characteristic is −10 dB or less from 30 GHz to 40 GHz which is a predetermined operating band, and good characteristics are obtained.

  As described above, the first embodiment of the present invention has an effect that the frequency fluctuation due to the C variation can be reduced. In this way, by suppressing the deterioration of the RF characteristics with respect to C variation, the yield of the microwave amplifier can be improved and the cost can be reduced.

  In order to realize the present embodiment with an actual circuit, the lumped constant C may be replaced with an MIM capacitor or the like, and the lumped constant L may be replaced with a chip inductor, a transmission line, or the like. The transmission line can be realized by a microstrip line or a coplanar line. All circuit elements and amplifying elements can be manufactured by integrating and integrating with MMIC, or can be manufactured by dividing into several parts.

  In this embodiment, the electrical length of the transmission line 6 is set to be 90 ° or more and 180 ° or less, but is not limited thereto, and is not less than 270 ° or more and 360 ° or less (−90 ° + 180 ° × n) or more. The same effect can be obtained when the electrical length is (180 ° × n) or less (n is a positive integer).

Embodiment 2. FIG.
5 is a circuit configuration diagram showing a microwave amplifier according to a second embodiment of the present invention. In FIG. 5, 1 is an FET, which is an amplifying element, 2a is an input matching circuit, 3 is an output matching circuit, 4, 6-L, 7 are inductors, and 5, 6-C, 8 are capacitors. An impedance transformation circuit 9 is configured by connecting the inductor 4 and the capacitor 5 in an inverted L shape. The inductor 6-L and the capacitor 6-C constitute a first series resonance circuit, and the inductor 7 and the capacitor 8 constitute a second series resonance circuit 10. The resonance frequency of the first series resonance circuit is set higher than the center frequency of the operation band of the microwave amplifier, and the resonance frequency of the second series resonance circuit is set lower than the center frequency of the operation band of the microwave amplifier. Has been. Capacitors 5, 6-C, 8 are made of MIM capacitors.

  Next, the operation will be described. In FIG. 5, the resonance frequency of the first series resonance circuit is set higher than the center frequency of the operation band of the microwave amplifier, and the resonance frequency of the second series resonance circuit is the center frequency of the operation band of the microwave amplifier. Is set lower. Therefore, at the center frequency, the first and second series resonant circuits have impedances equivalent to the capacitor and the inductor, respectively. That is, the inverted L-shaped circuit including the first series resonant circuit and the second series resonant circuit 10 also operates as an impedance transformation circuit.

  Therefore, the input matching circuit 2 in FIG. 5 together with the impedance transformation circuit 9 composed of the inductor 4 and the capacitor 5 is a circuit in which the impedance transformation circuits are cascaded in multiple stages, and a wide band characteristic can be obtained.

  Now, it is known that a transmission line having an electrical length of 180 ° at a certain frequency can be approximated by a series resonance circuit of a capacitor and an inductor having the frequency as a resonance frequency. If a negative inductor is further connected in series to this series resonance circuit, the inductance of the inductor cancels and decreases, resulting in a series resonance circuit with a higher resonance frequency.

  The first series resonance circuit in FIG. 5 has a resonance frequency higher than the center frequency, and is approximately equivalent to a circuit in which a transmission line having an electrical length of 180 ° and a negative inductor are connected in series. That is, FIG. 5 is equivalent to the circuit of the first embodiment shown in FIG. Therefore, FIG. 5 also has an effect that the frequency fluctuation due to C variation can be reduced as in the first embodiment.

  In FIG. 5, the capacitor 6 -C used in the first series resonance circuit can be configured by a capacitor other than the MIM capacitor that is not easily affected by process manufacturing variations.

Embodiment 3 FIG.
FIG. 6 is a circuit diagram showing a microwave amplifier according to the third embodiment of the present invention. In FIG. 6, 1 is an FET, 2b is an input matching circuit, 3 is an output matching circuit, 7 and 12 are inductors, 8 and 11 are capacitors, 6 is a transmission line, and the capacitors 11 and 12 are connected in an inverted L shape. Thus, the impedance transformation circuit 9 is configured. A series resonant circuit 10 is constituted by the inductor 7 and the capacitor 8 and is connected between the transmission line 6 and the ground. The capacitors 8 and 11 are made of MIM capacitors.

  In FIG. 6, the inverted L-type circuit of the impedance transformation circuit 9 has a configuration in which a capacitor 11 is connected to a series element and an inductor 12 is connected to a parallel element. Further, the resonance frequency of the series resonance circuit 10 is set lower than the band center frequency so that the series resonance circuit 10 operates as an inductor at the band center frequency. Also in this case, since the impedance transformation circuit 9 including the capacitor 11 and the inductor 12 and the impedance transformation circuit including the transmission line 6 and the series resonance circuit 10 are connected in multiple stages, wideband characteristics can be obtained.

  FIG. 7 shows the calculation result of the input reflection characteristic when the capacitance value of the capacitor varies ± 20% in the circuit of FIG. In FIG. 7, for example, the fluctuation due to C variation of the lower limit frequency at which the reflection characteristic becomes −10 dB is about 1.7 GHz in the range of this calculation. Therefore, even in this case, it can be suppressed to a small value with respect to the value in FIG.

  In addition, even if there is C variation in the range shown in the specified operating band of 30 GHz to 40 GHz, the reflection characteristics are suppressed to -8 dB or less, and although it is larger than -10 dB, almost good characteristics are obtained. It has been.

  As described above, the configuration as described above has an effect that the frequency fluctuation due to C variation can be reduced as in the first embodiment.

  Furthermore, in FIG. 6, the impedance transformation circuit 9 uses a C-L high-pass circuit configuration. For this reason, unlike FIG. 1, the inductor 12 which comprises the impedance transformation circuit 9 can be shunt-connected with respect to a signal track | line. In order to manufacture the inductor 12, it is generally necessary to use a long line. However, since the inductor 12 is shunt-connected to the signal line, it is perpendicular to the signal line having a relatively large space in terms of chip arrangement. Can be stretched. For this reason, it can be formed without increasing the length in the signal line direction, and the chip can be miniaturized as a whole.

Embodiment 4 FIG.
FIG. 8 is a circuit configuration diagram showing a microwave amplifier according to the fourth embodiment of the present invention. In FIG. 8, 1 is an FET, 2 and 13 are input matching circuits, 3 is an output matching circuit, 4, 7, 14, and 17 are inductors, 5, 8, 15, and 18 are capacitors, and 6 and 16 are transmission lines. The inductor 4 and the capacitor 5 constitute an impedance transformation circuit 9, the inductor 14 and the capacitor 15 constitute an impedance transformation circuit 19, the inductor 7 and the capacitor 8 constitute a series resonance circuit 10, and the inductor 17 and the capacitor 18 constitute a series resonance circuit 20. It is configured. Capacitors 5, 8, 15 and 18 are made of MIM capacitors.

  In FIG. 8, the input matching circuit 2 and the input matching circuit 13 are connected in cascade. Each of the input matching circuits 2 and 13 has the same configuration as that of the first embodiment. Therefore, there is an effect that frequency fluctuation due to C variation can be reduced.

  Furthermore, the number of matching circuits is increased and the number of stages is increased, so that the impedance transformation circuit can be connected in more stages and the bandwidth can be further increased.

  In the present embodiment, the input matching circuit 2 and the input matching circuit 13 may have any configuration from among the input matching circuits having the configurations shown in the first to third embodiments, for example. Alternatively, for example, the input matching circuits shown in the first to third embodiments may be connected using circuits having different circuit configurations. Even when the same circuit configuration is used, the element values of the corresponding elements may be different.

Embodiment 5 FIG.
FIG. 9 is a circuit configuration diagram showing a microwave amplifier according to the fifth embodiment of the present invention. In the figure, symbols having symmetry are generally added with symbols like 1a and 1b. Both 1a and 1b are collectively referred to as 1 as appropriate. In addition, for a line that shows the same operation even if it is divided like a transmission line, generally, the divided lines are added with symbols like 6-1 and 6-2, and both are collectively referred to as 6 as appropriate. Call. In FIG. 9, 1 is an FET, 2c is an input matching circuit, 3 is an output matching circuit, 4, 7 are inductors, 5 and 8 are capacitors, 6 is a transmission line, and an impedance transformation circuit 9 is formed from the inductor 4 and the capacitor 5. The series resonance circuit 10 is configured by the inductor 7 and the capacitor 8. Capacitors 5 and 8 are made of MIM capacitors.

  In FIG. 9, the transmission line 6 having an electrical length of 90 ° or more and 180 ° or less at the band center frequency is divided into two transmission lines 6-1 and 6-2 with the total electrical length being equal. Further, the transmission line 6-2 is further divided into two transmission lines 6-2a having the same electrical length and twice the characteristic impedance, and the transmission line 6-2b. The two FETs 1a and 1b and the two impedance transformation circuits 9a and 9b are combined in parallel by using the connection point of the transmission line 6-2a, the transmission line 6-2b, and the transmission line 6-1 as a branch point. doing. Other configurations are the same as those in the first embodiment.

  The configuration as described above also has the effect of reducing the frequency fluctuation due to C variation as in the first embodiment. Furthermore, there is an effect that the output can be increased by synthesizing FET1 in parallel as in the present embodiment.

  9 shows a configuration in which a branch point is provided in the middle of the transmission line 6 and two FETs 1a and 1b are combined in parallel. However, in this embodiment, the branch point is not limited to that of the FET 1 and the impedance transformation circuit 9. It may be provided between them, or may be provided before the input side of the series resonant circuit 10. Moreover, you may provide anywhere. Further, three or more FETs 1 that are amplification elements may be synthesized in parallel. A configuration in which a plurality of matching circuits as shown in the fourth embodiment are connected in cascade can also be applied to this embodiment.

  In the first to fourth embodiments described above, the present invention has been described by taking the input matching circuit as an example. However, the present invention is not limited to this, and can be applied to an output matching circuit, and may be applied to both an input matching circuit and an output matching circuit. When applied to an output matching circuit, the matching circuits shown in the first to fourth embodiments are symmetrical, and are connected in order from the output side of the FET 1 to the series resonant circuit 10 such as the impedance transformation circuit 9 and the transmission line 6. do it. When the present invention is used in either an input matching circuit or an output matching circuit, the other matching circuit is not necessarily required. Further, the amplifying element is not limited to the FET 1, and any other type of transistor or any amplifying element having an amplifying function may be used.

1, 1a, 1b FET
2, 2a, 2b, 2c, 13 Input matching circuit 3 Output matching circuit 4, 4a, 4b, 6-L, 7, 12, 14, 17, 21 Inductors 5, 5a, 5b, 6-C, 8, 11, 15, 18, 22 Capacitors 6, 6-1a, 6-1b, 6-2, 16 Transmission lines 9, 9a, 9b, 19 Impedance transformation circuit 10, 20 Series resonance circuit

Claims (8)

A microwave amplifier comprising an amplifying element and a matching circuit connected to the amplifying element, and performing an amplifying operation in a predetermined operating band corresponding to the matching circuit,
The matching circuit includes:
A first impedance transformation circuit including a capacitor connected to the amplifying element;
It is connected in series to the first impedance transformation circuit, and has an electrical length of (−90 ° + 180 ° × n) to (180 ° × n) (n is a positive integer) at the frequency of the predetermined operating band. A second impedance transformation circuit including a transmission line and a series resonance circuit including a capacitor connected in parallel to the other end of the transmission line and having a resonance frequency lower than the frequency of the predetermined operating band;
A microwave amplifier comprising: A microwave amplifier comprising an amplifying element and a matching circuit connected to the amplifying element, and performing an amplifying operation in a predetermined operating band corresponding to the matching circuit,
The matching circuit includes:
A first impedance transformation circuit including a capacitor connected to the amplifying element;
A first series resonance circuit connected in series to the first impedance transformation circuit and having a resonance frequency higher than the frequency of the predetermined operating band, and a capacitor connected in parallel to the other end of the first series resonance circuit A second impedance transformation circuit having a second series resonance circuit having a resonance frequency lower than the frequency of the predetermined operating band,
A microwave amplifier comprising:   3. The microwave amplifier according to claim 1, wherein the first impedance transformation circuit includes an inductor connected in series to the amplification element and a capacitor connected in parallel to the inductor. 4.   3. The microwave amplifier according to claim 1, wherein the first impedance transformation circuit includes a capacitor connected in series to the amplifying element and an inductor connected in parallel to the capacitor.   5. The microwave amplifier according to claim 1, wherein a plurality of the matching circuits are connected in cascade.   6. The microwave amplifier according to claim 1, wherein a plurality of the amplifying elements are combined in parallel.   The microwave amplifier according to claim 1, wherein the matching circuit is connected to an input unit and / or an output unit of the amplification element.   The microwave amplifier according to any one of claims 1 to 7, wherein the previous frequency of the predetermined operation band is a center frequency of the predetermined operation band.

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