JP4871334B2 - OFDM signal combining receiver - Google Patents

Description

  The present invention relates to a receiver for OFDM signal synthesis for digital broadcasting or digital transmission using the OFDM system, and in particular, for countermeasures against fading and interference waves that are problematic when receiving radio waves in digital broadcasting or wireless LAN, etc. The present invention relates to a receiving apparatus for combining OFDM signals to which an adaptive array antenna technique or a diversity receiving technique is applied.

  As an adaptive array technology for OFDM signals, for example, those described in Patent Documents 1 and 2 are known, and these are all intended to be applied to a broadcast wave relay device. Since an interference canceling apparatus for relaying broadcast waves using these techniques is a transmission-side facility, it is required to obtain a highly accurate interference canceling characteristic rather than performing processing with a low calculation amount. However, this interference canceling apparatus for relaying broadcast waves is not assumed to be used in an extremely poor reception environment. In particular, the service area of the SFN (Single Frequency Network) environment is an environment that is affected by multipath waves that have a high level and are within the GI (Guard Interval) but have a long delay time. There is a problem that there are things that cannot be obtained. Another problem is that the number of weighting factors to be optimized is the product of the number of subcarriers and the number of branches, which increases the amount of calculation.

  On the other hand, assuming that the above-described adaptive array technology for OFDM signals is applied to a receiver, in order to reduce the cost, the receiver has a simpler configuration and a smaller amount of calculation with the required bits. It is required that an error rate (BER: Bit Error Rate) be obtained, and that an operation can be performed even in a worse reception environment.

  In a broadcast wave relay device, the minimum requirement is that a required BER is obtained after interference removal, and it is required how to suppress the deterioration of transmission characteristics due to interference. On the other hand, the receiving device in the service area only needs to obtain a required BER after interference cancellation. As described above, the requirements for the OFDM signal adaptive array technology are greatly different between the broadcast wave relay device and the reception device in the service area.

  On the other hand, Non-Patent Documents 1 and 2 describe a pre-FFT type adaptive array technology for OFDM signals. The technique of Non-Patent Document 1 assumes mobile reception, and can suppress all incoming waves other than the desired wave. The technique of Non-Patent Document 2 assumes fixed reception, and can suppress only multipath waves whose delay time exceeds GI. All of these can be processed with a small amount of calculation.

Japanese Patent No. 3759448 JP 2005-295506 A Satoshi Hori, Nobuyoshi Kikuma, Naoki Inagaki, “MMSE Adaptive Array Using Guard Interval in OFDM”, Theory of Science, J85-B (9): 1608-1615, Sep 2002 Satoshi Hori, Nobuyoshi Kikuma, Naoki Inagaki, “MMSE Adaptive Array for Fixed Reception for Suppressing Only Incoming Waves Exceeding Guard Interval”, IEICE, J86-B (9): 1934-1940, 2003

  However, the techniques described in Non-Patent Documents 1 and 2 both use the feature of the OFDM signal that the GI is the same as the rear part of the effective symbol. For this reason, when the interference wave is the same system as the desired wave and the GI ratio is the same, there is a problem that the interference wave may not be suppressed. For example, when the symbol synchronization positions of the desired wave and the interference wave coincide with each other, there is a problem that the desired wave and the interference wave cannot be distinguished, and the interference wave cannot be suppressed.

  The present invention has been made to solve such a problem, and an object of the present invention is to be able to perform processing with a low calculation amount and to effectively suppress interference waves even in a poor reception environment. An object of the present invention is to provide a signal synthesis receiver.

In order to achieve the above object, an invention according to claim 1 is an OFDM signal combining receiver for receiving an OFDM wave by an array antenna including a plurality of array elements and outputting a bit string, the received OFDM wave An equivalent baseband signal by weighting in the time domain to generate an array synthesized signal, an FFT unit for converting the array synthesized signal into a carrier symbol which is a frequency domain signal, and a pilot signal A channel estimation unit that estimates a channel response, a channel equalization unit that performs channel equalization by dividing the carrier symbol by the channel response, and a weighting factor that is used for array synthesis by weighting in the time domain is controlled. A weighting factor control unit, and the weighting factor control unit includes the channel etc. A symbol reproducing unit that reproduces a symbol by demapping and remapping a subsequent carrier symbol to generate a reproduced carrier symbol, a pilot signal inserting unit that inserts a pilot signal into the reproduced carrier symbol, and the pilot signal are inserted An IFFT unit that converts the regenerated carrier symbol into a time domain signal, the time domain signal as a reference signal, and an optimization so that an error between the array composite signal and the reference signal is minimized, and the weight A weighting factor calculation unit for obtaining a coefficient.

The invention of claim 2 is the receiving device for OFDM signal combining according to claim 1, wherein the weighting factor controller, the reproduction carrier symbol pilot signal is inserted by the pilot signal inserter, the channel response The IFFT unit converts the multiplication result into a time domain signal.

Further, the invention of claim 3 is the OFDM signal combining receiver according to claim 1 , wherein the weighting factor control unit calculates a modulation error ratio based on the carrier symbol after the channel equalization and the reproduced carrier symbol. A modulation error ratio calculation unit for calculating, and the pilot signal insertion unit, based on the modulation error ratio, a predetermined pilot signal or a new pilot signal obtained by multiplying the pilot signal by a predetermined constant, It is inserted into the reproduction carrier symbol.

According to a fourth aspect of the present invention, in the OFDM signal synthesizing receiver according to the third aspect , the pilot signal insertion unit has the modulation error ratio below a predetermined threshold value, and the carrier symbol. A new pilot signal obtained by multiplying a predetermined pilot signal by a predetermined constant is inserted into the regenerated carrier symbol when the symbol number of is a multiple of a predetermined symbol interval. And

  As described above, according to the present invention, the array combining unit performs weighted combining in the time domain to generate an array combined signal, and the channel equalizing unit generates carrier symbols after channel equalization in the frequency domain. The weighting factor control unit uses a time domain signal obtained by IFFT of a predetermined pilot signal or regenerated carrier symbol as a reference signal, and sets a weighting factor used for array synthesis between the array synthesis signal and the reference signal. So that the error is minimized. As a result, the number of weighting factors to be optimized is only the number of branches to be array-synthesized, so that the amount of calculation is lower than that of a conventional apparatus that requires the number of products of the number of subcarriers and the number of branches. Further, since the channel estimation unit directly estimates the channel response using the pilot signal, the amount of calculation is similarly reduced. In addition, since array synthesis and channel equalization can be performed with such a low calculation amount, it is possible to effectively suppress interference waves even in a poor reception environment.

The best mode for carrying out the present invention will be described below with reference to the drawings.
[Example 1]
First, Example 1 will be described. FIG. 1 is a block diagram illustrating the configuration of the OFDM signal combining receiver according to the first embodiment. This OFDM signal synthesis receiver 1 includes a frequency conversion unit 11, an A / D conversion unit 12, an orthogonal demodulation unit 13, an array synthesis unit 14, a GI removal unit 15, an FFT unit 16, a channel estimation unit 17, a channel equalization unit. 18, a demapping unit 19, a parallel-serial conversion unit 20, and a weight coefficient control unit 21. Further, the array synthesis unit 14 includes a multiplication unit 141, an addition unit 142, and a complex conjugate unit 143 corresponding to the number of array elements. The weight coefficient control unit 21 includes GI removal units 214, weight coefficient calculation units 30, pilot signal generation units 211, multiplication units 212, and IFFT units 213 corresponding to the number of array elements.

  Frequency converters 11 corresponding to the number of array elements frequency-convert OFDM signals received via antenna 10 into IF signals. The IF signals output from the frequency converters 11 corresponding to the number of array elements are respectively input to the A / D converter 12. The A / D converters 12 corresponding to the number of array elements A / D convert the IF signals input from the frequency converter 11 into digital IF signals. The digital IF signals output from the A / D converters 12 corresponding to the number of array elements are respectively input to the orthogonal demodulator 13. The number of orthogonal demodulation units 13 corresponding to the number of array elements orthogonally demodulates the digital IF signal input from the A / D conversion unit 12 to generate an equivalent baseband signal. The equivalent baseband signals output from the orthogonal demodulation units 13 corresponding to the number of array elements are divided into two, one being input to the multiplication unit 141 of the array synthesis unit 14 and the other being input to the GI removal unit 214 of the weight coefficient control unit 21.

  The multiplier 141 corresponding to the number of array elements in the array synthesizer 14 multiplies the equivalent baseband signal input from the orthogonal demodulator 13 by the complex conjugate value of the weight coefficient input from the complex conjugate unit 143. The equivalent baseband signal multiplied by the complex conjugate value of the weighting coefficient output from the multiplier 141 corresponding to the number of array elements is input to the adder 142. Adder 142 adds the equivalent baseband signal multiplied by the complex conjugate value of the weighting coefficient output from multiplier 141 for the number of array elements, and generates an array composite signal. The array combined signal output from the adding unit 142 is input to the GI removing unit 15.

  The GI removal unit 15 removes the GI from the array composite signal input from the addition unit 142, and extracts a signal having a time corresponding to the effective symbol period. The array combined signal in the effective symbol period output from the GI removal unit 15 is input to the FFT unit 16.

  The FFT unit 16 converts the array composite signal in the effective symbol period output from the GI removal unit 15 into a carrier symbol which is a frequency domain signal by FFT. The carrier symbols output from the FFT unit 16 are divided into two, one being input to the channel equalization unit 18 and the other being input to the channel estimation unit 17.

  The channel estimation unit 17 estimates a channel response from the carrier symbol output from the FFT unit 16. The channel response output from the channel estimation unit 17 is divided into two, one being input to the channel equalization unit 18 and the other being input to the multiplication unit 212 of the weight coefficient control unit 21.

  The channel equalization unit 18 performs channel equalization by dividing the carrier symbol input from the FFT unit 16 by the channel response input from the channel estimation unit 17. The carrier symbol after channel equalization output from the channel equalization unit 18 is input to the demapping unit 19.

  The demapping unit 19 converts the carrier symbol after channel equalization output from the channel equalization unit 18 into a parallel signal by demapping. The parallel signal output from the demapping unit 19 is input to the parallel / serial conversion unit 20. The parallel-serial conversion unit 20 converts the parallel signal input from the demapping unit 19 into a serial signal, and outputs a bit string.

  On the other hand, the pilot signal generation unit 211 of the weight coefficient control unit 21 generates a predetermined pilot signal. The pilot signal output from pilot signal generator 211 is input to multiplier 212.

  Multiplier 212 multiplies the pilot signal input from pilot signal generator 211 by the channel response input from channel estimator 17. The pilot signal output by the multiplier 212 and multiplied by the channel response is input to the IFFT unit 213.

  IFFT section 213 converts the pilot signal output by multiplication section 212 and multiplied by the channel response into a time domain signal, and generates a time domain signal including only the pilot signal. The time domain signal including only the pilot signal output from IFFT section 213 is input to weighting coefficient calculating section 30. Here, the time domain signal including only the pilot signal output from the IFFT unit 213 is treated as a reference signal by the weighting factor calculation unit 30.

  The GI removal units 214 corresponding to the number of array elements remove GIs from the equivalent baseband signals input from the respective quadrature demodulation units 13 and extract an equivalent baseband signal having a time corresponding to an effective symbol period. The equivalent baseband signals in the effective symbol period output from the GI removal units 214 corresponding to the number of array elements are input to the weighting factor calculation unit 30.

  The weighting factor calculation unit 30 includes only the equivalent baseband signal in the effective symbol period input from the GI removal units 214 for the number of array elements and the pilot signal input from the IFFT unit 213 and multiplied by the channel response. A weighting factor is calculated using a reference signal that is a time domain signal. Specifically, the weight coefficient calculation unit 30 minimizes the square error (MMSE) so that the square error between the array synthesized signal obtained by weighting the input equivalent baseband signal and the input reference signal is minimized. ) To calculate the weight coefficient. The weighting factor output from the weighting factor calculating unit 30 is input to the complex conjugate unit 143 of the array combining unit 14.

  The complex conjugate unit 143 of the array synthesis unit 14 calculates the complex conjugate value of the weighting factor input from the weighting factor calculation unit 30. The complex conjugate value of the weighting coefficient output from the array synthesis unit 14 is input to the multiplication units 141 corresponding to the number of array elements.

  As described above, according to the OFDM signal combining receiver 1 of the first embodiment, the array combining unit 14 performs combining using the weighting coefficient in the time domain to generate an array combined signal, and the channel estimation unit 17 The channel response is estimated using the pilot signal, the channel equalization unit 18 channel equalizes the carrier symbol in the frequency domain, and the weight coefficient control unit 21 multiplies the predetermined pilot signal by the channel response, and this multiplication A time domain signal obtained by IFFT of the resulting signal is used as a reference signal, and a weighting factor used for array synthesis is obtained by optimization so that an error between the array synthesis signal and the reference signal is minimized. did. As a result, the number of weighting factors to be optimized is the number of branches for array synthesis. In the conventional apparatus, the number of weight coefficients to be optimized requires the number of products of the number of subcarriers and the number of branches. However, the OFDM signal combining receiver 1 according to the first embodiment requires a smaller amount of calculation than the conventional apparatus. . Further, since the channel estimation unit 17 directly estimates the channel response, the amount of calculation is similarly reduced. In addition, since array synthesis and channel equalization are performed with such a low calculation amount, it is possible to effectively suppress interference waves even in a poor reception environment.

[Example 2]
Next, Example 2 will be described. FIG. 2 is a block diagram illustrating a configuration of the OFDM signal combining receiver according to the second embodiment. This OFDM signal synthesis receiver 2 includes a frequency conversion unit 11, an A / D conversion unit 12, an orthogonal demodulation unit 13, an array synthesis unit 14, a GI removal unit 15, an FFT unit 16, a channel estimation unit 17, a channel equalization unit. 18, a symbol reproduction unit 23, a parallel / serial conversion unit 20, and a weight coefficient control unit 22. Further, the array synthesis unit 14 includes a multiplication unit 141, an addition unit 142, and a complex conjugate unit 143 corresponding to the number of array elements. The symbol reproduction unit 23 includes a demapping unit 19 and a remapping unit 221. The weighting factor control unit 22 includes a delay unit 226 for the number of array elements, a GI removal unit 227 for the number of array elements, a weighting factor calculation unit 228, a remapping unit 221, and a modulation error ratio (MER) modulation unit (MER). , A pilot signal insertion unit 223, a multiplication unit 224, and an IFFT unit 225 are provided.

  Comparing the OFDM signal combining receiver 1 of the first embodiment shown in FIG. 1 and the OFDM signal combining receiver 2 of the second embodiment shown in FIG. 2, both apparatuses have frequency converters 11 corresponding to the number of array elements. , An A / D converter 12 and an orthogonal demodulator 13, and an array synthesizer 14, a GI remover 15, an FFT unit 16, a channel estimator 17, a channel equalizer 18, a demapping unit 19, and a parallel serial conversion It is the same in that the unit 20 is provided. In both apparatuses, the weight coefficient control units 21 and 22 include GI removal units 214 and 227 corresponding to the number of array elements, multiplication units 212 and 224, IFFT units 213 and 225, and weight coefficient calculation units 30 and 228, respectively. Are the same.

  On the other hand, the OFDM signal combining receiver 1 includes the pilot signal generator 211 in the weighting factor control unit 21, whereas the OFDM signal combining receiver 2 in the symbol reproduction unit 23 and the weighting factor control unit 22. , Except that a remapping unit 221, a modulation error ratio calculation unit 222, and a pilot signal insertion unit 223 are provided. The receiving apparatus 2 for OFDM signal synthesis is different in that the weight coefficient control unit 22 includes delay units 226 corresponding to the number of array elements.

  Frequency conversion unit 11, A / D conversion unit 12, orthogonal demodulation unit 13, array synthesis unit 14, GI removal unit 15, FFT unit 16, channel estimation unit 17, channel equalization unit 18, demapping unit 19, and parallel serial conversion The unit 20 is as described in the first embodiment.

  Here, the equivalent baseband signals output from the orthogonal demodulation units 13 corresponding to the number of array elements are divided into two, one being input to the multiplication unit 141 of the array synthesis unit 14 and the other being input to the delay unit 226 of the weight coefficient control unit 22. The Also, the channel equalized carrier symbols output from the channel equalization unit 18 are divided into two, one to the demapping unit 19 of the symbol reproduction unit 23 and the other to the modulation error ratio calculation unit 222 of the weight coefficient control unit 22. Entered. Further, the parallel signal output from the demapping unit 19 of the symbol reproduction unit 23 is divided into two, one being input to the parallel-serial conversion unit 20 and the other being input to the remapping unit 221.

  The remapping unit 221 remaps the parallel signal input from the demapping unit 19 to generate a regenerated carrier symbol. The reproduced carrier symbols output from the remapping unit 221 are divided into two, one being input to the modulation error ratio calculating unit 222 and the other being input to the pilot signal inserting unit 223.

  The modulation error ratio calculation unit 222 calculates the modulation error ratio using the channel equalized carrier symbol input from the channel equalization unit 18 and the regenerated carrier symbol input from the remapping unit 221. The modulation error ratio output from the modulation error ratio calculation unit 222 is input to the pilot signal insertion unit 223.

  Pilot signal insertion section 223 generates a new playback carrier symbol including a known pilot signal, based on the playback carrier symbol input from remapping section 221 and the modulation error ratio input from modulation error ratio calculation section 222. . Specifically, pilot signal insertion section 223 generates a known pilot signal, and replaces the regenerated carrier symbol assigned to the pilot signal among the input regenerated carrier symbols with the generated known pilot signal. In this case, the input modulation error ratio is compared with a predetermined threshold value and replaced when the modulation error ratio is lower than the threshold value and the symbol number is an integral multiple of a predetermined symbol interval. The pilot signal is multiplied by a predetermined constant. The regenerated carrier symbol output from pilot signal insertion section 223 is input to multiplication section 224.

  Multiplier 224 multiplies the reproduced carrier symbol input from pilot signal inserter 223 by the channel response input from channel estimator 17. The regenerated carrier symbol multiplied by the channel response output from the multiplier 224 is input to the IFFT unit 225.

  The IFFT unit 225 converts the reproduction carrier symbol input from the multiplication unit 224 and multiplied by the channel response into a signal in the time domain. The time domain signal output from the IFFT unit 225 is input to the weight coefficient calculation unit 228. Here, the time domain signal output from the IFFT unit 225 is handled as a reference signal in the weighting factor calculation unit 228.

  The delay units 226 corresponding to the number of array elements delay the equivalent baseband signals input from the respective quadrature demodulation units 13 by a predetermined time. The predetermined time is a time required for an operation until the equivalent baseband signal output from the quadrature demodulator 13 is array-synthesized by the array synthesizer 14 and the reference signal is output from the IFFT unit 225. Say. The delayed equivalent baseband signals output from the delay units 226 corresponding to the number of array elements are input to the GI removal unit 227, respectively.

  The GI removal unit 227 removes the GI from the delayed equivalent baseband signal input from each delay unit 226, and extracts an equivalent baseband signal having a time corresponding to an effective symbol period. The equivalent baseband signals in the effective symbol period output from the GI removal units 227 for the number of array elements are input to the weight coefficient calculation unit 228.

  The weighting factor calculation unit 228 is a time domain signal multiplied by an equivalent baseband signal in an effective symbol period input from the GI removal unit 227 for the number of array elements and a channel response input from the IFFT unit 225. A weighting factor is calculated using the reference signal. Specifically, the weighting factor calculation unit 228 uses the least square error norm so that the square error between the array synthesized signal obtained by weighting the input equivalent baseband signal and the input reference signal is minimized. To calculate the weighting coefficient. The weighting factor output from the weighting factor calculation unit 228 is input to the complex conjugate unit 143 of the array synthesis unit 14.

  As described above, according to the OFDM signal combining receiver 2 of the second embodiment, the array combining unit 14 performs combining using the weighting coefficient in the time domain to generate an array combined signal, and the channel estimation unit 17 The channel response is estimated using the pilot signal, and the channel equalization unit 18 equalizes the carrier symbol in the frequency domain. Also, the weight coefficient control unit 22 demaps and remaps the channel equalized carrier symbol to generate a regenerated carrier symbol, and inserts a predetermined pilot signal into the regenerated carrier symbol to multiply the channel response. A time domain signal obtained by IFFT of the multiplication result is used as a reference signal, and a weighting factor used for array synthesis is obtained by optimization so that an error between the array synthesis signal and the reference signal is minimized. I made it. Thus, as in the case of the first embodiment, the number of weighting factors to be optimized is the number of branches to be array-synthesized, and therefore the weighting factor requires a lower calculation amount than in the prior art. Further, since the channel estimation unit 17 directly estimates the channel response, the amount of calculation is similarly reduced. In addition, since array synthesis and channel equalization are performed with such a low calculation amount, it is possible to effectively suppress interference waves even in a poor reception environment.

  Further, according to the OFDM signal combining receiver 2 of the second embodiment, the IFFT section 225 converts the re-mapped regenerated carrier symbol and the predetermined pilot signal into the time domain, thereby generating a reference signal. Then, the weighting factor calculation unit 228 calculates a weighting factor optimized so as to minimize an error between the array synthesized signal obtained by synthesizing the equivalent baseband signal input from the GI removing unit 227 and the reference signal. . Thus, the weight coefficient calculation unit 228 allows the data carrier other than the pilot signal to interfere with the desired wave even when the symbol synchronization position of the desired wave and the interference wave and the 4-symbol period in which the pilot signal is inserted into the same subcarrier match. Since it is different from the wave, a weighting factor that can distinguish the desired wave and the interference wave can be calculated.

  Further, according to the OFDM signal combining receiver 2 of the second embodiment, the pilot signal inserting unit 223 has the modulation error ratio calculated by the modulation error ratio calculating unit 222 when inserting the pilot signal into the reproduction carrier signal. A pilot signal multiplied by a predetermined constant is inserted when the value is below the threshold and the symbol number is an integral multiple of a predetermined symbol interval. In this case, the reference signal generated by IFFT section 225 is a signal based on a pilot signal having an amplitude different from that of the original pilot signal. Then, the weighting factor calculation unit 228 uses an array synthesis signal obtained by synthesizing the equivalent baseband signal input from the GI removal unit 227 and a reference signal based on a pilot signal having an amplitude different from that of the original pilot signal. Thus, a weighting factor optimized so as to minimize the error of these signals is calculated. As a result, the weighting factor calculation unit 228 uses a signal in which the amplitude of the pilot signal in the interference wave included in the received signal is different from the amplitude of the pilot signal in the reference signal. Can be calculated. On the other hand, when the pilot signal is inserted without multiplying a predetermined constant, the symbol synchronization position of the desired wave and the interference wave and the 4 symbol period in which the pilot signal is inserted into the same subcarrier match, and the desired wave When the power ratio of the interference wave is very small, the amplitude of the pilot signal in the interference wave included in the received signal matches the amplitude of the pilot signal in the reference signal, so that the weight that can distinguish the desired wave from the interference wave The coefficient could not be calculated. Therefore, if the symbol synchronization position of the desired wave and the interference wave and the period of the four symbols in which the pilot signals are inserted into the same subcarrier match and the power ratio between the desired wave and the interference wave is very small, the interference with the desired wave Waves could not be distinguished and interference waves could not be suppressed. However, according to the OFDM signal combining receiver 2 of the second embodiment, this problem can be solved.

  For the details of the components in the OFDM signal combining receiver 1 of the first embodiment and the OFDM signal combining receiver 2 of the second embodiment configured as described above, ISDB-T (Integrated Services Digital Broadcasting-Terrestrial) is applied. The case will be described below as an example.

ここで、ｔは時刻、Ｌはブランチ数を示す。また、上付きのＴは転置を示す。 [Array composition section]
First, the array combining unit 14 shown in FIGS. 1 and 2 will be described. Vectors composed of equivalent baseband signals output from the orthogonal demodulation units 13 corresponding to the number of array elements are shown below.

Here, t represents time and L represents the number of branches. Superscript T indicates transposition.

これにより、アレー合成部１４は、以下に示す演算を行い、アレー合成信号を出力する。

ここで、上付きのＨは複素共役転置を示す。なお、前記式（３）には、複素共役部１４３による処理が含まれる。 A vector composed of the weighting coefficients output from the weighting coefficient calculating unit 30 is shown below.

Thereby, the array synthesizing unit 14 performs the following calculation and outputs an array synthesized signal.

Here, the superscript H indicates complex conjugate transpose. The equation (3) includes processing by the complex conjugate unit 143.

[Channel estimation section]
Next, the channel estimation unit 17 shown in FIGS. 1 and 2 will be described. FIG. 3 is a block diagram showing the configuration of the channel estimation unit 17. The channel estimation unit 17 includes a pilot extraction unit 171, a pilot generation unit 172, a division unit 173, and an interpolation unit 174.

  The carrier symbol output from the FFT unit 16 is input to the pilot extraction unit 171 of the channel estimation unit 17. Pilot extraction section 171 extracts a pilot signal transmitted as a carrier symbol having a predetermined symbol number and subcarrier number from the input carrier symbols. The pilot signal output from pilot extraction section 171 is input to division section 173 as a received pilot signal.

  Pilot generator 172 generates a pilot signal having a predetermined amplitude and phase. The pilot signal output from pilot generator 172 is input to divider 173.

  Dividing section 173 divides the received pilot signal input from pilot extracting section 171 by the pilot signal input from pilot generating section 172 to obtain the channel response in the symbol and subcarrier in which the pilot signal is transmitted. The channel response in the pilot signal output from division unit 173 is input to interpolation unit 174.

  Interpolation section 174 interpolates the channel response in the symbols and subcarriers transmitted from pilot signal, which are input from division section 173, in the symbol direction and subcarrier direction, and calculates the channel responses in all subcarriers of the OFDM signal. Output.

ただし、ｍｏｄは剰余を示す。以下、前記式（４）を満足するｉ，ｋをそれぞれｉ_ｐ，ｋ_ｐとする。 In the ISDB-T system, subcarriers allocated to a pilot signal SP (Scattered Pilot) satisfy the following equation, where the symbol number is i and the subcarrier number is k.

However, mod indicates a remainder. Hereinafter, i and k satisfying the above-mentioned formula (4) are assumed to be i _p and k _p , respectively.

Here, the pilot extraction unit 171 shown in FIG. 3 is an SP extraction unit, and the pilot generation unit 172 is an SP generation unit. The received SP signal extracted by the SP extraction unit is X _{ip, kp,} and the SP signal generated by the SP generation unit, that is, the SP signal generated and transmitted by the ISDB-T modulator on the transmission side (hereinafter simply referred to as “SP signal”). _Assuming that “transmission SP signal” is S _{ip, kp} , the channel response U _{ip, kp at} symbol number i _p and subcarrier number k _p is expressed by the following equation.

  Here, the method of calculating the channel response using the SP signal employed in the ISDB-T system as a reference signal has been described. However, any signal that has a known amplitude and phase and that can be generated on the receiving side can be used. Similarly, it can be used as a reference signal for channel response calculation. That is, the present invention is not limited to using the SP signal as the reference signal in calculating the channel response.

  By the way, when the channel response is calculated using the SP signal, the channel responses for all symbols and subcarriers cannot be directly calculated. In order to calculate channel responses in all symbols and subcarriers, it is necessary to perform interpolation processing in the symbol and subcarrier directions.

また、線形補間法を用いる場合、以下の式により補間処理を行う。

For interpolation in the symbol direction, for example, the following latest value holding method or linear interpolation method can be used, and interpolation processing can be performed by the following equation. When the latest value holding method is used, interpolation processing is performed using the following formula.

Further, when the linear interpolation method is used, interpolation processing is performed according to the following formula.

On the other hand, for interpolation in the subcarrier direction, for example, the following linear interpolation method can be used, and interpolation processing can be performed by the following equation.

[Channel equalization section]
Next, the channel equalizer 18 shown in FIGS. 1 and 2 will be described. FIG. 4 is a block diagram showing the configuration of the channel equalization unit 18. The channel equalization unit 18 includes a division unit 181.

ここで、Ｚ_ｋはチャネル等化後のキャリヤシンボルを示す。 The carrier symbol output from the FFT unit 16 and the channel response output from the channel estimation unit 17 are input to the division unit 181 of the channel equalization unit 18. The division unit 181 of the channel equalization unit 18 uses the channel response U _k input from the channel estimation unit 17 for the carrier symbol X _k input from the FFT unit 16 for each subcarrier of the OFDM signal, as in the following equation. Channel equalization is performed by dividing.

Here, Z _k represents a carrier symbol after channel equalization.

[Symbol playback unit (demapping unit, remapping unit)]
Next, the demapping unit 19 and the remapping unit 221 of the symbol reproduction unit 23 shown in FIG. 2 will be described.

ただし、Ｔ_ｋは送信キャリヤシンボル、Ｎ_ｋは雑音成分を示す。 The demapping unit 19 estimates the transmitted carrier symbol from the channel equalized carrier symbol input from the channel equalizing unit 18 and extracts a parallel signal composed of a plurality of bits. The remapping unit 221 performs carrier modulation on the parallel signal (a parallel signal composed of a plurality of bits) input from the demapping unit 19 to reproduce a carrier symbol. Here, the formula (9) can be expressed by the following formula.

Here, T _k represents a transmission carrier symbol, and N _k represents a noise component.

ここで、ｄｅｃはマッピングおよび再マッピングを示す関数であり、具体的には、与えられたキャリヤシンボルとの間のノルムが最も小さい送信キャリヤシンボルを返す関数である。すなわち、シンボル再生部２３は、チャネル等化後のキャリヤシンボルをデマッピング部１９にてデマッピングし、再マッピング部２２１にて再マッピングすることにより、送信キャリヤシンボルを得ることができる。 Therefore, when the noise component N _k is sufficiently small, it can be expressed by the following equation.

Here, dec is a function indicating mapping and remapping, and specifically, a function that returns a transmission carrier symbol having the smallest norm with respect to a given carrier symbol. That is, the symbol reproduction unit 23 can obtain a transmission carrier symbol by demapping the channel equalized carrier symbol by the demapping unit 19 and remapping by the remapping unit 221.

ここで、Ｋは全サブキャリヤ数を示す。なお、変調誤差比の詳細については、下記の非特許文献を参照されたい。
ETR 290：Measurement guidelines for DVB Systems, ETSI Technical Report, may 1997 [Modulation error ratio calculation unit]
Next, the modulation error ratio calculation unit 222 shown in FIG. 2 will be described. The modulation error ratio calculation unit 222 uses the channel equalized carrier symbol Z _k input from the channel equalization unit 18 and the remapped regenerated carrier symbol D _k input from the remapping unit 221. A modulation error ratio MER defined by the following equation is calculated.

Here, K indicates the total number of subcarriers. Refer to the following non-patent document for details of the modulation error ratio.
ETR 290: Measurement guidelines for DVB Systems, ETSI Technical Report, may 1997

[Pilot signal insertion section]
Next, the pilot signal insertion unit 223 shown in FIG. 2 will be described. The pilot signal insertion unit 223 replaces the carrier symbol assigned to the pilot signal among the regenerated carrier symbols D _k input from the remapping unit 221 with a carrier symbol S _k having a predetermined amplitude and phase. That is, the processing shown in the following formula is performed.

〔perturbation〕
Here, when the pilot signal insertion unit 223 performs the processing of the equation (13), when the symbol synchronization position and the 4-symbol period in which the SP is inserted into the same subcarrier completely match in the desired wave and the interference wave The interference wave may not be removed satisfactorily. In order to cope with this, the pilot signal insertion unit 223 performs processing shown in the following equation.

The modulation error ratio obtained by the equation (12) indicates the certainty as the estimated value of the transmission symbol of the carrier symbol, in other words, the signal quality after interference removal. When the modulation error ratio MER is less than a predetermined threshold th and the symbol number i is an integral multiple of a predetermined symbol interval m, a pilot signal carrier having a predetermined amplitude and phase The symbol S _k is multiplied by a constant α ′.

  According to the experiments by the inventors of the present application, even if the symbol synchronization position and the 4-symbol period in which the SP is inserted in the same subcarrier completely match in the desired wave and the interference wave, for example, th = 19 dB , Α ′ = 1/2 and m = 3, the interference wave can be satisfactorily removed.

  When the modulation error ratio MER is less than a predetermined threshold th and the symbol number i is an integral multiple of a predetermined symbol interval m, the pilot signal inserting unit 223 includes the pilot included in the regenerated carrier symbol. A reference signal is generated based on a signal obtained by replacing the carrier symbol assigned to the signal with a carrier symbol of a pilot signal having a small amplitude. Then, the weight coefficient calculation unit 228 generates an array combined signal obtained by combining the equivalent baseband signal input from the GI removal unit 227 and a reference signal based on the signal replaced with a carrier symbol of a pilot signal having a small amplitude. And calculating a weighting factor optimized so as to minimize the error of these signals. As a result, the weighting factor calculation unit 228 can distinguish between the desired wave and the interference wave, and can calculate a weighting factor by which the array synthesis unit 14 can remove the interference wave.

[Pilot signal generator]
Next, the pilot signal generation unit 211 shown in FIG. 1 will be described. Pilot signal generation section 211 generates a carrier symbol having a predetermined amplitude and phase at a predetermined symbol number and subcarrier number, and at the same time, a carrier having a symbol number and subcarrier number not allocated to the pilot signal. Set the symbol to zero.

[Frequency characteristics multiplier]
Next, the frequency characteristic multiplier that is the multiplier 212 shown in FIG. 1 and the multiplier 224 shown in FIG. 2 will be described. Multiplier 212 receives a pilot signal (carrier symbol) from pilot signal generator 211. Multiplier 224 inputs a regenerated carrier symbol (carrier symbol) from pilot signal inserting unit 223.

に、チャネル推定部１７から入力されるチャネル応答Ｕ_ｋを乗算する。

ここで、Ｒ_ｋは、チャネル応答が乗じられたキャリヤシンボルを示し、周波数領域で表現した参照信号である。 Then, the multipliers 212 and 224 input the carrier symbol

Is multiplied by the channel response U _k input from the channel estimation unit 17.

Here, R _k represents a carrier symbol multiplied by a channel response, and is a reference signal expressed in the frequency domain.

  Since the reference signal in the frequency domain output from the multipliers 212 and 224 is a signal calculated using the channel response, the reference signal includes the channel response, and the weight coefficient calculation units 30 and 228 include the channel response. A weighting factor is calculated using a reference signal expressed in a time domain. Therefore, even in an environment where a multipath of a desired wave is received, the weighting factor calculation units 30 and 228 can calculate a weighting factor with which the array synthesis unit 14 can remove the interference wave. That is, the interference wave is removed by the array combining unit 14, and the frequency characteristic distortion due to multipath is equalized by the channel equalizing unit 18. As a result, the OFDM signal combining receivers 1 and 2 can receive the OFDM signal satisfactorily.

ここで、ｒ（ｔ）は参照信号を示す。 [IFFT part]
Next, the IFFT unit 213 shown in FIG. 1 and the IFFT unit 225 shown in FIG. 2 will be described. IFFT sections 213 and 225 convert carrier symbols R _k input from multiplication sections 212 and 224, which are frequency characteristic multiplication sections, into time-domain signals using IFFT, and generate reference signals. The IFFT units 213 and 225 perform processing shown in the following formula.

Here, r (t) represents a reference signal.

[Delay part]
Next, the delay unit 226 shown in FIG. 2 will be described. The delay unit 226 outputs an equivalent baseband signal from the orthogonal demodulation unit 13, and then the array synthesis unit 14, the GI removal unit 15, the FFT unit 16, the channel equalization unit 18, the demapping unit 19, the remapping unit 221, A delay corresponding to the difference between the time required for processing by pilot signal insertion unit 223, multiplication unit 224, and IFFT unit 225, and IFFT unit 225 to output a reference signal, and the time required for processing by GI removal unit 227 To output an equivalent baseband signal.

[Weight coefficient calculation unit]
Next, the weighting factor calculation unit 30 shown in FIG. 1 and the weighting factor calculation unit 228 shown in FIG. 2 will be described. FIG. 5 is a block diagram showing the configuration of the weight coefficient calculation units 30 and 228. The weight coefficient calculation units 30 and 228 include an autocorrelation matrix calculation unit 301, an inverse matrix calculation unit 302, a cross correlation vector calculation unit 303, and a multiplication unit 304.

また、アレー合成信号を以下の式に示す。

The weighting coefficient calculating units 30 and 228 are input from the IFFT unit 225 and the array combined signal obtained by weighting and combining the input signal vectors composed of equivalent baseband signals input from the GI removing units 214 and 227 for the number of array elements. A weighting coefficient vector that minimizes an error from the reference signal is generated. The input signal vector is shown in the following equation.

The array composite signal is shown in the following equation.

および前記式（３）に示した

は、ＧＩ期間および有効シンボル期間の等価ベースバンド信号を示しているのに対し、前記式（１９）に示したｘ（ｔ）および前記式（２０）に示したｙ（ｔ）は、ＧＩが除去された、有効シンボル期間のみの等価ベースバンド信号を示している。 The weighting factor calculation units 30 and 228 optimize the weighting factor so that the square error between the array synthesis signal of the equation (20) and the reference signal r (t) is minimized. Here, the above equation (1)

And the above formula (3)

Represents the equivalent baseband signal of the GI period and the effective symbol period, whereas x (t) shown in the equation (19) and y (t) shown in the equation (20) The equivalent baseband signal with only the effective symbol period removed is shown.

ここで、Ｅ［・］は期待値演算を示す。 The square error is defined by the following equation.

Here, E [•] indicates an expected value calculation.

ここで、Ｒ_ｘｘはｘ（ｔ）の自己相関行列、ｒ_ｘｒはｘ（ｔ）とｒ（ｔ）の相互相関ベクトルを示す。 The weighting coefficient w _opt for minimizing the square error of the equation (21) is expressed by the following equation.

Here, R _xx represents an autocorrelation matrix of x (t), and r _xr represents a cross-correlation vector of x (t) and r (t).

係数更新間隔を例えば１シンボル間隔として、これらを更新する。ただし、上付きの＊は複素共役を示す。 Here, the autocorrelation matrix and the cross-correlation vector are respectively expressed by the following equations,

These are updated by setting the coefficient update interval to, for example, one symbol interval. However, the superscript * indicates a complex conjugate.

ここで、ｎは係数更新時間を示す。また、λは、０≦λ＜１を満たす適応係数を示し、忘却係数と呼ばれる。さらに、前記式（２５）および（２６）における右辺の第２項の期待値演算は、係数更新間隔、例えば１シンボルのうちで有効シンボル期間に相当する期間における期待値を示す。 That is, the autocorrelation matrix calculation unit 301 performs the following equation (25) using the input signal vector x (t), and calculates the self-phase matrix R _xx (n). Further, the cross-correlation vector calculation unit 303 performs processing of the following equation (26) using the input signal vector x (t) and the reference signal r (t) that are input, and _obtains the cross-correlation vector r _xr (n). calculate.

Here, n indicates a coefficient update time. Λ represents an adaptive coefficient that satisfies 0 ≦ λ <1 and is called a forgetting coefficient. Further, the expected value calculation of the second term on the right side in the equations (25) and (26) indicates an expected value in a coefficient update interval, for example, a period corresponding to an effective symbol period in one symbol.

The inverse matrix calculation unit 302 calculates an inverse matrix R _xx ⁻¹ (n) of the self-phase matrix R _xx (n) calculated by the autocorrelation matrix calculation unit 301. The multiplying unit 304 multiplies the inverse matrix R _xx ⁻¹ (n) of the self-phase matrix calculated by the inverse matrix calculating unit 302 and the cross correlation vector r _xr (n) calculated by the cross correlation vector calculating unit 303. Then, the weight coefficient vector w is calculated. Therefore, the weight coefficient calculation units 30 and 228 can obtain the weight coefficient vector w by the following expression.

〔Experimental result〕
Next, the experimental result of the interference removal characteristic obtained by computer simulation will be described. FIG. 6A is a diagram illustrating interference cancellation characteristics with respect to symbol synchronization positions in a conventional Pre-FFT type OFDM signal combining receiver (OFDM signal combining receiver described in Non-Patent Document 1 described above). is there. FIG. 6B is a diagram showing interference cancellation characteristics with respect to symbol synchronization positions in the OFDM signal combining receiver 2 according to the embodiment of the present invention. These experimental results indicate that both the desired wave and the interference wave are based on ISDB-T, and when the GI ratio is 1/8 and the same, after the interference removal with respect to the difference in symbol synchronization position between the desired wave and the interference wave. The modulation error ratio is shown as the interference cancellation characteristic. Also, the received C / N is 25 dB, the D / U of the interference wave is 10 dB, the desired wave comes from 0 degree, and the interference wave comes from 50 degree, and these signals are received by a two-element array antenna. The interval was half wavelength. The difference in symbol synchronization position is shown on the horizontal axis, and the modulation error ratio after interference removal is shown on the vertical axis.

  According to the experimental result of the conventional Pre-FFT OFDM signal combining receiver shown in FIG. 6A, when the difference in symbol synchronization position is an integral multiple of the symbol length, the interference cancellation characteristic is degraded. I understand that. On the other hand, according to the experimental result of the OFDM signal combining receiver 2 according to the embodiment of the present invention shown in FIG. 6B, the interference is satisfactorily removed regardless of the difference in symbol synchronization position. Recognize.

  FIG. 7 shows interference cancellation characteristics in a multipath environment of a conventional OFDM signal synthesis receiver (OFDM signal synthesis receiver described in Patent Document 2) and an OFDM signal synthesis receiver 2 according to an embodiment of the present invention. FIG. This experimental result shows the modulation error ratio after interference removal for the multipath D / U as an interference removal characteristic in an environment where a multipath wave of the desired wave is received in addition to the desired wave and the interference wave. The received C / N is 45 dB, the interference wave D / U is 10 dB, the desired wave is 0 degrees, the interference wave is 50 degrees, and the desired multipath wave is -60 degrees. Are received by a two-element array antenna, and the interval is set to a half wavelength. The multipath wave has a delay time of 120 μs (about 95% of the GI length 126 μs), the D / U of the multipath wave is shown on the horizontal axis, and the modulation error ratio after interference removal is shown on the vertical axis.

  According to the experimental results shown in FIG. 7, when the multipath D / U is 4 dB or more, the conventional OFDM signal combining receiver is higher than the OFDM signal combining receiver 2 according to the embodiment of the present invention. It can be seen that the modulation error ratio is obtained. On the other hand, when the multipath D / U is 3 dB or less, interference is satisfactorily removed in the OFDM signal combining receiver 2 according to the embodiment of the present invention, whereas in the conventional OFDM signal combining receiver, It can be seen that the reception characteristics are degraded.

  As described above, in a broadcast wave relay device, the minimum requirement is that a required BER is obtained after interference cancellation, and it is required how to suppress deterioration of transmission characteristics due to interference, whereas reception in a service area is required. In the apparatus for use, a required BER may be obtained after interference removal. That is, when it is assumed that the OFDM signal combining receiver is applied to a receiver, it is only necessary to obtain a required BER after removing interference, and it is not necessary to obtain a lower BER if the required BER is exceeded. . Therefore, according to the experimental results shown in FIG. 7, when the OFDM signal synthesis receiver 2 according to the embodiment of the present invention is compared with the conventional OFDM signal synthesis receiver, the OFDM signal synthesis according to the embodiment of the present invention is compared. It can be seen that the receiving apparatus 2 is more resistant to a low D / U multipath environment.

  As described above, according to the OFDM signal combining receivers 1 and 2, the array combining unit 14 performs weighted combining in the time domain to generate an array combined signal, and the channel equalizing unit 18 is complex in the frequency domain. Division is performed to generate equalized carrier symbols, and weight coefficient control units 21 and 22 perform predetermined pilot signals (in the case of OFDM signal combining receiver 1) or regenerated carrier symbols (OFDM signal combining reception). In the case of apparatus 2, the time domain signal obtained by multiplying and remodulating the channel response is used as a reference signal, and the weighting factor used for the array synthesis is obtained by optimization according to the least square error criterion. . As a result, processing can be performed with a low calculation amount, and interference waves can be effectively suppressed even in a poor reception environment.

  In OFDM signal combining receiver 1 shown in FIG. 1, multiplication section 212 multiplies the pilot signal by the channel response, and weighting coefficient calculation section 30 outputs a signal obtained by IFFT of IFFT section 213. Handled as a reference signal. Similarly, in OFDM signal combining receiver 2 shown in FIG. 2, multiplication section 224 multiplies the reproduced carrier symbol after the pilot signal insertion by the channel response, and weight coefficient calculation section 228 outputs the multiplication result to IFFT section 225. The IFFT signal is handled as a reference signal. In this case, IFFT section 213 performs IFFT on the pilot signal that has not been multiplied by the channel response, IFFT section 225 performs IFFT on the reproduced carrier symbol that has not been multiplied by the channel response, and weight coefficient calculation sections 30 and 228 A signal subjected to IFFT without including the channel response may be handled as a reference signal.

It is a block diagram which shows the structure of the receiver for OFDM signal synthesis | combination by the 1st Embodiment (Example 1) of this invention. It is a block diagram which shows the structure of the receiver for OFDM signal synthesis by the 2nd Embodiment (Example 2) of this invention. It is a block diagram which shows the structure of a channel estimation part. It is a block diagram which shows the structure of a channel equalization part. It is a block diagram which shows the structure of a weighting coefficient calculation part. (A) is a figure which shows the interference removal characteristic with respect to a symbol synchronous position in the conventional receiver for pre-FFT type | mold OFDM signal synthesis | combination. (B) is a figure which shows the interference removal characteristic with respect to a symbol synchronous position in the receiver for OFDM signal synthesis by embodiment of this invention. It is a figure which shows the interference removal characteristic in the multipath environment of the receiver for conventional Pre-FFT type OFDM signal synthesis and the receiver for OFDM signal synthesis according to the embodiment of the present invention.

Explanation of symbols

1, 2 OFDM signal combining receiver 10 Antenna 11 Frequency converting unit 12 A / D converting unit 13 Orthogonal demodulating unit 14 Array combining unit 15, 214, 227 GI removing unit 16 FFT unit 17 Channel estimating unit 18 Channel equalizing unit 19 Demapping unit 20 Parallel serial conversion unit 21, 22 Weight coefficient control unit 23 Symbol reproduction unit 30, 228 Weight coefficient calculation unit 141, 212, 224, 304 Multiplication unit 142 Addition unit 143 Complex conjugate unit 171 Pilot extraction unit 172 Pilot generation unit 173, 181 Division unit 174 Interpolation unit 211 Pilot signal generation unit 213, 225 IFFT unit 221 Remapping unit 222 Modulation error ratio calculation unit 223 Pilot signal insertion unit 226 Delay unit 301 Autocorrelation matrix calculation unit 302 Inverse matrix calculation unit 303 Cross correlation Vector calculator

Claims (4)

An OFDM signal combining receiver that receives an OFDM wave by an array antenna including a plurality of array elements and outputs a bit string,
An array combining unit that combines the equivalent baseband signals of the received OFDM waves by weighting in the time domain and generates an array combined signal;
An FFT unit for converting the array combined signal into a carrier symbol which is a frequency domain signal;
A channel estimation unit for estimating a channel response using a pilot signal;
A channel equalizer for performing channel equalization by dividing the carrier symbol by the channel response;
A weighting factor control unit that controls a weighting factor used for array synthesis by weighting in the time domain,
The weight coefficient control unit
A symbol reproducing unit for reproducing a symbol by demapping and remapping the carrier symbol after the channel equalization, and generating a reproduced carrier symbol;
A pilot signal insertion unit for inserting a pilot signal into the regenerated carrier symbol;
An IFFT unit for converting the regenerated carrier symbol into which the pilot signal is inserted into a time domain signal;
A weighting factor calculating unit that obtains the weighting factor by optimizing the time domain signal as a reference signal and optimizing the error between the array combined signal and the reference signal to be a minimum. OFDM signal combining receiver. The weight coefficient control unit
The reproduction carrier symbol pilot signal is inserted by the pilot signal insertion unit includes a multiplying unit for multiplying the channel response,
The OFDM signal combining receiver according to claim 1 , wherein the IFFT unit converts the multiplication result into a signal in a time domain. The weight coefficient control unit
A modulation error ratio calculation unit for calculating a modulation error ratio based on the carrier symbol after channel equalization and the reproduced carrier symbol;
The pilot signal inserting unit inserts a predetermined pilot signal or a new pilot signal obtained by multiplying the pilot signal by a predetermined constant based on the modulation error ratio into the regenerated carrier symbol; The OFDM signal synthesizing receiver according to claim 1 . When the pilot signal insertion unit has the modulation error ratio below a predetermined threshold value and the symbol number of the carrier symbol is an integral multiple of a predetermined symbol interval, 4. The OFDM signal combining receiver according to claim 3 , wherein a new pilot signal obtained by multiplying a signal by a predetermined constant is inserted into the reproduction carrier symbol.

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