JP4868341B2 - Method and system for use in an antenna system - Google Patents

Abstract

A method includes transmitting at least two space-time coded signals in respective beams of a multi-beam antenna array, measuring a channel impulse response for each space-time coded signal at a second station, and sending an indicia of a selected set of least attenuated signals from the second station to the first station. The multi-beam antenna array is associated with a first station. The beams transmit a signature code embedded in each respective space-time coded signal, and the signature codes are orthogonal so that the second station can separate and measure the channel impulse response corresponding to each space-time coded signal. The space-time coded signals include the selected set of least attenuated signals and a remaining set of most attenuated signals. <IMAGE>

Description

としての12素子受信信号ベクトルに形成される。次に、受信信号ベクトル

の複素共役転置が行ベクトル

として形成され、受信信号の空間共分散行列

がステップS213（図16）で算出される。受信信号ベクトル

の長さが素子12個分であるとき、受信信号の空間共分散行列

は12x12行列になる。
【００６０】
アレイ・ステアリング・ベクトル

は、フェーズド・アレイ・アンテナの各放射素子ごとに１つのベクトル要素を有する列ベクトルである。たとえば、フェーズド・アレイ・アンテナが12個の放射素子を含む場合（たとえば、２分の１ダイポール）、アレイ・ステアリング・ベクトル

は12個のベクトル素子を含む。アレイ・ステアリング・ベクトル

は、図７の定数Ｃであり、フェーズド・アレイ・アンテナのビームを方位角θの方へ向けるのに用いられる。各ベクトル要素は次式によって与えられる。
【００６１】
【数１】

上式で、ｋは、2πを波長で割った値であり、ｍは、フェーズド・アレイ・アンテナの放射素子に関連する数を定める０からＭ（たとえば、12素子アンテナの場合は０から11）の指数であり、ｄは、フェーズド・アレイ・アンテナの放射素子間の離隔距離であり（好ましくは波長の２分の１）、θは、形成されるアンテナ・ビームの方位角である。
【００６２】
受信ビームにおける受信信号の到来角θを形成するように完全なベクトルが組み合わされるように、アレイ・ステアリング・ベクトル

の各ベクトル要素は、図７に示されているように定数Ｃの対応するベクトル要素である。この場合、θは、フェーズド・アレイ・アンテナの好都合な基準方向に対する角度である。アレイ・ステアリング・ベクトル

の複素共役転置は行ベクトル

である。
【００６３】
積

は、この場合も、フェーズド・アレイ・アンテナの各放射素子ごとに１つのベクトル要素を有する列ベクトルである。積

は、到来角θでの角パワー・スペクトルP(θ)の値を与えるようにステップS214（図16）で求められる単一の点、すなわちスカラである。したがって、角パワー・スペクトルP(θ)は、図18のG1に示されており、次式のように算出される。
【００６４】
【数２】

上式で、

はアレイ・ステアリング・ベクトルであり、

は受信信号ベクトルであり、

は受信信号の空間共分散行列であり、Ｈは複素共役転置を示す。
【００６５】
アレイ・ステアリング・ベクトルを算出するための上述の数式では、波長の２分の１の間隔を置いて配置された放射素子が直線的に配置されると仮定している。しかし、当業者には、曲線パスに沿って配置された放射素子に関するアレイ・ステアリング・ベクトルをどのように算出すべきかが理解されよう。有利なことに、図８に示されているアンテナ・システムで、３つのわずかに「湾曲した」アンテナ・アレイを使用することができる。実際には、アンテナ・アレイは、円を形成するほどかなり「湾曲」していてよい（たとえば、図６）。当業者には、放射素子のこのようなかなり湾曲したアレイに関するアレイ・ステアリング・ベクトルの計算において、好都合なことに、アレイ・ステアリング・ベクトルの振幅制御と位相制御が使用されることが理解されよう。
【００６６】
性能を向上させるために、繰り返し得られた測定値を平均することによって角パワー・スペクトルが求められる。図17において、ステップS211で、アレイ・ステアリング・ベクトルが準備され、アンテナ・ビームの向きが定められる。ステップS215AおよびS215Bを反復することによって複数回の測定が行われる。このループ内では、ステップ216で受信信号ベクトル

が繰り返し測定され、ステップS217で共分散行列Ｒが繰り返し求められ保存される。次いで、ステップS218で平均共分散行列が求められ、ステップS214で角パワー・スペクトルP(θ)が求められる。このように平均を求めると、所定の各方向θについての時間間隔にわたって数回繰り返される。このように、高速フェージング現象が平均される。この期間は、移動遠隔局２がこの平均期間中に位置するビームを変化させるのに十分な程度に移動遠隔局２の位置が変化しないほど短い期間でなければならない。この期間は好ましくは、高速フェージング効果を平均するチャネル・コヒーレンシ時間より長い。チャネル・コヒーレンシ時間は、厳密かつ普遍的には定義されていないが、ドップラー拡散の逆数に比例しかつ概ね等しい時間とみなしてよい。
【００６７】
ドップラー拡散はより厳密に定義されている。基地局と移動遠隔局との間の相対速度によって、受信周波数は送信周波数に対して物理的にずれる。ドップラー拡散はこの周波数シフトの２倍である。たとえば、ドップラー周波数シフトは、相対速度と波長の比である（可能性のある単位では、メートル／秒をメートルで割った比やフィート／秒をフィートで割った比など）。移動遠隔局が13.9メートル／秒（約50km/h）で走行しており、波長が約0.15mである場合（たとえば、毎秒300000000メートルに等しい光速を有する2000MHzの信号）、ドップラー周波数シフトは92.7Hzであり、ドップラー拡散は185Hzであり、チャネル・コヒーレンシ時間は約5.4ミリ秒である。相対速度が毎秒40m（約144km/h）であるとき、チャネル・コヒーレンシ時間が約1.9ミリ秒であり、相対速度が毎秒1m（約3.6km/h）であるとき、チャネル・コヒーレンシ時間が約75ミリ秒であることを容易に検証することができる。
【００６８】
平均時間間隔は好ましくは、ドップラー拡散の逆数よりも大きく、かつ予想される角速度で移動する移動局が基地局アンテナ・システムのビーム幅の２分の１移動する時間よりも小さい値に設定される。基地局は、遠隔局のレンジを知っているか、またはレンジを信号強度から推定することができる。基地局は、所定の速度までの速度で移動できる移動局と通信するように構成されている。移動局が基地局の周りを半径方向に移動している場合、この速度をレンジで割った値を角速度とみなすことができる。平均間隔を、ビーム幅の２分の１を角速度で割った値に設定すると、移動遠隔局２がこの平均期間中に位置するビームを変化させるのに十分な程度に移動遠隔局２の位置が変化しない時間の推定値を得ることができる。
【００６９】
パワーP(θ)が平均される期間は通常、チャネル・コヒーレンシ時間よりもずっと長い。たとえば、マルチパス反射の発生率が高い環境（たとえば、都会環境）で動作する広帯域CDMAシステムでは、平均期間はタイム・スロット数十個分になることがある。マルチパス反射の発生率が高い室内環境の場合、移動がずっと低速になり、平均期間がずっと長くなる可能性がある。
【００７０】
基地局は、角パワー・スペクトルを算出し、パワー・スペクトルにおいて鋭いピークが示されているか否かを判定する。鋭いピークが示されているとき、各ピークの角位置が求められる。パワー・スペクトルが拡散しており、鋭いピークが示されていないとき、基地局は、まず受信角パワー・スペクトルが所定のしきい値（図18のG2）を超える角度を求めることによって角拡散ASを求める。しきい値は、基地局１によって検出される無線環境（たとえば、信号密度）に基づいて適応させることもできる。
【００７１】
角パワー・スペクトルの鋭いピークは、たとえば、２しきい値試験を用いることによって検出することができる。たとえば、パワー・スペクトルが第１のしきい値G3を超える第１の連続的な角範囲（度単位またはラジアン単位）を求める。次いで、パワー・スペクトルが（第１のしきい値G3よりも小さい）第２のしきい値G2を超える第２の連続的な角範囲を求める。第１の角範囲を第２の角範囲で割った比が所定の値よりも小さいときは、ピークが示されている。
【００７２】
ピークが示されているときは、角ダイバーシチ管理（すなわち、ビームの到来方向の管理）が呼び出され、かつ場合によってはビーム幅管理が呼び出される。スペクトル・ピークの鋭さは、角パワー・スペクトルを２つのしきい値と比較することによって求めることができる。たとえば、図18では、３つのピークがしきい値G2を超えているが、しきい値G3を超えているピークは２つだけである。しきい値G2に従って求められた単一のピークの角拡散は、しきい値G3に従って求められた単一のピークの角拡散よりも広い。G3によって求められた単一のピークの角拡散とG2によって求められた拡散との比は、ピークの鋭さの尺度である。あるいは、角パワー・スペクトルにおいて、パス３および５の方向を示すしきい値を超えるピークが多くても２つになるまで、角パワー・スペクトルを測定するためのしきい値を適宜移動させることができる。たとえば、角パワー・スペクトルにおいて２つの鋭いピークが発生しており、基地局が２つのビームを送信するとき、基地局はこれらのピークの方向（すなわち、パワー・スペクトルがしきい値G3を超える２つの異なる角方向）を、パス３および５（図１）の角方向に定める。これを到来ダイバーシチ角度と呼ぶ。基地局は、追尾可能なビームをそれぞれパス３および５に沿った方向に向けるか、またはそれぞれパス３および５に沿った方向を向いた一定のビームを選択する。当業者には、角ダイバーシチ管理を２つよりも多くのビームにどのように拡張するかが理解されよう。
【００７３】
場合によっては、角パワー・スペクトルは、角パワー・スペクトルのそれぞれのピークに対応する３つ以上の角位置を含む。基地局は、２つのビームを有するとき、（１）同一チャネル・ユーザがシステム全体で同一チャネル干渉を最低限に抑えるように位置する角度を避けることに基づいて、または（２）送信局の増幅器におけるパワー分散のバランスを取るように、３つ以上の角位置から第１および第２の角位置を選択する。
【００７４】
フェーズド・アレイ・アンテナにおけるビーム幅は一般に、ビーム・ステアリング・ベクトル（たとえば、図７のベクトルＣ）の要素の振幅を調節することによって選択可能である。アンテナ・システムが、制御可能なビーム幅を有するフェーズド・アレイ・アンテナを含んでおり、スペクトル・ピークが鋭いとき、基地局は、それぞれパス３および５に沿った方向に送信パワーを集中させるように、アンテナ・システムを仮定してビームをできるだけ狭い幅に設定するか、またはできるだけ幅の狭いビームを選択する。スペクトル・パワー・ピークが鋭いので、パス３および５は良好な送信特性を有すると予想される。
【００７５】
一方、角パワー・スペクトルが非常に拡散しており、ピークが弱いかまたは示されないときは、パワー・スペクトルがしきい値（たとえば、図18のG2）を超える角範囲か、または少なくとも、角パワー・スペクトルがしきい値を超えるピークをカバーするのに必要な連続的な角範囲に基づいて、概略的な角窓が求められる。このような場合、本発明の好ましい態様では、ダウン・リンク送信に用いられるすべてのビームのビーム幅の和が角拡散ASに概ね等しくなるようにビームが選択される。
【００７６】
アンテナ・システムが、制御可能なビーム幅を有するフェーズド・アレイ・アンテナを含んでいるが、スペクトル・ピークが鋭くないとき、基地局はまず、パワー・スペクトルの、しきい値を超える角範囲か、または少なくとも、角パワー・スペクトルがしきい値を超えるピークをカバーするのに必要な連続的な角範囲を角拡散として求める。次いで、基地局は、ビームのビーム幅を概ね角拡散をカバーするように設定するか、またはビームのビーム幅として、概ね角拡散をカバーする幅を選択する。これを角パワー・ダイバーシチ管理またはビーム幅管理と呼ぶ。たとえば、角拡散をカバーすることを試みる２ビーム基地局は、両方のビームのビーム幅として角範囲の約２分の１を選択し、かつ基地局は、２つのビームを実質的に角拡散をカバーするように向きにする。
【００７７】
当業者には理解されるように、より多くのビームへの拡張は容易である。たとえば、基地局が４ビーム基地局でビーム空間時間符号化を行う機能を有するとき、角拡散の約４分の１のビーム幅が各ビームに選択される。このようにして、ダウン・リンク送信はチャネルに空間的に整合する。最大角ダイバーシチ利得が得られるように直交ビームのカバレージをチャネルの角拡散に一致させると有利である。しかし、通常２つから４つのビームが適切である。
【００７８】
基地局が（６コーナ・アンテナと同様に）複数の一定のビームを持つアンテナ・システムを有しているとき、および角パワー・スペクトルが拡散しており、角拡散ASが単一のビームのビーム幅を超えているとき、本発明の望ましい変形態様では、無線チャネルによりうまく整合するように２つの隣接するビームが単一のより幅の広いビームに組み合わされる（２つの60°ビームが単一の120°ビームに組み合わされる）。このような場合、２つの隣接するビームは、同じパイロット符号または直交化符号を使用する単一のより幅の広いビームとして用いられる。一定ビーム基地局では、生成できるビームの数Ｍが、高いビーム解像度を実現できるほど大きい（たとえば、M > 4、好ましくは少なくとも８）と有利である。チャネルによりうまく整合するのにより幅の広いチャネルが必要であるときは、２つの隣接するビームを組み合わせることができる。
【００７９】
本発明は、アンテナ・システムがフェーズ・アレイ・アンテナ（たとえば、図７のアンテナ20および図10のアンテナ40）でデータ・ビーム・フォーミング技術を使用する基地局にうまく適合する。デジタル・ビーム・フォーミング技術を用いた場合、利用可能な角拡散に従っていくつかの要素で零重み付けを用いることによって、アンテナ・アレイ内の要素の見掛けの数（すなわち、見掛けの開口寸法）を電子的に調整することができる。このように、基地局によって、角拡散に一致するようにビーム幅を容易に適応させることができる。このビーム幅制御は開ループ制御システムとして働く。
【００８０】
他の態様では、角パワー・スペクトルがある大きな角範囲においてしきい値を超えるときにビーム・ホッピング技術が使用される。ビーム・ホッピング技術とは、角拡散を順次カバーする技術である。たとえば、任意の１つのタイム・スロット内の送信ビームが角拡散をカバーしないとき、この角拡散を以後のタイム・スロットの間にカバーすることができる。角拡散が120°（すなわち、４つのビームの幅）をカバーする場合に30°ビームを形成することのできる２ビーム基地局を有する例示的なシステムを考える。ビーム・ホッピング・システムでは、基地局は、120°角拡散の第１の60°セクタをカバーするように第１のタイム・スロットの間の送信用に２つの30°ビームを形成し、120°角拡散の残りの60°セクタをカバーするように第２のタイム・スロットの間の送信用に他の２つの30°ビームを形成する。
【００８１】
ビーム・ホッピングは、大きな角拡散を有する無線環境における性能を大幅に向上させる。周波数分割２重携帯無線システムでは、角拡散が大きくなると、１つには、送信方向を最適に選択する際の角度の不確かさが増すために、ダウン・リンク性能が低下することが知られている。周波数分割２重システムでは、良好なパワー送信能力（減衰が弱い）を有すると判定されたアップ・リンク方向が、ダウン・リンク送信の場合に、搬送波周波数が互いに異なるために深いフェージングを起こす可能性がある。
【００８２】
無線環境で角拡散が大きい場合、ダウン・リンク送信用の可能な方向の数も多くなる。２つの最良の方向を選択するのではなく、角パワー・スペクトルがしきい値を超える、場合によっては良好なすべての方向をカバーするようにダウン・リンク・ビームを順次形成することによって、空間ダイバーシチが実現される。これは、角拡散がセクタ全体またはセル全体をカバーすることがあるマイクロセルまたはピコセルで特に重要である。
【００８３】
遠隔局２が一定の可動性または低い可動性を有する場合、ビーム・ホッピングは２つの最良の方向を選択することに勝る他の利点を有する。最良の２つの方向が多数の連続的なバーストのビーム送信方向として選択されると、選択された方向の選択が誤っている場合（たとえば、アップ・リンクが良好でもダウン・リンクで深いフェージングが起こっている）に（データの喪失に関して）かなりのペナルティが生じる。しかし、一群の可能な方向にわたってビームをホッピングさせることによって、深いフェージングを起こしていることが判明した１方向からのデータの喪失は限られた持続時間（たとえば、１タイム・スロットのみ）しか起こらない。この角ダイバーシチは、不良な送信方向を選択することによって生じるエラーを「白化」する傾向がある。
【００８４】
さらに、ビーム・ホッピング送信の間に生成される他の遠隔局への同一チャネル干渉は、送信信号の空間拡散によって白化される傾向がある。高ビット・レート接続は高ビーム・パワーを用いて行われるため、同一チャネル干渉は、高いデータ・ビット・レートが必要であるときに特に厄介である。高ビット・レート接続で大量のビーム・パワーが使用される場合、基地局がビーム選択に非ホッピング方式を用いるときに目立つ色の干渉が起こる（一様に分散されない）。
【００８５】
図19において、本発明の他の態様は、図14を参照して説明したように基地局210および遠隔局230を含んでいる。本態様では、基地局210はそれぞれの供給信号CH1およびCH2にそれぞれの重みW1およびW2を加える重み付け増幅器102および104を含んでいる。本態様では、重みW1およびW2は、複素数であるか、またはアンテナ16および18から送信される信号の振幅と位相の両方を調節する少なくとも位相・振幅対である。あるいは、重み付けされた信号を指向性アンテナ106および108から送信することができる。図19は、アンテナをアップ・リンク受信モードでもダウン・リンク送信モードでも使用できるようにアンテナを二重化するように重み付け増幅器とそれぞれのアンテナとの間に結合されたダイプレクサ16Dおよび18Dを示している。しかし、アップ・リンク信号を受信するのに別個の基地局アンテナを用いてよい。
【００８６】
好ましい変形形態では、１本のアンテナは、それに対応する重みが1+j0（または振幅=1、位相=0°）に設定された基準として用いられる。他方の重みは、基準重みに対して決定される。一般に、基地局210は、各々がアンテナ、ダイプレクサ、重み付け増幅器、およびすべての関連するエンコーダを有する２つ以上のチャネルを使用することができる。Ｍが送信側アンテナの数である場合、求める必要のあるのは差分情報（すなわち、重み）だけであるので、求めなければならない重みの数はM-1である。以下の説明では、一般性を失わずに、１つの複素数重みを求めるだけで済むように２本の送信側アンテナ（M=2）に焦点を当てる。
【００８７】
図19において、遠隔局230は、遠隔局アンテナ232と、ダイプレクサ233を通して遠隔局アンテナ232に結合された遠隔局受信機234と、信号測定回路238と、プロセッサ240とを含んでいる。受信機234は、遠隔局230が第１および第２の送信アンテナから第１および第２の信号を受信するための回路を構成する。信号測定回路238およびプロセッサ240および本明細書で説明する制御モジュールは、遠隔局230が、受信された第１および第２の信号に基づいてチャネル状態情報を得、チャネル状態情報を複数のチャネル状態情報セグメントに区分するための回路を構成する。信号測定回路238は、複数の直交アンテナのそれぞれから受信された信号強度（および位相）を測定し、プロセッサ240はチャネル状態情報を得る。信号測定回路238は、受信された瞬間信号強度（および位相）を測定し、あるいは他の変形態様では、基準時間に、平均された受信信号強度および位相を測定する。
【００８８】
プロセッサは、信号測定回路238から与えられた情報からチャネル状態情報を得る。プロセッサは、それぞれの異なるアンテナから受信された信号から基準信号を選択する。プロセッサは、複数のアンテナのそれぞれについて、信号測定回路238によって求められた受信信号強度（および位相）を選択された基準信号強度（および位相）で割る。この比は、複素数（すなわち位相／振幅対）の比として求められる。基準アンテナに関する比は、定義により1+j0である。アンテナが２本である場合、送信すべき比は１つだけであり、すなわち、基準アンテナの比は一定の基準である。
【００８９】
プロセッサ240は、正規化された比からチャネル状態情報を得る。各比は、振幅と角度情報の両方を含む。このプロセスの目的は、２本のアンテナ（またはそれよりも多くのアンテナ）から送信された信号の位相を、それらが遠隔局230で建設的に補強されるように調整する。建設的な補強を確実に行うには、各アンテナから送信される信号を基準アンテナに対して位相を遅れさせるかまたは進めることが望ましい。たとえば、第１のアンテナ16が基準アンテナである場合、第２のアンテナ18から受信される信号に関する比の角度部分がさらに調べられる。この角度が基準アンテナに対して45°進んでいる場合、遠隔局230で建設的な補強を実現するには第２のアンテナ18の送信機で45°の遅延を導入する必要がある。したがって、プロセッサ240は、初期送信信号の位相に所望の追加の遅延を足し、この足し算の結果が360より大きい場合には360を引くことによって、遠隔局230で建設的な補強を実現するのに必要な位相の遅れまたは進みの量を求める。この位相角は次いで、チャネル状態情報の一部として送信される位相角になる。
【００９０】
プロセッサ240は、チャネル状態情報の振幅部分も求める。ここでの目的は、アンテナから遠隔局230までの最良の経路（すなわち、減衰が最も弱いパス）を有するアンテナを強調することである。すべてのアンテナから送信される総パワーは、ここでは一定とみなしてよい。チャネル状態情報の振幅部分によって解決すべき問題は、総送信パワーをどのように分割すべきかである。
【００９１】
プロセッサ240は、このために、各アンテナごとに、受信されたパワーを基準信号で受信されたパワーで割った比を算出することによってチャネル利得（減衰の逆数）を測定する。受信されたパワーは、信号測定回路238によって測定された信号強度の二乗である（すなわち、P_i = (a_i)²。この式で、a_iはアンテナｉからの信号強度である）。それぞれの異なるアンテナまたはアンテナ・ビームを介して送信される信号は、信号パワーP_TXで送信される信号上に変調されたこの信号固有の互いに直交するパイロット符号を含んでいる。遠隔局は、複合チャネル・インパルス応答、すなわちH_i = a_iexp(φ_i)を、受信された信号を受信された基準信号で割った比として測定する。この場合、φ_iは、測定中の信号の相対位相であり、a_iは相対信号強度である。次いで、P_iがa_iの二乗として求められる。各チャネルごとの相対チャネル応答は、受信パワーに関して測定される。アップ・リンク発信チャネルにおいて振幅フィードバック情報用に１ビットのみが予約されている場合、このビットは好ましくは、減衰が最も弱いパスを有するアンテナによって遠隔局230に総パワーの80%を送信し、減衰が最も激しいパスを有するアンテナによって総パワーの20%を送信するよう命令する。
【００９２】
アップ・リンク発信チャネルにおいて振幅フィードバック情報用に２ビットが予約されている場合、これらのビットは４つの振幅状態を定義することができる。たとえば、プロセッサ240は、アンテナ16からのパス減衰とアンテナ18からのパス減衰との比を算出し、次いで、この比がとることのできる所定の範囲の値に従ってこの比をスライスする。このスライシング・プロセスでは、４つのサブレンジが定義され、算出された比が４つのレンジのうちのどれに適合するかが識別される。各サブレンジは、２本のアンテナ16および18によって送信された総パワーの所望の分割を、たとえば、それぞれ85%/15%、60%/40%、40%/60%、および15%/85%と定義する。したがって、この２ビットは、これらの分割のうちの１つを、２本のアンテナによって送信された総パワーにおける所望の分割として符号化する。
【００９３】
当業者には、これらの教示を考慮すれば、チャネル状態情報の振幅部分を様々な手段によって算出できることが理解されよう。本明細書ではテーブル参照手段について説明するが、送信すべき総パワーの分割を算出する他の手段も同等である。パワー分割を定義するのに３ビット以上を用いてよいことが理解されよう。
【００９４】
プロセッサ240はまた、チャネル状態情報（上述の振幅部分および位相角部分を含む）を構成に基づいて複数のチャネル状態情報セグメントに区分する。遠隔局230は、複数のチャネル状態情報セグメントを基地局210に送信する送信機242をさらに含んでいる。
【００９５】
送信すべきチャネル状態情報は、位相・振幅情報の形の複素係数であり、アップ・リンク発信チャネルにおける対応するスロットで搬送されるいくつかのセグメント（Ｎ個のセグメント）で遠隔局230から基地局210に送信する必要がある。Ｎ個のスロットのN1およびN2（N = N1 + N2）への区画は、最初のN1個のスロットが位相情報を搬送し、残りのN2個のスロットが振幅情報を搬送するように行われる。基本的に、N1およびN2は任意に選択できるが、これらのパラメータの共通の値はN1 = N2 = N/2であってよい。各スロットが対応する情報セグメントの搬送用にＫビットを予約すると仮定する。位相は次式の精度で分解することができる。
【００９６】
【数３】

振幅は次式の精度で分解することができる。
【００９７】
【数４】

上式で、A_maxは最大振幅である。
【００９８】
たとえば、スロットの数Ｎが６であり、N1およびN2のそれぞれに３つのスロットが予約されていると仮定する。スロット当たりビット数Ｋが１であると仮定し、最大振幅A_maxが3 Vであると仮定する。そうすると、位相および振幅の精度はφ_min = 45°であり、振幅A_minは0.375 Vである。しかし、スロット当たりビット数Ｋを２に増やした場合、送信できる位相および振幅の精度はφ_min = 5.6°であり、振幅A_minは0.05Vである。
【００９９】
一般に、厳密なチャネル状態情報の量子化形態または打切り形態は、打切り形態の各ビットがアップ・リンク発信チャネルで利用可能なビットの数に厳密に一致するように形成される。打切り態様は位相セグメントφ_i（i = 1からN1）に区分され、各セグメントは、最上位ビット（MSB）が第１のセグメントで送信され、最下位ビット（LSB）が最後のセグメントで送信されるように階層順に送信される。同様に、各振幅セグメントA_i（i = 1からN2）は、次のチャネル状態情報（比）の量子化セグメントまたは打切りセグメントを含んでおり、かつ各振幅セグメントは階層順に送信される。
【０１００】
本発明の本態様では、ダウン・リンク・ビームを形成する際に用いられる位相角精度および振幅精度が改善されるため、移動通信のダウン・リンク性能が向上する。この態様は、低可動性環境に特に適しており、かつ室内環境および歩行者環境における高ビット・レート用途に適している。この態様は、ラップトップ・コンピュータ用の高ビット・レート無線データ用途に特に適している。
【０１０１】
たとえば、遠隔局が毎秒1m（毎時3.6km）の速度ｖで移動しており、搬送波周波数が２ギガヘルツ（λ=0.15m）であると仮定する。最大ドップラー周波数f_Dはv/λであり、チャネル・コヒーレンシ時間T_Cは次式のように算出される。
【０１０２】
【数５】
T_C = 1/(2f_D) = λ/(2v) = 75ミリ秒
チャネル状態情報は、TC/10に等しい期間にわたって安定である（ほぼ一定）と仮定することができ、したがって、チャネル状態情報をこの7.5ミリ秒の安定した期間の間に遠隔局230から基地局210に送信することができる。広帯域CDMA（WCDMA）標準で、スロット持続時間は0.625ミリ秒と定められているので、12個のスロットを用いてチャネル状態情報を基地局に送り返すことができる。
【０１０３】
チャネル状態情報をアップ・リンク・スロットに詰める方法はいくつかある。表１は、スロット当たり１ビットのみ（K=1）に基づく例を示している。表１において、位相角情報と振幅情報の両方に３ビット精度が用いられている。位相角は最初の６つのスロットで送信され、振幅情報は最後の６つのスロットで送信される。どちらの場合も、最上位ビットが最初に送信される。スロット１では、３ビット位相角の最上位ビットが送信される。スロット２では、信頼性を向上させるために同じビットが繰り返される。その後、残りの位相角が送信される。振幅情報ビットも同様に送信される。第１のビットは、１ビットの場合と同様に位相角を180°の精度で示す。スロット３の後で、位相角が、２ビットの場合と同様に90°の精度で送信され、スロット５の後で、位相角が、３ビットの場合と同様に45°の精度で送信される。位相角がチャネルのコヒーレンシ時間の間に約360°変化すると仮定した場合、上記の例では、位相角は、12個のスロットを送信するのに必要な7.5ミリ秒の期間に約36°変化する。これは、３ビット・データ（45°）で利用可能な位相精度に相当する。
【０１０４】
スロット７の後で、振幅情報が、１ビットの場合と同様に最大振幅の0.5の精度で送信される。スロット９の後で、振幅情報が、２ビットの場合と同様に最大振幅の0.25の精度で送信され、スロット11の後で、振幅情報が、３ビットの場合と同様に最大振幅の0.125の精度で送信される。
【０１０５】
【表１】
チャネル状態情報を基地局に送信するためのフォーマット

【０１０６】
一般に、位相情報は振幅情報よりも重要である。最適な最大比合成は、振幅情報フィードバックがない場合に使用される等利得合成よりも約1dB優れているに過ぎず、したがって、位相ビット（N1）への割当てを増やし振幅ビット（N2）への割当てを減らすと有利である。たとえば、フィードバック・チャネル状態情報を3.125ミリ秒の冗長性なしにWCDMAフォーマットで送信できるように３位相ビットおよび２振幅ビットを割り当てることができる。
【０１０７】
許容フィードバック能力（たとえば、１ビット／スロット以上）とフィードバック信頼性（たとえば、反復ビットまたは冗長ビット）とフィードバック精度（たとえば、位相角ビットおよび振幅ビットの数）との兼ね合せは用途および環境に応じて決まる。たとえば、冗長性エラー検査を行えるように公知のSECDED（信号エラー訂正、２重エラー検出）フォーマットの３ビット検査符号を８ビットの情報に付加することができる。当業者には、これらの教示を考慮して、フィードバック能力、フィードバック信頼性、およびフィードバック精度を用途および環境にどのように整合すべきかが理解されよう。
【０１０８】
プロセッサ240（図19）は、チャネル状態情報をシステム・モードによって定められるフォーマットに従って複数のチャネル状態情報セグメントに区分する。実際には、システムは、各々が異なるフォーマットを定める複数のモードを備えてよい。たとえば、１つのモードにより、互いに等しい振幅を命令する位相角情報のみを各アンテナに送信し、他のモードにより、３ビットの位相角情報および１ビットの振幅情報を送信することができる。次いで、送信機242は、アップ・リンク発信チャネルにおける複数のチャネル状態情報を符号化し、符号化された情報をダイプレクサ233およびアンテナ232を通して基地局210に送信する。
【０１０９】
この態様の一変形態様では、アップ・リンク発信チャネルにおけるチャネル状態情報を表すのに１ビットからたとえば20ビットを必要とするいくつかのモードがある。この変形態様では、プロセッサ240は、各更新間の変化に基づいて、チャネル状態情報が変化する速度を求める。この速度が遅く、遠隔局が低速で移動しているか、または静止していることを示しているときは、それに応じてフィードバック・モードが、チャネル状態情報のより多くのデータ・ビットを基地局に送信できるようにするモードに変更される。しかし、チャネル状態情報が急速に変化し、遠隔局が高速に移動していることを示すときは、それに応じてフィードバック・モードが、チャネル状態情報更新のたびに送信されるビットを少なくするモードに変更される。
【０１１０】
基地局210は、アップ・リンク発信チャネルで符号化された情報を受信し、受信機／検出器220で複数のチャネル状態情報セグメントを復号する。次いで、プロセッサ220Pは、受信された複数のチャネル状態情報セグメントからチャネル状態情報を再構成し、重みW1およびW2を生成する。重みW1およびW2は、再構成されたチャネル状態情報に基づいて、それぞれ第１および第２のアンテナ16および18に送られる第１および第２の供給信号CH1およびCH2を重み付けするように、それぞれの増幅器102および104に供給される。
【０１１１】
この態様の２つの変形態様はプロセッサ220Pで実現することができる。第１に、プロセッサは、増幅器102および104に加えられる重みW1およびW2を形成する前に、すべてのセグメントを収集して総チャネル状態情報を再構成することができる。あるいは、チャネル状態情報として、基地局にまず位相角が送信され、位相角セグメント内ではまず最上位ビットが送信される。W1およびW2の値は、増幅器102および104へのフィードバックをより早く行うために各ビットが受信されるたびにプロセッサ内で更新することができる。これによって、事実上より大きなフィードバック帯域幅が得られる。
【０１１２】
図20において、プロセッサ240上で実施される方法は、通常ソフトウェア・モジュールおよび／または論理を有するプロセッサで実施されるいくつかの段階を含む。しかし、当業者には、ASICまたは他のカスタム回路を用いたプロセッサでこれらの段階を実施できることが理解されよう。
【０１１３】
段階S2で、プロセッサは、複数のアンテナのそれぞれについて、信号測定回路238によって求められた受信信号強度および位相（複素数）を受信する。段階S4で、プロセッサは、受信された信号のうちの１つを基準信号として選択する。この選択は任意であってよく、あるいは最大の位相遅れを有する信号（遅れさせる必要がある可能性が最も低い）を選択してもよい。段階S6で、プロセッサは、信号測定回路238によって求められた受信信号強度および位相（複素数）を、受信された基準信号強度および位相（複素数）で割る。基準アンテナに関する比は、定義により1+j0である。アンテナが２本である場合、求めて送信すべき比は１つだけであり、基準アンテナの比は一定の基準である。
【０１１４】
段階S8（図20）で、プロセッサ240は、遠隔局230で建設的な補強を行うために各送信側アンテナで必要とされる位相の遅れまたは進みの量を求める。最大の遅れを持つ信号として基準信号を選択する場合、残りの信号はアンテナにおいて遅延を加えることによって基準信号との位相合わせを行うことができる。段階S8で、必要な追加の遅延が求められるが、非基準信号の位相に追加の位相遅延を足した位相が360°よりも大きくなった場合は360が引かれる。次いで、この位相角は、チャネル状態情報の一部として送信される位相角になる。当業者には、これらの教示を考慮して、アップ・リンク発信チャネルでチャネル・インパルス応答の位相角だけを送信すればよくなるように段階S8を基地局で実行できることが理解されよう。
【０１１５】
段階S10で、送信分布（各送信アンテナ間での総パワーの割当て）を定めるためのパワー管理情報が得られる。当業者には、これらの教示を考慮して、チャネル状態情報の振幅部を様々な手段によって計算できることが理解されよう。本明細書では、テーブル参照手段について説明するが、送信すべき総パワーの分割を算出する他の手段も同等である。
【０１１６】
たとえば、各アンテナからの信号の相対振幅および相対位相を、基地局でさらに処理できるようにアップ・リンク発信チャネルで送信することができる。あるいは、遠隔局は、段階S10で所望のパワー分布を示す信号を生成することができる。アップ・リンク発信チャネルで振幅フィードバック情報用に１ビットしか予約されていない場合、このビットは好ましくは、減衰が最も弱いパスを有するアンテナから遠隔局230に総パワーの80%を送信し、減衰が最も激しいパスを有するアンテナから総パワーの20%を送信するよう命令する。アップ・リンク発信チャネルで振幅フィードバック情報用に２ビットが予約されている場合、これらのビットは４つの振幅サブレンジを定義することができる。たとえば、それぞれ85%/15%、60%/40%、40%/60%、および15%/85%を定義することができる。したがって、この２ビットは、これらのサブレンジの一方を、２つのアンテナによって送信される総パワーにおける所望の分割として符号化する。当業者には、より多くのアンテナへの拡張、またはチャネル状態情報の振幅部分を表すのにより多くのビットを用いるようにする拡張が明らかであろう。テーブル参照またはその他の手段の厳密な性質は、アップ・リンク・フォーマットにおいてチャネル状態情報の振幅部分を保持するために予約されるビットの数に依存する。
【０１１７】
段階S12で、チャネル状態情報が区分され、本明細書で説明するフォーマット（たとえば、表１）に変更される。段階S14で、各セグメントは順次、アップ・リンク発信チャネルで基地局に送信される。そこから、アンテナに対するそれぞれの重みが回復され、増幅器102および104（図19）に加えられる。
【０１１８】
アップ・リンク通信およびダウン・リンク通信が様々な周波数を介して行われる周波数分割二重化システムでは、２つの方向がそれぞれの異なる周波数に基づく方向であるため、アップ・リンク情報からダウン・リンク・チャネル状態を厳密に求めることは不可能である。本システムは、ダウン・リンク・データからダウン・リンク・チャネル状態を測定し、次いで、送信されるダウン・リンク信号の振幅および位相を調整するコマンドをアップ・リンク発信チャネルで送信するという利点を有している。
【０１１９】
図21において、基地局のアンテナ１は、セクタ・カバレージ型のアンテナである。アンテナ１は信号を直接パス３上で遠隔局２に送信する。しかし、別のマルチパス信号が電波散乱体４から反射され、マルチパス５上を進む。その結果、遠隔局２は、この信号の２つのレプリカを互いにわずかに異なる時間に受信する。図22において、２つのレプリカは、時間nTおよび時間nT+τで受信される信号として示されており、この場合、τは、直接パス３に対するマルチパス５の追加の長さによって起こる追加の時間遅延である。マルチパス遅延は、２つのパス上で受信される２つの信号間に破壊的な干渉を引き起こすような遅延であってよい。他の電波散乱体によってさらに多くのマルチパス信号を生成されることがある。
【０１２０】
従来のレーキ受信機は、ローカル信号（たとえば、CDMA信号の拡散符号）と、それぞれの異なる遅延を用いて受信された信号のレプリカを含む受信信号とを相関付ける。正しい遅延を用いた場合、各信号は、エネルギーを補強するようにコヒーレントに合成される。ローカル信号（たとえば、所望の拡散符号）が所望の信号路からの信号と相関付けされると、ローカル信号は他のあらゆる信号レプリカ（たとえば、それぞれの異なる遅延を有する信号路からの信号レプリカ）とも相関付けされる。他の信号レプリカとの相関に対応する項は不要な項であり、システムの性能を劣化させる傾向がある。不要な相関項は、それぞれの異なる符号を有する様々なユーザ間の直交性を失わせ、その結果、同一チャネル・ユーザが互いに干渉し始める。この劣化効果は、高ビット・レート・リンクで通常用いられる短い拡散符号の場合に顕著になる。
【０１２１】
本発明は、レーキ受信機を従来とは異なるように動作させる。本発明は、ビーム・フォーミングを用いて、それぞれの異なる信号路を分離し、すべての信号レプリカが受信機に同時に到来するように各信号レプリカ（たとえば、各ビーム）に事前送信時間ずれ補償を施す。このようにして、受信機は、実際には複数のパス（たとえば、図１のパス３および５）上で複数の信号を受信しコヒーレントに合成するにもかかわらず、１タップ・チャネルによってのみ処理された信号を受信するように見える。これによって、直交性が失われるのが避けられ、他の場合にはシステム性能を劣化させる恐れのある相互相関項を最小限に抑えるかまたは無くす。
【０１２２】
本発明の態様において、所望のデータは２つ以上の空間−時間符号化信号に含められる。各信号は、固有の互いに直交する署名符号によって識別される。空間−時間符号化信号の１つが他の空間−時間符号化信号に対して著しく遅延した場合、署名符号の直交性が低下することがある。最も短いパスの信号を、より長いパスの信号が遠隔局２に到来するのと同時に遠隔局２に到来するように遅延させることが好ましい。
【０１２３】
図23において、例示的なシステムはアンテナ１および遠隔局２を含んでいる。例示的なアンテナ１は、バトラー・マトリックス・マルチビーム・アンテナ・アレイまたは他の任意のマルチビーム・アンテナ・アレイであってよい。この例の所望のデータは、２つの空間−時間符号化信号I2およびI5として符号化される。空間−時間符号化信号I2およびI5はそれぞれビームD2およびD5で送信される。ビームD5は信号I5を直接パス３上で遠隔局２に送信する。ビームD2は信号I2を間接マルチパス５上で遠隔局２に送信する。
【０１２４】
図26に、空間−時間符号化信号I2およびI5を生成する例示的なエンコーダが示されている。図26は図２に類似している。ただし、図２のアンテナ16および18は図23のマルチビーム・アンテナで置き換えられており、プログラム可能な遅延線（たとえば、選択可能なマルチタップ遅延線）がマルチプライヤ14とマルチビーム・アンテナとの間に結合されている。マルチプライヤ12は、マルチプライヤ14によって信号CH2に符号化される署名符号と互いに直交する署名符号（OC）を有する信号CH1を符号化する。署名符号は、様々な形態で直交するトレーニング系列、パイロット符号、または拡散系列であってよい。遠隔局２は、これらの署名符号が互いに直交するかぎり、これらの署名符号を用いて、ビームD2から間接パスで受信された信号から、ビームD5から直接パスで受信された信号を分離する。当業者には、図23および図26に示されている２つのビームおよび対応する空間−時間符号化信号を２つよりも多くのビームに一般化することができ、すべての信号の時間同期をとるために追加のプログラム可能な遅延線が必要になることがあることが理解されよう。
【０１２５】
図24および図25に示されているように、ビームD5からの直接信号は、ビームD2からの間接信号が受信されるよりも時間τだけ早く遠隔局２で受信される。署名符号間に最良の直交性を維持するには、各信号を時間的に揃えることが望ましい。受信機（後述のように場合によっては基地局の受信機であり、場合によっては遠隔局２の受信機である）は、各信号を揃えるのに必要な時間遅延τを求める。遠隔局２で最後に受信される信号（たとえば、信号I2）を基準空間−時間符号化信号とみなすことができる。次に、残りの信号を少なくとも１つの残りの空間−時間符号化信号（たとえば、信号I5）とみなすことができる。この態様では、少なくとも１つの残りの空間−時間符号化信号は、送信される前に基地局のプログラム可能な遅延線（図26参照）で遅延させられる。各信号は、少なくとも１つの残りの空間−時間符号化信号が、遠隔局で受信されるときに基準信号と時間的に揃うように十分な遅延によって遅延させられる。図23に示す例では、遠隔局２で最後に受信される信号は、マルチパス５が長いために信号I2である。信号I5は、信号I2が遠隔局２に到来するのと同時に遠隔局２に到来するように遅延させる必要がある。
【０１２６】
空間−時間ダイバーシチ技術（図２）でもビーム−空間ダイバーシチ（図23）でも、上述のように遠隔受信機が信号CH1およびCH2を分離することが重要である。これは、様々な形式の直交署名符号を用いることによって行われる。２つのパス、すなわち直接パス３およびマルチパス５からの信号が遠隔局２に到来する到来時間の差を遅延拡散と呼ぶ。遅延拡散が存在しないか、または最小限であるとき、署名符号の直交性が保たれる。しかし、署名符号のかなりの遅延拡散が存在する周波数選択チャネルでは、各チャネル間の直交性が失われることがあり、遠隔局２がそれぞれのチャネルで搬送される信号を分離することは困難になる。たいていの一般的な符号化系列は、各署名符号間の所与の位相関係についてのみ小さい値かまたは零を有し、他の位相関係については非零を有する非理想的な相互相関関数（CCF）を特徴とする。
【０１２７】
マルチパス・チャネル上で遠隔局２に送信される複数の空間−時間ダイバーシチ信号はそれぞれの異なる遅延を受ける。位相がずれる所与の位置でのCCFの値は通常、非零であり、位置ごとに異なるので、無線チャネルによって課されるそれぞれの異なるパス遅延の、送信信号に対する効果として、遠隔局２が各信号を分離するのに用いる署名符号間の直交性が低下する。直交性が失われると、他の場合には無線通信システムにおける基地局と遠隔局との間の信号の空間−時間符号送信によって得られるダイバーシチ利得が低下する。
【０１２８】
本態様において、基地局に関連するマルチビーム・アンテナ・アレイは、マルチビーム・アンテナ・アレイの複数のビームのそれぞれで関心対象の遠隔局からアップ・リンク信号を受信する。アップ・リンク信号は、パイロット信号でも、アップ・リンク発信チャネルでも、信号の送信元を関心対象の遠隔局として識別する他の任意のアップ・リンク・チャネルであってもよい。アップ・リンク信号は、マルチビーム・アンテナ・アレイの対応する複数のビームで受信される無線信号から導かれる複数の信号として受信される。
【０１２９】
基地局の受信機は、複数の受信信号のそれぞれについて、署名符号により関心対象の特定の遠隔局から発信された信号成分として識別される信号成分を分離する。複数のビームの各ビームの受信信号成分は、基準ビームの信号成分に対する特定の時間遅延または遅延拡散における関心対象の特定の遠隔局の識別された信号のレプリカを含む。基地局の受信機は、それぞれのビームからの複数の信号成分を処理し、最後に受信された信号成分と、それぞれのビームから受信された他の信号成分のそれぞれを、基準ビームで受信された信号成分と揃えるのに必要な遅延拡散とを含むビームを基準ビームとして識別する。基地局が複数の遠隔局に対処するとき、このプロセスは、各遠隔局または選択された遠隔局に対して繰り返すことができる。選択される遠隔局は、高い送信パワーを有する遠隔局であってよい。高い送信パワーは、たとえば高データ・レート要件によって必要とされることがある。
【０１３０】
図27は、図23に示す８ビーム基地局システムと同様の16ビーム基地局システムの代表的なチャネル・インパルス応答すなわち遅延分布プロファイル300を示している。基地局は、マルチビーム・アンテナの各ビームに関連する遅延拡散τを測定する。しきい値を超えた信号強度を有する受信信号の場合、「ｘ」は、瞬間的なかつ／または平均された信号強度が所与のしきい値を超えていることを示す。それぞれ304、306、および308に示された方向D3、D6、およびD12は、（たとえば、遅延τ₄からτ₆にわたる）最小遅延拡散を有する信号を含む。いくつかの潜在的な方向が利用可能である場合、生じた干渉の白化、基地局によって用いられる複数のパワー増幅器におけるパワーの均等な分散、平均を超えた干渉が同一チャネル・ユーザに対して起こる可能性がある方向の回避のような追加の基準に基づいて、利用可能な方向のうちの好ましい方向が選択される。たとえば、高パワー・ビームによって照明される領域内に１人または複数の低ビット・レート・ユーザが位置している場合、高パワー・ビームによってこれらの低ビット・レート・ユーザに干渉が起こる可能性がある。好ましい状況によっては、より有効な干渉白化を実現するためにビーム・ホッピングを施してもよい。
【０１３１】
動作時には、基地局は、最小遅延拡散を有する方向を選択する。たとえば、基地局は、少なくとも２つのビームのうちの対応するビームで少なくとも２つの空間−時間符号化信号を送信するために、マルチビーム・アンテナ・アレイによって形成できる複数のビームのうちの少なくとも２つのビームを選択する。少なくとも２つのビームは、基準ビームおよび少なくとも１つの残りのビームを含む。基地局はまた、プログラム可能な遅延線をプログラムする際に用いられる、少なくとも１つの残りのビームの各ビームに対応する時間遅延を、遅延分布プロファイル300から求める。
【０１３２】
基地局は、基準空間−時間符号化信号および少なくとも１つの残りの空間−時間符号化信号を形成するように、少なくとも２つの空間−時間符号化信号の各信号を、少なくとも２つの空間−時間符号化信号に符号化された他の各署名符号と互いに直交する署名符号で符号化する（図26の12および14を参照されたい）。図23の例では、基準空間−時間符号化信号を信号I2とみなすことができ、少なくとも１つの残りの空間−時間符号化信号を信号I5とみなすことができる。しかし、当業者には、これらの教示を考慮して、本態様を２つよりも多くの空間−時間符号化信号に拡張するにはどうすべきかが理解されよう。
【０１３３】
基地局は、少なくとも１つの残りの空間−時間符号化信号の各信号を遅延させ、少なくとも１つの遅延空間−時間符号化信号（たとえば、図26の信号I5）を形成する。基地局は次いで、基準空間−時間符号化信号（たとえば、信号I2）と少なくとも１つの遅延空間−時間符号化信号（たとえば、信号I5）の両方が遠隔局２に同時に到来するように、少なくとも２つのビームの各ビームで基準空間−時間符号化信号および少なくとも１つの遅延空間−時間符号化信号を送信する。
【０１３４】
本態様は、遠隔局から基地局へのフィードバック・チャネルには依存しない。その代わりに、通常の発信信号のアップ・リンク測定値のみから基地局によって送信方向が選択される。アップ・リンク・チャネル応答を長期間にわたって平均して高速フェージングを軽減することによって、ダウン・リンク・チャネル応答のパワー応答を推定することができる。示されるアップ・リンク・チャネルとダウン・リンク・チャネルはパワーの点で相反する。
【０１３５】
しかし、周波数分割二重化（FDD）システムでは、フィードバック測定により、複雑さが増すことを犠牲にして結果を向上させることができる。アップ・リンク通信およびダウン・リンク通信が様々な周波数を介して行われる周波数分割二重化システムでは、２つの方向がそれぞれの異なる周波数に基づく方向であるため、アップ・リンク情報からダウン・リンク・チャネル状態を厳密に求めることは不可能である。
【０１３６】
上述の態様では、基地局がダウン・リンク・チャネル応答の代わりにアップ・リンク・チャネル応答を測定する態様について説明した。完全なダウン・リンク・チャネル・インパルス応答を得るには、ダウン・リンク・チャネルを直接測定し、ダウン・リンク・チャネル状態をフィードバック・チャネルで、測定を行う遠隔局から、測定値（たとえば、遅延分布プロファイル300）を必要とする基地局に送信する必要がある。
【０１３７】
遠隔局は、基地局における方向選択および遅延に必要な計算を行うのではなく、これらの機能に関与するか、またはこれらの機能を実行する。合意された標準信号が、互いに直交するパイロット系列またはトレーニング系列や拡散符号のような、各ビームに符号化された識別子すなわち署名符号と共に、基地局からすべての遠隔局に送信される。遠隔局は次いで、チャネル・インパルス応答（たとえば、遅延分布プロファイル300）を測定し、送信の好ましい方向および遅延を基地局に知らせる。
【０１３８】
当業者には、これらの教示を考慮して、チャネル性能を２段階プロセスで測定することができる。第１の段階で、基地局は、アップ・リンク・チャネルのインパルス応答の推定値を求め、この推定値をダウン・リンク・チャネルのインパルス応答の代わりに使用する。次いで、基地局は、第１の推定プロセスによって示される遅延を少なくとも１つの残りの空間−時間符号化信号に加える。
【０１３９】
第２の段階で、ダウン・リンク・チャネルが直接測定される。合意された標準信号が、互いに直交するパイロット系列またはトレーニング系列や拡散符号のような、各ビームに符号化された識別子すなわち署名符号と共に、基地局からすべての遠隔局に送信される。遠隔局は次いで、チャネル・インパルス応答（たとえば、遅延分布プロファイル300）を測定し、送信の好ましい方向および遅延をフィードバック・チャネル上で基地局に知らせる。
【０１４０】
図28において、セット・アップ・プロセスS200によってアップ・リンク・チャネル応答が測定され、測定された遅延が、ダウン・リンク・チャネル送信を制御するように設定される。プロセスS200は、チャネル応答を測定する段階S202と、使用すべきビームを選択する段階S204と、選択されたビームに対する時間遅延を求める段階S206と、基地局の可変遅延線（図26参照）を、求められた遅延を課すように構成する段階S208とを含む。可変遅延線は、素子間に複数のタップが配置された一定遅延素子のシーケンスで構成することができる。遅延線は、スイッチを用いて様々なタップを出力として選択することによって変化させられる。段階S204で、基地局は、基地局に関連するマルチビーム・アンテナ・アレイによって形成された複数のビームのうちの少なくとも２つのビームを選択する（ただし、図23および図26には２つのビームしか示されていない）。これらのビームにおいて、空間−時間エンコーダによって生成された対応する少なくとも２つの空間−時間符号化信号が送信される（ただし、図23および図26には２つのビームしか示されていない）。少なくとも２つのビームは、基準ビームおよび少なくとも１つの残りのビームを含む。段階S206で、基地局は、少なくとも１つの残りのビームの各ビームに対応する時間遅延を求める。段階S208で、基地局は、少なくとも１つの残りのビームの各ビームに対応する時間遅延を可変遅延線に設定する。各可変遅延線は、マルチビーム・アンテナ・アレイと空間−時間エンコーダとの間に結合されている（図26参照）。
【０１４１】
図29において、タイム・アライン・プロセスS220により、段階S222で、選択された各ビームの空間−時間符号化信号が、すべての他のビームに直交する署名符号でマーク付けされ、段階S224で、選択されたビームが、求められた遅延拡散に従って送信され、段階S226で、遅延した信号が基地局に送信される。段階S222で、基地局は、基準空間−時間符号化信号および少なくとも１つの残りの空間−時間符号化信号を形成するように、少なくとも２つの空間−時間符号化信号の各信号を、少なくとも２つの空間−時間符号化信号に符号化された他の各署名符号と互いに直交する署名符号で符号化する。段階S224で、基地局は、少なくとも１つの残りの空間−時間符号化信号の各信号をそれぞれの可変遅延線で遅延させ、少なくとも１つの遅延空間−時間符号化信号を形成する。段階S226で、基地局は、基準空間−時間符号化信号および少なくとも１つの遅延空間−時間符号化信号を少なくとも２つのビームの各ビームで送信する。
【０１４２】
図30において、フィードバック・プロセスS240を用いる遠隔局は、ダウン・リンク複合チャネル状態情報を測定し、この情報を基地局に送り返す。 プロセスS240は、基地局に関連するアンテナ・システムから少なくとも２つの識別子署名（たとえば、それぞれの異なるパイロット信号）を受信する段階S242と、受信された信号に基づいて複合チャネル状態情報を得る段階S244と、複合チャネル状態情報を複数のチャネル状態情報セグメントに区分する段階S246と、複数のチャネル状態情報セグメントを系列として基地局に送信する段階S248とを含む。セグメントの系列は、位相角の最下位ビットよりも前に位相角の最上位ビットを送信する。セグメントの系列は、振幅の最下位ビットよりも前に振幅の最上位ビットを送信する。セグメントの系列は、位相角のビットを、同じレベルの有意性を持つ対応する振幅ビットよりも前に送信する。チャネル・インパルス応答測定値をフィードバックするために、各ビーム（またはアンテナ）を、他のすべてのパイロット署名に直交する固有のパイロット署名に関連付ける必要があることに留意されたい。
【０１４３】
当業者には、これらの教示を考慮して、電気回路、特殊な特定用途向け集積回路（ASIC）、あるいはソフトウェア・プログラムを実行するかまたはデータ・テーブルを使用するコンピュータまたはプロセッサで様々なシステム構成要素を実現できることが理解されよう。たとえば、図４、図５、図11、または図12のエンコーダ10、マルチプライヤ12、14、および増幅器102、104は、性能要件に応じて回路またはASIC、あるいは場合によってソフトウェア制御プロセッサで実現してよい。図11のビーム・フォーマ40は通常、回路またはASICで実現され、変換器101、103およびマルチプライヤ105、107は通常、回路またはASICで実現されるが、ソフトウェア制御プロセッサで実現してもよい。図14の様々な基地局構成要素212、214、216、218、220、および222ならびに様々な遠隔局構成要素232、234、238、240、および242は、回路またはASICで実現されるが、ソフトウェア制御プロセッサで実現してもよい。図19の様々な基地局構成要素16D、18D、102、104、220、および220Pならびに様々な遠隔局構成要素232、233、234、238、240、および242は、回路またはASICで実現されるが、ソフトウェア制御プロセッサで実現してもよい。当業者には、本明細書で説明した様々な機能を、性能要件に応じて回路、ASIC、またはソフトウェア制御プロセッサで実現できることが理解されよう。
【０１４４】
ダウン・リンク性能を向上させる新規の閉ループ・フィードバック・システムの（例示的なものであり、制限的なものではない）好ましい態様について説明したが、当業者には、上記の教示を考慮して修正および変形を加えてよいことに留意されたい。添付の特許請求の範囲によって定義される本発明の範囲および趣旨内の変更を、開示された本発明の特定の態様に加えてよいことを理解されたい。
【０１４５】
したがって、特許法によって必要とされる詳細事項と共に本発明について説明したが、開封特許状によって保護される請求される所望のものは添付の請求の範囲に記載されている。
【図面の簡単な説明】
【図１】 本発明が使用される無線環境の概略図である。
【図２】 既知の基地局のブロック図である。
【図３】 既知の空間時間エンコーダのブロック図である。
【図４】 本発明の態様による基地局装置のブロック図である。
【図５】 本発明の他の態様による基地局装置のブロック図である。
【図６】 既知の６コーナ・アンテナ・システムの概略図である。
【図７】 既知のフェーズ・アレイ・アンテナの概略図である。
【図８】 例示的な３セクタ・アンテナ・システムの平面図における概略図である。
【図９】 既知の「バトラー・マトリックス」アンテナの概略図である。
【図１０】 ２重ビーム・フェーズ・アレイ・アンテナの概略図である。
【図１１】 本発明の他の態様による基地局装置のブロック図である。
【図１２】 本発明の他の態様によるTDMA基地局装置のブロック図である。
【図１３】 本発明による閉ループ・ビーム・パワー管理システムのブロック図である。
【図１４】 本発明による無線システムのブロック図である。
【図１５〜１７】 本発明による角パワー・スペクトルを求める方法のフローチャートである。
【図１８】 本発明によって受信されかつ／または算出された角パワー・スペクトルのグラフである。
【図１９】 本発明の態様のブロック図である。
【図２０】 本発明によるフィードバック制御方法のフローチャートである。
【図２１】 本発明によってセクタ・カバレージ・アンテナを用いて処理されたマルチパス信号を示す概略図である。
【図２２】 遠隔局によって受信された図21の直接・マルチパス信号を示すグラフである。
【図２３】 あるセクタをカバーするマルチビーム・アンテナを用いて本発明によって処理されたマルチパス信号を示す概略図である。
【図２４】 遠隔局によって受信された図21または図23の直接信号および直接信号の遅延されたレプリカを示すグラフである。
【図２５】 遠隔局によって受信された図21または図23のマルチパス信号を示すグラフである。
【図２６】 本発明の態様によるプログラム可能な遅延線を有する基地局装置のブロック図である。
【図２７】 本発明による遅延分散プロファイルを示すグラフである。
【図２８】 本発明によるセットアップ方法のフローチャートである。
【図２９】 本発明によるタイム・アライン方法のフローチャートである。
【図３０】 本発明によるフィードバック方法のフローチャートである。[0001]
Background of the Invention
Field of Invention
The present invention relates to a system for controlling downlink signal transmission from a base station to a remote station in a portable radio system. In particular, the present invention relates to a closed loop phase and amplitude control system that adjusts the phase and amplitude of a downlink transmission signal.
[0002]
Explanation of related technology
The mobile phone system is operated in an environment that causes multipath or reflection of the signal, particularly in an urban environment. In FIG. 1, a base station transmitter 1 broadcasts the signal along a direct path 3 to a remote station 2 (often a mobile station). However, due to the presence of the tall building 4, the transmitter 1 also broadcasts its signal along the indirect path 5 to the remote station 2, and therefore indirect to the remote station 2 and in the direction that the path 3 comes directly to the remote station 2. An angular spread AS is produced between the directions in which the path 5 arrives. The direct path 3 and the indirect path 5 are recombined at the remote station 2 and the signals overlap constructively and destructively, resulting in random or seemingly fading and blackout zones.
[0003]
In order to weaken the effects of multipath, known systems use space-time transmit diversity techniques. In FIG. 2, a known transmitter includes a space-time transmit diversity encoder 10, composite multipliers 12 and 14, and antennas 16 and 18. The space-time transmit diversity encoder 10 receives the input signal S_INTo process two channel signals CH₁And CH₂Get. Multipliers 12 and 14 have two channel signals CH₁And CH₂The same orthogonalization code OC is added to the two channels and the input signal S_INIdentify as a channel that contains information about. However, different orthogonal identifiers (for example, pilot sequences and training sequences) are added to various antenna signals so that the remote station can individually identify the signals from the two antennas. The multiplexed channel signals are transmitted on respective antennas 16 and 18 that are substantially separated by a distance (eg, 20 wavelengths). Such antennas arranged at intervals are referred to as diversity antennas. In a multipath environment, severe fading occurs when various propagation paths are destructively added at the receiving antenna. When diversity antenna is used, signal CH₁And signal CH₂Are less likely to cause deep fading. This is because the two signals are likely to propagate on different paths such as multipaths 3 and 5. The diversity antenna may be an omnidirectional antenna that is directed to an antenna sector having overlapping sectors. When diversity antennas are spaced sufficiently apart, these antennas can be considered orthogonal to each other because they propagate the signal in an uncorrelated channel (ie, path).
[0004]
Input signal S_INIs the two symbols S₁And S₂Are carried in time consecutively, i.e. the first symbol is carried in a symbol slot between 0 and T, and the second symbol is carried in a symbol slot between T and 2T. In FIG. 3, an exemplary encoder 10 uses a QPSK modulation technique and includes a time alignment register 20 and a holding register 22 that holds two symbols. Baseband carrier signal SBBC is inverted in inverter 24 to produce a negative baseband carrier-SBBC. The QPSK modulator 26 has the symbol S₁Is encoded onto the baseband carrier signal SBBC to generate a modulated first symbol, and the QPSK modulator 28₁Is encoded on the negative baseband carrier signal -SBBC to produce the modulated conjugate of the first symbol. The QPSK modulator 30 has the symbol S₂Is encoded onto the baseband carrier signal SBBC to generate a modulated second symbol, and the QPSK modulator 32 generates the symbol S₂Is encoded on the negative baseband carrier signal -SBBC to produce the modulated conjugate of the second symbol. The modulated conjugate of the second signal is inverted in inverter 34 to produce a negative modulated conjugate of the second symbol. The analog mux 36 has a signal on CH1 [S₁, -S₂ ^*In the first symbol time slot (ie, from 0 to T, FIG. 2), the modulated first symbol is switched to the first channel signal and the second symbol time slot During the slot (ie, from T to 2T, FIG. 2), the negative modulated conjugate of the second signal is switched to the first channel signal. The analog mux 38 has a signal on CH2 [S₂, S₁ ^*In the first symbol time slot (ie, from 0 to T, FIG. 2), the modulated second symbol is switched to the second channel signal, and the second symbol time slot During the slot (ie, from T to 2T, FIG. 2), the modulated conjugate of the first signal is switched to the second channel signal.
[0005]
In FIG. 2, the code OC is used as a CDMA spreading function that separates the two signals transmitted from antennas 16 and 18 from other signals that may cause co-channel interference. It consists of one code added to 14. Multipliers 12 and 14 multiplex the first and second channel signals before being transmitted through antennas 16 and 18. To simplify the diagram, the RF up converter is not shown.
[0006]
At remote station 2, the receiver receives the signals from both antennas 16 and 18 on a single antenna, down-converts these signals, despreads these signals using the code OC, Recovered as composite signals of channels CH1 and CH2 transmitted from antennas 16 and 18, respectively. In the first symbol time slot between 0 and T, the composite QPSK modulated signal R₁Is received (R₁ = k₁₁S₁ + k₁₂S₂), In the second symbol time slot between T and 2T, the composite QPSK modulated signal R₂Is received (R₂ = -k_{twenty one}S₂ * + k_{twenty two}S₁*, Asterisk indicates complex conjugate) Constant k₁₁Is the transmission line constant from the first antenna 16 to the remote station 2 during the first time slot and the constant k₁₂Is the transmission line constant from the second antenna 18 to the remote station 2 during the first time slot and the constant k_{twenty one}Is the transmission line constant from the first antenna 16 to the remote station 2 during the second time slot and the constant k_{twenty two}Is the transmission line constant from the second antenna 18 to the remote station 2 during the second time slot. The receiver reverses the channel and soft symbol S₁'And S₂'Recover. In this case, the following equation is established.
S₁'= k₁₁R₁ + k₁₂R₂And S₂'= k_{twenty one}R₂ ^* + k_{twenty two}R₁ ^*.
[0007]
In this space-time encoder technique, the first and second symbols are redundantly transmitted from separate antennas. The first symbol is encoded to be transmitted in both the first symbol time slot and the second symbol time slot, and the second symbol is also encoded in the first symbol time slot and the second symbol time slot. Encoded to be transmitted in both symbol time slots. The effect of this symbol recovery technique is that when interleaving is also used, fading or drop-out regions that can appear during one symbol time slot are less likely to appear in both time slots. That is. Interleaving is used before space-time coding to weaken the temporal correlation of adjacent bits. The received signal is both time slots R₁And R₂Since the received signal is recovered during this period, the fading effect is weakened.
[0008]
However, the prior art benefits from managing power and phase independently for individual beams transmitted by various diversity antennas to increase the spectral efficiency at the base station while minimizing co-channel interference Is not used. The prior art does not take advantage of the spatial power management of beams sent independently of each other to increase spectral efficiency at the base station while minimizing co-channel interference.
[0009]
Summary of the Invention
An object of the present invention is to improve the downlink performance of a portable radio system. Another objective is to minimize the undesirable effects of fading and drop-out.
[0010]
These and other objectives include receiving at least two space-time encoded signals from an antenna system associated with the first station, and combining channel state information based on the received space-time encoded signals. And obtaining the method.
[0011]
These and other objectives are for the method to transmit at least two space-time encoded signals on each beam of the multi-beam antenna array, and for each space-time encoded signal at a second station. It is realized by other aspects, including measuring a channel impulse response and transmitting a signal indicative of a selected set of weakest attenuated signals. A multi-beam antenna array is associated with the first station. The beam transmits a signature code embedded in each space-time encoded signal, and the signature codes are measured by the second station by separating and measuring the channel impulse response corresponding to each space-time encoded signal. They are orthogonal to each other so that they can. The space-time encoded signal includes a selected set of least attenuated signals and the remaining set of most attenuated signals.
[0012]
These and other objects are formed by a multi-beam antenna array associated with a first station for transmission of at least two corresponding space-time encoded signals generated by a space-time encoder. Selecting at least two of the plurality of beams; determining a time delay associated with each of the at least two space-time encoded signals received in each beam; This is accomplished by other aspects including setting a time delay corresponding to each beam in a variable delay line coupled between the antenna array and the space-time encoder.
[0013]
  The method used by one embodiment is:
  Receiving at least two space-time encoded signals from an antenna system associated with a first communication station;
  Obtaining complex channel state information based on the received space-time encoded signal;
  Partitioning the complex channel state information into a plurality of channel state information segments;
  Transmitting the complex channel state information to the first communication station;
  HaveThe plurality of channel state information segments include one or more phase segments and one or more amplitude segments;
In the step of transmitting the complex channel state information to the first communication station;Transmitting the one or more phase segments before transmitting the one or more amplitude segmentsA method of sequentially transmitting the plurality of channel state information segments.
  Hereinafter, the present invention will be described in detail with reference to the accompanying drawings.
[0014]
Detailed Description of the Preferred Embodiment
In order to increase the spectral efficiency of transmission from the base station while minimizing co-channel interference, we have developed a method to perform power management independently for each beam transmitted by various antennas of the diversity antenna, and the beam A space-time encoder technology has been developed to take advantage of the incoming diversity angle and the spatial power management of the beams sent independently of each other. Beam space-time techniques differ from known space-time encoder techniques by utilizing power beamwidth management and arrival diversity angles with two or more independently transmitted orthogonal beams. Use vertical deflection (in case of two beams), use different pilot codes for each beam in CDMA system as well as common CDMA spread spectrum code for all beams, and for each beam in CDMA system without pilot code Identify different orthogonal beams by the receiver by using different spreading spectrum codes and using different training sequences (eg pilot codes) multiplexed on each beam in a TDMA system Can do. Those skilled in the art will appreciate that other orthogonal beam technologies not described above, or receivers at remote stations, provide means for individually identifying individual beams and recovering the signals they carry. It will be appreciated that there are techniques using various combinations of
[0015]
Power management techniques that transmit different powers in different orthogonal beams minimize co-channel interference even when this power management control is applied to beams directed to overlapping sectors of diversity antennas or omnidirectional beams. By suppressing the spectral efficiency, the spectral efficiency at the base station is improved throughout the system. However, in the case of orthogonally coded beams directed in various directions, further improvements are realized by the spatial power management of beams sent independently of each other. The relatively poor downlink performance in a wireless environment with large angular spread is significantly improved by applying the beam space time encoder technology described herein.
[0016]
In FIG. 4, a first aspect of improved transmitter 100 (referred to as diversity antenna power management) includes a known space-time transmit diversity encoder 10 and composite multipliers 12 and 14. The improved transmitter 100 further includes scaling amplifiers 102 and 104 and diverse antennas 16 and 18. In a CDMA system, multipliers 12, 14 add different spread spectrum codes to the various beams so that the receiver of remote station 2 can identify each beam individually.
[0017]
Multipliers 12, 14 are added with separate distinguishable spreading codes in a CDMA system as described herein to form orthogonal beams, but diversity by any means for forming orthogonal beams. It will be appreciated that individual power management of transmissions from antennas (ie, superimposed coverage), or separate power management of transmissions from controllable directional antennas for this, is possible. For example, in a CDMA system where multipliers 12 and 14 have the same spreading code, another set of multipliers 12 'and 14' (not shown) can be used to add a pilot code to the channel signal. The multipliers 12 'and 14' are then provided with orthogonal pilot codes so that the receiver of the remote station 2 can identify each beam individually. In other variations, antennas 16 and 18 are orthogonally deflected (eg, deflected to tilt ± 45 ° relative to vertical or some other reference), but otherwise have the same sector. It consists of a single exciter antenna with two elements arranged to generate two beams to cover. Although such beams are orthogonal to each other, transmissions on their respective signal paths are subject to uncorrected fading.
[0018]
Scaling control signals SA1 and SA2 individually control the amplification or attenuation performed by separate scaling amplifiers 102 and 104, respectively. Scaling control signals SA1 and SA2 may be complex signals having real components of scale amplitude, or imaginary components of phase shift, or both real and imaginary components of both scale amplitude and phase shift. It will be appreciated that amplification may be applied at the output of encoder 10, or before or after multipliers 12 and 14, or within antennas 16 and 18.
[0019]
Antennas 16, 18 are diversity antennas that cover overlapping sectors or are omnidirectional. This first aspect differs from known space-time coding systems in that the power transmitted in each beam is controlled individually by SA1 and SA2.
[0020]
In FIG. 5, a second aspect of the improved transmitter 100 (referred to as angular spectrum power management) includes a known space time transmit diversity encoder 10 and composite multipliers 12 and 14. The improved transmitter 100 further includes scaling amplifiers 102 and 104 and direction control antennas 106 and 108. Unlike antennas 16 and 18 of FIG. 2, directional antennas 106 and 108 are directed toward direct path 3 and indirect path 5 (FIG. 1) or angular spread AS, ie, the angular power spectrum. It is directed in some other direction that covers a portion that exceeds the threshold described herein. In the CDMA system, the multipliers 12 and 14 each transmit the various beams to the remote station 2 so that the receiver of the remote station 2 can individually identify the angular beams as described with respect to the first aspect using the diversity antenna. Add different spread spectrum codes or use other means. Scaling control signals SA1 and SA2 individually control amplification or attenuation performed by separate scaling amplifiers 102 and 104, respectively. Scaling control signals SA1 and SA2 may be complex signals having real components of scale amplitude, or imaginary components of phase shift, or both real and imaginary components of both scale amplitude and phase shift. It will be appreciated that amplification may be applied at the output of encoder 10, or before or after multipliers 12 and 14, or within antennas 106 and 108. Multipliers 12, 14 are added with separate spreading codes in a CDMA system as described herein to form orthogonal beams, but direction control (ie, by means of forming orthogonal beams) It will be appreciated that individual power management of transmissions from the antennas (directions selected as described herein) is possible.
[0021]
In a third aspect (referred to as directional diversity, not shown separately), no differential amplification is performed, and both channels CH1 and CH2 have balanced equal amplification, but their signals are directed to direction control antenna 106. The amplifiers 102 and 104 of FIG. 5 have been removed from the transmitter 100 so that they are transmitted directionally through and.
[0022]
There are several ways to implement a directional control antenna. In FIG. 6, a known six-way control antenna 6 includes six identical site corner antennas such as corner antenna 8 which are arranged in a circle and are all shown in plan view. Each corner antenna 8 includes a single half-wave dipole 12 as an exciter element and a corner reflector 14. Each corner antenna 8 illuminates a beam width of 60 ° in the plan view. A 6-diversity antenna system 6 is presented that provides angular position information indicating the azimuth angle from the base station to the remote station based on the received signal strength at 820 MHz (Rhee, Sang-Bin, “Angle Based on Signal Strength”. Vehicle Location in Angular Sectors Based On Signal StrengthIEEE Trans. Veh. Technol.VT-27, 244-258, November 1978). Such co-site antennas have 360 ° coverage with 3 sectors (120 ° antenna), 4 sectors (90 ° antenna), 5 sectors (72 ° antenna), 8 sectors (45 ° antenna), Or it can be divided into any convenient number of sectors feasible.
[0023]
In the second and third aspects of the present invention, a directional control antenna system is used in the portable radio transmitter 1 (FIG. 1). A directional antenna system is defined as a system that can generate two or more distinct and individually controllable beams. The system is arranged to produce two or more beams (eg, two identifiable that are deflected to tilt ± 45 ° with respect to the vertical, but otherwise cover the same sector It may be a single antenna having two or more exciter elements (positioned to produce a uniform beam). The system may be a multi-antenna system that generates a beam that covers various sectors. For example, the directional control antenna system may advantageously be a six corner system such as the antenna system shown in FIG. The direction control antenna system is used in a reception mode to determine the angular position of the remote station 2 based on the signal transmitted from the remote station 2. The two sectors with the strongest received signal are identified as likely directions of arrival for direct path 3 and indirect path 5 (see FIG. 1). As antennas for illuminating the two sectors, the directional antennas 106 and 108 (FIGS. 4 and 5) of the second and third aspects of the present invention are selected. Alternatively, as will be described later, the respective arrival directions can be obtained based on the calculation of the angular power spectrum.
[0024]
In FIG. 7, a known trackable beam phased array antenna 20 includes an array of exciter elements 22 (eg, half-wave dipoles) spaced from a ground plane or reflector plane 24. . Although FIG. 7 shows eight radiating elements, more or fewer elements may be used. Each exciter element 22 is given a signal from a corresponding phase shifter 26. Each phase shifter 26 changes the phase of the signal S and attenuates (or amplifies) the amplitude according to the corresponding individual control portion of the control signal C. For example, the control signal C includes 8 phase shift parameters and 8 attenuation parameters. Each phase and amplitude parameter individually adjusts the phase and amplitude radiated from the corresponding one of the eight exciter elements of the antenna 20. The angular beam width of such an antenna is limited by the ratio of the wavelength of the emitted signal divided by the aperture dimension D. However, by adjusting the signal amplitude on the exciter element 22 distributed across the antenna with what is called a weighting function, the beam is broadened, the center of the beam is flattened, and / or side lobes are suppressed. So that the beam can be shaped. By adjusting the phase gradient in the exciter elements across the antenna, the beam can be electronically directed in the control direction.
[0025]
In the second and third variations, the antenna system for transmitter 1 (FIG. 1) includes a plurality of phased array antennas 20 configured as a multi-antenna system. In FIG. 8, an exemplary multi-antenna system faces the outer, equally spaced angular directions, such that three phased array antennas 20 are formed as a base station antenna system. May include three antennas (considered as phased array antenna 20). Each antenna 20 is configured to cover a 120 ° sector. The base station finds the remote station by electronically scanning the antenna 20. The amplitude weight for each radiating element is preferably set to a maximum value, all equal so that the antenna produces its narrowest beam (the most directional beam). The receive beam first calculates the phase parameter for the control signal C representing the phase gradient across the antenna to obtain the desired beam point, and then controls the antenna 20 to point in the desired direction by: Scanned in stages. Second, the receiver of transmitter 1 (FIG. 1) detects any received signal strength. The steps of facing the receive beam and detecting the signal strength are repeated at each of several beam positions until the entire sector covered by antenna 20 is scanned. In this way, the angular position of the remote station 2 is determined with an accuracy limited only by the minimum possible beam width of the antenna 20. After the positions of direct path 3 and indirect path 5 are determined to be in different sectors (eg, 120 ° sectors), antennas 106 and 108 (FIG. 5) are connected to the antenna system by direct path 3 and indirect. A phase gradient that determines a beam that is selected from a plurality of antennas 20 closest to the path 5 and that faces the angular positions of the direct path 3 and the indirect path 5 within a sector covered by each selected antenna 20 is determined. Alternatively, when paths 3 and 5 are in a single sector, the antenna system can form two beams in this single sector (see discussion below regarding FIG. 10). Two transmit beams can be formed in this single sector so as to be directed along paths 3 and 5.
[0026]
In FIG. 9, the antenna system 30 includes four radiating elements 32 spaced from a ground plane or reflector plane 34. Each radiating element or exciter element 32 is provided with a signal from a known Butler matrix 36. The Butler matrix combines the radiation from the four exciter elements 32 to produce signals S1, S2, S3, S4, B2, B3, and B4 that are oriented in a certain angular direction. And realize the phase shift function and synthesis function acting on S4. In general, a Butler matrix performs a Fourier processing function that configures M radiating elements to form M constant orthogonal beams (“square bins”). For example, in the antenna system 30, the signal S1 is transmitted only by the first beam B1, the signal S2 is transmitted only by the second beam B2, the signal S3 is transmitted only by the third beam B4, and the signal S4 is Only the fourth beam B4 is transmitted. Using the switching matrix, the desired signal (eg, signals CH1 and CH2 in FIG. 5) is sent to one of the lines for signals S1, S2, S3, and S4, from which each beam B1, B2, B3. , And sent in B4.
[0027]
In a variation of the second and third aspects, the antenna system for transmitter 1 (FIG. 1) includes a plurality of “Butler Matrix” antennas 30 configured as a multi-antenna system. In FIG. 8, an exemplary multi-antenna system is arranged to face the outer, equally spaced angular directions so that a “Butler Matrix” antenna 30 is formed as a base station antenna system. 3 antennas (referred to herein as “Butler Matrix” antennas 30). Each antenna 30 is configured to cover a 120 ° sector with, for example, four beams. The base station finds the remote station by electronically switching the four beams (each 30 °) of each of the three antennas 30 and detecting the received signal strength. In this way, the angular position of the remote station 2 is obtained with an accuracy of one beam width of the antenna 30. After the positions of the direct path 3 and the indirect path 5 are determined, when the direct path 3 and the indirect path 5 are located in different sectors, the antennas 106 and 108 (FIG. 5) are connected to the transmitter 1 ( The antenna system for FIG. 1) is selected from two different “Butler Matrix” antennas 30. The two specific “Butler Matrix” antennas 30 are selected to cover the sector closest to the direct path 3 and the indirect path 5, and from there the most strict to the path within each selected antenna 30. A specific beam is selected that matches. Alternatively, different beams of the same “Butler Matrix” antenna 30 may be selected as the antennas 106 and 108. Within the sector covered by each antenna 30, the beam that faces the angular position of each of the direct path 3 and the indirect path 5 is selected by a switch matrix (not shown).
[0028]
In FIG. 10, antenna 40 is a modification of phased array antenna 20 that produces two independently trackable and shapeable beams. Antenna 40 includes an array of exciter elements 42 (eg, a half-wave dipole) spaced from a ground plane or reflector plane 44. Although FIG. 10 shows eight radiating elements, more or fewer elements may be used. However, unlike the antenna 20, each exciter element of the antenna 40 is given a signal from the corresponding adder 48. Each adder 48 superimposes (eg, adds) signals from two corresponding phase shifters 46-1 and 46-2. All phase shifters 46-1 form a first phase shifter bank and all phase shifters 46-2 form a second phase shifter bank. Each phase shifter 46-1 of the first bank changes the phase of the signal S1 and attenuates (or amplifies) the amplitude according to the corresponding individual control portion of the control signal C1. For example, the control signal C1 includes eight phase shift parameters and eight attenuation parameters that individually adjust the phase and amplitude output from the corresponding phase shifter 46-1. Correspondingly, each phase shifter 46-2 in the second bank changes the phase of the signal S2 and attenuates (or amplifies) the amplitude according to the corresponding individual control portion of the control signal C2. For example, the control signal C2 includes eight phase shift parameters and eight attenuation parameters that individually adjust the phase and amplitude output from the corresponding phase shifter 46-2. Adder 48 combines the outputs of respective phase shifters 46-1 and 46-2 and provides the combined signal to radiating element. In this way, the control signal C1 adjusts the first beam that emits the signal S1, and the control signal C2 simultaneously adjusts the second beam that emits the signal S2.
[0029]
In a variation of the second and third aspects, the antenna system for transmitter 1 (FIG. 1) includes a plurality of phased array antennas 40 configured as a multi-antenna system. In FIG. 8, an exemplary multi-antenna system is directed to an outer, equally spaced angular orientation such that three phased array antennas 40 are formed as a base station antenna system. Includes three antennas (referred to herein as phased array antennas 40). Each antenna 40 is configured to cover a 120 ° sector with two independently shaped and trackable beams. The base station finds the remote station by electronically scanning the beam of antenna 40 as discussed above with respect to antenna 20 (FIG. 7). After the positions of direct path 3 and indirect path 5 are determined, antennas 106 and 108 (FIG. 5) are selected and selected from a plurality of antennas 40 closest to direct path 3 and indirect path 5 of the antenna system. In addition, a phase gradient that determines a beam facing the angular position of the direct path 3 and the indirect path 5 is determined in the sector covered by each antenna 40.
[0030]
Alternatively, different beams of the same dual beam antenna 40 may be selected as the antennas 106 and 108. In FIG. 11, antennas 106 and 108 (FIG. 5) are implemented in separate beams of dual beam antenna 40 (ie, beams 1 and 2) and scale the amplitude coefficients of control signals C1 and C2 (FIG. 10). By doing so, the scaling amplifiers 102 and 104 (of FIG. 5) are not required.
[0031]
In a fourth aspect, the base station uses a time division multiple access (TDMA) transmitter rather than a spread spectrum CDMA transmitter. In FIG. 12, the training sequence TS1 is modulated by the QPSK modulator 101 and sent from there to the first input of the multiplexer 105, and the training sequence TS2 is modulated by the QPSK modulator 103 and from there the first of the multiplexer 107 Sent to the input. The training sequences TS1 and TS2 are orthogonal to each other and constitute a means by which the remote station 2 can identify each beam in the same way that the pilot code helps to distinguish the beams in the CDMA system. In the TDMA system, multipliers 12 and 14 (of FIGS. 4, 5 and 11) are omitted and channel signals CH1 and CH2 are sent to the second inputs of multiplexers 105 and 107, respectively. In this fourth aspect, amplifiers 102 and 104 amplify or attenuate the outputs of respective multiplexers 105 and 107 independently. The outputs of amplifiers 102 and 104 are sent to the antenna system (through an up converter or the like (not shown)). This antenna system can realize the overlapping coverage of the diversity antennas 16 and 18 (FIG. 4) as in the first mode, or can implement the directional antennas 106 and 108 (as in the second and third modes). The direction control coverage of FIGS. 5 and 11) can also be realized. Further, in the case of directional control coverage, a modification is possible in which power management is eliminated, the amplifiers 102 and 104 are omitted, and the beam from the directional antennas 106 and 108 is used to track angle (beam) diversity. A data slot in a time division system may include, for example, 58 data bits followed by a 26 bit training sequence followed by 58 data bits, as in the GSM system. The training sequence is a signal S to the remote station 2 so that the remote station can identify each beam individually._INAnd identify the source of the individual beams. In this way, the remote station 2 receives the two beams separately using the training sequence instead of using the orthogonal spreading code OC as in the CDMA system.
[0032]
Although two beams are discussed, it is easy to extend to higher order coding techniques that use more beams. For example, to recover the first symbol from the encoded channel signal, four symbols (S₁, S₂, S_Three, S_Four) Is encoded into four channel signals (CH1, CH2, CH3, CH4) in four symbol time slots. The four channel signals are then transmitted from the base station with four beams, each beam corresponding to a channel signal of channel signals CH1, CH2, CH3, and CH4. Although this specification discusses QPSK modulation techniques, extensions to other PSK modulation techniques are easy, and extensions to other modulation techniques (eg, QAM) are available as well.
[0033]
In FIG. 13, a closed loop control system for managing transmission power is shown as process S10. In step S102, the base station selects a power level to be transmitted from each antenna. For example, in a two antenna system, the base station is measured at the total power (ie, P1 + P2) defined by a conventional power control loop (eg, a control loop typically used in a CDMA system) and at remote station 2. The powers P1 and P2 are selected based on the relative power (ie, P1 / P2) determined by the power control coefficient. In step S104, a value representing the selected transmission power level is transmitted to the remote station over the outgoing channel. In step S106, the power level received from each antenna radiation pattern at the remote station is measured and a corresponding power control factor is determined. As a power control coefficient for each antenna radiation pattern, a value proportional to a value obtained by dividing the reception power at the remote station 2 by the transmission power indicated by the power level value transmitted to the remote station through the transmission channel is obtained at the remote station 2. Desired. In step S106, the power control coefficient is transmitted from the remote station to the base station over the transmission channel. In step S108, the power control coefficients from step S106 are compared for each antenna. In step S110, an adjustment value of the transmission signal power is obtained according to the comparison in step S108. This adjustment is made to increase the transmission power transmitted on the channel having the preferred transmission quality and to decrease the transmission power of the channel having the insufficient transmission quality. Then, in step S102, at the start of the cycle, the base station selects the adjusted transmit power and forms the basis for the power to be transmitted from the antenna during the next cycle of closed loop beam power management. The loop cycle delay may be one time slot as in the third generation TDMA system.
[0034]
Alternatively, the remote station compares (in step S108) the power control factor for each antenna obtained from step S106, and then provides power factor indicator information to be transmitted from the remote station to the base station on the uplink outgoing channel. calculate. For example, the power control factor ratio (eg, P1 / P2 for two antennas) can be advantageously calculated as power factor indicator information and transmitted in the uplink direction. Alternatively, the power coefficient indicator information may be a value obtained by quantizing the ratio (for example, a single bit indicating whether P1> P2).
[0035]
Alternatively, at step S104, the selected transmit power is stored for the closed loop control system cycle time. For example, in a two antenna system, the base station is measured at the total power (ie, P1 + P2) defined by a conventional power control loop (eg, the control loop normally used in a CDMA system) and the remote station 2 The powers P1 and P2 are selected based on the relative power (ie, P1 / P2) determined by the power control coefficient. In step S106, the power level received from each antenna radiated power at the remote station is measured at the remote station 2 and transmitted as a power control factor from the remote station 2 to the base station 1 on the uplink outgoing channel. The power control coefficient is normalized with respect to each transmission power stored in step S104. In step S108, the normalized power control coefficient obtained from step S106 is compared for each antenna at the base station. In step S110, an adjustment value of the transmission signal power is obtained according to the comparison in step S108. Then, in step S102, at the start of the cycle, the base station selects the adjusted transmit power and forms the basis for the power to be transmitted from the antenna during the next cycle of closed loop beam power management.
[0036]
In FIG. 14, a portable radio system having a closed loop beam power management controller includes a base station 210 and a remote station 230. Base station 210 forms first and second radiation patterns with space-time encoder 212 and antenna system 216, respectively, that encode the stream of symbols as first and second space-time encoded signals. Transmitter 214 for transmitting the first and second space-time encoded signals from the antenna system at first and second initial transmission power, respectively, and base station reception for receiving power coefficient indicator information from the remote station And a power management controller 222 for determining first and second adjusted transmission powers based on first and second initial transmission patterns and power coefficient indicator information, respectively.
[0037]
The antenna system 216 may include a plurality of antennas, each antenna being an antenna that generates a substantially omnidirectional radiation pattern or a radiation pattern directed to a sector. The omnidirectional antennas are advantageously spaced apart. The antenna system 216 can form the first and second radiation patterns as orthogonal radiation patterns that can be individually received at a remote station. Alternatively, transmitter 214 processes the first and second space-time encoded signals so that the signals transmitted from the antenna system are orthogonal to each other and can be received individually at the remote station. Includes circuitry.
[0038]
The antenna system 216 can generate multiple beams (ie, a multi-beam antenna), and the base station includes an antenna controller 218 that controls the multi-beam antenna to form multiple beams. . In one aspect, the multi-beam antenna may be a multi-port Butler matrix antenna, in which case the transmitter 214 may be configured with first and second respectively based on first and second adjusted transmit powers, respectively. An amplifier that scales the first and second space-time encoded signals to form a plurality of scaled space-time encoded signals, and the antenna controller 218 includes first and second scaled spaces. A switch for coupling the time-coded signal to the first and second input ports of the Butler matrix antenna, respectively, to form first and second beams, respectively;
[0039]
Alternatively, the multi-beam antenna includes a phased array antenna system, and the antenna controller 218 includes a beam steering controller that forms first and second weighting functions. The beam steering controller inputs the first and second weighting functions to the phased array antenna system and is based on the first and second adjusted transmit powers without the scaling amplifier in the transmitter 214, respectively. To scale the antenna gains of the first and second beams, respectively. The phased array antenna system may include a multiple beam phased array antenna (eg, 40 in FIG. 10) or multiple phased array antennas (eg, 20 in FIG. 7).
[0040]
In some aspects, the power factor indicator information includes first and second power control factors, and the base station receiver 220 receives the uplink transmission information and includes first and second in the uplink transmission information. Detect the value of the power control coefficient.
[0041]
When the indicated first path attenuation characteristic (ie, the first power control coefficient) is less noticeable than the indicated second path attenuation characteristic (ie, the second power control coefficient), the power management controller 222 Includes a circuit (eg, a logic mechanism or processor) that determines that the adjusted transmit power is greater than the second adjusted transmit power.
[0042]
Remote station 230 includes remote station receiver 234, detector 236, power measurement circuit 238, and processor 240. Receiver 234, detector 236, power measurement circuit 238, and processor 240 provide power that remote station 230 has received from the first radiation pattern and measured by circuit 238, and initial transmit power determined by detector 236. The circuit is configured to be able to determine the indicated path attenuation characteristic based on According to this circuit, remote station 230 determines the indicated first path attenuation characteristic for the first radiation pattern of antenna system 216 and the indicated second path attenuation characteristic for the second radiation pattern of system 216. can do. This is because these two radiation patterns can be received separately. Detector 236 determines the initial transmit power, power measurement circuit 238 measures the power received from the radiation pattern received by receiver 234, and processor 240 divides the received power by the value of the initial transmit power. A value proportional to the obtained value is obtained as a power control coefficient. The power measurement circuit 238 measures the received instantaneous power, or in other embodiments, measures the average received power, or in other embodiments, measures both and is averaged with the received instantaneous power. A combination of received powers. The remote station 230 further includes a transmitter 242 that transmits the value of the power coefficient indicator information or the values of the indicated first and second path attenuation characteristics to the base station.
[0043]
In a variant, the processor 240 forms power factor indicator information as a ratio of the indicated first path attenuation characteristic divided by the indicated second path attenuation characteristic. In other variations, the processor 240 forms power factor indicator information having a first value when the indicated first path attenuation characteristic is less noticeable than the indicated second path decay characteristic, and Power coefficient indicator information having a second value is formed when a path attenuation characteristic of 1 is more prominent than a second path attenuation characteristic shown.
[0044]
In an exemplary aspect, the base station transmits a first signal from a first antenna at a first predetermined signal power P1, and a receiver at remote station 2 is received from a first antenna at the remote station. The power is obtained as the first power control coefficient PCC1. The base station also transmits a second signal from the second antenna with a predetermined signal power P2, and the receiver of the remote station 2 uses the second power control for the power received from the second antenna of the remote station. Calculated as coefficient PCC2.
[0045]
Both the first signal and the second signal are transmitted simultaneously from the first and second antennas, respectively, at respective predetermined power levels in normal operation. The transmit power is an orthogonal training sequence that can be used at the remote station 2 by the multipliers 12 and 14 (FIGS. 4, 5, and 11), respectively, or at the TDMA base station (FIG. 12). Can be distinguished by using. The receiver of remote station 2 obtains the signal power received from each antenna and transmits the uplink power using the values representing these received signal powers as separate power control coefficients PCC1 and PCC2 or relative power control coefficients PCC1 / PCC2. A part of the data is transmitted to the base station.
[0046]
In a preferred aspect, the base station first transmits signals from multiple antennas at selected powers that may not be equal to each other in normal operation (S102). In one variation, the base station transmits a power level selected from each of the plurality of antennas as the power level transmitted on the downlink outgoing channel. The remote station (1) receives the selected power level of the base station (S104), (2) determines the signal power received from the antenna (S106), and (3) is transmitted from each antenna of the base station. The received power is compared with the power received at the remote station, and the relative attenuation in the downlink path is determined as the ratio of the received power and the corresponding transmitted power (S108). The remote station uses this ratio obtained for each antenna as a power control coefficient and sends it back to the base station as uplink transmission data. The base station then adjusts the power that can be transmitted from each antenna according to the relative attenuation determined for all other downlink transmissions (S110).
[0047]
In another variation, (1) the remote station determines the signal power received from the antenna as a power control coefficient (S106), and (2) the remote station transmits the power control coefficient to the base station using uplink transmission data. Send back. The base station then (1) adjusts the closed loop time delay when it receives the power control factor from the remote station 2 (S104), and (2) the power transmitted from each antenna of the base station at the remote station. The relative attenuation in the downlink path is determined relative to the received power (S108), and (3) transmitted from each antenna of the base station according to the relative attenuation determined for all other downlink transmissions. Adjust the possible power (S110).
[0048]
In either variation, the power that can be transmitted from the antenna is greater for the antenna associated with the path that was determined to have less path attenuation. For example, the indicated path attenuation characteristic is advantageously determined as the ratio of the power received at the remote station 2 and the power transmitted from the base station 1. Thus, little or no power is transmitted on paths that are not well received by the remote station 2, while more power is transmitted on paths that are well received by the remote station 2. In many multipath environments, reception at the remote station 2 is hardly improved even if the power transmitted on the path with excessive attenuation increases, and this increase in power is the same as occurs at other remote stations. Contributes to channel channel interference. In order to improve the overall portable radio system, the path with the weakest attenuation is given maximum transmit beam power. For each beam, the base station adjusts the amplitude parameters in the control scaling signals SA1 and SA2 (FIGS. 4 and 5) or in the control signal C (FIG. 6) or the control signals C1 and C2 (FIG. 9). The power transmitted from each antenna is adjusted by adjusting the overall antenna gain.
[0049]
In this closed loop power control method aspect, the remote station determines which antenna (or beam) is associated with the lowest attenuation path. The remote station sends a signal back to the base station on the uplink outgoing path indicating which antenna (or beam) is preferred (ie, the least attenuated). The remote station preferably determines the preferred antenna to save the number of bits transmitted on this uplink outgoing path and indicates this by a single bit (ie, “0” indicates antenna 16 "1" means that the antenna 18 is preferable (see FIG. 4). The base station receives this single bit indicator and applies it to determine a predetermined relative power balance. For example, if 80% of the total power is added to antenna 16 (for example, when this is the preferred antenna), 20% of the total power is added to antenna 18, 100% of the total power is added to antenna 16, and antenna 18 has Performance improves consistently over no power applied. Thus, the base station receives the single bit relative power indicator and uses 80% / 20% for the “1” indicator bit as the relative power P1 / P2 for antennas 16 and 18 and “0” indicator For bits, select 20% / 80%.
[0050]
In a slowly changing wireless environment, the coefficient (or any associated channel information) can be analyzed as a segment, and each segment (including fewer bits than the entire coefficient) can receive more uplink time The slot can be used to transmit uplink data to the base station. Within a segment (possibly multiple TDMA time slots), the most significant bits are preferably transferred first, and these course values are gradually updated to be more precise with successive bits. Conversely, in a rapidly changing wireless environment, the average exponent of all coefficients is transmitted on the uplink (or estimated according to the outgoing symbol), and then only the most significant bits of the coefficients are transmitted (ie, the lower bits A special reserved outgoing symbol may indicate that one or more other compression formats for uplink transmission of the coefficients are being used. In extreme cases, it indicates that the power control factor is 1 when the downlink channel is good (eg, 80% of the total power transmission) and the power control factor is low when the associated channel is not appropriate. Only one bit is transmitted in the uplink direction, indicating zero (eg, only 20% of full power transmission).
[0051]
This closed loop control for beam power management is self-adaptive. If a power control factor that causes beam power overcompensation is uplinked to the base station, the closed loop control system corrects this during the next closed loop control cycle. Those skilled in the art will appreciate that other data compensation techniques can be used in the uplink transmission to make adjustments to the rapidly changing wireless environment. Similarly, those skilled in the art will appreciate that the remote station, not the base station, can calculate commands to the base station to increase or decrease the power in a particular beam.
[0052]
In other variations suitable for a slowly varying wireless environment, the first and second beams may be transmitted sequentially at each predetermined power level of the beam in a calibration mode. In such a variation, once the remote station does not have to use the orthogonal code OC or the orthogonal pilot signal to determine from which beam the received signal strength (eg, power control factor) is received. Only one beam is transmitted at a time. After the channel attenuation is determined, the signal S using beam space-time coding techniques._INIs sent.
[0053]
In addition to using amplifiers 102 and 104 or beam gain in the phased array antenna to control closed-loop power management, other aspects utilize angular diversity management and / or beam width management to control power management. Omitted. Still other aspects utilize both angle diversity management and / or beam width management, and power management.
[0054]
The performance of beam space-time coding techniques depends at least in part on the angular spread AS that characterizes the wireless environment and how the base station adapts each beam to match this angular spread. Downlink performance is generally improved when the downlink beam is directed to an angle of arrival where a sharp peak occurs in the angular power spectrum of the signal from the remote station. A sharp peak indicates good transmission along the path shown (eg, the likely direction of paths 3 and 5). However, sharp peaks are not always found. When the angular power spectrum spreads and no sharp peaks are found, the angular spread AS is estimated and multiple beams used for downlink transmission are assigned to cover the angular spread in general. In this way, the downlink transmission is spatially matched to the total channel determined by the angular rate.
[0055]
The circuit for measuring the angular power spectrum includes a receiver 220 (FIG. 14) and a signal and data processing circuit as required to determine the angular power spectrum and the peak of such spectrum as described below. Is included. Further, to direct the beam direction towards the detected angular position, the antenna controller 218 calculates an array steering vector that is input to the antenna system 216 (FIG. 14). When an excessive number of peaks are detected in the angular power spectrum, the power management controller 222 (FIG. 14) selects the angular direction to be used to form the beam. The power management controller 222 can select the beam direction towards a particular angle (ie, peak) of the incoming path, or the power management controller 222 can select the beam direction to cover the detected angular velocity, and In some cases, the beam width can be selected. The selected direction is provided to the antenna controller 218 to form a beam command to the antenna system.
[0056]
In systems using frequency division duplexing, uplink and downlink transmissions are performed at different frequencies. It is not guaranteed that the peak measured in the uplink power spectrum will occur at an angle that corresponds to the angle with good transmission performance in the downlink direction. However, the use of angular diversity management or beam width management increases the likelihood of good downlink transmission.
[0057]
Whether it is angular diversity management or beam width management, it is necessary to measure the angular power spectrum in some form. The remote station broadcasts the uplink signal in its normal operation (eg, outgoing operation), the base station antenna system receives the signal, and the base station receives the angular power spectrum (ie, the azimuth angle in the plan view). The received power as a function of. FIG. 18 is a graph showing the angular position of the signal power received from the remote station 2. In FIG. 18, the discrete power management at each of the 12 angular positions is illustrated as management based on 12 fixed position antenna beams directed at 30 ° intervals in the antenna system for base station 1, for example. ing. An exemplary 12 beam antenna system includes three Butler matrix antennas arranged in a triangle so that each Butler matrix antenna forms a 12 beam antenna system that forms four beams. It's okay. Although a 12 beam antenna system is discussed in this example, it will be appreciated that any number of beams in the antenna system can be applied to the present invention (eg, 24 beams, etc.).
[0058]
Alternatively, the antenna system can form 12 beams, each phased array antenna forming a trackable beam with a 30 ° beam width to allow scanning over 4 beam positions. It may include three phased array antennas arranged in a triangle to form an antenna system. A 12 beam antenna system may include any type of 12 antennas having a 30 ° beamwidth and arranged in 30 ° increments around a 360 ° sector. Although a 12 beam antenna system is discussed in this example, it will be appreciated that any number of beams in the antenna system can be applied to the present invention (eg, 24 beams, etc.).
[0059]
An antenna system based on a phased array antenna is a more interpolated angular power spectrum (eg, by tracking the antenna beam to point to the number of angular positions necessary to generate the angular power spectrum (eg, Gives the opportunity to generate G1) in FIG. The power management controller 222 (FIG. 14) generates an angular power spectrum in process S20 (FIG. 15) by scanning θ in steps S20A and S20B and determining angular power in step S21. If the angle θ is given, the power management controller determines the antenna orientation by causing the antenna controller 218 (FIG. 14) to calculate the array steering vector (step S211 in FIG. 16). The phased array antenna then receives signals from each radiating element of the remote station 2 at the receiver 220 (FIG. 14) and forms a signal vector at step S212 of FIG. . Each radiating element is preferably positioned at a half wavelength distance from an adjacent element. For example, if the phased array antenna includes 12 radiating elements (only 8 radiating elements are shown in antenna 20 of FIG. 7), the signal received at each of the 12 radiating elements is Sampled to form a measurement signal vector. The sampled signal is preferably a composite value having amplitude and phase information. The signal from each of the 12 radiating elements is a column vector

As a 12-element received signal vector. Next, the received signal vector

Complex conjugate transpose of is a row vector

The spatial covariance matrix of the received signal, formed as

Is calculated in step S213 (FIG. 16). Receive signal vector

When the length of is 12 elements, the spatial covariance matrix of the received signal

Becomes a 12x12 matrix.
[0060]
Array steering vector

Is a column vector with one vector element for each radiating element of the phased array antenna. For example, if the phased array antenna contains 12 radiating elements (eg, a half dipole), the array steering vector

Contains 12 vector elements. Array steering vector

Is the constant C in FIG. 7 and is used to direct the beam of the phased array antenna towards the azimuth angle θ. Each vector element is given by:
[0061]
[Expression 1]

Where k is 2π divided by wavelength, and m is a number from 0 to M that defines the number associated with the radiating elements of the phased array antenna (eg, 0 to 11 for a 12 element antenna). Where d is the separation between the radiating elements of the phased array antenna (preferably one-half wavelength) and θ is the azimuth angle of the formed antenna beam.
[0062]
Array steering vector so that the complete vector is combined to form the angle of arrival θ of the received signal in the receive beam

Are vector elements corresponding to a constant C as shown in FIG. In this case, θ is the angle with respect to the preferred reference direction of the phased array antenna. Array steering vector

Complex conjugate transpose of is a row vector

It is.
[0063]
product

Is again a column vector with one vector element for each radiating element of the phased array antenna. product

Is a single point, ie a scalar, determined in step S214 (FIG. 16) to give the value of the angular power spectrum P (θ) at the angle of arrival θ. Therefore, the angular power spectrum P (θ) is indicated by G1 in FIG. 18, and is calculated as follows:
[0064]
[Expression 2]

Where

Is the array steering vector,

Is the received signal vector,

Is the spatial covariance matrix of the received signal, and H is the complex conjugate transpose.
[0065]
In the above formula for calculating the array steering vector, it is assumed that radiating elements arranged at intervals of one-half wavelength are arranged linearly. However, those skilled in the art will understand how to calculate the array steering vector for radiating elements arranged along a curved path. Advantageously, three slightly “curved” antenna arrays can be used in the antenna system shown in FIG. In practice, the antenna array may be fairly “curved” to form a circle (eg, FIG. 6). Those skilled in the art will appreciate that array steering vector amplitude control and phase control are advantageously used in the calculation of array steering vectors for such highly curved arrays of radiating elements. .
[0066]
In order to improve performance, the angular power spectrum is determined by averaging repeated measurements. In FIG. 17, in step S211, an array steering vector is prepared and the direction of the antenna beam is determined. Multiple measurements are performed by repeating steps S215A and S215B. Within this loop, in step 216 the received signal vector

Are repeatedly measured, and the covariance matrix R is repeatedly obtained and stored in step S217. Next, an average covariance matrix is obtained in step S218, and an angular power spectrum P (θ) is obtained in step S214. When the average is obtained in this way, it is repeated several times over a time interval for each predetermined direction θ. Thus, the fast fading phenomenon is averaged. This period must be short enough that the position of the mobile remote station 2 does not change enough to allow the mobile remote station 2 to change the beam located during this average period. This period is preferably longer than the channel coherency time that averages the fast fading effect. The channel coherency time is not strictly and universally defined, but may be regarded as a time that is proportional to and approximately equal to the inverse of Doppler spread.
[0067]
Doppler diffusion is more strictly defined. Due to the relative speed between the base station and the mobile remote station, the reception frequency is physically shifted with respect to the transmission frequency. Doppler spread is twice this frequency shift. For example, Doppler frequency shift is the ratio of relative velocity to wavelength (possible units such as ratio of meters / second divided by meters or ratio of feet / second divided by feet). If the mobile remote station is traveling at 13.9 meters / second (about 50 km / h) and the wavelength is about 0.15 m (for example, a 2000 MHz signal with a speed of light equal to 300 million meters per second), the Doppler frequency shift is 92.7 Hz The Doppler spread is 185 Hz and the channel coherency time is about 5.4 milliseconds. When the relative speed is 40 m / sec (about 144 km / h), the channel coherency time is about 1.9 ms, and when the relative speed is 1 m / sec (about 3.6 km / h), the channel coherency time is about 75 It can be easily verified that it is milliseconds.
[0068]
The average time interval is preferably set to a value that is greater than the reciprocal of Doppler spread and less than the time that a mobile station moving at the expected angular velocity travels one half the beam width of the base station antenna system. . The base station knows the range of the remote station or can estimate the range from the signal strength. The base station is configured to communicate with a mobile station that can move at speeds up to a predetermined speed. When the mobile station is moving around the base station in the radial direction, the value obtained by dividing this speed by the range can be regarded as the angular speed. If the average interval is set to one half of the beam width divided by the angular velocity, the position of the mobile remote station 2 is sufficient to allow the mobile remote station 2 to change the beam located during this average period. An estimate of time that does not change can be obtained.
[0069]
The period during which power P (θ) is averaged is typically much longer than the channel coherency time. For example, in a wideband CDMA system operating in an environment with a high incidence of multipath reflections (eg, an urban environment), the average period may be tens of time slots. For indoor environments where the incidence of multipath reflections is high, movement can be much slower and the average duration can be much longer.
[0070]
The base station calculates an angular power spectrum and determines whether a sharp peak is shown in the power spectrum. When sharp peaks are shown, the angular position of each peak is determined. When the power spectrum is spread and no sharp peak is shown, the base station first determines the angle spread AS by finding the angle at which the received angle power spectrum exceeds a predetermined threshold (G2 in Figure 18). Ask for. The threshold can also be adapted based on the radio environment (eg, signal density) detected by the base station 1.
[0071]
Sharp peaks in the angular power spectrum can be detected, for example, by using a two-threshold test. For example, determine a first continuous angular range (in degrees or radians) where the power spectrum exceeds a first threshold G3. A second continuous angular range is then determined in which the power spectrum exceeds a second threshold G2 (which is smaller than the first threshold G3). A peak is indicated when the ratio of the first angular range divided by the second angular range is less than a predetermined value.
[0072]
When a peak is indicated, angular diversity management (ie, management of beam arrival direction) is invoked and possibly beam width management is invoked. The sharpness of the spectral peak can be determined by comparing the angular power spectrum to two threshold values. For example, in FIG. 18, three peaks exceed threshold G2, but only two peaks exceed threshold G3. The angular diffusion of the single peak determined according to the threshold value G2 is wider than the angular diffusion of the single peak determined according to the threshold value G3. The ratio of the angular spread of a single peak determined by G3 to the diffusion determined by G2 is a measure of peak sharpness. Alternatively, in the angular power spectrum, the threshold for measuring the angular power spectrum may be appropriately moved until there are at most two peaks exceeding the threshold indicating the directions of paths 3 and 5 it can. For example, when two sharp peaks occur in the angular power spectrum, and the base station transmits two beams, the base station is in the direction of these peaks (ie, the power spectrum exceeds the threshold G3 2 Two different angular directions) are defined in the angular directions of passes 3 and 5 (FIG. 1). This is called the arrival diversity angle. The base station directs the trackable beam in a direction along paths 3 and 5, respectively, or selects a constant beam that points in a direction along paths 3 and 5, respectively. Those skilled in the art will understand how to extend angular diversity management to more than two beams.
[0073]
In some cases, the angular power spectrum includes more than two angular positions corresponding to each peak of the angular power spectrum. When the base station has two beams, it is based on (1) avoiding the angle where co-channel users are positioned to minimize co-channel interference throughout the system, or (2) the amplifier of the transmitting station The first and second angular positions are selected from three or more angular positions so as to balance the power distribution at.
[0074]
The beam width in a phased array antenna is generally selectable by adjusting the amplitude of the elements of the beam steering vector (eg, vector C in FIG. 7). When the antenna system includes a phased array antenna with a controllable beamwidth and the spectral peak is sharp, the base station will concentrate the transmit power in the direction along paths 3 and 5, respectively. , Assuming the antenna system, set the beam as narrow as possible, or select the beam as narrow as possible. Since the spectral power peaks are sharp, paths 3 and 5 are expected to have good transmission characteristics.
[0075]
On the other hand, if the angular power spectrum is very diffuse and the peak is weak or not shown, then the angular spectrum where the power spectrum exceeds a threshold (eg, G2 in Figure 18), or at least the angular power A rough angular window is determined based on the continuous angular range necessary to cover the peaks where the spectrum exceeds the threshold. In such a case, in a preferred embodiment of the invention, the beams are selected such that the sum of the beam widths of all the beams used for downlink transmission is approximately equal to the angular spread AS.
[0076]
When the antenna system includes a phased array antenna with a controllable beam width, but the spectral peak is not sharp, the base station first determines whether the power spectrum has an angular range that exceeds a threshold, Or at least, the continuous angular range required to cover the peak whose angular power spectrum exceeds the threshold is determined as angular diffusion. The base station then sets the beam width of the beam to approximately cover angular spread, or selects the width that substantially covers angular spread as the beam width of the beam. This is called angular power diversity management or beam width management. For example, a two-beam base station attempting to cover angular spread will select approximately one half of the angular range as the beam width of both beams, and the base station will substantially reduce the angular spread of the two beams. Turn to cover.
[0077]
As will be appreciated by those skilled in the art, expansion to more beams is easy. For example, when the base station has the function of performing beam space-time encoding with a 4-beam base station, a beam width of about one quarter of the angular spread is selected for each beam. In this way, the downlink transmission is spatially matched to the channel. It is advantageous to match the orthogonal beam coverage to the angular spread of the channel so that maximum angle diversity gain is obtained. However, usually 2 to 4 beams are appropriate.
[0078]
When the base station has an antenna system with multiple constant beams (similar to a six-corner antenna), and the angular power spectrum is spread and the angular spread AS is a single beam beam When exceeding the width, in a desirable variant of the invention, two adjacent beams are combined into a single wider beam to better match the wireless channel (two 60 ° beams are combined into a single Combined with 120 ° beam). In such a case, two adjacent beams are used as a single wider beam using the same pilot code or orthogonalization code. In a constant beam base station, it is advantageous if the number M of beams that can be generated is large enough to achieve a high beam resolution (eg M> 4, preferably at least 8). When a wider channel is needed to better match the channel, two adjacent beams can be combined.
[0079]
The present invention is well suited for base stations in which the antenna system uses data beamforming techniques with phased array antennas (eg, antenna 20 of FIG. 7 and antenna 40 of FIG. 10). When using digital beamforming techniques, the apparent number of elements in the antenna array (ie, the apparent aperture size) can be electronically determined by using zero weighting on some elements according to the available angular spread. Can be adjusted. In this way, the base station can easily adapt the beam width to match the angular spread. This beam width control acts as an open loop control system.
[0080]
In another aspect, beam hopping techniques are used when the angular power spectrum exceeds a threshold in a large angular range. The beam hopping technique is a technique that sequentially covers angular diffusion. For example, when a transmit beam in any one time slot does not cover angular spread, this angular spread can be covered during subsequent time slots. Consider an exemplary system with a two-beam base station that can form a 30 ° beam when the angular spread covers 120 ° (ie, the width of four beams). In a beam hopping system, the base station forms two 30 ° beams for transmission during the first time slot to cover the first 60 ° sector with 120 ° angular spread, Two other 30 ° beams are formed for transmission during the second time slot to cover the remaining 60 ° sectors of angular spread.
[0081]
Beam hopping greatly improves performance in wireless environments with large angular spread. In frequency division duplex portable radio systems, it is known that, when angular spread is increased, down link performance is degraded due in part to increased angular uncertainty in optimally selecting the transmission direction. Yes. In a frequency division duplex system, the uplink direction determined to have good power transmission capability (weak attenuation) can cause deep fading due to different carrier frequencies in the case of downlink transmission There is.
[0082]
If the angular spread is large in a wireless environment, the number of possible directions for downlink transmission also increases. Rather than selecting the two best directions, spatial diversity is achieved by sequentially forming the downlink beam to cover all possible directions where the angular power spectrum exceeds the threshold and in some cases good Is realized. This is particularly important in microcells or picocells where angular spread can cover entire sectors or entire cells.
[0083]
If the remote station 2 has constant or low mobility, beam hopping has other advantages over choosing the two best directions. If the best two directions are selected as the beam transmission direction for a number of consecutive bursts, if the selected direction is wrongly selected (for example, the uplink is good but the fading occurs on the downlink) A significant penalty (with respect to data loss). However, by hopping the beam over a group of possible directions, loss of data from one direction that has been found to cause deep fading only occurs for a limited duration (eg, only one time slot). . This angular diversity tends to “whiten” errors caused by selecting a bad transmission direction.
[0084]
In addition, co-channel interference to other remote stations generated during beam hopping transmission tends to be whitened by the spatial spreading of the transmitted signal. Since high bit rate connections are made with high beam power, co-channel interference is particularly troublesome when high data bit rates are required. When a large amount of beam power is used on a high bit rate connection, noticeable color interference occurs when the base station uses a non-hopping scheme for beam selection (not uniformly distributed).
[0085]
In FIG. 19, another aspect of the present invention includes a base station 210 and a remote station 230 as described with reference to FIG. In this aspect, base station 210 includes weighting amplifiers 102 and 104 that apply respective weights W1 and W2 to respective supply signals CH1 and CH2. In this aspect, the weights W1 and W2 are complex or at least a phase / amplitude pair that adjusts both the amplitude and phase of the signals transmitted from the antennas 16 and 18. Alternatively, weighted signals can be transmitted from directional antennas 106 and 108. FIG. 19 shows diplexers 16D and 18D coupled between the weighting amplifiers and the respective antennas to duplex the antennas so that the antennas can be used in both uplink and downlink transmission modes. However, a separate base station antenna may be used to receive the uplink signal.
[0086]
In a preferred variant, one antenna is used as a reference with a corresponding weight set to 1 + j0 (or amplitude = 1, phase = 0 °). The other weight is determined relative to the reference weight. In general, base station 210 may use more than one channel, each with an antenna, diplexer, weighting amplifier, and all associated encoders. When M is the number of transmitting antennas, since only the difference information (ie, weight) needs to be obtained, the number of weights that must be obtained is M-1. The following description focuses on two transmitting antennas (M = 2) so that only one complex weight need be obtained without losing generality.
[0087]
In FIG. 19, the remote station 230 includes a remote station antenna 232, a remote station receiver 234 coupled to the remote station antenna 232 through a diplexer 233, a signal measurement circuit 238, and a processor 240. Receiver 234 forms a circuit for remote station 230 to receive the first and second signals from the first and second transmit antennas. The signal measurement circuit 238 and the processor 240 and the control module described herein allow the remote station 230 to obtain channel state information based on the received first and second signals and to obtain the channel state information from a plurality of channel states. A circuit for dividing into information segments is configured. The signal measurement circuit 238 measures the signal strength (and phase) received from each of the plurality of orthogonal antennas, and the processor 240 obtains channel state information. The signal measurement circuit 238 measures the received instantaneous signal strength (and phase), or in other variations, measures the average received signal strength and phase at a reference time.
[0088]
The processor obtains channel state information from the information provided from the signal measurement circuit 238. The processor selects a reference signal from signals received from each different antenna. The processor divides the received signal strength (and phase) determined by the signal measurement circuit 238 by the selected reference signal strength (and phase) for each of the plurality of antennas. This ratio is determined as the ratio of complex numbers (ie phase / amplitude pairs). The ratio for the reference antenna is 1 + j0 by definition. If there are two antennas, there is only one ratio to be transmitted, i.e. the ratio of the reference antenna is a constant reference.
[0089]
The processor 240 obtains channel state information from the normalized ratio. Each ratio includes both amplitude and angle information. The purpose of this process is to adjust the phase of the signals transmitted from the two antennas (or more) so that they are constructively augmented at the remote station 230. To ensure constructive reinforcement, it is desirable to delay or advance the phase of the signal transmitted from each antenna relative to the reference antenna. For example, if the first antenna 16 is a reference antenna, the angular portion of the ratio for the signal received from the second antenna 18 is further examined. If this angle is 45 ° ahead of the reference antenna, a 45 ° delay must be introduced at the transmitter of the second antenna 18 to achieve constructive reinforcement at the remote station 230. Thus, the processor 240 adds a desired additional delay to the phase of the initial transmission signal and subtracts 360 if the result of this addition is greater than 360 to achieve constructive reinforcement at the remote station 230. Determine the amount of phase delay or advance required. This phase angle then becomes the phase angle transmitted as part of the channel state information.
[0090]
The processor 240 also determines the amplitude portion of the channel state information. The purpose here is to emphasize the antenna that has the best path from the antenna to the remote station 230 (ie, the path with the weakest attenuation). The total power transmitted from all antennas may be considered constant here. The problem to be solved by the amplitude part of the channel state information is how to divide the total transmit power.
[0091]
For this purpose, the processor 240 measures the channel gain (reciprocal of attenuation) for each antenna by calculating the ratio of the received power divided by the power received with the reference signal. The received power is the square of the signal strength measured by the signal measurement circuit 238 (ie, P_i = (a_i)². In this formula, a_iIs the signal strength from antenna i). The signal transmitted through each different antenna or antenna beam is the signal power P_TXThis signal contains mutually orthogonal pilot codes modulated on the signal transmitted at. The remote station has a composite channel impulse response, i.e. H_i = a_iexp (φ_i) As the ratio of the received signal divided by the received reference signal. In this case, φ_iIs the relative phase of the signal under measurement and a_iIs the relative signal strength. Then P_iIs a_iIs calculated as the square of. The relative channel response for each channel is measured with respect to the received power. If only one bit is reserved for amplitude feedback information on the uplink outgoing channel, this bit preferably transmits 80% of the total power to the remote station 230 via the antenna with the path with the weakest attenuation and attenuates Orders 20% of the total power to be transmitted by the antenna with the most intense path.
[0092]
If 2 bits are reserved for amplitude feedback information in the uplink outgoing channel, these bits can define 4 amplitude states. For example, the processor 240 calculates the ratio of the path attenuation from the antenna 16 to the path attenuation from the antenna 18, and then slices this ratio according to a predetermined range of values that this ratio can take. In this slicing process, four sub-ranges are defined to identify which of the four ranges the calculated ratio fits. Each subrange provides a desired division of the total power transmitted by the two antennas 16 and 18, eg, 85% / 15%, 60% / 40%, 40% / 60%, and 15% / 85%, respectively. It is defined as The two bits thus encode one of these divisions as the desired division in the total power transmitted by the two antennas.
[0093]
Those skilled in the art will appreciate that in view of these teachings, the amplitude portion of the channel state information can be calculated by various means. In this specification, the table reference means will be described, but other means for calculating the division of the total power to be transmitted are also equivalent. It will be appreciated that more than two bits may be used to define the power split.
[0094]
The processor 240 also partitions channel state information (including the amplitude and phase angle portions described above) into a plurality of channel state information segments based on the configuration. Remote station 230 further includes a transmitter 242 that transmits a plurality of channel state information segments to base station 210.
[0095]
The channel state information to be transmitted is a complex coefficient in the form of phase and amplitude information and is transmitted from the remote station 230 to the base station in several segments (N segments) carried in corresponding slots in the uplink outgoing channel. Need to send to 210. The partitioning of N slots into N1 and N2 (N = N1 + N2) is done such that the first N1 slots carry phase information and the remaining N2 slots carry amplitude information. Basically, N1 and N2 can be chosen arbitrarily, but a common value for these parameters may be N1 = N2 = N / 2. Assume that each slot reserves K bits for carrying the corresponding information segment. The phase can be resolved with the accuracy of:
[0096]
[Equation 3]

The amplitude can be resolved with the accuracy of:
[0097]
[Expression 4]

Where A_maxIs the maximum amplitude.
[0098]
For example, assume that the number N of slots is 6, and that 3 slots are reserved for each of N1 and N2. Assuming that the number of bits K per slot is 1, the maximum amplitude A_maxIs 3 V. Then the phase and amplitude accuracy is φ_min = 45 °, amplitude A_minIs 0.375 V. However, when the number of bits K per slot is increased to 2, the accuracy of the phase and amplitude that can be transmitted is φ_min = 5.6 °, amplitude A_minIs 0.05V.
[0099]
In general, the quantized or truncated form of strict channel state information is formed such that each bit of the truncated form exactly matches the number of bits available on the uplink outgoing channel. Termination mode is phase segment φ_i(I = 1 to N1), each segment is sent in hierarchical order with the most significant bit (MSB) sent in the first segment and the least significant bit (LSB) sent in the last segment The Similarly, each amplitude segment A_i(I = 1 to N2) includes the quantized segment or truncation segment of the next channel state information (ratio), and each amplitude segment is transmitted in hierarchical order.
[0100]
In this aspect of the invention, the phase angle accuracy and amplitude accuracy used in forming the downlink beam is improved, thereby improving the downlink performance of mobile communications. This aspect is particularly suitable for low mobility environments and is suitable for high bit rate applications in indoor and pedestrian environments. This aspect is particularly suitable for high bit rate wireless data applications for laptop computers.
[0101]
For example, assume that the remote station is moving at a speed v of 1 m / s (3.6 km / h) and the carrier frequency is 2 gigahertz (λ = 0.15 m). Maximum Doppler frequency f_DIs v / λ and the channel coherency time T_CIs calculated as:
[0102]
[Equation 5]
T_C = 1 / (2f_D) = λ / (2v) = 75 milliseconds
The channel state information can be assumed to be stable (almost constant) for a period equal to TC / 10, so that the channel state information is transferred from the remote station 230 to the base station 210 during this 7.5 millisecond stable period. Can be sent to. The wideband CDMA (WCDMA) standard defines a slot duration of 0.625 milliseconds, so that channel state information can be sent back to the base station using 12 slots.
[0103]
There are several ways to pack the channel state information into the uplink slot. Table 1 shows an example based on only one bit per slot (K = 1). In Table 1, 3-bit precision is used for both phase angle information and amplitude information. The phase angle is transmitted in the first six slots, and the amplitude information is transmitted in the last six slots. In both cases, the most significant bit is transmitted first. In slot 1, the most significant bit of the 3-bit phase angle is transmitted. In slot 2, the same bits are repeated to improve reliability. Thereafter, the remaining phase angle is transmitted. The amplitude information bit is transmitted in the same manner. The first bit indicates the phase angle with an accuracy of 180 ° as in the case of 1 bit. After slot 3, the phase angle is transmitted with an accuracy of 90 ° as in the case of 2 bits, and after slot 5, the phase angle is transmitted with an accuracy of 45 ° as in the case of 3 bits. . Assuming that the phase angle changes approximately 360 ° during the channel coherency time, in the above example, the phase angle changes approximately 36 ° in the 7.5 millisecond period required to transmit 12 slots. . This corresponds to the phase accuracy available with 3 bit data (45 °).
[0104]
After slot 7, the amplitude information is transmitted with an accuracy of 0.5 of the maximum amplitude as in the case of 1 bit. After slot 9, the amplitude information is transmitted with an accuracy of 0.25 of the maximum amplitude as in the case of 2 bits, and after slot 11, the accuracy of the amplitude of 0.125 of the maximum amplitude as in the case of 3 bits. Sent by.
[0105]
[Table 1]
Format for transmitting channel state information to the base station

[0106]
In general, phase information is more important than amplitude information. Optimal maximum ratio combining is only about 1 dB better than equal gain combining used in the absence of amplitude information feedback, thus increasing the allocation to the phase bit (N1) and to the amplitude bit (N2) It is advantageous to reduce the allocation. For example, three phase bits and two amplitude bits can be assigned so that feedback channel state information can be transmitted in WCDMA format without 3.125 ms of redundancy.
[0107]
The combination of allowable feedback capability (eg, 1 bit / slot or more), feedback reliability (eg, repeat or redundant bits) and feedback accuracy (eg, number of phase angle bits and amplitude bits) depends on the application and environment Determined. For example, a known SECDED (signal error correction, double error detection) format 3-bit check code can be added to 8-bit information so that a redundancy error check can be performed. One of ordinary skill in the art will understand how to match feedback capabilities, feedback reliability, and feedback accuracy to the application and environment in view of these teachings.
[0108]
The processor 240 (FIG. 19) partitions the channel state information into a plurality of channel state information segments according to a format defined by the system mode. In practice, the system may have multiple modes, each defining a different format. For example, in one mode, only phase angle information commanding equal amplitudes can be transmitted to each antenna, and in other modes, 3-bit phase angle information and 1-bit amplitude information can be transmitted. Transmitter 242 then encodes a plurality of channel state information in the uplink outgoing channel and transmits the encoded information to base station 210 through diplexer 233 and antenna 232.
[0109]
In a variation of this aspect, there are several modes that require from 1 bit to 20 bits, for example, to represent channel state information in the uplink outgoing channel. In this variation, the processor 240 determines the rate at which the channel state information changes based on the change between each update. When this speed is slow, indicating that the remote station is moving slowly or stationary, the feedback mode will accordingly send more data bits of channel state information to the base station. The mode is changed to enable transmission. However, when the channel state information changes rapidly and indicates that the remote station is moving at high speed, the feedback mode is accordingly changed to a mode that reduces the number of bits transmitted each time the channel state information is updated. Be changed.
[0110]
Base station 210 receives information encoded on the uplink outgoing channel, and receiver / detector 220 decodes a plurality of channel state information segments. Processor 220P then reconstructs channel state information from the received plurality of channel state information segments and generates weights W1 and W2. The weights W1 and W2 are respectively weighted to weight the first and second supply signals CH1 and CH2 sent to the first and second antennas 16 and 18, respectively, based on the reconfigured channel state information. Provided to amplifiers 102 and 104.
[0111]
Two variations of this embodiment can be implemented with the processor 220P. First, the processor can collect all segments and reconstruct total channel state information before forming weights W1 and W2 applied to amplifiers 102 and 104. Alternatively, as the channel state information, the phase angle is first transmitted to the base station, and the most significant bit is first transmitted within the phase angle segment. The values of W1 and W2 can be updated in the processor as each bit is received to provide faster feedback to amplifiers 102 and 104. This results in a substantially larger feedback bandwidth.
[0112]
In FIG. 20, the method implemented on processor 240 typically includes several steps implemented on a processor having software modules and / or logic. However, those skilled in the art will appreciate that these steps can be implemented in a processor using an ASIC or other custom circuit.
[0113]
In step S2, the processor receives the received signal strength and phase (complex number) obtained by the signal measurement circuit 238 for each of the plurality of antennas. In step S4, the processor selects one of the received signals as a reference signal. This selection may be arbitrary, or the signal with the greatest phase lag (the least likely to need to be delayed) may be selected. In step S6, the processor divides the received signal strength and phase (complex number) determined by the signal measurement circuit 238 by the received reference signal strength and phase (complex number). The ratio for the reference antenna is 1 + j0 by definition. When there are two antennas, there is only one ratio to be obtained and the reference antenna ratio is a constant reference.
[0114]
In step S8 (FIG. 20), the processor 240 determines the amount of phase lag or advance required at each transmitting antenna to provide constructive reinforcement at the remote station 230. When the reference signal is selected as the signal having the maximum delay, the remaining signal can be phase-matched with the reference signal by adding a delay at the antenna. In step S8, the necessary additional delay is determined, but 360 is subtracted if the phase of the non-reference signal plus the additional phase delay is greater than 360 °. This phase angle is then the phase angle transmitted as part of the channel state information. Those skilled in the art will appreciate that in view of these teachings, step S8 can be performed at the base station so that only the phase angle of the channel impulse response need be transmitted on the uplink outgoing channel.
[0115]
In step S10, power management information for determining a transmission distribution (assignment of total power among the transmission antennas) is obtained. Those skilled in the art will appreciate that the amplitude portion of the channel state information can be calculated by various means in view of these teachings. In this specification, the table reference means will be described, but other means for calculating the division of the total power to be transmitted are also equivalent.
[0116]
For example, the relative amplitude and phase of the signal from each antenna can be transmitted on the uplink outgoing channel for further processing at the base station. Alternatively, the remote station can generate a signal indicating the desired power distribution in step S10. If only one bit is reserved for amplitude feedback information on the uplink outgoing channel, this bit preferably transmits 80% of the total power from the antenna with the weakest path to the remote station 230. Command the antenna with the most intense path to transmit 20% of the total power. If two bits are reserved for amplitude feedback information on the uplink outgoing channel, these bits can define four amplitude subranges. For example, 85% / 15%, 60% / 40%, 40% / 60%, and 15% / 85% can be defined, respectively. The two bits thus encode one of these subranges as the desired division in the total power transmitted by the two antennas. Those skilled in the art will appreciate extensions to more antennas, or extensions that use more bits to represent the amplitude portion of the channel state information. The exact nature of the table look-up or other means depends on the number of bits reserved to hold the amplitude portion of the channel state information in the uplink format.
[0117]
In step S12, the channel state information is partitioned and changed to the format described herein (eg, Table 1). In step S14, each segment is sequentially transmitted to the base station on the uplink outgoing channel. From there, the respective weights for the antennas are recovered and added to amplifiers 102 and 104 (FIG. 19).
[0118]
In frequency division duplex systems where uplink and downlink communications are performed over different frequencies, the two directions are based on different frequencies, so the uplink channel state is determined from the uplink information. It is impossible to determine exactly. The system has the advantage of measuring the downlink channel condition from the downlink data and then sending commands on the uplink outgoing channel to adjust the amplitude and phase of the transmitted downlink signal. is doing.
[0119]
In FIG. 21, the antenna 1 of the base station is a sector coverage type antenna. Antenna 1 transmits the signal directly to remote station 2 over path 3. However, another multipath signal is reflected from the radio wave scatterer 4 and travels on the multipath 5. As a result, the remote station 2 receives two replicas of this signal at slightly different times. In FIG. 22, the two replicas are shown as signals received at time nT and time nT + τ, where τ is the additional time caused by the additional length of multipath 5 relative to direct path 3 It is a delay. The multipath delay may be a delay that causes destructive interference between the two signals received on the two paths. More multipath signals may be generated by other radio scatterers.
[0120]
Conventional rake receivers correlate local signals (e.g., CDMA signal spreading codes) with received signals that include replicas of signals received using different delays. With the correct delay, each signal is coherently combined to reinforce energy. When a local signal (eg, a desired spreading code) is correlated with a signal from a desired signal path, the local signal is also associated with any other signal replica (eg, a signal replica from a signal path with a different delay). Correlated. Terms corresponding to correlations with other signal replicas are unnecessary terms and tend to degrade system performance. Unnecessary correlation terms cause loss of orthogonality between various users with different codes, so that co-channel users begin to interfere with each other. This degradation effect is noticeable in the case of short spreading codes normally used on high bit rate links.
[0121]
The present invention operates the rake receiver differently than before. The present invention uses beam forming to separate different signal paths and apply pre-transmission time offset compensation to each signal replica (eg, each beam) so that all signal replicas arrive at the receiver simultaneously. . In this way, the receiver actually processes multiple signals (eg, paths 3 and 5 in FIG. 1) and processes them only with a one-tap channel, even though they are coherently combined. Appears to receive the received signal. This avoids loss of orthogonality and minimizes or eliminates cross-correlation terms that may otherwise degrade system performance.
[0122]
In aspects of the invention, the desired data is included in more than one space-time encoded signal. Each signal is identified by a unique mutually orthogonal signature code. If one of the space-time encoded signals is significantly delayed with respect to the other space-time encoded signals, the orthogonality of the signature code may be reduced. Preferably, the shortest path signal is delayed so that the longer path signal arrives at the remote station 2 at the same time.
[0123]
In FIG. 23, the exemplary system includes antenna 1 and remote station 2. The exemplary antenna 1 may be a Butler matrix multibeam antenna array or any other multibeam antenna array. The desired data in this example is encoded as two space-time encoded signals I2 and I5. Space-time encoded signals I2 and I5 are transmitted in beams D2 and D5, respectively. Beam D5 transmits signal I5 directly over path 3 to remote station 2. Beam D2 transmits signal I2 to remote station 2 over indirect multipath 5.
[0124]
FIG. 26 illustrates an exemplary encoder that generates space-time encoded signals I2 and I5. FIG. 26 is similar to FIG. However, antennas 16 and 18 in FIG. 2 have been replaced with the multi-beam antenna of FIG. 23, and a programmable delay line (eg, a selectable multi-tap delay line) is provided between multiplier 14 and the multi-beam antenna. Coupled between. The multiplier 12 encodes a signal CH1 having a signature code (OC) that is orthogonal to the signature code encoded by the multiplier 14 into the signal CH2. The signature code may be a training sequence, a pilot code, or a spreading sequence that are orthogonal in various forms. As long as these signature codes are orthogonal to each other, the remote station 2 uses these signature codes to separate the signal received on the direct path from the beam D5 from the signal received on the indirect path from the beam D2. One skilled in the art can generalize the two beams and corresponding space-time encoded signals shown in FIG. 23 and FIG. 26 to more than two beams, and time synchronization of all signals. It will be appreciated that additional programmable delay lines may be required to take.
[0125]
As shown in FIGS. 24 and 25, the direct signal from beam D5 is received at remote station 2 earlier by time τ than the indirect signal from beam D2 is received. In order to maintain the best orthogonality between signature codes, it is desirable to align each signal in time. A receiver (sometimes a base station receiver, and sometimes a remote station 2 receiver, as described below) determines the time delay τ required to align the signals. The last signal received at remote station 2 (eg, signal I2) can be considered a reference space-time encoded signal. The remaining signal can then be considered as at least one remaining space-time encoded signal (eg, signal I5). In this aspect, at least one remaining space-time encoded signal is delayed on a base station programmable delay line (see FIG. 26) before being transmitted. Each signal is delayed by a sufficient delay so that at least one remaining space-time encoded signal is aligned in time with the reference signal when received at the remote station. In the example shown in FIG. 23, the last signal received by the remote station 2 is the signal I2 because the multipath 5 is long. Signal I5 needs to be delayed so that signal I2 arrives at remote station 2 at the same time as signal I2 arrives at remote station 2.
[0126]
In both space-time diversity techniques (FIG. 2) and beam-space diversity (FIG. 23), it is important that the remote receiver separate signals CH1 and CH2 as described above. This is done by using various forms of orthogonal signature codes. The difference in arrival time at which signals from the two paths, that is, the direct path 3 and the multipath 5 arrive at the remote station 2 is called delay spread. When delay spread is not present or minimal, signature code orthogonality is preserved. However, in frequency selective channels where there is significant delay spread of the signature code, orthogonality between each channel may be lost, making it difficult for the remote station 2 to separate the signals carried on each channel. . Most common coding sequences have a small value or zero only for a given phase relationship between each signature code, and a non-ideal cross-correlation function (CCF) with non-zeros for the other phase relationships. ).
[0127]
The multiple space-time diversity signals transmitted to the remote station 2 over the multipath channel are each subjected to different delays. Since the value of CCF at a given position out of phase is typically non-zero and varies from position to position, the effect of each different path delay imposed by the radio channel on the transmitted signal is as follows: The orthogonality between the signature codes used to separate the signals is reduced. Loss of orthogonality reduces the diversity gain that would otherwise be obtained by space-time code transmission of signals between the base station and the remote station in a wireless communication system.
[0128]
In this aspect, a multi-beam antenna array associated with a base station receives uplink signals from a remote station of interest on each of the plurality of beams of the multi-beam antenna array. The uplink signal may be a pilot signal, an uplink outgoing channel, or any other uplink channel that identifies the source of the signal as a remote station of interest. The uplink signal is received as a plurality of signals derived from radio signals received on the corresponding plurality of beams of the multi-beam antenna array.
[0129]
The base station receiver separates, for each of the plurality of received signals, a signal component identified as a signal component originating from a particular remote station of interest by a signature code. The received signal component of each beam of the plurality of beams includes an identified signal replica of a particular remote station of interest in a particular time delay or delay spread relative to the reference beam signal component. The base station receiver processes multiple signal components from each beam, and receives the last received signal component and each of the other signal components received from each beam with a reference beam. The beam containing the delay spread necessary to align with the signal component is identified as the reference beam. When the base station serves multiple remote stations, this process can be repeated for each remote station or selected remote stations. The selected remote station may be a remote station having a high transmission power. High transmit power may be required, for example, due to high data rate requirements.
[0130]
FIG. 27 shows a typical channel impulse response or delay distribution profile 300 of a 16 beam base station system similar to the 8 beam base station system shown in FIG. The base station measures the delay spread τ associated with each beam of the multi-beam antenna. For a received signal having a signal strength that exceeds a threshold, an “x” indicates that the instantaneous and / or averaged signal strength exceeds a given threshold. The directions D3, D6, and D12 shown in 304, 306, and 308, respectively, (for example, delay τ_FourTo τ₆Including signals with minimal delay spread. When several potential directions are available, whitening of the resulting interference, equal distribution of power in multiple power amplifiers used by the base station, interference above average occurs for co-channel users Based on additional criteria such as avoiding possible directions, a preferred one of the available directions is selected. For example, if one or more low bit rate users are located in an area illuminated by a high power beam, the high power beam may cause interference to these low bit rate users There is. Depending on the preferred situation, beam hopping may be applied to achieve more effective interference whitening.
[0131]
In operation, the base station selects the direction with the minimum delay spread. For example, the base station may transmit at least two space-time encoded signals with corresponding ones of the at least two beams, at least two of the plurality of beams that can be formed by the multi-beam antenna array. Select a beam. The at least two beams include a reference beam and at least one remaining beam. The base station also determines from the delay distribution profile 300 a time delay corresponding to each of the at least one remaining beam that is used in programming the programmable delay line.
[0132]
The base station converts each signal of the at least two space-time encoded signals into at least two space-time codes so as to form a reference space-time encoded signal and at least one remaining space-time encoded signal. It encodes with the signature code orthogonal to each other signature code encoded in the encoded signal (see 12 and 14 in FIG. 26). In the example of FIG. 23, the reference space-time encoded signal can be considered as signal I2, and at least one remaining space-time encoded signal can be considered as signal I5. However, those skilled in the art will understand how to extend this aspect to more than two space-time encoded signals in view of these teachings.
[0133]
The base station delays each signal of the at least one remaining space-time encoded signal to form at least one delayed space-time encoded signal (eg, signal I5 in FIG. 26). The base station then selects at least 2 so that both the reference space-time encoded signal (eg, signal I2) and at least one delayed space-time encoded signal (eg, signal I5) arrive at the remote station 2 simultaneously. A reference space-time encoded signal and at least one delayed space-time encoded signal are transmitted in each beam of the two beams.
[0134]
This aspect does not depend on the feedback channel from the remote station to the base station. Instead, the transmission direction is selected by the base station only from the uplink measurements of normal outgoing signals. By averaging the uplink channel response over a long period to mitigate fast fading, the power response of the downlink channel response can be estimated. The uplink and downlink channels shown are contradictory in terms of power.
[0135]
However, in frequency division duplex (FDD) systems, feedback measurements can improve results at the expense of increased complexity. In frequency division duplex systems where uplink and downlink communications are performed over different frequencies, the two directions are based on different frequencies, so the uplink channel state is determined from the uplink information. It is impossible to determine exactly.
[0136]
In the above aspect, an aspect has been described in which the base station measures the uplink channel response instead of the downlink channel response. To obtain a complete downlink channel impulse response, measure the downlink channel directly and the downlink channel state with the feedback channel from the remote station performing the measurement (eg, delay The distribution profile 300) needs to be transmitted to the base station that needs it.
[0137]
The remote station is involved in or performs these functions rather than performing the calculations necessary for direction selection and delay at the base station. An agreed standard signal is transmitted from the base station to all remote stations, together with identifiers or signature codes encoded in each beam, such as mutually orthogonal pilot sequences or training sequences and spreading codes. The remote station then measures the channel impulse response (eg, delay distribution profile 300) and informs the base station of the preferred direction of transmission and delay.
[0138]
One of ordinary skill in the art, in view of these teachings, can measure channel performance in a two-step process. In the first stage, the base station determines an estimate of the impulse response of the uplink channel and uses this estimate instead of the impulse response of the downlink channel. The base station then adds the delay indicated by the first estimation process to the at least one remaining space-time encoded signal.
[0139]
In the second stage, the downlink channel is measured directly. An agreed standard signal is transmitted from the base station to all remote stations, together with identifiers or signature codes encoded in each beam, such as mutually orthogonal pilot sequences or training sequences and spreading codes. The remote station then measures the channel impulse response (eg, delay distribution profile 300) and informs the base station on the feedback channel of the preferred direction and delay of transmission.
[0140]
In FIG. 28, the uplink channel response is measured by the setup process S200, and the measured delay is set to control the downlink channel transmission. Process S200 includes measuring a channel response S202, selecting a beam to be used S204, determining a time delay for the selected beam S206, and a base station variable delay line (see FIG. 26). And S208 configured to impose the determined delay. The variable delay line can be constituted by a sequence of constant delay elements in which a plurality of taps are arranged between the elements. The delay line can be changed by selecting various taps as outputs using a switch. In step S204, the base station selects at least two of the plurality of beams formed by the multi-beam antenna array associated with the base station (however, only two beams are shown in FIGS. 23 and 26). Not shown). In these beams, corresponding at least two space-time encoded signals generated by the space-time encoder are transmitted (however, only two beams are shown in FIGS. 23 and 26). The at least two beams include a reference beam and at least one remaining beam. In step S206, the base station determines a time delay corresponding to each of the at least one remaining beam. In step S208, the base station sets a time delay corresponding to each of the at least one remaining beam in the variable delay line. Each variable delay line is coupled between a multi-beam antenna array and a space-time encoder (see FIG. 26).
[0141]
In FIG. 29, the time-alignment process S220 marks, in step S222, the space-time encoded signal of each selected beam is marked with a signature code that is orthogonal to all other beams and is selected in step S224. The transmitted beam is transmitted according to the obtained delay spread, and the delayed signal is transmitted to the base station in step S226. In step S222, the base station converts each signal of the at least two space-time encoded signals to at least two to form a reference space-time encoded signal and at least one remaining space-time encoded signal. Encoding is performed with a signature code orthogonal to each other signature code encoded in the space-time encoded signal. In step S224, the base station delays each signal of at least one remaining space-time encoded signal by a respective variable delay line to form at least one delayed space-time encoded signal. In step S226, the base station transmits a reference space-time encoded signal and at least one delayed space-time encoded signal in each of the at least two beams.
[0142]
In FIG. 30, the remote station using feedback process S240 measures the downlink composite channel state information and sends this information back to the base station. Process S240 receives at least two identifier signatures (eg, different pilot signals) from an antenna system associated with the base station, and obtains composite channel state information based on the received signals S244 Step S246 for dividing the composite channel state information into a plurality of channel state information segments, and Step S248 for transmitting the plurality of channel state information segments as sequences to the base station. The sequence of segments transmits the most significant bits of the phase angle before the least significant bits of the phase angle. The sequence of segments transmits the most significant bit of amplitude before the least significant bit of amplitude. The sequence of segments transmits the phase angle bits before the corresponding amplitude bits with the same level of significance. Note that in order to feed back channel impulse response measurements, each beam (or antenna) must be associated with a unique pilot signature that is orthogonal to all other pilot signatures.
[0143]
Those skilled in the art, in light of these teachings, have various system configurations with electrical circuits, special application specific integrated circuits (ASICs), or computers or processors that execute software programs or use data tables. It will be understood that the element can be realized. For example, the encoder 10, multipliers 12, 14, and amplifiers 102, 104 of FIG. 4, 5, 11, or 12 can be implemented with a circuit or ASIC, or possibly a software controlled processor, depending on performance requirements. Good. The beam former 40 of FIG. 11 is typically implemented with a circuit or ASIC, and the transducers 101, 103 and multipliers 105, 107 are typically implemented with a circuit or ASIC, but may be implemented with a software control processor. The various base station components 212, 214, 216, 218, 220, and 222 and the various remote station components 232, 234, 238, 240, and 242 of FIG. 14 are implemented in circuitry or ASIC, but are software It may be realized by a control processor. Although the various base station components 16D, 18D, 102, 104, 220, and 220P of FIG. 19 and the various remote station components 232, 233, 234, 238, 240, and 242 are implemented in circuits or ASICs. It may be realized by a software control processor. Those skilled in the art will appreciate that the various functions described herein can be implemented in a circuit, ASIC, or software controlled processor, depending on performance requirements.
[0144]
Having described a preferred embodiment of the novel closed-loop feedback system that improves downlink performance (exemplary and not limiting), those skilled in the art will make modifications in view of the above teachings. Note that and variations may be made. It should be understood that changes within the scope and spirit of the invention as defined by the appended claims may be made to the particular embodiments of the invention disclosed.
[0145]
Thus, while the invention has been described in conjunction with the details required by patent law, what is claimed desired protected by an open patent is set forth in the appended claims.
[Brief description of the drawings]
FIG. 1 is a schematic diagram of a wireless environment in which the present invention is used.
FIG. 2 is a block diagram of a known base station.
FIG. 3 is a block diagram of a known space-time encoder.
FIG. 4 is a block diagram of a base station apparatus according to an aspect of the present invention.
FIG. 5 is a block diagram of a base station apparatus according to another aspect of the present invention.
FIG. 6 is a schematic diagram of a known six-corner antenna system.
FIG. 7 is a schematic diagram of a known phased array antenna.
FIG. 8 is a schematic diagram in plan view of an exemplary three sector antenna system.
FIG. 9 is a schematic diagram of a known “Butler Matrix” antenna.
FIG. 10 is a schematic diagram of a dual beam phased array antenna.
FIG. 11 is a block diagram of a base station apparatus according to another aspect of the present invention.
FIG. 12 is a block diagram of a TDMA base station apparatus according to another aspect of the present invention.
FIG. 13 is a block diagram of a closed loop beam power management system according to the present invention.
FIG. 14 is a block diagram of a wireless system according to the present invention.
FIGS. 15-17 are flowcharts of a method for determining an angular power spectrum in accordance with the present invention.
FIG. 18 is a graph of angular power spectrum received and / or calculated according to the present invention.
FIG. 19 is a block diagram of aspects of the present invention.
FIG. 20 is a flowchart of a feedback control method according to the present invention.
FIG. 21 is a schematic diagram illustrating a multipath signal processed using a sector coverage antenna in accordance with the present invention.
22 is a graph showing the direct multipath signal of FIG. 21 received by a remote station.
FIG. 23 is a schematic diagram illustrating a multipath signal processed by the present invention using a multi-beam antenna covering a certain sector.
FIG. 24 is a graph illustrating the direct signal of FIG. 21 or FIG. 23 and a delayed replica of the direct signal received by a remote station.
FIG. 25 is a graph showing the multipath signal of FIG. 21 or FIG. 23 received by a remote station.
FIG. 26 is a block diagram of a base station apparatus having a programmable delay line according to an aspect of the present invention.
FIG. 27 is a graph showing a delay dispersion profile according to the present invention.
FIG. 28 is a flowchart of a setup method according to the present invention.
FIG. 29 is a flowchart of a time alignment method according to the present invention.
FIG. 30 is a flowchart of a feedback method according to the present invention.

Claims (38)

Receiving at least two space-time encoded signals from an antenna system associated with a first communication station;
Obtaining complex channel state information based on the received space-time encoded signal;
Partitioning the complex channel state information into a plurality of channel state information segments;
Transmitting the complex channel state information to the first communication station, wherein the plurality of channel state information segments include one or more phase segments and one or more amplitude segments;
In the step of transmitting the complex channel state information to the first communication station, the plurality of channel state information segments to transmit the one or more phase segments before transmitting the one or more amplitude segments. Send in order. The method of claim 1 , wherein in the step of transmitting the complex channel state information to the first communication station, a most significant phase segment is transmitted before a least significant phase segment. Said step of partitioning channel state information into a plurality of channel state information segments;
Determining the number of phase bits assigned to the phase information according to the operating mode;
Rounding and truncating the correction phase angle to fit the number of phase bits;
Determining the number of amplitude bits assigned to the amplitude information according to the operating mode;
The method of claim 1 including rounding down the correction amplitude according to the number of amplitude bits.   2. The most significant bit of the correction phase angle is transmitted prior to transmitting the most significant bit of the correction amplitude in the step of transmitting the plurality of channel state information segments to the first communication station. The method described.   The most significant bit of the correction phase angle is transmitted before transmitting the least significant bit of the correction phase angle in the step of transmitting the plurality of channel state information segments to the first communication station. The method according to 1. Receiving the plurality of channel state information segments;
Reconstructing complex channel state information from the received plurality of channel state information segments;
The method of claim 1, further comprising: weighting first and second supply signals sent to first and second antennas, respectively, based on the reconstructed complex channel state information.   The method of claim 1, wherein in the step of transmitting the channel state information segment to the first communication station, a plurality of channel state information segments are sequentially transmitted over a period based on channel coherency time. The antenna system includes a multi-beam antenna array;
Receiving the first and second space-time encoded signals from the first and second beams of the multi-beam antenna array, respectively, in the step of receiving;
The method of claim 1, wherein in the step of obtaining complex channel state information, the complex channel state information is obtained based on received first and second space-time encoded signals. The first communication station determines an angular power spectrum of a signal from a second communication station that defines first and second peaks at first and second angular positions, respectively.
Transmitting first and second space-time encoded signals by the first and second beams, respectively, such that the first and second beams are directed toward the first and second angular positions, respectively. The method of claim 8, further comprising: The antenna system includes a multi-beam antenna array, the method comprising:
Transmitting at least two space-time encoded signals in each beam of a multi-beam antenna array together with a signature code encoded in each signal of at least two space-time encoded signals, the signature comprising: Each of the codes is substantially orthogonal so that the second communication station can separate and measure the channel impulse response corresponding to each space-time encoded signal;
Said second communication station measuring a channel impulse response for each of the space-time encoded signals including a selected set of least attenuated signals and the remaining set of most attenuated signals; ,
The method of claim 1, further comprising: transmitting from the second communication station to the first communication station a signal indicative of the selected set of least attenuated signals. The antenna system includes first and second diversity antennas, the first and second diversity antennas being spatially separated by first and second orthogonal deflection antennas and at least one wavelength. And receiving the first and second space-time encoded signals from the first and second diversity antennas, respectively, and receiving the first and second diversity antennas, respectively,
The method of claim 1, wherein the step of obtaining complex channel state information obtains complex channel state information based on the received first and second space-time encoded signals. The antenna system includes a plurality of diversity antennas spatially separated from each other by at least one wavelength;
At least two signature codes embedded in each space-time encoded signal and with a substantially orthogonal signature code so that the second communication station can separate and measure the channel impulse response corresponding to each space-time encoded signal Transmitting a space-time encoded signal on each of a plurality of diversity antennas;
The channel impulse response of each space-time encoded signal of the space-time encoded signal including the selected set of least attenuated signals and the remaining set of most attenuated signals is represented as a second communication station. Measuring stage,
The method of claim 1, further comprising: transmitting from the second communication station to the first communication station a signal indicative of the selected set of least attenuated signals. The antenna system includes first and second diversity antennas, the first diversity antenna being deflected to be orthogonal to the second diversity antenna;
Embedded in the first and second space-time encoded signals, respectively, so that the second communication station can separate and measure the channel impulse response corresponding to each of the first and second space-time encoded signals Transmitting first and second space-time encoded signals with first and second diversity antennas, respectively, together with first and second signature codes that are substantially orthogonal;
Measuring a channel impulse response of each space-time encoded signal of the first and second space-time encoded signals including the weakest attenuated signal and the most attenuated signal at a second communication station;
The method of claim 1, further comprising: transmitting a signal indicating the weakest attenuation signal from the second communication station to the first communication station.   The first and second space-time encoded signals are embedded in the first and second space-time encoded signals, respectively, and are substantially orthogonal so that the second communication station can separate the complex signal into the first and second space-time encoded signals. Transmitting the first and second space-time encoded signals together with the first and second signature codes, the step of receiving at the second communication station in the first and second space- The method of claim 1, wherein the time encoded signal is received as a complex signal.   The method of claim 1, wherein the complex channel state information includes at least one weight including amplitude information and phase angle information.   The step of obtaining complex channel state information comprises: a first space transmitted from the first antenna such that the first and second space-time encoded signals are constructively augmented at the second communication station; Determining a corrected phase angle for adjusting a first phase of the time-encoded signal with respect to a second phase of the second space-time encoded signal transmitted from the second antenna. The method according to 1. Determining the corrected phase angle comprises:
Measuring a first phase angle defined by a first phase;
Measuring a second phase angle defined by the second phase;
And determining a corrected phase angle defined as a difference between the second phase angle and the first phase angle. Receiving at least two space-time encoded signals by a second communication station by respective beams of a multi-beam antenna array associated with the first communication station, the beams being in each space- Transmitting a signature code embedded in the time-coded signal, the signature code being orthogonal so that the second communication station can separate and measure the channel impulse response corresponding to each space-time coded signal , The stage,
Determining from the received space-time encoded signal a selected set of least attenuated signals and a remaining set of most attenuated signals;
Transmitting from the second communication station to the first communication station a signal indicative of the selected set of least attenuated signals. Selecting at least two beams transmitted from the first communication station based on signals received from the second communication station;
Transmitting at least two space-time encoded signals with the selected at least two beams;
Obtaining complex channel state information based on the received space-time encoded signal;
19. The method of claim 18, further comprising: transmitting complex channel state information to the first communication station. A system including a remote station, wherein the remote station is
A receiver for receiving at least two space-time encoded signals from an antenna system;
A processor that determines complex channel state information from the received space-time encoded signal and divides the complex channel state information into a plurality of channel state information segments;
A transmitter for transmitting complex channel state information to a base station, wherein the plurality of channel state information segments include one or more phase segments and one or more amplitude segments;
When the transmitter transmits complex channel state information to the base station, the transmitter transmits the plurality of channel state information segments to transmit the one or more phase segments before transmitting the one or more amplitude segments. A system that sends in order. 21. The system of claim 20, wherein when transmitting the complex channel state information to the base station , a most significant phase segment is transmitted before a least significant phase segment. A processor that divides channel state information into a plurality of channel state information segments,
A processing unit for determining the number of phase bits assigned to the phase information according to the operation mode;
A processing unit that rounds off the correction phase angle to fit the number of phase bits;
A processing unit for determining the number of amplitude bits assigned to the amplitude information according to the operation mode;
21. The system according to claim 20, wherein the system controls a processing unit that rounds off the correction amplitude according to the number of amplitude bits.   21. The system of claim 20, wherein the transmitter transmits the most significant bit of the correction phase angle prior to transmitting the most significant bit of the correction amplitude.   21. The system of claim 20, wherein the transmitter transmits the most significant bit of the correction phase angle prior to transmitting the least significant bit of the correction phase angle. The base station includes a receiver for receiving a plurality of channel state information segments;
The base station further comprises a processor for reconstructing complex channel state information from the received plurality of channel state information segments;
21. The circuit of claim 20, wherein the base station processor includes circuitry for weighting first and second feed signals sent to first and second antennas, respectively, based on reconstructed complex channel state information. system.   21. The system of claim 20, wherein the transmitter sequentially transmits a plurality of channel state information segments over a period based on channel coherency time. The system includes a multi-beam antenna array;
The receiver receives first and second space-time encoded signals from first and second beams, respectively, of a multi-beam antenna array;
21. The system of claim 20, wherein the processor obtains complex channel state information based on received first and second space-time encoded signals. The base station is
A multi-beam antenna array;
A circuit for determining an angular power spectrum of a signal transmitted from the remote station that defines first and second peaks at first and second angular positions, respectively.
First and second space-time by each of the first and second beams of the multi-beam antenna array such that the first and second beams are directed toward the first and second angular positions, respectively. 28. The system of claim 27, comprising a circuit for transmitting the encoded signal. The base station includes an antenna system that is a multi-beam antenna array;
The base station is encoded into each signal of at least two space-time encoded signals so that the remote station can separate and measure the channel impulse response corresponding to each space-time encoded signal. Circuitry for transmitting at least two space-time encoded signals with orthogonal signature codes in each beam of the multi-beam antenna array;
The remote station determines the channel impulse response of each space-time encoded signal of the space-time encoded signal including the selected set of least attenuated signals and the remaining set of most attenuated signals. Including circuitry to measure at remote stations,
21. The system of claim 20, wherein the remote station transmitter transmits a signal from the remote station to the base station indicating a selected set of weakest attenuated signals. The base station includes an antenna system, the antenna system being one of first and second orthogonally polarized antennas and first and second antennas spatially separated by at least one wavelength. Including first and second diversity antennas;
The receiver receives first and second space-time encoded signals from first and second diversity antennas, respectively;
21. The system of claim 20, wherein the processor obtains complex channel state information based on received first and second space-time encoded signals. The base station includes an antenna system, the antenna system including a plurality of diversity antennas spatially separated from each other by at least one wavelength;
With a signature code that is substantially orthogonal so that the base station is embedded in each space-time encoded signal and the remote station can separate and measure the channel impulse response corresponding to each space-time encoded signal; Further comprising a circuit for transmitting at least two space-time encoded signals on each of the plurality of diversity antennas;
The remote station determines the channel impulse response of each space-time encoded signal of the space-time encoded signal including the selected set of least attenuated signals and the remaining set of most attenuated signals. Including circuitry to measure at remote stations,
21. The system of claim 20, wherein the remote station transmitter includes circuitry for transmitting a signal indicative of a selected set of weakly attenuated signals from the remote station to the base station. The base station includes an antenna system, the antenna system includes first and second diversity antennas, and the first diversity antenna is deflected to be orthogonal to the second diversity antenna. And
The base station is embedded in each of the first and second space-time encoded signals, and the remote station separates channel impulse responses corresponding to the first and second space-time encoded signals, respectively. Further comprising circuitry for transmitting the first and second space-time encoded signals with the first and second diversity antennas, respectively, along with first and second signature codes that are substantially orthogonal so as to be measurable,
The remote station measures the channel impulse response of each space-time encoded signal of the first and second space-time encoded signals including the weakest and most attenuated signals at the remote station. Including the circuit,
21. The system of claim 20, wherein the remote station transmitter includes circuitry for transmitting a signal indicative of the weakest attenuated signal from the remote station to the base station. The base station includes an antenna system and a transmitter coupled to the antenna system, wherein the base station transmitter is embedded in first and second space-time encoded signals, respectively, and the remote station is The first and second space-time encoded signals together with the substantially orthogonal first and second signature codes so that the complex signal can be separated into the first and second space-time encoded signals, Sent by the system,
21. The system of claim 20, wherein the remote station receiver includes circuitry for receiving the first and second space-time encoded signals as a composite signal.   21. The system of claim 20, wherein the complex channel state information includes at least one weight that includes phase angle information. The system includes first and second antennas;
A first space-time signal transmitted from a first antenna such that the processor obtaining the complex channel state information constructively augments the first and second space-time encoded signals at the remote station; 21. A circuit for determining a corrected phase angle for adjusting the first phase of the encoded signal relative to the second phase of the second space-time encoded signal transmitted from the second antenna. The described system. The circuit for determining the corrected phase angle is:
A processing unit for measuring a first phase angle defined by the first phase;
A processing unit for measuring a second phase angle defined by the second phase;
36. The system according to claim 35, further comprising: a processing unit that obtains a corrected phase angle defined as a difference between the second phase angle and the first phase angle. A system including a base station, wherein the base station is
A multi-beam antenna array;
A transmitter for transmitting at least two space-time encoded signals by each beam of the multi-beam antenna array, the beam transmitting a signature code embedded in each space-time encoded signal; Each of the signature codes is substantially orthogonal so that the remote station can separate and measure the channel impulse response corresponding to each space-time encoded signal;
The remote station includes a receiver and a processor that measures the channel impulse response of each space-time encoded signal of the space-time encoded signal, the processor being a selected set of weakest attenuated sets. Determining the signal and the remaining set of most attenuated signals from the received space-time encoded signal;
The remote station further includes a transmitter that transmits a signal from the remote station to the base station indicative of a selected set of least attenuated signals. The base station includes a processor that selects at least two beams transmitted from the base station based on signals received from the remote station;
Said base station further comprising circuitry for transmitting at least two space-time encoded signals in the selected at least two beams;
The remote station processor includes circuitry for obtaining complex channel state information based on a received space-time encoded signal;
38. The system of claim 37, wherein the remote station transmitter includes circuitry for transmitting complex channel state information to the base station.

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