JP4808221B2 - High frequency modulation / demodulation multiphase rectifier - Google Patents
High frequency modulation / demodulation multiphase rectifier Download PDFInfo
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- JP4808221B2 JP4808221B2 JP2007550161A JP2007550161A JP4808221B2 JP 4808221 B2 JP4808221 B2 JP 4808221B2 JP 2007550161 A JP2007550161 A JP 2007550161A JP 2007550161 A JP2007550161 A JP 2007550161A JP 4808221 B2 JP4808221 B2 JP 4808221B2
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
- H02M1/4208—Arrangements for improving power factor of AC input
- H02M1/4216—Arrangements for improving power factor of AC input operating from a three-phase input voltage
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
- H02M7/02—Conversion of ac power input into dc power output without possibility of reversal
- H02M7/04—Conversion of ac power input into dc power output without possibility of reversal by static converters
- H02M7/12—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
- H02M7/21—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
- H02M7/217—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
- H02M7/23—Conversion of ac power input into dc power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only arranged for operation in parallel
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
- H02M1/00—Details of apparatus for conversion
- H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
- H02M1/4208—Arrangements for improving power factor of AC input
- H02M1/4283—Arrangements for improving power factor of AC input by adding a controlled rectifier in parallel to a first rectifier feeding a smoothing capacitor
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- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
- Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
- Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
- Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
Description
【技術分野】
【０００１】
この出願の発明は、交流電源側に流出する高調波電流と高周波雑音を低減しながら小型軽量化、高効率化を実現する高周波変復調多相整流装置に関するものである。
【背景技術】
【０００２】
一般に、数ＫＷ以上の交流電力を直流電力に変換する整流装置や、これにＤＣ−ＡＣインバータ、ＤＣ−ＤＣコンバータ等を付加して回転機器や各種産業機器の動力源として用いる代表的な変換装置に三相全波整流回路がある。
近年、これらパワーエレクトロニクス機器から発生する高調波電流が配電線を通して他の電気機器に障害を与えるという問題が地球規模で認識され、各政府機関主導の元にＩＥＣ（国際電気標準会議）の規制値をガイドラインとして規定している。
図１は、代表的な三相全波整流回路の抵抗負荷時における交流入力電流波形と、含まれる高次高調波成分および全高調波歪（ＴＨＤ：Total Harmonic Distortion）の例を示したものであるが、容量性負荷が加わると、この値は更に大きくなる。
このような電源の公害を除くために、この十数年の間に受動（Passive）形、能動（Active）形およびその複合形など２０種類を越す各種の回路構成法が公表されているが、コスト、過負荷耐量、小型軽量化、効率、高調波歪量、高周波雑音（輻射電磁波）、装置寿命（ＭＴＢＦ）等の観点からそれぞれ特長や問題点がある。
図２に、従来の高力率整流回路の代表例を示す。図２−（１）から図２−（３）までは、いわゆる受動形の整流回路で、低周波交流電圧を変圧器により多相に変換、整流するものである。これらの受動形の整流回路は、原理的にスイッチング雑音は発生せず、最近話題の多い電力線通信に対しての整合性が良く、過負荷耐量、装置寿命等の信頼性が高い反面、変圧器の容積、重量等から小型、軽量化は困難である。
図２−（１）の１２相整流回路は簡易な高調波対策として用いられてきたが、純抵抗負荷において１４％前後の全高調波歪（以下、ＴＨＤとも記す）があり、前述のガイドラインに適合させるにはフィルタの追加が必要である。また、直流出力電力の大よそ１／３に相当する電力容量の位相差変圧器を必要とする。
最近、図２−（２）に示した１８相整流回路が提案された（特開２００３−８８１２４号公報）。この１８相整流回路では、９％弱のＴＨＤで、しかも９相全波（１８パルス）用単巻変圧器の電力容量が１２相用に比べ２０％減で済む。これに１２相用よりも小規模のフィルタを付加することによりＴＨＤ３％以下が可能になる。しかし、この場合でも低周波変圧器を使用するため、１０ＫＷ直流出力時で２５〜４０Ｋｇの重量、１０〜１５リットルの容量を要する。
図２−（３）は、図２−（１）の整流回路の１２相パルス出力電圧のリップルと、入力高調波電流を低減するために、出力容量の５％程度の補助インバータを付加して主高調波（主として１１次、１３次）を打消す方法を用いた整流回路である。この整流回路では、ＴＨＤが５％程度まで改善されるが、主変圧器が絶縁型のため容積、重量が大となる。また、高電圧用に限られるため低圧用には別の回路構成が必要になるなどの課題がある。
図２−（４）は、良く知られた６パルス昇圧形の力率改善回路（ＰＦＣ）である。この力率改善回路では主回路構成が単純化されているが、駆動回路が複雑であり、直流出力電流値に対して数倍の電流定格値を持つ半導体スイッチが必要であり、更に電流連続モードの駆動を行ってもなお、給電系に対して厳重なノイズフィルタが必要で、大電力になるほど十分な減衰量を得るフィルタの設計が困難になる。現在、市場で使用されている力率改善回路のＴＨＤは５％以内の規格を満足しているものが多いが、スイッチング方式を用いる力率改善回路は高周波雑音の発生が大きい。
図２−（５）は、三相全波整流回路を主とする既存の工場設備の受電端で高調波対策を行う場合に使われることが多い回路である。この回路は、図２−（４）の整流回路に比べて主直流電流分の負担がないので能動フィルタの電力容量が少なくて済み、一見合理的で総合経済性の点からも好ましく見える。しかし実際の整流器出力側には容量性負荷、トルク変動の大きい電動機等の存在によって、抵抗負荷時の３０％前後の高調波を大幅に越す高調波を発生するものもあり、既存設備の負荷状況に応じた電力容量の能動フィルタが必要になる。この場合の高周波雑音成分は図２−（４）の整流回路より相対的に大きいため、装置内におけるノイズフィルタの比率が大きくなる。
図２−（６）は、前途の高周波雑音を極力低減する能動フィルタ方式の代表例で、直流電流は三相全波整流回路により供給し、高調波の５次または７次を受動フィルタで打消し、残りの高調波を直流出力電力の６％程度の電力容量を持つ能動フィルタによって除去するので、能動形力率改善回路の中では最も高周波雑音が少ないがＴＨＤは４％どまりとなる。
以上に述べたＡＣ−ＤＣコンバータの他にも電流不連続モードの簡易型やＤＣ−ＤＣコンバータとのスイッチ共用による経済化型など多種、多様な回路構成が発表されているが、それぞれ一長一短があり、製造業者または使用者側で使用目的、環境に応じて回路方式を選択しているのが実情である。従って、三相入力の高力率整流回路に課せられる課題のうち、小型軽量化、低雑音、高効率化および低ＴＨＤを同時に解決する整流回路の実現が望まれていた。
発明の開示
［０００３］
この出願の発明は、以上のような従来技術の実情に鑑みてなされたもので、小型軽量化、低雑音、高効率化および低ＴＨＤを同時に達成する高周波変復調多相整流装置を提供することを課題とする。
この出願の発明は、上記課題を解決するものとして、第１には、三相交流電源から入力した三相交流を直流に変換する直結主三相全波整流器と、これに並列して設けられる三組のリング変調波電力発生器と三相複巻線高周波多相変換変圧器と相数に対応して設けられる複数のリング変調波復調器兼補助三相全波整流器を有する高調波補正回路を備え、前記三相複巻線高周波多相変換変圧器の各相一次側巻線はリング変調波電力発生器の出力とそれぞれ接続され、前記三相複巻線高周波多相変換変圧器の二次側から、主巻線と複数の補助巻線との組み合わせ接続により、リング変調された多相交流出力電圧の内の一組を多相出力端子から取出し、前記複数のリング変調波復調器兼補助三相全波整流器に入力させ、前記高調波補正回路の直流出力を前記直結主三相全波整流器の直流出力と並列接続し、三相交流電源側から見て等価的に６ｎ相（ｎは３〜７の整数）の多相全波整流回路を構成することを特徴とする高周波変復調多相整流装置を提供する。
また、第２には、三相流電源に接続される三組のリング変調波電力発生器と三相複巻線高周波多相変換変圧器と相数に対応して設けられる複数のリング変調波復調器兼三相全波整流器を有する高調波補正回路を備え、前記三相複巻線高周波多相変換変圧器の各相一次側巻線はリング変調波電力発生器の出力とそれぞれ接続され、前記三相複巻線高周波多相変換変圧器の二次側から、主巻線と複数の補助巻線との組み合わせ接続により、リング変調された多相交流出力電圧の内の一組を多相出力端子から取出し、前記複数のリング変調波復調器兼三相全波整流器に入力させ、前記高調波補正回路の直流出力を直接負荷に供給し、三相交流電源と負荷側が前記高周波多相変換変圧器により絶縁されることを特徴とする高周波変復調多相整流装置を提供する。
また、第３には、上記第１または第２の発明において、前記リング変調波電力発生器の時比率制御により直流出力電圧を連続調整可能としたことを特徴とする高周波変復調多相整流装置を提供する。
さらに、第４には、上記第１または第２の発明において、多相整流回路の主整流回路をアクティブ素子で構成し、一方、主整流回路と全く同一の回路配置をホトモス・スイッチと補助電源で構成し、ホトモス・スイッチの出力により任意の多相全波整流回路の同期整流を部分的または全面的に実現することを特徴とする高周波変復調多相整流装置を提供する。
この出願の発明に係る高周波変復調多相整流装置によれば、交流電源側に流出する高調波電流を１〜２％程度に抑えながら、小型、軽量高効率をも実現できる。電力機器は元来各種システム機器の一部を分担しているにすぎないが、今後システムで多く用いられるデジタル素子は現在の５Ｖ、３．３Ｖ系から１．８Ｖ系を経て将来は１Ｖに近い論理レベルでの高速動作を目標に開発が進められている。将来のパワーエレクトロニクス機器はこれら低圧大電流負荷における小型化のために高効率化が進行しているが、これと並行して高調波雑音の極力少ない電力機器が安定なシステム構成のために必要であり、この出願の発明はこの課題に十分応えることができる。
【図面の簡単な説明】
【０００４】
【図１】図１は、従来の三相全波整流回路における電流波形と高調波成分を示す図である。
【図２】図２は、従来の高力率整流回路の代表例を示す図である。
【図３】図３は、一般的な多相整流回路における高調波成分とフィルタ効果及び変圧器比容量を示す図である。
【図４】図４は、この出願の発明に係る第１実施形態の高周波変復調多相整流装置の構成を模式的に示す図である。
【図５】図５は、図４の高周波変復調多相整流装置の主要構成要素であるリング変調波電力発生器の回路例を示す図である。
【図６】図６は、１８パルス整流時の交流入力電流波形とリング変調波形を示す図である。
【図７】図７は、時比率を高めたこの出願の発明による方式と通常のＰＷＭ制御波形との脈動成分の相違を比較した図である。
【図８】図８は、合成された１８パルス三相入力電流と主整流回路及び補助整流回路が分担する各位相と振幅の相対関係を示す図である。
【図９】図９は、図８の交流入力電流中最大値を示すＩ５のベクトル図による求め方の説明図である。
【図１０】図１０は、図８で示された交流入力電流と三相各相電流の相対関係を示す図である。
【図１１】図１１は、図４の高周波変復調多相整流装置を交流入力、直流出力間で絶縁した場合の各部電流分布を示す図である。
【図１２】図１２は、３０相整流回路を構成するこの出願の発明に係る第２実施形態高周波変復調多相整流装置を示す図である。
【図１３】図１３は、３０パルス整流波形における入力フィルタの効果を示す図である。
【図１４】図１４は、低電圧、大電流向絶縁型のこの出願の発明に係る第３実施形態の高周波変復調多相整流装置を示す図である。
【図１５】図１５は、主整流ダイオードの代わりに同期整流を行う場合の駆動回路を示す図である。
【発明を実施するための最良の形態】
【０００５】
以下、この出願の発明を実施形態により詳細に説明する。
説明の便宜上、先ず低ＴＨＤ、低雑音化のために低周波で多相整流を行った場合の全高調波歪率（ＴＨＤ）とその主成分、並直列フィルタ容量、フィルタ付加時のＴＨＤ、多相変換変圧器容量、入出力電力比の関連を図３に要約した。図３の表中の１項におけるＴＨＤの数値は実測値で、フーリエ級数から算出される理論値よりも１０％前後低い値を示すが、これは配電系の電源インピーダンスと装置容量との兼ね合いで変わるので、パルス数との相対的な改善度に着目して評価する。
現在の交流機器の多くは、測定器などの特殊機器を除けば電圧歪率、電流歪率共にＴＨＤ５％以下、各高調波単独成分含有率３％以下が一般的な規格として流通している。従って、入力フィルタ無しで上記規格を満足するには図３の表の実測値から３０パルス以上の多相化が必要であり、１８パルス、２４パルスでは、図４に例示したＬＣフィルタ（２）を付加すれば規格を満足する。
一方、６パルス、１２パルスでは図２−（４）、（５）に示した能動形でないと規格が満足されない事が多く、図２−（６）は能動形フィルタ用スイッチの容量を削減するために受動形フィルタを併用した例である。
いずれにしても内、外の学術論文等に発表されている内容から見る限り、過渡現象を含む広範囲の負荷電流、負荷力率等の変動に対してＴＨＤ４％程度までが能動形の実力値と推定される。その上、能動形では原理上パルス幅変動の大きいＰＷＭ制御を用いるので、大電力の機器では３０ＭＨｚ帯域までの雑音スペクトラム電力を抑えきれず、通信機器や医療機器等に影響を与える事もある。
これに対して図２−（１）、（２）に示した受動形は原理的に高周波スイッチングを行っておらず、また図２−（３）では部分的にスイッチングを行ってはいるが、１１次、１３次の高調波を打消す目的のため直流出力電力の５％程度の電力で固定パルス位相に近い制御が行われる。従って、能動形のような大きな雑音電力の発生はない。その反面、図２に示した多相変換用変圧器が必要になり、小型軽量化の達成が困難になる。
この出願の発明に係る高周波変復調多相整流装置は、入力側三相交流の多相化を図２−（１）、（２）および（３）のような低周波変圧器の代わりに高周波変圧器を用いて実現した処に大きい特徴がある。この目的のため、通信技術で用いられてきたリング変調技術を電力分野に導入した。すなわち、リング変調波電力は三相または単相三個の複巻線高周波変圧器の一次巻線に加えられ、二次側の主巻線と複数の補助（三次）巻線との接続構成により、変調波形に含まれる低周波成分の多相変換を行い、更に復調回路を構成するダイオード自体が多相全波整流器を兼ねている。また、電子化に伴って生じるスイッチング雑音を極小とするため、オフ期間をほとんど生じないスイッチ駆動を行っている。すなわち、従来のＰＷＭ制御スイッチング方式に比べ片側５０％に近い時比率で各スイッチを駆動する事により、交流入力側電流の連続性がほぼ保たれ、高周波雑音成分を数分の一に低減する。
この出願の発明は、能動形ＰＦＣと同等の小型、軽量化を保ちながら、能動形では実現困難な１〜２％のＴＨＤと高周波帯域雑音電力を低減できる高周波変復調多相整流装置に関するもので、その具体例を以下に述べる。
図４に、この出願の発明による第１実施形態の高周波変復調多相整流装置の構成を示す。この実施形態の整流装置は、１８相（パルス）整流装置の一例であるが、これは多相変換の中では回路規模が小さく、説明が容易なために便宜上選んだものであり、２４、３０、３６及び４２相整流装置となっても、基本原理に変わりはない。
図４において、（１）は直列リアクトルで、ＬＣ形同次フィルタ（２）と共に高調波を抑える役割をする。（３）は直結三相全波整流器で全直流出力電流の１／３を分担する。（４ａ）、（４ｂ）及び（４ｃ）はリング変調波復調器兼補助三相全波整流器で、（４ａ）は全直流出力電流の１／３を分担し、（４ｂ）、（４ｃ）は、それぞれ全直流出力電流の１／６ずつを分担する。（５）は通常の直流リップル・フィルタ、（６）は入力三相交流の位相を保持するリング変調波による高周波多相変換変圧器、（７ａ）、（７ｂ）及び（７ｃ）は図５−（ａ）、（ｃ）及び（ｄ）に詳しく例示したリング変調波電力発生器で、各巻線に生じる電圧波形は巻数比に比例した図５−（ｂ）の形となる。なお図５において（Ｓ１）〜（Ｓ８）はスイッチを示す。（８ａ）、（９ａ）及び（１０ａ）はそれぞれ三相入力Ｒ−Ｓ相の位相変調電圧を変圧する一次、二次及び三次巻線であり、（８ｂ）、（９ｂ）及び（１０ｂ）はそれぞれ三相入力Ｓ−Ｔ相の位相変調電圧を変圧する一次、二次及び三次巻線であり、（８ｃ）、（９ｃ）及び（１０ｃ）はそれぞれ三相入力Ｔ−Ｒ相の位相変調電圧を変圧する一次、二次及び三次巻線である。（１１）は前記三組のリング変調波電力発生器（７ａ）、（７ｂ）及び（７ｃ）のゲート駆動回路、（１２ａ）は直流出力＋端子、（１２ｂ）は直流出力−端子、（１３）はスナバコンデンサで、リング変調波復調器兼補助三相全波整流器（４ａ）、（４ｂ）及び（４ｃ）のリング変調波復調ダイオードとの組み合わせにより無損失スナバ回路を構成する。
上述したように、入力交流電力はＲ，Ｓ，Ｔ各相間電圧がリング変調波電力発生器（７ａ）、（７ｂ）及び（７ｃ）によって１０ＫＨｚ以上の固定周波数で変調され、高周波多相変換変圧器（６）の一次巻線（８ａ）、（８ｂ）及び（８ｃ）に加えられる。高周波多相変換変圧器（６）の二次巻線（９ａ）、（９ｂ）及び（９ｃ）と三次巻線（１０ａ）、（１０ｂ）及び（１０ｃ）とを、例えば図４中に示されたように接続する事により、(1)端子（以下、便宜上図面中の丸囲み数字は括弧付数字で表す）から(9)端子まで９相の低周波交流位相で包絡線（エンベロープ）変調されたリング変調波形が発生する。
図６−（ｂ）、（ｃ）及び（ｄ）は試作装置の波形例を示したもので、無損失スナバの効果により、高周波パルスにスパイク・ノイズの発生がなく、入力交流電流波形図６−（ａ）も低周波変圧器による多相交流変換時の電流波形と大差ない。
図５−（ａ）、（ｃ）中に示したスイッチ（Ｓ１）、（Ｓ２）は通常用いられているＭＯＳ−ＦＥＴを背面接続して双方向スイッチとして用いているが、ＩＧＢＴの背面接続や最近開発が進展している双方向ＩＧＢＴであっても良い。また、ゲート駆動回路（１１）の駆動パルス幅は、時比率５０％−（デッドタイム）の固定とし、パルス幅変調を行わない場合、全波整流出力波形は図７−（ａ）に示したように、高周波雑音成分の少ない電圧、電流波形となり、通常のパルス幅変調を行う図７−（ｂ）に比べ、雑音成分が数分の一に低下することがわかる。ゲート駆動回路（１１）は図５−（ａ）のスイッチ（Ｓ１）、（Ｓ２）、（Ｓ７）及び（Ｓ８）を同時に、また逆位相時はスイッチ（Ｓ３）、（Ｓ４）、（Ｓ５）及び（Ｓ６）を同時に駆動する。この動作は、図４中のリング変調波電力発生器（７ａ）、（７ｂ）及び（７ｃ）の全ブロックについて同位相で駆動し、パルス位相を揃える事により正常動作が確保される。
図４の高周波変復調多相整流装置の整流動作は若干複雑であるため、図８及び図９を用いて説明する。図８−（ｃ）に高周波多相変換変圧器（６）の二次側巻線端子間電圧と変調波位相の関係を低周波成分のみで表現した。この１８パルス位相の内から代表して(1)、(2)、(3)及び(4)のパルス位相における変圧器二次巻線の電圧ベクトルを図８−（ｂ）の太い矢印で示した。この内(3)の位相時の各相電流はＲ相では（Ｉ１＋Ｉ３）、Ｓ相では（Ｉ０−Ｉ４−Ｉ３）、Ｔ相では（Ｉ２＋Ｉ４）の電流が流れ、変圧器各二次巻線には矢印の点線で示したＩ１、Ｉ２、Ｉ３及びＩ４が流れる。ここでＩ０は図４の補助整流器入力電流であり、その振幅値は１８パルス整流直流出力に等しく、時比率１／１８の正、負各１パルスの交流成分であり、Ｉ１、Ｉ２、Ｉ３及びＩ４は変圧器の各二次巻線を位相(3)の期間に流れる電流値であり、その値は図８−（ａ）の表中に示されたように、二次巻線のコイル間電圧から容易に算出される。図８−（ｂ）で矢印(2)の電圧ベクトル時はＩ１とＩ２、Ｉ３とＩ４が入替わるのみであり、この場合、Ｒ１相とＴ１相の電流も入替わる。
これらの様子を時間軸で示したのが図８−（ａ）の入力交流電流波形であり、図中の(1)、(2)、(3)及び(4)が図８−（ｂ）及び図８−（ｃ）の位相番号に対応する。図１０は三相入力各相電流の関係を明確化するために前記位相関係を追加したものであり、加えて(1)、(4)の位相時にはリング変調波電力発生器（７ａ）、（７ｂ）及び（７ｃ）、高周波多相変換変圧器（６）及びリング変調波復調器兼補助三相全波整流器（４ａ）、（４ｂ）及び（４ｃ）で構成される高周波多相整流器よりインピーダンスの低い直結三相全波整流器（３）が負荷電流を分担する。従って(1)、(4)の区間は前記高周波多相整流器の負担が減り、整流装置全体の効率が改善される。反面、直結三相全波整流器（３）の存在により、整流装置全体で見た入出力間の絶縁は確保できなくなる。
図１１は図４から直結三相全波整流器（３）を取り除いて、１８相整流回路を構成した場合の各部電流分布を示す。直結三相全波整流器（３）で負担した正負各２パルス分（図１０の正、負パルスＩ０）を整流器（４ａ）が兼ね、図１１の素子導通区間のように分担する事により、入出力絶縁型高周波変復調多相整流装置が実現する。なお本例では（４ａ）の三相全波ブリッジが全体の出力の２／３を負担し、（４ｂ）、（４ｃ）が各１／６の出力を分担しているが、これは高周波変圧器の二次側巻線容量を低減するためであり、リング変調波復調器兼補助三相全波整流器（４ａ）、（４ｂ）及び（４ｃ）の各ブリッジを均等に１／３ずつ負荷分担するように、高周波変圧器を設計しても一向に差し支えない。ただし、この場合は同一直流出力に対して高周波変圧器の二次側巻線容量が約２０％増加する。
次に、この出願の発明に係る第２実施形態の高周波変復調多相整流装置について説明する。この実施形態の整流装置は、非絶縁型３０相整流装置の例であり、図１２にその回路接続例を示した。図１２において図４と同様な要素には同じ符号を付してある。この実施形態の整流装置と図４の１８相整流装置との相違点は、変圧器二次巻線の多相化（９相から１５相）とそれに伴い補助三相整流器（４ｄ）及び（４ｅ）が追加され、各部電流値配分が変わった点にある。また、変圧器二次巻線の接続方法は図中の点線で示したような接続（この場合二次巻線容量が数％増える）をはじめ、２〜３の変形があるが、その取捨選択は通常公知の技術である。
図１３に図１２に示した回路の電流波形を示す。図で（Ａ）は直列リアクトル（１）、同次フィルタ（２）（５０Ｈｚ×３０＝１．５ＫＨｚに共振）が共にない場合で約４％のＴＨＤを示し、（Ｂ）は３％電圧降下する直列リアクトル（１）のみ挿入した場合で２．６％のＴＨＤを示し、（Ｃ）は同じ直列リアクトル（１）と同次フィルタ（２）を加えた場合に１％のＴＨＤが得られる事を確認している。この値は、従来の如何なる高力率整流回路も実現困難であったので、この出願の発明の特徴を示している。またこのときの三相直列リアクトル（１）の容量は直流出力電力の３．７％、同次フィルタ（２）のリアクトル容量は単相当たり直流出力電力の１．２％、共振用コンデンサはデルタ接続時０．５μＦ／相／直流出力１ＫＷ当たりであり、整流装置全体の容積、重量及びコストに占める比重は少ない。
なお、図１２の回路で直結三相全波整流器（３）を取り除くと、図１１の１８相整流装置と同様に、３０相整流装置全体は交流入力と直流出力とが高周波多相変換変圧器（６）によって絶縁される。この場合、直結三相全波整流器（３）の入、出力電流（２／１５）1/2Ｉ、（３／１５）Ｉが三相全波整流器（４ａ）に加わり、三相全波整流器（４ａ）の入、出力電流分担分は（６／１５）1/2Ｉ、（６／１５）Ｉとなる。また、高周波多相変換変圧器（６）とリング変調波電力発生器（７ａ）、（７ｂ）及び（７ｃ）の容量が（５／４）1/2倍増加する以外は変わりないので、具体例は省略する。なお、入出力回路が絶縁されると、当然高周波変圧器の一次、二次及び三次巻線比によって電圧比が自由に選べる。
図１４は１２Ｖ１０００Ａレベルの電源高周波低減形整流装置の１８パルス整流例を示した。リング変調波電力発生器（７ａ）、（７ｂ）及び（７ｃ）に図５−（ｄ）の回路を用い、スイッチ（Ｓ９）と（Ｓ１０）を５０％−（デッドタイム）の時比率で駆動し、スイッチ（Ｓ１１）と（Ｓ１２）も同じ時比率ではあるが公知の位相差ＰＷＭ制御によって常にリング変調波電力発生器（７ａ）、（７ｂ）及び（７ｃ）の電源インピーダンスを低く抑える事により、安定な動作を行う事が出来る。この場合、位相差制御によって直流出力は１０〜１００％可変にできるが、この場合には交流電源側の雑音用フィルタの追加が必要になる。なお、整流ダイオードにショットキ・バリア形式を用いた場合でも全体の効率を９０％以上に保つ事が可能になる。
２４パルス、３６パルス及び４２パルス等についても原理的には高周波多相変換変圧器（６）の二次および三次巻線の組み合わせと、出力側復調器兼整流ダイオードの追加により、容易に実現可能である事が明らかであるので、これら例についての説明は省略した。
図１５は更に高効率化を図るため、主整流ダイオードの代わりに同期整流を行う場合の駆動回路を示した。この例では最も単純な３相全波について示したが、低周波から高周波までの入力電源、更に６、１２、２４、３０、３６および４２相の多相交流電源で、半波、全波の整流回路すべてにおいて、主整流素子の構成と全く同一形態の補助整流回路をホト・モス・スイッチ（Photo MOS Switch）で構成し、配置の対応が等しい主スイッチをホト・モス・スイッチで駆動する事により、単純な回路構成で同期整流を実現できる。
図中のＱ1〜Ｑ6は主スイッチ、ＰＳ1〜ＰＳ6はホト・モス・スイッチ、Ｅ1〜Ｅ6は駆動用電源を示す。なおダミー抵抗ＲLによってホト・ダイオードに流れる電流を調整する。
以上の構成を採る事により、通常、ショットキーバリヤーダイオードの０．６Ｖ程度の電圧降下を０．１Ｖ以下に低減でき、低圧、大電流用整流回路の効率を改善できる。なお、総合経済性の見地から、全部の整流素子でなく電流分担の大きい素子のみを同期整流としてもよい。
また、この出願の発明の装置構成では大容量電解コンデンサを用いないので長寿命が期待でき、更に電源投入時の突入電流は極めて少ない。【Technical field】
[0001]
The invention of this application relates to a high-frequency modulation / demodulation multiphase rectifier that realizes a reduction in size and weight and an increase in efficiency while reducing harmonic current and high-frequency noise flowing out to the AC power supply side.
[Background]
[0002]
Generally, a rectifier that converts AC power of several kilowatts or more into DC power, and a typical converter that is used as a power source for rotating equipment and various industrial equipment by adding a DC-AC inverter, DC-DC converter, etc. Has a three-phase full-wave rectifier circuit.
In recent years, the problem that harmonic currents generated from these power electronics devices can interfere with other electrical devices through distribution lines has been recognized on a global scale, and IEC (International Electrotechnical Commission) regulatory values are led by government agencies. As a guideline.
Fig. 1 shows an example of the AC input current waveform of a typical three-phase full-wave rectifier circuit under resistive load, and the included higher-order harmonic components and total harmonic distortion (THD: Total Harmonic Distortion). Although there is a capacitive load, this value becomes even larger.
In order to eliminate such power pollution, over 20 types of circuit configuration methods have been announced over the past 10 years, including passive, active and composite types. Each has its own features and problems from the viewpoints of cost, overload capability, reduction in size and weight, efficiency, harmonic distortion, high frequency noise (radiated electromagnetic wave), device life (MTBF), and the like.
FIG. 2 shows a typical example of a conventional high power factor rectifier circuit. FIG. 2- (1) to FIG. 2- (3) are so-called passive rectifier circuits that convert a low-frequency AC voltage into multiple phases and rectify it using a transformer. These passive rectifier circuits do not generate switching noise in principle, have good compatibility with power line communication, which has recently been a hot topic, and have high reliability such as overload capability and device life, but transformers Because of its volume, weight, etc., it is difficult to reduce the size and weight.
Although the 12-phase rectifier circuit of Fig.2- (1) has been used as a simple countermeasure against harmonics, there is a total harmonic distortion of about 14% (hereinafter also referred to as THD) in a pure resistance load. It is necessary to add a filter to adapt. In addition, a phase difference transformer having a power capacity corresponding to approximately 1/3 of the DC output power is required.
Recently, an 18-phase rectifier circuit shown in FIG. 2- (2) has been proposed (Japanese Patent Laid-Open No. 2003-88124). In this 18-phase rectifier circuit, the THD is less than 9%, and the power capacity of the single-phase transformer for 9-phase full wave (18 pulses) can be reduced by 20% compared to that for 12-phase. By adding a filter smaller than that for 12-phase to this, THD of 3% or less becomes possible. However, since a low-frequency transformer is used even in this case, a weight of 25 to 40 kg and a capacity of 10 to 15 liters are required at the time of 10 kW DC output.
Fig. 2- (3) shows a 12-phase pulse output voltage ripple of the rectifier circuit of Fig. 2- (1) and an auxiliary inverter of about 5% of the output capacity to reduce the input harmonic current. This is a rectifier circuit using a method of canceling main harmonics (mainly 11th and 13th). In this rectifier circuit, the THD is improved to about 5%, but the main transformer is an insulating type, so that the volume and weight are increased. In addition, since it is limited to high voltage, there is a problem that another circuit configuration is required for low voltage.
FIG. 2- (4) shows a well-known 6-pulse boost type power factor correction circuit (PFC). In this power factor correction circuit, the main circuit configuration is simplified, but the drive circuit is complicated, a semiconductor switch having a current rating value several times the DC output current value is required, and the current continuous mode However, a severe noise filter is required for the power supply system, and it becomes difficult to design a filter that obtains a sufficient amount of attenuation as the power increases. Currently, many THDs of power factor correction circuits used in the market satisfy a standard of 5% or less, but power factor improvement circuits using a switching method generate a large amount of high frequency noise.
Fig. 2- (5) is a circuit that is often used when taking countermeasures against harmonics at the receiving end of existing factory equipment mainly consisting of a three-phase full-wave rectifier circuit. Since this circuit is not burdened with the main DC current compared to the rectifier circuit of FIG. 2- (4), the power capacity of the active filter can be reduced, and it looks reasonable and preferable from the viewpoint of overall economy. However, on the actual rectifier output side, there are some that generate harmonics that greatly exceed the harmonics of around 30% at the time of resistance load due to the presence of capacitive loads, motors with large torque fluctuations, etc. Therefore, an active filter having a power capacity corresponding to the above is required. Since the high-frequency noise component in this case is relatively larger than that of the rectifier circuit of FIG. 2- (4), the ratio of the noise filter in the apparatus is increased.
Fig. 2- (6) is a typical example of an active filter system that reduces the high-frequency noise as much as possible. DC current is supplied by a three-phase full-wave rectifier circuit, and the fifth or seventh harmonic is canceled by a passive filter. The remaining harmonics are removed by an active filter having a power capacity of about 6% of the DC output power. Therefore, the active power factor correction circuit has the least high frequency noise, but the THD is only 4%.
In addition to the AC-DC converters described above, a wide variety of circuit configurations have been announced, such as a simple current discontinuous mode type and an economical type by sharing a switch with the DC-DC converter, but each has advantages and disadvantages. The actual situation is that the manufacturer or the user selects the circuit system according to the purpose of use and the environment. Therefore, among the problems imposed on the three-phase input high power factor rectifier circuit, it has been desired to realize a rectifier circuit that simultaneously solves the reduction in size and weight, low noise, high efficiency, and low THD.
Disclosure of the invention
[0003]
The invention of this application was made in view of the actual situation of the prior art as described above, and provides a high-frequency modulation / demodulation multiphase rectifier that simultaneously achieves a reduction in size and weight, low noise, high efficiency, and low THD. Let it be an issue.
In order to solve the above problems, the invention of this application is firstly provided in parallel with a direct-coupled main three-phase full-wave rectifier that converts three-phase alternating current input from a three-phase alternating current power source into direct current. Harmonic correction circuit having three sets of ring modulation wave power generator, three-phase double-winding high-frequency multi-phase conversion transformer, and a plurality of ring modulation wave demodulator and auxiliary three-phase full-wave rectifiers corresponding to the number of phases Each phase primary side winding of the three-phase multi-winding high-frequency multi-phase conversion transformer is connected to an output of a ring-modulated wave power generator, respectively, From the secondary side, a combination of a main winding and a plurality of auxiliary windings is used to take out one set of ring-modulated polyphase AC output voltages from the polyphase output terminal, Input to the auxiliary three-phase full-wave rectifier, DC output of the harmonic correction circuit It is connected in parallel with the DC output of the directly connected main three-phase full-wave rectifier, and equivalently constitutes a 6n-phase (n is an integer of 3 to 7) multi-phase full-wave rectifier circuit when viewed from the three-phase AC power supply side. A high-frequency modulation / demodulation multiphase rectifier characterized by the above is provided.
Second, three sets of ring modulation wave power generators connected to a three-phase power source, a three-phase multi-winding high-frequency multi-phase conversion transformer, and a plurality of ring modulation waves provided corresponding to the number of phases. A harmonic correction circuit having a demodulator and a three-phase full-wave rectifier, each phase primary winding of the three-phase double-winding high-frequency multi-phase conversion transformer is connected to the output of the ring-modulated wave power generator, From the secondary side of the three-phase multi-winding high-frequency multi-phase conversion transformer, a combination of a main winding and a plurality of auxiliary windings is used to connect one set of ring-modulated multi-phase AC output voltages to a multi-phase. Take out from the output terminal, input to the plurality of ring modulation wave demodulator and three-phase full-wave rectifier, directly supply the DC output of the harmonic correction circuit to the load, the three-phase AC power supply and the load side is the high-frequency multi-phase conversion A high-frequency modulation / demodulation multiphase rectifier characterized by being insulated by a transformer Subjected to.
According to a third aspect of the present invention, there is provided the high-frequency modulation / demodulation multiphase rectifier according to the first or second invention, wherein the DC output voltage can be continuously adjusted by the time ratio control of the ring modulation wave power generator. provide.
Further, fourthly, in the first or second invention, the main rectifier circuit of the multiphase rectifier circuit is constituted by an active element, while the same circuit arrangement as that of the main rectifier circuit is formed by a photomoss switch and an auxiliary power source. The high-frequency modulation / demodulation multiphase rectifier is realized by partially or fully realizing synchronous rectification of an arbitrary multiphase full-wave rectifier circuit by an output of a photomoss switch.
According to the high frequency modulation / demodulation multiphase rectifier according to the invention of this application, it is possible to realize a small size, light weight and high efficiency while suppressing the harmonic current flowing out to the AC power source side to about 1 to 2%. Electric power devices originally share only a part of various system devices, but digital elements that will be used in many systems in the future will be close to 1V in the future through the current 5V, 3.3V system to 1.8V system. Development is progressing with the goal of high-speed operation at the logic level. In the future, power electronics devices are becoming more efficient due to the miniaturization of these low-voltage, high-current loads, but in parallel, power devices with as little harmonic noise as possible are required for a stable system configuration. Yes, the invention of this application can sufficiently meet this problem.
[Brief description of the drawings]
[0004]
FIG. 1 is a diagram showing current waveforms and harmonic components in a conventional three-phase full-wave rectifier circuit.
FIG. 2 is a diagram showing a typical example of a conventional high power factor rectifier circuit.
FIG. 3 is a diagram illustrating a harmonic component, a filter effect, and a transformer specific capacity in a general polyphase rectifier circuit.
FIG. 4 is a diagram schematically showing the configuration of the high-frequency modulation / demodulation multiphase rectifier according to the first embodiment of the present invention.
5 is a diagram illustrating a circuit example of a ring modulated wave power generator that is a main component of the high-frequency modulation / demodulation multiphase rectifier of FIG. 4;
FIG. 6 is a diagram showing an AC input current waveform and a ring modulation waveform during 18-pulse rectification.
FIG. 7 is a diagram comparing the difference in pulsation component between a method according to the invention of this application with an increased time ratio and a normal PWM control waveform.
FIG. 8 is a diagram showing a relative relationship between the synthesized 18-pulse three-phase input current and each phase and amplitude shared by the main rectifier circuit and the auxiliary rectifier circuit.
FIG. 9 is an explanatory diagram of how to obtain the maximum value in the AC input current of FIG. 8 using a vector diagram of I5.
FIG. 10 is a diagram showing a relative relationship between the AC input current and the three-phase currents shown in FIG.
11 is a diagram showing a current distribution of each part when the high-frequency modulation / demodulation multiphase rectifier of FIG. 4 is insulated between an alternating current input and a direct current output.
FIG. 12 is a diagram showing a second embodiment of a high-frequency modulation / demodulation multiphase rectifier according to the invention of this application that constitutes a 30-phase rectifier circuit;
FIG. 13 is a diagram illustrating the effect of an input filter on a 30-pulse rectified waveform.
FIG. 14 is a diagram showing a high-frequency modulation / demodulation multiphase rectifier according to a third embodiment of the invention of this application of a low voltage and large current insulation type.
FIG. 15 is a diagram showing a drive circuit in the case of performing synchronous rectification instead of the main rectifier diode.
BEST MODE FOR CARRYING OUT THE INVENTION
[0005]
Hereinafter, the invention of this application will be described in detail by embodiments.
For convenience of explanation, first, the total harmonic distortion factor (THD) and its main components, the parallel series filter capacity, the THD when the filter is added, and the many when the multiphase rectification is performed at a low frequency for low THD and low noise. The relationship between the phase conversion transformer capacity and the input / output power ratio is summarized in FIG. The numerical value of THD in the
Many current AC devices, except for special devices such as measuring instruments, have a voltage distortion rate and a current distortion rate of 5% or less for THD and 3% or less for each harmonic individual component as general standards. Therefore, in order to satisfy the above-mentioned standard without an input filter, it is necessary to increase the number of phases to 30 pulses or more from the measured values in the table of FIG. 3, and for 18 pulses and 24 pulses, the LC filter (2) illustrated in FIG. Adds to satisfy the standard.
On the other hand, in 6 pulses and 12 pulses, the standard is often not satisfied unless the active type is shown in FIGS. 2- (4) and (5), and FIG. 2- (6) reduces the capacity of the active filter switch. Therefore, this is an example in which a passive filter is used together.
In any case, as far as it is seen from the contents published in other academic papers etc., up to about 4% THD is the active ability value for fluctuations in a wide range of load current and load power factor including transient phenomena. Presumed. In addition, since the active type uses PWM control with a large pulse width variation in principle, a high-power device cannot suppress the noise spectrum power up to the 30 MHz band, which may affect communication devices, medical devices, and the like.
On the other hand, the passive type shown in FIGS. 2- (1) and (2) does not perform high-frequency switching in principle, and partially switches in FIG. 2- (3). For the purpose of canceling the 11th and 13th harmonics, control close to the fixed pulse phase is performed with power of about 5% of the DC output power. Therefore, there is no generation of large noise power as in the active type. On the other hand, the transformer for multiphase conversion shown in FIG. 2 is required, and it is difficult to achieve a reduction in size and weight.
The high-frequency modulation / demodulation multi-phase rectifier according to the invention of this application uses a high-frequency transformation instead of a low-frequency transformer as shown in FIGS. 2- (1), (2) and (3). There is a big feature in the place that was realized by using the vessel. For this purpose, the ring modulation technology that has been used in communication technology has been introduced in the electric power field. In other words, the ring modulated wave power is applied to the primary winding of three-phase or single-phase three-winding high-frequency transformer, and depending on the connection configuration of the secondary side main winding and multiple auxiliary (tertiary) windings The low-frequency component included in the modulation waveform is subjected to multiphase conversion, and the diode constituting the demodulation circuit itself also serves as a multiphase full-wave rectifier. In addition, in order to minimize the switching noise caused by the digitization, switch driving is performed with almost no off period. That is, by driving each switch at a duty ratio close to 50% on one side compared to the conventional PWM control switching method, the continuity of the AC input side current is substantially maintained, and the high frequency noise component is reduced to a fraction.
The invention of this application relates to a high-frequency modulation / demodulation multiphase rectifier capable of reducing 1-2% THD and high-frequency band noise power, which is difficult to achieve with the active type, while maintaining the same small size and light weight as the active type PFC. Specific examples are described below.
FIG. 4 shows the configuration of the high-frequency modulation / demodulation multiphase rectifier according to the first embodiment of the present invention. The rectifier of this embodiment is an example of an 18-phase (pulse) rectifier, but this is selected for convenience because the circuit scale is small in the multi-phase conversion and is easy to explain. Even if it becomes a 36 and 42 phase rectifier, the basic principle does not change.
In FIG. 4, (1) is a series reactor, which plays a role of suppressing harmonics together with the LC type homogeneous filter (2). (3) is a direct-coupled three-phase full-wave rectifier that shares 1/3 of the total DC output current. (4a), (4b) and (4c) are ring modulation wave demodulator and auxiliary three-phase full-wave rectifier, (4a)
As described above, the input AC power is modulated at a fixed frequency of 10 KHz or more by the ring-modulated wave power generators (7a), (7b) and (7c), and the R, S, T interphase voltages are modulated. Is added to the primary windings (8a), (8b) and (8c) of the vessel (6). The secondary windings (9a), (9b) and (9c) and the tertiary windings (10a), (10b) and (10c) of the high-frequency multi-phase conversion transformer (6) are shown, for example, in FIG. By connecting as shown above, the envelope (envelope) is modulated with 9 low-frequency AC phases from the terminal (1) (for convenience, the encircled numbers in the drawing are shown in parentheses) to the terminal (9). A ring modulation waveform is generated.
6 (b), 6 (c) and 6 (d) show waveform examples of the prototype device. Due to the effect of the lossless snubber, no spike noise is generated in the high frequency pulse, and the waveform of the input AC current waveform. -(A) is not much different from the current waveform at the time of polyphase AC conversion by the low-frequency transformer.
The switches (S1) and (S2) shown in FIGS. 5 (a) and 5 (c) are used as bidirectional switches by connecting a commonly used MOS-FET to the back. It may be a bidirectional IGBT that has been developed recently. Further, the drive pulse width of the gate drive circuit (11) is fixed at a 50% duty ratio (dead time), and when pulse width modulation is not performed, the full-wave rectified output waveform is shown in FIG. 7- (a). Thus, it can be seen that the voltage and current waveforms have a small high-frequency noise component, and the noise component is reduced by a fraction of that in FIG. 7- (b) where normal pulse width modulation is performed. The gate drive circuit (11) switches the switches (S1), (S2), (S7), and (S8) of FIG. 5- (a) at the same time, and switches (S3), (S4), (S5) in the opposite phase. And (S6) are simultaneously driven. In this operation, all the blocks of the ring modulated wave power generators (7a), (7b) and (7c) in FIG. 4 are driven in the same phase, and normal operation is ensured by aligning the pulse phases.
Since the rectifying operation of the high-frequency modulation / demodulation multiphase rectifier of FIG. 4 is slightly complicated, it will be described with reference to FIGS. FIG. 8C illustrates the relationship between the voltage between the secondary winding terminals of the high-frequency multi-phase conversion transformer (6) and the modulation wave phase using only the low-frequency component. The voltage vector of the transformer secondary winding at the pulse phases (1), (2), (3) and (4) is represented by the thick arrows in FIG. It was. Each of the phase currents during phase (3) is (I1 + I3) in the R phase, (I0-I4-I3) in the S phase, and (I2 + I4) in the T phase, and flows in each secondary winding of the transformer. Flows through I1, I2, I3, and I4 indicated by dotted dotted lines. Here, I0 is the input current of the auxiliary rectifier in FIG. 4 and its amplitude value is equal to the 18-pulse rectified DC output, and is an alternating current component of positive and negative 1 pulse each having a time ratio of 1/18, and I1, I2, I3 and I4 is a current value flowing through each secondary winding of the transformer during the phase (3), and the value is between the coils of the secondary winding as shown in the table of FIG. 8- (a). It is easily calculated from the voltage. In the voltage vector indicated by the arrow (2) in FIG. 8B, only the currents I1 and I2 and I3 and I4 are switched. In this case, the currents in the R1 phase and the T1 phase are also switched.
These modes are shown on the time axis in the input AC current waveform in Fig. 8- (a), and (1), (2), (3) and (4) in Fig. 8- (b) are shown in Fig. 8- (b). And corresponding to the phase numbers in FIG. FIG. 10 is a diagram in which the above-described phase relationship is added in order to clarify the relationship between the currents of the three-phase input phases. In addition, during the phases (1) and (4), the ring modulated wave power generator (7a), ( 7b) and (7c), high frequency multiphase conversion transformer (6) and ring modulated wave demodulator / auxiliary three phase full wave rectifier (4a), impedance from high frequency multiphase rectifier composed of (4b) and (4c) Low direct-coupled three-phase full-wave rectifier (3) shares the load current. Therefore, in the sections (1) and (4), the burden on the high-frequency multiphase rectifier is reduced, and the efficiency of the entire rectifier is improved. On the other hand, due to the presence of the direct-coupled three-phase full-wave rectifier (3), it is not possible to ensure insulation between the input and output as seen in the entire rectifier.
FIG. 11 shows the current distribution of each part when the 18-phase rectifier circuit is configured by removing the direct-coupled three-phase full-wave rectifier (3) from FIG. The rectifier (4a) doubles as two positive and negative pulses (positive and negative pulses I0 in FIG. 10) borne by the direct-coupled three-phase full-wave rectifier (3). An output insulation type high frequency modulation / demodulation multiphase rectifier is realized. In this example, the three-phase full-wave bridge (4a) bears 2/3 of the total output, and (4b) and (4c)
Next, a high frequency modulation / demodulation multiphase rectifier according to a second embodiment of the present invention will be described. The rectifier of this embodiment is an example of a non-insulated 30-phase rectifier, and its circuit connection example is shown in FIG. In FIG. 12, the same elements as those in FIG. The difference between the rectifier of this embodiment and the 18-phase rectifier of FIG. 4 is that the transformer secondary winding is multi-phased (9 to 15 phases) and accompanying this, the auxiliary three-phase rectifiers (4d) and (4e). ) Is added, and the current value distribution of each part has changed. In addition, the transformer secondary winding is connected in the manner shown by the dotted line in the figure (in this case, the secondary winding capacity increases by several percent), and there are a few variations. Is a generally known technique.
FIG. 13 shows a current waveform of the circuit shown in FIG. In the figure, (A) shows a THD of about 4% when both the series reactor (1) and the homogeneous filter (2) (resonant at 50 Hz × 30 = 1.5 KHz) are not present, and (B) shows a 3% voltage drop. When only the series reactor (1) is inserted, 2.6% THD is shown, and (C) shows that 1% THD is obtained when the same series reactor (1) and the same order filter (2) are added. Have confirmed. This value is characteristic of the invention of this application because it has been difficult to realize any conventional high power factor rectifier circuit. At this time, the capacity of the three-phase series reactor (1) is 3.7% of the DC output power, the reactor capacity of the homogeneous filter (2) is 1.2% of the DC output power per single phase, and the resonance capacitor is delta. When connected, it is per 0.5 μF / phase /
When the directly connected three-phase full-wave rectifier (3) is removed from the circuit of FIG. 12, the entire 30-phase rectifier has a high-frequency multi-phase conversion transformer in which the AC input and the DC output are similar to the 18-phase rectifier of FIG. Insulated by (6). In this case, the input and output current (2/15) of the direct-coupled three-phase full-wave rectifier (3) 1/2 I, (3/15) I is added to the three-phase full-wave rectifier (4a), and the input and output current share of the three-phase full-wave rectifier (4a) is (6/15) 1/2 I, (6/15) I. The capacity of the high-frequency multiphase conversion transformer (6) and the ring modulated wave power generators (7a), (7b) and (7c) is (5/4). 1/2 Since it does not change except that it doubles, a specific example is omitted. When the input / output circuit is insulated, the voltage ratio can be freely selected according to the primary, secondary, and tertiary winding ratio of the high-frequency transformer.
FIG. 14 shows an example of 18 pulse rectification of a 12V 1000 A level power source high frequency reduction type rectifier. Using the circuit of FIG. 5- (d) for the ring modulated wave power generators (7a), (7b) and (7c), the switches (S9) and (S10) are driven at a time ratio of 50%-(dead time). The switches (S11) and (S12) have the same duty ratio, but the power supply impedance of the ring modulated wave power generators (7a), (7b) and (7c) is always kept low by the known phase difference PWM control. Stable operation can be performed. In this case, the DC output can be made variable by 10 to 100% by phase difference control, but in this case, an additional noise filter on the AC power supply side is required. Even when the Schottky barrier type is used for the rectifier diode, the overall efficiency can be maintained at 90% or more.
In principle, 24 pulses, 36 pulses, 42 pulses, etc. can be easily realized by combining the secondary and tertiary windings of the high-frequency multiphase transformer (6) and adding an output demodulator / rectifier diode. Since it is clear that these are examples, explanations of these examples are omitted.
FIG. 15 shows a drive circuit for performing synchronous rectification instead of the main rectifier diode in order to further increase the efficiency. In this example, the simplest three-phase full wave is shown. However, the input power source from low frequency to high frequency, and the multiphase AC power source of 6, 12, 24, 30, 36 and 42 phase, In all rectifier circuits, an auxiliary rectifier circuit that has exactly the same configuration as the main rectifier element is composed of a photo MOS switch, and the main switch with the same arrangement is driven by the photo mos switch. Thus, synchronous rectification can be realized with a simple circuit configuration.
Q in the figure 1 ~ Q 6 Is the main switch, PS 1 ~ PS 6 Is a photo moss switch, E 1 ~ E 6 Indicates a driving power source. Dummy resistor R L To adjust the current through the photodiode.
By adopting the above configuration, the voltage drop of about 0.6 V of the Schottky barrier diode can be normally reduced to 0.1 V or less, and the efficiency of the low-voltage, large-current rectifier circuit can be improved. Note that, from the standpoint of total economy, not all rectifying elements but only elements having a large current share may be used for synchronous rectification.
Further, in the device configuration of the invention of this application, since a large-capacity electrolytic capacitor is not used, a long life can be expected, and furthermore, the inrush current when power is turned on is extremely small.
Claims (4)
これに並列して設けられる三組のリング変調波電力発生器と三相複巻線高周波多相変換変圧器と相数に対応して設けられる複数のリング変調波復調器兼補助三相全波整流器を有する高調波補正回路を備え、
前記三相複巻線高周波多相変換変圧器の各相一次側巻線はリング変調波電力発生器の出力とそれぞれ接続され、
前記三相複巻線高周波多相変換変圧器の二次側から、主巻線と複数の補助巻線との組み合わせ接続により、リング変調された多相交流出力電圧の内の一組を多相出力端子から取出し、前記複数のリング変調波復調器兼補助三相全波整流器に入力させ、
前記高調波補正回路の直流出力を前記直結主三相全波整流器の直流出力と並列接続し、
三相交流電源側から見て等価的に６ｎ相（ｎは３〜７の整数）の多相全波整流回路を構成することを特徴とする高周波変復調多相整流装置。A direct-coupled main three-phase full-wave rectifier that converts three-phase AC input from a three-phase AC power source into DC,
Three sets of ring-modulated wave power generators and three-phase multi-winding high-frequency multi-phase conversion transformers provided in parallel, and a plurality of ring-modulated wave demodulator and auxiliary three-phase full-waves provided corresponding to the number of phases A harmonic correction circuit having a rectifier is provided,
Each phase primary winding of the three-phase multi-winding high-frequency multi-phase conversion transformer is connected to the output of the ring modulated wave power generator,
From the secondary side of the three-phase multi-winding high-frequency multi-phase conversion transformer, a combination of a main winding and a plurality of auxiliary windings is used to connect one set of ring-modulated multi-phase AC output voltages to a multi-phase. Take out from the output terminal, input to the ring modulated wave demodulator and auxiliary three-phase full-wave rectifier,
The DC output of the harmonic correction circuit is connected in parallel with the DC output of the directly connected main three-phase full-wave rectifier,
A high-frequency modulation / demodulation multiphase rectifier comprising a 6n-phase (n is an integer of 3 to 7) multiphase full-wave rectifier circuit equivalently as viewed from the three-phase AC power supply side.
前記三相複巻線高周波多相変換変圧器の各相一次側巻線はリング変調波電力発生器の出力とそれぞれ接続され、
前記三相複巻線高周波多相変換変圧器の二次側から、主巻線と複数の補助巻線との組み合わせ接続により、リング変調された多相交流出力電圧の内の一組を多相出力端子から取出し、前記複数のリング変調波復調器兼三相全波整流器に入力させ、
前記高調波補正回路の直流出力を直接負荷に供給し、
三相交流電源と負荷側が前記高周波多相変換変圧器により絶縁されることを特徴とする高周波変復調多相整流装置。Three sets of ring modulated wave power generators and three-phase double-winding high-frequency multi-phase conversion transformers connected to a three-phase AC power source and multiple ring-modulated wave demodulator and three-phase full wave A harmonic correction circuit having a rectifier is provided,
Each phase primary winding of the three-phase multi-winding high-frequency multi-phase conversion transformer is connected to the output of the ring modulated wave power generator,
From the secondary side of the three-phase multi-winding high-frequency multi-phase conversion transformer, a combination of a main winding and a plurality of auxiliary windings is used to connect one set of ring-modulated multi-phase AC output voltages to a multi-phase. Take out from the output terminal, input to the ring modulated wave demodulator and three-phase full-wave rectifier,
Supply the direct current output of the harmonic correction circuit directly to the load,
A high-frequency modulation / demodulation multiphase rectifier, wherein a three-phase AC power supply and a load side are insulated by the high-frequency multiphase conversion transformer.
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| JPS56145776A (en) * | 1980-03-24 | 1981-11-12 | Hisao Matsumoto | 3-phase frequency converter improved in ac current waveform |
| JP2000050634A (en) * | 1998-07-29 | 2000-02-18 | Mitsubishi Electric Corp | Alternating current, direct current power conversion device |
| JP2000354376A (en) * | 1999-06-10 | 2000-12-19 | I Hitsutsu Kenkyusho:Kk | Ac voltage regulator |
| JP2003088124A (en) * | 2001-09-06 | 2003-03-20 | Toshiba Corp | Rectifier |
| JP2003304680A (en) * | 2002-04-09 | 2003-10-24 | Chiyoda:Kk | Two-way step-up/down converter both for alternating current and for direct current |
| JP2004215469A (en) * | 2003-01-09 | 2004-07-29 | Renesas Technology Corp | Switching power supply and semiconductor integrated circuit for controlling power supply |
| JP2005287251A (en) * | 2004-03-30 | 2005-10-13 | Kawasaki Heavy Ind Ltd | Rectifying device |
-
2006
- 2006-12-11 WO PCT/JP2006/324646 patent/WO2007069556A1/en active Application Filing
- 2006-12-11 CN CN2006800468068A patent/CN101331671B/en active Active
- 2006-12-11 JP JP2007550161A patent/JP4808221B2/en active Active
Patent Citations (7)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JPS56145776A (en) * | 1980-03-24 | 1981-11-12 | Hisao Matsumoto | 3-phase frequency converter improved in ac current waveform |
| JP2000050634A (en) * | 1998-07-29 | 2000-02-18 | Mitsubishi Electric Corp | Alternating current, direct current power conversion device |
| JP2000354376A (en) * | 1999-06-10 | 2000-12-19 | I Hitsutsu Kenkyusho:Kk | Ac voltage regulator |
| JP2003088124A (en) * | 2001-09-06 | 2003-03-20 | Toshiba Corp | Rectifier |
| JP2003304680A (en) * | 2002-04-09 | 2003-10-24 | Chiyoda:Kk | Two-way step-up/down converter both for alternating current and for direct current |
| JP2004215469A (en) * | 2003-01-09 | 2004-07-29 | Renesas Technology Corp | Switching power supply and semiconductor integrated circuit for controlling power supply |
| JP2005287251A (en) * | 2004-03-30 | 2005-10-13 | Kawasaki Heavy Ind Ltd | Rectifying device |
Also Published As
| Publication number | Publication date |
|---|---|
| CN101331671A (en) | 2008-12-24 |
| WO2007069556A1 (en) | 2007-06-21 |
| JPWO2007069556A1 (en) | 2009-05-21 |
| CN101331671B (en) | 2012-06-06 |
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