JP4789431B2 - Underwater detector - Google Patents

Description

  The present invention relates to an underwater detection device such as a scanning sonar that obtains underwater information based on transmission / reception of ultrasonic waves, and more particularly to a technique for removing the influence of interference waves.

  Scanning sonar used in fishing grounds (hereinafter simply referred to as “sonar”) transmits ultrasonic waves from a transducer with multiple transducers into the water and analyzes the echo signals received by each transducer. This is a device for obtaining underwater information. Patent Document 1 and Patent Document 2 described later describe the configuration of such a scanning sonar. By the way, signals received by the scanning sonar include ultrasonic waves (hereinafter referred to as “interference waves”) transmitted from sonars of other ships operating in the same fishing ground, in addition to echo signals from fish schools. When this interference wave is received by the transmitter / receiver, an image of the interference wave appears on the display portion of the sonar, and this image obscures the image of the echo signal from the fish school, making it difficult to detect the fish school. In the case of interference waves received directly from other ships, it appears as a relatively clear image on the screen with high intensity, so it is easy for the occupant to identify the interference waves, but they are received after they have been reflected once on the seabed. In the case of the interference wave to be generated, since the intensity is small, it is generally difficult for the occupant to distinguish the image of the interference wave and the image of the school of fish on the screen.

  As a measure for removing the influence of the interference wave as described above, it is conceivable to use a narrow-band vibrator having a narrow passband having a frequency characteristic (fo is a center frequency) as shown in FIG. In this way, the influence of interference waves having frequencies outside the passband can be removed. However, in order to remove interference from other ships that transmit ultrasonic waves of the same frequency, the transmission frequency of the ship must be changed from the center frequency fo of the pass band to another frequency f within the pass band. Don't be. In this case, as can be seen from FIG. 12, the level of the echo signal decreases. That is, the conventional sonar using the narrow band transducer cannot remove the interference wave without deteriorating the sensitivity to the echo signal. Therefore, in order to avoid the influence of interference waves, a plurality of types of sonar with different bands are required so that the sonars do not receive ultrasonic waves transmitted from each other. For this reason, there exists a problem that the cost which manufactures and maintenance of a sonar will increase.

  On the other hand, in recent years, a sonar using a wide-band vibrator having a frequency characteristic (fo is a center frequency) as shown in FIG. In this wideband sonar, when interference waves from other ships appear on the image, the band of the filter to be applied to the received signal and the frequency of the transmission signal are linked, that is, the center frequency of the filter is set to the transmission signal. By switching so as to match the frequency, it is possible to remove the influence of the interference wave without reducing the echo reception sensitivity.

Japanese Patent Laid-Open No. 2003-84060 (paragraphs 0003 to 0005, 0021 to 0025, FIGS. 1 and 10) JP 2001-99914 (paragraph 0067, FIG. 12)

  However, in the above-described broadband sonar, when the power of the interference wave is larger than the power of the echo signal and the interference wave is saturated at the front stage of the filter, the power of the echo signal is lowered and the SN ratio of the echo signal is deteriorated. There is a problem. This will be described below.

  FIG. 14 is a block diagram showing a part of the receiving system circuit in the broadband sonar. The illustration of the transmission system circuit is omitted. 70 is a broadband transducer for transmitting and receiving ultrasonic waves, 71 is an analog amplifier (hereinafter simply referred to as “amplifier”) for amplifying a signal received by the broadband transducer 70, and 72 is a predetermined band of signals output from the amplifier 71. A filter 73 that passes only the above signal 73 is an A / D converter that performs A / D conversion on the signal output from the filter 72. FIG. 14 shows only one broadband oscillator 70, but in actuality, only a few hundred oscillators 70 are provided, and 71 to 73 circuits are provided for each oscillator. Comes with.

  In the above configuration, when the broadband transducer 70 receives an echo (frequency f1) reflected by a fish school or the like and an interference wave (frequency f2) from another ship, the signal level of the interference wave is much higher than the signal level of the echo. Is larger, the interference wave signal on which the echo signal is superimposed is input to the amplifier 71. However, since the interference wave is saturated before the filter 72 due to amplification by the amplifier 71, the echo waveform cannot be normally extracted from the interference wave, and the power of the echo signal is greatly reduced. As a result, the S / N ratio of the echo signal is lowered and the detection performance is deteriorated. As the effective band of the vibrator becomes wider, it becomes easier to receive interference waves from a larger number of devices (for example, fish detectors installed separately from sonar), and this deterioration in detection performance becomes more significant.

  On the other hand, if an analog filter for removing the interference wave is arranged in front of the amplifier 71, the amplifier 71 does not saturate the interference wave, so that the echo signal can be taken out normally. It is possible to prevent a decrease in the SN ratio due to power reduction. In this case, since the frequency of the interference wave is indefinite, it is necessary to use an analog programmable filter having a variable pass band as the analog filter provided in the preceding stage of the amplifier 71. However, in the case of a typical sonar, hundreds to thousands of resonators are used, and using an analog programmable filter for each resonator significantly increases the cost and circuit scale. . Therefore, this method is not practical.

On the other hand, in place of the above-described analog programmable filter, it is conceivable to provide a digital programmable filter 74 having a variable pass band at the subsequent stage of the A / D converter 73 as shown in FIG. The filter 74 can be configured by a single IC group common to each transducer by processing the received signal from each transducer in a time-sharing manner, and can suppress an increase in cost. However, in this case, it is necessary to lower the amplification factor of the amplifier 71 so that the interference wave is not saturated in the amplifier 71. On the other hand, in the A / D converter 73, as shown in FIG. 16, when the sampling value of the echo signal is quantized, the sampling value and the quantization threshold values A ₁ , A ₂ , A ₃ ,. The generation of quantization noise is unavoidable, but the power of this quantization noise is determined by the characteristics of the A / D converter 73 and does not depend on the amplification factor of the amplifier 71. Therefore, when the amplification factor of the amplifier 71 is lowered as described above, only the power of the echo signal is reduced although the quantization noise is not reduced. As a result, the SN ratio of the echo signal is lowered and the detection performance is deteriorated. There's a problem.

  The present invention solves the above-described problems, and an object of the present invention is to realize an underwater detection device capable of removing the influence of interference waves without degrading the detection performance at a low cost. It is in.

An underwater detection apparatus according to the present invention is an underwater detection apparatus that obtains underwater information by transmitting ultrasonic waves from a transducer having a plurality of transducers into the water and analyzing echo signals received by the transducers. There are a plurality of each oscillator , amplifying the output of the oscillator, amplifiers having different amplification factors, a multiplexer that sequentially switches the outputs of these amplifiers, and the output of this multiplexer, And a data selection unit for selecting an output corresponding to the amplifier having the largest amplification factor, and an amplitude phase compensation unit for correcting the amplitude discontinuity and the phase discontinuity for the output selected by the data selection unit, And a beam forming unit that forms a received beam by beam forming processing based on the output of the amplitude phase compensation unit .

By doing so, in the time range in which only a weak echo signal is included in the signal received by the transducer, an echo signal is obtained with a good S / N ratio by using the output data of the amplifier having the largest amplification factor. be able to. On the other hand, by using the output data of the amplifier with the largest amplification factor in the range where saturation does not occur, in the time range where a large power interference wave is superimposed on the weak echo signal received by the transducer, the echo signal without lowering the power, only interference Watarunami can be removed.

The underwater detection device according to the first aspect of the present invention is selected by an A / D converter that generates complex data for the output of each amplifier based on the output of the multiplexer and outputs the complex data to the data selection unit, and the data selection unit. And a digital programmable filter having a variable pass band that allows a frequency component of a predetermined band of the output to pass therethrough. The amplitude phase compensator corrects the amplitude discontinuity and the phase discontinuity by multiplying the output of the digital programmable filter by a complex correction coefficient.

  By doing so, in the time range in which only a weak echo signal is included in the signal received by the transducer, an echo signal is obtained with a good S / N ratio by using the output data of the amplifier having the largest amplification factor. be able to. On the other hand, by using the output data of the amplifier with the largest amplification factor in the range where saturation does not occur, in the time range where a large power interference wave is superimposed on the weak echo signal received by the transducer, the echo signal Only the interference wave can be removed by the digital programmable filter without reducing the power. Further, since no analog programmable filter is used, there is no problem that the cost is significantly increased. Furthermore, since the multiplexer and the amplitude / phase compensation unit are provided, it is not necessary to use individual A / D converters for a plurality of amplifiers, so that the cost can be further reduced.

  In the underwater detection device of the first embodiment, for example, the following signal processing is performed. The A / D converter outputs real part data and imaginary part data of complex data corresponding to the output of each amplifier by sampling in a predetermined cycle. The data selection unit has a buffer memory for holding a complex data string composed of M + 1 complex data corresponding to each amplifier, where M is the order of the digital programmable filter, and is held in the memory. In addition, each of the complex data in each complex data string is detected with the maximum absolute value, and among the detected maximum values, complex data having an absolute value equal to or less than a predetermined threshold is selected, and the selected complex data The data having the maximum absolute value is detected from the data, and a complex data string corresponding to the amplifier having the maximum value is output. The digital programmable filter calculates a filter coefficient according to the specified center frequency and passband width value, and executes a predetermined operation on the complex data string output from the data selection unit using the filter coefficient. And output the calculation result.

をそれぞれ保持するバッファメモリを有する。データ選択部は、各アンプ番号ｋについて、
｛｜Ｚ_ｋ［ｎ］｜，｜Ｚ_ｋ［ｎ＋１］｜，…，｜Ｚ_ｋ［ｎ＋Ｍ］｜｝
の中の最大値ｍａｘ_ｋ（ｋ＝１，２，３，…，Ｎ）、すなわち各複素データ列における複素データの中で絶対値が最大のものをそれぞれ検出する。次に、検出した最大値のうち、所定の閾値Ｓ以下の絶対値をもつｍａｘ_ｋを選択し、さらに、選択したｍａｘ_ｋの中で最大のものを検出して、当該最大値に対応するアンプを同定する。そして、同定したアンプのアンプ番号をｋ０として複素データ列
｛Ｚ_ｋ０［ｎ］，Ｚ_ｋ０［ｎ＋１］，…，Ｚ_ｋ０［ｎ＋Ｍ］｝
を出力する。
デジタル・プログラマブル・フィルタは、指定された中心周波数と通過帯域幅の値に応じてフィルタ係数ｈ［ｍ］（ｍ＝０，１，２，…，Ｍ）を算出し、次に、データ選択部から出力される上記複素データ列とフィルタ係数との畳み込み演算

を実行して、その演算結果Ｙ［ｎ］を出力する。 As a more specific example, when the number of each analog amplifier is k (k = 1, 2, 3,..., N), the A / D converter performs sampling of each amplifier by sampling at a predetermined period. Complex data for output Z _k [n] = I _k [n] + jQ _k [n] (n = 0, 1, 2,...)
Real part data I _k [n] and imaginary part data Q _k [n] are output.
The data selection unit is a complex data string composed of M + 1 complex data corresponding to each amplifier, where M is the order of the digital programmable filter.

Each have a buffer memory. The data selection unit, for each amplifier number k,
{| Z _k [n] |, | Z _k [n + 1] |, ..., | Z _k [n + M] |}
, Max _k (k = 1, 2, 3,..., N), that is, complex data in each complex data string having the maximum absolute value is detected. Next, among the detected maximum values, max _k having an absolute value equal to or less than a predetermined threshold S is selected, and the maximum one of the selected max _k is detected, and an amplifier corresponding to the maximum value Is identified. Then, the amplifier number of the identified amplifier is k0, and a complex data string {Z _k0 [n], Z _k0 [n + 1],..., Z _k0 [n + M]}
Is output.
The digital programmable filter calculates a filter coefficient h [m] (m = 0, 1, 2,..., M) according to the specified center frequency and pass bandwidth value, and then selects a data selection unit. Convolution of the complex data sequence output from the filter coefficient

And the operation result Y [n] is output.

  In the underwater detection device according to the first aspect, when all the absolute values of complex data detected from each complex data sequence are larger than the threshold value, the data selection unit outputs a complex data sequence corresponding to the amplifier having the smallest amplification factor. What should I do? By doing so, it is possible to minimize the influence of the saturation of the interference wave in the amplifier.

In the underwater detection device according to the second aspect of the present invention, the amplitude / phase compensation unit is provided in front of the digital programmable filter. The underwater detection device according to the second embodiment receives an A / D converter that generates complex data for the output of each amplifier based on the output of the multiplexer and outputs the complex data to the data selection unit, and the output of the amplitude phase compensation unit. And a digital programmable filter having a variable pass band for passing the frequency component of the predetermined band of the output. The amplitude phase compensator corrects the amplitude discontinuity and the phase discontinuity by multiplying the output of the data selection unit by a correction coefficient composed of a complex number.

  Even in such a second form of underwater detection device, in the time range in which only a weak echo signal is included in the signal received by the transducer, the output data of the amplifier having the largest amplification factor can be used to obtain a good SN An echo signal can be obtained by the ratio. On the other hand, by using the output data of the amplifier with the largest amplification factor in the range where saturation does not occur, in the time range where a large power interference wave is superimposed on the weak echo signal received by the transducer, the echo signal Only the interference wave can be removed by the digital programmable filter without reducing the power. Further, since no analog programmable filter is used, there is no problem that the cost is significantly increased. Furthermore, since the multiplexer and the amplitude / phase compensation unit are provided, it is not necessary to use individual A / D converters for a plurality of amplifiers, so that the cost can be further reduced.

  In the underwater detection device of the second form, for example, the following signal processing is performed. The A / D converter outputs real part data and imaginary part data of complex data corresponding to the output of each amplifier by sampling in a predetermined cycle. The data selection unit has a buffer memory for holding one piece of complex data corresponding to each amplifier, and among the pieces of complex data held in the memory, complex data having an absolute value equal to or less than a predetermined threshold value Select. Then, the selected complex data having the largest absolute value is detected and the complex data is output. The digital programmable filter calculates a filter coefficient according to the specified center frequency and passband width value, and executes a predetermined operation on the complex data output from the amplitude phase compensation unit using the filter coefficient. And output the calculation result.

  As a result, since the buffer memory only needs to hold one complex data for each amplifier, the selection control unit does not need to extract the maximum data from the complex data string for each amplifier. Accordingly, it is possible to simultaneously reduce the capacity of the buffer memory and the processing amount of the selection control unit.

  Also in the second embodiment, when all the absolute values of complex data are larger than the threshold value, the data selection unit may output complex data corresponding to an amplifier having the smallest amplification factor. By doing so, it is possible to minimize the influence of the saturation of the interference wave in the amplifier.

  Further, in the second embodiment, when the level of the DC offset included in the output of each amplifier is not sufficiently smaller than the level of the echo signal, a digital high-pass signal is inserted between the A / D converter and the data selection unit. A DC offset can be removed by providing a filter.

According to the underwater detection system of the present invention, regardless of the level of presence and interference waves of interference waves, always, the kill in that automatically selects the output of the amplifier with the optimal gain saturation does not occur, Interference waves can be removed without degrading the detection performance. Further, since an analog programmable filter is not used, there is no problem that the cost is significantly increased. Furthermore, since the multiplexer and the amplitude / phase compensation unit are provided, it is not necessary to use individual A / D converters for a plurality of amplifiers, so that the cost can be further reduced.

  FIG. 1 is a block diagram of a scanning sonar that is an embodiment of the underwater detection device according to the present invention. Here, only the reception system block is shown, and the transmission system block is omitted.

  Reference numeral 1 denotes a plurality of broadband vibrators (hereinafter referred to as “vibrators”) having a predetermined effective band, which convert incident ultrasonic waves into electrical signals. Each transducer 1 is provided on the surface of a cylindrical or spherical transducer, for example. A signal processing circuit 101 is composed of 2 to 12 blocks described below. Reference numeral 2 denotes an analog bandpass filter (hereinafter referred to as “bandpass filter”), which removes components other than the effective band of the vibrator 1 from the electrical signal extracted from the vibrator 1. Reference numerals 31 to 33 denote analog amplifiers (hereinafter referred to as “amplifiers”) that amplify the output of the bandpass filter 2, and the amplification factor G 1 of the amplifier 31, the amplification factor G 2 of the amplifier 32, and the amplification factor of the amplifier 33. G3 is different from each other. In the following description, it is assumed that G1 <G2 <G3. The subscript (1, 2, 3) of G is referred to as an amplifier number and is represented by k.

  Reference numeral 4 denotes an analog multiplexer (hereinafter referred to as “multiplexer”), which sequentially switches the outputs of the three amplifiers 31 to 33 and outputs them to the A / D converter 5. The A / D converter 5 converts the output of the multiplexer 4 into a digital signal, and outputs complex data corresponding to the amplifiers 31 to 33 by time division sampling, as will be described in detail later. A data selection unit 8 includes a selection control unit 6 and a buffer memory 7, and selects the optimum complex data output from the A / D converter 5. Reference numeral 9 denotes a digital programmable filter having a variable pass band, and passes the frequency component of a predetermined band by performing an operation on the complex data selected by the data selection unit 8 using a predetermined filter coefficient. Let Reference numeral 12 denotes an amplitude phase compensator, which includes a correction coefficient memory 10 and a multiplier 11, and corrects the amplitude discontinuity and the phase discontinuity in the output of the digital programmable filter 9. Detailed operations of the data selection unit 8, the digital programmable filter 9, and the amplitude / phase compensation unit 12 will be described later.

  In FIG. 1, reference numerals 102, 103,... 10 n are signal processing circuits having the same configuration as the signal processing circuit 101. That is, each of the vibrators 1 is provided with a signal processing circuit including the above-described 2 to 12 blocks. However, the digital programmable filter 9 is set so that one filter 9 corresponds to one transducer 1 for convenience because the pass band is set corresponding to each transducer 1. However, in practice, the filter 9 is formed of a single IC group common to each transducer 1 in order to process the received signal from each transducer 1 in a time division manner.

  Reference numeral 13 denotes a gain adjustment unit that is super-existing with respect to the output of the amplitude phase compensation unit 12 so that the amplitude of the echo signal from the same target is constant without depending on the distance between the transducer and the target. A process of multiplying by a TVG (Time Variable Gain) coefficient that increases in accordance with the elapsed time after transmitting the sound wave is performed. A beam forming unit 14 forms a received beam by performing a known beam forming process on the received signals of each channel processed by the signal processing circuits 101, 102,... 10n and the gain adjusting unit 13. . An image processing unit 15 generates echo image data based on the received beam data generated by the beam forming unit 14. Reference numeral 16 denotes a display unit composed of a color liquid crystal display, for example, which displays the video data generated by the video processing unit 15 on the screen.

  Next, the operation of the scanning sonar having the above configuration will be described in detail. Each transducer 1 receives an output from a transmission circuit (not shown) and transmits ultrasonic waves of a predetermined frequency into water. The transmission beam at this time is emitted from the transmitter / receiver toward all directions in the water at a predetermined tilt angle (declining angle) to form an umbrella-shaped beam. The transmitted ultrasonic waves are reflected by the underwater fish school and the bottom of the water and are received by each transducer 1 as an echo. Each transducer 1 converts the received echo into an electrical signal. The reception signal output from each transducer 1 is input to signal processing circuits 101, 102,. Since the signal processing circuit has the same configuration as described above, the signal processing circuit 101 will be described below.

  The received signal from the vibrator 1 is subjected to removal of components other than the effective band of the vibrator 1 by the bandpass filter 2 and is simultaneously input to the three amplifiers 31 to 33. The amplifier 31 amplifies the input reception signal with the amplification factor G1, the amplifier 32 amplifies the input reception signal with the amplification factor G2, and the amplifier 33 converts the input reception signal into the input reception signal. On the other hand, amplification is performed at an amplification factor G3. As described above, since the amplification factor has a relationship of G1 <G2 <G3, the output of the amplifier 31 is the smallest and the output of the amplifier 33 is the largest. Each output of the amplifiers 31 to 33 is input to the multiplexer 4. The multiplexer 4 switches the outputs of the three amplifiers 31 to 33 in the order of 31 → 32 → 33 → 31 → 32 → 33... And outputs it to the A / D converter 5.

  The A / D converter 5 performs time division sampling on the outputs of the amplifiers 31 to 33 to generate complex data for each amplifier output. FIG. 2 is a diagram for explaining time-division sampling. 2A shows the output signal waveform of the amplifier 31, FIG. 2B shows the output signal waveform of the amplifier 32, and FIG. 2C shows the output signal waveform of the amplifier 33, respectively. Here, as an example, a case where the A / D converter 5 performs sampling once every 0.25 period of the transmission signal is taken as an example.

When the output signal of each amplifier is complex data, the imaginary part data Q ₁ [0] of the output signal of the amplifier 31 is extracted at time t1, and the real part data I ₁ [0] is extracted at time t2. At time t3, imaginary part data Q ₂ [0] of the output signal of the amplifier 32 is extracted, and at time t4, real part data I ₂ [0] is extracted. At time t5, imaginary part data Q ₃ [0] of the output signal of the amplifier 33 is extracted, and at time t6, real part data I ₃ [0] is extracted.

At time t7, imaginary part data Q ₁ [1] of the output signal of the amplifier 31 is extracted, and at time t8, real part data I ₁ [1] is extracted. The imaginary part data Q ₂ [1] of the output signal of the amplifier 32 is extracted at time t9, and the real part data I ₂ [1] is extracted at time t10. The imaginary part data Q ₃ [1] of the output signal of the amplifier 33 is extracted at time t11, and the real part data I ₃ [1] is extracted at time t12.

At time t13, imaginary part data Q ₁ [2] of the output signal of the amplifier 31 is extracted, and at time t14, real part data I ₁ [2] is extracted. At time t15, the imaginary part data Q ₂ [2] of the output signal of the amplifier 32 is extracted, and at time t16, real part data I ₂ [2] is extracted. At time t17, imaginary part data Q ₃ [2] of the output signal of the amplifier 33 is extracted, and at time t18, real part data I ₃ [2] is extracted.

In the same manner, the A / D converter 5 performs complex data on the output signals of the amplifiers 31 to 33 by performing time-division sampling as described above.
Z _k [n] = I _k [n] + jQ _k [n] (n = 0, 1, 2,...)
Real part data I _k [n] and imaginary part data Q _k [n] are output.

The complex data of each output signal generated by the A / D converter 5 in this way is sequentially stored in the buffer memory 7 of the data selection unit 8. Here, when the order of the digital programmable filter 9 is M (M = filter length−1), the buffer memory 7 stores M + 1 pieces of complex data corresponding to the outputs of the amplifiers 31 to 33, respectively. Is done. That is, the buffer memory 7 holds the following complex data string.
{Z ₁ [n], Z ₁ [n + 1],..., Z ₁ [n + M]}
{Z ₂ [n], Z ₂ [n + 1],..., Z ₂ [n + M]}
{Z ₃ [n], Z ₃ [n + 1],..., Z ₃ [n + M]}

Now, assuming that M = 3 for the sake of simplicity, four pieces of each complex data are stored in the buffer memory 7 as shown in FIG. 71 is an area where complex data corresponding to the amplifier 31 (k = 1) is stored, 72 is an area where complex data corresponding to the amplifier 32 (k = 2) is stored, and 73 is an amplifier 33 (k = 3). This is the area where the corresponding complex data is stored. FIG. 3A shows a complex data string in the buffer memory 7 when n = 0.
{Z ₁ [0], Z ₁ [1], Z ₁ [2], Z ₁ [3]}
{Z ₂ [0], Z ₂ [1], Z ₂ [2], Z ₂ [3]}
{Z ₃ [0], Z ₃ [1], Z ₃ [2], Z ₃ [3]}
Shows a state where is held. For example, Z ₁ [0] is complex data composed of I ₁ [0] and Q ₁ [0] in FIG. 2A, and Z ₁ [1] is I ₁ [1] and Q ₁ [1]. ] (And so on). Z ₂ [0] is complex data consisting of I ₂ [0] and Q ₂ [0] in FIG. 2B, and Z ₂ [1] is I ₂ [1] and Q ₂ [1]. ] (And so on). Z ₃ [0] is complex data composed of I ₃ [0] and Q ₃ [0] in FIG. 2C, and Z ₃ [1] is I ₃ [1] and Q ₃ [1]. ] (And so on).

When the selection process described later is completed for these complex data, the process shifts to n = 1, and the buffer memory 7 stores a complex data string as shown in FIG.
{Z ₁ [1], Z ₁ [2], Z ₁ [3], Z ₁ [4]}
{Z ₂ [1], Z ₂ [2], Z ₂ [3], Z ₂ [4]}
{Z ₃ [1], Z ₃ [2], Z ₃ [3], Z ₃ [4]}
Is retained. When the selection process described later is completed for these complex data, the process shifts to n = 2, and the buffer memory 7 stores a complex data string as shown in FIG.
{Z ₁ [2], Z ₁ [3], Z ₁ [4], Z ₁ [5]}
{Z ₂ [2], Z ₂ [3], Z ₂ [4], Z ₂ [5]}
{Z ₃ [2], Z ₃ [3], Z ₃ [4], Z ₃ [5]}
Is retained. Similarly, the buffer memory 7 sequentially holds the complex data obtained by time division sampling for each amplifier.

The selection control unit 6 reads the complex data held in the buffer memory 7 and identifies the amplifier number k0 of the optimum amplifier (a non-saturated amplifier with the highest amplification factor) according to the following procedure. That is, first, for each amplifier number k,
{| Z _k [n] |, | Z _k [n + 1] |, ..., | Z _k [n + M] |}
The maximum value max _k (k = 1, 2, 3) is detected. As a result, for example, as shown in FIG. 4, when n = 0, Z ₁ [2] is detected as the maximum value max ₁ corresponding to the amplifier 31 (k = 1) and corresponds to the amplifier 32 (k = 2). It is detected _Z 2 [1] as the maximum value max _{2, Z} 3 [3] as the maximum value max ₃ corresponding to the amplifier 33 (k = 3) is detected.

Next, max _k that takes a value equal to or less than a predetermined threshold is selected. Here, as shown in FIG. 5, an upper limit value S that the output of the A / D converter 5 can take is used as a threshold value. However, this is an example, and a value slightly smaller than the upper limit value S may be used as the threshold value. The fact that max _k exceeds the upper limit value S means that the amplifier output is saturated. Therefore, selecting a value equal to or lower than the upper limit value S means that only the non-saturated amplifier output is taken out. . Here, since max _k is an absolute value of a complex number, it can be expressed as a vector length on a complex plane as shown in FIG. The upper limit value S is represented by a circle with a radius S (broken line). In FIG. 6, as an example, max ₁ <S, max ₂ <S, and S <max ₃ are set. Therefore, in this case, max ₁ and max ₂ are selected. Next, the selection control unit 6 extracts the selected max _k having the maximum value, and outputs the amplifier number taking the maximum value as k0. As described above, since the amplification factor of the amplifier 31 to 33 have a relationship of G1 <G2 <G3, when comparing the value of max ₁ and max _2, towards max ₂ is greater than max _1. Therefore, max ₂ is extracted as the maximum value, and k0 = 2 is output as the optimum amplifier number.

FIG. 7 is a table showing the relationship between the values of max _{1 to} max ₃ and the amplifier number k0, taking the case where S = 31 as an example. For _Lee, all max 1 ~max ₃ is smaller than the upper limit value S = 31, because it becomes a selection candidate all three, the is the largest value among them is max _3, this max ₃ “3” is determined as the corresponding amplifier number. In the case of (b), max ₁ and max ₂ are smaller than the upper limit value S = 31, and max ₃ is larger than the upper limit value S = 31. Therefore, max ₁ and max ₂ are selection candidates. Then, max ₂ having a large value is adopted, and “2” is determined as the amplifier number corresponding to max ₂ . In the case of c, since only max ₁ is smaller than the upper limit value S = 31 and max ₂ and max ₃ are larger than the upper limit value S = 31, max ₁ is adopted, and “1” is set as the amplifier number corresponding to this max ₁ Is determined. In the case of d, all of max _{1 to} max ₃ are larger than S = 31 and are in a saturated state. In this case, “1” is unconditionally selected as the amplifier number k0. That is, the amplifier number of the amplifier 31 having the smallest amplification factor is employed. The reason for this is to minimize the influence of the saturation of the interference wave in the amplifier by selecting the smallest data in the saturated state.

As described above, the maximum value max _k in each column is detected for the complex data sequence for each amplifier held in the buffer memory 7, and this maximum value is compared with the upper limit value S. By selecting the largest value max _k among the certain maximum values max _k , it is possible to identify an amplifier that is not saturated and has the largest amplification factor. The selection control unit 6 outputs the amplifier number k0 to the buffer memory 7. The buffer memory 7 receives this amplifier number k0 and receives a complex data string { _Zk0 [n], _Zk0 [n + 1],..., _Zk0 [n + M]} corresponding to _k0 .
Is output to the digital programmable filter 9.

The digital programmable filter 9 is an FIR (Finite Impulse Response) filter of order M, and performs the following arithmetic processing on the complex data selected by the data selection unit 8 to obtain a predetermined band. Pass frequency components. First, the filter coefficient h [m] (m = 0, 1, 2,..., M) is calculated according to the center frequency and the pass bandwidth value designated by the operator using the input device (not shown). As described above, in this embodiment, for simplicity of explanation, M = 3 and the filter length (M + 1) is 4, so that four filter coefficients h [0] to h [3] are calculated. . Next, a convolution operation (product-sum operation) is performed on the complex data output from the data selection unit 8 and the filter coefficient, and the operation result Y [n] is output. This convolution operation is defined by the following equation.

FIG. 8 is an example of an arithmetic circuit that performs a convolution operation, and shows a circuit when M = 3 for simplicity. The arithmetic circuit includes delay units 51-53, multipliers 54-57, and adders 58-60. FIG. 8 illustrates a convolution operation of complex data of Z [0] to Z [3] and filter coefficients h [0] to h [3].
Y [0] = h [3] · Z [0] + h [2] · Z [1]
+ H [1] · Z [2] + h [0] · Z [3]
Is output.

  The calculation result obtained by the digital programmable filter 9 is input to the amplitude phase compensation unit 12. In the correction coefficient memory 10 of the amplitude phase compensation unit 12, correction coefficients corresponding to the amplification factors of the amplifiers 31 to 33 are stored in advance. This correction coefficient consists of complex numbers. The correction coefficient memory 10 receives the amplifier number k0 output from the selection control unit 6 and outputs a correction coefficient corresponding to the amplifier number k0. The multiplier 11 multiplies the output of the digital programmable filter 9 by the correction coefficient output from the correction coefficient memory 10. As a result, the amplitude discontinuity due to the difference between the amplification factors G1 to G3 of the amplifiers 31 to 33 and the phase discontinuity due to the difference in sampling time with respect to the outputs of the amplifiers 31 to 33 are corrected.

For example, when the amplitude and phase of the output of the amplifier 31 are used as a reference and it is desired to match the amplitude and phase to this, if the amplifier number selected by the data selection unit 8 is k0 = 2, the amplitude of the output of the amplifier 32 is The output of the filter 9 is multiplied by a correction coefficient C ₁ that corrects the phase to the amplitude and phase equivalent to the amplifier 31. If the number of the amplifier selected by the data selection unit 8 is k0 = 3, the correction coefficient C ₂ for correcting the amplitude and phase of the output of the amplifier 33 to the amplitude and phase equivalent to the amplifier 31 is set in the filter 9. Multiply the output. If the number of the amplifier selected by the data selection unit 8 is k0 = 1, no correction is necessary, and the output of the filter 9 is left as it is. Therefore, the correction coefficient memory 10 may hold at least two correction coefficients.

Since the output of the digital programmable filter 9 is a complex number represented by Z = R · e ^jθ (R: absolute value, θ: declination), the amplitude can be corrected by adjusting the value of R. The phase can be corrected by adjusting the value of θ. Therefore, the absolute value and the declination value of the correction coefficient (complex number) are selected as values for correcting the amplification factor and the sampling timing, respectively, and the correction coefficient is multiplied by the output of the filter 9 so that the amplitude and phase are properly set. Can be corrected.

  The complex data whose amplitude and phase are corrected in this way is input to the gain adjusting unit 13. As described above, the gain adjustment unit 13 performs a process of multiplying the output of the amplitude phase compensation unit 12 by the TVG coefficient that increases with the elapsed time after the ultrasonic transmission. Thereby, the amplitude of the echo signal is kept constant regardless of the distance between the transducer and the target. Since the gain adjusting unit 13 is composed of a known circuit and is not a main part of the present invention, the description of the circuit details is omitted.

  The output of the gain adjusting unit 13 is input to the beam forming unit 14. As described above, the beam forming unit 14 performs a beam forming process on the reception signal of each channel to form a received beam. In this case, since the received signal has been corrected by the amplitude / phase compensator 12 so as not to cause discontinuity in the amplitude and phase of the received signal, the received beam is normally formed. Since the beam forming unit 14 is also composed of a known circuit and is not a main part of the present invention, the detailed description of the circuit is omitted.

  The output of the beam forming unit 14 is input to the video processing unit 15. As described above, the video processing unit 15 generates echo video data based on the received beam data generated by the beam forming unit 14. Since the video processing unit 15 is also composed of a known circuit and is not a main part of the present invention, description of details of the circuit is omitted. The video data generated by the video processing unit 15 is output to the display unit 16, and an echo video is displayed on the display unit 16.

  In the embodiment described above, in the time range where only the weak echo signal is included in the signal received by the vibrator 1, the echo signal can be obtained with a good S / N ratio by using the output data of the amplifier having the largest amplification factor. Can be obtained. On the other hand, in the time range in which a high-power interference wave is superimposed on the weak echo signal received by the vibrator 1, the echo signal is obtained by using the output data of the amplifier having the highest amplification factor in a range that does not reach saturation. Only the interference wave can be removed by the digital programmable filter 9 without reducing the power of the signal.

  That is, regardless of the presence or absence of interference waves and the level of interference waves, the output of an amplifier having an optimum amplification factor that does not always cause saturation can be automatically selected by the data selection unit 8 and increases with time. Since the interference wave can be removed before the gain adjusting unit 13 that multiplies the coefficient to be detected, the interference wave can be removed without deteriorating the detection performance. Further, since an analog programmable filter is not used, there is no problem that the cost is significantly increased. Furthermore, since the analog multiplexer 4 and the amplitude / phase compensation unit 12 are provided, it is not necessary to provide an individual A / D converter for each of the plurality of amplifiers 31 to 33, so that the cost can be further suppressed.

  FIG. 9 is a block diagram of a scanning sonar according to another embodiment of the present invention. Here, only the reception system block is shown, and the transmission system block is omitted. 9 differs from FIG. 1 in the signal processing circuits 101, 102,... In FIG. 1 in that the amplitude / phase compensation unit 12 is provided in the subsequent stage of the digital programmable filter 9, whereas the signal in FIG. In the processing circuits 201, 202,..., The amplitude / phase compensation unit 12 is provided in the preceding stage of the digital programmable filter 9. As a result, in FIG. 9, the amplitude phase compensation unit 12 corrects the amplitude discontinuity and the phase discontinuity with respect to the output selected by the data selection unit 8, and the corrected output is converted into a digital programmable filter. 9, the interference wave is removed here, and only the frequency component in the predetermined band is extracted. Since other configurations are basically the same as those in FIG. 1, the same components as those in FIG. 1 are denoted by the same reference numerals.

  9, the A / D converter 5 performs time-division sampling in the manner described with reference to FIG. 2 and outputs real part data and imaginary part data of complex data with respect to the outputs of the amplifiers 31 to 33. The data selection unit 8 holds one piece of complex data corresponding to each of the amplifiers 31 to 33 in the buffer memory 7, and the upper limit value that the output of the A / D converter 5 can take from among the held complex data. Select complex data with the following absolute values: Then, the selected complex data having the largest absolute value is detected and the complex data is output. The digital programmable filter 9 calculates a filter coefficient according to the specified center frequency and pass bandwidth value, and performs a convolution operation on the complex data output from the amplitude phase compensation unit 12 using the filter coefficient. Then, the calculation result is output. The output of the filter 9 is given to the gain adjusting unit 13, and thereafter the same processing as in the case of FIG. 1 is performed.

  In the case of FIG. 1, in order to correct the amplitude and the phase after performing the convolution operation on the output of the data selection unit 8 by the digital programmable filter 9, the buffer memory 7 is M + 1 for each amplifier (FIG. It is necessary to hold complex data (4 in the example of 3), and the selection control unit 6 also needs to extract the maximum one from the M + 1 complex data (FIG. 4). On the other hand, in the case of FIG. 9, since the amplitude and phase are directly corrected for the output of the data selection unit 8, as shown in FIG. 10, the buffer memory 7 has one buffer memory for each amplifier. It is only necessary to hold complex data. This eliminates the need for the selection control unit 6 to extract the maximum data from the complex data string for each amplifier. Therefore, the capacity of the buffer memory 7 and the processing amount of the selection control unit 6 can be reduced at the same time. In FIG. 9, since the amplitude phase compensation unit 12 is in the preceding stage of the digital programmable filter 9, if the level of the DC offset included in the outputs of the amplifiers 31 to 33 is large, the amplitude correction in the amplitude phase compensation unit 12 is performed. However, if the level of the DC offset is sufficiently smaller than the level of the echo signal, there is no problem in amplitude correction in the amplitude / phase compensation unit 12, and this embodiment does not affect FIG. The same effect as in the case of can be obtained.

  Even when the level of the DC offset included in the outputs of the amplifiers 31 to 33 is larger than the level of the echo signal, the DC offset can be removed by using the digital high-pass filter. FIG. 11 shows this embodiment, and is a block diagram illustrating a part of the signal processing circuit 201 of FIG. As shown in FIG. 11, in this embodiment, a digital high-pass filter 80 is provided between the A / D converter 5 and the data selection unit 8. The digital high-pass filter 80 acts on each of the real part data string and the imaginary part data string of the complex data string with respect to the outputs of the amplifiers 31 to 33, and removes the DC offset included in each data string. Further, like the digital programmable filter 9, the digital high-pass filter 80 is formed of a single IC group common to each transducer by processing the received signal from each transducer in a time-sharing manner. can do.

  In the above embodiment, the three amplifiers 31 to 33 are used as analog amplifiers. However, in the present invention, an arbitrary plurality of analog amplifiers can be used. As the number of analog amplifiers increases, the SN ratio of the echo signal increases. The average value is improved.

  Further, in the above embodiment, the gain adjusting unit 13 is disposed in the front stage of the beam forming unit 14, but the gain adjusting unit 13 may be disposed in the subsequent stage of the beam forming unit 14.

  In the above embodiment, the complex data is obtained by performing sampling once every 0.25 period of the transmission signal in the A / D converter 5, but instead of this, a known Hilbert converter or A quadrature detector can also be used in combination with the A / D converter 5.

In the above embodiment, the bandpass filter 2 is provided between the broadband vibrator 1 and the amplifiers 31 to 33. However, the filter 2 may be omitted . Et al is, in light of the purpose of the removal of the interference wave, gain adjustment section 13 is not essential in the present invention.

In the above embodiment, an example of applying the present invention to a scanning sonar, the present invention is underwater detection system other than scanning sonar, for example, Zona cross fan beam system as disclosed in Patent Document 2 before supra it can also be applied to the over, and so on.

It is a block diagram of the scanning sonar which is one Embodiment of this invention. It is a figure explaining time division sampling. It is a figure which shows the content of the buffer memory. It is a figure which shows the content of the buffer memory. It is a figure explaining the upper limit of the output of an A / D converter. It is the figure which displayed the complex data in which an absolute value becomes the maximum as a vector. 4 is a table showing the relationship between the maximum absolute value of complex data and the amplifier number for each case. It is an example of the arithmetic circuit which performs a convolution operation. It is a block diagram of the scanning sonar which concerns on other embodiment of this invention. It is a figure which shows the content of the buffer memory in other embodiment. It is a partial block diagram which shows the modification of FIG. It is a figure which shows the frequency characteristic of a narrow band vibrator. It is a figure which shows the frequency characteristic of a wideband vibrator. It is a block diagram of the receiving system circuit in the conventional broadband sonar. It is a block diagram which shows the other example of the receiving system circuit in the conventional broadband sonar. It is a figure explaining quantization noise.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Broadband vibrator 2 Analog band pass filter 31-33 Analog amplifier 4 Analog multiplexer 5 A / D converter 6 Selection control part 7 Buffer memory 8 Data selection part 9 Digital programmable filter 10 Correction coefficient memory 11 Multiplier 12 Amplitude and phase compensation unit 13 Gain adjustment unit 14 Beam forming unit 15 Video processing unit 16 Display unit 80 Digital high-pass filter 101, 102, 103, 10n Signal processing circuit 201, 202, 203, 20n Signal processing circuit

Claims (9)

In an underwater detection device that obtains underwater information by transmitting ultrasonic waves from a transducer equipped with a plurality of transducers into the water and analyzing echo signals received by the transducers,
A plurality of amplifiers are provided for each vibrator , amplifying the output of the vibrator, and amplifiers having different amplification factors,
A multiplexer that sequentially switches the outputs of the plurality of amplifiers;
A data selection unit that selects an output corresponding to an amplifier that is non-saturated and has the largest amplification factor from the output of the multiplexer;
An amplitude phase compensation unit that corrects an amplitude discontinuity and a phase discontinuity for the output selected by the data selection unit;
A beam forming unit that forms a received beam by beam forming processing based on an output of the amplitude phase compensation unit;
An underwater detection device characterized by comprising: The underwater detection device according to claim 1 ,
An A / D converter that generates complex data for the output of each amplifier based on the output of the multiplexer and outputs the complex data to the data selector;
An output selected by the data selection unit is input, and a digital programmable filter with a variable pass band that passes a frequency component of a predetermined band of the output, and
Further comprising
The underwater detection device, wherein the amplitude phase compensation unit corrects an amplitude discontinuity and a phase discontinuity by multiplying the output of the digital programmable filter by a complex correction coefficient . The underwater detection device according to claim 2,
The A / D converter outputs real part data and imaginary part data of complex data with respect to the output of each amplifier by sampling at a predetermined period,
The data selection unit has a buffer memory for holding a complex data string composed of M + 1 complex data corresponding to each amplifier, where M is the order of the digital programmable filter, and the memory includes Detects the complex data with the maximum absolute value among the complex data in each stored complex data string, and selects the complex data having the absolute value below the predetermined threshold from the detected maximum values, and then selects The complex data having the maximum absolute value is detected, and the complex data string corresponding to the amplifier having the maximum value is output.
The digital programmable filter calculates a filter coefficient according to a specified center frequency and pass bandwidth value, and performs a predetermined operation using the filter coefficient for the complex data string output from the data selection unit. And an underwater detection device that outputs the calculation result. The underwater detection device according to claim 3,
The underwater detection, wherein the data selection unit outputs a complex data sequence corresponding to an amplifier having the smallest amplification factor when all the absolute values of complex data detected from the complex data sequences are larger than the threshold value. apparatus. The underwater detection device according to claim 1 ,
An A / D converter that generates complex data for the output of each amplifier based on the output of the multiplexer and outputs the complex data to the data selector;
A digital programmable filter having a variable pass band, which receives an output of the amplitude phase compensation unit and passes a frequency component of a predetermined band of the output;
Further comprising
The underwater detection device, wherein the amplitude phase compensation unit corrects an amplitude discontinuity and a phase discontinuity by multiplying an output of the data selection unit by a complex correction coefficient . The underwater detection device according to claim 5,
The A / D converter outputs real part data and imaginary part data of complex data with respect to the output of each amplifier by sampling at a predetermined period,
The data selection unit has a buffer memory for holding one piece of complex data corresponding to each amplifier, and among the pieces of complex data held in the memory, complex data having an absolute value equal to or less than a predetermined threshold value. Select the data, find the one with the largest absolute value from the selected complex data, output the complex data,
The digital programmable filter calculates a filter coefficient according to a specified center frequency and pass bandwidth value, and performs a predetermined operation on the complex data output from the amplitude phase compensation unit using the filter coefficient. And an underwater detection device that outputs the calculation result. The underwater detection device according to claim 6,
The underwater detection device, wherein the data selection unit outputs complex data corresponding to an amplifier having the smallest amplification factor when the absolute values of the complex data are all greater than the threshold value. In the underwater detection apparatus in any one of Claim 5 thru | or 7,
An underwater detection apparatus, wherein a digital high-pass filter is provided between the A / D converter and the data selection unit. In the underwater detection device according to claim 2 or 5,
After transmitting the ultrasonic wave to the output of the amplitude phase compensation unit so that the amplitude of the echo signal from the same target becomes constant without depending on the distance between the transducer and the target An underwater detection apparatus, further comprising a gain adjustment unit that multiplies a coefficient that increases in accordance with the elapsed time.

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