JP4727210B2 - Positioning control device for moving body and laser processing device - Google Patents

Description

  The present invention relates to a positioning control device for a moving body that controls the position of the moving body while detecting the position of the moving body in order to cause the moving body to follow a specified target value, and to move by such a positioning control device for the moving body. The present invention relates to a laser processing apparatus that controls the position of a body.

  High-speed and high-accuracy positioning devices for moving bodies are widely used in industry such as precision machine tools, computer storage devices, and semiconductor manufacturing devices. For example, in a laser processing apparatus that performs drilling in the manufacturing process of a printed wiring board, a laser beam positioning control device for sequentially irradiating a plurality of processing positions of a workpiece with laser light is necessary, and high processing throughput is required. In order to realize high-precision machining, a galvanometer mirror control device is often used.

  The laser processing apparatus is usually a numerical control (NC) apparatus having a hierarchical control structure, and the galvano mirror control apparatus is included in the lowest hierarchy. In the control device of the upper layer (hereinafter referred to as “upper control”), the processing order is optimized to realize high throughput based on the CAM (Computer Aided Manufacturing) data of the printed wiring board, and the hole position coordinates are processed. Are described in the NC program in the order in which they are performed.

  Such NC programs are created in advance, and when machining starts, the host control converts the hole position coordinates in the program one after another and sends time-series angle command data to the galvanometer mirror control device. To do. In order to machine a hole into a perfect circle, it is necessary to irradiate the laser beam after the galvano mirror is stationary at the angle commanded by the angle command data. This is done synchronously inside the control.

  The main components of the galvanometer mirror control device are a galvanometer mirror that is a moving body, a rotary actuator that changes the position or angle of the galvanometer mirror within a range of about ± 15 ° at maximum, and a control circuit that feeds back the angle of the galvanometer mirror.

  As the rotary actuator, an electromagnetic actuator that generates a driving torque on the electromagnetic principle is often used, and the galvanometer mirror is fixed to the rotary shaft of the rotary actuator as a support. In addition to the galvanometer mirror, a sensor, a movable coil, or a movable magnet is attached to the rotating shaft.

  The rotation angle of the galvanometer mirror is detected by a sensor, and the angle detection data is sent to a feedback control circuit. The feedback control circuit is realized by analog control composed of an operational amplifier or digital control firmware combining a microprocessor and a program. One-time positioning of the galvanometer mirror is an angular stroke, and there are variations from the order of 0.01 ° to about 30 °, and the positioning time is from less than 1 ms to several ms.

  The galvanomirror control device receives one angle command data as a step input signal and performs one positioning operation. That is, the galvanometer mirror is rotated based on one received angle command data. When the galvanometer mirror starts rotating, integral compensation is performed to make the mirror angle coincide with the angle command data without error. In this compensation, the value obtained by subtracting the angle detection data from the angle command data, that is, the tracking error signal is time-integrated. Furthermore, in order for the feedback loop of the galvanomirror control device to operate stably, it is necessary to sufficiently increase the phase margin and gain margin of the one-round transfer function. For this reason, stabilization compensation or phase advance compensation by the angular velocity signal acts by differentiating the angle detection data or using a so-called state observer. These control methods are well known as the basis of feedback control theory (Non-Patent Document 1).

  In order to shorten the positioning time of the galvanometer mirror, a technique for widening the feedback loop is used. That is, in the above-described electromagnetic actuator, a galvanometer mirror, a sensor, or the like attached to the rotating shaft acts as an inertial load, so that shaft torsional vibration may occur at high speed operation. Usually, since a plurality of torsional vibration modes exist in a region of several kHz or more, the feedback loop is broadened by a vibration mode stabilization compensator. This torsional vibration stabilization compensator estimates and feeds back the torsional vibration mode state quantity (Patent Documents 1 and 2).

  Hereinafter, a technique based on Patent Document 2 will be described.

  FIG. 7 is a block diagram of a conventional galvanometer mirror control apparatus.

  This galvanometer mirror control device is realized by digital control firmware using a microprocessor (not shown), and includes an integral compensator 2, a proportional compensator 3, a speed observer 4, a torsional vibration stabilization compensator 5, and a loop gain 10. The processing related to is described in a part of the program executed by the microprocessor. The processing calculation is executed at every discrete time (hereinafter referred to as “discrete time”) having a constant sample period.

  One galvanometer mirror (not shown) is attached to the rotary shaft of the rotary actuator 1. The angle of the galvanometer mirror (that is, the rotation angle of the rotary shaft of the rotary actuator 1) is the control amount signal 11 of the control device. Further, a rotary encoder (not shown) built in the rotary actuator 1 outputs angle detection data 9 at each discrete time. One or more torsional vibration modes exist on the rotary shaft of the rotary actuator 1 that holds the galvanometer mirror or the rotary encoder. In the illustrated case, the primary torsional vibration mode is compensated.

  The current detection signal id output from the current control circuit 7 that controls the drive current of the rotary actuator 1 is input to the speed observer 4 and the torsional vibration stabilization compensator (torsional primary compensation circuit) 5.

  Next, the operation of the conventional galvanometer mirror control device will be described.

  FIG. 8 is a flowchart showing the operation of the conventional galvanometer mirror control apparatus.

  When the timer interruption occurs (start), the time-series angle command data 8 transmitted from the host control device is received and the angle detection data 9 is fetched (steps S10 and S20). Then, in order to set the value of the deviation signal in the steady state to 0, the deviation signal (following error signal) between the angle command 8 and the angle detection data 9 is calculated and integrated by the integral compensator 2 (steps S30 and S40). In order to maintain the stability of the servo mechanism, compensation values are calculated by the proportional compensator 3, the speed observer 4, and the torsional vibration stabilization compensator 5, respectively (steps S50, 60, and 70). The proportional compensator 3 multiplies the angle detection data 9 by a proportional gain. The speed observer 4 calculates an estimated value of the angular speed of the mirror. Further, the torsional vibration stabilization compensator 5 calculates a state estimation amount of angular displacement and angular velocity in the torsional vibration mode.

  Then, the output value obtained in steps S50 to S70 is subtracted from the output value in step S40, the result is multiplied by a coefficient (loop gain calculation) (step S80), and the result is passed through the D / A converter 6. An analog signal is input to the current control circuit 7 (step S90).

  The current control circuit 7 supplies a drive current corresponding to the input signal to the rotary actuator 1.

In addition, there is also a method of suppressing residual vibration by filtering the angle command data for the natural vibration of the motor. (Patent Document 3)
These methods are combined with the above-described integral compensation and phase lead compensation to form a feedback loop. As for the characteristics of the feedback loop, the time required for one positioning time (positioning time) satisfies the target specification, and the overshoot and residual torsional vibration included in the transient response (settling response) near the target angle are within the allowable range. Adjusted to fit.

  By the way, when positioning at high speed, a large current is required at the time of acceleration until reaching the target position and at a constant speed or at the time of deceleration from the accelerated state to the target position, the coil generates heat, the temperature rises, and the coil resistance value increases.

  Magnets with high coercive force generally have significant changes in characteristics due to temperature changes, and the torque constant changes greatly due to changes in environmental temperature and self-heating. For this reason, when the temperature of the magnet rises, the overshoot of the settling response increases and causes an increase in positioning time. Further, since the frequency of the torsional vibration mode changes due to the temperature rise, an error occurs with the frequency set by the torsional vibration stabilization compensator 5, resulting in residual torsional vibration, resulting in poor drilling position accuracy. Conceivable.

  Thus, a change in temperature due to self-heating or the like can occur in many precision positioning control devices. For example, in a hard disk device, a voltage between voice coil motor (VCM) terminals during positioning (following) on a desired track. And VCM current is measured by the voltage detection circuit between the VCM terminals and the VCM current detection circuit, and is taken into the drive control unit via an analog-digital (AD) converter and the coil resistance value is calculated. Estimate and correct the loop gain (Patent Document 4), or detect the current flowing through the VCM by applying a voltage to the VCM for a certain period of time and calculate the temperature inside the VCM and the actuator detected by the temperature sensor Taking into account the thermal gradient from the VCM to the temperature detection means using the nearby ambient temperature Calculating the temperature of the sexual body, when the magnetic circuit characteristic due to temperature change is changed to modify by changing the gain of the transfer characteristic of the electrical system (Patent Document 5).

In addition, in an electric power steering system mounted on a vehicle, a temperature is estimated by a temperature sensor and a motor current, and both an error of a motor circuit resistance and an error of a motor constant are accumulated and corrected from the estimated temperature. Some are provided with one or more conditional integrators (Patent Document 6).
JP 2002-40357 A JP 2002-40358 A JP 2003-43404 A JP 2002-367307 A Japanese Patent No. 335827715 JP 2003-116291 A Toru Katayama, “Basics of Feedback Control”, Asakura Shoten, May 20, 1987, Chapters 6-7

  When the torsional vibration frequency hardly changes, the torsional vibration of the galvanometer mirror can be suppressed by the techniques of Patent Documents 1 to 3.

  However, when the temperature of the actuator changes due to self-heating or the like, the torsional vibration frequency also changes accordingly. However, Patent Documents 1 to 3 show means for suppressing torsional vibration when the torsional vibration frequency changes. It has not been.

  Further, in order to improve the throughput of the laser processing apparatus, it is necessary to irradiate the laser beam immediately after the galvanometer mirror enters the allowable range of the settling response amplitude and to move to the next positioning operation.

  However, Patent Documents 1 to 3 do not show a means for dealing with a change in the characteristic of the settling response accompanying a change in the temperature of the actuator. For this reason, the positioning accuracy is lowered, and the time interval of the step signal of the angle command pattern (hereinafter referred to as “command interval”) cannot be shortened.

  In the case of a hard disk device, the amount of displacement of the magnetic head with respect to the center of the target data track is the control amount, and the target value after switching the control mode to the high-accuracy positioning mode is always a constant value zero. The coil resistance value can be obtained by calculation.

 On the other hand, in the case of the galvanomirror control device, the angle command pattern that becomes the target value changes one after another as a step signal, so that it is difficult to use the above calculation method.

  SUMMARY OF THE INVENTION An object of the present invention is to solve the above-described problems of the prior art and to provide a moving body positioning apparatus capable of positioning a moving body at high speed and with high accuracy even when the temperature of the actuator changes, and such movement. It is in providing the laser processing apparatus which controls the position of a moving body by the body positioning control apparatus.

  The inventor of the present application believes that the overshoot or vibrational settling response of the galvano mirror that occurs when the galvano mirror is controlled by a conventional galvano mirror control device is a change in torque constant caused by a temperature change accompanying self-heating of the actuator. It was found that this is the influence of frequency change in torsional vibration mode.

Based on the above knowledge, the first means of the present invention is a positioning of a moving body comprising a current control circuit that controls the driving current of the actuator and a position sensor that detects the position of the moving body driven by the actuator. In the control device, the internal parameter estimating means for estimating the torque constant and the coil resistance inside the actuator, the estimating means for estimating the coil temperature from the coil resistance, and the correction of the torsional vibration frequency based on the coil temperature estimated by the estimating means a torsional vibration frequency estimation means for estimating an amount, based on the torque constant is estimated, and the loop gain correction means for correcting the loop gain so settling response is within the settling range, twisting previously set as servo parameters The torsional vibration frequency to which the correction amount of the estimated torsional vibration frequency is added to the vibration frequency A positive means, and correcting means for correcting the twisting torsional fixed in vibration frequency modifying means oscillation frequency and the loop gain and the drive current is corrected by the loop gain modifying means, comprising the. The positioning control device of the moving object, the coil temperature estimated by the estimating means is for indicating the internal temperature of the actuator comprising the coils, the coil temperature is previously stored in memory the coil resistance value at reference temperature The current coil temperature when the estimated coil resistance value estimated by the internal parameter estimating means is Rc, the temperature coefficient is α, the coil resistance value at the reference temperature is R0, and the coil temperature at the reference temperature is T0. It is preferable that T1 is obtained by the relational expression T1 = (Rc−R0) / αR0 + T0. In any one of the above-described positioning control devices for a moving body, it is preferable that the torque constant is estimated by an internal parameter estimation unit from a digital value signal obtained by converting a coil applied voltage signal and a digital value signal obtained by converting a drive current. . Further, in any one of the movable body positioning control devices described above, the movable body is preferably a galvanometer mirror.

According to a second aspect of the present invention, there is provided a laser processing apparatus comprising the moving body positioning apparatus and a laser output means for outputting a laser, wherein the laser output from the laser output means is positioned by a galvano mirror. Thus, the positioning control device controls the angle of the galvanometer mirror .

  According to the present invention, changes in torque constant and torsional vibration frequency due to temperature rise inside the actuator can be corrected immediately, so that positioning accuracy of the moving body can be improved.

  Hereinafter, a case where the present invention is applied to a galvanomirror control device will be described.

  FIG. 1 is a block diagram of a galvanomirror control device according to an embodiment of the present invention. Components having the same or the same functions as those in FIG.

  This galvanometer mirror control device is realized by digital control firmware using a microprocessor (not shown), and includes an integral compensator 2, a proportional compensator 3, a speed observer 4, a torsional vibration stabilization compensator 5, and a loop gain 10. The processing related to the internal parameter estimator 17 of the current control circuit, the temperature estimator 20 and the torsional vibration frequency temperature dependency table 12 is described in a part of the program executed by the microprocessor. And the said process is performed for every sporadic time.

  In the torsional vibration stabilization compensator 5, a reference torsional vibration frequency is set in advance as a servo parameter, and the torsional vibration frequency is set from the correction amount of the torsional vibration frequency output from the torsional vibration frequency temperature dependency table 12. Correct the value according to the current coil temperature. The torsional vibration stabilization compensator 5 is discretized in order to realize the techniques of Patent Documents 1 and 2 using a microprocessor.

  In the loop gain 10, a coefficient for amplifying the manipulated variable 21 is corrected according to the estimated torque constant 18 output from the internal parameter estimator 17 of the current control circuit.

  The AD converter 13 converts the coil application voltage signal Va in the current control circuit 7 into a digital value signal Vd.

  The AD converter 15 converts the current detection signal ia corresponding to the drive current output from the current control circuit 7 into a digital value signal id.

  The digital value signals Vd and id output from the AD converter 13 and the AD converter 15 are input to the internal parameter estimator 17 of the current control circuit.

  The internal parameter estimator 17 of the current control circuit outputs an estimated torque constant 18 and an estimated coil resistance value 19 which are internal parameters estimated for each sampling period.

  The temperature estimator 20 estimates the current coil internal temperature from the input estimated coil resistance value 19 and inputs the coil internal temperature to the torsional vibration frequency temperature dependency table 12.

  The current control circuit 7 will be further described.

  FIG. 2 is a circuit diagram of the current control circuit 7 in FIG.

  The value of the inductance of the coil 1c of the rotary actuator 1 surrounded by the dotted line in the figure is Lc, and the value of the resistance is Rc.

  A subtraction circuit including the resistor 51, the resistor 52, the resistor 78, and the operational amplifier 53 calculates a difference between the current command 39 and the current detection signal ia.

  The amplifier circuit composed of the resistor 54, the resistor 55, and the operational amplifier 56 multiplies the output of the operational amplifier 53 by a coefficient.

  A power amplifier output voltage 79 output from the first power amplifier including the resistor 57, the capacitor 58, the resistor 59, the capacitor 60, and the operational amplifier 61 is applied to one end of the coil 1c. The power amplifier output voltage 80 output from the second power amplifier including the resistor 64, the capacitor 65, the resistor 66, the capacitor 63, and the operational amplifier 62 is applied to the other end of the coil 1c. The voltage of the power amplifier output voltage 80 is opposite in sign to the power amplifier output voltage 79, and a coil application voltage signal Va obtained by subtracting the power amplifier output voltage 80 from the power amplifier output voltage 79 is applied to both ends of the coil 1c.

  The current sensor 69 detects the current flowing through the coil 1c.

  A current detection circuit including the resistor 70, the operational amplifier 71, the resistor 72, the resistor 73, the resistor 74, the capacitor 75, the resistor 76, and the operational amplifier 77 converts the current detected by the current sensor 69 into a voltage signal. The output of this current detection signal circuit is the current detection signal ia.

  FIG. 3 is a block diagram of FIG.

The mechanical characteristics are configured as secondary system characteristics.
The gain 90 is a gain obtained from the circuit resistance value from the subtraction of the current command 39 and the current detection signal ia to the coil application voltage signal Va. The transfer function 91 is a transfer characteristic from the coil applied voltage signal Va to the current detection signal ia for the series connection of the inductance Lc and the coil resistance Rc. Further, the torque constant Kt, the moment of inertia Jc, the gain 94, the gain 95, the integrator 96, and the integrator 97 are constituent elements of the secondary system characteristics. The output of the integrator 96 in the secondary system characteristic multiplied by the counter electromotive force coefficient Kb is added to the coil applied voltage signal Va as a counter electromotive force.

  The current detection gain Kid is a characteristic from the current flowing through the coil to the current detection signal ia.

The transfer function from the coil applied voltage signal Va to the current detection signal ia in FIG. 2 is expressed by Equation 1, where Jc is the moment of inertia, ω is the breakpoint frequency of the rotary actuator, and ζ is the damping coefficient.

  From Expression 1, it can be seen that the transfer function from the coil applied voltage signal Va to the current detection signal ia is expressed by a third order.

  In this embodiment, the coil internal temperature is estimated by estimating the coefficient of the transfer function of Equation 1.

  Next, the procedure in which the internal parameter estimator 17 of the current control circuit estimates the coefficient of the transfer function expressed by Equation 1 will be described.

  In order to estimate the internal parameters of the current control circuit constituting Equation 1 with a microprocessor, it is necessary to convert from a continuous time system to a discrete time system.

  As a means for converting from a continuous time system to a discrete time system, there is a bilinear transformation or the like. However, the higher the order, the more difficult the coefficient correspondence with the continuous time system becomes.

  Therefore, in this embodiment, a third-order ARX (auto-regressive exogenous) model using a delta operator that receives the coil application voltage signal Vd and outputs the current detection signal id is employed as the current control circuit model.

  The delta operator operator δ is one of conversion means for converting a continuous-time system into a discrete-time system, and has an advantage that consistency with the Laplace operator s becomes higher as the sampling frequency Ts is shorter than bilinear transformation or the like. Yes, each coefficient of the transfer function of Equation 1 and the identified coefficient correspond to each other.

Here, the definition of the delta operator will be described. If the sampling period is Ts, the symbol δ representing the delta operator is expressed by Equation 2.

Since z in Equation 2 represents the forward difference operator with respect to the sampling period Ts in the discrete time system model, the meaning of the operator δ is the same as performing the differential operation represented by Equation 3.

However,
i: Index representing discrete time f (i): a certain function f ′ (i): a derivative of a certain function f (i).

The transfer function when the single input / output continuous time system is discretized with the sampling period Ts using this delta operator is expressed by Expression 4 using Expression 5 and Expression 6.

However,
uν (i): Discrete time coil applied voltage signal yi (i): Discrete time current detection signal ε (i): Error including observation noise at discrete time.

Here, when θ and φ (i) are expressed by the right side of Expression 7 and Expression 8, Expression 4 is expressed by Expression 9.

  Parameter identification means obtaining θ in Equation 7 which is each coefficient vector of the denominator and numerator equations.

Therefore, if φ (i) can be obtained, each coefficient of the transfer function can be calculated from Expression 9. However, the approximate derivative that is a component of φ (i) is easily affected by noise. Therefore, when Expression 10 is introduced as a stable third-order filter of the same order as A (δ) and Expression 4 is modified, Expression 11 is obtained.

Here, when θf and φf (i) are defined on the right side of Equation 12 and Equation 13 using Equation 14 to Equation 18, Equation 19 holds.

The element of Expression 15 can be generated by a three-dimensional state space model expressed by Expression 20, and can be expressed by Expression 21 using Expression 23 and Expression 24. Similarly, φu (i) can be expressed by Equation 22.

  Here, the evaluation function J is defined by Expression 25.

  Then, identification processing is performed by sequentially calculating a parameter that minimizes the error using the successive least square method (RLS method).

The specific calculation process is the same as that of the general weighted sequential least square method, and detailed expression expansion from Expression 25 is omitted, but P (i) is a covariance matrix and λ is a forgetting coefficient. The time update formula (i = 1, 2,..., N) shown in Formula 27 is calculated.

  Therefore, θf can be obtained from the coil application voltage signal that is an input signal and the current detection signal that is an output signal by Expression 20, Expression 25, and Expression 26.

  As described above, since a2, a1, a0, b2, b1, and b0, which are components of θf, correspond to the coefficients of the transfer function of Equation 1, the estimated coil resistance value Rc and the estimated torque constant 18 are expressed by the equation It is calculated by Equation 28 and Equation 29 using 1 and θf.

  Considering SI units, the back electromotive force coefficient and the estimated torque constant 18 are the same (Kt = Kb). Therefore, if the square root of the right side of Equation 28 is taken, the estimated torque constant 18 is obtained.

As described above, the internal parameter estimator 17 of the current control circuit outputs the estimated torque constant 18 and the estimated coil resistance value Rc obtained by Expressions 28 and 29.

  Next, the temperature estimator 20 will be described.

Assuming that the coil temperature at the reference temperature is T0, the coil resistance value at the reference temperature is R0, and the temperature coefficient is α, the current coil temperature T1 can be obtained by Equation 30, so the coil resistance value R0 at the reference temperature Is previously stored in the memory.

  When estimating the temperature of the magnet instead of the current coil temperature T1, it is possible to estimate using the estimated torque constant 18 by examining changes in the torque constant Kt due to temperature in advance.

  Next, a servo parameter correction method will be described.

  In this embodiment, the first-order mode torsional vibration frequency Ω1 and the torque constant Kt, which are servo parameters, are corrected.

(1) Correction of the first-order mode torsional vibration frequency Ω1 Here, the relationship between the coil temperature and the first-order mode torsional vibration frequency Ω1 is examined in advance, and the current coil temperature that is the output of the temperature estimator 20 is checked. Based on the above, the torsional vibration frequency Ω1 in the first mode is calculated.

  The relationship between the coil temperature and the first-order mode torsional vibration frequency Ω1 is obtained as follows.

  That is, a constant temperature and humidity chamber is used to keep the ambient temperature constant, the drive signal is applied with the coil temperature of the rotary actuator 1 being the same as the ambient temperature, and the torsional vibration frequency Ω1 of the primary mode at that time is measured. To do. This is repeatedly measured at a certain temperature interval (for example, every 10 degrees).

  The relationship between the torsional vibration frequency Ω1 and the temperature may be stored in a memory as a table, but here, the measured data is linearly approximated and stored in the form of Equation 31. In Equation 31, ΔΩ1 is the correction amount of the torsional vibration frequency Ω1, and α1 is the temperature coefficient of the torsional vibration primary mode.

Then, the correction amount ΔΩ1 of the torsional vibration frequency obtained by the equation 31 is added to the torsional vibration frequency Ω1 that is a servo parameter set in the torsional vibration stabilization compensator 5, and an error due to a temperature change is reduced.

Expression 32 is a discrete state equation of the torsional vibration stabilization compensator 5, and expression 36 is an output equation of the torsional vibration stabilization compensator 5. The elements of Expression 32 and Expression 36 are as shown in Expression 33 to Expression 35, and the coefficients of Expression 32 are as shown in Expression 37 to Expression 48.

However,
u (i): manipulated variable 21
y (i): output θ1 (i) of torsional vibration stabilization compensator 5: state estimation amount [rad] of angular displacement of torsional vibration primary mode
ω1 (t): torsional vibration first-order angular velocity state estimation amount [rad / s]
Ad: Discretized matrix Bd1: State equation X (i) Bd1: Discrete matrix Bd2: State equation u (i-1) Discrete matrix Cd: State equation u (i) Discrete matrix Cd: Output Discretized matrix exp related to X (i) of the equation: natural logarithm ζ1: damping coefficient of torsional vibration primary mode Ω1: natural angular frequency of the torsional vibration primary mode [rad / s]
L: u (i) time delay [s]
k1: Mode constant of torsional vibration primary mode Note that when controlling torsional vibration modes of other orders, the servo parameters relating to the order can be corrected in the same manner as described above.

  In addition, the servo parameters can be corrected by measuring in advance fluctuations due to temperature with respect to the damping coefficient and mode constant of the torsional vibration mode.

  Further, in this embodiment, the torsional vibration stabilization compensator 5 in the feedback loop has been described. However, a feed is provided in which a residual vibration is reduced by providing a filter and performing a filtering process on the angle command. The present invention can also be applied to forward compensation (in this case, a parameter related to residual vibration of the filter is corrected by estimating fluctuation due to temperature).

(2) Correction of Torque Constant Kt For the change of the torque constant Kt due to the temperature change, the loop gain 10 is increased or decreased using the estimated torque constant 18 so that the settling response satisfies the setting range specification.

  Specifically, the denominator of the loop gain is the estimated torque constant 18, and the numerator is the nominal value of the torque constant Kt. In this way, for example, when the estimated torque constant 18 becomes smaller than the nominal value of the torque constant Kt, the loop gain 10 increases by the ratio between the two.

  Next, the operation of this embodiment will be described.

  FIG. 4 is a flowchart showing a control procedure in the present invention.

  Since steps S10 to 90 are the same as the conventional procedure described with reference to FIG.

  When the current command 39 is output from the DA converter 6 in step S90, the coil application voltage signal Va and the current detection signal ia at this time are taken out from the current control circuit 7, and digitally converted by the AD converter 13 and the AD converter 15. It is converted into a value signal (procedure S200). The internal parameter estimator 17 of the current control circuit calculates φy (i) and φu (i) from Equation 21 and Equation 22 using the input coil applied voltage signal Vd and the current detection signal id (step S210). . In addition, P (i), which is a covariance matrix, is calculated by Expression 26 (step S220). Then, φf (i) is calculated from the obtained P (i), φf (i) and Equation 27 (step S230). By calculating θf (i), each coefficient of the transfer function of Equation 1 is obtained.

  Next, an estimated coil resistance value 19 is obtained from Equation 28 using a0 and b0 that are components of θf (i) obtained above (step S240). Further, the estimated torque constant 18 is calculated based on Expression 29 using a1, a0, b2, b1, b0 that are components of θf (i) (step S250).

  Next, the current coil temperature is calculated by Equation 30 using the estimated coil resistance value 19 (procedure S260), and the correction amount ΔΩ1 of the torsional vibration frequency Ω1 is calculated by Equation 31 (procedure S270).

  Then, using the correction amount ΔΩ1 of the torsional vibration frequency, Ω1 in Equations 38 to 45 is corrected to ΔΩ1 + Ω1 (step S280).

  Then, the loop gain 10 is corrected using the estimated torque constant 18 obtained above (step S290).

  The corrected torsional vibration frequency Ω1 and the loop gain 10 are reflected in the servo processing of the next sampling period.

  FIG. 5 is a diagram showing a positioning response curve of the moving body according to the present invention, where the vertical axis represents position deviation and the horizontal axis represents time. The thick line in the figure is the case of the present invention, and the conventional case is shown by a broken line for comparison. In the figure, the alternate long and short dash line indicates the allowable setting range (hereinafter referred to as “setting range”).

  As shown in the figure, when the present invention is applied, when the target position is reached, the moving body is quickly positioned within the settling range, and a good settling response is realized with little residual vibration.

  On the other hand, in the case of the prior art, the overshoot increases as the temperature rises, and the positioning time is delayed. Further, residual vibration is caused by a change in the torsional vibration frequency.

  As described above, the galvanometer mirror is desired without being affected by temperature changes such as self-heating by correcting the torsional vibration frequency of the torsional vibration stabilization compensator 5 and correcting the loop gain 10 from the parameter identification of the present invention. Can be positioned with high accuracy.

  Next, the case where the present invention is applied to a laser processing apparatus for processing a printed wiring board will be described.

  FIG. 6 is a diagram showing an outline of a laser processing apparatus to which the present invention is applied.

  The laser light source 301 oscillates a laser and emits laser light. Laser light irradiation is controlled by a command from the host control.

  The galvano scanner 302 positions the galvanometer mirror at an angle commanded by the angle command data in order to irradiate the laser beam emitted from the laser light source 301 to the processing position.

  The galvano scanner 303 positions the galvanometer mirror at an angle commanded by the angle command data in order to irradiate the laser beam reflected by the galvanometer scanner 302 to the processing position.

  Here, the control devices of the galvano scanners 302 and 303 correct the torsional vibration frequency and the loop gain with the configuration shown in FIG.

  The fθ lens 304 condenses the laser light reflected by the galvano scanner 303 and irradiates the processing position of the printed wiring board 306.

  The XY table 305 moves while holding the printed wiring board 306. When the processing within the angular range in which the galvano scanner 302 and the galvano scanner 303 can be operated ends, the galvano scanner 302 moves to the next processing position.

  By applying the present invention to the laser processing apparatus, the galvano scanner 302 and the galvano scanner 303 can suppress overshoot and residual vibration due to temperature rise, and the positioning time and positioning accuracy can be improved.

  Therefore, since the galvanometer mirror that reflects the laser light is positioned at the processing position at high speed and with high accuracy, the processing throughput and processing accuracy of the laser processing apparatus can be improved.

  By the way, although it is considered that the breakpoint frequency ω and the damping coefficient ζ of the rotary actuator are caused by friction of the bearing of the rotary actuator, using the internal parameter estimator 17 of the current control circuit according to the present invention, In addition to the estimated coil resistance value 19 and the estimated torque constant 18, the coil inductance Lc, the break-point frequency ω of the rotary actuator, or the damping coefficient ζ can be obtained in the same manner using equations 46 to 48.

  Therefore, a disturbance observer can be constructed using the rotary actuator breakpoint frequency ω and the damping coefficient ζ estimated by the internal parameter estimator 17 of the current control circuit. If it does in this way, torque disturbance resulting from friction can be canceled and high-speed and highly accurate positioning can be performed.

It is a block diagram of the galvanometer mirror control device concerning the present invention. FIG. 3 is a circuit diagram of a current control circuit according to the present invention. FIG. 2 is a block diagram. It is a flowchart which shows the control procedure in this invention. It is a figure which shows the effect of this invention. It is a figure which shows the outline of the laser processing apparatus to which this invention is applied. It is a block diagram of the conventional galvanometer mirror control apparatus. It is a flowchart which shows operation | movement of the conventional galvanometer mirror control apparatus.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Rotary actuator 1c Coil 7 Current control circuit 10 Loop gain 17 Internal parameter estimator of current control circuit 18 Estimated torque constant 19 Estimated coil resistance value id Current detection signal Vd Coil applied voltage signal 20 Temperature estimator Ω1 Torsional vibration frequency ΔΩ1 Torsion Correction amount of vibration frequency

Claims (5)

In a movable body positioning control device comprising: a current control circuit that controls a drive current of the actuator; and a position sensor that detects a position of the movable body driven by the actuator.
An internal parameter estimating means for estimating a torque constant and a coil resistance inside the actuator;
And estimating means for estimating the coil resistance or Rako yl temperature,
A torsional vibration frequency estimating means for estimating a correction amount of the torsional vibration frequency based on the coil temperature estimated by the estimating means;
Based on the torque constant that is estimated, and the loop gain correction means for correcting the loop gain so settling response is within the settling range,
Advance servo parameters to torsional vibration frequency is set as the added amount of correction estimated by the torsional vibration frequency torsional vibration frequency correcting means,
Correction means for correcting the drive current by the torsional vibration frequency corrected by the torsional vibration frequency correction means and the loop gain corrected by the loop gain correction means;
A positioning control apparatus for a moving body, comprising: Wherein the coil temperature estimated by the estimating means is for indicating the internal temperature of the actuator comprising the coils, the coil temperature is previously stored in a memory coil resistance value at the reference temperature, the internal parameter As the current coil temperature T1 where Rc is the estimated coil resistance value estimated by the estimating means, α is the temperature coefficient, R0 is the coil resistance value at the reference temperature, and T0 is the coil temperature at the reference temperature, T1 = ( 2. The moving body positioning control device according to claim 1, wherein the positioning control device is obtained by a relational expression of Rc−R0) / αR0 + T0.   3. The moving body according to claim 1, wherein the torque constant is estimated by the internal parameter estimating means from a digital value signal obtained by converting a coil applied voltage signal and a digital value signal obtained by converting the drive current. Positioning control device.   The said moving body is a galvanometer mirror, The positioning device of the moving body in any one of Claims 1-3 characterized by the above-mentioned.   5. A laser processing apparatus comprising: a moving body positioning apparatus according to claim 4; and laser output means for outputting a laser, wherein the laser output from the laser output means is positioned by the galvanometer mirror, wherein the positioning is performed. The control device controls the angle of the galvanometer mirror.

Images