JP4719102B2 - Propagation path estimation device and path diversity reception device - Google Patents

Description

  The present invention relates to a propagation path estimation device and a path diversity reception device, and more particularly to a propagation path estimation device that estimates a propagation path of a radio wave including a plurality of pilot carriers in a signal band and performs path diversity reception using the estimated propagation path. And a path diversity receiver.

  In recent years, in addition to digital terrestrial broadcasting, OFDM (Orthogonal Frequency Division Multiplexing) signal formats have been adopted for program material transmission apparatuses (Field Pick-up Units) for mobile relay, and for marathon relay and wireless Used in cameras. An OFDM signal has a certain degree of resistance to frequency selective fading by arranging a plurality of subcarriers orthogonal to each other on the frequency axis.

  On the other hand, techniques such as spatial diversity reception, adaptive antennas (see, for example, Non-Patent Document 1), interference wave cancellers, and the like have been developed for the purpose of reducing the influence of interference waves such as delayed waves that cause frequency selective fading. . When developing these receivers, in order to grasp or use the characteristics of the propagation environment to be used, the delay profile, the DU ratio that is the ratio of the signal power of the desired wave and the signal power of the delayed wave, the direction of arrival, etc. It is necessary to measure representative propagation parameters.

  A propagation analysis apparatus that measures a delay profile of an OFDM signal is generally obtained by dividing a pilot carrier extracted from a received signal by a known reference pilot carrier and performing an IFFT calculation, as described in Patent Document 1, for example. Various methods based on this method have been proposed.

  FIG. 1 shows a general block diagram of a propagation analyzer for obtaining a delay profile. In the figure, the received OFDM signal is mixed with the local signal from the local signal generator 2 by the mixer 1, and the intermediate frequency signal filtered through the low-pass filter (LPF) is converted into a digital signal by the AD converter 3. Symbol synchronization is established by performing orthogonal demodulation in the orthogonal demodulation unit 4 and performing guard interval correlation in the guard interval correlation calculation unit 5. Thereafter, after the effective symbol is cut out by the window position control unit 6, an FFT (Fast Fourier Transform) calculation unit 7 performs an FFT calculation process to convert the signal into a frequency domain signal.

  The CP extraction unit 8 extracts a pilot carrier inserted for equalization from the signal converted into the frequency domain, and the complex division unit 9 performs complex division on the pilot carrier by the reference pilot carrier output from the reference CP generation unit 10. The delay profile can be obtained by performing IFFT calculation on the propagation path characteristics (amplitude and phase fluctuations received in the propagation path) obtained by the delay profile measurement unit 11.

  Further, the linear interpolation unit 12 estimates the channel characteristics of the data carrier between the pilot carriers by linear interpolation from the channel characteristics of the pilot carriers on both sides of the data carrier, and supplies them to the demodulation unit 13. The demodulator 13 demodulates the data carrier supplied from the FFT calculator 7. The carrier signal-to-noise ratio measurement unit 14 measures the ratio between the power of the noise floor other than the carrier and the power of each carrier.

  By the way, in order to obtain an accurate delay profile, generally, precise symbol synchronization and average processing over a plurality of symbols are required. Therefore, a delay profile measurement method combined with synchronization means as disclosed in Patent Document 2 is proposed. Has been.

  For estimation of the direction of arrival of radio waves, beamformer methods, linear prediction methods, MUSIC (Multiple Signal Classification) and ESPRIT (Estimation of Signal Parameters via Rotational Innovation Techniques) are methods that do not use a reference signal such as a pilot signal. Proposed. In addition, there is a method for accurately obtaining only the desired wave direction using a guard interval period in which the same waveform as the predetermined length section at the end of the effective symbol section of the OFDM signal is used (see Patent Document 3).

  As a method using a reference signal, for example, as described in Patent Document 4, a pilot carrier is extracted from each symbol of an OFDM signal received by two antennas, and the influence of a delayed wave or the like is suppressed, thereby reducing the distance between antennas. Some of them detect the phase difference and estimate only the direction of arrival of the desired wave having the maximum received power with high accuracy.

On the other hand, as a technique for efficiently receiving each path component, there is a CDMA (Code Division Multiple Access) type path diversity reception technique used in a mobile communication system such as a mobile phone. This is a communication technology based on the spread spectrum technology. A path search is performed using a spread code, the timing of each path component having a different delay time is matched, and then synthesized in the same phase in the time domain. To do. In the CDMA system, the narrowband signal itself is spread with a unique pseudo-random code. Therefore, by detecting and synthesizing the path timing, the power ratio of the desired wave to interference and thermal noise is improved, and the reception characteristics are improved. it can.
JP 2002-335226 A JP 2000-134176 A JP 2005-348278 A JP 2005-140768 A Akizumi Kimura Shibuya “Examination of OFDM Adaptive Array for Digital Terrestrial Broadcasting” Video Media Society Technical Report, Vol. 27, no. 17, p25-28 "Portable OFDM digital wireless transmission system for transmitting television broadcast program material" ARIB STD-B33 "Transmission system for digital terrestrial television broadcasting" ARIB STD-B31

  In the method for measuring a delay profile of an OFDM signal described in Patent Document 1, even when only highly accurate delay time information of one path component is obtained, an IFFT operation is performed after dividing a pilot carrier extracted for each symbol by a reference pilot. Thus, it is necessary to obtain and extract the delay profile in the entire measurable time range, and there is a problem that the calculation load increases when the number of symbols to be averaged increases in order to improve accuracy.

  When the received power fluctuates, the absolute level of the delay profile also fluctuates.Therefore, in order to distinguish the effective path component from the side lobe component and extract only the effective path component, it is necessary to standardize the level separately and set the threshold value. There was a problem that setting was troublesome. Further, there is a problem that the propagation parameters such as the observed desired wave and the arrival direction of each delayed wave cannot be simultaneously and accurately obtained only by the conventional delay profile measuring method.

  With regard to direction estimation of incoming waves, methods such as MUSIC and ESPRIT that do not use a reference signal have low correlation with each other such as different signal sources, and the arrival directions of incoming waves within (number of antennas −1) should be determined with high accuracy. Although it is possible, it is generally difficult to accurately estimate the arrival direction of a coherent incoming wave having a high correlation such as a reflected wave or an incoming wave exceeding (the number of antennas −1) with a small number of receiving systems.

  On the other hand, as described in Patent Document 4, there is a method for accurately estimating only the arrival direction of a desired wave having the maximum received power using a pilot carrier that is a reference signal. Path component propagation parameters (delay time of each delayed wave, correlation phase, DU ratio, direction of arrival) could not be obtained simultaneously.

  On the other hand, as a technique for improving the reception characteristics of the OFDM signal, a technique for reducing the influence of multiple paths using spatial diversity reception, and each path component other than the desired wave is regarded as an interference wave and canceled by equalization. Although a technology has been proposed, it is difficult to implement a technology that actively uses and improves reception characteristics, such as separating and recombining each path component, as in CDMA path diversity reception. There was a problem.

  The present invention has been made in view of the above points, and it is an object of the present invention to provide a propagation path estimation device and a path diversity reception device that can simultaneously obtain a propagation parameter for each propagation path with a small amount of computation and with high accuracy. To do.

A propagation path estimation apparatus of the present invention is a propagation path estimation apparatus that receives a radio wave including a pilot carrier and estimates a propagation path parameter.
Extracting means for extracting a pilot carrier from a signal in a frequency domain obtained by converting a signal received by an antenna;
Phase rotation means for sequentially applying a predetermined amount of phase rotation to a reference pilot carrier prepared in advance or a reception pilot carrier extracted by the extraction means;
Correlation calculating means for obtaining a cross-correlation value by performing a correlation calculation of the reference pilot carrier or the received pilot carrier whose phase is rotated by the phase rotating means and the received pilot carrier or the reference pilot carrier which is not phase-rotated,
A delay profile is generated from the cross-correlation value to obtain a cross-correlation value for the desired wave and the delayed wave, and from the cross-correlation value, a signal delay time, a correlation phase, and a ratio of the desired wave and the delayed wave for each propagation path Propagation parameter calculation means for calculating a certain DU ratio ;
Averaging means for averaging the cross-correlation values obtained by the correlation calculating means over a plurality of symbols and supplying the averaged values to the propagation parameter calculating means;
Have
The propagation parameter calculation means roughly specifies a peak point of an absolute value of a cross-correlation value exceeding a preset threshold value from a cross-correlation value of the first one symbol or several symbols as a propagation path, and cross-correlates before and after the peak point. By specifying the peak point of the absolute value of the cross-correlation value obtained by averaging the correlation values over a plurality of symbols as the propagation path, it is possible to obtain the propagation parameter for each propagation path simultaneously and with high accuracy with a small amount of calculation.

Further, the path diversity receiving device of the present invention, a plurality of extraction means for extracting a pilot carrier from a signal in a frequency domain obtained by converting a signal received by a plurality of antennas,
Phase rotation means for sequentially applying a predetermined amount of phase rotation to a reference pilot carrier prepared in advance;
A plurality of correlation calculation means for performing a correlation calculation of the reference pilot carrier phase-rotated by the phase rotation means and the received pilot carrier obtained by the plurality of extraction means to obtain cross-correlation values of a plurality of systems;
A delay profile is generated from the cross-correlation values of the plurality of systems to obtain a propagation path of a desired wave and a delay wave, and a signal delay time for each propagation path is determined from the cross-correlation value of the one propagation path of the plurality of systems. Calculating a correlation phase, a DU ratio that is a ratio of the desired wave and the delayed wave, and calculating a correlation phase of a signal for each propagation path from cross-correlation values of the propagation paths of a plurality of systems other than the one system Propagation parameter calculation means;
A propagation path estimation device having arrival direction detection means for detecting the arrival direction of a signal for each propagation path from a plurality of correlation phases of signals for each propagation path obtained by the plurality of propagation parameter calculation means ;
Phase rotation means for rotating the phase of each signal received by the plurality of antennas based on the direction of arrival of the signal for each propagation path;
First combining means for separating and combining the plurality of phase-rotated signals for each propagation path;
The reception characteristics can be improved by having second combining means for combining the combined signal separated for each propagation path in the same phase based on the delay time and the correlation phase of the signal for each propagation path.

In the path diversity receiver,
The second combining means can combine the desired wave and the delayed wave of the propagation path by weighting based on the DU ratio.

Further, the path diversity receiving device of the present invention, a plurality of extraction means for extracting a pilot carrier from a signal in a frequency domain obtained by converting a signal received by a plurality of antennas,
Phase rotation means for sequentially applying a predetermined amount of phase rotation to a reference pilot carrier prepared in advance;
A plurality of correlation calculation means for performing a correlation calculation of the reference pilot carrier phase-rotated by the phase rotation means and the received pilot carrier obtained by the plurality of extraction means to obtain cross-correlation values of a plurality of systems;
A delay profile is generated from the cross-correlation values of the plurality of systems to obtain a propagation path of a desired wave and a delay wave, and a signal delay time for each propagation path is determined from the cross-correlation value of the one propagation path of the plurality of systems. Calculating a correlation phase, a DU ratio that is a ratio of the desired wave and the delayed wave, and calculating a correlation phase of a signal for each propagation path from cross-correlation values of the propagation paths of a plurality of systems other than the one system Propagation parameter calculation means;
Arrival direction detection means for detecting the arrival direction of a signal for each propagation path from a plurality of correlation phases of signals for each propagation path obtained by the plurality of propagation parameter calculation means;
A propagation path estimation device having
Weight vector estimation means for generating weight vectors for separating signals of a plurality of propagation paths from the cross-correlation values for each of the plurality of antennas obtained by the propagation path estimation device;
Signal separation means for separating the signals of the plurality of propagation paths by multiplying the signals received by the plurality of antennas by the weight vector and combining the signals;
It is possible to have path synthesizing means for equalizing each signal separated for each propagation path, or synthesizing after making the same phase based on the delay time and correlation phase of the signal for each propagation path.

A path diversity receiver according to the present invention includes the propagation path estimator according to any one of claims 1 to 3,
A propagation path for separating the signal for each propagation path in the frequency domain using the delay time and correlation phase of the signal for each propagation path obtained by the propagation parameter calculation means from the frequency domain signal obtained by converting the signal received by the antenna Separating means;
By having combining means for combining the signals for each of the propagation paths separated by the propagation path separation means in the same phase, it is possible to improve reception characteristics.

In the path diversity receiver,
The combining means can combine the desired wave and the delayed wave of the propagation path by weighting based on the DU ratio.

  According to the present invention, it is possible to obtain a propagation parameter for each propagation path at the same time with a small amount of computation and with high accuracy, and it is possible to improve reception characteristics by receiving path diversity.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

  In the present embodiment, the transmission / reception number is assumed to be an OFDM signal in a format defined in Non-Patent Document 2. As shown in FIG. 2, this type of OFDM signal has a pilot carrier (CP: Continuous Pilot) as a reference for channel equalization at every eight carriers in the frequency direction and always in the same position in the symbol direction. 1) Inserted after BPSK modulation by a value of a pseudo-random sequence (PN sequence) using the equation as a generator polynomial.

g (x) = x ¹¹ + x ² +1 (1)
Among the propagation parameters obtained by the present invention, the delay time of the signal that has passed through each path, the correlation phase, and the DU ratio that is the ratio of the signal power of the desired wave and the signal power of the delayed wave can be estimated with one system of received signal. is there. Here, the correlation phase indicates the phase of the cross-correlation peak value between the reference pilot carrier and the reception pilot carrier, as will be described later.

  On the other hand, in order to estimate the direction of arrival of the signal that has passed through each path, for example, the phase difference from the antenna located at a position half a wavelength away from the carrier wave is used, so at least two received signals are required. .

<Detection of delay time, correlation phase, DU ratio>
First, a method for obtaining the delay time, the correlation phase, and the DU ratio will be described.

  FIG. 3 shows a block diagram of an embodiment of the propagation path estimation apparatus of the present invention. In the figure, the signal received by the antenna 20 is down-converted to a frequency capable of digital signal processing by the frequency converter 21, converted to a digital signal by the AD converter 22, demodulated by the orthogonal demodulator 23, and converted into an IQ signal. To separate. Here, the quadrature demodulation is performed by digital signal processing after AD conversion, but it may be converted into a digital signal by AD conversion after being separated into IQ signals by quadrature demodulation.

  Next, symbol synchronization is established by the symbol synchronization unit 24 to establish symbol synchronization, and one effective symbol of the desired wave is cut out. The FFT calculation unit 25 performs FFT calculation and supplies it to the demodulation unit 26. It supplies to CP extraction part 27, and extracts P pilot carriers (CP) in total. The CP extracted in this way is hereinafter referred to as a reception pilot carrier.

On the other hand, for each of all P reference pilot carriers generated by the reference CP generation unit 30 on the receiving side, the phase rotation unit 31 performs complex multiplication by the values Z _{m and p} shown in the equation (2) to perform phase rotation. Are stored in the memory 32 in advance as m-th set of phase rotation reference pilot carriers.

Z _{m, p} = exp (j · 2πp · m / N) (2)
Here, p is a pilot carrier number that takes an integer value from 0 to P-1, and N can be arbitrarily set by the number of divisions, for example, a value such as 1024. Also, m is an integer value from 0 to N−1, and all N sets of phase rotation reference pilot carriers obtained by multiplying the reference pilot carrier by the above phase rotation are stored in the memory 32.

That is, when the reference CP is given by {r ₀ , r ₁ ,..., R _P-1 }, as the m-th set of phase rotation reference pilot carriers,
{R ₀ , r ₁ e ^{j2π1m / N} ,..., R _P-1 ^{j2π (P−1) m / N} } are stored in the memory 32.

  Here, the phase rotation reference pilot carrier is stored in the memory 32 in advance. However, the reference pilot carrier is generated every time reception is performed, the phase rotation processing is performed, and the phase rotation reference pilot carrier is generated. It can be omitted.

  In addition, although the phase rotation is given to the reference pilot carrier, the same result can be obtained even if the phase rotation is given to the reception pilot carrier side without changing the reference pilot carrier.

  The correlation calculation unit 33 sequentially performs cross-correlation calculations on the received pilot carriers extracted from the CP extraction unit 27 and all N sets of phase rotation reference pilot carriers output from the memory 32. The cross-correlation value holding unit 34 Hold N sets of cross-correlation values. FIG. 4 is an example in which the horizontal axis is plotted as m and the vertical axis is plotted as the absolute value | γ (m) | of the cross-correlation value, which is called a delay profile.

  The position of the signal that has passed through the effective path, which is a propagation path that can be detected on the delay profile, is roughly determined from the position indicated by the peak point. The calculation of the cross-correlation value γ (m) is performed using equation (3).

ここで、ｘ_ｐはパイロットキャリア番号がｐの受信パイロットキャリアであり、ｙ_ｍ，ｐは第ｍ組でパイロットキャリア番号がｐの位相回転基準パイロットキャリアである。また、ｘ_ａｖ，ｙ_ｍａｖはそれぞれ各組のｐ個にわたるｘ_ｐ，ｙ_ｍ，ｐそれぞれの平均値である。

Here, _xp is a received pilot carrier having a pilot carrier number p, and ym _{, p} are phase rotation reference pilot carriers having a pilot carrier number _p in the m-th set. Further, x _av and y _mav are average values of x _p , _ym , and p over p in each group _, respectively.

  Further, the above correlation calculation is performed in the frequency domain, but due to the definition of Fourier transform expressed by the equation (4), the time shift in the time domain corresponds to the phase rotation in the frequency domain.

このため、図５の変形例のブロック図に示すように、受信パイロットキャリアと基準パイロットキャリアをＩＦＦＴ演算部４１，４２でＩＦＦＴ演算して、それぞれ時間領域に変換した後、基準パイロットキャリア（または、受信パイロットキャリア）を時間ずらし部４３にてサンプル時間単位でずらしてメモリ４４に格納し、相関演算部４５で相関演算を行うことで、図４の遅延プロファイルと同様の結果を得ることができる。

For this reason, as shown in the block diagram of the modified example of FIG. 5, after the received pilot carrier and the reference pilot carrier are IFFT-calculated by the IFFT calculators 41 and 42 and converted into the time domain, respectively, the reference pilot carrier (or The reception pilot carrier) is shifted in units of sample time by the time shifting unit 43 and stored in the memory 44, and the correlation calculation is performed by the correlation calculation unit 45, whereby the same result as the delay profile of FIG. 4 can be obtained.

  In the following description, a method using phase rotation in the frequency domain shown in FIG. 3 will be described. Each peak point of the delay profile shown in FIG. 4 from the cross-correlation value holding unit 34 represents the reception timing of each path, and since the absolute value indicates the strength of the correlation, a preset threshold value (fixed value) is set. As an effective path, which is a propagation path that can detect a path exceeding the limit, it will be analyzed later.

  In order to increase accuracy, the above correlation calculation processing is performed over a plurality of symbols and averaged in the symbol direction to obtain a highly accurate delay profile. In the present invention, however, a measurable time range (m = 0 to N− 1) The purpose is not to obtain all delay profiles, but to obtain propagation parameters for each effective path.

Therefore, in the present invention, the signal having passed through the effective path is roughly identified from the peak point of the absolute value | γ (m) | of the cross-correlation value exceeding the preset threshold value by only a few symbols from the first symbol, Only the partial cross-correlation values before and after the peak point of the absolute value of the cross-correlation values (solid line part in FIG. 6) are obtained by averaging over a plurality of symbols, thereby accurately passing the effective path while reducing the amount of calculation. The cross-correlation values γ (m _d ), γ (m _u1 ), and γ (m _u2 ) for each signal can be extracted from the peak points.

  In the present invention using the correlation calculation, the maximum value of the cross-correlation value when the correlation is completely obtained is 1 and the minimum value of the cross-correlation value when the correlation is not completely obtained is 0 according to the equation (3). Therefore, even when averaging over a plurality of symbols, it is possible to easily set a threshold value (fixed value), and it is easy to extract a signal through an effective path regardless of signal power.

The above averaging process is performed by the averaging unit 35 in FIG. 3, and the effective path detection unit 36 sets the largest peak point exceeding a preset threshold as the correlation value of the desired wave, and sets the other peak points as the delayed wave. The effective path delay time detector 37 obtains the time difference from the desired wave as the delay time of each delayed wave. For example, in the delay profile shown in FIG. 6, the delay time (m _u1 -m _d ) and the delay time (m _u2 -m _d ) are obtained.

The effective path phase detector 38 obtains the correlation phase of each effective path component from the phase of the cross-correlation value at the peak point. For example, when the correlation value of the peak point of the desired wave is represented by equation (5), the correlation phase of the desired signal is estimated phi _d.

相関位相に関しても、ピーク点を求めるときと同様にして、シンボル方向にわたって平均をとることで、精度を高めることができる。ピーク点の相互相関値から得られる相関位相は、後述する方法による有効パスを通過した各信号の到来方向の推定や、同位相で合成するパスダイバーシチ受信に必要となる情報である。

As for the correlation phase, the accuracy can be improved by taking the average over the symbol direction in the same manner as when obtaining the peak point. The correlation phase obtained from the cross-correlation value at the peak point is information necessary for estimating the arrival direction of each signal that has passed through the effective path by the method described later, and for receiving path diversity combined at the same phase.

On the other hand, the effective path DU ratio detection unit 39 can roughly determine the DU ratio of each delayed wave as the ratio of the absolute values of the cross-correlation values with the desired wave. For example, the DU ratio of the first delayed wave (with the highest received power among the delayed waves) is obtained by | γ (m _d ) | / | γ (m _u1 ) |, and the DU ratio of the second delayed wave is | γ (m _d ) | / | γ (m _u2 ) |

<Effective path DU ratio detection>
In order to obtain the DU ratio of each delayed wave more accurately, the delay time information obtained by the effective path delay time detection unit 37 and the method previously proposed by the present applicant in Japanese Patent Application No. 2006-107182 are combined. This method can be obtained with high accuracy.

  In the method of the prior application, in order from the delayed wave with the larger DU ratio, the phase rotation amount due to the delay time is estimated from the distribution on the constellation of the received pilot carrier, and the signal passing through each effective path is separated. In the course of going, the arrival direction, delay time, phase rotation amount, DU ratio, and the like are obtained, and the relative reception level of each delayed wave can be obtained accurately.

  FIG. 7 is a block diagram for obtaining the DU ratio of each delay wave when three waves of the desired wave, the first delay wave, and the second delay wave are effective paths. In FIG. 3, an antenna 20, a frequency converter 21, an AD conversion unit 22, an orthogonal demodulation unit 23, a symbol synchronization unit 24, an FFT calculation unit 25, a CP extraction unit 27, and an effective path delay time detection unit 37 are shown in FIG. The description is omitted.

  In FIG. 7, the reception pilot carrier extracted by the CP extraction unit 27 is averaged for each signal value by the averaging unit 50, and the desired wave signal power detection unit 51 detects the square of the average value as the desired wave signal power. .

  On the other hand, an average value obtained for each signal value from each received pilot carrier is subtracted by the subtractor 52. This corresponds to removing the desired wave signal component from the received pilot carrier as described in the prior application. The signal input to the FFT operation unit 25 is an effective symbol in which symbol synchronization is completely achieved with the desired wave signal, and there is no phase rotation due to FFT window position shift of the desired wave.

  Subsequently, the phase shift unit 53 gives a phase rotation corresponding to the delay time of the first delay wave obtained by the effective path delay time detection unit 37 to the signal from which the desired wave signal component output from the subtraction unit 52 is removed. Compensate. The first delayed wave has the highest received power among the delayed waves, and can be determined, for example, by the absolute value of the cross-correlation value of the obtained effective path. As a result, symbol synchronization is achieved with the first delayed wave, and a received pilot carrier without phase rotation due to FFT window position shift of the first delayed wave is obtained.

  Further, the received pilot carrier symbol-synchronized with the first delayed wave is averaged for each signal value by the averaging unit 54, and the square of the average value is calculated by the first delayed wave signal power detecting unit 55. Detect as power.

  On the other hand, the average value obtained for each signal value from each received pilot carrier is subtracted by the subtracting unit 56. This corresponds to removing the first delayed wave signal component from the received pilot carrier. Finally, the phase corresponding to the delay time from the first delayed wave of the second delayed wave obtained by the effective path delay time detecting unit 37 to the signal from which the first delayed wave signal component output from the subtracting unit 56 is removed. The rotation is given by the phase shifter 57 to compensate. As a result, symbol synchronization is achieved with the second delayed wave, and a received pilot carrier without phase rotation due to FFT window position shift of the second delayed wave is obtained.

  Further, for the received pilot carrier symbol-synchronized with the second delayed wave, the second delayed wave signal is obtained by using the square of the average value for each signal value obtained by the averaging unit 58 as the signal power of the second delayed wave. It is detected by the power detection unit 59. With respect to each signal power detected as described above, the effective path DU ratio detection unit 60 can detect the DU ratio of each delayed wave.

<Effective path arrival direction detection>
Next, a method according to the present invention for estimating the direction of arrival of signals that have passed through each effective path will be described. In the direction-of-arrival estimation method proposed in the present invention, for example, another system of received signals in which an antenna is disposed at a position half a wavelength away from the carrier wave is necessary.

  FIG. 8 shows a block diagram of an embodiment of the arrival direction estimation apparatus of the present invention. Here, the system used to determine the delay time, correlation phase, and DU ratio of the signal that has passed through each effective path is the first receiving system, and is located at a position half a wavelength away from the antenna of the first receiving system. The direction of arrival is estimated using the system in which the antenna is arranged as the second receiving system.

  In the second receiving system, as in the first receiving system, the received signal is frequency-converted by the frequency converter, converted to a digital signal by the AD converter, and then separated into an IQ signal by digital orthogonal demodulation. . However, it is assumed that the same local oscillator output as that of the first receiving system is distributed and used, and the AD converter also performs sampling at the same sampling frequency. If such a condition is satisfied, the process of establishing symbol synchronization may be performed by extracting effective symbols using the timing established by the symbol synchronization unit 24 of the first reception system. The symbol synchronizer can be omitted.

  After the FFT calculation unit 62 performs an FFT operation on the effective symbols cut out by the second reception system, the CP extraction unit 63 extracts the received pilot carrier.

  Next, similarly to the processing performed in the first reception system, the cross-correlation calculation between the phase rotation reference pilot carrier generated by the reference CP generation unit 30 and rotated in phase by the phase rotation unit 31 and the reception pilot carrier is performed. This is performed by the correlation calculation unit 64. However, since the generation of the phase rotation reference pilot carrier is the same as that generated in the first reception system, as shown in FIG. 8, the reference CP generation unit, phase rotation unit, memory of the second reception system The 32 parts can be omitted. Further, in the second receiving system, based on the position of the peak point of the cross-correlation value indicating the signal that has passed through each effective path specified in the first receiving system, the phase rotation that gives the phase rotation corresponding to only that position By performing only the cross-correlation calculation with the reference CP, the cross-correlation value of the signal that has passed through each effective path can be obtained, so that the calculation amount can be greatly reduced. Here, the cross-correlation value of the second reception system does not necessarily become the peak point at the position of the peak point of the cross-correlation value of the first reception system.

For example, assuming that the first delayed wave that has passed through the effective path obtained in the first receiving system is at a position corresponding to m = _mu1 , and that the cross-correlation value is expressed by equation (6),

図９に示すように、第２の受信系統では、全Ｐ個の基準パイロットキャリア（ｐ＝０〜Ｐ−１）それぞれに対し、ｅｘｐ（ｊ・２πｐ・ｍ_ｕ１／Ｎ）を複素乗算して位相回転を与えた位相回転基準パイロットキャリアと、受信パイロットキャリアの相互相関演算だけを行えばよく、ｍ≠ｍ_ｕ１以外の点についての演算は行う必要がない。

As shown in FIG. 9, in the second receiving system, all P reference pilot carriers (p = 0 to P-1) are multiplied by exp (j · 2πp · m _u1 / N) in a complex manner. It is only necessary to perform the cross-correlation calculation between the phase rotation reference pilot carrier to which the phase rotation is applied and the reception pilot carrier, and it is not necessary to perform calculations for points other than m ≠ _mu1 .

If the cross-correlation value is expressed by equation (7) from the calculation result for m = _mu1 obtained in the second receiving system,

位相差△φ_ｕ１＝φ_ｕ１，２−φ_ｕ１，１が、各アンテナで受信した第１遅延波のアンテナ間位相差を示す。ここで、φ_ｕ１，２はφ_ｕ１，１にアンテナ間路長差に相当する位相差が加わったものに相当する。

The phase difference _{Δφ u1} = φ _u1,2 −φ _u1,1 indicates the inter-antenna phase difference of the first delayed wave received by each antenna. Here, φ _u1,2 corresponds to φ _u1,1 added with a phase difference corresponding to the path length difference between the antennas.

The effective path phase detector 65 shown in FIG. 8 detects the correlation phase φ _u1 , 2 of the first delayed wave of the second reception system and supplies it to the effective path phase difference detector 66. The effective path phase difference detection unit 66 has a phase difference Δ between the correlation phase φ _u1,2 of the first delay wave of the second reception system and the correlation phase φ _u1,1 of the first delay wave of the first reception system. Find _φu1 . Effective path arrival direction detection unit 67 obtains the DOA theta _u1 from the phase difference △ phi _u1.

FIG. 10 shows a general relationship when the antenna element interval is d, the radio wave arrival direction is θ, and the inter-antenna phase difference is Δφ. When the phase difference is _{Δφ u1} and the antenna interval is a half wavelength of the carrier frequency, the arrival direction of the radio wave can be estimated by the equation (8).

θ _u1 = sin ⁻¹ (Δφ _u1 / π) (8)
Similarly to the method described in the first receiving system, the correlation phase of the first delayed wave received in the second receiving system is obtained by averaging the cross-correlation values not only in one symbol but in the symbol direction. It can be determined with higher accuracy.

  The arrival direction is obtained in the same manner not only for the first delay wave but also for signals that have passed through other effective paths such as the desired wave and the second delay wave. The calculation can be obtained with a small calculation amount at the same time. In addition, the number of paths that can be estimated is not limited, and the arrival directions can be individually obtained for all path components that can be separated at the cross-correlation peak points.

<Effective path arrival direction detection (3 reception systems)>
In the above description, the number of receiving systems for estimating the direction of arrival is two, but the accuracy can be further improved by preparing three or more receiving systems and spatially averaging them.

  FIG. 11 shows an antenna having an antenna arranged at the same distance as the antenna interval (half wavelength of the carrier wave) of the first and second receiving systems from the antenna of the second receiving system, and having the same configuration as that of the second receiving system. The block diagram of embodiment which added 3 receiving systems is shown.

  In the figure, similarly to the processing performed in the second receiving system, based on the position of the peak point of the cross-correlation value indicating each effective path obtained in the first receiving system, the phase rotation corresponding to that position is performed. Only the cross-correlation operation between the given phase rotation reference pilot carrier and the received pilot carrier is performed, and the correlation phase of each signal in the third reception system is obtained.

  Thereafter, the effective path phase difference detection unit 68 obtains the phase difference between the antennas of the second and third receiving systems, and the phase difference averaging processing unit 69 averages the phase difference between the antennas of the first and second receiving systems. Thus, the phase difference between the antennas of each signal can be accurately obtained, and the accuracy of the arrival direction estimation in the effective path arrival direction detection unit 70 can be improved.

<Space-separated path diversity 1>
Next, using the propagation parameters estimated above, spatially separated paths that are spatially separated and received after passing through each path having a delay time equal to or shorter than the guard interval period received by the array antenna and then synthesized in the same phase A first embodiment of the diversity receiver will be described with reference to the block diagram of FIG.

  FIG. 12 has four receiving systems as an example, receives the desired wave and the first delay wave spatially separated, combines the phases of the respective signals, and then synthesizes the desired wave and the first delay wave. This is a multi-beam adaptive array antenna.

  In the figure, the signals received by the four antennas 80a to 80d, each of which is located at half the wavelength of the carrier wave, are converted to digital signal processing frequencies and then converted to digital signals by the AD converters 81a to 81d. To do. Each received signal is branched into two and one is input to the path estimation unit 82. The path estimation unit 82 has the same configuration as that shown in FIG. 11, and obtains propagation parameters such as the arrival direction, delay time, correlation phase, and DU ratio of the signals that have passed through each path as described above. The input to the path estimation unit 82 uses all four signals, but only two adjacent signals may be used.

  Next, the other branched signal is supplied to the phase shifters 83a1, 83a2-83d1, 83d2. The phase shifters 83a1, 83a2 to 83d1, 83d2 perform phase rotation based on the arrival direction information of the desired wave and the first delayed wave obtained by the effective path arrival direction detection unit 70 of the path estimation unit 82 on each signal by digital signal processing. To give the same phase. The synthesizer 84 synthesizes the first delayed wave with the same phase, and the synthesizer 85 synthesizes the desired wave with the same phase. The above processing corresponds to forming the directivity of the antenna individually in the arrival direction of each signal.

  Further, in order to realize the path diversity reception of the desired wave and the first delayed wave, first, the effective path of the path estimation unit 82 is received for the desired wave signal received separately from the first delayed wave with individual antenna directivity. A delay corresponding to the delay amount of the first delay wave estimated by the delay time detector 37 is provided by the delay unit 86 to compensate. The delay given here is realized by, for example, a tapped delay line (TDL).

  Next, since the propagation path is different between the desired wave and the first delayed wave, and the phase rotation of all carriers is the same between the desired wave and the first delayed wave due to the respective fading fluctuations, the effective path of the path estimation unit 82 A phase rotation corresponding to the difference in correlation phase between the desired wave and the first delayed wave estimated by the phase detector 38 (referred to as the relative phase between paths) is applied to the desired wave by the phase shifter 87 to compensate.

  Further, by combining the signal power ratio obtained by the effective path DU ratio detecting unit 39 of the path estimating unit 82 as the weighting coefficients h1 and h2 of the multiplying units 88 and 89 by the combining unit 90, the maximum ratio combining path diversity. Reception is possible.

  As described above, in an OFDM signal composed of a plurality of orthogonal subcarriers, in order to realize path diversity in which the signals that have passed through each path are combined in phase in the time domain, phase rotation (delay of delay) is achieved. Insertion) and phase rotation due to fading in the propagation path must be individually compensated.

  However, by using the configuration shown in FIG. 13, these phase rotation compensations can be simultaneously performed for each carrier. In FIG. 13, for the first delayed wave and desired wave signals combined with individual antenna directivities, one symbol worth extracted at the same symbol cutout position (assumed to be symbol-synchronized with the desired wave here). The OFDM signal is subjected to an FFT calculation in each of the FFT calculation units 91 and 92 and converted to the frequency domain.

  FIG. 14A shows the constellation (16QAM modulation) of the data carrier of the desired wave. On the other hand, a delayed wave signal obtained by cutting out a symbol at a position synchronized with the desired wave signal shows a constellation in which each carrier is individually rotated in phase as shown in FIG.

In order to compensate for this phase rotation, the phase rotation amount corresponding to the delay amount is compensated based on the delay time information obtained by the effective path delay time detection unit 37 of the path estimation unit 82. That is, the phase shifter 93 compensates the desired wave by applying phase rotation due to delay and phase rotation due to fading in the propagation path. When the delay time of the delayed wave is Δm _u (= m _u −m _d ), the phase rotation compensation amount Δφ _{mu, k} given to the received pilot carrier of the carrier number _k is expressed by equation (9). The constellation after this compensation is shown in FIG.

Δφ _{mu, k} = 2πk · Δm _u / N (9)
Next, the phase difference information obtained by the effective path phase detector 38 of the path estimator 82 to compensate for the phase difference (Δφ shown in FIG. 14C) with the desired wave caused by fading fluctuation in the propagation path. The amount of phase rotation is compensated based on FIG. 14D shows a constellation when this compensation is performed.

In the above description, although divided into two steps for convenience, the phase rotation amount corresponding to the delay time and the phase difference Δφ caused by the fading fluctuation in the propagation path are combined to perform phase compensation for each carrier at a time. be able to. Equation (10) shows the phase rotation compensation amount Δφ _k for each carrier when the compensation is performed simultaneously.

Δφ _k = Δφ _{mu, k} + Δφ (10)
With the above processing, the desired wave in FIG. 14A and the delayed wave in FIG. 14D can be synthesized in phase for each carrier in the frequency domain. Further, by using the weighting coefficients h1 and h2 as the power ratio of the signal obtained by the effective path DU ratio detection unit 39 of the path estimation unit 82, it is possible to receive path diversity reception of maximum ratio combining.

  In the description of FIG. 12 and FIG. 13, for the sake of simplicity, the configuration is such that the desired wave and one delayed wave are combined. However, the signal can be expanded by branching each signal according to the number of paths for which path diversity is attempted.

  FIG. 15 shows a block diagram when the number of paths to be combined is three. The signals received by the four antennas 80a to 80d, each of which is arranged at a half-wave distance of the carrier wave, are converted to digital signal processing frequencies, and then converted to digital signals by the AD converters 81a to 81d. Each received signal is branched into two and one is input to the path estimation unit 82.

  The other branched signal is supplied to the phase shifters 83a1, 83a2, 83a3 to 83d1, 83d2, and 83d3. The phase shifters 83a1, 83a2, 83a3 to 83d1, 83d2, and 83d3 are based on the arrival direction information of the desired wave, the first delay wave, and the second delay wave obtained by the effective path arrival direction detection unit 70 of the path estimation unit 82. Rotation is set to the same phase given to each signal by digital signal processing. The synthesizer 84 synthesizes the first delayed wave with the same phase, the synthesizer 85 synthesizes the desired wave with the same phase, and the synthesizer 95 synthesizes the second delayed wave with the same phase.

  Furthermore, the second delay wave estimated by the effective path delay time detection unit 37 of the path estimation unit 82 is obtained for the desired wave signal received separately from the first delay wave and the second delay wave with individual antenna directivity. A delay corresponding to the delay amount is given by the delay unit 96 to compensate. Further, the delay unit 97 compensates the signal of the first delayed wave with a delay corresponding to the delay amount with the second delayed wave.

  Next, a phase rotation corresponding to the difference (inter-path relative phase) of the correlation phase between the desired wave and the second delayed wave estimated by the effective path phase detection unit 38 of the path estimation unit 82 is given to the desired wave by the phase shift unit 98. The phase shift corresponding to the difference in correlation phase between the first delay wave and the second delay wave (inter-path relative phase) is applied to the first delay wave by the phase shifter 99 to compensate.

  Further, by combining the power ratio of the signal obtained by the effective path DU ratio detecting unit 39 of the path estimating unit 82 as the weighting coefficients h1, h2, and h3 of the multiplying units 100, 101, and 102, the combining unit 103 combines them. Synthetic path diversity reception is possible.

<Space separation type path diversity 2>
FIG. 16 shows a spatial separation type in which a signal passing through each path having a delay time equal to or shorter than the guard interval period received by the array antenna is spatially separated and then combined in the same phase using the propagation parameter estimated above. The block diagram of 2nd Embodiment of a path diversity receiver is shown.

  In FIG. 16, the signals received by the four antennas 80a to 80d, each of which is located at half the wavelength of the carrier wave, are converted to digital signal processing frequencies, and then converted to digital signals by the AD converters 81a to 81d. To do. The digital signal is converted into a frequency domain signal by the FFT calculation units 140a to 140d, branched into two, and one of the signals is input to the path estimation unit 141. The path estimator 141 has the same configuration as that shown in FIG. 11, and obtains propagation parameters such as the arrival direction, delay time, correlation phase, and DU ratio of the signal in the frequency domain that has passed through each path as described above.

In the weight vector estimation unit 142, the column vector h _k whose component is the cross-correlation value for each reception system of the k-th path specified by the path estimation unit 141 is used as the array response vector of the k-th path. Further, a pseudo correlation matrix R _xx is generated from the array response vector obtained for each propagation path using the equation (11), and a weight vector W for separating the signals of the respective propagation paths using the equation (12). _{Find k} . Here, sigma _n ² I is the noise matrix is added to the diagonal elements are the correlation matrix with a positive value is _regularized, transposed vector of _h ^{k *} is the conjugate vector of _{h k, h} ^{k T} is _{h k,} R _xx ⁻¹ is an inverse matrix of R _xx .

ウエートベクトル推定部１４２の出力するウエートベクトルＷ_１，Ｗ_２は希望波と第１遅延波を受信するためのウエートベクトルで、乗算部１４３ａ１，１４３ａ２〜１４３ｄ１，１４３ｄ２に供給される。

The weight vectors W ₁ and W ₂ output from the weight vector estimation unit 142 are weight vectors for receiving the desired wave and the first delay wave, and are supplied to the multiplication units 143a1, 143a2 to 143d1, 143d2.

Multiplying unit 143a1~143d1 multiplies the weight vector _{W 1} to an output signal of the FFT calculation unit 140a-d, the output signal of the multiplier unit 143a1~143d1 the desired wave is synthesized is supplied to the combining unit 145. Multiplying unit 143a2~143d2 multiplies the weight vector _{W 2} to the output signal of the FFT calculation unit 140a-d, the output signal of the multiplying unit 143a2~143d2 first delayed wave is synthesized is supplied to the combining unit 146.

  The output signals of the synthesis units 145 and 146 are equalized by the equalization units 147 and 148, and then supplied to the multiplication units 149 and 150. The weight ratio of the desired wave and the first delay wave obtained by the path estimation unit 141 is weighted. The coefficients h1 and h2 are multiplied and synthesized by the synthesis unit 151 and output.

In the second embodiment, the weight vector W _k forms a directional null in a path other than the desired path to be received and separated, so that each signal is more clearly separated spatially than in the first embodiment. Can be received.

  FIG. 17 shows a block diagram of a modification of the second embodiment of the space-separated path diversity receiver. In this modification, the path diversity reception of the desired wave and the first delay wave is realized in the time domain. That is, the output signals of the AD conversion units 81a to 81d are branched into two without being subjected to the FFT operation, and are supplied to the path estimation unit 141 and the multiplication units 143a1, 143a2 to 143d1, and 143d2.

  The first delayed wave output from the combining unit 146 is compensated for a delay corresponding to the delay amount of the first delayed wave estimated by the path estimating unit 82 by the delay unit 152, and the desired wave and the first delayed wave are output by the phase shift unit 153. After a phase rotation corresponding to the difference of the correlation phase is given, the synthesis unit 154 synthesizes the desired wave output from the synthesis unit 145. Thereafter, the signal is converted into a frequency domain signal by the FFT operation unit 155 and output.

<Frequency domain separation type path diversity>
Next, a description will be given of an embodiment of path diversity reception that separates a plurality of paths that have undergone fading fluctuations independent from each other within a single received signal and recombines them in the same phase without spatially separating the paths. To do. In this type of path diversity, in general, in the DS (Direct Sequence) method of spread spectrum technology, each path component is temporally separated using a spreading code having a high-speed chip rate, and signals passing through each path are separated. This can be realized by spreading and demodulating the signal.

  However, in the OFDM signal of the format defined in Non-Patent Document 2, the delay wave signal other than the desired wave received by the same receiving system can be reduced by the equalization process, but it is separated and the same. A so-called rake reception method for improving reception characteristics by combining in phase has not been proposed so far.

  In the present invention, among the propagation parameters obtained by the estimated path estimation, each signal is separated and re-synthesized in the frequency domain based on the delay time and phase difference information of the signal that has passed through each path. This is called diversity.

  FIG. 18 is a block diagram for explaining the principle of a frequency domain separation type path diversity receiving apparatus that synthesizes a desired wave and a first delay wave.

  In the figure, the received signal is converted into a digital signal, symbol synchronization is established by the symbol synchronization unit 24, one effective symbol of the desired wave is cut out, converted into a signal a in the frequency domain by the FFT operation unit 25, and then into two systems. Branch. An example of the constellation after the FFT operation is shown in FIG. 20A to 20F correspond to data carrier constellation (16QAM modulation) in each part of FIG.

  One of the branched signals a is supplied to the equalizing unit 110 in the same way as in the normal demodulation operation, and a desired wave signal c is obtained in which the influence of the propagation path characteristics (including the influence of the delayed wave) is canceled. It is done. An example of constellation after desired wave signal equalization is shown in FIG. However, only the signal value of the data carrier is displayed in the constellation.

  A block diagram of the equalization unit 110 is shown in FIG. In the figure, a CP extraction unit 111 extracts a pilot carrier inserted for equalization from a signal converted into a frequency domain, and a complex pilot unit 112 outputs the pilot carrier to the reference CP generation unit 113 as a reference pilot. By performing complex division with the carrier, propagation path characteristics (amplitude and phase fluctuations received in the propagation path) are obtained.

  The carrier filter 114 estimates and interpolates the propagation path characteristics of the data carrier between the pilot carriers, and then supplies it to the complex division unit 115. The complex division unit 115 obtains a desired wave signal c in which the influence of the propagation path characteristic is canceled by performing complex division with the propagation path characteristic of the data carrier estimated data carrier. Up to this point, the Institute of Television Engineers Technical Report ITE Technical Report Vol. 20, no. 53 pp. 55, “Examination of Adaptive Equalization Method in OFDM Demodulation” and the like. The amplitude averaging unit 116 calculates and outputs the average of the absolute values of the complex signal values for each carrier obtained by the complex division unit 112.

  In FIG. 18, the other of the branched signal a is supplied to the phase shifter 120 and compensated by applying a phase rotation based on the correlation phase of the desired wave obtained by the effective path phase detector 38 of the path estimator 82. FIG. 20B shows a constellation after phase compensation. Normally, an offset such as a direct current component is removed before the FFT computation unit 25 performs the FFT computation. If there is still a residual component, the offset removal unit 121 removes it.

  In each of the modulation schemes such as 16QAM and 64QAM, the offset removing unit 121 estimates the offset by, for example, averaging the signal value for each carrier because the signal value of each carrier is mapped around the origin. Thereafter, the estimated offset is subtracted from the signal value of each carrier to obtain the signal b shown in FIG. 20B from which the offset is removed. Here, for the sake of simplicity, it is assumed that the signal supplied to the FFT calculation unit 25 is an effective symbol that is extracted in synchronization with the desired wave and in synchronization with the desired wave.

  Next, the signal d shown in FIG. 20D is obtained by performing symbol determination on the output c of the equalization unit from the signal that has been subjected to phase rotation and offset compensation by the symbol determination unit 122 and correcting the level by the level correction unit 123. Is supplied to the subtracting unit 124 for subtracting for each carrier. The symbol determination unit 122 shown here determines and outputs the closest transmission signal value for each carrier in order to remove residual errors and noise. On the other hand, the level correction unit 123 performs processing for returning the signal equalized to the transmission signal level by the equalization unit 110 to the reception signal level before equalization. This can be realized by multiplying the signal value output from the symbol determination unit 122 by the average absolute value of the complex signal values for each carrier output from the amplitude averaging unit 116 of the equalization unit 110.

  By subtraction of the subtracting unit 124, the signal e shown in FIG. 20E corresponding to the delayed wave obtained by removing the signal component corresponding to the desired wave from the received signal is obtained.

  Furthermore, the delayed wave signal f shown in FIG. 20F is demodulated by performing equalization processing on the output of the subtracting unit 124 by the equalizing unit 125 having the same configuration as that in FIG. Thereafter, the complex signal value for each carrier output from the amplitude averaging unit 116 of each of the equalization units 110 and 125 is respectively added to the desired wave signal c shown in FIG. 20C and the delayed wave signal f shown in FIG. Multiplying the average value of absolute values (that is, signal power) as a weighting factor by the multipliers 126 and 127, and recombining each signal for each carrier by the adder 128 enables maximum ratio combining path diversity reception. Become.

  It is also possible to provide a threshold value for the DU ratio and not consider it as an effective path when the set threshold value is exceeded, and set the weighting factor to 0 so as not to contribute to demodulation. Further, in FIG. 18, the weighting coefficient is determined from the signal power obtained by the equalization units 110 and 125, but based on the DU ratio of the delayed wave estimated by the effective path DU ratio detection unit 39 of the path estimation unit 82. Weighting may be performed.

  FIG. 21 is a block diagram of an embodiment of a frequency domain separation type path diversity receiving apparatus that combines a desired wave, a first delay wave, and a second delay wave. Here, the first delayed wave indicates a larger received power than the second delayed wave regardless of the delay time.

  First, when the received signal in the time domain before the FFT calculation is expressed in a discrete time format, it is expressed by equation (13).

上式右辺の第１項から第３項は、それぞれ希望波、第１遅延波、第２遅延波に相当する項を示し、ａ_ｋはキャリア番号ｋの送信信号、ｂ_ｍは受信信号の時刻ｍに関する信号サンプリング値、ρ_ｄ，ρ_ｕ１，ρ_ｕ２は希望波、第１遅延波、第２遅延波それぞれの信号振幅に関するフェージング変動、φ_ｄ，φ_ｕ１，φ_ｕ２はそれぞれの位相に関するフェージング変動、ｍ_ｕ１，ｍ_ｕ２は第１遅延波、第２遅延波それぞれの遅延時間、ＮはＦＦＴのポイント数とする。ただし、簡単のため熱雑音は省略した。

The first to third terms on the right side of the above equation indicate terms corresponding to the desired wave, the first delay wave, and the second delay wave, respectively, a _k is the transmission signal of carrier number k, and b _m is the time of the reception signal m is a signal sampling value, ρ _d , ρ _u1 , and ρ _u2 are fading fluctuations relating to the signal amplitudes of the desired wave, the first delay wave, and the second delay wave, and φ _d , φ _u1 , and φ _u2 are fading fluctuations relating to the respective phases. , M _u1 and m _u2 are the delay times of the first delay wave and the second delay wave, and N is the number of FFT points. However, thermal noise was omitted for simplicity.

  When the signal corresponding to the above equation is converted into the frequency domain by the FFT calculation unit 25, the signal of the carrier number k is expressed by the equation (14).

次に、（１４）式で表される信号に対し、移相部１２０でパス推定部８２の有効パス位相検出部３８で得られた希望波の相関位相からの位相回転量φ_ｄに基づいた補償を行い（ｅ^−ｊφｄを乗算）、さらに等化部１１０、シンボル判定部１２２、レベル補正部１２３を経て得られた信号ρ_ｄａ_ｋを減算部１２４で減算すると、（１４）式の右辺第１項の希望波成分が除去された（１５）式で示される信号が得られる。ここで、直流成分などによるオフセットはないものとする。

Next, based on the phase rotation amount φ _d from the correlation phase of the desired wave obtained by the effective path phase detection unit 38 of the path estimation unit 82 by the phase shift unit 120 with respect to the signal represented by the equation (14). Compensation is performed (multiplication by e− ^jφd ), and the signal ρ _{d ak} obtained through the equalization unit 110, symbol determination unit 122, and level correction unit 123 is subtracted by the subtraction unit 124, then the right side of the equation (14) A signal represented by equation (15) from which the desired wave component of the first term is removed is obtained. Here, it is assumed that there is no offset due to a DC component or the like.

得られた信号を等化部１２５に供給して等化処理を行うことで、第１遅延波の信号を再生することができる。一方、第１遅延波の信号を除去し、第２遅延波の信号を再生するため、パス推定部８２の有効パス遅延時間検出部３７と有効パス位相検出部３８で得られた第１遅延波の遅延時間に相当する位相回転と相関位相差からの位相回転量を合わせた

By supplying the obtained signal to the equalization unit 125 and performing equalization processing, the signal of the first delay wave can be reproduced. On the other hand, the first delay wave obtained by the effective path delay time detection unit 37 and the effective path phase detection unit 38 of the path estimation unit 82 is used to remove the first delay wave signal and reproduce the second delay wave signal. The phase rotation corresponding to the delay time and the phase rotation amount from the correlation phase difference were combined

を移相部１３０で乗算して補償を行った後オフセット除去部１３３でオフセット除去を行い、さらに等化部１２５、シンボル判定部１３１、レベル補正部１３２を経て得られた、もう一方の信号ρ_ｕ１ａ_ｋを減算部１３４で減算すると、（１５）式の右辺第１項の第１遅延波成分が除去された（１６）式で示される信号が得られる。

Is compensated by the phase shifter 130, the offset is removed by the offset remover 133, and the other signal ρ obtained through the equalizer 125, the symbol determiner 131, and the level corrector 132 is obtained. _{When u1} a _k is subtracted by the subtracting unit 134, a signal represented by Expression (16) from which the first delayed wave component of the first term on the right side of Expression (15) is removed is obtained.

得られた信号を等化部１３５に入力して等化処理を行うことで、第２遅延波の信号を再生することができる。

By inputting the obtained signal to the equalization unit 135 and performing equalization processing, the signal of the second delayed wave can be reproduced.

  Through the above processing, the desired wave, the first delay wave, and the second delay wave are separated from each other and can have the same phase. Therefore, the multipliers 126, 127, and 136 are assigned appropriate weights h1, h2, and h3, respectively. In addition, by recombining by the adding unit 137, it is possible to perform path diversity reception that combines the OFDM signal with three paths.

  As described above, by using the weighting coefficients h1, h2, and h3 as the power ratio of the signal obtained by the amplitude averaging unit 116 or the effective path DU ratio detecting unit 39, it is possible to receive path diversity reception of maximum ratio combining. .

<Pilot carrier (SP)>
In the above embodiment, the OFDM signal of the format defined in Non-Patent Document 2 has been described. However, a pilot carrier (SP: Scattered Pilot) included in a terrestrial digital television broadcast wave defined in Non-Patent Document 3 is described. However, although the position of the inserted carrier is different, the present invention can be easily applied by a method similar to that of the pilot carrier (CP).

  FIG. 21 shows the arrangement of pilot carriers (SP). The arrangement of the pilot carriers (SP) satisfies the relationship of equation (15), where n is the symbol number and k is the carrier number.

k mod 12 = 3 (n mod 4) (15)
However, mod indicates a remainder. The carrier symbol to be transmitted is a value forming a pseudo-random sequence using the equation (16) as a generator polynomial, and is BPSK modulated in the same manner as the pilot carrier CP, and the amplitude and phase are known on the receiving side.

g (x) = x ¹¹ + x ⁹ +1 (16)
As shown in FIG. 22, the pilot carrier (SP) is different from the pilot carrier (CP), and the carrier number of the pilot carrier (SP) to be inserted is different in four consecutive symbols. However, interpolation interpolation, that is, processing is performed once every four symbols for a carrier in which no pilot carrier (SP) exists, so that the pilot carrier is always transmitted to the same carrier (every three carriers) in the symbol direction in the same manner as the pilot carrier CP. It can be considered that (SP) is inserted. Further, although the detection time range of the delayed wave is narrowed, correlation calculation may be performed for each symbol using a pilot carrier (SP) of a 4-symbol sequence.

  Therefore, the embodiment of the present invention has been described using the signal of the format defined in Non-Patent Document 2, but the digital terrestrial television broadcasting defined in Non-Patent Document 3 is performed by performing processing once for four symbols. It can also be applied to waves. In the description, FFT or IFFT is used for conversion between the time domain and the frequency domain. However, DFT (Discrete Fourier Transform) or IDFT (Inverse Discrete Fourier Transform) can also be used.

  In this way, the threshold for detecting the effective path can be easily set, and the delay time, correlation phase, and DU ratio of the detected effective path can be obtained simultaneously and with high accuracy with a small amount of calculation. . In addition, it is possible to accurately estimate the arrival directions of all effective paths that can be detected by at least two antennas, and to separate each path component of the OFDM signal in the spatial domain or frequency domain based on the estimation result The reception characteristics can be improved by combining the components in the same phase.

  The CP extraction unit 27 corresponds to the extraction unit described in the claims, the phase rotation unit 31 corresponds to the phase rotation unit, the correlation calculation unit 33 corresponds to the correlation calculation unit, the cross correlation value holding unit 34, the effective path The detection unit 36 to the effective path DU ratio detection unit 39 correspond to the propagation parameter calculation unit, the averaging unit 35 corresponds to the averaging unit, the effective path arrival direction detection unit 67 corresponds to the arrival direction detection unit, and 83a1 to 83a1. 83d2 corresponds to the phase rotation unit, the synthesis units 84 and 85 correspond to the first synthesis unit, the delay unit 86 to the synthesis unit 90 correspond to the second synthesis unit, and the path estimation units 82 and 141 and the weight vector The estimation unit 142 corresponds to weight vector estimation means, the multiplication units 143a1, 143a2 to 143d1, 143d2, the synthesis units 145, 146 correspond to signal separation means, the equalization units 147, 148, and the multiplication unit 14 , 150, combining unit 151, a delay unit 152, the phase shift unit 153 corresponds to the path combining means, the equalizer 110 to multiplying section 127 corresponds to the propagation path separating means, the adding unit 128 is equivalent to the combining means.

It is a general block diagram of a propagation analysis device for obtaining a delay profile. It is a figure for demonstrating the arrangement | sequence of a pilot carrier (CP). It is a block diagram of one Embodiment of the propagation path estimation apparatus of this invention. It is a figure which shows a delay profile. It is a block diagram of the modification of one Embodiment of the propagation path estimation apparatus of this invention. It is a figure for demonstrating a partial correlation calculation. It is a block diagram of an effective path DU ratio detection unit. It is a block diagram of one embodiment of an arrival direction estimating device of the present invention. It is a figure for demonstrating a partial correlation calculation. It is a figure which shows the relationship between a phase difference and the arrival direction of a desired wave. It is a block diagram of the arrival direction estimation apparatus of 3 reception systems. 1 is a block diagram of a first embodiment of a space-separated path diversity receiver. FIG. It is a block diagram of the modification of 1st Embodiment of a space separation type | formula path diversity receiver. It is a figure which shows the constellation of the data carrier in FIG. It is a block diagram of a space-separated path diversity receiver when the number of paths to be combined is 3. It is a block diagram of 2nd Embodiment of a space-separated path diversity receiver. It is a block diagram of the modification of 2nd Embodiment of a space separation type | formula path diversity receiver. Block diagram for explaining the principle of a frequency domain separation type path diversity receiving apparatus that synthesizes a desired wave and a first delay wave It is a block diagram of an equalization part. It is a figure which shows the constellation of the data carrier in FIG. It is a block diagram of one Embodiment of the frequency domain separation type | formula path diversity receiver which synthesize | combines a desired wave, a 1st delay wave, and a 2nd delay wave. It is a figure for demonstrating the arrangement | sequence of a pilot carrier (SP).

Explanation of symbols

20 Antenna 21 Frequency Converter 22 AD Converter 23 Orthogonal Demodulator 24 Symbol Synchronizer 25 FFT Calculator 26 Demodulator 27, 63 CP Extractor 30 Reference CP Generator 31 Phase Rotator 32, 44 Memory 33, 45, 64 Correlation Calculation unit 34 Cross-correlation value holding unit 35, 50, 58 Averaging unit 36 Effective path detection unit 37 Effective path delay time detection unit 38, 65 Effective path phase detection unit 39, 60 Effective path DU ratio detection unit 41, 42 IFFT calculation Unit 43 Time shift unit 45 Correlation calculation unit 51 Desired wave signal power detection unit 52, 56 Subtraction unit 53, 57 Phase shift unit 55 First delay wave signal power detection unit 59 Second delay wave signal power detection unit 66, 68 Effective path Inter-phase difference detector 67, 70 Effective path arrival direction detector 69 Phase difference average processor 80a-80d Antenna 81a-81d AD conversion unit 82 Path estimation unit 83a1, 83a2-83d1, 83d2, 87, 93 Phase shift unit 84, 85, 90 Synthesis unit 86, 96, 97 Delay unit 88, 89, 100-102 Multiplication unit 91, 92 FFT calculation unit 110, 125, 135 Equalization unit 120, 130 Phase shift unit 121, 133 Offset removal unit 122, 131 Symbol determination unit 123, 132 Level correction unit 124, 134 Subtraction unit 126, 127, 136 Multiplication unit 128, 137 Addition unit

Claims (6)

A propagation path estimation device that receives a radio wave including a pilot carrier and estimates a propagation path parameter,
Extracting means for extracting a pilot carrier from a signal in a frequency domain obtained by converting a signal received by an antenna;
Phase rotation means for sequentially applying a predetermined amount of phase rotation to a reference pilot carrier prepared in advance or a reception pilot carrier extracted by the extraction means;
Correlation calculating means for obtaining a cross-correlation value by performing a correlation calculation of the reference pilot carrier or the received pilot carrier whose phase is rotated by the phase rotating means and the received pilot carrier or the reference pilot carrier which is not phase-rotated,
A delay profile is generated from the cross-correlation value to obtain a cross-correlation value for the desired wave and the delayed wave, and from the cross-correlation value, a signal delay time, a correlation phase, and a ratio of the desired wave and the delayed wave for each propagation path Propagation parameter calculation means for calculating a certain DU ratio ;
Averaging means for averaging the cross-correlation values obtained by the correlation calculating means over a plurality of symbols and supplying the averaged values to the propagation parameter calculating means;
Have
The propagation parameter calculation means roughly specifies a peak point of an absolute value of a cross-correlation value exceeding a preset threshold value from a cross-correlation value of the first one symbol or several symbols as a propagation path, and cross-correlates before and after the peak point. A propagation path estimation device characterized in that a peak point of an absolute value of a cross-correlation value obtained by averaging correlation values over a plurality of symbols is specified as a propagation path. A plurality of extraction means for extracting a pilot carrier from a signal in a frequency domain obtained by converting a signal received by a plurality of antennas;
Phase rotation means for sequentially applying a predetermined amount of phase rotation to a reference pilot carrier prepared in advance;
A plurality of correlation calculation means for performing a correlation calculation of the reference pilot carrier phase-rotated by the phase rotation means and the received pilot carrier obtained by the plurality of extraction means to obtain cross-correlation values of a plurality of systems;
A delay profile is generated from the cross-correlation values of the plurality of systems to obtain a propagation path of a desired wave and a delay wave, and a signal delay time for each propagation path is determined from the cross-correlation value of the one propagation path of the plurality of systems. Calculating a correlation phase, a DU ratio that is a ratio of the desired wave and the delayed wave, and calculating a correlation phase of a signal for each propagation path from cross-correlation values of the propagation paths of a plurality of systems other than the one system Propagation parameter calculation means;
A propagation path estimation apparatus for <br/> have a arrival direction detection means for detecting the direction of arrival of signals for each propagation path from the correlation phase of a plurality of systems of propagation paths for each of the signals obtained by the plurality of transmission parameter calculating means,
Phase rotation means for rotating the phase of each signal received by the plurality of antennas based on the direction of arrival of the signal for each propagation path;
First combining means for separating and combining the plurality of phase-rotated signals for each propagation path;
A second combining unit configured to combine the combined signal separated for each propagation path in the same phase based on a delay time and a correlation phase of the signal for each propagation path;
A path diversity receiver characterized by comprising: The path diversity receiving device according to claim 2 ,
The path diversity receiver characterized in that the second combining means combines the desired wave and the delayed wave of the propagation path by weighting based on the DU ratio. A plurality of extraction means for extracting a pilot carrier from a signal in a frequency domain obtained by converting a signal received by a plurality of antennas;
Phase rotation means for sequentially applying a predetermined amount of phase rotation to a reference pilot carrier prepared in advance;
A plurality of correlation calculation means for performing a correlation calculation of the reference pilot carrier phase-rotated by the phase rotation means and the received pilot carrier obtained by the plurality of extraction means to obtain cross-correlation values of a plurality of systems;
A delay profile is generated from the cross-correlation values of the plurality of systems to obtain a propagation path of a desired wave and a delay wave, and a signal delay time for each propagation path is determined from the cross-correlation value of the one propagation path of the plurality of systems. Calculating a correlation phase, a DU ratio that is a ratio of the desired wave and the delayed wave, and calculating a correlation phase of a signal for each propagation path from cross-correlation values of the propagation paths of a plurality of systems other than the one system Propagation parameter calculation means;
Arrival direction detection means for detecting the arrival direction of a signal for each propagation path from a plurality of correlation phases of signals for each propagation path obtained by the plurality of propagation parameter calculation means;
A propagation path estimation device having
Weight vector estimation means for generating weight vectors for separating signals of a plurality of propagation paths from the cross-correlation values for each of the plurality of antennas obtained by the propagation path estimation device;
Signal separation means for separating the signals of the plurality of propagation paths by multiplying the signals received by the plurality of antennas by the weight vector and combining the signals;
Path diversity means comprising path combining means for equalizing each signal separated for each propagation path, or combining after equalizing based on the delay time and correlation phase of the signal for each propagation path Receiver device. A propagation path estimation device according to claim 1 ;
A propagation path for separating the signal for each propagation path in the frequency domain using the delay time and correlation phase of the signal for each propagation path obtained by the propagation parameter calculation means from the frequency domain signal obtained by converting the signal received by the antenna Separating means;
A path diversity receiving apparatus comprising combining means for combining the signals for each of the propagation paths separated by the propagation path separating means in the same phase. The path diversity receiver according to claim 5 , wherein
The path diversity receiver characterized in that the combining unit combines each desired wave and delayed wave of the propagation path by weighting based on the DU ratio.

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