JP4611199B2 - Hardware architecture for processing Galileo alternating binary offset carrier (AltBOC) signals - Google Patents

Abstract

A GNSS receiver tracks the AltBOC ( 15,10 ), or composite E5a and E5b, codes using hardware that locally generates the complex composite signal by combining separately generated real and the imaginary components of the complex signal. To track the dataless composite pilot code signals that are on the quadrature channel of the AltBOC signal, the receiver operates PRN code generators that produce replica E5a and E5b PRN codes and square wave generators that generate the real and imaginary components of the upper and lower subcarriers, and combines the signals to produce a locally generated complex composite code. The receiver removes the complex composite code from the received signal by multiplying the received signal, which has been downconverted to baseband I and Q signal components, by the locally generated complex composite code. The receiver then uses the results, which are correlated I and Q prompt signal values, to estimate the center frequency carrier phase angle tracking error. The error signal is used to control a numerically controlled oscillator that operates in a conventional manner, to correct the phase angle of the locally generated center frequency carrier. The receiver also uses early and late versions of the locally generated complex composite pilot code in a DLL, and aligns the locally generated composite pilot code with the received composite pilot code by minimizing the corresponding DLL error signal. Once the receiver is tracking the composite pilot code, the receiver determines its pseudorange and global position in a conventional manner. The receiver also uses a separate set of correlators to align locally generated versions of the in-phase composite PRN codes with the in-phase channel codes in the received signal, and thereafter, recover the data that is modulated thereon.

Description

  The present invention relates generally to GNSS receivers, and more particularly to receivers that operate on Galileo AltBOC satellite signals.

  Global Navigation Satellite System (GNSS) receivers, such as GPS receivers, determine their global position based on orbiting GPS and signals received from other satellites. A GPS satellite, for example, transmits signals using two carriers: an L1 carrier at 1575.42 MHz and an L2 carrier at 1227.60 MHz. Each carrier is modulated by at least a binary pseudorandom (PRN) code that consists of a seemingly random sequence of 1s and 0s that repeats periodically. The 1s and 0s in the PRN code are referred to as “code chips” and the transitions in the 1 to 0 or 0 to 1 codes that occur in “code chip time” are referred to as “bit transitions”. Each GPS satellite uses a unique PRN code, so the GPS receiver can associate the received signal with a particular satellite by determining which PRN code is included in the signal. .

  The GPS receiver calculates the difference between the time when the satellite transmits its signal and the time when the receiver receives the signal. The receiver then calculates its distance from the satellite, or “pseudorange”, based on the associated time difference. Using pseudoranges from at least four satellites, the receiver determines its global position.

  To determine the time difference, the GPS receiver synchronizes the locally generated PRN code with the PRN code in the received signal by aligning the code chip with each of the codes. The GPS receiver then determines how much the locally generated PRN code has been shifted in time from the known timing of the satellite PRN code at the time of transmission and calculates the associated pseudorange. The more closely the GPS receiver aligns the locally generated PRN code with the PRN code in the received signal, the more GPS receivers are associated with the associated time difference and pseudorange, then their total. The position of the earth can be determined more accurately.

  The code synchronization operation includes acquisition of the satellite PRN code and tracking of the code. To obtain a PRN code, a GPS receiver generally makes a series of correlation measurements that are separated in time by a code chip. After acquisition, the GPS receiver tracks the received code. It generally includes “early-minus-late” correlation measurements, ie, (i) correlation measurements associated with early versions of PRN codes and locally generated PRN codes in the received signal; , (Ii) measure the difference between the correlation measurement associated with the delayed version of the PRN code and the local PRN code in the received signal. The GPS receiver then uses an early minus late measurement in a delay lock loop (DLL) that generates an error signal proportional to the misalignment between the local and received PRN codes. The error signal is then used to control a PRN code generator that shifts the local PRN code to substantially minimize the DLL error signal.

  The GPS receiver also typically aligns the satellite carrier with the local carrier using correlation measurements associated with a punctual version of the local PRN code. To do this, the receiver uses a carrier tracking phase lock loop.

  GPS receivers have not only line-of-sight or direct-path satellite signals, but also multi-path signals, which are signals that travel along different paths and are reflected to the receiver from the earth, waters, nearby buildings, etc. Receive. The multipath signal arrives at the GPS receiver after the direct path signal and is combined with the direct path signal to produce a distorted received signal. This distortion of the received signal adversely affects the code synchronization operation because the correlation measurement that measures the correlation between the local PRN code and the received signal is the total received signal containing its multipath component. Because it is based on. The distortion may be such that the GPS receiver attempts to synchronize to the multipath signal instead of the direct path signal. This is particularly true for multipath signals having code bit transitions that occur close to the time that code bit transitions occur in the direct path signal.

  One method for more accurately synchronizing the received signal with the locally generated PRN code is assigned to a common assignee and is incorporated herein by reference. Using the “narrow correlator” discussed in US Pat. Nos. 416, 5,390,207 and 5,495,499. It has been determined that narrowing the delay space between early and late correlation measurements substantially reduces the negative effects of noise and multipath signal distortion in early minus late measurements.

  The delay space is narrowed so that the noise is correlated in early and late correlation measurements. Also, narrow correlators are essentially spaced closer to the correlation peaks associated with punctual PRN code correlation measurements than the contribution of many multipath signals. Thus, the early minus late correlation measurements made by these correlators are distorted much less than would be the case if they made a larger interval around the peak. The closer the correlator is located to the correlation peak, the more adversely affected the multipath signal on the correlation measurement. However, the delay space cannot be made so narrow that the DLL cannot lock to the satellite PRN and then cannot maintain code lock. If not, the receiver cannot track the PRN code in the received signal without iteratively taking the time to relock to the code.

  The L1 carrier is modulated by two PRN codes: a 1.023 MHz C / A code and a 10.23 MHz P code. The L2 carrier is modulated by a P code. In general, a GPS receiver configured according to the above-mentioned patent obtains a satellite signal using a locally generated C / A code and a locally generated L1 carrier. After acquisition, the receiver uses the narrow correlator in the DLL and the punctual correlator in the carrier tracking loop to generate the locally generated C / A code and the L1 carrier, the C / A code in the received signal, and Synchronize with the L1 carrier. The receiver may then use the C / A code tracking information to track the C / A code and the L1 and / or L2 P codes that have a known timing relationship with each other.

  In a newer generation of GPS satellites, the L2 carrier is also modulated by a C / A code that is modulated by a 10.23 MHz square wave. Hereafter, the square wave modulated C / A code, referred to as the split C / A code, has a maximum in its power spectrum at a ± 10 MHz offset from the L2 carrier or in the null of the power spectrum of the P code. The split C / A code can thus be selectively disturbed as needed without disturbing the L2 P code.

  The autocorrelation function associated with the split C / A code is an envelope that corresponds to the autocorrelation of the 1.023 MHz C / A code, and a number of peaks in the envelope that correspond to the autocorrelation of the 10.23 MHz square wave. Have. Thus, there are 20 peaks in the two chip C / A code envelopes, ie, there is a square wave autocorrelation peak for each 0.1 C / A code chip. The multiple peaks associated with the square wave are each relatively narrow, thus providing the accuracy to track the increased code, assuming that the DLL tracks the correct narrow peak.

  A locally generated 10.23 MHz that can be thought of as a 20.46 MHz square wave code, as discussed in US Pat. No. 6,184,822, assigned to a common assignee and incorporated herein by reference. There are advantages to obtaining and tracking split C / A codes by separately aligning the phase of the square wave and locally generated 1.023 MHz C / A code with the received signal. The receiver first aligns the phase of the locally generated square wave code with the received signal and tracks one of the multiple peaks of the autocorrelation function of the split C / A code. It then shifts the phase of the locally generated C / A code relative to the phase of the locally generated square wave code, aligns the local and received C / A code, and divides C Position the correlator on the central peak of / A. The receiver then tracks the center peak directly with the locally generated split C / A code.

  The European Commission and the European Space Agency (ESA) are developing a GNSS known as Galileo. The Galileo satellite uses the proposed modulation known as alternating binary offset carrier wave (AltBOC) as a decoded signal with a center frequency of 1195.795 MHz as E5a band (1176.45 MHz) and E5b band (1207.14 MHz). Transmit the signal at. The generation of AltBOC signals is described in the Galileo Signal Task Force document “Technical Annex to the Galileo SRD Signal Plan” Draft 1, July 18, 2001, ref # STF-annexSRD-2001 / 003, in its entirety. Is hereby incorporated by reference. Like GPS satellites, each GNSS satellite transmits its own PRN code, and the GNSS receiver can thus associate the received signal with a particular satellite. Thus, the GNSS receiver determines the respective pseudoranges based on the difference between the time when the satellite transmitted the signal and the time when the receiver received the AltBOC signal.

A standard binary offset carrier (BOC) is generated by a sine wave sin (w ₀ t) that shifts the frequency of the signal to both the upper sideband (sideband) and the corresponding lower sideband (sideband). Modulate the time domain signal. BOC modulation achieves a frequency shift using a square wave or sine wave (sin (w ₀ t)) and is generally represented as BOC (fs, fc), where fs is a subcarrier (square wave). Frequency and fc is the spreading code chipping rate. The 1.023 MHz factor is usually omitted from the notation for clarity, and BOC (15.345 MHz, 10.23 MHz) modulation is represented as BOC (15, 10). BOC modulation, which produces a signal similar to the above-described split C / A code, for example, allows a single spread or PRN code on each of the in-phase and quadrature carriers.

^Modulation of the time domain signal with the complex exponential function ^ew0t shifts the frequency of the signal only to the upper sideband. The goal of AltBOC modulation is to generate E5a and E5b bands modulated by complex exponential functions or sub-carriers, respectively, in a coherent manner so that the signal can be received as a wideband “BOC-like signal”. The E5a and E5b bands each have an associated in-phase and quadrature spread, or PRN, code, the E5a code is shifted to the lower sideband, and the E5b code is shifted to the upper sideband. Each E5a and E5b quadrature carrier is modulated by a pilot signal without data, and each in-phase carrier is modulated by both a PRN code and a data signal. The GNSS receiver may track either E5a code or E5b code in a manner similar to the above-described split C / A code tracking.

  However, there is an advantage in both tracking accuracy and multipath mitigation associated with tracking the decoded E5a and E5b signals, ie, tracking the broadband AltBOC coherent signal. Each in-phase and quadrature carrier of the decoded signal is modulated by a complex spreading code, and thus the in-phase and quadrature channels each contain contributions from both the real and imaginary signal components of the E5a and E5b codes. The theoretical analysis of the decoding tracking operation has been performed using high-level mathematics. Thus, associated receivers that inherently reproduce high levels of mathematical behavior are expected to be complex and expensive.

  One proposed receiver uses the same lookup table that Galileo satellites use to generate signals for transmission, ie, the table corresponding to the basic phase shift keying (PSK) spreading code, and AltBOC Generate a local version of the decryption code. The proposed receiver therefore not only has to maintain a large look-up table for each of the codes transmitted by each Galileo satellite, but the receiver also A complex circuit that controls entry to the lookup table must be operated every time a chip is received. The tables are getting larger and more complicated to enter when different pilot codes are used on the E5a and E5b bands, as intended now.

  The present invention uses AltBOC (15, 10), ie, composites E5a and E5b, using hardware to locally generate a complex composite signal by combining the separately generated real and imaginary components of the complex signal. A GNSS receiver that tracks the code. For example, to track a dataless composite pilot code signal that is on the quadrature channel of an AltBOC signal, the receiver uses a local version of the composite pilot code as a combination of locally generated real and imaginary pilot signal components. Is generated. Thus, the receiver operates a PRN code generator that generates replica E5a and E5b PRN codes, and a square wave generator that generates the real and imaginary components of the upper and lower subcarriers.

  The receiver multiplies the received signal that has been down-converted (frequency down-converted) into baseband I and Q signal components by a locally generated complex composite code to complex the received signal. Remove compound code. The receiver then evaluates the center frequency carrier phase angle tracking error using the result, which is the correlated I and Q acceleration signal values. The error signal is used to control a numerically controlled oscillator operating in the normal manner to correct the phase angle of the locally generated center frequency carrier. The receiver also uses the early and late versions of the locally generated complex composite pilot code in the DLL and minimizes the corresponding DLL error signal to minimize the locally generated composite signal. The pilot code is aligned with the received composite pilot code.

  When the receiver is tracking the composite pilot code, the receiver determines its pseudorange and global position in the usual manner. Further, as described in more detail below, the receiver uses a separate set of correlators to convert the locally generated version of the in-phase composite PRN code and the in-phase channel code in the received signal. Align and then collect the data modulated on it.

  The present invention will be described below with reference to the accompanying drawings.

  The Galileo AltBOC modulation scheme is an “BOC (15,10) signal with E5a and E5b bands with their respective spreads, or PRNs, codes on the AltBOC (15,10) signals, ie their in-phase and quadrature carriers 10) Generate a state "signal. The AltBOC (15,10) signal has a center carrier frequency of 1191.795 MHz with an E5a band (1176.45 MHz) as the lower sideband and an E5b band (1207.14 MHz) as the upper sideband and 15.345 MHz. Subcarrier frequency.

  AltBOC (15,10) signal is constant including E5a and E5b spread, or PRN, code and data composite on the same phase channel, and E5a and E5b dataless PRN, or pilot, code composite on the quadrature channel. Is generated on the satellite as an envelope signal. FIG. 1 shows the frequency spectrum of the AltBOC PRN quadrature channel sequence.

  The idealized normalized autocorrelation function for the AltBOC (15,10) signal is shown in FIG. The envelope 100 of the autocorrelation function 111 is the autocorrelation function of the 10.23 MHz chipping rate signal, and the multiple peaks of the autocorrelation function 111 are associated with a 15.3454 MHz subcarrier that can be considered a complex square wave code. is doing.

  The operation of the GNSS receiver 10 when tracking AltBOC (15,10) Galileo satellite signals is described below. Section 1 describes the operation for tracking the quadrature dataless pilot code as a composite code. Section 2 describes the operation for retrieving E5a and E5b data from the in-phase composite data code. The following description assumes that the receiver has acquired a center frequency carrier using a normal carrier tracking loop (not shown).

ここに、ｃ_１は同相Ｅ５ｂコードであり、ｃ_２は同相Ｅ５ａコードであり、ｃ_３は直角位相Ｅ５ｂコードであり、ｃ_４は直角位相Ｅ５ａコードであり、そしてＥ５ｂ及びＥ５ａは、それぞれ、上部搬送波ｅｒ（ｔ）、及びｅｒ（ｔ）の共役複素数である下部搬送波ｅｒ^＊（ｔ）上で変調される。上部搬送波ｅｒ（ｔ）は、

であり、ここに、ｃｒ（ｔ）＝符号（ｃｏｓ（２πｆ_ｓｔ））、ｓｒ（ｔ）＝符号（ｓｉｎ（２πｆ_ｓｔ））であり、ｆ_ｓは、副搬送波周波数である。下部搬送波ｅｒ^＊（ｔ）は、

である。ＡｌｔＢＯＣ（１５，１０）信号の直角位相チャンネル上にある複合パイロット・コードは、Ｅ５ａ及びＥ５ｂ直角位相コードｃ_４（ｔ）及びｃ_３（ｔ）を含む。直角位相チャンネル信号に対する表現ｘ_ｑ（ｔ）は、同相チャンネル・コードをゼロにセットし、搬送波に対する表現を置換することにより式（１）から導出される。 Section 1. Compound Pilot Code Tracking The AltBOC signal is given by:

Where c ₁ is an in-phase E5b code, c ₂ is an in-phase E5a code, c ₃ is a quadrature E5b code, c ₄ is a quadrature E5a code, and E5b and E5a are respectively upper The carrier wave er (t) and the lower carrier wave er ^* (t), which is a conjugate complex number of er (t), are modulated. The upper carrier wave er (t) is

Where cr (t) = sign (cos (2πf _s t)), sr (t) = sign (sin (2πf _s t)), and f _s is the subcarrier frequency. The lower carrier er ^* (t) is

It is. The composite pilot code on the quadrature channel of the AltBOC (15,10) signal includes E5a and E5b quadrature codes c ₄ (t) and c ₃ (t). The representation x _q (t) for the quadrature channel signal is derived from equation (1) by setting the in-phase channel code to zero and replacing the representation for the carrier.

式（２）の項は、次に、実数成分及び虚数成分に分離され得る。

受信器は、図４を参照して一層詳細に説明するように、局部的に発生された実数及び虚数信号成分を結合することによって、複素複合パイロット・コードの局部バージョンを生成する。受信器は、次に、以下の図５を参照して一層詳細に説明するように、局部的に発生された複合パイロット・コードを、受信された信号における対応の複合パイロット・コードと相関させる。受信器は、次に、従来の態様で、関連の擬似距離とその全地球位置とを決定する。

The terms of equation (2) can then be separated into real and imaginary components.

The receiver generates a local version of the complex composite pilot code by combining locally generated real and imaginary signal components, as described in more detail with reference to FIG. The receiver then correlates the locally generated composite pilot code with the corresponding composite pilot code in the received signal, as described in more detail with reference to FIG. 5 below. The receiver then determines the associated pseudorange and its global position in a conventional manner.

  Referring now to FIG. 3, the GNSS receiver 10 receives a signal via the antenna 12 that includes an AltBOC composite code transmitted by all of the satellites in view. The received signal is provided to a down converter 14 which, in the normal manner, converts the received signal to an intermediate having a frequency compatible with the analog to digital converter 18. Convert to frequency (“IF”) signal.

  The IF signal is then applied to an IF band filter 16 having a band at the desired center carrier frequency. The bandwidth of the filter 16 should be wide enough to allow the first harmonic of the AltBOC composite pilot code, i.e., about 1192 MHz, to pass. The wide bandwidth results in a relatively sharp bit transition in the received code and thus a fairly well defined correlation peak.

The analog-to-digital converter 18 samples the filtered IF signal at a rate that satisfies the Nyquist theorem and generates corresponding digital in-phase (I) and quadrature (Q) signal samples in a known manner. The I and Q digital signal samples are supplied to a Doppler cancellation processor 20 that operates in a known manner to generate _baseband I _baseband and Q _baseband samples by rotating the signal according to an estimate of the center frequency carrier phase angle. The evaluation of the carrier phase angle is based in part on the signal generated by the carrier numerically controlled oscillator (“carrier NCO”) 30 that is adjusted according to the carrier phase error tracking signal generated by the correlation subsystem 22. . The operation of the correlator subsystem is described below with reference to FIG.

The I _baseband and Q _baseband samples are then provided to a correlator subsystem 22 that provides an early representation of the locally generated composite pilot code generated by the composite code generator 24. Correlation measurements are made by multiplying the samples by accelerated and delayed or early minus late versions. The operation of the composite code generator and correlator subsystem is described below with reference to FIGS. 4 and 5, respectively. I and Q correlation measurements associated with early, accelerated and late or early minus late versions of the local composite pilot code accumulate (accumulate) each I and Q measurement separately over a given interval To the integration and dump circuit 26. At the end of each interval, the integration and dump circuit 26 provides the controller 40 with the results of the respective I and Q accumulated values, ie, the I and Q correlation signals. The controller then controls the carrier NCO 30 and the composite code generator 24 to align the locally generated composite pilot code with the corresponding code in the received signal.

  The GNSS receiver 10 tracks the AltBOC (15, 10) signal using a locally generated composite pilot code generated from locally generated real and imaginary composite signal components. The operations performed by the composite code generator 24 to generate the locally generated composite pilot code component will now be described in detail with reference to FIG.

Composite code generator 24 includes _{c 3} and _c 4 PRN code generators 242 and 243 generate local versions of the E5a and E5b · PRN code for a given GNSS satellite, respectively. The code generator 24 further includes two square wave generators 244 and 245 that generate cr and sr values corresponding to the real and imaginary components of the upper and lower carriers er (t) and er ^* (t). . As described in more detail below with reference to FIG. . . The correlation signal is used to control the transitions of the cr and sr square waves that can be considered as respective code patterns, and the relative timing of the locally generated c ₃ and c ₄ code chips.

である。以下に一層詳細に説明するように、相関器サブシステム２２は、受信された複合パイロット・コードを、局部的に発生された複合パイロット・コードで乗算する。その結果に基づいて、制御器４０は、ＰＲＮコード及び方形波発生器２４２−２４５を調整して、局部コードを受信されたコードに整列させる。 Composite code generator 24, _{c 3} and _{c 4} code chips and added together in the adder 240, the multiplier 246 the additional value, the sign (cos (πf _s t)) multiplied by the value of cr is The real component of the locally generated composite pilot code is then generated. The real component of the composite pilot code is hereinafter referred to as “I _pilot ”. Composite code generator converts the c ₃ code chips in an inverter 248, the converted c ₃ code chips added in adder 250 to the corresponding c ₄ code chips, and the multiplier results 252 in by multiplying by sr is the sign _{(sin (πf s t))} , and generates the imaginary component of the composite pilot code. The imaginary component of the locally generated composite pilot code is hereinafter referred to as “Q _pilot ”. The local replica of the composite pilot code then totals

It is. As will be described in more detail below, correlator subsystem 22 multiplies the received composite pilot code by a locally generated composite pilot code. Based on the result, the controller 40 adjusts the PRN code and square wave generator 242-245 to align the local code with the received code.

  Now, with reference to FIG. 5, the operation of the correlator subsystem 22 will be described with respect to the operation associated with the enhanced version of the locally generated composite pilot code. The receiver operates in a known manner to generate an associated DLL error signal, operates as a part of a delay lock loop or DLL, and early and late or early minus of locally generated composite pilot codes. Includes similar circuitry for later versions.

項を乗算して開いて実数成分と虚数成分を分けると以下の相関信号が生成される。

Correlation subsystem 22 multiplies together two complex signals: a locally generated composite pilot code and a received composite signal. The correlator subsystem thus performs the following operations.

When the term component is multiplied and opened to separate the real component and the imaginary component, the following correlation signal is generated.

As shown in FIG. 5, the correlation subsystem includes _baseband signals I _baseband and Q _baseband provided by Doppler cancellation processor 20 and locally generated real and imaginary signal components I _pilot provided by code generator 24. And Q _pilot to generate the real and imaginary components of the correlation signal. Correlation subsystem 22 multiplies the I _baseband signal by the I _pilot signal at multiplier 502 and multiplies the Q _baseband signal by the Q _pilot signal at multiplier 510. Adder 506 then adds the two products together and provides the result to integration and dump circuit 516. The integration and dump circuit 516 accumulates the sum generated by the adder 506 and generates a corresponding real component, i _prompt , signal at the appropriate time. To generate the imaginary component, the correlation subsystem multiplies the Q _baseband signal by the I _pilot signal at multiplier 508 and multiplies the I _baseband signal by the Q _pilot signal at multiplier 504. The product generated in multiplier 504 is converted by inverter 512 and added to the product generated by multiplier 508 in adder 514. Adder 514 then provides the sum to integration and dump circuit 518, which accumulates the sum and generates a corresponding Q _prompt signal at the appropriate time.

Controller 40 (FIG. 3) processes the I _prompt and Q _prompt signals to determine the center carrier tracking phase error as the arctangent of Q _prompt / I _prompt . The phase error signal is then used in a known manner to control the carrier NCO 30 that controls the Doppler removal processor 20.

  As described above, the controller 40 also receives early and delayed or early minus delayed I and Q correlation signals. Based on these signals, controller 40 adjusts generators 242-245 to align the local composite code in the received code, thus minimizing the associated DLL error signal.

Section 2. Data Recovery from Composite In-phase Signal The AltBOC (15,10) signal contains both data and spreading codes on the E5a and E5b in-phase channels. The E5a in-phase channel carries data transmitted at a specific data rate, and the E5b in-phase channel carries different data transmitted at different data rates. Data transitions on the E5a and E5b in-phase channels, however, occur at corresponding times. The GNSS receiver 10 acquires and tracks the AltBOC (15, 10) signal using the composite pilot code as described above. After carrier removal, the receiver recovers data from the composite in-phase signal using a separate set of correlators, as described in more detail below with reference to FIG.

であり、ここに、副搬送波は、上式中で、対応の矩形関数ｃｒ（ｔ）±ｓｒ（ｔ）の代わりにシヌソイド（正弦曲線）として含まれており、さしあたり、ｃ_１（ｔ）及びｃ_２（ｔ）にはデータが無い（are free of data）と仮定している。上式中の項

は、上述のセクション１で述べた直角位相拡散、すなわちＰＲＮ、コードに対する表現と同様であるということに留意されたし。 An AltBOC (15,10) complex in-phase baseband signal, ie, a signal with a quadrature pilot code set to zero is

Where the sub-carrier is included in the above equation as a sinusoid instead of the corresponding rectangular function cr (t) ± sr (t), and for the time being c ₁ (t) and It is assumed that c ₂ (t) has no data (are free of data). The term in the above formula

Note that is similar to the representation for quadrature spreading, ie PRN, code, described in section 1 above.

であり、乗算項は、

であり、ここに、Ｒ_ｋは、信号ｋに対する自己相関関数を表す。クロス項は、Ｅ５ａ及びＥ５ｂ同相拡散コードが特にクロス相関値を有するよう設計されているので、事前検知間隔に渡ってフィルタリングされる。クロス項を除去し、複素指数関数を展開すれば、

となる。 If the baseband in-phase signal is correlated with a local replica of the composite in-phase spreading code, the result is

And the multiplication term is

Where R _k represents the autocorrelation function for signal k. The cross terms are filtered over the pre-detection interval because the E5a and E5b in-phase spreading codes are specifically designed to have cross correlation values. If we remove the cross term and expand the complex exponential function,

It becomes.

対応の相関関数が図６に示されている。これは、図２に示されている複合直角位相パイロット・コードに対する相関関数と同様であることに留意されたし。 In the case of a signal with no data (data free), the autocorrelation functions R ₁ and R ₂ are equal and the representation is simplified as follows:

The corresponding correlation function is shown in FIG. Note that this is similar to the correlation function for the composite quadrature pilot code shown in FIG.

If the assumption that there is no data in the E5a and E5b in-phase codes c ₁ (t) and c ₂ (t) is removed, the maximum normalized ideal of the individual correlation functions R ₁ and R ₂ The values and their sums and differences are shown in the table of FIG. Thus, the original data sequence can be recovered when the receiver recovers both (R ₁ + R ₂ ) and (R ₂ −R ₁ ) from the composite in-phase signal.

は、複合直角成分チャンネル信号の自己相関関数と同様である自己相関関数を有する。しかしながら、図８に示されるように、同相複合信号のＲ_１−Ｒ_２信号成分

は、同様の自己相関関数を有しておらず、その理由は、τがゼロに行くとき正弦（ｓｉｎ）項がゼロに行くからである。 The receiver may recover (R ₁ + R ₂ ) data directly from the in-phase I _prompt signal. However, the recovery of (R ₂ -R ₁ ) data is not as direct (straight forward). As shown in FIG. 6, the R ₁ + R ₂ component of the in-phase composite signal, ie,

Has an autocorrelation function that is similar to the autocorrelation function of the composite quadrature component channel signal. However, as shown in FIG. 8, the R ₁ -R ₂ signal components of the in-phase composite signal

Does not have a similar autocorrelation function because the sine term goes to zero when τ goes to zero.

に対応する相関ピークを追跡し、（Ｒ_２−Ｒ_１）データ・シーケンスは、次に、減少されたパワーでＱ_{ｐｒｏｍｐｔ}相関器から読取られ得る。 To compensate for the sine term that goes to zero when τ goes to zero, the correlation action can be offset so that the action is

The (R ₂ -R ₁ ) data sequence can then be read from the Q _prompt correlator with reduced power.

を生成する信号を発生し得る。局部同相複合信号は、従って、

となる。 Alternatively, a circuit for generating a local version of the composite signal may instead, the autocorrelation function for (R 2 _{-R 1)} term

Can be generated. The local in-phase composite signal is therefore

It becomes.

を生成する。（Ｒ_２―Ｒ_１）データを回収するために、受信器は、局部信号

を生成する。 In order to demodulate data using this method, the receiver has two combinations of c ₁ (t) and c ₂ (t) spreading codes, ie R ₁ + R ₂ data and R ₂ −R ₁ data, respectively. Corresponding bonds are generated locally. In order to collect (R ₁ + R ₂ ) data, the receiver

Is generated. (R ₂ -R ₁ ) To collect data, the receiver

Is generated.

Referring now to FIG. 9, the real and imaginary components of the R ₂ −R ₁ and R ₁ + R ₂ combinations are locally generated by a code generator 54 that can be part of the local composite code generator 24 (FIG. 3). Is done. To generate the real component of the R ₁ + R ₂ combination, the code generator adds the c ₁ and c ₂ codes together in adder 540 and multiplies the result by square wave code cr in multiplier 546. . To generate the imaginary component of the R ₁ + R ₂ combination, the code generator adds the c ₁ code and the inverted c ₂ code together at adder 550 and the result is a square wave at multiplier 562. Multiply by code sr. The generator also multiplies the sum c ₁ + c ₂ generated by the adder 540 by the square wave code sr in the multiplier 560 to generate an imaginary component of the R ₂ -R ₁ combination. The generator further multiplies the total c ₁ -c ₂ generated by the adder 550 by the square wave code cr in the multiplier 522 to generate a real component of the R ₂ -R ₁ code.

The receiver then recovers the data from the composite in-phase code using the locally generated real and imaginary components of the R ₁ + R ₂ and R ₂ −R ₁ combinations.

Now, referring to FIG. 10, the system multiplies the in-phase _baseband signal I _baseband by the real component of R ₁ + R _{2 in} the multiplier 602, and multiplies the quadrature _baseband signal Q _baseband in the multiplier 606 by R ₁ + R. Multiply by ₂ imaginary components. The products generated by multipliers 602 and 606 are then added together in adder 608 and the sum is provided to integration and dump circuit 615. To generate a correlation signal related to R ₁ -R ₂ , the correlation subsystem multiplies the quadrature _baseband signal Q _baseband by the imaginary component of R ₂ -R ₁ in multiplier 610. Further, the system multiplies the real component of the _baseband signal I _{baseband by} the real component of R ₂ -R _{1 in} the multiplier 604 code. The two sums are added together in adder 612 and provided to integration and dump circuit 616. Integration and dump circuits 615 and 616 accumulate the correlation values generated by _summers 608 and 612, respectively, and generate R ₁ + R 2 _prompt and R ₁ −R 2 _prompt signals at the appropriate times. The result is then used to collect data according to the chart of FIG.

  The circuits of FIGS. 5 and 10 can be combined as shown in FIG. 11 to generate a system that tracks the quadrature AltBOC composite code and recovers E5a and E5b data from the in-phase AltBOC composite code.

FIG. 6 shows the frequency spectrum of an AltBOC (15,10) quadrature channel sequence. FIG. 2 shows a normalized autocorrelation function associated with the signal shown in FIG. FIG. 2 is a functional block diagram of one channel for a GNSS receiver. FIG. 4 is a functional block diagram of a local code generator included in the receiver of FIG. 3. FIG. 5 is a functional block diagram of a correlator subsystem included in the receiver of FIG. FIG. 5 shows an autocorrelation function associated with an AltBOC in-phase channel. It is a chart figure of an ideal autocorrelation value. FIG. 6 shows another autocorrelation function associated with an AltBOC in-phase channel. It is a functional block diagram of a local code generator. It is a functional block diagram of a correlator subsystem. FIG. 11 is a functional block diagram combining the correlator subsystems of FIGS. 5 and 10.

Explanation of symbols

10 GNSS receiver 12 Antenna 14 Down converter (down converter)
16 IF Band Filter 18 Analog to Digital Converter 20 Doppler Removal Processor 22 Correlator Subsystem 24 Composite Code Generator 26 Integral and Dump Circuit 30 Carrier Numerically Controlled Oscillator (Carrier NCO)
40 controller

Claims (11)

A receiver for use with a Earth navigation satellite system that transmits an alternating binary offset carrier wave, ie an AltBOC signal,
Station unit generated code (c3, c4) to the upper and lower carrier (er, er *) real and imaginary components of (cr, sr) of the local version of the AltBOC composite code obtained by combining A local composite code generator (24) for generating real and imaginary code components (Ipilot, Qpilot) ;
In the station unit to generated real and imaginary composite code components, baseband in-phase and quadrature components of the received signal (Ibaseband, Qbaseband) by combining products produced by multiplying the locally A correlation subsystem (22, 26) that generates a correlation signal (Iprompt, Qprompt) by correlating the generated composite code with the composite code in the received AltBOC signal;
A controller that adjusts the local composite code generator to align the local composite code with the corresponding composite code in the AltBOC signal received based on the correlation signal, the time when the AltBOC composite code is transmitted; The controller (40) having means for determining a global position based on a timing difference from a time at which the AltBOC composite code is received;
A receiver that is configured to include: The local composite code generator (24)
The upper and lower carrier (er, er *) real and imaginary components of (cr, sr) square wave code generator for generating a a (244, 245),
A first PRN code generator (243) for generating a first code (c3) modulated on the upper carrier (er) ;
A second PRN code generator (242) for generating a second code (c4) modulated on the lower carrier (er *) ;
The real and imaginary components of upper and lower carrier (er, er *) (cr , sr) and the first and second codes (c3, c4) and coupled to a real and imaginary of the local composite code Adders (240, 250) and multipliers (246, 252) for generating components (Ipilot, Qpilot) ;
2. The receiver according to claim 1, comprising: The receiver according to claim 1 or 2, wherein the generated local composite code (Ipilot, Qpilot) corresponds to a dataless composite pilot code . The local composite code generator (24) further includes a first autocorrelation function (R1) associated with the third code (c1) and a second autocorrelation function associated with the fourth code (c2). (R2) and a means for generating a combination (I _{R1 + R2} , Q _{R1 + R2} , I _{R1 − R2} , Q _{R1 − R2} ) respectively corresponding to the sum and difference with
The correlation subsystem (22) further includes means for generating a combination correlation signal ((R1 + R2) prompt, (R1-R2) prompt) corresponding to each combination,
Wherein the controller (40) is further configured with a means for recovering the combined correlation signals or data from the receiver according to any one of claims 1 to 3. One or more of the adders (240, 250), said code (c3, c4) optionally one or more of claims and an inverter (248) is configured for reversing 4 Receiver as described in. The local composite code generator is
A third PRN code generator (542) for generating a third code (c1) modulated on the upper carrier (er);
A fourth PRN code generator (543) for generating a fourth code (c2) modulated on the lower carrier wave (er *);
One or more adders (540, 548, 550) that combine the third code and the fourth code to produce an associated sum;
Each sum is multiplied separately by the first and second square waves, and the real and imaginary components (I _{R1 + R2} , ... ) Of each associated said combined correlation signal ((R1 + R2) prompt, (R1-R2) prompt) One or more multipliers (546, 552, 560, 562) generating Q _{R1 + R2} , I _{R1 - R2} , Q _{R1 - R2} )
Furthermore, the receiver as described in any one of Claim 3 to 5 comprised . A method for determining a global position from an AC binary offset carrier wave, ie, an AltBOC, signal received from a Earth Navigation satellite system, comprising:
Real number of local version of AltBOC composite code obtained by combining the real and imaginary components (cr, sr) of the upper and lower carriers (er, er *) with the locally generated code (c3, c4) Generating imaginary code components (Ipilot, Qpilot) ;
And generating in-phase and quadrature components of the received the AltBOC signal,
By combining the products generated by multiplying the baseband in-phase and quadrature components (Ibaseband, Qbaseband) of the received AltBOC signal with the locally generated real and imaginary composite code components locally Correlating the generated composite code with the composite code in the received AltBOC signal to generate an associated correlation signal (Iprompt, Qprompt) ;
A step of adjusting the local composite code generator, the local composite code are aligned with the corresponding composite code in the received said AltBOC signal based on the correlation signal,
Time and the received by AltBOC composite code is transmitted, determining a global position based on timing differences between times the AltBOC composite code is received,
Having a method. Generating a local version of the AltBOC composite code comprises :
Generating a square wave (cr, sr) that correspond to real and imaginary components of the upper and lower carrier (er, er *),
Generating a first code (c3) modulated on the upper carrier;
Generating a second code (c4) modulated on the lower carrier;
Selectively combine the first and second codes and multiply the result by the real and imaginary components of the upper and lower carriers to generate the real and imaginary components (Ipilot, Qpilot) of the local composite code And steps to
The method according to claim 7 having. The step of selectively coupling the first and second code has a step of generating a first and a second sum of the respectively associated with the real and imaginary components (Ipilot, Qpilot), the first sum of Corresponds to the inverted first code (c3) plus the second code (c4) , the second sum is the addition of the first and second two codes that are not inverted The method of claim 8 corresponding to: 10. The method according to any one of claims 7 to 9, wherein the generated local composite code (Ipilot, Qpilot) corresponds to a dataless composite pilot code . Generating a third code (c1) modulated on the upper carrier;
Generating a fourth code (c2) modulated on the lower carrier;
Bond _R 1 + _{R 2} and the real and imaginary components of the binding _{R 2 -R 1 (I R1 +} R2, Q R1 + R2, I R1 - R2, Q R1 - R2) in order to obtain, using said first and second square-wave Generating two combinations of the third and fourth codes ;
Associating said bonds to obtain a combined correlation signal ((R1 + R2) prompt, (R1-R2) prompt) corresponding to each bond;
Recovering data from the combined correlation signal;
Further comprising
R ₁ is an autocorrelation function associated with the third code;
Wherein R ₂ is The method according to any one of claims 7 the autocorrelation function 9 associated with the fourth code.

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