JP4558548B2 - Microstrip antenna, radio module, radio system, and microstrip antenna control method - Google Patents

Description

  The present invention relates to a microstrip antenna, a radio module, a radio system, and a control method for a microstrip antenna, and more particularly, a microstrip antenna, a radio module, a radio system, and a micro that can handle multiple frequencies and can control directivity. The present invention relates to a strip antenna control method.

  Conventionally, omnidirectional antennas having an isotropic radiation intensity distribution have been used in wireless communications such as satellite communications and mobile communications.

  However, when wireless communication using an omnidirectional antenna is performed in the same area, radio waves generated by other radio waves may become noise sources for other wireless communication. When noise occurs in wireless communication, the S / N (signal to noise ratio) between the amplitude of a signal desired to be received at an arbitrary point and the amplitude of the noise signal at that point decreases, and the transmission error rate increases. The demerit that it will occur. Moreover, since high output transmission is required to maintain communication quality, power consumption must be increased.

  In recent years, radio wave ecology where radio waves are regarded as finite resources has been proposed, and directional antennas that can efficiently use the frequency and power of radio waves have attracted attention. A directional antenna is an antenna that selectively transmits and receives only radio waves in a specific direction.

  By using this directional antenna, radio waves in the same area can be spatially separated, and the communication capacity in the area can be dramatically improved. Moreover, since the antenna gain in the direction of the maximum radiation angle of the beam is larger than that of the omnidirectional antenna, the directivity control antenna can improve reception power and S / N when used on the reception side. Furthermore, if the null point, which is a point where the radiation intensity is low, is directed to the noise source, the reception intensity of the interference wave / interference wave can be reduced, so that the S / N can be improved and the transmission error rate can be reduced. Become. Furthermore, if a directional antenna is used on the transmission side of wireless communication, the antenna gain in the maximum radiation angle direction is large, so that transmission power can be reduced and reduction of power consumption can be expected.

  A normal propagation environment is a multipath fading environment, and a change in electric field strength (fading) of radio waves occurs. In order to fundamentally solve this multipath fading, it is necessary to minimize the propagation path. From this point of view, the directional antenna is regarded as important.

  In particular, at present, a plurality of wireless standards such as PDC, FOMA, CDMA2000, PHS, etc. are used for mobile phones, IEEE802.11a and Bluetooth are used for wireless LANs, and GPS, VICS, ETC, etc. are used simultaneously for ITS. In the future, it is expected that an environment in which multiple standards coexist will continue. Therefore, a multi-band (multi-frequency compatible) directional antenna capable of supporting a plurality of wireless standards with a single device has attracted attention.

  The followings have been proposed as techniques related to such directional antennas that can handle multiple frequencies.

  In Patent Document 1, a dipole antenna is configured by connecting metal pieces with a PIN diode, and an antenna device that controls directivity by changing the resonance length by applying a bias to the PIN diode to switch between conduction and cutoff is proposed. Has been.

  Further, Patent Document 2 proposes an antenna device in which one feeding point and a plurality of ground points are provided in an antenna element, and the directivity is controlled by changing the resonance length by switching the ground point with a corresponding switch. .

  In Patent Document 3, a phase shifter is provided in an RF front-end circuit, and a phased array antenna is used, and the radiation beam pattern of the antenna is electronically variably controlled based on high-precision detection information of the vehicle position. An antenna device has been proposed.

Non-Patent Document 1 proposes an antenna device in which element elements arranged in a matrix are connected by FET switches and excited at one feeding point by a balanced current. In this antenna device, the shape of the radiating element can be changed by the ON / OFF combination of the FET switch, and the frequency and directivity can be switched.
JP 2000-236209 A JP 2002-261533 A JP 2000-156606 A The GTRI Prototype Reconfigurable Aperture Antenna (2003 IEEE Antenna and Propagation Society International Digest, p683-686)

  However, the above technique has the following problems.

  In the antenna device described in Patent Document 1, it is necessary to excite the dipole antenna with a balanced current. However, the line used in the RF circuit often uses an unbalanced current such as a microstrip line or a coplanar line. If it is necessary, a balun must be provided between the antenna and the track. In general, since the balun has a narrow band, it cannot cope with a plurality of frequencies, and one balun is required for each frequency. Therefore, in order to cope with multiband, it is necessary to arrange baluns in the vicinity of the feeding point of the dipole antenna by the number of multibands, and the number of multibands is limited by the installation area of the baluns. Therefore, it can cope with a low frequency band such as a dual band, but cannot deal with a multiband with a large frequency band.

  Further, in the antenna device described in Patent Document 2, since the short-circuit point is switched by the switch, the input impedance of the antenna changes at each frequency. Therefore, the ground point can be moved only within a matching range, and the frequency range that can be varied in multiband cannot be increased. Actually, the variable frequency band disclosed in Patent Document 2 is 1.55 to 2.2 GHz, which is as small as 30% with respect to the center frequency of 1.8 GHz. For this reason, it is possible to cope with a case where a relatively close frequency band such as a cellular phone is used, but not to use a 2.4 GHz band and a 5 GHz band as in a wireless LAN.

  Further, the antenna devices described in Patent Document 1 and Patent Document 2 have no description of directivity control, and it is considered that directivity control cannot be performed.

  In addition, the antenna device described in Patent Document 3 has not proposed any method for multi-frequency (corresponding to multi-frequency) of one phased array antenna, so that directivity control and multi-frequency can be realized at the same time. It is necessary to prepare the required number of fuzzed array antennas for the required number of bands. For this reason, the cost becomes high, and it is not realistic in a wireless LAN or the like.

  Further, in the antenna device of Non-Patent Document 1, the operation circuit becomes complicated because the ON / OFF of the FET switch is selected using a genetic algorithm. In addition, since there is only one feeding point and the input impedance of the radiating element needs to be matched with the characteristic impedance of the feeding line, the frequency range in which the degree of freedom of the shape of the radiating element can be varied is limited. Actually, the variable frequency range reported in the above-mentioned document was about 1 to 2 GHz, which was enough to accommodate a bandwidth of ± 50%. In addition, since a balun is required to connect to a microstrip line or coplanar line widely used in an RF circuit because excitation is performed with a balanced current, the number of multibands is limited by the installation area of the balun. There is also.

  The present invention has been made in view of the above-described problems, and can cope with multiple frequencies and can control directivity, and further has a wide corresponding variable frequency band. An object of the present invention is to provide a radio system and a method for controlling a microstrip antenna.

The microstrip antenna according to the present invention is
  With the main plate,
  A feed element disposed above the ground plane;
  A first parasitic element arranged in a matrix around the feeding element;
  A conductor connecting each element of the feeding element and the first parasitic element;
  First switch means provided between the ground plate and the first parasitic element, for switching whether to short-circuit the first parasitic element to the ground plate,
  Corresponding to high frequency by resonating the feeding element alone,
  Corresponding to low frequency by resonating the first radiating element composed of the feeding element and the first parasitic element arranged in a matrix,
  The antenna directivity is controlled by switching the first switch means at the time of the high frequency response.

The microstrip antenna according to the present invention is
  With the main plate,
  A feed element disposed above the ground plane;
  A first parasitic element arranged in a matrix around the feeding element;
  A plurality of second parasitic elements arranged in a matrix around the first parasitic element arranged in the matrix; and
  A conductor connecting each of the feeding element, the first parasitic element, and the second parasitic element; and
  A first switch means provided between the ground plane and the first parasitic element, for switching whether or not the first parasitic element is short-circuited to the ground plane;
  Second switch means provided between the ground plane and the second parasitic element, and switching whether to short-circuit the second parasitic element to the ground plane,
  The directivity of the antenna is controlled at a plurality of frequencies by switching the first switch means and the second switch means.

The method of controlling the microstrip antenna according to the present invention includes:
  With the main plate,
  A feed element disposed above the ground plane;
  A first parasitic element arranged in a matrix around the feeding element;
  A conductor connecting each element of the feeding element and the first parasitic element;
  A method of controlling a microstrip antenna, comprising: first switch means provided between the ground plane and the first parasitic element, for switching whether or not the first parasitic element is short-circuited to the ground plane. Because
  Resonating the power feeding element alone to cope with high frequency,
  Resonating the first radiating element composed of the feeding element and the first parasitic element arranged in a matrix to correspond to a low frequency,
  The antenna directivity is controlled by switching the first switch means at the time of the high frequency response.

The method of controlling the microstrip antenna according to the present invention includes:
  With the main plate,
  A feed element disposed above the ground plane;
  A first parasitic element arranged in a matrix around the feeding element;
  A conductor connecting each element of the feeding element and the first parasitic element;
  A first switch means provided between the ground plane and the first parasitic element, for switching whether or not the first parasitic element is short-circuited to the ground plane;
  A parasitic second radiating element that is arranged in a matrix around the first radiating element and resonates at the same frequency as the first radiating element;
  And a second switch unit that is provided between the ground plane and the second radiating element and switches between whether or not the second radiating element is short-circuited to the ground plane. And
  Resonating the power feeding element alone to cope with high frequency,
  Resonating the first radiating element composed of the feeding element and the first parasitic element arranged in a matrix to correspond to a low frequency,
  At the time of the high frequency response, the directivity of the antenna is controlled by switching the first switch means,
  The antenna directivity is controlled by switching the second switch means when the low frequency is supported.

The method of controlling the microstrip antenna according to the present invention includes:
  With the main plate,
  A feed element disposed above the ground plane;
  A first parasitic element arranged in a matrix around the feeding element;
  A plurality of second parasitic elements arranged in a matrix around the first parasitic element arranged in the matrix; and
  A conductor connecting each of the feeding element, the first parasitic element, and the second parasitic element; and
  A first switch means provided between the ground plane and the first parasitic element, for switching whether or not the first parasitic element is short-circuited to the ground plane;
  A control method for a microstrip antenna, comprising: a second switch means provided between the ground plane and the second parasitic element, for switching whether or not the second parasitic element is short-circuited to the ground plane. Because
  The directivity of the antenna is controlled at a plurality of frequencies by switching the first switch means and the second switch means.

  According to the present invention, the microstrip antenna can be adapted to multiple frequencies. Also, directivity can be controlled in the microstrip antenna. Furthermore, the corresponding variable frequency band can be increased.

  Embodiments according to the present invention will be described below. In the following description, using the coordinate system shown in FIG. 18, the position in the space is represented by orthogonal coordinates with three axes of XYZ, the rotation angle from the X axis to the Y axis is φ, and the Z axis is the XY plane. The rotation angle to is expressed as θ. The positive direction of the Z axis represents the zenith direction, and the XY plane represents the horizontal plane. The X axis represents an axis perpendicular to the excitation direction of the feed element (H plane direction), and the Y axis represents an axis parallel to the excitation direction of the feed element (E plane direction).

<First Embodiment>
The microstrip antenna (MSA) of the first embodiment will be described. The MSA of the present embodiment is a two-frequency antenna that operates in two modes of low frequency and high frequency, and can switch directivity in the high frequency mode.

  1 and 2 show the configuration of the MSA of this embodiment. 1 is a top view of the MSA, and FIG. 2 is a cross-sectional view of the MSA.

  The MSA includes a ground plane 1, a dielectric 2, a feeding element 3, a first parasitic element 4, a minute conductor 5, a feeding point 6, a first switch 7, a first substrate 8, It comprises a matching circuit 9 and a feeder line 10.

  A ground plane 1 is provided on the first substrate 8, and a dielectric 2 is provided on the ground plane 1. On top of the dielectric 2, a rectangular feed element 3 and a first parasitic element 4 arranged in a matrix with the feed element 3 as the center are stacked. The parasitic element 4 has the same shape as the rectangular feeding element 3. Further, between the adjacent feeding element 3 and the first parasitic element 4 and between the adjacent first parasitic elements 4 are connected by a minute conductor 5 having a width shorter than the width of the opposing sides. Yes.

  The feeding element 3 is provided with a feeding point 6. The feed point 6 is connected to the matching circuit 9 via a via hole 19 penetrating the dielectric 2, the ground plane 1, and the first substrate 8 and the feed line 10. Note that the feeder line 10 is formed of a microstrip line.

  The first parasitic element 4 is provided with a first switch 7. The first switch 7 switches between a conduction / cutoff state between the first parasitic element 4 and the ground plane 1. When the first switch 7 is in a conductive state, the first parasitic element 4 is short-circuited with the ground plane 1, and when the first switch 7 is in a disconnected state, the first parasitic element 4 is released from the ground plane. The first switch 7 is switched via a bias line.

  In the following description, a radiation system including the feed element 3, the first parasitic element 4, and the minute conductor 5 is referred to as “first radiation element 20”. The first radiating element 20 corresponds to a portion surrounded by a dotted line in FIG.

  In the MSA of the present embodiment, the resonance mode of the first radiating element 20 includes a resonance mode in which the length L1 of the feeding element 3 in the Y direction is approximately a half wavelength, and the length of the first radiating element 20 in the Y direction. There are two types of resonance modes, the length L2 of which is approximately a half wavelength. This will be specifically described below.

  In the MSA of the present embodiment, when all the first switches 7 of the first parasitic element 4 of the first radiating element 20 are shut off, the length L2 of the first radiating element 20 in the Y direction is approximately set. The resonance mode is a half wavelength. Here, this resonance mode is referred to as a “low frequency side resonance mode”. Actually, the resonance frequency shifts by several tens of percent to the low frequency side because the current path of the standing wave generated in the first radiating element 20 becomes longer by the minute conductor 5.

  When an RF signal (radio frequency signal) coinciding with the resonance mode on the low frequency side is given to the power feeding element 3, it resonates at low frequency. Therefore, it becomes possible to make the 1st radiation element 20 respond | correspond to a low frequency.

  On the other hand, when an RF signal having a half-wavelength L1 in the Y direction of the feed element 3 is applied to the feed element 3, the RF signal is reflected by the side of the feed element 3 in the X direction, and L1 has a half wavelength. Standing waves are generated. This resonance mode is referred to as a “high-frequency side resonance mode”. The actual resonance frequency is shifted depending on the width of the minute conductor 5 connected to the power feeding element 3.

  At this time, in the first parasitic element connected to the feed element 3, a current having the same frequency as that of the feed element 3 is excited by mutual coupling between the feed element 3 and the first radiating element 20 is It becomes possible to cope with high frequency. When all the first switches 7 are shut off, that is, when the first parasitic element 4 is opened from the ground plane, the radiation pattern is the radiation of the feeding element 3 and the eight first parasitic elements 4. Since the pattern is a composite shape, the beam is directed toward the zenith.

  Next, a directivity control method in the resonance mode on the high frequency side will be described. The directivity control is performed by switching the short-circuit state of the eight first power supply elements 5 surrounding the power supply element 3 in the first radiating element 20, in other words, switching the conduction / cut-off state of the first switch 7. Made in

  For example, in the resonance mode on the high frequency side, only one parasitic element 5 is released from the ground plane 1 and the other seven first parasitic elements 4 are short-circuited to the ground plane 1. When the combination of conduction / interruption is selected, the currents of the seven first parasitic elements 4 short-circuited to the ground plane 1 are greatly reduced, but the power is fed to the first parasitic element 4 opened from the ground plane 1. A large current flows through the element 3.

  Therefore, the radiation pattern in the first radiating element 20 is greatly influenced by the parasitic element 5 opened from the grounding plate 1 and the parasitic element 5 opened from the ground plane 1. The beam is emitted in a tilted form.

  As described above, in the MSA of the present embodiment, the first radiating element 20 has two resonance modes of the low frequency side resonance mode and the high frequency side resonance mode. It becomes possible. In the resonance mode on the high frequency side, it is possible to perform beam switching, that is, directivity control, depending on the short-circuit state of the first parasitic element 4.

  In the MSA of this embodiment, the ground plate 1 is a Cu plate, the dielectric 2 is a Teflon (registered trademark) glass fiber substrate having a relative dielectric constant of 2.2, the feeding element 3 is a 4.4 mm square Cu plate, The parasitic element 4 is a 4.4 mm square Cu plate, the minute conductor 5 is a 800 μm wide Cu plate, and the first parasitic element surrounding the feeding element 3 in a matrix is disposed at intervals of 0.4 mm. When arranged, the beam can be tilted by 20 to 35 ° in the θ direction at φ = 0, 45, 90, 135, 180, 225, 270, 315 °, etc., and beam switching was realized.

  Note that the input impedance of the MSA includes an input impedance Ra at the antenna end and an input impedance Rin at inset power feeding. This value can be calculated from the following equation.

Each parameter of the above equation is as follows. FIG. 3 shows what each parameter suggests.
L: Length of side in resonance direction W: Length of side perpendicular to resonance direction L ′: Inset position of feeding point εr: Dielectric constant of dielectric

From the above formula, it can be seen that the input impedance Ra at the antenna end is proportional to (L / W) ² , and the input impedance Rin at the inset power supply varies depending on the value of (L ′ / L).

  The above formula can also be applied to the MSA of this embodiment.

  In the MSA of the present embodiment, (L ′ / L) differs greatly between the resonance mode on the high frequency side that resonates with the power feeding element 3 alone and the resonance mode on the low frequency side that resonates with the first radiation element 20. The input impedance is not the same value. Therefore, when the low-frequency resonance mode is executed by matching the input impedance of the feed element 3 with the characteristic impedance (50Ω) of the feed line 10, the input impedance of the antenna greatly deviates from the characteristic impedance of the feed line 10, so that reflection is large. turn into.

  Therefore, in the MSA of this embodiment, the matching circuit 9 is provided to solve the reflection problem. Hereinafter, the matching circuit will be described.

  The configuration of the matching circuit 9 will be described with reference to FIG. The matching circuit 9 is composed of a circuit in which a spiral inductor 11 and a capacitor 12 are combined in a π type. Note that the changeover switch 13 switches the connection line between the matching circuit 9 and the 50Ω feed line 10.

  Next, switching control between the matching circuit 9 and the feeder line 10 will be described.

  For example, when the feed point 6 is arranged so that the input impedance of the feed element 3 is 50Ω, in the resonance mode on the high frequency side, the changeover switch 13 is placed on the 50Ω feed line 10 side to directly connect the antenna and the RF circuit. As a result, in the resonance mode on the high frequency side, even when the input impedance of the antenna is switched to directivity, the input impedance of the feed element 3 is almost the same value, so that reflection can be suppressed and good transmission / reception can be performed. Become.

  On the other hand, in the resonance mode on the low frequency side, the selector switch 13 is placed on the matching circuit 9 side to connect the antenna and the RF circuit. At this time, by appropriately selecting the L and C constants of the matching circuit 9, the input impedance of the first radiating element 20 can be matched to 50Ω, so that reflection is suppressed even in the resonance mode on the low frequency side. And good transmission / reception can be performed.

  By providing the matching circuit 9 in the MSA in this manner, reflection can be suppressed and good transmission / reception can be performed at both frequencies.

  The matching circuit 9 has a π-type circuit configuration, but is not limited to the π-type, and may have a T-type or L-type circuit configuration. Further, not only a spiral inductor and a capacitor but also a combination of a λ / 4 transformer and a phase shifter may be used.

  In addition, when using a matching circuit that can change the constant, matching is possible by directly connecting the matching circuit to the antenna and appropriately selecting the constant according to the input impedance of the antenna on the high frequency side / low frequency side, Good transmission / reception can be performed.

  In the MSA of the present embodiment, a PIN diode can be used for the first switch 7. Since the PIN diode can be obtained at a low price and can satisfactorily block high-frequency signals up to 20 GHz, a directivity control antenna corresponding to 2.4 GHz or 5 GHz wireless LAN or a portable terminal can be configured.

  The PIN diode is partly cut off the ground plane 1 on the back surface of the dielectric 2 and mounted in the cut area. At this time, one end of the PIN diode is connected to the ground plane 1 and the other end is connected to the first parasitic element 4 through a via hole penetrating the dielectric 2. The bias line for controlling the PIN diode is wired on the first substrate 8 and connected to the PIN diode by a via hole penetrating the first substrate 8.

  A MEMS switch can also be used as the first switch 7. The MEMS switch can satisfactorily cut off high-frequency signals up to about 100 GHz and has a small insertion loss, so that it constitutes a directivity control antenna for two frequencies for high frequencies such as submillimeter waves to millimeter waves. be able to.

  The MEMS switch is mounted on the lower surface of the dielectric 2 in the same manner as the PIN diode described above. Further, after forming the MEMS switch on the first substrate 8 by a bulk micromachine process or a surface micromachine process, the first substrate 8 is attached to the lower surface of the dielectric 2, and one end of the MEMS switch is connected to the ground plane 1. The other end can also be mounted by connecting to the first parasitic element 4 through a via hole penetrating the dielectric 2. When such a method is used, a bias line for controlling the MEMS switch can be wired on the first substrate 8.

  In the present embodiment, the feeding element 3 and the first parasitic element 4 have the same shape. However, when the feeding element 3 alone resonates, the first parasitic element 4 is excited by mutual coupling between the elements. As long as it is possible, it is not necessary to have the same shape, and it is not necessary to have the same shape.

<Second Embodiment>
A microstrip antenna (MSA) according to a second embodiment will be described.

  In the MSA according to the first embodiment, the adjacent feeding element 3 and the first parasitic element 4 and the adjacent first parasitic elements 4 are connected by one minute conductor 5. Large voids are generated inside. Therefore, in the resonance mode on the low frequency side that resonates with the first radiating element 20, the bandwidth of the air gap is reduced due to the air gap.

  It is desirable to eliminate the air gap in order to improve the above bandwidth problem. However, if the air gap is completely eliminated, the first radiating element 20 becomes a pure square, and the current is reflected by the sides of the power feeding element 3. Therefore, the resonance mode on the high frequency side cannot be obtained.

  Therefore, in the MSA of the present embodiment, as shown in FIG. 5, there are a plurality of gaps between the adjacent parasitic elements 3 and the first parasitic elements 4 and between the adjacent first parasitic elements 4 (in the figure). Two) the small conductors 5 are connected.

  With this configuration, in the resonance mode on the low frequency side, the gap inside the first radiating element 20 is reduced and the bandwidth is improved. Further, in the resonance mode on the high frequency side, since the side of the feed element 3 exists inside the first radiating element 20, a relatively large standing wave can be obtained, and the minute conductor 5 is compared with one MSA. Thus, a decrease in antenna gain in the resonance mode on the high frequency side can be reduced.

  A specific comparative example will be described. 3. Ground plate 1 made of a Cu layer, dielectric 2 made of a Teflon (registered trademark) glass fiber substrate having a relative dielectric constant of 2.2, a 4.4 mm square feed element 3 made of a Cu layer, and a Cu layer made of 4. The first parasitic elements 4 each having a width of 4 mm and the minute conductor 5 made of Cu having a width of 800 μm and the first parasitic elements surrounding the feeder elements 3 in a matrix are arranged at intervals of 0.4 mm. In the MSA, the bandwidth in the resonance mode on the low frequency side of the MSA (MSA of the first embodiment / FIG. 1) having one microconductor 5 between the elements, and two microconductors 5 between the elements When comparing the bandwidth in the resonance mode on the low frequency side of a certain MSA (MSA of the present embodiment / FIG. 5), a comparison result that the latter bandwidth is about 15% larger was obtained.

  In the MSA of the present embodiment, the number and width of the minute conductors 5 are determined according to the required antenna bandwidth, antenna gain, and the like.

<Third Embodiment>
A microstrip antenna (MSA) according to a third embodiment will be described.

  In the MSA of the first embodiment, the tilt angle of the beam in the resonance mode on the high frequency side is almost determined by the feed element 3 and the first parasitic element 4 opened from the ground plane 1. That is, it is strongly influenced by the pitch interval between the feed element 3 and the first parasitic element 4. Further, since the resonance frequency of the resonance mode on the low frequency side is substantially determined by L2 of the first radiating element 20, this is also strongly influenced by the pitch interval between the feeding element 3 and the first parasitic element 4.

  Therefore, it is difficult to independently control the tilt angle of the beam and the resonance frequency in the resonance mode on the low frequency side in the resonance mode on the high frequency side.

  Therefore, in the MSA of the present embodiment, as shown in FIG. 6, a microconductor that is bent between adjacent feeding elements 3 and first parasitic elements 4 and between adjacent first parasitic elements 4. Connect with 5. In FIG. 6, micro conductors bent at right angles at two locations are used.

  With this configuration, the current path of the first radiating element 20 can be lengthened while the pitch interval between the feeding element 3 and the first parasitic element 4 is kept constant. The resonance frequency on the low frequency side can be lowered without changing the tilt angle of the beam. That is, the tilt angle of the beam in the resonance mode on the high frequency side and the resonance frequency in the resonance mode on the low frequency side can be controlled almost independently.

  A specific comparative example will be described. 3. Ground plate 1 made of a Cu layer, dielectric 2 made of a Teflon (registered trademark) glass fiber substrate having a relative dielectric constant of 2.2, a 4.4 mm square feed element 3 made of a Cu layer, and a Cu layer made of 4. The first parasitic elements 4 each having a width of 4 mm and the minute conductor 5 made of Cu having a width of 800 μm and the first parasitic elements surrounding the feeder elements 3 in a matrix are arranged at intervals of 0.4 mm. In the MSA, an MSA having one microconductor 5 (MSA of the first embodiment / FIG. 1) and an MSA having three microconductors 5 bent at right angles at two locations (MSA / FIG. Of the present embodiment) In comparison with 6), the comparison result was obtained that the resonance frequency on the lower frequency side of the latter MSA was lower by about 15%.

  The bent shape of the minute conductor 5 is not limited, and the current path of the first radiating element 20 can be lengthened while the pitch interval between the feeding element 3 and the first parasitic element 4 is kept constant. Any shape can be used.

<Fourth Embodiment>
A microstrip antenna (MSA) according to a fourth embodiment will be described. 7 and 8 show the configuration of the MSA of this embodiment. 7 is a top view of the MSA, and FIG. 8 is a cross-sectional view of the MSA.

  The MSA of the present embodiment includes a first radiating element 20 of the MSA of the first embodiment and an unpowered “second radiating element 21” that resonates at substantially the same frequency as the first radiating element 20. The second radiating elements 21 are arranged in a matrix with the first radiating elements 20 as the center. The second radiating element 21 is composed of a single element. Further, the second radiating element 21 is provided with a second switch 14 connected to the ground plane 1, and switching control of the conduction / cutoff state of the second switch 14 is performed via a bias line.

  In the MSA of the present embodiment, when all the first switches 7 that short-circuit the first parasitic element 4 of the first radiating element 20 are shut off, the length L2 of the first radiating element 20 in the Y direction is set to A resonance mode on the low frequency side having a half wavelength is generated. Therefore, the RF element corresponding to the resonance mode on the low frequency side can be provided to the feeding element 3 to realize low frequency response.

  Further, in the MSA of the present embodiment, the first radiating element 20 is surrounded by the second radiating element 21 having substantially the same resonance frequency as that of the first radiating element 20, and therefore, the first radiating element 20 in the resonance mode on the low frequency side. Due to the mutual coupling between the radiating element 20 and the second radiating element 21, a current having the same resonance frequency (resonance mode on the low frequency side) as that of the first radiating element 20 is induced in the second radiating element 21. As a result, the resonance mode radiation pattern on the low frequency side is a combined form of the first radiation element 20 and the second radiation element 21. Since the second radiating element 21 is provided with the second switch 14 that is short-circuited with the ground plane 1, one of the pair of second radiating elements 21 arranged symmetrically with respect to the first radiating element 20 is selected. By short-circuiting with the ground plane 1, the radiation pattern can be tilted.

  On the other hand, when the RF signal having the half length of the length L1 in the Y direction of the feed element 3 is given to the MSA of the present embodiment to the feed element 3, the RF signal is substantially reflected on the side of the feed element 3 in the X direction. Therefore, a resonance mode on the high frequency side can be realized. In this resonance mode, since the feed element 3 is surrounded by the eight first parasitic elements 4, the first switch 7 controls the short-circuiting of the first parasitic element 4 to control the high frequency side. Directivity switching in the resonance mode can be realized.

  Note that the second radiating element 21 and the feed element 3 have greatly different pitches and resonance frequencies, and the mutual coupling between the elements is small. For this reason, since the second radiating element 21 hardly has a resonance mode on the high frequency side, the influence of the second radiating element 21 on the radiation pattern is very small.

  Further, when the second radiating element 21 is short-circuited to the ground plane 1 by the second switch 14, the current of the second radiating element 21 is further suppressed. Therefore, by performing the short circuit control of the second radiating element 21, it is possible to further reduce the influence of the second radiating element 21 on the directivity switching of the resonance mode on the high frequency side.

  As described above, in the MSA of the present embodiment, the first radiating element 20 in which the feeding element 3 and the first parasitic element 4 are connected by the minute conductor 5 and integrated has two resonance modes. High frequency 2 frequency response can be realized. In the resonance mode on the high frequency side, the switching of the beam, that is, the directivity can be controlled by the combination of conduction / cutoff of the eight first switches 7. Even in the resonance mode on the low frequency side, beam switching, that is, directivity control can be performed by the combination of conduction / cutoff of the eight second switches 14.

  In the MSA of this embodiment, the ground plate 1 is a Cu plate, the dielectric 2 is a Teflon (registered trademark) glass fiber substrate having a relative dielectric constant of 2.2, the feeding element 3 is a 4.4 mm square Cu plate, The parasitic element 4 is formed of a 4.4 mm square Cu plate, the minute conductor 5 is formed of a 800 μm wide Cu plate, and the first parasitic elements 4 are arranged at a pitch of 4.8 mm. Further, a second radiating element 21 of 14.0 mm square (the same size as the first radiating element 20) is arranged around the first radiating element 20 with an interval width of 1.0 mm. In this case, depending on the conduction / cutoff combination of the second switch 14, the beam is directed 20 to 30 in the θ direction at φ = 0, 45, 90, 135, 180, 225, 270, 315 °, etc. ° Can be tilted and beam switching is realized Came. Further, when all the second switches 14 are turned on and the second radiating element 21 is short-circuited with the ground plane 1, φ = 0, 45, 90, 135, depending on the combination of conduction / cutoff of the first switch 7. The beam can be tilted by 20 to 35 ° in the θ direction at 180, 225, 270, 315 °, etc., and beam switching can be realized even in the resonance mode on the high frequency side.

  Note that a PIN diode can be used for the second switch 14. Since the PIN diode can be obtained at a low price and can satisfactorily block high-frequency signals up to 20 GHz, a directivity control antenna corresponding to 2.4 GHz or 5 GHz wireless LAN or a portable terminal can be configured.

  The PIN diode is partly cut off the ground plane 1 on the back surface of the dielectric 2 and mounted in the cut area. At this time, one end of the PIN diode is connected to the ground plane 1 and the other end is connected to the second radiating element 21 through a via hole penetrating the dielectric 2. The bias line for controlling the PIN diode is wired on the first substrate 8 and connected to the PIN diode by a via hole penetrating the first substrate 8.

  A MEMS switch can also be used as the second switch 14. The MEMS switch can satisfactorily cut off high-frequency signals up to about 100 GHz and has a small insertion loss, so that it constitutes a directivity control antenna for two frequencies for high frequencies such as submillimeter waves to millimeter waves. be able to.

  The MEMS switch is mounted on the lower surface of the dielectric 2 in the same manner as the PIN diode described above. Further, after forming the MEMS switch on the first substrate 8 by a bulk micromachine process or a surface micromachine process, the first substrate 8 is attached to the lower surface of the dielectric 2, and one end of the MEMS switch is connected to the ground plane 1. The other end can be mounted by connecting to the second radiating element 21 through a via hole penetrating the dielectric 2. When such a method is used, a bias line for controlling the MEMS switch can be wired on the first substrate 8.

  In the above description, the shape of the second radiating element 21 is 14.0 mm square. However, the current is efficiently induced by the mutual coupling between the first radiating element 20 and the first radiating element 20. The present invention is not limited to this as long as it has substantially the same resonance frequency as the element 20.

  In the above description, the feeding element 3 and the first parasitic element 4 have the same shape. However, when the feeding element 3 alone resonates, the first parasitic element 4 is As long as they can be excited, they do not have to be the same, and may have substantially the same shape.

  Further, in this embodiment, there is one microconductor 5 that connects each element of the first radiating element 20, but the microconductor 5 that connects each element as in the MSA of the second embodiment. May be plural. With this configuration, the bandwidth in the resonance mode on the low frequency side can be improved. In addition, the microconductor 5 may be bent as in the third embodiment. With this configuration, the tilt angle of the high frequency beam and the resonance frequency on the low frequency side can be controlled almost independently.

<Fifth Embodiment>
A microstrip antenna (MSA) according to a fifth embodiment will be described. 9 and 10 show the configuration of the MSA of this embodiment. FIG. 9 is a top view of the MSA, and FIG. 10 is a cross-sectional view of the MSA.

  The MSA of the present embodiment is the same as the MSA of the fourth embodiment except that the second radiating element 21 has the second parasitic elements 15 arranged in a matrix and the elements are connected by the minute conductors 5. Yes. That is, the second radiating element 21 has substantially the same arrangement configuration as the first radiating element 20. The second radiating element 21 is provided with a plurality (three in FIG. 8) of second switches 14 that are short-circuited with the ground plane 1.

  In the structure of the present embodiment, since the shape of the second radiating element 21 is substantially the same as that of the first radiating element 20, the second radiating element 21 has a resonance mode on the low frequency side of the first radiating element 20. Will have the same resonance mode. Therefore, when the first radiating element 20 is excited in the resonance mode on the low frequency side, a large current is induced in the second radiating element 21 due to mutual coupling between the first radiating element 20 and the first radiating element 20. The combination of conduction / interruption of the second switch 14 makes it possible to efficiently tilt the beam in the resonance mode on the low frequency side.

  In the MSA of this embodiment, the ground plate 1 is a Cu plate, the dielectric 2 is a Teflon (registered trademark) glass fiber substrate having a relative dielectric constant of 2.2, the feeding element 3 is a 4.4 mm square Cu plate, The parasitic element 4 is made of a 4.4 mm square Cu plate, the minute conductor 5 is made of a 800 μm wide Cu plate, and the first parasitic elements 4 are arranged at intervals of 0.4 mm. A radiating element 20 is formed, and further, a second radiating element 21 in which 4.4 mm square second parasitic elements 15 made of a Cu layer are arranged in a 3 × 3 matrix at intervals of 0.4 mm, 1 is arranged around the radiating element 20 with an interval width of 1.0 mm, φ = 0, 45, 90, 135, 180, 225, depending on the combination of conduction / cutoff of the second switch 14. 270, 315 degrees, etc., the beam is up to 25-3 in the θ direction The beam can be switched in the resonance mode on the low frequency side by tilting 5 °.

  Further, in this configuration, the second radiating element 21 is provided with three second switches 14, and the current induced in the second radiating element 21 can be adjusted by changing the number of switches to be conducted. Fine adjustment of the tilt angle of the beam in the θ direction becomes possible.

  From the above description, according to the present configuration, a current can be efficiently induced in the second radiating element 21 in the resonance mode on the low frequency side, and the second radiating element 21 is short-circuit controlled by the second switch 14. Therefore, directivity switching can be realized efficiently. Further, by providing the second radiating element 21 with a plurality of second switches 14, finer directivity control is possible in the resonance mode on the low frequency side.

  In the above description, the second switch number 14 is three, but the second switch number 14 is not limited to three. Further, the location where the second switch 14 is provided is not limited to the location shown in FIG. Further, although the arrangement shapes of the first radiating element 20 and the second radiating element 21 are substantially the same, the second radiating element 21 is efficient in the resonance mode on the low frequency side that resonates with the first radiating element 20. If they are excited, they need not be substantially the same.

<Embodiment 6>
A microstrip antenna (MSA) according to a sixth embodiment will be described. The MSA of this embodiment is a three-frequency antenna that operates in three modes of low frequency, intermediate frequency, and high frequency, and can switch directivity in each frequency band.

  11 and 12 show the configuration of the MSA of this embodiment. 11 shows a top view of the MSA, and FIG. 12 shows a cross-sectional view of the MSA.

  The MSA includes a ground plane 1, a dielectric 2, a feeding element 3, a first parasitic element 4, a minute conductor 5, a feeding point 6, a first switch 7, a first substrate 8, The matching circuit 9, the feeder line 10, the second switch 14, and the second parasitic element 15 are configured.

  A ground plane 1 is provided on the first substrate 8, and a dielectric 2 is provided on the ground plane 1. On top of the dielectric 2, a rectangular feed element 3 and a first parasitic element 4 arranged in a matrix with the feed element 3 as the center are stacked. The parasitic element 5 has the same shape as the rectangular feeding element 3. Further, between the adjacent feeding element 3 and the first parasitic element 4 and between the adjacent first parasitic elements 4 are connected by a minute conductor 5 having a width shorter than the width of the opposing sides. Yes.

  The feeding element 3 is provided with a feeding point 6. The feed point 6 is connected to the matching circuit 9 via a via hole 19 penetrating the dielectric 2, the ground plane 1, and the first substrate 8 and the feed line 10. Note that the feeder line 10 is formed of a microstrip line. The first parasitic element 4 is provided with a first switch 7. The first switch 7 switches between a conduction / cutoff state between the first parasitic element 4 and the ground plane 1. When the first switch 7 is in a conductive state, the first parasitic element 4 is short-circuited with the ground plane 1, and when the first switch 7 is in a disconnected state, the first parasitic element 4 is released from the ground plane. The feeding element 3, the first parasitic element 4, and the minute conductor 5 constitute a first radiating element 20.

  A second parasitic element 15 is stacked on the dielectric 2 so as to surround the first radiating elements 20 in a matrix. The shape of the second parasitic element 15 is substantially the same as that of the first parasitic element 4. The second parasitic element 15 surrounds the first radiating element 20 for six turns. Further, the adjacent first parasitic element 4 and the second parasitic element 15 and the adjacent second parasitic element 15 are connected by the minute conductor 5 having a width shorter than the width of the opposing side. Has been.

  The second parasitic element 15 is provided with a second switch 14. The second switch 14 switches the conduction / cut-off state between the second parasitic element 15 and the ground plane 1. Note that the switch selection of the first switch 7 and the second switch 14 is made via a bias line.

  Next, resonance mode control for each frequency in the MSA of the present embodiment will be described.

  First, control in the resonance mode on the high frequency side will be described. In the MSA of the present embodiment, when an RF signal having a half-wavelength L1 in the Y direction of the feed element 3 is applied to the feed element 3, the RF signal is substantially reflected on the side of the feed element 3 in the X direction. A resonance mode on the high frequency side can be realized. In addition, since the feed element 3 is surrounded by the eight first parasitic elements 4, the first switch 7 performs short-circuit control on the first parasitic element 4, so that the resonance mode on the high frequency side is achieved. The directivity can be switched.

  Furthermore, in the MSA of the present embodiment, the second parasitic element 15 has substantially the same shape as the first parasitic element 4 and the feeder element 3, and thus the second parasitic element 15 is close to the first radiating element 20. The parasitic element 15 is excited in the resonance mode on the high frequency side by mutual coupling between the feed element 3 and the first parasitic element 4 that are excited in the resonance mode on the high frequency side. For this reason, the second parasitic element 15 close to the first radiating element 20 can slightly control the resonance mode radiation pattern on the high frequency side. As a specific example, when the second parasitic element 15 close to the first parasitic element 4 opened from the ground plane 1 is opened from the ground plane 1, the tilt angle of the beam can be enhanced.

  Next, the resonance mode on the intermediate frequency side will be described. In the MSA of the present embodiment, since the feed element 3 and the first parasitic element 4 are connected by the minute conductor 5 to form the first radiating element 20, when all the first switches 7 are cut off, A resonance mode in an intermediate frequency band having a half-wavelength L2 in the Y direction of the first radiating element is generated. For this reason, when an RF signal that matches the resonance mode in the intermediate frequency band is given to the feed element 3, resonance occurs in the intermediate frequency band, so that it can correspond to the intermediate frequency.

  Here, since the first radiating element 20 is surrounded by the second parasitic element 15 and the second parasitic elements 15 are connected to each other by the minute conductor 5, the first radiating element 20 is provided. A region of 3 × 3 elements close to is subjected to strong inter-element coupling with the first radiating element 20 and is excited in an intermediate frequency resonance mode. When this 3 × 3 element region is regarded as the second radiating element 21, as shown in FIG. 13, eight second radiating elements 21 made of 3 × 3 elements are arranged in a matrix around the first radiating element 20. It can be considered that it is arranged.

  Therefore, directivity switching can be performed even in the intermediate frequency band by performing short-circuit control of the second radiating element 21 using the second switch 14.

  In addition, since the second radiating element 21 is provided with a plurality of second switches 14, by combining conduction / cutoff of the second switch 14 connected to the same second radiating element 21, It is possible to finely adjust the tilt angle of the beam in the θ direction.

  Next, the resonance mode on the low frequency side will be described. In the MSA of the present embodiment, since the first radiating element 20 and the second parasitic element 15 are connected by the minute conductor 5, all the first switches 7 are cut off, and the first radiating element 20 is used. When the 16 second parasitic elements 15 adjacent to the base plate 1 are opened from the ground plane 1, it functions as a “third radiating element 22” having a shape of 5 × 5 shown in FIG. A resonance mode on the low frequency side in which the length L3 in the direction is approximately a half wavelength occurs. Therefore, when an RF signal that matches the resonance mode on the low frequency side is given to the power feeding element 3, resonance occurs on the low frequency side, so that low frequency response can be realized.

  The third radiating element 22 is surrounded by a second parasitic element 15 having the same shape as that of the first parasitic element 4, and the second parasitic elements 15 are minute conductors 5. It is connected. Therefore, the “fourth radiating element 23” having a 5 × 5 shape adjacent to the third radiating element 22 is excited in the resonance mode on the low frequency side due to strong mutual coupling with the third radiating element 22. Is done.

  The state of the above-mentioned MSA is that a radiating element (fourth radiating element 23) composed of parasitic elements of the same size is arranged around a radiating element (third radiating element 22) including a 5 × 5 size feeding element. It can be regarded as a shape arranged in a matrix of eight. Therefore, by controlling the second switch 14 connected to the fourth radiating element 23 and performing the short-circuit control of the fourth radiating element 23 adjacent to the third radiating element 22, directivity can be achieved even on the low frequency side. Switching can be performed.

  In addition, since the plurality of second switches 14 are provided in the fourth radiating element 23, the combination of conduction / cut-off of the second switches 14 connected to the same fourth radiating element 23 makes θ It becomes possible to finely adjust the tilt angle of the beam in the direction.

  Thus, in this embodiment, since it operates in three modes of low frequency, intermediate frequency, and high frequency, it is possible to cope with three resonance modes. In addition, the directivity can be switched in each frequency band.

  11, the first radiating element 20 includes a ground plane 1 made of a Cu layer, a dielectric 2 made of a Teflon (registered trademark) glass fiber substrate having a relative dielectric constant of 2.2, and a Cu layer. A first feed element 3 surrounding the feed element 3 in a matrix using a 4 mm square feed element 3, a 4.4 mm square first parasitic element 4 made of a Cu layer, and a minute conductor 5 made of Cu having a width of 800 μm. The parasitic elements are arranged at intervals of 0.4 mm, and the periphery of the first radiation element 20 is matrixed by a second parasitic element 15 of 4.4 mm square made of a Cu layer at an element interval of 0.4 mm. 6 rounds, and when the adjacent first parasitic element 4 and the second parasitic element 15 and between the second parasitic elements 15 are connected by the micro conductor 5 having a width of 800 μm made of Cu. In the first switch 7 Depending on the combination of conduction / cutoff of the second switch 14, in the resonance mode on the high frequency side, φ = 0, 45, 90, 135, 180, 225, 270, 315 °, etc., and the beam is 20-40 in the θ direction. I was able to tilt. Further, by the combination of conduction / cutoff of the second switch 14, even in the resonance mode of the intermediate frequency band, φ = 0, 45, 90, 135, 180, 225, 270, 315 °, etc., and the beam is 25 in the θ direction. It was possible to tilt ~ 35 °. Further, by the combination of conduction / cutoff of the second switch 14, even in the resonance mode on the low frequency side, the beam is directed in the θ direction at φ = 0, 45, 90, 135, 180, 225, 270, 315 °, etc. It was possible to tilt 15-30 °. In other words, XY two-dimensional beam switching and directivity switching could be performed in the three frequency bands of low frequency, intermediate frequency, and high frequency.

  Since the MSA of the present embodiment is compatible with three frequencies as described above, the L ′ / L of the radiating element is greatly different in each frequency band, so that there is a possibility that good transmission / reception cannot be performed. Therefore, adjustment is performed using the matching circuit 9 whose constant can be changed. This will be specifically described below.

  The configuration of the matching circuit 9 will be described with reference to FIG. The matching circuit 9 is composed of a circuit in which a varactor diode 16 and a spiral inductor 11 are assembled in a π type by CLC. Switching between the matching circuit 9 and the 50Ω feed line 10 is performed by a changeover switch 13.

  For example, when the feed point 3 is arranged so that the input impedance of the feed element 3 is 50Ω, in the resonance mode on the high frequency side, the changeover switch 13 is placed on the 50Ω feed line 10 side to directly connect the antenna and the RF circuit. . As a result, in the resonance mode on the high frequency side, the input impedance of the antenna is almost equal to the input impedance of the feed element 3 even if the directivity is switched, and thus reflection is suppressed and good transmission / reception is possible.

  In the resonance mode on the intermediate frequency side and the resonance mode on the low frequency side, the changeover switch 13 is placed on the matching circuit 9 side, and the antenna and the RF circuit are connected via the matching circuit 9. If the input impedance of the first radiating element 20 and the input impedance of the third radiating element 22 are matched to 50Ω by appropriately selecting C of the varactor diode of the matching circuit 9, the resonance mode on the intermediate frequency side, the low frequency In both resonance modes, reflection is suppressed and good transmission / reception is possible.

  As described above, by using the matching circuit 9 that can change the constant, good transmission and reception can be performed for all three frequencies.

  The matching circuit 9 described above is a π-type matching circuit, but is not limited to such a type of matching circuit, and may be a T-type or L-type matching circuit. Further, not only the combination of the varactor diode 16 and the spiral inductor 11, but also a combination of a phase shifter, an inductor, and a capacitor may be used.

  In the above description, the varactor diode 16 is used as an element that can change the constant. However, the present invention is not limited to this, and a variable inductor by MEMS may be used.

  In addition, when there is a space, a fixed matching circuit for the intermediate frequency side and a fixed matching circuit for the low frequency side are provided, respectively, and a 50Ω feed line and an intermediate frequency are set in each frequency band by a changeover switch. The two matching circuits for the low frequency side may be switched.

  In the MSA of the present embodiment, PIN diodes can be used for the first switch 7 and the second switch 14. Since the PIN diode can be obtained at a low cost and can cut off high-frequency signals well up to 20 GHz, a directivity control antenna for 2.4 GHz or 5 GHz wireless LAN or a mobile terminal can be configured.

  The PIN diode is partly cut off the ground plane 1 on the back surface of the dielectric 2 and mounted in the cut area. At this time, one end of the PIN diode is connected to the ground plane 1 and the other end is connected to the first parasitic element 4 and the second parasitic element 15 through a via hole penetrating the dielectric 2. The bias line for controlling the PIN diode is wired on the first substrate 8 and connected to the PIN diode by a via hole penetrating the first substrate 8.

  Also, MEMS switches can be used as the first switch 7 and the second switch 14. The MEMS switch can satisfactorily cut off a high-frequency signal up to about 100 GHz and has a small insertion loss. Therefore, the MEMS switch constitutes a directivity control antenna corresponding to a high frequency such as a submillimeter wave to a millimeter wave. be able to.

  The MEMS switch is mounted on the lower surface of the dielectric 2 in the same manner as the PIN diode described above. Further, after forming the MEMS switch on the first substrate 8 by a bulk micromachine process or a surface micromachine process, the first substrate 8 is attached to the lower surface of the dielectric 2, and one end of the MEMS switch is connected to the ground plane 1. The other end can also be mounted by connecting to the first parasitic element 4 and the second parasitic element 15 via a via hole penetrating the dielectric 2. When such a method is used, a bias line for controlling the MEMS switch can be wired on the first substrate 8.

  Further, in the MSA of the present embodiment, similarly to the MSA of the second embodiment, if the adjacent elements are connected by a plurality of microconductors 5, the first / second / third / fourth radiating elements are provided. The internal gap can be reduced, and the bandwidth of the resonance mode on the intermediate frequency side and the resonance mode on the low frequency side can be improved.

  Further, by using the microconductor 5 having a bent structure as in the MSA of the third embodiment, the current path becomes longer and the frequency is lowered in the resonance mode on the intermediate frequency side and the resonance mode on the low frequency side. It is possible to control the tilt angle of the side beam and the resonance frequency on the intermediate frequency side / low frequency side almost independently.

  In the MSA of the present embodiment, the feed element 3, the first parasitic element 4, and the second parasitic element 15 have the same shape. However, when the feed element 3 alone resonates, Since the parasitic element 4 and the second parasitic element 15 need only be excited by mutual coupling between the elements, they need not be the same and may have substantially the same shape.

  In the MSA of the present embodiment, the first radiating element 20 has a structure in which the second parasitic element 15 surrounds six circumferences, so that it corresponds to three frequencies. However, the first radiating element 20 is further increased. If the second parasitic element 15 surrounds the second parasitic element 15 in a matrix, the directivity can be switched in more frequency bands.

  Further, in the MSA of the present embodiment, the second switch 14 that is short-circuited to the ground plane 1 is provided in all of the second parasitic elements 15, but depending on the required tilt angle of the antenna, a part of the second switch 14 is provided. The second switch 14 may be provided only in the parasitic element.

<Seventh Embodiment>
Next, a radio module using the above MSA will be described as a seventh embodiment of the present invention. The wireless module of the present embodiment uses the MSA of the sixth embodiment as the MSA to be mounted.

  This will be specifically described below. FIG. 16 is a diagram for explaining the radio module, and shows a radio module using the MSA of the sixth embodiment.

  The wireless module has a structure in which an MSA is provided on a second substrate 17 made of a porous polyimide substrate. A front-end circuit 18 made of a chip component is configured on the second substrate 17, and the front-end circuit 18 and the feeder line 10 are connected by a via hole 19 that penetrates the second substrate 17.

  Since the wireless module of the present embodiment uses the MSA of the sixth embodiment, three-frequency control is possible and directivity control at each frequency is possible. Therefore, since the maximum radiation angle of the antenna can be directed in the direction of the desired wave in each frequency band, a large gain can be obtained and good transmission / reception can be performed. Moreover, since it can respond | correspond to 3 frequencies, it can respond | correspond to several standards with one radio | wireless module, and it can contribute to size reduction of a radio | wireless module.

  In the above description, the wireless module using the MSA of the sixth embodiment has been described. However, the MSA of the first to fifth embodiments may be used.

  In the above description, only the front end circuit 18 is mounted on the second substrate 17, but both the front end circuit and the baseband circuit may be mounted. Further, only a part of the front end circuit and the baseband circuit may be mounted. With this configuration, it is possible to reduce the size of the wireless module.

<Eighth Embodiment>
Next, a wireless system using the wireless module of the seventh embodiment will be described as an eighth embodiment of the present invention. FIG. 17 shows a circuit configuration of the wireless system of the present embodiment, which uses a wireless module (FIG. 16) using the MSA of the sixth embodiment.

  The wireless system of the present embodiment includes an MSA, a matching circuit 9, a front end circuit 18, a control circuit 31, a first switch bias generation circuit 32, a second switch bias generation circuit 33, and a switching switch bias. The circuit includes a generation circuit 34, a barracuda diode bias generation circuit 35, a transmission / reception changeover switch bias generation circuit 36, and a transmission / reception changeover switch 37.

  In the wireless system of the present embodiment, the control circuit 31 sets the conduction / cutoff combination of the first switch 7 and the second switch 14 so that a desired frequency and directivity can be achieved.

  If it is compatible with a high frequency, a predetermined bias is generated by applying a control signal A from the control circuit 31 to the first switch bias generation circuit 32 and a control signal B to the second switch bias generation circuit 33, The first switch 7 and the second switch 14 are turned on / off, and the first parasitic element 4 and the second parasitic element 15 that are short-circuited with the ground plane are switched to perform two-dimensional XY beam switching.

  Further, if it is compatible with the intermediate frequency, the control signal A is supplied from the control circuit 31 to the first switch bias generation circuit 32 to generate a predetermined bias, and the first switch 7 is turned off to make the first parasitic power supply. The element 4 is released from the ground plane 1 to form the first radiating element 20, and the control signal B is supplied from the control circuit 31 to the second switch bias generation circuit 33 to generate a predetermined bias. The switch 14 is cut off or turned on to open / short-circuit the second radiating element 21 composed of the 3 × 3 second parasitic element 15 and the minute conductor 5 with the ground plane 1 to perform beam switching in two dimensions XY.

  In addition, if low frequency is supported, a predetermined bias is generated by applying a control signal A from the control circuit 31 to the first switch bias generation circuit 32 and a control signal B to the second switch bias generation circuit 33. Thus, the first switch 7 and the second switch 14 connected to the second parasitic element 15 adjacent to the first radiating element 20 are cut off to form the third radiating element 22. Then, the second switch 14 connected to the other second parasitic element 15 is cut off / conducted, and the 5 × 5 second parasitic element 21 adjacent to the third radiating element 22 is minutely connected. The fourth radiating element 23 made of the conductor 5 is opened / short-circuited with the ground plane 1 to perform beam switching in two dimensions of XY.

  For impedance matching at three frequencies, a predetermined bias is generated by supplying a control signal C from the control circuit 31 to the changeover switch bias generation circuit 34, and the changeover control of the changeover switch 13 of the matching circuit 9 is performed. Specifically, the antenna is connected to the front end circuit 18 via the feeder line 10 for high frequency, and the antenna is connected to the matching circuit 9 and connected to the front end circuit 18 for intermediate frequency and low frequency.

  For intermediate frequency / low frequency, the control signal C is also applied to the varactor diode bias generation circuit 35 to generate a predetermined bias, and the constant of the varactor diode 16 of the matching circuit 9 is changed to change the intermediate frequency band / low frequency. Match the input impedance of the antenna on the side to 50Ω.

  Further, in order to switch between transmission and reception modes, a predetermined bias is generated by applying a control signal D from the control circuit 31 to the transmission / reception changeover switch bias generation circuit 36, and transmission / reception is changed over by the transmission / reception changeover switch 37. The front-end circuit is electrically connected to perform communication.

  As described above, the wireless system of this embodiment can perform directivity control corresponding to three frequencies. Therefore, one radio system can support three frequency standards, and the radio system itself can be miniaturized. In addition, since directivity control is possible, high gain can be realized by directing a radiation pattern in a desired direction, and favorable transmission and reception can be performed. Furthermore, since the frequency and directivity can be switched depending on the radio wave condition, it is possible to constantly perform good transmission / reception.

  The above-described wireless system is an example, and is not limited to such a form. The bias generation circuit is omitted, and the first switch 7, the second switch 14, the changeover switch 13, and the transmission / reception changeover switch 37 are used. The conduction / shut-off control may be performed by directly supplying the control signals A / B / C / D.

It is a top view of MSA of a 1st embodiment. It is sectional drawing of MSA of 1st Embodiment. It is an enlarged view of a feed element. It is a figure for demonstrating a matching circuit. It is a figure which shows MSA of 2nd Embodiment. It is a figure which shows MSA of 3rd Embodiment. It is a top view of MSA of a 4th embodiment. It is sectional drawing of MSA of 4th Embodiment. It is a top view of MSA of a 5th embodiment. It is sectional drawing of MSA of 5th Embodiment. It is a top view of MSA of a 6th embodiment. It is sectional drawing of MSA of 6th Embodiment. It is a figure for demonstrating correspondence with an intermediate frequency. It is a figure for demonstrating low frequency response. It is a figure for demonstrating a matching circuit. It is a figure which shows a wireless module. It is a figure which shows a radio | wireless system. It is a figure which shows a coordinate system.

Explanation of symbols

DESCRIPTION OF SYMBOLS 3 Feeding element 4 1st parasitic element 5 Micro conductor 7 1st switch 9 Matching circuit 14 2nd switch 15 2nd parasitic element 20 1st radiation element 21 2nd radiation element 22 3rd radiation Element 23 Fourth radiating element 31 Control circuit

Claims (30)

With the main plate,
A feed element disposed above the ground plane ;
A first parasitic element arranged in a matrix around the feeding element;
A conductor connecting each element of the feeding element and the first parasitic element;
First switch means provided between the ground plate and the first parasitic element, for switching whether to short-circuit the first parasitic element to the ground plate ,
Corresponding to high frequency by resonating the feeding element alone,
It corresponds to a low frequency by resonating a first radiating element constituted by the first parasitic element disposed on the feed element and a matrix,
The high frequency to the corresponding time, microstrip antenna and controls the directivity of the antenna by switching the first switch means.   The microstrip antenna according to claim 1, wherein the feeding element and the first parasitic element have substantially the same shape.   3. The microstrip antenna according to claim 1, wherein a width of the conductor is shorter than a side opposite to each element to which the conductor is connected. 4.   The microstrip antenna according to any one of claims 1 to 3, wherein the feed element is connected to a matching circuit.   The microstrip antenna according to any one of claims 1 to 4, wherein the first switch means is a PIN diode.   The microstrip antenna according to any one of claims 1 to 4, wherein the first switch means is a MEMS switch.   The microstrip antenna according to any one of claims 1 to 6, wherein a plurality of the conductors are provided between the elements to which the conductors are connected.   The microstrip antenna according to claim 1, wherein the conductor has a bent shape. A parasitic second radiating element that is arranged in a matrix around the first radiating element and resonates at the same frequency as the first radiating element;
A second switch means provided between the ground plate and the second radiating element, for switching whether to short-circuit the second radiating element to the ground plate ;
Wherein the low frequency response when the microstrip antenna according to any one of claims 1 to 8, characterized in that to control the directivity of the antenna by switching the second switch means.   10. The microstrip antenna according to claim 9, wherein the second switch means is a PIN diode.   10. The microstrip antenna according to claim 9, wherein the second switch means is a MEMS switch.   The microstrip antenna according to any one of claims 9 to 11, wherein the second radiating element includes one second parasitic element.   The said 2nd radiation | emission element is comprised from the some 2nd parasitic element and the said conductor which connects between the said parasitic elements, The any one of Claim 9 to 11 characterized by the above-mentioned. The described microstrip antenna.   The microstrip antenna according to claim 13, wherein at least one of the plurality of second parasitic elements includes the second switch means.   The microstrip antenna according to claim 13 or 14, wherein the second radiating element has the same arrangement configuration as the first radiating element. With the main plate,
A feed element disposed above the ground plane ;
A first parasitic element arranged in a matrix around the feeding element;
A plurality of second parasitic elements arranged in a matrix around the first parasitic element arranged in the matrix; and
A conductor connecting each of the feeding element, the first parasitic element, and the second parasitic element; and
Provided between said base plate first passive element, a first switching means for switching whether to short-circuit the first parasitic elements該地plate,
It provided between the base plate and the second parasitic element, and a second switching means for switching whether to short-circuit the second parasitic elements該地plate, and
A microstrip antenna, wherein the directivity of the antenna is controlled at a plurality of frequencies by switching between the first switch means and the second switch means.   The microstrip antenna according to claim 16, wherein the feeding element, the first parasitic element, and the second parasitic element have substantially the same shape.   18. The microstrip antenna according to claim 16, wherein a width of the conductor is shorter than a side opposite to each element to which the conductor is connected.   The microstrip antenna according to any one of claims 16 to 18, wherein the feeding element is connected to a matching circuit.   The microstrip antenna according to any one of claims 16 to 19, wherein the first switch means is a PIN diode.   The microstrip antenna according to any one of claims 16 to 19, wherein the first switch means is a MEMS switch.   The microstrip antenna according to any one of claims 16 to 21, wherein the second switch means is a PIN diode.   The microstrip antenna according to any one of claims 16 to 21, wherein the second switch means is a MEMS switch.   The microstrip antenna according to any one of claims 16 to 23, wherein a plurality of the conductors are provided between the elements to which the conductors are connected.   The microstrip antenna according to any one of claims 16 to 24, wherein the conductor has a bent shape.   A wireless module using the microstrip antenna according to any one of claims 1 to 25.   27. A wireless system using the wireless module according to claim 26. With the main plate,
A power feeding element disposed above the ground plane ;
A first parasitic element arranged in a matrix around the feeding element;
A conductor connecting each element of the feeding element and the first parasitic element;
A control method for a microstrip antenna, comprising: first switch means provided between the ground plane and the first parasitic element, for switching whether to short-circuit the first parasitic element to the ground plane. Because
By making the power feeding element alone resonate,
Resonating the first radiating element composed of the feeding element and the first parasitic element arranged in a matrix to correspond to a low frequency,
The high frequency to the corresponding time, the control method of the microstrip antenna and controlling the antenna directivity by switching the first switch means. With the main plate,
A feed element disposed above the ground plane ;
A first parasitic element arranged in a matrix around the feeding element;
A conductor connecting each element of the feeding element and the first parasitic element;
Provided between said base plate first passive element, a first switching means for switching whether to short-circuit the first parasitic elements該地plate,
A parasitic second radiating element that is arranged in a matrix around the first radiating element and resonates at the same frequency as the first radiating element;
Provided between the base plate and the second radiating element, a control method of a microstrip antenna having a second switching means for switching whether to short-circuit the second radiating element to該地plate, the And
Resonating the power feeding element alone to cope with high frequency,
Resonating the first radiating element composed of the feeding element and the first parasitic element arranged in a matrix to correspond to a low frequency,
Wherein the frequency corresponding time, and controls the directivity of the antenna by switching the first switch means,
A microstrip antenna control method, wherein the directivity of the antenna is controlled by switching the second switch means at the time of the low frequency response. With the main plate,
A feed element disposed above the ground plane ;
A first parasitic element arranged in a matrix around the feeding element;
A plurality of second parasitic elements arranged in a matrix around the first parasitic element arranged in the matrix; and
A conductor connecting each of the feeding element, the first parasitic element, and the second parasitic element; and
Provided between said base plate first passive element, a first switching means for switching whether to short-circuit the first parasitic elements該地plate,
Provided between the base plate and the second parasitic element, a control method of a microstrip antenna having a second switching means for switching whether to short-circuit the second parasitic elements該地plate, the Because
A method of controlling a microstrip antenna, wherein the directivity of an antenna is controlled at a plurality of frequencies by switching between the first switch means and the second switch means.

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