JP4240486B2 - Input / output isolation type DC-DC converter device - Google Patents

Description

  The present invention relates to an input / output isolation type DC-DC converter device, and more particularly to an input / output isolation type DC-DC converter device that changes a switching control state in accordance with an input power supply voltage.

In an idle stop vehicle or a hybrid vehicle that performs idle stop, a dual power supply vehicle power supply device equipped with a high voltage power supply system including a high voltage battery and a generator and a low voltage power supply system including an electric load and a low voltage battery. Is known. In this two-power-source vehicle power supply device, various types of DC-DC converters are employed to transmit power from a high-voltage power supply system to a low-voltage power supply system. As an example of the dual power supply type vehicle power supply device, there is the following Patent Document 1 filed by the present applicant.
JP, 2002-345161, A In this two power supply type vehicle power supply device, the input side circuit which changes the input power supply voltage applied from a direct-current power supply intermittently into an alternating voltage, and outputs it by switching a switching element, and this alternating voltage Output side circuit that rectifies and outputs output voltage to the outside, and input / output insulation having a transformer that transmits AC power from the input side circuit to the output side circuit while electrically isolating the output side circuit from the input side circuit A separate DC-DC converter device is preferably employed. In order to converge the output voltage of the input / output isolation type DC-DC converter device to the target voltage, a controller (control circuit) that controls the switching element of the input side circuit based on the difference between them is employed. As an example of this control format, PWM (pulse width modulation) control is the most common. For example, the following Patent Documents 2 and 3 have been proposed as this type of input / output isolation type DC-DC converter device. JP 2003-102175 A USP 5291382 In order to improve the safety of this input / output isolation type DC-DC converter device, various electrical states of each part of the device or detection signals corresponding thereto are transmitted to the controller and reflected in the control of the switching element by the controller. It is generally done. As the detection of the electrical state, it is particularly important to detect an abnormal state of a high-voltage power supply system that has high energy and high safety requirements.

  However, when the above-described conventional input / output isolation type DC-DC converter apparatus is used for a two-power-source vehicle power supply apparatus, the controller for controlling the switching element of the DC-DC converter is the potential level of the low-voltage power supply system. Usually, it is driven by a power supply voltage with reference to.

  Therefore, in this case, the input side circuit of the input / output isolation type DC-DC converter device is a high voltage power supply system, and the detection circuit for detecting the electrical state of the input side circuit side has the potential level of the high voltage power supply system. It is driven by a reference power supply voltage. As a result, it is necessary to provide an input / output isolation circuit such as a photocoupler or pulse transformer in order to transmit a signal related to the electrical state of the input side circuit of the input / output isolation type DC-DC converter device to the controller. It becomes.

  However, the conventional input / output isolation circuit that transmits signals while electrically isolating the input and output, such as a photocoupler, is expensive, and such an input / output isolation circuit converts the transmission signal into a pulse signal format. Therefore, transmitting a plurality of types of status signals detected by the input-side circuit of the input / output isolation type DC-DC converter to the controller is accompanied by an increase in the circuit load, which is a large cost burden. It was.

  The present invention has been made in view of the above problems, and it is an object of the present invention to provide an input / output isolation type DC-DC converter device that can use detection signals relating to various operational safety while suppressing an increase in circuit burden. It is aimed.

An input / output isolation type DC-DC converter device according to the present invention includes an input side circuit that intermittently switches a switching element to convert an input power supply voltage applied from a DC power source into an AC voltage and outputs the AC voltage, and rectifies the AC voltage. DC-DC converter having an output side circuit for outputting output voltage to the outside, and a transformer for transmitting AC power from the input side circuit to the output side circuit while electrically isolating the output side circuit from the input side circuit A circuit, an input state monitor circuit for outputting a pulse signal having a pulse width or an on-duty ratio corresponding to the detected magnitude of the input power supply voltage through an input / output isolation circuit, and an output voltage of the DC-DC converter circuit The power supply voltage is based on the potential level of the input power supply, and is responsive to the input power supply voltage and the output voltage of the DC-DC converter circuit. And a controller for controlling the switching elements Te, the input status monitor circuit, when detecting an abnormal state of the input-side circuit, and changing the pulse period of the pulse signal in which the input status monitor circuit outputs, The controller determines an abnormal state of the input side circuit by determining the change of the pulse period.

  That is, according to the present invention, the input side of the input / output isolation type DC-DC converter device is connected to the controller supplied with the power supply voltage of the potential reference of the output side circuit of the input / output isolation type DC-DC converter. An input that generates a pulse signal having a pulse width or an on-duty ratio corresponding to the magnitude of the input power supply voltage applied to the circuit, and transmits this pulse signal to the controller through an input / output isolation circuit, which is usually a photocoupler. Install a condition monitor circuit.

  Thus, the controller can detect the magnitude of the input power supply voltage applied to the input / output isolation type DC-DC converter device from the DC power supply on the input side.

  Further, according to the present invention, by detecting other electrical states of the input side circuit of the input / output isolation type DC-DC converter device, the carrier period (carrier frequency) of the pulse signal is changed according to the detected state. Send to controller. As a result, a plurality of electrical states of the input side circuit of the input / output isolation / separation type DC-DC converter device can be provided without providing a plurality of input / output isolation / separation type signal transmission systems requiring expensive photocouplers and pulse signal transmission / reception circuits on both sides thereof. Can be transmitted to the controller on the output side. Therefore, according to the present invention, it is possible to realize an input / output isolation type DC-DC converter device that can use detection signals related to various operational safety while suppressing an increase in circuit burden.

  In a preferred embodiment, the switching element of the input side circuit of the input / output isolation type DC-DC converter device is PWM (zero) so that the deviation between the output voltage of the input / output isolation type DC-DC converter device and its target voltage is zero. Pulse width modulation). Hereinafter, it is assumed that when the duty ratio (referred to as the on-duty ratio) of the switching element is increased, the output voltage of the input / output isolation type DC-DC converter device is increased. At this time, the input state monitoring circuit generates a pulse signal whose duty ratio (or pulse width) decreases when the input power supply voltage applied to the input / output isolation type DC-DC converter device from the DC power supply on the input side increases. It shall be. In this way, by limiting the duty ratio (or pulse width) of the switching element within the range of the duty ratio (or pulse width) of the pulse signal, the input / output isolation type DC-DC converter device is input. Thus, the withstand voltage protection of the switching element due to the change of the input power supply voltage can be easily performed. This detail is described in the embodiment.

  In a preferred aspect, the input-side circuit opens and closes an inverter circuit that incorporates the switching element and converts the input power supply voltage to the AC voltage, and an electrical connection between the DC power supply and the inverter circuit. A main power switch, and a sub power switch and an inrush current limiting resistor connected in series with each other, and an inrush current limiting resistor circuit connected in parallel to the main power switch. The pulse cycle is changed when an abnormal current state in which the current flowing through the sub power switch exceeds a predetermined threshold value when an off command is issued to the power switch is detected. In this way, in addition to the above effects, an abnormality of the inrush current limiting resistor circuit can be detected at an early stage while suppressing an increase in circuit load.

  In a preferred aspect, the input state monitor circuit changes the pulse period when detecting an overvoltage input state in which the input power supply voltage exceeds a predetermined threshold value. In this way, it is possible to detect an excessive increase in input power supply voltage while suppressing an increase in circuit load.

  In a preferred aspect, the input state monitoring circuit includes an A mode in which the leakage current abnormal state is detected and the overvoltage input state is not detected, the leakage current abnormal state is not detected, and the overvoltage input state is detected. B mode in which no leakage current is detected, the leakage current abnormal state is not detected, the C mode in which the overvoltage input state is detected, the leakage current abnormal state is detected, and the overvoltage input state is detected And a different pulse period is assigned to each mode. In this way, the input power supply voltage and a plurality of types of abnormal conditions can be transmitted from the input side circuit to the controller by a single input / output isolation circuit, preventing an increase in circuit burden and ensuring device safety. Can be further improved.

  In a preferred aspect, the input state monitoring circuit transmits a PWM pulse signal having a duty ratio or a pulse width that decreases with an increase in the input power supply voltage to the controller in the A and B modes, and the controller The PWM pulse signal applied to the switching element of the DC-DC converter for making the output voltage of the DC-DC converter circuit constant, the duty ratio or pulse width of the PWM pulse signal received from the input state monitor circuit It is characterized by being limited within the range. In this way, the DC-DC converter circuit can be easily driven within the safe duty ratio range by using the safe duty ratio range received from the input state monitor circuit.

  In a preferred aspect, the input state monitor circuit includes a first pulse signal having a pulse interval that changes in response to an increase in the input power supply voltage, a first period indicating a first abnormal state, and the input Either a pulse interval that changes in accordance with an increase in power supply voltage or a second pulse signal having a second period indicating a second abnormal state is selected and output, and the first pulse signal The pulse falling edge and the pulse falling edge of the second pulse signal are set at the same timing. In this way, even when the mode is switched, that is, when the first pulse signal and the second pulse signal are switched, the pulse interval count can be accurately performed, and the reliability of the detection of the input power supply voltage is further improved. be able to.

  In a preferred aspect, the input state monitor circuit includes a third pulse signal indicating a third abnormal state with a pulse interval that does not change according to an increase in the input power supply voltage, the first signal, and the second signal. And the pulse period of the third pulse signal is set to be longer than the pulse period of the first pulse signal or the second signal, and the first pulse signal or the second pulse signal The pulse interval of the pulse signal received by the controller at the time of transition to the third pulse signal is set in a range not corresponding to the input power supply voltage. In this way, since the pulse interval always deviates from the possible range of the input power supply voltage when shifting from the first pulse signal or the second pulse signal to the third pulse signal, the third abnormal state is detected early. In addition, the input power supply voltage is not erroneously detected.

  In a preferred aspect, the input state monitoring circuit includes a third pulse signal, a first signal, and a second signal that indicate a third abnormal state with a pulse interval that does not change according to an increase in the input power supply voltage. And the pulse interval of the third pulse signal is set to be longer than the maximum value of the pulse interval of the first pulse signal or the second signal.

  In this way, the third abnormal state occurs when the pulse interval of the pulse signal to be counted exceeds the maximum value of the pulse interval between the first pulse signal and the second pulse signal (maximum value of the input power supply voltage). This abnormality can be quickly dealt with and no erroneous detection of the input power supply voltage occurs.

  A preferred embodiment of an input / output isolation type DC-DC converter device of the present invention will be described below with reference to the drawings. However, the present invention is not limited to the following embodiments, and it is obvious that the technical idea of the present invention may be realized by a combination of other known techniques or techniques having functions equivalent thereto.

  FIG. 1 shows a block circuit diagram of the vehicle power supply device of the two power supply system of this embodiment.

(Description of the entire circuit)
The input / output isolation type DC-DC converter apparatus of Example 1 will be described with reference to FIG.

  DESCRIPTION OF SYMBOLS 1 is a high voltage battery (DC power supply said by this invention), 2 is a 1st main power switch, 4 is an inrush current limiting resistor, 5 is a 2nd main power switch, 6 is a smoothing capacitor, 7 is a three phase A three-phase inverter circuit that feeds three-phase AC power to the AC motor M, 8 is a sub power switch composed of a MOS transistor, and 9 is a voltage conversion circuit that applies a low-voltage power supply voltage to the three-phase inverter circuit 7.

  The voltage conversion circuit 9 includes an input / output isolation type DC-DC converter circuit 90 having a built-in transformer, a controller 91 for controlling the DC-DC converter circuit 90, an input state monitor circuit 92 for detecting circuit protection information and outputting the information to the controller 91, a circuit It has one photocoupler 93 that transmits the protection information from the input state monitor circuit 92 to the controller 91 so that the input and output can be isolated.

  The positive terminal of the battery 1 is connected to the high potential side power input terminal of the voltage conversion circuit 9 through the main power switch 2, and the negative terminal of the battery 1 is connected to the high potential side power input terminal of the voltage conversion circuit 9 through the main power switch 5. It is connected to the. The sub power switch 8 and the inrush current limiting resistor 4 are connected in series to form an inrush current limiting circuit, and this inrush current limiting circuit is connected in parallel with the main power switch 2.

  The DC-DC converter circuit 90 can be composed of various known DC-DC converter circuits with a built-in transformer. The DC-DC converter circuit employed in this embodiment will be described later.

  The controller 91 sets the switching element built in the DC-DC converter circuit 90 within a range allowed by the circuit protection information so that the input output voltage Vo of the DC-DC converter circuit 90 becomes a predetermined target level. PWM control. The controller 91 is supplied with power from a constant voltage circuit (not shown). The constant voltage circuit forms a DC voltage based on the potential level of the battery 1 with the output voltage of the DC-DC converter circuit 90 or the potential level of a low voltage battery (not shown) as a reference potential, and supplies the power voltage to the controller 91. Apply as When the power supply voltage for the controller 91 is formed by the power of the low voltage battery described above, the low potential level of the low voltage battery is equal to the potential level of the low potential line of the pair of output terminals of the DC-DC converter circuit 90. It is preferred that By creating the power supply voltage of the controller 91 based on the potential level of the output voltage Vo of the DC-DC converter circuit 90, various circuit processes of the controller 91 are simplified.

  The input state monitor circuit 92 receives the voltage drop ΔV of the inrush current limiting resistor 4 and the input power supply voltage Vi of the DC-DC converter circuit 90. The input state monitor circuit 92 outputs the circuit protection information formed based on the input information to the controller 91 as a pulse signal in the PWM (pulse width modulation) format. The input state monitor circuit 92 is supplied with power from a constant voltage circuit (not shown). This constant voltage circuit forms a DC voltage having a required magnitude with reference to the potential level of the battery 1 and applies it to the input state monitor circuit 92 as a power supply voltage. By creating the power supply voltage of the input state monitor circuit 92 based on the potential level of the battery 1, the processing of the voltage drop ΔV and the input power supply voltage Vi input as input information to the input state monitor circuit 92 is simplified.

  The photocoupler 93 performs signal transmission from the input state monitor circuit 92 whose power supply voltage levels are independent from each other to the controller 91 while performing input / output isolation.

  In order to input the DC power of the battery 1 to the voltage conversion circuit 9, first, the main power switch 5 and the sub power switch 8 are turned on, and the smoothing capacitor 6 is charged to a predetermined level through the inrush current limiting resistor 4. The main power switch 2 is turned on. Thereby, it is possible to prevent a large inrush current from flowing into the circuit, particularly the smoothing capacitor 6.

  In FIG. 1, the auxiliary power switch 8 and the inrush current limiting resistor 4 are disposed on the positive terminal side of the battery 1, but may be disposed on the negative terminal side of the battery 1. Further, the sub power switch 8 may be an IGBT, a bipolar transistor, or an electromagnetic relay in addition to the MOS transistor shown in FIG.

  When DC power is input from the battery 1 to the voltage conversion circuit 9, the DC-DC converter circuit 90 performs PWM control of the built-in switching element to converge the output voltage Vo to the target voltage, and supplies power to the electric load.

(Description of DC-DC converter circuit 90)
Next, the DC-DC converter circuit 90 employed in this embodiment will be described with reference to FIG. First, the circuit configuration will be described.

  The DC-DC converter circuit 90 includes transformers T1, T2, switching elements Q1, Q2, Q3, Q4, and capacitors C1, C2, C3. The duty ratio of the switching elements Q1 to Q4 is PWM-controlled by the controller 91 based on the deviation between the output voltage Vo and the target voltage value. The transformer T1 has primary coils W1, W2 and a secondary coil W3, and the transformer T2 has primary coils W4, W5 and a secondary coil W6. The capacitor C3 is an output side smoothing capacitor for reducing ripple. Switching elements Q3 and Q4 constitute a rectifier circuit. Reference numeral 10 denotes an input terminal on the high potential side of the DC-DC converter circuit 90. Reference numeral 20 denotes an input terminal on the low potential side of the DC-DC converter circuit 90, which is also a connection point between the main switch Q1 and the capacitor C1. Reference numeral 30 denotes a connection point between the capacitor C1, the capacitor C2, and the primary coil W2, and reference numeral 40 denotes a connection point between the main switch Q1, the sub switch Q2, the primary coil W4, and the primary coil W5. The coils W1, W2, W4, W5, the capacitors C1, C2, and the switching elements Q1, Q2 constitute the input side circuit 11 together with the main power switches 2, 5, the inrush current limiting resistor 4, and the sub power switch 8. . The coils W3 and W6, the output smoothing capacitor C3, and the switching elements Q3 and Q4 constitute an output side circuit 21.

  A circuit portion belonging to the DC-DC converter circuit 90 in the input side circuit 11 will be described. If the dead time is ignored, the main switch Q1 and the sub switch Q2 are alternately operated, and the main switch Q1 is turned on when the circuit current flows other than the current passing through the parasitic diode D of the MOS transistor constituting the switching elements Q1 and Q2. It is easy to think based on the main switch circuit formed when the sub switch Q2 is turned on and the sub switch circuit formed when the sub switch Q2 is turned on. The main switch circuit includes a connection point 10, primary coils W1, W4, a main switch Q1, a connection point 20, a first circuit unit connecting the battery 1, and a connection point 20, a capacitor C1, coils W2, W5, a main switch Q1, And a second circuit portion connecting the connection points 20. The sub-switch circuit includes a connection point 10, coils W1, W4, connection point 40, sub-switch Q2, capacitor C2, connection point 30, capacitor C1, connection point 20, third circuit unit connecting battery 1, and connection point 40. The sub-switch Q2, the capacitor C2, the coils W2 and W5, and the fourth circuit portion connecting the connection point 40 are included. In addition, a capacitor C1 charging circuit is formed from the battery 1 to the battery 1 through the primary coils W1 and W4, the primary coils W2 and W5, and the capacitor C1.

  The output side circuit 21 will be described. The output side circuit of this embodiment is constituted by a conventionally known synchronous rectification smoothing circuit. Since the output switches Q3 and Q4 operate in principle, they are divided into a fifth circuit portion formed when the output switch Q3 is turned on and a sixth circuit portion formed when the output switch Q4 is turned on. Can do. The fifth circuit unit is a circuit connecting the connection point 60, the output switch Q3, the coil W6, the capacitor C3, and the connection point 60. The sixth circuit unit is the connection point 60, the output switch Q4, the coil W3, the capacitor C3, and the connection. A circuit connecting the points 60. Reference numeral 50 denotes an output terminal of the output side circuit 21. Either or both of the output switches Q3 and Q4 may be replaced with a diode. The output switch Q3 has almost the same operation state as the main switch Q1, and the output switch Q3 has the same operation state as the sub switch Q2. The operation timing of the switching elements Q1 to Q4 is shown in FIG. 3, and the voltage change of the coils W1 to W6 is schematically shown in FIG.

  The operation of the above-described DC-DC converter circuit 90 will be described.

  As shown in FIG. 3, the switching elements Q1 and Q3 operate synchronously, the switching elements Q2 and Q4 operate synchronously, the switching elements Q1 and Q2 operate complementarily, and the switching elements Q3 and Q4 operate complementary. A state in which the main switch Q1 is on and the sub switch Q2 is off is referred to as mode 1, and a state in which the main switch Q1 is off and the sub switch Q2 is on is referred to as mode 2. The input current from the battery 1 is called i1, the charging / discharging current of the capacitor C1 is called i2, and the charging / discharging current of the capacitor C2 is called i3. In mode 1, switching elements Q1, Q3 are turned on, and switching elements Q2, Q4 are turned off. In mode 2, switching elements Q1, Q3 are turned off, and switching elements Q2, Q4 are turned on.

  Mode 1 in which the main switch Q1 is turned on and the sub switch Q2 is turned off will be described.

  When the main switch Q1 is turned on, the current i1 that has passed through the primary coils W1 and W4 from the battery 1 is commutated from the flow toward the capacitor C1 via the primary coils W5 and W2 in the preceding mode 2 to be directly connected. Go to point 20. Thereby, the input current i1 increases with time, and in the transformer T2, a magnetomotive force (ampere turn) i1 of the primary coil W4 and a magnetomotive force (ampere turn) i2 of the primary coil W5 are formed. Further, the capacitor C1, which is charged in mode 2 to be described later and has a high voltage, is discharged with the current i2 through the primary coils W2, W5 and the main switch Q1. This current i2 becomes a flow that increases with time in the discharge direction. In this embodiment, the direction of the magnetic flux formed by the magnetomotive force (ampere turn) i1 of the primary coil W4 is made equal to the direction of the magnetic flux formed by the magnetomotive force (ampere turn) i2 of the primary coil W5. As a result, a magnetic flux φ2 corresponding to the sum (i1 + i2) of these magnetomotive forces (ampere turns) is formed. A secondary voltage V6 having a magnitude corresponding to the change of the magnetic flux φ2 is formed in the secondary coil W6. The winding direction of the secondary coil W6 is the direction in which the current i3 is output in mode 1. If the load 3 is regarded as a resistance, a current i3 having a magnitude proportional to the secondary voltage V6 flows. Therefore, ideally, this current i3 becomes a substantially direct current having a predetermined amplitude during this mode period. In response to the outflow of the current i3 from the transformer T2, the current i1 flowing through the primary coil W4 and the current i2 flowing through the primary coil W5 increase. That is, since the current i1 flowing through the primary coil W4 and the current i2 flowing through the primary coil W5 correspond to the exciting current and current i3 of the transformer T2, the exciting current of the transformer T2 is subtracted from the current i2 flowing through the primary coil W5. And the current i1 flowing through the primary coil W4 are output from the secondary coil W6 as a current i3. Eventually, during the period in which the main switch Q1 is on, in the transformer T2, an increase in magnetomotive force (ampere turn) in the first direction due to an increase in current in the primary coil W4 and a current in the discharge direction of the primary coil W5. A secondary voltage is formed in the secondary coil W6 by the combined magnetomotive force (ampere turn), which is the sum of the increase in magnetomotive force (ampere turn) due to the increase in current, and the current i3 is output. At this time, the transformer T1 operates as a choke coil. That is, during the period in which the main switch Q1 is on, in the transformer T2, an increase in magnetic flux due to an increase in current in the primary coil W4 and an increase in magnetic flux due to an increase in current in the charging direction of the primary coil W5 that promotes the increase in magnetic flux. The current i3 is output from the secondary coil W6, and the power energy for outputting the current i3 is supplied from the primary coils W4 and W5.

  A description will be given with reference to mode 2 in which the main switch Q1 is turned off and the sub switch Q2 is turned on.

  The current i1 from the battery 1 through the primary coils W1 and W4 to the connection point 40 tends to decrease due to an increase in the potential at the connection point 40 due to the main switch Q1 being turned off. A current i2 'flowing from the connection point 40 through the primary coils W5 and W2 charges the capacitor C1. The current ic2 flowing from the battery 1 through the primary coils W1 and W4 flows through the capacitor C2 and charges the capacitor C1. Therefore, the current i1 is equal to the sum of i2 'and ic2, that is, the charging current i2 of the capacitor C1. At this time, the magnetic flux in a certain direction of the transformer T1 decreases due to the decrease of the current i1 of the primary coil W1. At this time, the current change in the primary coil W2, that is, the change from the discharge current i2 shown in FIG. 7 to the charge current i2 ′ shown in FIG. 8 is directed to the direction of the transformer T1 due to the decrease in the current i1 of the primary coil W1. The direction is to promote the reduction of the magnetic flux. As a result, the secondary coil W3 outputs the current i4 due to the change (decrease from increase) of the current i1 in the primary coil W1 and the current change (i2 ′ in the opposite direction from i2) in the primary coil W2. A voltage V3 is generated. This voltage is a substantially DC voltage having a magnitude proportional to the rate of change of magnetic flux due to the current change. The winding direction of secondary coil W3 is the direction in which current i4 is output in mode 2. If the load 3 is a resistance, a current i4 proportional to the magnitude of the voltage V4 flows. Ideally, this current i4 becomes a substantially direct current having a predetermined amplitude during this mode period. In response to the outflow of the current i4 from the transformer T1, the current i1 flowing through the primary coil W1 and the current i2 ′ flowing through the primary coil W2 increase, so that the exciting current of the transformer T1 from the current i2 ′ flowing through the primary coil W2 And the current i1 flowing through the primary coil W1 are output as the current i4 from the secondary coil W3. Eventually, during the period in which the main switch Q1 is off, in the transformer T1, it is caused by a decrease in magnetomotive force (ampere turn) due to a decrease in the current of the primary coil W1, and an increase in current in the charging direction of the capacitor C1 of the primary coil W2. A secondary voltage is formed in the secondary coil W3 by the combined magnetomotive force (ampere turn), which is the sum of the increase in magnetic force (ampere turn), and the current i4 is output. At this time, the transformer T2 operates as a choke coil. That is, during the period in which the main switch Q1 is off, the transformer T1 has a magnetic flux reduction due to the current reduction of the primary coil W1, and a magnetic flux reduction due to a current change in the discharge direction of the primary coil W2 that promotes the magnetic flux reduction. The current i4 is output from the secondary coil W3, and the power for outputting the current i4 is supplied from the primary coils W1 and W2. Capacitor C2 and sub switch Q2 are essentially a clamp circuit that prevents the generation of surge voltage when main switch Q1 is off. By increasing the duty ratio of the switching element Q1, the current flowing from the battery 1 into the DC-DC converter circuit 90 increases, and the output voltage Vo of the DC-DC converter circuit 90 increases. Therefore, the controller 91 reads the output voltage Vo of the DC-DC converter circuit 90, compares the output voltage Vo with a pre-stored target voltage, and when the output voltage Vo is smaller than the target voltage, the controller 91 When the on-duty ratio is increased and the output voltage Vo is higher than the target voltage, feedback PWM control is performed to decrease the on-duty ratio of the main switch Q1, and the output voltage Vo is converged to the target voltage.

(Duty ratio limiting operation of switching element based on input power supply voltage)
Next, the protection operation of the DC-DC converter circuit 90, particularly the switching element Q1, will be described.

  The voltage Vds between the source and the drain of the switching element Q1 is substantially equal to Vi / (1-D) when the input power supply voltage is Vi and the duty ratio (on-duty ratio) of the switching element Q1 is D. That is, the maximum value Vdsmax of the source-drain voltage Vds of the switching element Q1 does not exceed Vin / (1-Dmax) when the on-duty ratio D of the switching element Q1 at that time is Dmax. Therefore, if the maximum value Dmax of the on-duty ratio D is decreased as the input power supply voltage Vi increases, the maximum value Vdsmax is prevented from exceeding a constant value, that is, the breakdown voltage Vdsth between the source and drain of the switching element Q1. be able to. Of course, when the input power supply voltage Vi is small, the allowable maximum value of the duty ratio (Duty) can be set large, and when the input power supply voltage Vi is large, the allowable maximum value of the duty ratio (Duty) must be set small. FIG. 5 shows a combination of the allowable range (Duty change permission range) of the duty ratio (Duty) that allows the source-drain voltage Vds of the switching element Q1 to be equal to or lower than the maximum value Vdsmax and the input power supply voltage Vi.

  In FIG. 5, Vmax is a theoretical maximum withstand voltage value (corresponding to Vdsmax) of the switching element Q1, and Vxx is a practical maximum withstand voltage value obtained by multiplying Vmax by a predetermined safety factor (a constant smaller than 1). DL is the practical maximum value of the duty ratio (Duty) when the input power supply voltage Vi is the practical maximum withstand voltage value Vxx, and is determined from the maximum possible duty ratio (Duty) in view of the withstand voltage of the switching element Q1. The duty ratio is set to a small value by Δd. Similarly, DH is the practical maximum value of the duty ratio (Duty) when the input power supply voltage Vi is very small, and the predetermined duty ratio is determined from the maximum value of the duty ratio (Duty) theoretically possible in view of the breakdown voltage of the switching element Q1. The value is set to be smaller by Δd. As a result, it can be seen that the switching element Q1 can be used in the hatched area in FIG.

  An example of the duty ratio restriction control of the switching element Q1 according to the input power supply voltage Vi performed by the controller 91 will be described with reference to the flowchart shown in FIG.

  First, the input power supply voltage Vi is read (S200), and the duty corresponding to the input power supply voltage Vi is determined from a table indicating the relationship between the input power supply voltage Vi stored in advance and the maximum value Dmax of the allowable duty ratio (on-duty ratio) D. The maximum value Dmax of the ratio D is extracted (S202), and the current duty ratio D command is read (S204). As will be described later, the duty ratio of the pulse signal input from the input state monitor circuit 92 may be used as it is as the maximum value Dmax of the duty ratio D without using the above table. The pulse width of the pulse signal to be used may be adopted as the pulse width at the maximum value Dmax of the duty ratio D.

  Next, it is determined whether or not the current duty ratio D is larger than the maximum value Dmax of the duty ratio D (S206). If the current duty ratio D is larger, the maximum value Dmax is adopted as the next duty ratio D. The ratio D is adopted as the next duty ratio D (S210). By doing so, it is possible to prevent the breakdown voltage of the switching element Q1 from being suddenly increased by the input power supply voltage Vi. Of course, it can be naturally processed by a hardware circuit having an equivalent function instead of the software processing shown in FIG.

(Description of Input Status Monitor Circuit 92)
Next, an example of the input state monitor circuit 92 will be described with reference to FIG. However, it is a matter of course that the circuit function of the input state monitor circuit 92 may be realized by other hardware circuits or by software processing. For example, instead of providing a plurality of oscillators with different oscillation periods, the oscillation period of a common oscillator may be changed as necessary, or the oscillators may be oscillated only when oscillation is required, instead of always oscillating. Good.

  The input state monitor circuit 92 includes a leakage current detection circuit 100, an overvoltage detection circuit 200, a sawtooth pulse oscillator 300, 400, a rectangular wave pulse oscillator 500, a multiplexer 600, a resistance voltage dividing circuit 700, a comparator 800, various logic gates 900, 1000, 1100, 1200, 1300. The sawtooth pulse oscillators 300 and 400, the multiplexer 600, and the comparator 800 constitute a PWM pulse signal generation circuit, but the PWM pulse signal may be generated by another known circuit.

  The leakage current detection circuit 100 outputs a leakage current detection signal S1 having a high level when the leakage current of the sub power switch 8 is large, and a low level when the leakage current is small, to the multiplexer 600 and the NAND gate 1300. The overvoltage detection circuit 200 outputs an overvoltage detection signal S2 having a high level when the voltage of the battery 1 is abnormally high, and outputs a low level overvoltage detection signal S2 to the AND gate 1000 and the NAND gate 1300 otherwise, to the AND gate 900 through the NOT circuit 1400. Output. The sawtooth pulse oscillator 300 oscillates a sawtooth pulse at a period of 10 μsec, the sawtooth pulse oscillator 400 oscillates a sawtooth pulse at a period of 20 μsec, and the rectangular wave pulse oscillator 500 oscillates a rectangular wave pulse at a period of 40 μsec. The multiplexer 600 outputs the output of the oscillator 300 to the comparator 800 when the leakage current is small and the leakage current detection signal S1 is low level. When the leakage current is large and the leakage current detection signal S1 is high level, the multiplexer 600 The output is output to the comparator 800. As a result, when the leakage current is small, the comparator 800 outputs a duty ratio (Duty) proportional to the magnitude of the battery voltage (input power supply voltage) Vi at a cycle of 10 μsec. Further, the comparator 800 outputs a PWM pulse signal having a duty ratio (Duty) proportional to the magnitude of the battery voltage (input power supply voltage) Vi at a cycle of 20 μsec when the leakage current is large.

  The overvoltage detection circuit 200 outputs the PWM pulse signal from the comparator 800 to the AND gate 1200 through the OR circuit 1100 in order to open the AND gate 900 through the NOT circuit 1400 and close the AND gate 1000 when the voltage of the battery 1 is not abnormally high. To do. Since the NAND gate 1300 opens the AND gate 1200 when the leakage current is small and the voltage of the battery 1 is not abnormally high, the PWM pulse signal is output to the photocoupler 93 through the AND gate 1200 and a buffer amplifier (not shown). Further, when the voltage of the battery 1 is abnormally high, the overvoltage detection circuit 200 closes the AND gate 900 and opens the AND gate 1000, so that a rectangular wave pulse signal (duty ratio 50%) having a period of 40 μsec is generated from the oscillator 500. And output to the AND gate 1200. Since the NAND gate 1300 opens the AND gate 1200 when the leakage current is small and the voltage of the battery 1 is abnormally high, the rectangular pulse signal is output to the photocoupler 93 through the AND gate 1200 and a buffer amplifier (not shown).

  Next, when the leakage current is large and the voltage of the battery 1 is abnormally high, the output of the NAND gate 1300 becomes low level and the AND gate 1200 is closed, and the low level is output to the photocoupler 93, and the input state monitor Signal transmission from the circuit 92 to the controller 91 is stopped. By transmitting information from the input state monitor circuit 92 described above to the controller 91, the controller 91 receives a PWM pulse signal having a duty ratio corresponding to the magnitude of the input power supply voltage Vi, and at the same time, the leakage current of the sub power switch 8 and An abnormally high voltage state of the input power supply voltage Vi can be monitored only by using the single photocoupler 93.

(Description of Leakage Current Detection Circuit 100 and Overvoltage Detection Circuit 200)
Next, an example of the leakage current detection circuit 100 and the overvoltage detection circuit 200 will be described with reference to FIG. The inrush current limiting circuit may be provided in parallel with the main power switch 2 or the inrush current limiting circuit may be provided in parallel with the main power switch 5 as shown in FIG. In FIG. 8, X is a connection point between the sub power switch 8 and the inrush current limiting resistor 4, Y is a connection point between the sub power switch 8, the battery 1, and the main power switch 5, and Z is an inrush current limiting resistor 4. This is a connection point with the main power switch 5. In FIG. 8, 100 is a leakage current detection circuit, and 200 is an overvoltage detection circuit. A constant power supply voltage circuit 240a generates a power supply voltage based on a low potential level of the voltage of the battery 1, and applies the power supply voltage to an operational amplifier and a comparator which will be described later. Reference numeral 240b denotes a constant power supply voltage circuit that applies a power supply voltage to a voltage dividing circuit for generating a threshold voltage. The resistor voltage dividing circuit includes resistors 310 and 320 connected in series and resistors 210 and 220 connected in series. A power supply voltage is applied to the resistance voltage dividing circuit.

  The leakage current detection circuit 100 includes an inverting voltage amplification circuit 250 and a comparator 290. The inverting voltage amplification circuit 250 includes an operational amplifier type inverting voltage amplification circuit including an input resistor 260, a feedback resistor 270, and an operational amplifier 280. ing. The potential Vx at the connection point X is input to the negative input terminal of the operational amplifier 280 through the input resistor 260, and the positive input terminal of the operational amplifier 280 is connected to the connection point Z. The signal voltage Vs output from the operational amplifier 280 is input to the negative input terminal of the comparator 290, and the threshold voltage Vth is input to the positive input terminal of the comparator 290. This threshold voltage Vth is formed by a resistance voltage dividing circuit formed by connecting resistors 310 and 320 in series. The voltage amplification factor of the inverting voltage amplifier circuit 250 is defined by the resistance ratio between the input resistor 260 and the feedback resistor 270.

  The leakage current detection operation of the leakage current detection circuit 100 will be described assuming that the potential at the connection point Z is 0V. With the main power switch 2 turned on and the main power switch 5 and the sub power switch 8 turned off, the potential Vx at the connection point X is read into the negative input terminal of the operational amplifier 280 through the input resistor 260. Since the potential at the connection point Z is assumed to be 0V, the potential at the connection point X is relatively negative, and the signal voltage Vs of the operational amplifier 280 is +. The comparator 290 compares the signal voltage Vs with the threshold voltage Vth, and when the signal voltage Vs is larger than the threshold voltage Vth, a low level potential (for example, 0 V) indicating that the leakage current of the sub power switch 8 is abnormally large. ) Is output, otherwise a high level potential (positive potential) is output. According to this leakage current detection circuit 100, it is considered that the potential at the negative input terminal of the operational amplifier 280 is virtually grounded even though the signal potential Vx is a negative potential (when the connection point Z is set to 0V). In addition, since the output voltage Vs of the inverting voltage amplification circuit 250 swings to the positive side, a power supply voltage in the range of 0 to a predetermined positive potential Vd is applied to the inverting voltage amplification circuit 250 and the comparator 290 from the control power supply 240a. Can do.

  The overvoltage detection circuit 200 is a circuit that determines whether or not the input power supply voltage Vi exceeds a predetermined maximum allowable voltage value. The overvoltage detection circuit 200 includes a voltage dividing circuit including resistors 180 and 190, a voltage dividing circuit including resistors 210 and 220, and The comparator 230 is configured. The voltage of the battery 1 is divided by a voltage dividing circuit including resistors 180 and 190 and input to the + input terminal of the comparator 230. A resistance voltage dividing circuit including resistors 210 and 220 applies a threshold voltage to the negative input terminal of the comparator 230. The comparator 230 outputs a high level as an overvoltage determination signal that warns when the voltage of the battery 1, that is, the input power supply voltage Vi exceeds the threshold voltage. D1 and D2 are clamp diodes for preventing the signal voltage Vx input to the negative input terminal of the inverting amplifier 280 from becoming abnormally high or abnormally low due to some factor by the clamping action. .

  Next, the corresponding operation of the controller 91 based on the monitor information input through the photocoupler 93 will be described. Based on the single pulse signal input through the photocoupler 93, the controller 91 performs the voltage protection of the switching element Q1 and the monitoring of the sub power switch 8 in the following four different modes shown in FIG. The A mode is a case where the leakage current is large and the input power supply voltage Vi is not abnormally high. The controller 91 performs PWM control of the DC-DC converter circuit 90 and warns the outside of an increase in leakage current of the sub power switch 8. The B mode is a case where the leakage current is small and the input power supply voltage Vi is not abnormally high. The controller 91 performs PWM control of the DC-DC converter circuit 90 and does not warn of an increase in leakage current of the sub power switch 8 to the outside. The C mode is a case where the leakage current is small and the input power supply voltage Vi is abnormally high. When the input power supply voltage Vi is abnormally high, the voltage applied to the switching element Q1 may exceed the allowable range. Therefore, the controller 91 stops the PWM control of the DC-DC converter circuit 90 and externally It does not warn of an increase in leakage current of the sub power switch 8. The D mode is a case where the leakage current is large and the input power supply voltage Vi is abnormally high. The controller 91 stops the PWM control of the DC-DC converter circuit 90 and warns the outside of the increase in the leakage current of the sub power switch 8. Since such control by the controller 91 can be easily realized by microcomputer control or the like, the flowchart is not shown.

(Additional explanation of PWM control of switching element Q1)
Next, feedback PWM control of the switching element Q1 by the controller 91 in this embodiment will be additionally described. In this embodiment, the comparator 800 shown in FIG. 7 is set to output a PWM pulse signal having a duty ratio having a quantity relationship corresponding to the oblique line L1 shown in FIG. 5 with respect to the input power supply voltage Vi. Vxx corresponds to the threshold voltage whether or not the input power supply voltage Vi mentioned above is abnormally high. Thereby, the controller 91 forcibly inputs the PWM through the photocoupler 93 when the duty ratio of the PWM pulse signal output to the switching element Q1 exceeds the duty ratio of the PWM pulse signal input through the photocoupler 93. Only by limiting the duty ratio of the pulse signal, the voltage applied to the switching element Q1 can be within the limit. Further, when a rectangular wave pulse signal (duty ratio 50%) with a period of 40 μsec is received through the photocoupler 93 and when no pulse signal is received, switching of the switching element Q1 is stopped.

(Example effect)
According to this embodiment, an excessive state of the input power supply voltage Vi is detected and the PWM control of the switching element Q1 is prohibited. An excessive state of the input power supply voltage Vi is not detected and the switching element Q1 can be PWM controlled. In addition, the period of the pulse signal input to the controller 91 through the photocoupler 93 is set to be long. Thereby, the state in which the switching element Q1 can be controlled can be detected quickly.

  Further, in the D mode in which no pulse signal is input from the photocoupler 93, the duty ratio of the pulse signal is 0. Therefore, the duty ratio of the switching element Q1 that is always suppressed to be equal to or lower than the duty ratio of the pulse signal is the controller. Since it can be easily limited to 0 by a hardware circuit before the input power supply voltage Vi is excessively recognized by 91, the switching element Q1 is well protected.

(Modification 1)
In the above embodiment, a PWM pulse signal having a duty ratio corresponding to the detected magnitude of the input power supply voltage is transmitted to the controller 91 through the photocoupler (input / output insulation separation circuit) 93, and this PWM pulse is determined depending on the magnitude of the leakage current. The signal cycle is changed, but instead, a PWM pulse signal having a pulse width corresponding to the detected input power supply voltage is transmitted to the controller 91 through the photocoupler (input / output isolation circuit) 93 to cause leakage current. The period of the PWM pulse signal may be changed depending on the magnitude of the.

  It is obvious that a circuit for generating a pulse signal having a pulse width corresponding to the magnitude of the voltage and a circuit for changing the period of the pulse signal can be easily realized by a hardware circuit or software processing, without needing to be described. . In this case, the controller 91 limits the duty ratio of the PWM pulse signal that controls the switching element Q1 within the range of the pulse width extracted from the pulse signal received through the photocoupler 93. The pulse width of the pulse signal output from the input state monitor circuit 92 through the photocoupler 93 is the pulse width at the maximum allowable duty ratio of the switching element Q1. If it does in this way, there can exist an effect similar to the said Example.

(Modification 2)
In Modification 1, a PWM pulse signal having a pulse width (also referred to as a pulse width period) corresponding to the input voltage Vi in modes A and B is transmitted to the controller 91 through the photocoupler 93, and this PWM is determined depending on the magnitude of the leakage current. Changed the cycle of the pulse signal. In this modified embodiment 2, a PWM pulse signal having a pulse interval period (= pulse period−pulse width period) corresponding to (proportional to) the input voltage Vi in modes A and B is transmitted to the controller 91 through the photocoupler 93, and the leakage current The period of the PWM pulse signal can also be changed depending on the magnitude of (see FIG. 10).

(Modification 3)
When the input voltage Vi is transmitted according to the length of the pulse width period or the pulse interval period described above, the controller 91 erroneously sets the pulse width period or the pulse interval period of the received pulse signal by switching the period between the mode A and the mode B. It is necessary to avoid counting. Therefore, in this modification, the erroneous count by the controller 91 is prevented by synchronizing the pulse signal voltage Sa for mode A transmission and the pulse signal voltage Sb for mode B transmission in spite of the mode switching. .

  Hereinafter, an example will be specifically described with reference to FIG. The input state monitor circuit 92 forms the pulse signal voltage for mode A transmission by thinning out the odd-numbered (or even-numbered) pulse voltage of the pulse signal voltage for mode B transmission. In this way, the oscillator 300 and the multiplexer 600 can be omitted in FIG. However, when transmitting mode B, the pulse signal voltage that has been thinned out from the comparator 800 to the AND gate 900 is passed through a thinning-out circuit that thins out the odd-numbered (or even-numbered) pulse voltage output from the comparator 800. Output. Naturally, when transmitting the mode A, the pulse signal voltage may be directly output from the comparator 800 to the AND gate 900 by bypassing this thinning circuit. In FIG. 10, St is a sawtooth pulse voltage output from the oscillator 400. It should be noted that, instead of thinning out the rectangular wave pulse voltage obtained by binarizing the sawtooth pulse voltage, it is obvious that the pulse voltage may be thinned out with a sawtooth pulse voltage waveform in the same manner as described above.

  Next, an example of reading of the input voltage Vi by the controller 91 performed in the modification 3 will be described below. First, it counts from the pulse falling edge tx to the next pulse rising edge. If it is mode B, it is regarded as a signal corresponding to the input voltage Vi, and if it is mode A, 10 μs is subtracted therefrom and regarded as a signal corresponding to the input voltage Vi. Just do it. In either mode, the pulse interval start time of the controller 91 is generated at the same timing, so that the pulse interval time counting process that is the basis for calculating the input voltage Vi is simplified, and this is not mistaken.

  That is, in this modification, the pulse falling edge, which is the count start point of the pulse interval corresponding to the input voltage (input power supply voltage) Vi, is set to substantially the same timing with the two pulse signals, so the pulse signal switching In this case, the pulse interval can be accurately counted, and the reliability of detection of the input power supply voltage can be further improved.

(Modification 4)
In this modified embodiment 4, as shown in FIG. 10, the period of the C mode is made longer than the period of the A mode and the period of the B mode. Thereby, each period is not erroneously determined. Further, in this modification, when switching from mode A and mode B to mode C, the mode related to the input voltage Vi counted by the controller 91 is never the time corresponding to the input voltage Vi. A Set the pulse signal voltage timing for transmission. That is, for example, while the controller 91 counts the elapsed time from the falling point of the pulse signal voltage for mode A or mode B as the time corresponding to the input voltage Vi, the pulse of the pulse signal Sc for mode C transmission When the rising edge is input, the time corresponding to the input voltage Vi is determined thereby, and the controller 91 calculates an incorrect duty ratio. Therefore, in this modification, at the timing delayed beyond the value corresponding to the maximum value of the duty ratio from the falling edge of the pulse signal voltage for mode A or B transmission (when counting of the time corresponding to the input voltage Vi is started). Control is performed so that a rising edge of the pulse signal Sc for mode C transmission occurs. Thereby, the said problem can be solved.

  That is, the pulse period of the pulse signal Sc for mode C transmission is set longer than the pulse period of the pulse signal Sa for mode A transmission or the pulse signal Sb for mode B transmission, and the pulse interval Tc is longer than the pulse interval Tb. Therefore, the mode C can be detected at the time of transition from the pulse signal Sa for mode A transmission or the pulse signal Sb for mode B transmission to the pulse signal Sc for mode C transmission, and the input power supply voltage There is no false detection.

(Modification 5)
In this modified embodiment 5, as shown in FIG. 10, the C-mode pulse interval Tc is longer than the A-mode pulse interval Ta and the B-mode pulse interval Tb. Thereby, the pulse signal Sc for mode C transmission corresponds to the normal input voltage Vi when the controller 91 counts the time corresponding to the input voltage Vi from the pulse signal Sc for mode C transmission. Will never end in time. More specifically, referring to FIG. 10, the controller 91 starts counting the time corresponding to the input voltage Vi from the pulse falling edge tx, and determines the time corresponding to the input voltage Vi at the next pulse rising edge to be input. To do. In this embodiment, the pulse interval period Tc of the pulse signal Sc for mode C transmission (= the pulse period of the pulse signal voltage for mode A transmission) is the maximum value of the pulse interval period Tb of the pulse signal voltage Sa for mode A transmission. Therefore, the controller 91 does not mistake the pulse interval period counted by the controller 91 as the time corresponding to the input voltage Vi by transmitting the pulse signal Sc. Further, it becomes possible to quickly detect the transition to mode C.

  Note that, finally, when the pulse voltage is inverted by an inverter, the pulse width period becomes a pulse interval period, the pulse interval period becomes a pulse width period, and the pulse rising edge and the pulse falling edge are reversed. Such a change is naturally included in the category of the above embodiment.

It is a circuit diagram which shows the Example of the input-output insulation isolation type DC-DC converter apparatus of this invention. It is a circuit diagram of the DC-DC converter circuit of FIG. 3 is a timing chart showing an operation state of a switching element of the DC-DC converter circuit of FIG. 2. FIG. 3 is a schematic waveform diagram of each part voltage of the DC-DC converter circuit of FIG. 2. It is a figure which shows the safe duty ratio range of the switching element of FIG. It is a flowchart which shows the control for keeping the switching element of FIG. 2 in the safe duty ratio range. FIG. 2 is a circuit diagram illustrating an example of an input state monitor circuit of FIG. 1. It is a circuit diagram which shows an example of the leakage current detection circuit of FIG. 7, and an overvoltage detection circuit. It is a figure which shows the operation mode of the controller of FIG. It is a timing chart for demonstrating a deformation | transformation aspect.

Explanation of symbols

1 Battery (input power)
2 Main power switch 4 Inrush current limiting resistor 5 Main power switch 6 Smoothing capacitor 7 Three-phase inverter circuit 8 Sub power switch 9 Voltage conversion circuit 90 DC-DC converter circuit 91 Controller 92 Input state monitor circuit 93 Photocoupler 100 Current detection circuit 200 Overvoltage detection circuit

Claims (8)

An input side circuit that intermittently switches the switching element and converts an input power supply voltage applied from a DC power source into an AC voltage and outputs the output voltage, an output side circuit that rectifies the AC voltage and then outputs the output voltage to the outside, And a DC-DC converter circuit having a transformer that transforms the input side circuit from the input side circuit to the output side circuit and transmits AC power while electrically isolating the output side circuit from the input side circuit;
An input state monitor circuit that outputs a pulse signal having a pulse width or an on-duty ratio according to the detected magnitude of the input power supply voltage through an input / output isolation circuit;
A controller that operates with a power supply voltage based on a potential level of an output voltage of the DC-DC converter circuit and controls the switching element in accordance with the input power supply voltage and the output voltage of the DC-DC converter circuit;
With
The input state monitor circuit includes:
When an abnormal state of the input side circuit is detected, the pulse period of the pulse signal output by the input state monitor circuit is changed,
The controller is
An input / output insulation-separated DC-DC converter device, wherein an abnormal state of the input side circuit is determined by determining the change of the pulse period. In the input / output isolation type DC-DC converter device according to claim 1,
The input side circuit is:
An inverter circuit that incorporates the switching element and converts the input power supply voltage to the AC voltage; a main power switch that opens and closes an electrical connection between the DC power supply and the inverter circuit; An inrush current limiting resistor circuit configured by a power switch and an inrush current limiting resistor and connected in parallel to the main power switch;
The input state monitor circuit includes:
The input / output isolation type, wherein the pulse cycle is changed when an abnormal current state in which the current flowing through the sub power switch exceeds a predetermined threshold is detected when the sub power switch is turned off. DC-DC converter device. In the input / output isolation type DC-DC converter device according to claim 1,
The input state monitor circuit includes:
An input / output isolation type DC-DC converter device, wherein the pulse period is changed when an overvoltage input state in which the input power supply voltage exceeds a predetermined threshold is detected. In the input / output isolation type DC-DC converter device according to claim 2 and 3,
The input state monitor circuit includes:
The A mode in which the leakage current abnormal state is detected and the overvoltage input state is not detected, the B mode in which the leakage current abnormal state is not detected and the overvoltage input state is not detected, and the leakage current abnormal state Are detected, and the overvoltage input state is detected in the C mode, and the leakage current abnormal state is detected, and the overvoltage input state is detected in the D mode. An input / output isolation type DC-DC converter device characterized by assigning different pulse periods to each other. In the input / output isolation type DC-DC converter device according to claim 4,
The input state monitor circuit includes:
In the A and B modes, a PWM pulse signal having a duty ratio or a pulse width that decreases as the input power supply voltage increases is transmitted to the controller,
The controller is
The PWM control applied to the switching element of the DC-DC converter for making the output voltage of the DC-DC converter circuit constant is the duty ratio or pulse width of the PWM pulse signal received from the input state monitor circuit. An input / output isolation type DC-DC converter device characterized by being limited within a range. In the input / output isolation type DC-DC converter device according to claim 1,
The input state monitor circuit includes:
A first pulse signal having a pulse interval that changes in response to an increase in the input power supply voltage, and a first period indicating a first abnormal state;
A second pulse signal having a pulse interval that changes in response to an increase in the input power supply voltage and a second period that indicates a second abnormal state;
Select one of these to output,
The input / output isolation type DC-DC converter device characterized in that the pulse falling edge of the first pulse signal and the pulse falling edge of the second pulse signal are set at the same timing. In the input / output isolation type DC-DC converter device according to claim 6,
The input state monitor circuit has a pulse interval that does not change according to an increase in the input power supply voltage, and indicates any one of the third pulse signal indicating the third abnormal state, the first signal, and the second signal. Select and output
The pulse period of the third pulse signal is set longer than the pulse period of the first pulse signal or the second signal,
The pulse interval of the pulse signal received by the controller at the time of transition from the first or second pulse signal to the third pulse signal is set in a range not corresponding to the input power supply voltage. Insulation separation type DC-DC converter device. In the input / output isolation type DC-DC converter device according to claim 6,
The input state monitor circuit has a pulse interval that does not change according to an increase in the input power supply voltage, and indicates any one of the third pulse signal indicating the third abnormal state, the first signal, and the second signal. Select and output
The input / output isolation type DC-DC converter device characterized in that the pulse interval of the third pulse signal is set longer than the maximum value of the pulse interval of the first pulse signal or the second signal.

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