JP4127706B2 - transceiver - Google Patents

Description

  The present invention relates to a wireless device such as a portable wireless terminal.

  In recent years, portable wireless terminals (hereinafter referred to as portable terminals) have been in increasing demand, and high-performance products have been commercialized.

  Trends in commercialization include miniaturization that makes it easy for people to carry around, low power consumption that lowers current consumption and lengthens talk time, and high linearization that resists interference.

  Not all problems have been solved in providing such products to the market, and further research is currently underway to solve individual problems.

  Hereinafter, a conventional wireless device will be described with reference to FIGS.

  FIG. 35 shows a transmission / reception configuration of a conventional heterodyne radio.

  In the description of this wireless device, as a receiving system, one that is actively used as a configuration of orthogonal quasi-synchronous detection using an orthogonal demodulation method is given as an example. This wireless device is premised on using a linear modulation method (π / 4 shift QPSK modulation method) used in PHS, for example.

  In the case of this wireless device, the high-frequency signal received by the antenna 101 is given a predetermined gain by the high-frequency amplifier 102 for improving the NF of the wireless unit, and then passes through the high-frequency filter 103 for image suppression. The reference carrier signal supplied from the local oscillator 125 is multiplied by the frequency converter 104 and frequency-converted to an intermediate frequency. Thereafter, the frequency is converted to the baseband frequency by the quadrature demodulating unit including the frequency converters 105 and 106, the local oscillator 107, and the π / 2 phase shifter 108. That is, a baseband signal of ΙQ is obtained by the reference signal having a phase difference of π / 2 at the same frequency as the intermediate frequency transmitted from the local oscillator 107 and the two frequency converters 105 and 106. This ΙQ baseband signal is subjected to anti-aliasing processing (filtering) for the A / D converters 113 and 114 in the subsequent stages by the low-frequency filters 109 and 110. Note that channel selection may be performed by the low-frequency filters 109 and 110.

  The filtered IQ signal is given a predetermined gain by the low frequency amplifiers 111 and 112. Thereafter, the signals are converted into digital signals by the A / D converters 113 and 114 and demodulated into data signals by the detector 117. In FIG. 35, capacitors 119 to 124 following the frequency converters 105 and 106 are inserted for the purpose of removing DC (direct current) components.

  Next, a conventional radio transmission system will be described.

  The orthogonal digital signals Ich and Qch from the baseband signal generator are converted into analog signals by the respective digital / analog converters D / A 161 and 162. The converted signals are input to the root roll-off filters 159 and 160, so that unnecessary waves generated in the D / A 161 and 162 are removed and a first operation for removing interference between symbols is performed. Is done. The second operation after the first operation is performed by a roll-off filter of the reception system.

  By the quadrature modulator including the multipliers 157 and 158, the local oscillator 163 (synthesizer), the π / 2 phase shifter 163, and the adder 156, the output of the local oscillator is modulated by the baseband signal to generate an IF signal. .

  Since this IF signal is frequency-converted to the same frequency as the transmission frequency, the IF signal is input to the local oscillator 125 and the frequency converter 154 via the low-pass filter 155 and is frequency-converted to an RF signal. The RF signal is output from the antenna 101 via the bandpass filter 153, the variable attenuator 152, the power amplifier 151, the directional coupler 172, the bandpass filter 150, and the transmission / reception switch 170. The transmission power is sensed by the directional coupler 172 between the power amplifier 151 and the filter 150, and the transmission power control signal is transmitted from the control signal unit 171 based on the power detected by the power detector 173. Is input. The low-pass filter and the band-pass filters 155, 153, and 150 are used to avoid unnecessary radiation.

  Hereinafter, the problems of such a conventional wireless device will be described in the order of a reception system, a synthesizer (meaning the local oscillator described above), a transmission system, and an antenna.

  First, problems of the conventional receiving system will be described.

  There are two problems with the conventional reception system. The first problem is that the current consumption increases, and the second problem is that a DC offset (hereinafter referred to as DC offset) occurs during reception. This is to deteriorate the reception characteristics.

  First, the first problem will be described. Since the reception characteristics of the radio device must always satisfy the performance required for the system, it is designed to cope with the worst radio wave environment where the radio device is placed. An example of the worst radio wave environment mentioned here is a case where an unnecessary wave defined by, for example, intermodulation characteristics or adjacent channel selectivity exists. That is, when an unnecessary signal other than the desired signal exists in the system band, the level of the unnecessary signal is a maximum value that should satisfy the required bit error rate defined by the system.

  In general, in order to satisfy the standard value defined by the system even in the worst radio wave environment, the wireless device is always operated under the worst condition. Therefore, the wireless device is operated with the performance satisfying the worst condition other than the worst radio wave environment. In order to meet the standard even in the worst radio wave environment mentioned here, it is necessary to maintain the linearity of the radio reception system. In other words, it is necessary to reduce the distortion of the receiving system so as to satisfy the standard. As a result, this is related to the operating currents of the circuit blocks constituting the receiving system, mainly the low noise amplifier and the frequency converter.

  In general, in order to improve the linearity of a circuit, it is necessary to increase the operating current of the circuit. For this reason, a radio that always assumes the worst radio wave environment consumes more power than necessary. This is because the radio is not always in the worst radio wave environment, and most of the time when the radio is operating is outside the worst radio wave environment. Next, a second problem of the reception system in which a direct current offset (hereinafter referred to as DC offset) occurs during reception and the reception characteristics are deteriorated will be described.

  In general, in an active circuit such as the above-described frequency converter, low-frequency filter, low-frequency amplifier, etc., a DC component is generated by superimposing a desired signal on its output. This DC component is also generated by self-mixing.

  Here, the phenomenon of self-mixing will be described with reference to FIG.

  As shown in FIG. 36 (a), the output of the local oscillator 107 leaks and is reflected by the frequency converter 104, and the reflection component 401 is multiplied again with the local oscillator 107 by the frequency converter 105. Since the reflection component 401 and the signal oscillated from the local oscillator 107 have the same frequency, a DC component is generated at the output of the frequency converter 105. FIG. 37 shows the output (desired signal) 301 of the frequency converter 105 on which the DC output 305 is superimposed. A line denoted by reference numeral 304 indicates a thermal noise level.

  Since this DC offset output causes a significant deterioration in the reception error rate, a conventional AC receiver capacitor (reference numeral in FIG. 35) normally removes the DC output for the purpose of deleting this DC output. 119-124) are inserted into the baseband stage.

  In FIG. 38A, the DC output 305 can be removed by inserting a capacitor so that the AC coupled frequency characteristic becomes 302. However, a part 303 of the desired signal component in the desired signal 301 is deleted. That is, the notch 306 shown in FIG. This notch naturally occurs in the thermal noise 304 (307). Therefore, the C / N characteristics may be improved in some cases for an FSK signal having a high modulation index with a small desired signal component in the vicinity of DC.

  A method of removing DC offset by effectively using this AC couple has been proposed, and this method can be effectively used particularly in binary FSK having a high modulation index, which has been used in a conventional pager or the like. This is because the signal component near the DC is small, so that the attenuation of the signal component due to the AC couple is small.

  However, in FSK and quaternary FSK having a low modulation index used for high-speed data transmission in recent years, there are many signal components near DC, which is problematic in practical use.

  As described above, the DC offset generated in the receiving unit has been regarded as a problem in the heterodyne system. However, in the case of the direct conversion system currently used in the mobile communication field or the like, it becomes a particular problem. The problem of the DC offset in this direct conversion method has elements different from those of the heterodyne method, and will be described in detail below.

  A conventional direct conversion type radio (hereinafter referred to as a direct conversion radio) provides an external radio input signal (RF signal) and a local signal of the same frequency to a mixer, thereby providing a radio frequency signal. This is a radio device in a signal conversion form for directly converting to a baseband signal. The configuration of this direct conversion radio is shown in FIG.

  As shown in FIG. 39, the direct conversion radio device differs from the heterodyne system in that a local oscillator 130 that oscillates at the same frequency as the center frequency of the RF signal, frequency converters 2131 and 2132 as mixers, and a local oscillator And a phase shifter 2133 that shifts the phase of the local signal by π / 2 (shifts the phase), whereby the RF signal can be directly converted into a baseband signal.

  Next, the operation of this direct conversion radio will be described.

  As shown in FIG. 39, at the time of reception, the high frequency signal received by the antenna 101 is guided to the high frequency amplifier 102 via the transmission / reception changeover switch 170. The amplified signal is added to the frequency converters 2131 and 2132 together with the local signals from the local oscillator 130 and the phase shifter 2133. The frequency converters 2131 and 2132 are basically three-port devices and have three terminals of RF, LO, and baseband signals (terminal names are not shown). A high frequency signal is applied to RF and a local signal is applied to LO. As a result, a frequency-converted signal is generated at the baseband output terminal.

  By the way, a mathematically ideal mixer has infinite isolation between the respective terminals, and a signal applied to a specific terminal does not appear at other terminals.

  However, since the actual mixer, that is, the frequency converters 2131 and 2132 cannot obtain infinite isolation, the local signal generated by the local oscillator 130 is a path indicated by a broken line 402a as shown in FIG. To the antenna 101. Therefore, the direct conversion radio device radiates a local signal from the antenna 101 at the time of reception.

  The local signal radiated from the antenna 101 is reflected by an external reflector, which is also received by the antenna 101, and is input to the frequency converters 2131 and 2132 again by following the path of the broken line 402b in the figure. become. Since the signal re-input to each frequency converter 2131, 2132 and the local signal are signals of exactly the same frequency, the multiplication process which is the mixing function of the frequency converters 2131, 2132 performs the multiplication process of each frequency converter 2131, 2132. A DC component output is generated at the baseband output terminal. That is, a desired high-frequency signal is subjected to frequency conversion, and at the same time, a direct current component, that is, a DC offset is generated by multiplication of the own local signals.

  In addition, when reflection at the amplifier 102 and reflection at the filter 103 occur, a DC offset occurs along the paths indicated by the broken lines 403 and 404 in FIG. 36B, as in the case of the heterodyne method.

  The DC offset generated in the path indicated by the broken line 402a in FIG. 36B varies depending on the amount of reflection of the local signal from the outside, that is, the reflecting object around the antenna 101. Therefore, the DC offset occurs in the paths indicated by the broken lines 403 and 404. The problem is larger than the DC offset or the DC offset inherent in the active device.

  Among direct conversion radios, especially mobile phones that are subject to miniaturization are carried by people and carried in bags or pockets, so the status of reflectors outside the antenna 101 is constantly changing. And change. Therefore, the amount of reflection of the local signal is also time-varying, and the DC offset also changes from time to time, which cannot be suppressed and causes a decrease in reception sensitivity.

  In order to compensate for this DC offset, a capacitor is provided on the subsequent circuit. However, since the capacitance of the capacitor is fixed, the reception error rate deteriorates due to the transient response of the DC offset that is time-varying.

  As described above, the conventional receiving system is not able to cope with the two problems of low current consumption and improved reception characteristics at present, and the direct conversion method is more problematic than the heterodyne method.

  Next, a conventional radio synthesizer will be described.

  A conventional synthesizer uses a frequency synthesizer, which is a signal generator that can easily obtain a stable and accurate frequency signal by applying a control signal from the outside.

  The configuration of a conventional frequency synthesizer is shown in FIG.

  In the figure, 1101 is a highly stable reference oscillator such as a temperature compensated crystal oscillator (hereinafter referred to as TCXO), 1103 is a reference frequency divider, 1105 is a phase comparator, 1107 is a loop filter, and 1109 is a voltage controlled oscillator (hereinafter referred to as VCO). Reference numeral 1111 denotes a comparison frequency divider.

  Next, the operation of this frequency synthesizer will be described.

  In the case of this frequency synthesizer, the output of the highly stable reference oscillator 1101 is divided by the reference frequency divider 1103, and the divided signal is used as a reference signal. The phase of the reference signal and the signal obtained by dividing the output of the VCO 1109 by the comparison frequency divider 1111 are phase-compared by the phase comparator 1105, and a voltage corresponding to the phase difference between the two input signals is output. The loop filter 1107 smoothes the output voltage of the phase comparator 1105 and generates a control voltage for the VCO 1109. By configuring a phase locked loop (hereinafter referred to as PLL) in this way, an output signal with a stable frequency can be obtained from the VCO 1109.

  Now, when the frequency switching is performed by changing the frequency dividing number of the comparison frequency divider 1111 from N1 to N2, a voltage corresponding to the phase difference between the two signals is output from the phase comparator 1105, and the loop filter 1107 is output. Frequency synchronization and phase synchronization are performed via the. The time until synchronization at this time is determined by the natural angular frequency ωn of the loop determined by the loop filter 1107 and the damping coefficient ζ. Generally, when a natural angular frequency and a damping coefficient with a stable oscillation frequency and little noise are selected, the frequency switching time becomes long.

  By the way, since this type of frequency synthesizer requires low phase noise characteristics, it takes time to switch frequencies. For this reason, when the conventional frequency synthesizer is applied to a TDMA type radio device, there is a serious drawback in that an empty channel search cannot be performed using a gap between empty slots during a call.

  Next, a conventional radio transmission system will be described.

  In FIG. 35, the frequency converter 154, the variable attenuator 152, the power amplifier 151, the transmission power control circuit 171, the transmission / reception changeover switch 170 and the like can be easily made into an IC, and these parts can be miniaturized by the IC technology. I have been.

  However, the band-pass filters 150 and 153, the directional coupler 172, and the power detector 173 are difficult to be integrated into an IC, and thus it is necessary to mount components on the motherboard alone. For this reason, the mounting area increased. For example, the directional coupler is a chip component of about 5 mm × 5 mm. The power detector DET is a diode switch as shown in FIG. 41. However, when the mounting area of the diode D1, the capacitors C1 and C2, and the resistor RES is considered, the power detector DET is also 5 mm × 5 mm or more. For this reason, there existed a problem that the volume of the portable radio | wireless machine in which size reduction is essential will become large. Further, it is necessary to increase the output power of the power amplifier 151 so as to compensate for the loss of output power due to the directional coupler 172, and there is a disadvantage that the power consumption of the transmission system increases.

  Next, a conventional radio antenna will be described.

  Among the components that make up radios, miniaturization of radio circuits has been progressing from the viewpoint of improving portability, and in particular, the miniaturization of batteries and antennas has progressed rapidly. Absent. Therefore, when considering the whole radio, the antenna is one of the things that must be further reduced in size and thickness.

  On the other hand, in a portable radio device, the influence of the human body on the antenna is a problem. The human body absorbs or scatters radio frequency radio waves. Furthermore, the human body changes the operating impedance of the adjacent antenna. This is because the human body works as a radio wave absorber having a high dielectric constant in terms of high frequency. As a result, the radiation characteristics of the antenna are deteriorated by the human body.

  In addition, the ears tend to be closer to the antenna due to the recent miniaturization and thinning of radios. This proximity makes the deterioration of antenna characteristics due to the human body more serious.

  Here, the measurement results of the antenna characteristics of the conventional PHS terminal are shown in FIGS. The experimental data shown here is the result of experimenting with a model of a portable radio (PHS terminal) having a frequency of about 2 GHz.

  FIGS. 42 and 43 both show vertical polarization patterns in a horizontal plane in a state during a call. FIG. 43 is a diagram in the case of using an antenna and a housing that are smaller than those in FIG. Comparing the patterns of vertically polarized waves in FIG. 43 and FIG. 42, when using a small antenna and housing (FIG. 43), compared to using a normal size antenna and housing (FIG. 42). It can be seen that the gain deteriorates by about 8 dB.

  As described above, one of the causes of this deterioration is that the impedance of the antenna varies depending on the human body. Here, the state of deterioration will be described in detail. In the description, it is assumed that an antenna is used for transmission.

  In order to emit radio waves from an antenna, power must first be input to the antenna. The optimum condition for power input to the antenna is that the impedance of the feeder line and the impedance of the antenna have the same value. When the impedance of the antenna fluctuates from the optimum value, the power transmitted through the feed line is reflected at the input end of the antenna and returns to the transmission amplifier. Therefore, the power output from the amplifier is not input to the antenna. In addition, this reflection may adversely affect the oscillation of the amplifier in some cases.

  Hereinafter, what can be easily inferred as a method for solving the above-described problem and problems included in the solution will be described.

  The first thing that comes to mind as a method of suppressing reflection at the input end of an antenna is to make the antenna broadband. That is, even if the input impedance fluctuates due to the proximity of the human body, this method utilizes the fact that the fluctuation of the wideband antenna is smaller than that of the narrowband antenna. However, widening the antenna bandwidth requires increasing the volume of the antenna. Therefore, the method for widening the antenna causes a result which is contrary to the miniaturization of the antenna described in the beginning, and consequently the miniaturization of the entire radio.

  As another method, it is conceivable to adjust the impedance of the antenna so as to be optimal when it is brought close to a human body. However, this method is generally not a good method. This is because the portable wireless device is not always used when the human body is in proximity. Since a person carries a portable wireless device by hand or puts it in a bag, the usage state of the portable wireless device changes variously. The fluctuation amount of the impedance of the antenna changes depending on the use state. This is because the adjacent materials and distances differ depending on the use state. When the amount of variation varies in this way, it is very difficult to adjust the impedance of the antenna to an optimum state in all states.

  Furthermore, the ear is raised as the human body part closest to the antenna, but the size of the ear varies greatly between individuals. Antenna performance is also affected by individual differences in the size of the ear. The influence of the ear greatly fluctuates the impedance of the antenna. This is because the dielectric constant of the ear is as high as around 80, and when this is close to the antenna, the electrical length of the antenna changes greatly. The impedance of the antenna varies greatly depending on whether the ear is attached or not attached to the antenna, or whether the ear is attached or not. The relative position of the ear antenna depends on the size of the ear. In this way, even when trying to optimize the impedance when the human body is close to the human body, performance variation occurs depending on the person, such that an antenna optimally designed for one person is not optimal for another person.

  As methods other than the above two methods, there are some methods for optimally controlling the antenna matching circuit in accordance with the use state.

  One method is to change the antenna matching circuit by turning on / off the call button. This is putting a call button. In other words, it is established by the assumption that the antenna should be close to the ear because it is talking.

  However, in this case, there is an advantage that it can be realized with a simple configuration, but it cannot cope with the individual differences in the size of the ear described above.

  The second method is to examine the level of the reflected wave reflected from the antenna and change the matching circuit of the antenna according to the amount of reflection.

  However, in this case, it is necessary to insert a probe between the antenna and the radio circuit in order to check the amount of reflection. This probe causes a reflection loss or a conductor loss, resulting in a loss of high-frequency signals transmitted and received. cause.

  As described above, in the conventional radio device described above, in the reception system, the current consumption of the radio device must always flow the maximum current of use, and is consumed more than necessary. In addition, the DC offset removal by the AC couple also attenuates the desired signal component. Furthermore, there is a problem that there is a time-varying DC offset caused by a reflector outside the antenna that cannot be removed even by an AC couple, and deterioration of reception sensitivity cannot be suppressed.

  In addition, the synthesizer has a problem in that it is not possible to perform an empty channel search using an interval between empty channel slots during a call.

  Further, in the transmission system, circuit components other than the antenna, such as a directional coupler or a power detector, are large in mounting area, and further downsizing is difficult.

  In addition, when carrying a wireless device, antenna characteristics deteriorate due to the proximity of the antenna and the human body, and in order to improve this, there is a problem that the antenna must be enlarged or the user must be selected. there were.

  The present invention has been made to solve such problems, and a first object thereof is to reduce power consumption. A second object is to remove time-varying DC offset and improve reception sensitivity. Furthermore, the third object is to maintain the original performance of the antenna without increasing the size of the antenna and without selecting a user.

In order to achieve the above object, a radio device according to claim 1 is a frequency converter for converting an RF signal received by an antenna into an IF signal, and removes a signal having a predetermined frequency or more from the IF signal. A filter, an analog / digital converter that converts an IF signal from which a signal of a predetermined frequency or more has been removed to a digital signal, and / or between the frequency converter and the filter or between the filter and the analog / digital converter The capacitor includes a capacitor that removes a DC component included in the IF signal, and a compensation unit that multiplies the digital signal by coefficient data of a part of the inverse frequency characteristic of the capacitor . The radio of the invention according to claim 2 is characterized in that the coefficient data of a part of the inverse characteristic of the frequency characteristic of the capacitor is coefficient data of a frequency component not including a DC component.

According to a third aspect of the present invention, there is provided a wireless device according to the first or second aspect of the present invention, wherein the storage device stores the frequency characteristic data of the capacitor, and the reverse based on the data stored in the storage device. And generating means for generating coefficient data of a part of the characteristic, wherein the compensating means multiplies the digital signal by the coefficient data of the part of the inverse characteristic generated by the generating means.

According to a fourth aspect of the present invention, there is provided a radio apparatus according to the third aspect , wherein the radio apparatus according to the third aspect is configured to measure frequency characteristics of the capacitor to obtain data, and based on frequency characteristic data measured by the measurement means. And calculating means for calculating the inverse characteristic of the frequency characteristic of the capacitor and storing the coefficient data of a part of the inverse characteristic obtained by the calculation in the storage means.

  As described above, according to the present invention, in the radio reception system, the power in the system band and the power of the desired wave are detected, and the current consumption control is performed based on the result. It can be powered. In the case where a capacitor for AC coupling is provided in the receiving unit, since the deterioration of the frequency characteristic is corrected including the characteristic of the capacitor, the DC offset that deteriorates the reception error rate can be removed. Further, the reflected power returning from the reflector is detected by the directional coupler, and each unit that generates the DC offset is controlled based on the detected reflected power and transmission power, so that the time-varying DC offset can be removed.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

  FIG. 1 is a diagram showing a configuration of a reception system of a direct demodulation radio according to an embodiment of the present invention. Here, a direct demodulation wireless device will be described as an example of an embodiment of the present invention, but the present invention can also be applied to a heterodyne method or the like.

  In the figure, LNA is a low noise amplifier, BPF is a band pass filter, MIX is a frequency converter, BUFF 1 and 2 are buffer amplifiers, LPF 1 or BPF 1 and LPF 2 are low pass filters (or BPF 2 is a band pass filter), RSSI 1 and The RSSI 2 is a power detector, 10 is a subtractor (or a divider), 11 is a decision device (decision), 12 is a delay device (deley), and 13 is a current control means (current-contol). LPF1 or BPF1 is a filter having a characteristic of allowing only a signal band to pass, and RSSI1 detects power in the signal band. LPF2 or BPF2 is a filter having a characteristic of allowing the entire system band to pass, and RSSI2 detects power in the system band. Since the passbands of the LPF 2 and the BPF 2 need to cover all the frequencies in the system band, a filter that passes through the system band at least is required. For example, the delay device 12 delays the input signal by one frame or one slot from the time when the current control is determined, and outputs the delayed input signal. In order to reduce the current in the circuit block of the receiving system, it is necessary to detect that the radio is not currently present in the worst radio wave environment. In order to do this, RSSI1 and RSSI2 are provided. RSSI1 detects the power in the desired wave band, and RSSI2 detects the power in the system band. The determination device 11 determines whether or not the radio wave environment in which the radio is present is in the worst state by taking the difference or ratio between the power detected by the RSSI 1 and the power detected by the RSSI 2. Actually, there is no guarantee that the circuit will operate without any problem even if the current is reduced to, for example, about half when the radio wave environment is not worst, so the numerical value obtained as a result of taking the difference or ratio between RSSI1 and RSSI2 Are assigned to several stages, and a current is set for each of the assigned stages to flow through the low-noise amplifier LNA and the frequency converter MIX. The current set based on the difference or ratio between RSSI1 and RSSI2 is checked in advance for the distortion characteristics of the circuit block, and the obtained result is written in a table created on a memory (not shown) in the determination device 11. Shall. Then, it is assumed that a current flows to the low noise amplifier LNA and the frequency converter MIX with reference to the set current value in the table. When the number of current stages to be allocated is small, it is not always necessary to refer to the table.

  Next, the operation of this reception system will be described with reference to the flowchart of FIG.

  In describing the operation of this reception system, it is assumed that the reception slot is only once per frame, and control is performed using only that reception slot. For the sake of simplicity of explanation, only two levels of currents are assigned: normal current (for the worst radio wave environment) and low current mode.

  In the case of this reception system, as shown in FIG. 2, from the start of the reception slot (slot start) to the end of the reception slot (slot end), that is, after the slot end (S201), the RSSI 1 In addition to detecting the power, the power in the system band is detected by RSSI2, and each power detection result is subtracted (RSSI2-RSSI1) or divided (RSSI2 / RSSI1), and the subtraction result (or division result) has a set value It is determined whether or not it is greater than or equal to A (S202).

  Here, for example, when the subtraction result is equal to or greater than a certain set value A (YES in S202), the determination device determines that the radio wave environment is bad and determines that normal current is passed (normal mode) (S203). The determination result is applied to the next reception slot (S204).

  When the subtraction value (RSSI2-RSSI1) or the division value (RSSI2 / RSSI1) detected in the next reception slot is smaller than the set value Α (NO in S202), the determination device 11 is in the low current mode (low current mode). (S205), the currents of the low noise amplifier LNA and the frequency converter MIX are reduced (S206).

  In order to reduce the current, for example, a resistance value or a specified voltage in a bias circuit (not shown) for setting a bias current of the low noise amplifier LNA and the frequency converter MIX is changed. Similarly, the current is adjusted. In this embodiment, the case where the detection is performed one frame before has been described. However, the detection may be performed one slot before. In this case, the determination is made by taking the difference or ratio between the output level of the desired wave detected one frame before and the output within the current system band. In carrying out the invention, it is desirable to set the initial current setting to the normal current mode. However, if all the channels are free by the above-described free channel detection, there is clearly no unnecessary wave in the system. Since it can be estimated, the initial value may be set to the low current mode.

  Next, since the band of the LPF 2 and the BPF 2 is the band of the entire system, white noise (thermal noise) is increased by the band ratio compared to the band of the desired wave. When this white noise (thermal noise) increases, the determination device 11 may erroneously determine that there is a radio wave that exceeds a specified value in the system. In order to avoid this, it is necessary to correct the detected power value of RSSI2 by subtracting the power component of white noise corresponding to the expanded band from the detected power value obtained from RSSI2. As a correction method, the power component of white noise may be simply subtracted from the power value detected by RSSI2, and the power value detected by RSSI2 is corrected using a table in which the power of white noise is written. You may do it.

  FIG. 3 is a diagram showing another embodiment relating to the reception system of the wireless device of the present invention.

  As shown in FIG. 3, this receiving system is obtained by adding an I-channel multiplier 115, a Q-channel multiplier 116, a memory 118 as storage means, and the like to the configuration of the conventional receiving system in FIG. .

  In the memory 118, data of the reverse characteristic of the overall frequency characteristics of the AC couple of the baseband part from the frequency converter 105 to the A / D converter 113, and the frequency converter 106 to the A / D converter 114 are stored. The data of the reverse characteristics of the overall frequency characteristics of the AC couple of the baseband part up to are stored. Each inverse characteristic data is multiplied by a desired signal in each multiplier 115 and 116. The desired signal that is the output of the A / D converters 113 and 114) has a notch 306 as shown in FIG. 38B, but the multipliers 115 and 116 multiply the contents of the memory 118. Thus, the original signal is reproduced.

  Next, frequency characteristics stored in the memory 118 will be described with reference to FIGS.

  In FIG. 4A, reference numeral 801 denotes the frequency characteristics of the AC couple in the baseband part from the output of the frequency converter 105 to the output of the A / D converter 113. The frequency characteristic 801 of this AC couple is zero at direct current (DC). The inverse characteristic of the frequency characteristic 801 is a characteristic 802 shown in FIG. 4B, which is infinite at the DC frequency. This frequency characteristic (reverse characteristic) 802 cannot be stored as data in the memory 118 as it is.

  Therefore, the reverse characteristic 802 is used as the reverse characteristic by removing a predetermined level or more of the DC component according to the required compensation accuracy.

  In the example of FIG. 4B, a part 803 of the DC component of the AC couple inverse characteristic 802 is removed, and the remaining part is the inverse characteristic 804. Therefore, the desired wave after multiplying the output signals of the A / D converters 113 and 114 by the inverse characteristic 804 has a compensation error corresponding to the DC component of the inverse characteristic 804 being cut off. That is, as shown in FIG. 5, notches 701 and 702 are generated in the desired wave 301 and the thermal noise 304, respectively, due to the compensation error of the DC component censoring of the inverse characteristic 804.

  However, the unnecessary DC output is completely removed, and it is clear that the notch 701 of the signal component is improved as compared with the notch 306 shown in FIG. 38B before compensation. Since the DC offset output is basically different between the I channel and the Q channel, this self-compensation operation must be performed for both IQ channels. The advantage of this DC output self-compensation function is that even if the reference carrier frequency supplied from the local oscillator 107 is offset from a desired value in FIG.

  For example, when the frequency of the local oscillator 107 is offset from a desired value, the center frequency of the baseband desired signal output from the frequency converters 105 and 106 is offset from direct current. This is shown in FIG.

  As shown in FIG. 6, in the desired signal 601, the center frequency is offset from the DC component by the width indicated by reference numeral 602. This corresponds to a frequency offset from the desired frequency of the local oscillator 107. However, as can be seen from FIG. 6, even when the center frequency of the signal frequency-converted to baseband is offset from direct current, the frequency of the DC output 305 is not offset from direct current in principle. Therefore, the DC output 305 is completely removed by the AC couple frequency characteristics 302 of the capacitors 119 to 124. Thereafter, the inverse characteristic 804 stored in the memory 118 may be multiplied. Therefore, the DC output 305 can be removed without causing a notch in the desired signal 301 as in the case where the local oscillator 107 has no frequency offset. As the data of the inverse characteristic 804, data measured in advance may be stored in the memory 118.

  Next, another embodiment of the radio reception system according to the present invention will be described with reference to FIG. The reception system of this embodiment is obtained by adding a sweep oscillator 901, a changeover switch 902, and a calculator 907 to the reception system shown in FIG. The sweep oscillator 901 is connected to each circuit system of the I channel and the Q channel via the changeover switch 902. The sweep oscillator 901 sweeps up to a frequency range where the frequency characteristics 801 of the AC couple from the output of the frequency converters 105 and 106 to the output of the A / D converters 113 and 114 are flat. By this sweep operation, the frequency characteristics 801 of the AC couple of the I channel and the Q channel can be obtained and sent to the arithmetic unit 907 provided by branching the circuit downstream of the A / D converters 113 and 114. The computing unit 907 calculates the inverse characteristic 804 of FIG. 4 from the measured AC couple frequency characteristic 801 and stores it in the memory 118. The operation of measuring the frequency characteristics using the sweep oscillator 901 can be performed in a time interval in which the reception operation is not performed.

  As described above, according to this embodiment, even when the frequency characteristic 801 of the AC couple changes due to the temperature characteristic or the like, the inverse characteristic 804 can be obtained more flexibly and the DC offset compensation can be performed.

  Although this receiving system has been described as a heterodyne system, it can also be used in a direct modulation system. However, it cannot cope with the transient response of the time-varying DC offset caused by the reflector. However, depending on the radio system, this measure may provide sufficient characteristics.

  Next, a method for removing a time-varying DC offset caused by a reflector that is a problem in the direct conversion method will be described.

  FIG. 8 is a diagram showing a configuration of a direct conversion radio (hereinafter referred to as a direct conversion radio) which is an embodiment of the radio of the present invention.

  In the figure, 101 is an antenna, 170 is a transmission / reception selector switch, 102 is a high frequency amplifier, and 105 and 106 are frequency converters. The frequency converters 105 and 106 include a DC offset control terminal 132-1. Reference numeral 130 is a local oscillator, 131 is a π / 2 phase shifter, 109 is a baseband filter, and 111 and 112 are low-frequency amplifiers. The low frequency amplifiers 111 and 112 are provided with a DC offset control terminal 132-2. Reference numerals 135 and 136 denote baseband signal processing circuits each having a DC offset control terminal 132-3. Each baseband signal processing circuit 135, 136 is provided with analog / digital converters 113, 114 and addition / subtraction circuits 133, 134, respectively.

  Reference numeral 137 denotes a transmission system of this direct conversion radio. The transmission system 137 includes a bandpass filter 150, a directional coupler 172, a power amplifier 151, a variable attenuator 152, a power detector 173, a power control circuit 171, an adder 156, frequency converters 157 and 158, a low-pass filter. 159, 160, a digital / analog converter, a transmission signal generator 138, and the like. A DC offset control circuit 139 is connected to the directional coupler 172 and the control terminals 132-2, 132-2, and 132-3.

  Next, the operation of this direct conversion radio will be described. First, the basic operation of transmission will be described.

  In this direct conversion radio, at the time of transmission, the transmission / reception changeover switch 102 is switched to the transmission system 137 side. A transmission wave from the transmission signal generator 138 is amplified by the power amplifier 151 and transmitted from the antenna 101 via the directional coupler 172 and the transmission / reception changeover switch 170.

  Here, an explanation will be given by taking as an example a ΤDD system that uses the same frequency for transmission and reception. During transmission, reflected power from the antenna 101 is measured by the directional coupler 172. Further, the traveling power to the antenna 101 is also measured by the directional coupler 172. These measurement results are output to the DC offset control circuit 139. Then, the reflection coefficient of the antenna 101 at the time of transmission is obtained from the reflected power and the traveling power input in the DC offset control circuit 139. Since the same frequency is used for transmission and reception in the TDD scheme, it can be seen that when the reflected power during transmission is large, the reflection of the local signal from the antenna 101 during reception is large. Conversely, when the reflected power is low, the magnitude of reflection of the local signal at the time of reception is also small. For this reason, by performing the transmission operation, the reflected power at the time of transmission from the antenna 101 can be known, and thereby the amount of reflection of the local signal to the reception system can be obtained.

  When the reflected wave is large, the DC offset control circuit 139 sends a control signal having a polarity in a direction to reduce the DC offset and proportional to the reflected power to the DC offset control terminal 132-of the frequency converters 105 and 106. Output to 1. Thereby, the DC offset in the receiving system can be reduced.

  Thus, according to the direct conversion radio of this embodiment, when performing transmission / reception by the TDD method, the directional coupler 172 measures the traveling power to the antenna 101 and the reflected power from the antenna 101 during transmission, Since the reflection coefficient of the antenna 101 is obtained based on these and reflected in the reception system, the reception operation can be started in a state in which the DC offset due to the external reflector is reduced from the reception immediately after transmission. The reception sensitivity at the reception start time can be improved as compared with the case where no processing is performed. In addition, since reception can be performed from a state in which the DC offset is reduced, there is no need to provide a large number of AC coupling capacitors on the circuit.

  In the above description, the control for the frequency converters 105 and 106 has been described. Of the DC offset control terminals 132-2 and 132-3 provided in the low frequency amplifiers 111 and 112 and the analog / digital converters 113 and 114, By outputting the control signal to either one, it is possible to obtain the same effect as when the frequency converters 105 and 106 are controlled. This is because the DC offset may be compensated as necessary after the mixer, that is, after the frequency converters 105 and 106.

  Next, with reference to FIG. 9, the modification of the radio | wireless machine of embodiment of the said FIG. 8 is demonstrated. FIG. 9 is a view showing a modification of the embodiment of FIG.

  In this modification, a memory 141 is provided between the DC offset control circuit 139 and the directional coupler 172. This memory 141 generates a predetermined control voltage according to the reflected power or reflection coefficient from the antenna 101. Therefore, if a control voltage for reducing the DC offset is stored in advance according to the reflected power of the antenna 101, only the control voltage is input to the DC offset control circuit 139. The DC offset control circuit 139 Therefore, the calculation of the reflection coefficient is not performed, and the DC offset can be reduced in a shorter time than the embodiment of FIG.

  Next, another modification of the wireless device of the embodiment of FIG. 8 will be described with reference to FIG. FIG. 10 is a view showing a modification of the embodiment shown in FIG.

  This modification is an example in which the DC offset reduction processing is collectively performed by the baseband signal processing circuit 135 without providing the DC offset control circuit 139. In FIG. 10, only the baseband signal processing circuit 135 is illustrated, but the baseband signal processing circuit 136 is the same.

  In this case, the addition / subtraction circuit 133 adds the value according to the traveling power of the power amplifier 151 and the reflected power of the antenna 101 measured by the directional coupler 172 and the values output from the analog / digital converters 113 and 114. , 134 is used to reduce the DC offset. This is because subtracting or adding from the output values of the analog / digital converters 113 and 114 is equivalent to performing a DC value offset, that is, performing an analog DC offset.

  Thus, according to this modification, the same effect as described above can be obtained without connecting the special DC offset control circuit 139 to an analog circuit such as a frequency converter or a low-frequency amplifier. Note that although the TDD system has been described as an example of the communication system here, the same effect can be obtained by measuring the amount of reflection during transmission even when the same frequency band is used for transmission and reception.

  Next, a radio synthesizer will be described.

  FIG. 11 is a block diagram showing a first embodiment of the synthesizer of this radio.

  In the figure, 1101 is a reference oscillator, 1103 is a reference frequency divider, 1105 is a phase comparator, 1151 is a loop filter for normal mode, 1152 is a loop filter for high speed mode, 1153 is a switch, 1109 is a VCO, and 1111 is a comparison component. It is a peripheral. Since the basic loop operation is the same as that of a conventional synthesizer, its description is omitted.

  As shown in FIG. 11, the synthesizer includes a normal mode loop filter 1, a high speed mode loop filter 2, and a switch 1153. When the switch 1153 is switched to the normal mode loop filter 1, the loop is in the normal mode, and the output of the VCO 1109 has low phase noise characteristics, but the frequency switching time is long.

  On the other hand, when the switch 1153 is switched to the loop filter 2 for the high speed mode, the loop enters the high speed mode (empty channel search mode) and the phase noise characteristic of the output of the VCO 1109 deteriorates, but the frequency switching time becomes fast.

  Next, the operation of the synthesizer used in the wireless device according to the embodiment of the present invention will be described. FIG. 12 is a conceptual diagram showing a slot configuration of the TDMA system.

  In FIG. 12, 1160 is a frame and 1161 is a slot. Here, it is assumed that a call is performed using the reception slot R1 and the transmission slot Τ1. The frequency at which a call is made is assumed to be F1.

  In the case of this synthesizer, an empty channel search is performed as follows.

  First, during the period of the reception slot R1, it operates in a normal mode with good phase noise characteristics. Simultaneously with the end of the period of the reception slot R1, the loop is switched to the high speed mode by the switch 1153.

  Then, the idle channel search is performed by setting the synthesizer to a frequency F2 different from that of the call channel. At this time, since the loop is in the high-speed mode, the desired frequency is switched at a high speed, and a free channel search is quickly performed. When the empty channel search is completed at this frequency F2, the idle channel search is performed again after switching to another frequency F3. After such an operation is repeated and an empty channel search is performed at a plurality of frequencies, the frequency returns to the frequency F1 of the original speech channel before the transmission slot Τ1 arrives. At the same time, the mode is switched from the high-speed mode to the normal mode, and during the period of the transmission slot Τ1, the operation is again performed in the normal mode with good phase noise characteristics. By repeating the above operation, a free channel search can be performed during a call. When searching for an empty channel, since the loop is in the high-speed mode, the S / N of the synthesizer, that is, the phase noise characteristic is not good and the reception sensitivity is lowered, but the degree of whether there is an empty channel is determined. Therefore, the phase noise characteristics of the synthesizer necessary for this purpose are alleviated as compared with those during a call, so there is no problem even if an empty channel search is performed in the high speed mode. In the above description, the loop characteristic is switched by switching the frequency characteristic of the loop filter. However, the loop characteristic may be switched by switching the sensitivity of the phase comparator, for example. Further, all the empty channel searches may be performed within one slot, for example.

  Also, if all channels are free due to a high-speed free channel search, it can be determined that there is no interference wave near the system band, so this free channel search can be used as an operation to detect the radio wave environment. good. That is, as a result of the empty channel search, for example, if all the channels are free, it is possible to control to reduce the current consumption of the receiving system.

  In this case, the result of the empty channel search is input to the determination device 11 shown in FIG. 1, and the current mode is determined in the determination device 11 to control the current of the LNA and MIX.

  Next, the transmission system will be described with reference to FIGS.

  FIG. 13 is a basic conceptual diagram in the case where the power amplifier or the transmission / reception changeover switch is integrated into an IC, and FIG.

  In FIG. 13, reference numeral 1200 denotes an IC. When the IC 1200 is, for example, a power amplifier IC (hereinafter referred to as PΑ-IC), RF signals from the frequency converters 105 and 106 are input to the input terminal IN of the IC 1200. An RF signal amplified by the PA-IC 1200 is output to the output terminal OUT. This IC 1200 is provided with sensing means 1201 for sensing the output power, and a signal proportional to the sensed power from the sensing means 1201 is output from a terminal different from the output terminal OUT. It is output to a power detector DET or the like which is a signal processing means.

  In the same figure, when the IC 1200 is, for example, a transmission / reception changeover switch (hereinafter referred to as a T / R switch), the RF signal output from the power amplifier P is input to the input terminal IN of the IC 1200. An RF signal is output to the OUT terminal via the T / R switch. In the sensing means 1201, a signal proportional to the RF output power is sensed as in the PA-IC, and the sense output is output from a terminal different from the output terminal OUT.

  With reference to FIG. 14, another embodiment in which the embodiment of FIG. 13 is expanded will be described. In FIG. 14, sensing means 1201 is the same as that shown in FIG. 13, but in this case, the sensed signal is input to an output power detector DET which is a signal processing means formed in the same IC. The output power detector DET1202 converts output power into a low frequency by inputting a sensing signal, and outputs a signal proportional to the output power to a power control circuit (CONT) 171. Here, the reason why the output power detector DET1202 is described as the signal processing means is that the output power detector DET also performs signal processing such as frequency conversion.

  In the description of FIGS. 13 and 14, the PA-IC or the T / R switch IC does not assume a single IC, but may be an IC provided with at least one of them.

  FIG. 15 is a diagram showing a circuit that more specifically embodies the basic conceptual diagram related to the present proposal shown in FIG. 13, and shows an example using a PA-IC.

  The RF signal output from the frequency converters 105 and 106 is input to the input terminal IN and input to the power amplifier PΑ151. The output of the power amplifier PA151 is output to the band pass filter BPF at the next stage via the output terminal OUT.

  A power supply terminal VDD1 in the IC 1200 is a terminal that supplies power to the power amplifier PA151, and a ground terminal GND1 is a ground terminal of the power amplifier PA151. The reason why the external power supply terminal VDD and the external ground terminal GND are separated as the terminal VDD1 and the terminal GND1 is that when an IC having such a configuration is actually mounted, parasitic inductors, resistors, and capacitor components are generated between the circuits. It is for showing. That is, the impedance Zvdd 1203 is connected between the terminal VDD and the terminal VDD1 due to parasitics, and the impedance Zgnd 1204 is connected between the terminal GND and the terminal GND1 due to parasitics.

  In the figure, a terminal s1 is a terminal for extracting a signal proportional to the RF output power of the power amplifier PA151, and is connected to the power supply terminal VDD1 via the sensing means 1201 described in FIG. That is, in this example, a signal proportional to the multiplication of the parasitic impedance Zvdd1203 and the instantaneous current of the power amplifier P is observed at the power supply terminal VDD1. In general, the change in the instantaneous current of the power amplifier PA151 is proportional to the output power of the power amplifier PA151. Therefore, the AC component observed at the terminal VDD1 is proportional to the output power of the power amplifier PA151.

The sensing means 1201 will be described in detail. Considering that a signal proportional to the output power of the power amplifier PA151 can be obtained by obtaining an AC component observed at the terminal VDD1, For example, as shown in FIG. 16A, a capacitor C1 or the like may be provided between the terminal VDD1 and the terminal s1.
In this case, for example, the power detection circuit DET shown in the conventional example is connected to the next stage of the terminal s1. The capacitor C1 does not have directionality but can be used without any problem because a coupler having directionality is not necessarily required for the purpose of power detection. In the circuit configuration of FIG. 16A, a sensing means 1201 is configured with a simple element configuration in which only the capacitor C1 is provided, so that a signal proportional to the output power can be extracted. In addition, as the sensing means 1201, other elements such as a diode and a resistor may be used.

  By the way, depending on how the PA-IC is mounted, the values of the parasitic impedance Zvdd1203 and the parasitic impedance Zgnd1204 change, so the proportionality coefficient depends on the mounting. For this reason, the power detection value output from the power detection circuit DET may change depending on the implementation. Further, since the signal detected from the terminal VDD1 is about −50 dB of the RF signal output power and the signal level is relatively small, the low frequency detection signal output from the power detection circuit DET is used as the power control circuit (CONT). Too small to receive at 171.

  Therefore, as shown in FIG. 16B, a sensing means 1201 may be configured by connecting a variable gain high frequency amplifier (AMP) 1205 to the capacitor C1 and connecting it to the terminal s1.

  In this case, the variable gain high frequency amplifier (AMP) 1205 is connected to the next stage of the capacitor C1 in order to match the output signal of the power detection circuit DET with the dynamic range of the power control circuit (CONT) 171. The variable gain high frequency amplifier (AMP) 1205 is provided with a gain control terminal (gain adjustment) 1206, and a control signal is input to the gain control terminal (gain adjustment) terminal 1206 from the power control circuit (CONT) 171. By doing so, an output signal corresponding to the control signal is obtained, and fluctuations in the amplitude of the output power detection signal due to mounting can be suppressed. As a result, a stable power detection signal having a high signal level is input to the power control circuit (CONT) 171 so that the power control signal (CONT) 171 can sufficiently detect the power detection signal.

  In this way, in the circuit configuration of FIG. 16B, the power control circuit (CONT) 171 can sufficiently detect the power detection signal by using the variable gain high frequency amplifier (AMP) 1205 as a subordinate connection to the capacitor C1, An IC that can withstand practical use can be achieved.

  In addition, although new power consumption is produced by the variable gain high frequency amplifier (AMP) 1205 formed in the IC and the overall power consumption increases somewhat, this power consumption is the power consumed by the power amplifier (PA) 151. Since it is sufficiently small compared to, the power consumption of the transmission system is not a problem.

  Here, a specific internal configuration of the variable gain high frequency amplifier (AMP) 1205 will be described with reference to FIG.

  As shown in FIG. 17, VDD2 is a voltage source. The voltage source VDD2 has an anode connected to the base terminal of the transistor Q1 and a cathode connected to the ground terminal GND1 in the IC. The emitter terminal of the transistor Q1 is connected to the input terminal in and also connected to the ground terminal GND1 via the variable current source I1. A gain control signal from the gain control terminal (gain adjustment) 1206 is input to the variable current source I1. The collector of the transistor Q1 is connected to the power supply terminal VDD1 in the IC through a load impedance Z1. The collector of the transistor Q1 is connected to the base terminal of the buffer transistor Q2. The collector terminal of the transistor Q2 is connected to the power supply terminal VDD1. The emitter terminal of the transistor Q2 is connected to the ground terminal GND1 via the constant current source I2, and is also connected to the output terminal out via the capacitor C3 for DC block. In this circuit configuration, the gain G can be approximated by the following equation.

G = gm (Q1) × Z1
= I1, dc × Z1 / Vt (Formula 1)
In Equation 1, i1 and dc are currents flowing through the variable current source I1, and Vt is a thermal voltage.

  Therefore, the gain of the signal output from the variable gain high frequency amplifier (AMP) 1205 is adjusted by changing the current values i1 and dc of the variable current source I1 according to the gain control signal input from the gain control terminal (gain adjustment) 1206. can do.

  Next, some examples in which the T / R switch and the sensing means 1201 are manufactured as an IC based on the basic concept of FIG. 13 will be described.

  One of the most common circuits that make a T / R switch an IC is a single-pole-dual-throw (SPDT) switch. The basic circuit of this SPDTswitch IC is shown in FIG.

  In FIG. 18, Tin is a transmission system input terminal, and Rin is a reception system output terminal. The terminal ANT is an output terminal for the transmission system and an input terminal for the reception system, and is connected to the antenna 101. GND1 is a ground terminal of the IC. Complementary control signals are input to the terminals cont11 and cont2, and transmission / reception switching is performed by the control signals. When the terminal cont1 is high “Η” and the terminal cont2 is low “L”, the switch elements Q11 and Q12 are conductive, the switch elements Q10 and Q13 are open, and the signal input from the terminal ANT is the reception system output terminal Rin. Is output.

  On the other hand, when the terminal cont1 is low “L” and the terminal cont2 is high “Η”, the switch elements Q11 and Q12 are open, the switch elements Q10 and Q13 are conductive, and the signal input to the transmission system input terminal Tin is the terminal. Output to ANT.

  The SP / D switch IC is a T / R switch IC in which a sensing circuit as a sensing means for sensing a signal proportional to the transmission output power is added to the SPDT switch IC.

  As shown in FIG. 19, the T / Rswitch IC 1200 is obtained by connecting a sensing circuit 1201 between the source terminal of the switch element Q12 of FIG. 18 and the ground terminal GND1 in the IC. A specific configuration of the sensing circuit 1201 is shown in FIGS.

  As an example of the sensing circuit 1201, for example, as shown in FIG. 20, an impedance circuit Z is interposed between the source terminal of the switch element Q12 and the ground terminal GND1. The impedance circuit Z includes a resistor, a capacitor, an inductor, a series circuit or a parallel circuit thereof. In addition, out is an output terminal.

  Hereinafter, the operation of the IC 1200 will be described.

  At the time of transmission, cont1 is low “L”, cont2 is high “Η”, and the switch element Q12 is opened. However, since the RF signal is input to the transmission system input terminal Tin, the RF signal is output from the terminal out by the capacitor between the source and drain of the switch element Q12 or the series connection of the source and gate capacitor and the gate and drain capacitor. Leaks into. Since the leakage current flows through the impedance Z, a voltage proportional to the leakage current is generated. Since this leakage current is proportional to the power of the RF signal, a signal proportional to the RF power can be extracted from the terminal out by this method.

  Further, as another example of the sensing circuit 1201, as shown in FIG. 21, a variable gain high frequency amplifier (AMP) 1205 is inserted between the impedance circuit Z and the output terminal out. The variable gain high frequency amplifier (AMP) 1205 amplifies a signal generated in the impedance circuit Z, and gain adjustment is performed by inputting a gain control signal, for example, an applied voltage, from a gain control terminal (gain adjustment) 1206. .

  Next, a case where the power amplifier (PA) 151 portion is integrated into an IC, that is, a PA-IC will be described with reference to FIG. FIG. 22 shows a PA-IC in which a power amplifier (PA) 151, sensing means 1201 and detection means such as the output power detector DET1202 shown in FIG. 14 are formed on a one-chip IC. FIG. In the PA-IC 1200, a signal proportional to the power to be detected is output from the terminal d1 and input to the power control circuit (CONT) 171. Thus, by using the one-chip PA-IC 1200 including the sensing means 1201 and the output power detector DET 1202, the operation of sensing and detecting power can be performed inside the IC, so that the transmission system can be downsized. be able to.

  Next, a case where the T / R switch is integrated with the IC, that is, the T / Rswitch IC will be described with reference to FIG. FIG. 23 is a diagram showing a T / Rswitch IC in which a T / R switch circuit, a sensing means 1201, and an output power detector DET1202 are formed on a one-chip IC.

  In this case as well, a signal proportional to the output power is output from the terminal d1 as described above. The specific configuration of the sensing means 1201 is the same as that shown in FIGS. In addition, although not shown here, such an IC is not limited to the IC including the power amplifier PΑ or the T / R switch alone, and may be an IC including one of them.

  Thus, by using a one-chip T / R switch IC in which the sensing means 1201 and the output power detector DET 1202 are formed, an operation for sensing and detecting power can be performed inside the IC. It can be downsized.

  Furthermore, the power sensing means and the detection means shown in FIGS. 15, 19, 22, 23, etc. detect the power leaking to the power supply or the ground, for example, without being directly connected to the RF signal line. This eliminates the power loss taken by the transmitter and the like, thereby reducing the power consumption of the transmission system.

  FIG. 24 is a view showing one embodiment in which the power sensing and detection elements shown in FIG. 16 are formed on an IC chip. FIG. 24 (a) is a plan view, and FIG. 24 (b) is a view. It is AA sectional drawing of 24 (a).

  In FIG. 24, reference numeral 1401 denotes a power supply wiring using a metal layer of the second metal layer supplied to the power amplifier (PA) 151, which is formed on the IC surface layer. Reference numeral 1402 denotes a metal wiring of the first layer metal, which is formed in the inner layer of the IC. The power supply wiring 1401 and the metal wiring 1402 constitute a power sensing element 1402 (sensing means). 1202 is an output power detector DET (detect means), and 1404 is an insulating layer.

  When power sensing and detection means are provided in the IC as described above, in a normal IC, a metal layer is formed on the power wiring of the surface layer via an insulating layer to form a capacitance component in terms of low loss. There are many cases.

  However, in the case of this embodiment, the first-layer metal wiring 1402 formed in the inner layer under the power supply wiring 1401 on the surface layer is adjusted and capacitively coupled.

  In general, the power supply wiring 1401 is formed in a wide area in order to reduce the impedance of the power supply. Therefore, the area of the power detection capacitor formed by the first and second layers can be increased, and the leakage power can be easily picked up. This characteristic is advantageous for power detection. Note that the present invention can be applied without impairing the effect even when the metal layer assignment is reversed as in the prior art. For example, as shown in FIG. 25, the capacitance to be coupled is adjusted by making the line width of the metal wiring 1402 of the first layer metal formed in the inner layer narrower than that of the power supply wiring 1401. As shown in FIG. 26, the line width of the metal wiring 1402 of the first layer metal may be adjusted wider than that of the power supply wiring 1401. Further, as shown in FIG. 27, the line length of the metal wiring 1402 of the first layer metal may be adjusted. Further, as shown in FIG. 28, adjustment may be performed by changing the formation direction of the metal wiring 1402 of the first layer metal.

  Finally, another embodiment of the radio of the present invention, for example, an antenna portion of a portable radio will be described with reference to FIGS. FIG. 29 is a diagram showing a configuration of a portable radio apparatus according to an embodiment of the present invention.

  In FIG. 29, reference numeral 1500 denotes a housing, and 1501 denotes a radio circuit, which includes, for example, the frequency converters 157 and 158 and the variable attenuator 152 shown in FIG. Reference numeral 1509 is a transmission amplifier, 101 is an antenna, 1504 is an ammeter, 1503 is a control circuit, 1502 is a matching circuit, 1505 is a power supply circuit, and 1506 is a probe for current measurement.

  The power supply circuit 1505 supplies power to the wireless circuit 1501, the transmission amplifier 1509, and the control circuit 1503. The radio circuit 1501 generates an information signal mixed with modulation and transmission frequency based on the supplied power, and sends the information signal to the transmission amplifier 1509. The transmission amplifier 1509 amplifies the transmitted signal and transmits it to the antenna 101. The antenna 101 sends the amplified signal into the air, but part of it is sent back to the transmission amplifier 1509 as a reflected wave, and the gain and efficiency of the transmission amplifier 1509 vary. This variation causes a variation in current consumption. This variation is measured by an ammeter 1504 and the current level is sent to the control circuit 1503. There are cases where the current increases or decreases depending on how the transmission amplifier 1509 fluctuates. Here, one that changes monotonically is used. The control circuit 1501 reads the value of the signal sent from the ammeter 1504 and electrically adjusts the variable part on the matching circuit 1502. As the variable portion on the matching circuit 1502, a variable capacitor such as a semiconductor switch or a semiconductor varicap may be used.

  Hereinafter, it has been confirmed through experiments that the characteristics of the antenna 101 can be prevented from being deteriorated due to the state of use of the portable wireless device, which will be described in order.

  30 is a diagram showing a radio model created based on the circuit configuration of the portable radio shown in FIG. 29, and FIGS. 31 to 33 are graphs showing the results of measurement using the radio model shown in FIG.

  As shown in FIG. 30, the wireless device model includes a speaker 1511, a microphone 1512, an antenna cover 1514, and the like on the surface of the housing 1500, and a wireless circuit 1501 including a transmission amplifier 1509 in the housing 1500. A coiled antenna (helical antenna) 101 is provided in the antenna cover 1514.

  The power supply to the wireless circuit 1501 of this wireless device model was performed by connecting a power supply line to a casing 1500 from a constant voltage source 1510 installed outside. The current consumed by this wireless device model was measured by reading the fluctuation of an ammeter 1513 provided in the constant voltage source 1510. The matching circuit 1502 is simply simulated by directly changing the parameters of the antenna 101. The operating frequency of the wireless device model is around 2 GHz, and the size of the casing 1500 is about one wavelength in length, about one quarter wavelength in width, and about one-twentieth wavelength in thickness. The antenna 101 has a height of about one full wavelength.

  FIG. 31 shows the relationship between the use state of the wireless device model and current consumption. FIG. 32 shows the reflection coefficient at the input end of the antenna 101 for each use state. FIG. 33 is a diagram showing the relationship between the use state and the average power in the horizontal plane radiated from the radio device model. From FIG. 31 and FIG. 32, it can be seen that the reflection coefficient of the antenna 101 changes depending on the use state, and the current consumption of the transmission amplifier 1509 increases depending on the use state. Further, from FIG. 33, it can be seen that the radiated power from the antenna 101 decreases as the unit is held by hand and further becomes in a talking state. These phenomena can be explained by considering the antenna 101 as a load of the transmission amplifier 1509. That is, it is considered that the operating state of the power amplifier PΑ and the like has changed due to the fluctuation of the load, and as a result, the current consumption has increased. Moreover, the fluctuation of the load at this time is clearly caused by the human body.

  Subsequently, the antenna parameters were optimized while reading the current values. The current consumption is increasing from a single unit to a handheld unit as the telephone conversation state is further increased. Therefore, in order to bring it closer to the original single state, it was considered that the parameters should be reset so that the current consumption decreases. The antenna parameter used for adjustment was the antenna length. This is because the resonance frequency of the antenna 101 can be changed by changing the antenna length. When the antenna length was extended or shortened, it was found that the current consumption was reduced by reducing the antenna length. When the antenna radiated power was measured in this state, it was found that the radiated power also increased by about 2 dΒ as compared to before the antenna length was adjusted. Therefore, it has been experimentally confirmed that the characteristic degradation of the antenna 101 is reduced by the proposed method.

  In this experiment, the antenna length was adjusted. As an equivalent method, the characteristic of the matching circuit 1502 can be variably controlled. Its specific configuration is shown in FIG.

  In FIG. 34, 1521 is an antenna element shorter than a quarter wavelength, 1522 is a variable capacitor which is a part of the matching circuit, and 1523 is a path for preventing the control power supply 1526 from flowing directly into the high frequency source 1527. 1524, an inductance for preventing high frequency from flowing into the control circuit 1503, 1525, a variable resistor for controlling the voltage applied to the variable capacitor 1521, and 1528, a resistor.

  By changing the value of the variable resistor 1525, the value of the variable capacitor changes. If the value of the variable capacitance increases, the antenna 101 appears to be equivalently extended, thereby reducing the resonance frequency. If the value of the variable capacitance decreases, the antenna 101 appears to contract and the resonance frequency increases. In this way, the matching condition can be changed by changing the resonance frequency of the antenna 101.

  As described above, according to the portable wireless device of this embodiment, it is possible to improve the characteristics of the antenna when close to the human body.

  According to each of the embodiments described above, it is possible to reduce the current consumption and increase the efficiency of the portable wireless device. In addition, time-invariant and time-varying DC offsets that increase the error rate of the receiver can be eliminated. Further, the portable radio can be reduced in size. Further, it is possible to speed up the empty channel search.

The block diagram which detects the power in a system band and the power of a desired wave for the receiver part power consumption reduction by this invention. The figure which shows the control flow for the receiver part power consumption reduction by this proposal. The figure which shows the example of 1 structure of the receiver provided with the self-compensation function by this invention. The figure which shows the reverse characteristic of AC coupling used for the self-compensation of this invention. The figure which shows the desired wave after the self-compensation by this invention. The figure which shows DC offset output when a local oscillator frequency detunes. The figure which shows the structure of other embodiment of the receiver provided with the self-compensation function. The figure which shows the structure of the direct conversion radio | wireless machine of one Embodiment of this invention. The figure which shows the modification of the radio | wireless machine of embodiment of FIG. The figure which shows the other modification of the radio | wireless machine of embodiment of FIG. The figure which shows the structure of the synthesizer of the radio | wireless machine of one Embodiment of this invention. The figure explaining the high-speed empty channel search operation | movement of this synthesizer. The basic conceptual diagram at the time of providing a sensing means in front end high frequency IC. The basic conceptual diagram at the time of providing a sensing means and a power detector in front end high frequency IC. The figure which shows the structure of PA-IC of one Embodiment of this invention. (A) is a figure which shows an example of the sensing means (sensing means) of PA-IC of FIG. (B) is a figure which shows the other example of a sensing means (sensing means). The figure which shows an example of the variable gain control circuit shown in FIG.16 (b). The figure which shows the structure of a general T / R switch. The figure which shows the structure of T / RswitchIC of one Embodiment of this invention. The figure which shows an example of the sensing means. The figure which shows the other example of the sensing means (sensing means). The figure which shows the structure of PA-IC of one Embodiment of this invention. The figure which shows the structure of T / RswitchIC of one Embodiment of this invention. (A) is a top view which shows one embodiment at the time of forming the element for electric power sensing and detection shown in FIG. 16 on the IC chip. (B) is AA 'sectional drawing of Fig.24 (a). The figure which shows the example which made the line | wire width of the metal layer narrower than the power supply line, and adjusted the coupling capacity. The figure which shows the example which made the line | wire width of the metal layer wider than the power supply line, and adjusted the coupling capacity. The figure which shows the example which adjusted the coupling capacity | capacitance by changing the line length of a metal layer. The figure which shows the example which changed the formation direction of the metal layer and adjusted the coupling capacity. The figure which shows the structure of the portable radio | wireless machine of other embodiment of this invention. The external view of the radio | wireless machine model used for experiment. The figure which shows the relationship between the consumption current of the portable radio | wireless machine of FIG. 29, and an operation state. The figure which shows the relationship between the reflection coefficient of the antenna input end at the time of seeing from the feeder line side, and the operation state of a portable radio | wireless machine. The figure which shows the relationship between the horizontal plane average radiation gain of this portable radio | wireless machine, and an operation state. The figure which shows the structure of the matching circuit of an antenna. The figure which shows the structure of the conventional radio | wireless machine (heterodyne system). (A) is a figure which shows DC offset produced by reflection of LO signal output from the local oscillator. (B) is a figure which shows DC offset which arises other than (a). It is a figure which shows DC offset output. It is a figure which shows the influence on the desired signal by AC coupling. The figure which shows the structure of the conventional radio | wireless machine (direct conversion system). The block diagram which shows the synthesizer of the conventional radio | wireless machine. The figure which shows the conventional electric power detection circuit. The figure which shows the vertical polarized-wave radiation pattern of the PHS terminal at the time of a human body head mounting | wearing. FIG. 43 is a diagram showing a vertically polarized radiation pattern in the case of a PHS terminal that is smaller than the PHS terminal in FIG. 42.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 101 ... Antenna, 102 ... High frequency amplifier, 103 ... High frequency filter, 104, 105, 106, 157, 158 ... Frequency converter, 108, 131, 2133 ... Pi / 2 phase shifter, 109, 110, 159, 160 ... Low Frequency filter 111, 112 ... Low frequency amplifier, 113, 114 ... A / D converter, 115, 116, 2131, 2132 ... Multiplier, 117 ... Detector, 118, 141 ... Storage device, 119, 120, 121, 122, 123, 124 ... capacitor, 107, 125, 130 ... local oscillator, 126 ... frequency conversion unit, 132-1, 132-2, 132-3 ... DC offset control signal input terminal, 133, 134 ... adder or subtraction 137 ... transmission block, 138 ... transmission signal generator, 139 ... DC offset control circuit, 151 ... electricity Amplifiers 152... Variable attenuators 156. Modulator adders 161 and 162... Digital / analog converters 170. Transmission / reception changeover switches 171... Transmission power control circuit (CONT) 172. Desired signal, 302 ... AC couple frequency characteristics, 303 ... Signal deleted portion, 304 ... thermal noise, 305 ... DC output, 306,307 ... notch, 401, 402, 403, 404 ... reflection component, 601 ... desired signal, 701, 702 ... Notch, 801 ... Frequency characteristics of AC couple, 802 ... Reverse characteristic of AC couple, 803 ... Part of DC component of reverse characteristic, 804 ... Reverse characteristic, 901 ... Sweep oscillator, 902 ... Changeover switch, 903 ... I channel test Signal input, 904... Q channel test signal input, 905... I channel frequency characteristics, 906. Frequency characteristics, 907 ... arithmetic unit, 1101 ... reference oscillator, 1103 ... reference frequency divider, 1105 ... phase comparator, 1109 ... voltage controlled oscillator, 1111, ... comparison frequency divider, 1107, 1151 ... normal loop filter, 1152 ... High-speed loop filter, 1153 ... changeover switch, 1160 ... 1 frame used in the TDMA system, 1161 ... 1 slot used in the TDMA system, 1200 ... PA-IC (or transmission / reception changeover switch IC), 1201 ... sensing means (sensing) means) 1202... power detector DET (detect means), 1203... impedance Zvdd parasitic on power supply line, 1204... impedance Zgnd parasitic on ground line, 1205... variable gain high frequency amplifier (AMP), 1401. Surface layer) power line, 1402... First layer (Inner layer) metal layer, 1404 ... insulating layer, 1500 ... housing, 1501 ... radio circuit, 1504 ... ammeter, 1503 ... control circuit, 1502 ... matching circuit, 1505 ... power supply circuit, 1506 ... probe for current measurement, 1509 ... Transmission amplifier, 1520 ... Antenna element, 1521 ... Variable capacitance element, 1523 ... Capacitance element, 1524 ... Inductance element, 1525 ... Variable resistance element, 1526 ... DC power supply, 1527 ... High frequency wave source, 1510 ... Constant voltage source, 1511 ... Speaker , 1512 ... microphone, 1513 ... ammeter, 1514 ... antenna cover, PA ... power amplifier, T / R ... transmission / reception switch, AMP ... variable gain high frequency amplifier, Z ... impedance circuit, MIX ... frequency converter, LO ... local signal Generator, BPF ... band pass filter, CPL ... coupler, ... Signal detection circuit (DET), ANT ... antenna, D1 ... diode, RES ... resistor, Cn (n = integer) ... capacitor, Qn (n = integer) ... transistor, Zn (n = integer) ... impedance circuit, In ( n = integer) ... current source.

Claims (4)

  A frequency converter for converting an RF signal received by an antenna into an IF signal;
  A filter that removes a signal having a predetermined frequency or higher from the IF signal;
  An analog / digital converter that converts the IF signal from which the signal of the predetermined frequency or more has been removed into a digital signal;
  A capacitor that is inserted between at least one of the frequency converter and the filter or between the filter and the analog / digital converter and removes a DC component included in the IF signal;
  Compensation means for multiplying the digital signal by coefficient data of a part of the inverse characteristic of the frequency characteristic of the capacitor;
A wireless device characterized by comprising: The radio apparatus according to claim 1, wherein the coefficient data of a part of the inverse characteristic of the frequency characteristic of the capacitor is coefficient data of a frequency component not including a DC component.   Storage means for storing frequency characteristic data of the capacitor;
  Generating means for generating coefficient data of a part of the inverse characteristic based on the data stored in the storage means;
  The wireless device according to claim 1, wherein the compensation unit multiplies the digital signal by coefficient data of a part of the inverse characteristic generated by the generation unit.   Measuring means for measuring the frequency characteristics of the capacitor to obtain the data;
  An operation for calculating the inverse characteristic of the frequency characteristic of the capacitor based on the data of the frequency characteristic measured by the measuring unit and storing the coefficient data of a part of the inverse characteristic obtained by the calculation in the storage unit Means and
The wireless device according to claim 3, further comprising:

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