JP4024718B2 - Reception time measuring device and distance measuring device using the same - Google Patents

Description

【００４３】
ここで、ａは正の実数である（ただし、図４ではａ＝１としている）。
ｆ１（ｔ）においてｔ∈［−Ｔ／２，Ｔ／２］が信号レベル変化部分である。
ｔ＝±Ｔ／２の点はそれぞれ異なる２つの関数で表される波形で構成されている。即ち、信号レベル変化部分は異なる関数で表現される複数の波形の接続で構成されている。そして、接続点ｔ＝±Ｔ／２の点に１階微分で現れる不連続点が、図示のようにサブバンド分解フィルタ１３Ｂの出力Ｄ１〜Ｄ３の変化として検出される。
【００４４】
図５は、Ｍ＝２の場合、即ち、波形の２階微分で不連続点が現れるような信号レベル変化部分の波形例と、その波形に対する特異部分検出部１３のサブバンド分解フィルタ１３ＢのＤ１〜Ｄ３の出力例を示す。
【００４５】
波形ｆ２（ｔ）は次の数２の式で表せる。
【００４６】
【数２】

【００４７】
ｐは２以上の自然数である（ただし、図ではｐ＝３）
波形ｆ２（ｔ）でｔ∈［−Ｔ／２，Ｔ／２］が信号レベル変化部分である。
フィルタ等を使用せずに占有周波数帯幅を抑制するため、復調信号の信号レベル変化部分はそれ以外の部分と滑らかに連続していることが望ましい。具体的には、信号レベル変化部分の始点と終点で、復調信号の波形の１階微分の値が連続であることが望ましい。更に、信号レベル変化部分の波形を１階微分して得られる波形もその信号レベルが連続していることがより望ましい。波形ｆ２（ｔ）はそのような特性を実現でき、ｐ＝２以上では、ｆ２（ｔ）の１階微分の値は、信号レベル変化部分の始点と終点で零であり、始点と終点の間で連続する。そして、ｆ２（ｔ）の２階微分で、異なる２つの関数で表現される波形の接続点ｔ＝０に不連続が現れ、図示のようにサブバンド分解フィルタ１３Ｂの出力Ｄ１〜Ｄ３の変化として検出される。尚、ｐ＝２では２階微分によりｔ＝０の他にｔ＝±Ｔ／２にも不連続が生じ、ｐ＝３以上とすれば、図示のようにｔ＝±Ｔ／２では連続となり、ｔ＝０のみに不連続が生じる。
【００４８】
図６は、Ｍ＝３の場合、即ち、波形の３階微分で不連続点が現れるような信号レベル変化部分の波形例と、その波形に対する特異部分検出部１３のサブバンド分解フィルタ１３ＢのＤ１〜Ｄ３の出力例を示す。
【００４９】
波形ｆ３（ｔ）は次の数３の式で表せる。
【００５０】
【数３】

【００５１】
波形ｆ３（ｔ）は、図５の波形ｆ２（ｔ）と同様に、信号レベル変化部分の始点と終点で、復調信号の波形の１階微分の値が連続であり、信号レベル変化部分の波形を１階微分して得られる波形もその信号レベルが連続する特性を満たしている。更に、波形ｆ３（ｔ）は２階微分についても１階微分で得られる波形と同様の特性を有する。波形ｆ３（ｔ）でｔ∈［−Ｔ／２，Ｔ／２］が信号レベル変化部分である。そして、図示のように、接続点ｔ＝±Ｔ／２の点に３階微分で現れる不連続点が、サブバンド分解フィルタ１３Ｂの出力Ｄ１〜Ｄ３の変化として検出される。
【００５２】
送信装置１において、変調部３は例えば振幅変調や角度変調等の種々の変調方式が利用可能であり、上述したような特異部分（復調信号の波形のＭ階微分で現れる不連続点）が時間軸上で局所的に存在し且つ信号レベルが連続している復調信号を生じさせる送信信号を送信できればよい。受信装置１０の復調部１２は、送信装置１からの送信信号を復調できる構成とすることは言うまでもない。
【００５３】
図７に、図１に示す変調部３と復調部１２の構成例を示す。
図７の（Ａ）は変調部３を示し、（Ｂ）は復調部１２を示す。
図７（Ａ）において、変調信号ｆＭ（ｔ）は、変調信号生成部２で生成され、受信装置１０の復調部１２で本来生成されるべき復調信号と同じ信号であり、信号レベル変化部分が例えば上述した関数ｆ１（ｔ）、ｆ２（ｔ）又はｆ３（ｔ）等で構成される信号である。
【００５４】
図７（Ａ）、（Ｂ）の変調部３と復調部１２による変復調動作を説明する。
変調部３では、変調信号ｆＭ（ｔ）が入力すると、Ｖ／Ｆ変換器で中心周波数ｆｏを変調信号ｆＭ（ｔ）で周波数変調したベースバンド帯域の信号ｆｃｏ（ｔ）に変換し、この周波数変調信号ｆｃｏ（ｔ）を信号源ＲＦの変調用基準信号（周波数ｆｒ）により周波数変換器で無線帯域に周波数変換し、フィルタで不要な周波数帯域成分を除去して送信信号ｆＳＤ（ｔ）を生成し送信する。この送信信号ｆＳＤ（ｔ）が受信されて入力する図（Ｂ）の復調部１２では、送信信号ｆＳＤ（ｔ）を周波数変換器で信号源ＬＯの復調用基準信号（周波数ｆｒ）により周波数変換し、フィルタで不要な周波数帯域成分を除去してベースバンド帯域の周波数変換信号ｆＤＮ（ｔ）としてＦ／Ｖ変換器に入力し、Ｆ／Ｖ変換器で信号ｆＤＮ（ｔ）の周波数に比例した電圧レベルの周波数検波信号ｆＤＭ（ｔ）を生成する。この周波数検波信号ｆＤＭ（ｔ）は、例えば周波数ｆｏを電圧出力の基準として次式で表せる。
【００５５】
ｆＤＭ（ｔ）＝Ｄ・ｆＭ（ｔ） （Ｄは定数）
この式から復調信号としてｆＭ（ｔ）を得ることができる。従って、送信装置１は、特異部分（復調信号の波形のＭ階微分で現れる不連続点）が時間軸上で局所的に存在し且つ信号レベルが連続している復調信号を生じさせる送信信号を送信できる。
【００５６】
図８は、変調部３と復調部１２の別の構成例を示す。
図８の（Ａ）は変調部３を示し、（Ｂ）は復調部１２を示す。
図８は一般的な直交変復調回路であり、変調部３の動作は、変調信号ｆＭＩ（ｔ）をＩｓ入力とし、変調信号ｆＭＱ（ｔ）をＱｓ入力として、直交変調器で、信号源ＲＦからの変調用基準信号（周波数ｆｒ）をＩｓ入力とＱｓ入力で変調して信号ｆＱＭ（ｔ）を出力し、出力信号ｆＱＭ（ｔ）の不要周波数帯成分をフィルタで除去して送信信号ｆＳＤ（ｔ）を送信する。ここで、ｆＭＩ（ｔ）とｆＭＱ（ｔ）は、ｆＭＩ（ｔ）²＋ｆＭＱ（ｔ）²＝１が成立するように定め、次式とする。
【００５７】
ｆＭＩ（ｔ）＝ｆＭ（ｔ）
ｆＭＱ（ｔ）＝sin［cos^-1（ｆＭ（ｔ））］
ただし、（｜ｆＭ（ｔ）｜≦１）である。
【００５８】
ここで、ｆＭ（ｔ）は図７の変調信号と同様で、本来生成されるべき復調信号と同じ信号で、信号レベル変化部分が例えば上述した関数ｆ１（ｔ）、ｆ２（ｔ）又はｆ３（ｔ）等で構成される信号である。
【００５９】
また、フィルタから出力される送信信号ｆＳＤ（ｔ）がｆＱＭ（ｔ）と殆ど同等になるようにフィルタとｆＭ（ｔ）を定める。これにより、ｆＳＤ（ｔ）＝ｆＱＭ（ｔ）となる。
【００６０】
送信信号ｆＳＤ（ｔ）が受信されて入力する復調部１２では、直交復調器で、信号源ＬＯからの復調用基準信号（周波数ｆｒ）により受信信号を復調して、互いに直交関係にある直交復調信号ｆＩＲ（ｔ）とｆＱＲ（ｔ）を出力し、フィルタで不要な高周波成分を除去して信号ｆＤＩ（ｔ）とｆＤＱ（ｔ）を出力する。ここで、信号ｆＤＩ（ｔ）、ｆＤＱ（ｔ）は次式で表せる。
【００６１】
ｆＤＩ（ｔ）＝Ｅ′・cos［cos^-1（ｆＭ（ｔ））＋φ′］
ｆＤＱ（ｔ）＝Ｅ′・sin［cos^-1（ｆＭ（ｔ））＋φ′］
ここでＥ′は定数である。上式で、位相差φ′＝０を保持できるとき、
ｆＤＩ（ｔ）＝Ｅ′・ｆＭ（ｔ）
ｆＤＱ（ｔ）＝Ｅ′・（１−ｆＭ（ｔ）²）^1/2
となり、ｆＤＩ（ｔ）としてｆＭ（ｔ）を得ることができ、図８の直交変復調方式を採用した場合は、位相差φ′＝０を保持できる構成とすれば、特異部分検出用の復調信号としてｆＤＩ（ｔ）を用いることができる。
【００６２】
以下に、第１実施形態の計測動作について説明する。
送信装置１では、変調信号生成部２において、変調部３が図７（Ａ）の場合には変調信号ｆＭ（ｔ）を生成し、図８（Ａ）の場合にはｆＭＩ（ｔ）及びｆＭＱ（ｔ）を生成して変調部３に入力し、前述したように変調部３から送信信号ｆＳＤ（ｔ）を送信する。ここで、前記変調信号ｆＭ（ｔ）或いはｆＭＩ（ｔ）は、信号レベル変化部分が例えば前述した図４〜図６で示すような関数ｆ１（ｔ）、ｆ２（ｔ）又はｆ３（ｔ）等で構成される信号であり、送信信号ｆＳＤ（ｔ）は周波数帯幅が電波法等の占有周波数帯幅の規定を満たすものである。
【００６３】
受信装置１０では、前記送信信号ｆＳＤ（ｔ）を受信して受信時刻計測装置１１に入力する。受信時刻計測装置１１では、受信信号が復調部１２に入力し、前述したように復調部１２が図７（Ｂ）の場合には復調信号ｆＤＭ（ｔ）（＝ｆＭ（ｔ））を出力し、図８（Ｂ）の場合には復調信号としてｆＤＩ（ｔ）（＝ｆＭ（ｔ）；ただし、位相差φ′＝０が保持されているものとする）を出力する。復調信号ｆＤＭ（ｔ）或いはｆＤＩ（ｔ）は、図２に示す構成の特異部分検出部１３に入力し、出力Ｄ１を特異部分検出信号として出力する。ここで、復調信号の信号レベル変化部分が、例えば、１階微分で不連続点が現れる関数ｆ１（ｔ）で構成されている場合は特異部分検出信号として図４の出力Ｄ１が出力され、２階微分で不連続点が現れる関数ｆ２（ｔ）で構成されている場合は特異部分検出信号として図５の出力Ｄ１が出力され、３階微分で不連続点が現れる関数ｆ３（ｔ）で構成されている場合は特異部分検出信号として図６の出力Ｄ１が出力される。
【００６４】
特異部分検出部１３からの特異部分検出信号Ｄ１は、時刻計測部１４のレベル比較器１４Ａに入力して予め設定した閾値Ｖｔｈと比較され、特異部分検出信号Ｄ１の信号レベルが閾値Ｖｔｈ以上になれば、レベル比較器１４Ａから論理値１の信号が受信時刻算定部１４Ｃに入力する。受信時刻算定部１４Ｃには、計時部１４Ｂから時刻情報が逐次入力しており、受信時刻算定部１４Ｃは、レベル比較器１４Ａから論理値１の信号が入力したときの時刻を受信信号の受信時刻として通報する。尚、図５のように特異部分検出信号Ｄ１が負の値の場合には、図３の点線で示す絶対値化回路１４Ｄを設けてレベル比較器１４Ａに入力すればよい。
【００６５】
かかる構成によれば、信号レベルが連続している信号を用いて無線信号の占有周波数帯幅を拡大することなく、受信信号の受信時刻を高精度に検出することが可能となる。従って、有限の周波数資源を有効利用でき、また、情報伝送速度を低下させずに済み、実用的効果大である。
【００６６】
次に、本発明の第２実施形態について説明する。
無線通信においては、受信信号へのノイズ混入を配慮する必要があり、受信信号にノイズが混入した場合、復調信号に含まれる特異部分の検出を妨害したり、誤ってノイズを特異部分として検出する虞れがあり、時刻計測精度の悪化を招く。
【００６７】
第２実施形態は、ノイズによる誤計測を防止するよう構成したものであり、例えば、図１の特異部分検出部１３と時刻計測部１４との間に、誤計測防止部として図９の構成の相関演算部２０を設ける構成である。
【００６８】
図９において、相関演算部２０は、入力信号と相関用基準信号との相関演算を行い両者の類似度が高いときほど高レベルの相関信号を出力するものであり、ｎ個の遅延要素Ｄからなる遅延回路２１と、各遅延要素Ｄの出力Ｄｓ１〜Ｄｓｎと相関用基準信号から定める相関演算用の係数１〜係数ｎとをそれぞれ乗算するｎ個の乗算器２２−１〜２２−ｎと、ｎ個の乗算器２２−１〜２２−ｎの出力を加算して相関信号を出力する加算回路２３とで構成される。
【００６９】
次に、図９の相関演算部２０の動作を図１０を参照して説明する。
例えば復調信号が図１０（ａ）のように１階微分で不連続点が生じる波形（図４参照）で構成される信号レベル変化部分を含むもので、図中のＴｘを単位として同じ信号レベル変化部分の繰り返しで構成されるものとする。この場合、特異部分検出部１３からは図１０（ｂ）のような特異部分検出信号Ｄ１が出力される。特異部分検出信号Ｄ１は遅延回路２１を各遅延要素Ｄで遅延されながら伝搬し、伝搬中における各遅延要素Ｄの出力Ｄｓ１〜Ｄｓｎに係数１〜係数ｎを乗算器２２−１〜２２−ｎで掛け合わせる（尚、同図でＤｓ１〜Ｄｓｎは実際よりも粗い間隔として示してある）。その演算結果を加算回路２３で加算して相関信号として出力する。係数１〜係数ｎは、遅延回路２１の出力Ｄｓ１〜Ｄｓｎの出力パターンが本来生成されるべき特異部分検出信号の発生パターンであるときに加算回路２３から高レベルの相関信号が発生するよう入力信号と相関用基準信号の相関演算を実現するよう相関用基準信号から定める。ここで、相関用基準信号は、本来生成されるべき特異部分検出信号を類似度が高い信号と見なすように定める。これにより、特異部分検出信号Ｄ１が遅延回路２１に入力し、遅延回路２１の出力パターンが本来生成されるべき特異部分検出信号の発生パターンであれば、加算回路２３から高レベルの相関信号が発生し、その後は本来の発生パターンが継続する限り図１０（ｃ）に示すようにＴｘの周期で加算回路２３から時刻計測部１４における閾値Ｖｔｈ以上の高レベルの相関信号が発生する。ノイズの混入により特異検出信号Ｄ１の発生パターンが本来のパターンと異なると、加算回路２３からの相関信号レベルは閾値Ｖｔｈより低くなる。これにより、ノイズが存在するとレベル比較器１４Ａから論理値１の出力が発生せず、受信時刻を通報することはなく、ノイズによる誤計測を防止できる。
【００７０】
また、ノイズにより誤計測防止のため、図９の相関演算部２０に代えて、例えば復調部１２と特異部分検出部１３との間に誤計測防止部として図１１の構成のノイズ抑制部３０を設ける構成としてもよい。
【００７１】
図１１のノイズ抑制部３０は、加算回路３１と遅延回路３２とで構成され、入力信号を加算回路３１と遅延回路３２に入力する。遅延回路３２は、入力信号を所定時間遅延して加算回路３１に伝達する。加算回路３１は、入力信号と遅延回路３２の遅延出力を加算しその加算信号を特異部分検出部１３に入力する復調信号として出力する。ここで、ノイズ抑制部３０に入力する復調信号は、所定の信号波形（以下、復調信号ブロックとする）を単位とし、この復調信号ブロックの繰り返しで構成される信号とし、復調信号ブロックには信号レベル変化部分が１つ以上含まれるものとする。
【００７２】
かかる構成では、遅延回路３２の遅延時間を復調信号ブロックの繰り返し周期と同一に設定すれば、加算回路３１は、ノイズが存在しなければ時間軸上で連続する２つの復調信号ブロックを加算することになり、ノイズが存在すれば本来生成されるべき復調信号ブロック波形と異なる波形の加算信号となる。
【００７３】
尚、図１１のノイズ抑制部３０は、特異部分検出部１３と時刻計測部１４との間に設けてもよい。また、図９の相関演算部２０と図１１のノイズ抑制部３０を組み合わせてもよい。例えば、復調部１２と特異部分検出部１３との間にノイズ抑制部３０を設けると共に特異部分検出部１３と時刻計測部１４との間に相関演算部２０を設ける構成や特異部分検出部１３と時刻計測部１４との間に特異部分検出部１３側から順次ノイズ抑制部３０と相関演算部２０を設ける構成等が考えられる。
【００７４】
次に、本発明の第３実施形態について説明する。
復調部１２に図８の直交復調回路を用いる場合に、位相差φ′＝０の保持を必要とするが、このためには一般的に複雑な位相保持のための構成が必要となり、装置が複雑化する。
【００７５】
図１２に示す本発明の第３実施形態は、複雑な位相保持構成を不要とするものである。
図１２において、本実施形態の受信時刻計測装置４１の復調部４２は、図８（Ｂ）に示す構成である。また、特異部分検出部４３は、復調信号Ｉに含まれる特異部分を検出して特異部分検出信号Ｉを出力する第１検出部４３Ａ及び復調信号Ｑに含まれる特異部分を検出して特異部分検出信号Ｑを出力する第２検出部４３Ｂを備える。第１検出部４３Ａ及び第２検出部４３Ｂは、図２と同様の構成である。時刻計測部４４は、図３の構成に、相関演算部４５及び自乗和演算部４６を付加し、自乗和演算部４６の自乗和信号をレベル比較器１４Ａに入力する構成である。ここで、復調信号Ｉ及び特異部分検出信号Ｉが第１復調信号及び第１特異部分検出信号に相当し、復調信号Ｑ及び特異部分検出信号Ｑが第２復調信号及び第２特異部分検出信号に相当する。
【００７６】
相関演算部４５は、特異部分検出信号Ｉと相関用基準信号との相関演算を実行して相関信号Ｉを出力する第１演算部４５Ａ及び特異部分検出信号Ｑと相関用基準信号との相関演算を実行して相関信号Ｑを出力する第２演算部４５Ｂを備える。第１演算部４５Ａ及び第２演算部４５Ｂは、図９と同様の構成である。ここで、第１演算部４５Ａと第２演算部４５Ｂの相関用基準信号は同一の信号を用い、例えば、位相差φ′＝０のときに本来生成されるべき特異部分検出信号Ｉを用いる。相関信号Ｉが第１相関信号に相当し相関信号Ｑが第２相関信号に相当する。
【００７７】
自乗和演算部４６は、図１３に示すように、相関信号Ｉの自乗演算を行う乗算器４６Ａと、相関信号Ｑの自乗演算を行う乗算器４６Ｂと、両乗算器４６Ａ，４６Ｂの出力を加算して自乗和信号を出力する加算回路４６Ｃとを備えて構成される。
【００７８】
尚、ノイズを考慮しなければ相関演算部４５は不要である。
かかる構成では、復調信号Ｉ，Ｑとして
Ｉ＝ｆＤＩ（ｔ）＝Ｅ′・cos［cos^-1（ｆＭ（ｔ））＋φ′］
Ｑ＝ｆＤＱ（ｔ）＝Ｅ′・sin［cos^-1（ｆＭ（ｔ））＋φ′］
が出力される。
【００７９】
それぞれの復調信号Ｉ，Ｑには特異部分の情報が含まれており、特異部分検出部４３の各第１及び第２検出部４３Ａ，４３Ｂから例えば出力Ｄ１が前述のようにして特異部分検出信号Ｉ，Ｑとしてそれぞれ出力され、相関演算部４５の各第１及び第２演算部４５Ａ，４５Ｂから前述のようにしてそれぞれの加算信号が相関信号Ｉ，Ｑとして出力される。各相関信号Ｉ，Ｑは自乗和演算部４６の各乗算器４６Ａ，４６Ｂで自乗演算され、加算回路４６Ｃで加算された自乗和信号がレベル比較器１４Ａで閾値Ｖｔｈと比較され、受信時刻算定部１４Ｃで受信時刻が計測される。
【００８０】
かかる構成によれば、自乗和演算部４６で自乗和演算することにより、位相差φ′の影響を排除できる。また、相関演算部４５を設けることによりノイズの影響を抑制できる。従って、高精度に受信時刻を計測できる。
【００８１】
上述の各実施形態において、復調信号としてＰＮ符号を用いることができる。ＰＮ符号は、データ値の変化点を複数個持つので、この変化点の中の一部又は全てを前述した信号レベル変化部分の構成方法を用いて信号レベル変化部分として構成した復調信号とすればよい。
【００８２】
次に、復調信号をＰＮ符号で構成した場合の本発明の第４実施形態について説明する。
図１４は、本実施形態の受信時刻計測装置の構成図である。
【００８３】
図１４において、本実施形態の受信時刻計測装置５１は、復調部５２と、特異部分検出部５３と、第１時刻計測部５４と、逆拡散処理部５５と、第２時刻計測部５６と、計測値判定部５７とを備えて構成される。尚、前記復調部５２及び特異部分検出部５３は第１実施形態と同様の構成であり、第１時刻計測部５４は、図３のレベル比較器の前段に図９の相関演算部２０を設けた構成である。
【００８４】
前記逆拡散処理部５５は、ＰＮ符号で構成した復調信号をＡ／Ｄ変換器で離散化し、この離散化復調信号をディジタル整合フィルタで逆拡散処理して図１５（ａ）に示すような波形の逆拡散信号を出力する。前記第２時刻計測部５６は、図３の構成を有し、入力する前記逆拡散信号が所定の閾値以上になった時刻を計測して基準受信時刻として計測値判定部５７に入力する。計測値判定部５７は、第２時刻計測部５６から入力する基準受信時刻に基づいて第１時刻計測部５４から入力する受信時刻の正常／異常を判定して判定信号を出力する。尚、逆拡散処理部５５と第２時刻計測部５６による時刻計測方法は、前述したスペクトラム拡散通信を用いた計測方式であり、「スペクトラム拡散通信とその応用」（丸林他、電子情報通信学会）に記載されている。
【００８５】
次に、第４実施形態の動作を説明する。
復調部５２からの復調信号は、逆拡散処理部５５と特異部分検出部５３にそれぞれ入力する。特異部分検出部５３からはその出力Ｄ１が特異部分検出信号として発生し第１時刻計測部５４に入力する。第１時刻計測部５４では、相関演算部２０により特異部分検出信号と基準用相関信号との相関演算を実行して図１５（ｂ）に示す波形の相関信号が生成される。尚、ＰＮ符号で構成した復調信号の場合には、ＰＮ符号の周期で相関信号が高レベルとなる。この相関信号レベルと閾値をレベル比較器１４Ａで比較し、相関信号レベルが所定の閾値以上となったときの時刻を受信時刻として計測値判定部５７に出力する。計測値判定部５７は、前述のようにして逆拡散処理部５５、第２時刻計測部５６を経て入力する基準受信時刻に基づいて第１時刻計測部５４の受信時刻が正常／異常かを判定する。
【００８６】
図１５の逆拡散信号と相関信号は同じ復調信号から生成されるので、逆拡散信号と相関信号の高レベル発生時刻の間には一定の関係が存在する。即ち、第２時刻計測部５６からの基準受信時刻と受信時刻との間には一定の関係が存在する。また、スペクトル拡散通信は耐ノイズ性の高い通信方式であり、第２時刻計測部５６から得られる基準受信時刻は、第１時刻計測部５４から得られる受信時刻に比べて精度は劣るが耐ノイズ性に優れる。そこで、基準受信時刻に基づいて受信時刻が存在すべき時間軸上での範囲（以下、受信時刻存在範囲とする）を計測精度等を勘案して予め定める。例えば、基準受信時刻を中心としてＰＮ符号の前後１チップ幅を受信時刻存在範囲として定める。そして、計測値判定部５７では、第１時刻計測部５４の受信時刻が、前記受信時刻存在範囲内であるとき受信時刻は正しい値と判定して正常を通報し、前記受信時刻存在範囲外であるとき受信時刻は正しくない値と判定して異常を通報する。
【００８７】
かかる構成によれば、受信時刻計測の精度が基準受信時刻の計測精度より悪化することはなく、また、ノイズ増加による受信時刻の計測精度の悪化を検出できるので、受信時刻計測装置の信頼性が向上する。
【００８８】
尚、図１４に点線で示すように、第２時刻計測部５６の基準受信時刻も計測受信時刻情報として出力するよう構成してもよい。
復調部５２が図８の直交復調回路である場合は、基準受信時刻は位相差φ′に影響される。従って、復調信号Ｉ，Ｑをそれぞれ逆拡散処理する逆拡散処理部と、各逆拡散処理部からそれぞれ出力される各逆拡散信号を自乗和演算する自乗和演算部とを設け、自乗和演算部からの自乗和信号を第２時刻計測部５６に入力して基準受信時刻を計測すれば、基準受信時刻に対する位相差φ′の影響を排除できる。前記各逆拡散処理部は図１４の逆拡散処理部と同じ構成でよく、自乗和演算部は図１３の構成でよい。この場合、特異部分検出部５３及び第１時刻計測部５４は、図１２に示す第３実施形態の構成とする。
【００８９】
次に、図１４の第４実施形態における信号処理を軽減する場合の本発明の第５実施形態について説明する。
前述したように、第１時刻計測部５４から得られる受信時刻と第２時刻計測部５６から得られる基準受信時刻の間には一定の関係があり、基準受信時刻から受信時刻存在範囲を定めることができる。受信時刻は相関信号と時間軸上で１対１の関係にあり、相関信号は時間軸上で相関範囲の特異部分検出信号と１対１の関係にあり、特異部分検出信号は時間軸上で所定範囲の復調信号と１対１の関係にある。図１６は、受信時刻を求めるための特異部分検出処理と相関演算について、上述の関係に基づいてそれぞれ使用される信号データの範囲を模式的に示したものであり、図に示すように、受信時刻存在範囲に対応する復調信号範囲を定めることができる。受信時刻の計測処理を、図１６における復調信号範囲内の復調信号についてのみ行うようにすることで、受信時刻の計測処理を軽減できる。
【００９０】
図１７は、受信時刻計測処理を所定の復調信号範囲のみ行うようにした本発明の第５実施形態の要部構成図である。
図１７において、本実施形態の受信時刻計測装置は、図１４の特異部分検出部５３の前段に信号記憶部６０を設ける構成である。
【００９１】
信号記憶部６０は、復調信号を離散化復調信号に変換するＡ／Ｄ変換器６１と、離散化復調信号と時刻情報とを対応付けして記憶すると共に入力する基準受信時刻に基づいてその後の信号処理に使用するための復調信号範囲を定める記憶装置６２とを備える。
【００９２】
本実施形態の動作を説明する。
復調信号が信号記憶部６０に入力すると、Ａ／Ｄ変換器６１で離散化復調信号に変換され、記憶装置６２に順次入力する。記憶装置６２は、離散化復調信号が入力する毎に時刻情報と対応付けて記憶する。第２時刻計測部５６から基準受信時刻が入力すると、基準受信時刻に基づいて復調信号範囲を定め、記憶した離散化復調信号の中から前記復調信号範囲に該当するものを、入力した順番通りに読出復調信号として特異部分検出部５３に順次出力すると共に、対応付けて記憶した時刻情報を読出時刻情報として第１時刻計測部５４に順次出力する。特異部分検出部５３では、入力した離散化復調信号に基づく特異部分検出信号を第１時刻計測部５４に出力する。第１時刻計測部５４は、特異部分検出信号に基づく相関信号と読出時刻情報とから受信時刻を計測する。
【００９３】
かかる構成によれば、復調信号の所定の範囲だけ信号処理すればよいので、受信時刻計測のための信号処理を軽減できる。
尚、図１７の構成の場合、特異部分検出部５３の図２におけるＡ／Ｄ変換器は不要である。また、信号記憶部６０を特異部分検出部５３の後段に設けて特異部分検出信号範囲を定めて記憶するようにしてもよく、この場合は、特異部分検出部５３が図２の構成である場合には、信号記憶部６０のＡ／Ｄ変換器は不要である。
【００９４】
次に、上述した本発明の受信時刻計測装置を適用した本発明に係る距離計測装置の一実施形態の構成を図１８に示す。
図１８において、本実施形態の距離計測装置７０は、送信装置８０と受信装置９０から構成される。
【００９５】
前記送信装置８０は、変調信号生成部８１、変調部８２及び送信装置側計時部としての計時部８３を備える。変調信号生成部８１及び変調部８２は前述した受信時刻計測装置の場合と同様の構成である。計時部８３は、受信装置９０側の計時部９２と十分な精度で同期しているものである。
【００９６】
受信装置９０は、前述した本発明の受信時刻計測装置９１、受信装置側計時部としての計時部９２及び距離算出部９３を備える。距離算出部９３は、受信時刻計測装置９１からの受信時刻情報と計時部９２からの時刻情報とから送信装置８０と受信装置９０との間の距離を算出する。
【００９７】
次に、本実施形態の距離計測装置の距離計測動作について説明する。
送信装置８０は、計時部８３の時刻情報に基づいて予め定めた送信時刻毎に変調信号生成部８１で変調信号を生成して送信信号を送信する。送信信号を受信した受信装置９０では、受信時刻計測装置９１で受信時刻を計測し計測結果を距離算出部９３に通報する。距離算出部９３は、送信側の計時部８３と互いに同期する計時部９２の時刻情報に基づいて予め定めた送信時刻情報を得る。これにより、距離算出部９３は、（距離）＝（光速）×（（受信時刻）−（送信時刻））の演算式により距離を算出する。
【００９８】
尚、距離算出部９３において受信時刻と送信時刻を対応付けることができるよう、送信時刻は（計測予定の最大距離）／（光速）で算出される時間よりも十分間隔をあけて設定することが望ましい。
【００９９】
【発明の効果】
以上説明したように本発明の受信時刻計測装置によれば、特異部分が時間軸上で局所的に存在し且つ信号レベルが連続する復調信号の特異部分を検出して受信時刻を計測する構成としたので、信号レベルが連続する復調信号を利用して受信時刻を高精度に計測することが可能である。従って、無線信号の占有周波数帯幅を拡大せずに済み、有限な周波数資源を有効利用できると共に、無線通信における情報伝送速度を犠牲にしなくて済むようになる。
【０１００】
また、本発明の距離計測装置によれば、本発明の受信時刻計測装置を用いて送信装置と受信装置間の距離を計測するので、高精度に送信装置と受信装置間の距離を計測できる。
【図面の簡単な説明】
【図１】本発明に係る受信時刻計測装置の第１実施形態を示す構成図
【図２】特異部分検出部の構成図
【図３】時刻計測部の構成図
【図４】１階微分で不連続点が現れる信号レベル変化部分の波形例
【図５】２階微分で不連続点が現れる信号レベル変化部分の波形例
【図６】３階微分で不連続点が現れる信号レベル変化部分の波形例
【図７】変調部と復調部の構成例を示し、（Ａ）は変調部、（Ｂ）は復調部
【図８】変調部と復調部の別の構成例を示し、（Ａ）は変調部、（Ｂ）は復調部
【図９】本発明に係る受信時刻計測装置の第２実施形態における相関演算部の構成図
【図１０】相関演算部の動作説明図
【図１１】ノイズ抑制部の構成図
【図１２】本発明に係る受信時刻計測装置の第３実施形態の要部構成図
【図１３】自乗和演算部の構成図
【図１４】本発明に係る受信時刻計測装置の第４実施形態を示す構成図
【図１５】逆拡散信号と相関信号の時間軸上における関係を示す図
【図１６】受信時刻計測のために使用される信号データ範囲の関係を模式的に示した図
【図１７】本発明に係る受信時刻計測装置の第５実施形態を示す構成図
【図１８】本発明に係る距離計測装置の一実施形態を示す構成図
【図１９】無線通信における従来の送信信号生成例の説明図で、（Ａ）は信号レベルの不連続なデータ信号、（Ｂ）はローパスフィルタで高周波成分をカットした信号
【符号の説明】
１，８０ 送信装置
１０，９０ 受信装置
１１、９１ 受信時刻計測装置
１２、４２、５２ 復調部
１３、４３、５３ 特異部分検出部
１４、４４、５４ 時刻計測部
２０、４５ 相関演算部
３０ ノイズ抑制部
４６ 自乗和演算部
５５ 逆拡散処理部
５６ 第２時刻計測部
５７ 計測値判定部
６０ 信号記憶部
７０ 距離計測装置
９２ 計時部
９３ 距離算出部[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a reception time measurement device that measures the reception time of a transmission signal, and more particularly to a reception time measurement device that can measure the reception time with high accuracy without increasing the occupied frequency bandwidth. The present invention also relates to a distance measuring device using this reception time measuring device.
[0002]
[Prior art]
In a distance measuring device that calculates the distance by measuring the propagation time of a radio wave from a transmitting device to a receiving device, it is necessary to measure the reception time of the radio wave in the receiving device.
[0003]
Conventionally, there is a measurement method using spread spectrum communication as a reception time measurement method of radio waves (for example, see Non-Patent Document 1). Below, the case where the direct spreading | diffusion system by a PN code is used is demonstrated easily.
[0004]
The transmission apparatus spreads the data signal with a PN code to generate a baseband spread signal, modulates the spread signal, and transmits the modulated RF signal as a radio frequency (RF) band radio signal. The receiving device demodulates the received radio signal into a demodulated signal in the baseband, and demodulates the demodulated signal with a matched filter using the same PN code as that on the transmitting side. The matched filter output shows the maximum value when the demodulated signal is in phase with the PN code of the matched filter, and becomes almost zero when the phase of one chip or more of the PN code is shifted. For example, the time of occurrence of the maximum value is measured. The reception time.
[0005]
As another measurement method, there is a method using wavelet transform (see, for example, Patent Document 1). In this method, the timing information of the received signal is extracted using wavelet transform. Specifically, the time position of the signal level discontinuity point of the demodulated signal is detected. In the receiving apparatus, a demodulated signal having a signal level discontinuity is input to an orthogonal wavelet transformer that performs wavelet transformation. In general, since signal level discontinuities include high frequencies, the high-frequency signals of interest are filtered and extracted by the orthogonal wavelet transformer, and the time positions of the discontinuities are detected from the output of the orthogonal wavelet transformer. .
[0006]
[Non-Patent Document 1]
Marubayashi et al., "Spread Spectrum Communication and its Applications", IEICE
[Patent Document 1]
JP-A-8-70330
[0007]
[Problems to be solved by the invention]
By the way, the accuracy of the distance measurement depends on the accuracy of the reception time measurement, and in order to improve the measurement accuracy by the former method, the chip rate (frequency) of the PN code may be increased. However, since the occupied frequency bandwidth of the radio signal to be transmitted is normally limited by the Radio Law or the like, there is an upper limit on the chip rate that can be used, and it is difficult to improve the measurement accuracy beyond this upper limit.
[0008]
In the latter method, a level discontinuity point of the demodulated signal is targeted for detection, and a highly accurate reception time measurement can be realized. The condition is that the demodulated signal has a level discontinuity point. However, as described above, the level discontinuity point includes a high frequency. Therefore, in the case of wireless communication, the demodulated signal, and thus the occupied frequency bandwidth of the wireless signal, is increased, and there is a problem in terms of the efficiency of utilization of frequency resources. is there. If the occupied frequency bandwidth is to be kept within the limit, the transmission rate (information transmission speed) of the binary data signal must be reduced on the transmission side. For this reason, in wireless communication, normally, a discontinuous data signal having a signal level as shown in FIG. 19A is cut off by a low-pass filter on the transmission side as shown in FIG. 19B. By making a smooth signal free from level discontinuity and modulating and transmitting a carrier wave as a modulation signal, the occupied frequency bandwidth is suppressed and the information transmission rate is not lowered. In this case, since the level discontinuity does not exist in the demodulated signal (the same signal as FIG. 19B) generated by the demodulation processing of the receiving apparatus, the latter measurement method cannot be adopted. If the time can be detected with high accuracy using a signal with continuous signal levels, it is not necessary to increase the occupied frequency bandwidth of the radio signal, and it is not necessary to sacrifice the information transmission speed. Usefulness is high. However, there has not been proposed a device that can detect the time with high accuracy using a signal having continuous signal levels.
[0009]
The present invention has been made paying attention to the above problems, and an object of the present invention is to provide a reception time measuring apparatus capable of measuring a reception time with high accuracy using a signal having a continuous signal level. It is another object of the present invention to provide a distance measuring device using this reception time measuring device.
[0010]
[Means for Solving the Problems]
  For this reason, the invention of claim 1 is a reception time measuring device for measuring a reception time of a transmission signal from a transmission device,Input transmission signalDemodulateRecoveryA demodulator that outputs a modulated signal and a demodulated signal that is output from the demodulatorSpecialA singular part detection unit that detects a different part and outputs a singular part detection signal, and based on the singular part detection signal output by the singular part detection unitSendA first time measuring unit for measuring a signal reception time;The transmission signal is a signal that generates a demodulated signal in which the singular part is locally present on the time axis and the signal level is continuous, and the singular part is a discontinuity that appears as an M-th derivative of the waveform of the demodulated signal. It was set as the point.
[0011]
  In such a configuration, a signal that causes a demodulated signal having a singular part locally on the time axis and having a continuous signal level is transmitted from the transmitter. The demodulatorInput to sendThe signal is demodulated, and a demodulated signal in which the singular part is locally present on the time axis and the signal level is continuous is output. The singular part detector is a singular part included in the demodulated signal.In other words, discontinuous points appearing in the Mth derivative of the waveform of the demodulated signalAnd a singular part detection signal is output. The first time measurement unit measures and reports the reception time of the received signal based on the input singular part detection signal.
[0012]
  The demodulated signal is as in claim 2.It is preferable to have a signal level change portion configured by connecting a plurality of waveforms represented by different continuous functions, and the discontinuity point is a signal generated at a connection point of the waveform in the signal level change portion. Further claims3As described above, it is advantageous that the waveform obtained by the first-order differentiation of the waveform of the demodulated signal is continuous at least at the start point and the end point of the signal level changing portion.
[0013]
  SaidSingular part detectorClaims4As described above, a configuration is adopted in which discontinuous points appearing in the Mth derivative of the waveform of the demodulated signal are detected. Specifically, the claims5As described above, the singular part detection unit may be configured to generate a singular part detection signal by wavelet transform using a wavelet capable of detecting discontinuous points appearing in M-th order differentiation.
[0014]
  Claim6As described above, an erroneous measurement preventing unit that prevents erroneous measurement of the reception time due to noise present in the demodulated signal may be provided.
  With such a configuration, it is possible to prevent erroneous measurement by regarding the noise portion as a singular portion.
[0015]
  Specifically, the erroneous measurement prevention unit is claimed in the claims.7As described above, the correlation calculation unit performs a correlation calculation between the singular part detection signal and a predetermined reference signal for correlation, and outputs a higher level correlation signal as the similarity between the two signals is higher. The unit may be configured to measure a reception time based on a correlation signal from the correlation calculation unit.
[0016]
  In such a configuration, since the degree of similarity between the noise and the correlation reference signal is low, the correlation signal level becomes low and the reception time is not measured when noise is present.
  Claim8In the invention, when the demodulator is configured to output two first and second demodulated signals that are orthogonal to each other, the singular part detector includes the singular part included in the first and second demodulated signals. First and second detection units for detecting the portions and outputting first and second specific portion detection signals, respectively, wherein the first time measurement unit is output from the first and second detection units, respectively. A square sum calculation unit that outputs a square sum signal obtained by adding the square calculation results of the first and second singular part detection signals, and measures the reception time of the received signal based on the square sum signal from the square sum calculation unit It is good to have a configuration to do.
[0017]
  In such a configuration, the reception time measurement process can be executed without considering the phase difference when the received signal is demodulated by the square sum calculation unit.
  Claim9In the invention, the first time measurement unit performs a correlation calculation with a reference signal for correlation determined in advance with respect to the first and second singular part detection signals before the square sum calculation unit. The first and second correlation signals output from the square sum calculation unit are provided with a correlation calculation unit having first and second calculation units that output higher-level first and second correlation signals as the degree of similarity is higher. The reception time of the received signal may be measured by a square sum signal based on the above.
[0018]
  In such a configuration, the claims8In this configuration, the influence of noise can be eliminated.
  Claim10In the invention, the demodulated signal may be a PN sequence code signal. In this case, the claim11As described above, the despreading processing unit that despreads the demodulated signal and outputs the despreading signal, and the second time measurement that measures the reception time of the received signal based on the despreading signal from the despreading processing unit It is good to set it as a structure provided with a part.
[0019]
  With such a configuration, it is possible to perform reception time measurement processing using a spread spectrum communication system with excellent noise resistance.
  Claim12As described above, the reception time measured by the second time measurement unit is used as a reference reception time, and the existence range of the reception time measured by the first time measurement unit based on the reference reception time is determined in advance, and the first time It is good to set it as the structure provided with the measured value determination part which reports measurement time normality when the reception time measured by the measurement part is in the said existing range, and reports measurement time abnormality when it is outside the said existing range.
[0020]
In such a configuration, it becomes possible to detect a deterioration in accuracy of the reception time measured by the first time measurement unit due to the influence of noise based on the reference reception time that is hardly affected by noise. Moreover, it becomes possible to prevent the accuracy of the reception time measured by the first time measurement unit from deteriorating from the measurement accuracy of the reference reception time.
[0021]
  Claim12In the invention of claim13As described above, the reception time measured by the second time measurement unit is set as a reference reception time, and the demodulation signal range or the singular part detection signal range for performing time measurement processing by the first time measurement unit is set based on the reference reception time. The time measurement process may be performed by the first time measurement unit only for the demodulated signal range or the singular part detection signal range.
[0022]
  With this configuration, the time measurement process can be simplified.
  Claim14The invention is a distance measuring device for measuring a distance between a transmitting device and a receiving device, wherein the transmitting device includes a transmitting device side timing unit synchronized with the receiving device side timing unit, and the transmitting device side timing unit The transmission device is configured to transmit a transmission signal at a predetermined transmission time based on the time information.13From the reception time measurement device according to any one of the above, the reception device side timing unit, the transmission time information obtained from the time information of the reception device side timing unit, and the reception time measured by the reception time measurement device A distance calculating unit that calculates a distance between the transmitting device and the receiving device is provided.
[0023]
With this configuration, the distance between the transmission device and the reception device can be measured with high accuracy.
[0024]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, embodiments of the present invention will be described with reference to the drawings.
FIG. 1 is a block diagram showing a configuration of a communication apparatus to which a first embodiment of a reception time measuring apparatus according to the present invention is applied.
[0025]
In FIG. 1, the communication device includes a transmission device 1 that transmits a transmission signal and a reception device 10 that receives a transmission signal from the transmission device 1, and the reception device 10 receives a reception time of the transmission signal. A measuring device 11 is provided.
[0026]
The transmission device 1 is configured to transmit a transmission signal that generates a demodulated signal in which the singular part is locally present on the time axis and the signal level is continuous in the demodulation unit 12 of the reception time measuring device 11. A modulation signal generator 2 that D / A-converts the digital waveform data by a D / A converter to generate an analog modulation signal, and a modulation unit 3 that modulates the analog modulation signal and outputs a radio signal in the RF band. This radio signal is transmitted from the antenna.
[0027]
The reception time measuring device 11 of this embodiment demodulates a transmission signal input via an antenna and outputs a demodulated signal, and detects a singular part included in the demodulated signal by inputting the demodulated signal. A singular part detection unit 13 that outputs a singular part detection signal, and a time measurement unit 13 as a first time measurement unit that inputs the singular part detection signal and measures and reports the reception time of the received signal.
[0028]
The demodulator 12 outputs a demodulated signal in which the singular part is locally present on the time axis and the signal level is continuous. Here, the singular part is a discontinuous point that appears in the M-th derivative (M is a natural number) of the waveform of the demodulated signal.
[0029]
The singular part detection unit 13 detects a discontinuous point that appears in the Mth derivative (M is a natural number) of the waveform of the demodulated signal and outputs it as a singular part detection signal. Specifically, when wavelet transform capable of detecting discontinuous points appearing in M-order differentiation is performed, a singular part detection signal is generated and output from the wavelet transform result, and when discrete wavelet transform is used as the wavelet transform Is configured using a subband decomposition filter.
[0030]
FIG. 2 shows a configuration example of the singular part detection unit 13 using a subband decomposition filter that performs discrete wavelet transform.
In FIG. 2, the singular part detection unit 13 includes an A / D converter 13A that performs A / D conversion on the demodulated signal, and a sub-band decomposition filter 13B that is divided into four bands. The subband decomposition filter 13B includes three HPFs 13a to 13c, three LPFs 13d to 13f, and six samplers 13g to 13l.
[0031]
The demodulated signal is converted into a digital discrete demodulated signal by the A / D converter 13A and input to the subband decomposition filter 13B. The input discretized demodulated signal is input to the HPF 13a and LPF 13d, and each output is thinned out by the respective samplers 13g and 13h and down-sampled to ½. The output on the HPF 13a side at this time is D1. The output on the LPF 13d side is input to the subsequent HPF 13b and LPF 13e, each output is down-sampled by 1/2 by the respective samplers 13i and 13j, and the output on the HPF 13b side is D2. The output on the LPF 13e side is input to the subsequent HPF 13c and LPF 13f, the output is down-sampled to 1/2 by the respective samplers 13k and 13l, the output on the HPF 13c side is D3, and the output on the LPF 13f side is A. . One set of HPF and LPF divides the frequency band of the demodulated signal into two. Therefore, the subband decomposition filter 13B divides the discretized demodulated signal into four bands having a ratio of 4: 2: 1: 1 from the high frequency to the low frequency. The time measurement based on the detection of discontinuous points is more accurate as the frequency is higher. In this embodiment, the output D1 is output as a singular part detection signal.
[0032]
The wavelet transform is used in the field of signal processing and signal analysis. For example, the document “Singularity Detection and Processing with Wavelets” (S.Mallat and WLHwang: IEEE) Transactions on Information Theory, Vol. 38, No. 2, pp. 617-643 (1992-3)), “Wavelet analysis and filter bank” (G. Strang et al., Takahashi et al., Baifukan), “ "Wavelet Beginners Guide" (Hagiwara, Tokyo Denki University Press).
[0033]
  As shown in FIG. 3, for example, the time measuring unit 14 includes a level comparator 14A, a time measuring unit 14B, and a reception time calculating unit 14C. The level comparator 14A compares the signal level of the singular part detection signal (in this embodiment, the output D1 in FIG. 2) with a preset threshold value Vth, and when the signal level of the singular part detection signal becomes equal to or higher than the threshold value Vth. A signal having a logical value of 1 is output to the reception time calculation unit 14C. The timing unit 14B outputs the current time information to the reception time calculation unit 14C. The reception time calculation unit 14C performs level comparisonvesselThe reception time of the received signal is measured based on the time information when the signal of logical value 1 is input from 14A. When the singular part detection signal is positive or negative, as shown by the dotted line in FIG. 3, an absolute value converting circuit 14D may be provided to input the absolute value of the singular part detection signal to the level comparator 14A.
[0034]
Here, a demodulated signal that causes a discontinuous point by M-order differentiation will be described.
The waveform of the demodulated signal that generates a discontinuous point due to the Mth order differentiation is configured such that the signal level is continuous by connecting a plurality of waveforms represented by different continuous functions. As a result, discontinuous points appearing in the Mth order differentiation occur at connection points of a plurality of waveforms represented by different continuous functions. In the demodulated signal, a range in which the signal level changes with a plurality of waveforms represented by different continuous functions is referred to as a signal level changing portion.
[0035]
Specifically, the configuration method of the signal level changing portion is as follows.
The signal level changing part is represented by a variable t on the horizontal axis and is in the range of α ≦ t ≦ β. For example, t = γ1, γ2ε [α, β] (γ1 <γ2) Configure to occur. At this time, the signal level changing portion includes a waveform represented by a continuous function fa (t) defined in the range of α ≦ t ≦ γ1 and a continuous function fb (t) defined in the range of γ1 ≦ t ≦ γ2. And a waveform represented by a continuous function fc (t) defined in the range of γ2 ≦ t ≦ β, fa (t), fb (t) and fc ( t) is determined so as to satisfy at least the following conditions (1) to (4).
(1) fa (γ1) = fb (γ1), fb (γ2) = fc (γ2)
(2) fa (α) = fD1 (α), fc (β) = fD2 (β)
fD1 (t) and fD2 (t) are continuous functions representing the demodulated signals in the interval t ≦ α and t ≧ β, respectively.
(3) fa (t), fb (t), and fc (t) can be M-order differentiated by [α, γ1], [γ1, γ2], and [γ2, β], respectively.
(4) fa_- ^(M)(Γ1) ≠ fb₊ ^(M)(Γ1), fb_- ^(M)(Γ2) ≠ fc₊ ^(M)(Γ2)
f^(M)(T) is the derivative of the Mth derivative of the function f (t), and f_- ^(M)(Ρ) and f₊ ^(M)(Ρ) is the left derivative and the right derivative at t = ρ, respectively.
[0036]
The above (1) is a condition for preventing level discontinuity at the connection points γ1 and γ2 of each waveform of the signal level changing portion. The above (2) is a condition for preventing level discontinuity at the start point α and the end point β of the signal level changing portion. The continuity of the demodulated signal level is ensured by the above conditions (1) and (2). The above (3) means that fa (t), fb (t) and fc (t) are continuous functions having no level discontinuity, and the derivatives up to the Mth order are also continuous. The above (4) means that the M-th derivative of fa (t), fb (t), and fc (t) is not continuous at each connection point. Due to the above conditions (3) and (4), discontinuous points appearing in the M-th order differentiation of the waveform of the demodulated signal occur at connection points of a plurality of waveforms represented by different continuous functions.
[0037]
In order to make only one discontinuous point, γ1 = γ2 may be set. Further, there may be a point where the level does not become discontinuous due to the differentiation of the Mth floor among a plurality of connection points, and in that case, the left-side derivative and the right-side fine coefficient of the Mth floor at that connection point under the condition (4) above. The coefficients are equal.
[0038]
The difference between the function fa (t) (tε [α, γ1]) and the function fb (t) (tε [γ1, γ2]) is that when fa (t) is expanded to [γ1, γ2] as it is. , Ξ∈ [γ1, γ2] such that fa (ξ) ≠ fb (ξ) exists. Therefore, for example, even when the function fb (t) is a function fa (t−τ) obtained by shifting fa (t) by τ on the t-axis, when the functions fa (t) and fb (t) satisfy the above condition, Both functions are treated as different functions. Furthermore, when fa (t) and fb (t) are different functions, and fb (t) and fc (t) are different functions, fa (t) and fc (t) are independent of the above conditions. Treat as a different function.
[0039]
As described above, in wireless communication, the transmission signal has an occupied frequency bandwidth defined by the Radio Law or the like, so that the frequency bandwidth of the transmission signal is limited, and therefore the frequency bandwidth of the demodulated signal is also limited. If the frequency bandwidth is limited by a filter or the like, the singular part existing in the signal level changing part is affected, and the level of the singular part detection signal may be lowered. For this reason, it is desirable to configure the demodulated signal (and the modulation signal on the transmission side) so that the frequency band width of the transmission signal originally satisfies the restriction of the radio wave law, thereby eliminating the need for a filter for band restriction. Further, it is considered that the high frequency component of the signal is mainly due to the component where the signal level changes. Therefore, the function that gives the waveform of the signal level changing portion is determined so that the conditions (1) to (4) described above are satisfied and the transmission signal satisfies the occupied frequency bandwidth defined by the Radio Law or the like. desirable.
[0040]
4 to 6 show specific examples of waveforms for M = 1, 2, and 3. FIG. In each figure, a constant value of +1 or −1 is assumed except for the signal level changing portion.
FIG. 4 shows a waveform example of a signal level changing portion where M = 1, that is, a discontinuous point appears in the first-order differentiation of the waveform, and D1 of the subband decomposition filter 13B of the singular portion detection unit 13 for the waveform. -D3 output examples are shown.
[0041]
The waveform f1 (t) can be expressed by the following equation (1).
[0042]
[Expression 1]

[0043]
Here, a is a positive real number (provided that a = 1 in FIG. 4).
In f1 (t), tε [−T / 2, T / 2] is a signal level changing portion.
The point of t = ± T / 2 is composed of waveforms represented by two different functions. That is, the signal level changing portion is composed of a plurality of waveform connections expressed by different functions. Then, a discontinuous point that appears as a first-order differential at the connection point t = ± T / 2 is detected as a change in the outputs D1 to D3 of the subband decomposition filter 13B as illustrated.
[0044]
FIG. 5 shows a waveform example of a signal level changing part where M = 2, that is, a discontinuous point appears in the second derivative of the waveform, and D1 of the subband decomposition filter 13B of the singular part detection unit 13 for the waveform. -D3 output examples are shown.
[0045]
The waveform f2 (t) can be expressed by the following equation (2).
[0046]
[Expression 2]

[0047]
p is a natural number of 2 or more (however, p = 3 in the figure)
In the waveform f2 (t), tε [−T / 2, T / 2] is a signal level changing portion.
In order to suppress the occupied frequency bandwidth without using a filter or the like, it is desirable that the signal level changing portion of the demodulated signal is smoothly continuous with other portions. Specifically, it is desirable that the value of the first derivative of the waveform of the demodulated signal is continuous at the start point and end point of the signal level changing portion. Further, it is more desirable that the signal level of the waveform obtained by first-order differentiation of the waveform of the signal level changing portion is continuous. The waveform f2 (t) can realize such a characteristic. When p = 2 or more, the value of the first derivative of f2 (t) is zero at the start point and the end point of the signal level changing portion, and between the start point and the end point. Is continuous. Then, in the second derivative of f2 (t), a discontinuity appears at the connection point t = 0 of the waveform expressed by two different functions, and as shown in the figure, changes in the outputs D1 to D3 of the subband decomposition filter 13B Detected. In addition, when p = 2, discontinuity occurs in t = ± T / 2 in addition to t = 0 due to the second order differentiation. If p = 3 or more, it becomes continuous at t = ± T / 2 as shown in the figure. , Discontinuity occurs only at t = 0.
[0048]
FIG. 6 shows a waveform example of a signal level changing portion where M = 3, that is, a discontinuous point appears in the third-order differentiation of the waveform, and D1 of the subband decomposition filter 13B of the singular portion detection unit 13 for the waveform. -D3 output examples are shown.
[0049]
The waveform f3 (t) can be expressed by the following equation (3).
[0050]
[Equation 3]

[0051]
Similarly to the waveform f2 (t) in FIG. 5, the waveform f3 (t) has a continuous first-order differential value of the waveform of the demodulated signal at the start point and end point of the signal level change portion, and the waveform of the signal level change portion. The waveform obtained by first-order differentiation also satisfies the characteristic that the signal level is continuous. Further, the waveform f3 (t) has the same characteristics as the waveform obtained by the first-order differentiation for the second-order differentiation. In the waveform f3 (t), tε [−T / 2, T / 2] is a signal level changing portion. Then, as shown in the figure, a discontinuous point that appears as a third-order differentiation at a connection point t = ± T / 2 is detected as a change in the outputs D1 to D3 of the subband decomposition filter 13B.
[0052]
In the transmission apparatus 1, the modulation unit 3 can use various modulation methods such as amplitude modulation and angle modulation, for example, and the singular part (discontinuous point appearing in the M-th derivative of the waveform of the demodulated signal) is the time. It is only necessary to be able to transmit a transmission signal that generates a demodulated signal that exists locally on the axis and has a continuous signal level. Needless to say, the demodulator 12 of the receiving apparatus 10 is configured to demodulate the transmission signal from the transmitting apparatus 1.
[0053]
FIG. 7 shows a configuration example of the modulation unit 3 and the demodulation unit 12 shown in FIG.
7A shows the modulation unit 3, and FIG. 7B shows the demodulation unit 12.
In FIG. 7A, the modulated signal fM (t) is the same signal as the demodulated signal that is generated by the modulated signal generator 2 and should be originally generated by the demodulator 12 of the receiving apparatus 10, and the signal level change portion is the same. For example, it is a signal composed of the above-described functions f1 (t), f2 (t), f3 (t), and the like.
[0054]
A modulation / demodulation operation by the modulation unit 3 and the demodulation unit 12 in FIGS.
In the modulation unit 3, when the modulation signal fM (t) is input, the V / F converter converts the center frequency fo into a baseband signal fco (t) frequency-modulated with the modulation signal fM (t). The modulation signal fco (t) is converted into a radio band by a frequency converter using a modulation reference signal (frequency fr) of the signal source RF, and unnecessary frequency band components are removed by a filter to generate a transmission signal fSD (t). Then send. In the demodulator 12 shown in FIG. 5B, which receives and inputs the transmission signal fSD (t), the transmission signal fSD (t) is frequency-converted by a frequency converter using the demodulation reference signal (frequency fr) of the signal source LO. Then, unnecessary frequency band components are removed by the filter and input to the F / V converter as the baseband frequency conversion signal fDN (t), and the voltage proportional to the frequency of the signal fDN (t) by the F / V converter. A level frequency detection signal fDM (t) is generated. This frequency detection signal fDM (t) can be expressed by the following equation using the frequency fo as a reference for voltage output, for example.
[0055]
fDM (t) = D · fM (t) (D is a constant)
From this equation, fM (t) can be obtained as a demodulated signal. Therefore, the transmission apparatus 1 generates a transmission signal that generates a demodulated signal in which a singular part (a discontinuous point that appears in the Mth derivative of the waveform of the demodulated signal) exists locally on the time axis and the signal level is continuous. Can be sent.
[0056]
FIG. 8 shows another configuration example of the modulation unit 3 and the demodulation unit 12.
8A shows the modulation unit 3, and FIG. 8B shows the demodulation unit 12.
FIG. 8 shows a general quadrature modulation / demodulation circuit. The operation of the modulation unit 3 is as follows. The modulation signal fMI (t) is input as Is and the modulation signal fMQ (t) is input as Qs. The modulation reference signal (frequency fr) is modulated by Is input and Qs input to output a signal fQM (t), and unnecessary frequency band components of the output signal fQM (t) are removed by a filter to transmit a transmission signal fSD (t ). Here, fMI (t) and fMQ (t) are expressed as fMI (t)²+ FMQ (t)²= 1 is established, and the following equation is obtained.
[0057]
fMI (t) = fM (t)
fMQ (t) = sin [cos^-1(FM (t))]
However, (| fM (t) | ≦ 1).
[0058]
Here, fM (t) is the same as the modulation signal in FIG. 7 and is the same signal as the demodulated signal to be originally generated, and the signal level changing portion is, for example, the above-described functions f1 (t), f2 (t) or f3 ( t) and the like.
[0059]
Further, the filter and fM (t) are determined so that the transmission signal fSD (t) output from the filter is almost equal to fQM (t). As a result, fSD (t) = fQM (t).
[0060]
In the demodulator 12 that receives and inputs the transmission signal fSD (t), the orthogonal demodulator demodulates the received signal using the demodulation reference signal (frequency fr) from the signal source LO, and performs orthogonal demodulation that is orthogonal to each other. Signals fIR (t) and fQR (t) are output, unnecessary high frequency components are removed by a filter, and signals fDI (t) and fDQ (t) are output. Here, the signals fDI (t) and fDQ (t) can be expressed by the following equations.
[0061]
fDI (t) = E ′ · cos [cos^-1(FM (t)) + φ ′]
fDQ (t) = E ′ · sin [cos^-1(FM (t)) + φ ′]
Here, E ′ is a constant. When the phase difference φ ′ = 0 can be held in the above equation,
fDI (t) = E ′ · fM (t)
fDQ (t) = E ′ · (1−fM (t)²)^1/2
Thus, fM (t) can be obtained as fDI (t), and when the quadrature modulation / demodulation method shown in FIG. 8 is adopted, the demodulated signal for detecting the singular part can be obtained if the phase difference φ ′ = 0 is maintained. FDI (t) can be used as
[0062]
The measurement operation of the first embodiment will be described below.
In the transmission apparatus 1, in the modulation signal generation unit 2, the modulation unit 3 generates the modulation signal fM (t) in the case of FIG. 7A, and fMI (t) and fMQ in the case of FIG. 8A. (T) is generated and input to the modulation unit 3, and the transmission signal fSD (t) is transmitted from the modulation unit 3 as described above. Here, the modulation signal fM (t) or fMI (t) has a signal level changing portion such as the functions f1 (t), f2 (t), f3 (t), etc. as shown in FIGS. The transmission signal fSD (t) has a frequency bandwidth that satisfies the definition of the occupied frequency bandwidth such as the Radio Law.
[0063]
The receiving device 10 receives the transmission signal fSD (t) and inputs it to the reception time measuring device 11. In the reception time measuring device 11, the received signal is input to the demodulator 12, and as described above, the demodulator 12 outputs the demodulated signal fDM (t) (= fM (t)) in the case of FIG. In the case of FIG. 8B, fDI (t) (= fM (t); provided that the phase difference φ ′ = 0 is held) is output as the demodulated signal. The demodulated signal fDM (t) or fDI (t) is input to the singular part detection unit 13 having the configuration shown in FIG. 2, and the output D1 is output as a singular part detection signal. Here, when the signal level changing portion of the demodulated signal is composed of, for example, the function f1 (t) in which the discontinuity appears in the first-order differentiation, the output D1 of FIG. 5 is output as a singular part detection signal, and is composed of a function f3 (t) in which a discontinuity appears in the third order differentiation. If it is, the output D1 in FIG. 6 is output as the singular part detection signal.
[0064]
The singular part detection signal D1 from the singular part detection unit 13 is input to the level comparator 14A of the time measurement unit 14 and compared with a preset threshold value Vth, and the signal level of the singular part detection signal D1 becomes equal to or higher than the threshold value Vth. For example, a signal of logical value 1 is input from the level comparator 14A to the reception time calculation unit 14C. The time information is sequentially input to the reception time calculation unit 14C from the time measurement unit 14B, and the reception time calculation unit 14C uses the time when the signal of the logical value 1 is input from the level comparator 14A as the reception time of the reception signal. Report as. When the singular part detection signal D1 has a negative value as shown in FIG. 5, an absolute value conversion circuit 14D indicated by a dotted line in FIG. 3 may be provided and input to the level comparator 14A.
[0065]
According to such a configuration, it is possible to detect the reception time of the received signal with high accuracy without increasing the occupied frequency bandwidth of the radio signal using a signal having a continuous signal level. Therefore, a finite frequency resource can be used effectively, and the information transmission speed does not need to be reduced.
[0066]
Next, a second embodiment of the present invention will be described.
In wireless communication, it is necessary to consider noise mixing in the received signal. When noise is mixed in the received signal, the detection of the singular part included in the demodulated signal is disturbed, or the noise is erroneously detected as the singular part. There is a possibility that the time measurement accuracy will deteriorate.
[0067]
The second embodiment is configured to prevent erroneous measurement due to noise. For example, the configuration of FIG. 9 is used as an erroneous measurement prevention unit between the singular part detection unit 13 and the time measurement unit 14 of FIG. The correlation calculation unit 20 is provided.
[0068]
In FIG. 9, a correlation calculation unit 20 performs a correlation calculation between an input signal and a correlation reference signal, and outputs a higher level correlation signal as the similarity between the two is higher. A delay circuit 21, and n multipliers 22-1 to 22-n that respectively multiply the outputs Ds1 to Dsn of the delay elements D and the coefficients 1 to n for correlation calculation determined from the correlation reference signal, An adder circuit 23 that outputs the correlation signal by adding the outputs of the n multipliers 22-1 to 22-n.
[0069]
Next, the operation of the correlation calculation unit 20 in FIG. 9 will be described with reference to FIG.
For example, the demodulated signal includes a signal level changing portion composed of a waveform (see FIG. 4) in which a discontinuity occurs in the first order differentiation as shown in FIG. 10A, and the same signal level in units of Tx in the figure. It shall be composed of repeated change parts. In this case, a singular part detection signal D1 as shown in FIG. The singular part detection signal D1 propagates through the delay circuit 21 while being delayed by the delay elements D, and the multipliers 22-1 to 22-n apply the coefficients 1 to n to the outputs Ds1 to Dsn of the delay elements D being propagated. Multiply (Ds1 to Dsn in the figure are shown as coarser than actual). The calculation results are added by the adding circuit 23 and output as a correlation signal. The coefficients 1 to n are input signals so that a high-level correlation signal is generated from the adder circuit 23 when the output patterns of the outputs Ds1 to Dsn of the delay circuit 21 are generation patterns of singular part detection signals that should be generated originally. And the correlation reference signal are determined so as to realize the correlation calculation of the correlation reference signal. Here, the correlation reference signal is determined so that the singular part detection signal to be originally generated is regarded as a signal having high similarity. As a result, if the singular part detection signal D1 is input to the delay circuit 21 and the output pattern of the delay circuit 21 is the generation pattern of the singular part detection signal to be originally generated, a high level correlation signal is generated from the adder circuit 23. Then, as long as the original generation pattern continues, a high-level correlation signal equal to or higher than the threshold value Vth in the time measurement unit 14 is generated from the adder circuit 23 at a period of Tx as shown in FIG. If the generation pattern of the singular detection signal D1 differs from the original pattern due to noise mixing, the correlation signal level from the adder circuit 23 becomes lower than the threshold value Vth. Accordingly, when noise is present, the output of the logical value 1 is not generated from the level comparator 14A, the reception time is not reported, and erroneous measurement due to noise can be prevented.
[0070]
In order to prevent erroneous measurement due to noise, instead of the correlation calculation unit 20 in FIG. 9, for example, a noise suppression unit 30 having the configuration in FIG. 11 is provided as an erroneous measurement prevention unit between the demodulation unit 12 and the singular part detection unit 13. It is good also as a structure to provide.
[0071]
The noise suppression unit 30 in FIG. 11 includes an adder circuit 31 and a delay circuit 32, and inputs an input signal to the adder circuit 31 and the delay circuit 32. The delay circuit 32 transmits the input signal to the adder circuit 31 with a predetermined time delay. The adder circuit 31 adds the input signal and the delayed output of the delay circuit 32 and outputs the added signal as a demodulated signal that is input to the singular part detector 13. Here, the demodulated signal input to the noise suppression unit 30 is a signal composed of a predetermined signal waveform (hereinafter referred to as a demodulated signal block) as a unit, and this demodulated signal block is repeated. One or more level change parts are included.
[0072]
In such a configuration, if the delay time of the delay circuit 32 is set to be the same as the repetition cycle of the demodulated signal block, the adder circuit 31 adds two demodulated signal blocks that are continuous on the time axis if there is no noise. If noise is present, an addition signal having a waveform different from that of the demodulated signal block waveform to be originally generated is obtained.
[0073]
Note that the noise suppression unit 30 in FIG. 11 may be provided between the singular part detection unit 13 and the time measurement unit 14. Moreover, you may combine the correlation calculating part 20 of FIG. 9, and the noise suppression part 30 of FIG. For example, the noise suppression unit 30 is provided between the demodulation unit 12 and the singular part detection unit 13 and the correlation calculation unit 20 is provided between the singular part detection unit 13 and the time measurement unit 14. The structure etc. which provide the noise suppression part 30 and the correlation calculating part 20 sequentially from the singular part detection part 13 side between the time measurement parts 14 can be considered.
[0074]
Next, a third embodiment of the present invention will be described.
When the quadrature demodulation circuit of FIG. 8 is used for the demodulator 12, it is necessary to maintain the phase difference φ ′ = 0, but this generally requires a complicated configuration for maintaining the phase. To be complicated.
[0075]
The third embodiment of the present invention shown in FIG. 12 does not require a complicated phase holding configuration.
In FIG. 12, the demodulator 42 of the reception time measuring device 41 of the present embodiment has the configuration shown in FIG. Also, the singular part detection unit 43 detects a singular part included in the demodulated signal I and outputs a singular part detection signal I, and detects a singular part included in the demodulated signal Q by detecting the singular part. A second detector 43B that outputs the signal Q is provided. The first detection unit 43A and the second detection unit 43B have the same configuration as that in FIG. The time measurement unit 44 has a configuration in which a correlation calculation unit 45 and a square sum calculation unit 46 are added to the configuration of FIG. 3 and the square sum signal of the square sum calculation unit 46 is input to the level comparator 14A. Here, the demodulated signal I and the singular part detection signal I correspond to the first demodulated signal and the first singular part detection signal, and the demodulated signal Q and the singular part detection signal Q become the second demodulated signal and the second singular part detection signal. Equivalent to.
[0076]
The correlation calculation unit 45 performs a correlation calculation between the singular part detection signal I and the correlation reference signal and outputs a correlation signal I, and a correlation calculation between the singular part detection signal Q and the correlation reference signal. And a second operation unit 45B that outputs a correlation signal Q. The first calculation unit 45A and the second calculation unit 45B have the same configuration as that in FIG. Here, the correlation reference signals of the first calculation unit 45A and the second calculation unit 45B use the same signal, for example, the singular part detection signal I that should be generated when the phase difference φ ′ = 0. Correlation signal I corresponds to the first correlation signal, and correlation signal Q corresponds to the second correlation signal.
[0077]
As shown in FIG. 13, the sum of squares calculation unit 46 adds the outputs of the multiplier 46A that performs the square calculation of the correlation signal I, the multiplier 46B that performs the square calculation of the correlation signal Q, and the multipliers 46A and 46B. And an adder circuit 46C for outputting a square sum signal.
[0078]
If the noise is not taken into account, the correlation calculation unit 45 is unnecessary.
In such a configuration, the demodulated signals I and Q are
I = fDI (t) = E ′ · cos [cos^-1(FM (t)) + φ ′]
Q = fDQ (t) = E ′ · sin [cos^-1(FM (t)) + φ ′]
Is output.
[0079]
Each demodulated signal I, Q includes information on the singular part, and the output D1 from each of the first and second detection units 43A, 43B of the singular part detection unit 43 is, for example, the singular part detection signal as described above. I and Q are respectively output, and the respective addition signals are output as correlation signals I and Q from the first and second calculation units 45A and 45B of the correlation calculation unit 45 as described above. The correlation signals I and Q are squared by the multipliers 46A and 46B of the square sum calculator 46, and the square sum signal added by the adder circuit 46C is compared with the threshold value Vth by the level comparator 14A. The reception time is measured at 14C.
[0080]
According to such a configuration, the influence of the phase difference φ ′ can be eliminated by performing the square sum calculation by the square sum calculation unit 46. In addition, the influence of noise can be suppressed by providing the correlation calculation unit 45. Therefore, the reception time can be measured with high accuracy.
[0081]
  In each of the above-described embodiments, a PN code can be used as a demodulated signal. Since the PN code has a plurality of change points of data values, a part or all of the change points are described above.Using the signal level changing part configuration methodWhat is necessary is just to set it as the demodulated signal comprised as a signal level change part.
[0082]
Next, a fourth embodiment of the present invention in the case where the demodulated signal is composed of a PN code will be described.
FIG. 14 is a configuration diagram of the reception time measuring apparatus of the present embodiment.
[0083]
In FIG. 14, the reception time measuring device 51 of the present embodiment includes a demodulation unit 52, a singular part detection unit 53, a first time measurement unit 54, a despreading processing unit 55, a second time measurement unit 56, A measurement value determination unit 57 is provided. The demodulating unit 52 and the singular part detecting unit 53 have the same configuration as in the first embodiment, and the first time measuring unit 54 is provided with the correlation calculating unit 20 in FIG. 9 before the level comparator in FIG. It is a configuration.
[0084]
  The despreading processing unit 55 discretizes a demodulated signal composed of a PN code with an A / D converter, despreads the discretized demodulated signal with a digital matched filter, and has a waveform as shown in FIG. The despread signal is output. The second time measuring unit 56 has the configuration of FIG. 3, measures the time when the input despread signal is equal to or greater than a predetermined threshold value, and inputs the time as a reference reception time to the measured value determining unit 57. The measurement value determination unit 57 is a second time measurement unit.56Based on the reference reception time input from, normal / abnormal of the reception time input from the first time measuring unit 54 is determined and a determination signal is output. The despreading processing unit 55 and the second time measuring unit56Is a measurement method using spread spectrum communication as described above, and is described in “Spread spectrum communication and its application” (Marubayashi et al., IEICE).
[0085]
  Next, the operation of the fourth embodiment will be described.
  The demodulated signal from the demodulator 52 is input to the despreading processor 55 and the singular part detector 53, respectively. The output D1 is generated from the singular part detection unit 53 as a singular part detection signal and is input to the first time measurement unit 54. In the first time measurement unit 54, the correlation calculation unit 20 performs a correlation calculation between the singular part detection signal and the reference correlation signal to generate a correlation signal having a waveform shown in FIG. In the case of a demodulated signal composed of a PN code, the correlation signal becomes a high level in the period of the PN code. The correlation signal level and the threshold value are compared by the level comparator 14A, and the time when the correlation signal level becomes equal to or higher than the predetermined threshold value is output to the measurement value determination unit 57 as the reception time. Measurement value judgment unit57Determines whether the reception time of the first time measurement unit 54 is normal / abnormal based on the reference reception time input through the despreading processing unit 55 and the second time measurement unit 56 as described above.
[0086]
Since the despread signal and the correlation signal in FIG. 15 are generated from the same demodulated signal, there is a certain relationship between the despread signal and the high level generation time of the correlation signal. That is, there is a certain relationship between the reference reception time from the second time measurement unit 56 and the reception time. In addition, spread spectrum communication is a communication method with high noise resistance, and the reference reception time obtained from the second time measurement unit 56 is less accurate than the reception time obtained from the first time measurement unit 54, but noise resistance. Excellent in properties. Therefore, a range on the time axis where the reception time should exist (hereinafter referred to as a reception time existence range) is predetermined based on the reference reception time in consideration of measurement accuracy and the like. For example, the 1-chip width before and after the PN code around the reference reception time is defined as the reception time existence range. Then, in the measurement value determination unit 57, when the reception time of the first time measurement unit 54 is within the reception time existence range, the reception time is determined to be a correct value and reports normality. In some cases, the reception time is judged to be an incorrect value and an abnormality is reported.
[0087]
According to this configuration, the accuracy of the reception time measurement is not deteriorated compared to the measurement accuracy of the reference reception time, and the deterioration of the reception time measurement accuracy due to an increase in noise can be detected. improves.
[0088]
Note that, as indicated by a dotted line in FIG. 14, the reference reception time of the second time measurement unit 56 may also be output as measurement reception time information.
When the demodulation unit 52 is the quadrature demodulation circuit of FIG. 8, the reference reception time is affected by the phase difference φ ′. Therefore, a despreading processing unit that despreads each of the demodulated signals I and Q and a square sum calculation unit that calculates a sum of squares of each despread signal output from each despreading processing unit are provided. If the square sum signal from is input to the second time measuring unit 56 and the reference reception time is measured, the influence of the phase difference φ ′ on the reference reception time can be eliminated. Each of the despreading processing units may have the same configuration as the despreading processing unit of FIG. 14, and the square sum calculation unit may have the configuration of FIG. In this case, the singular part detection unit 53 and the first time measurement unit 54 have the configuration of the third embodiment shown in FIG.
[0089]
Next, a fifth embodiment of the present invention for reducing signal processing in the fourth embodiment of FIG. 14 will be described.
As described above, there is a certain relationship between the reception time obtained from the first time measurement unit 54 and the reference reception time obtained from the second time measurement unit 56, and the reception time existence range is determined from the reference reception time. Can do. The reception time has a one-to-one relationship with the correlation signal on the time axis, the correlation signal has a one-to-one relationship with the singular part detection signal in the correlation range on the time axis, and the singular part detection signal has a one-to-one relationship on the time axis. There is a one-to-one relationship with a demodulated signal in a predetermined range. FIG. 16 schematically shows a range of signal data used based on the above-described relationship for the singular part detection processing and correlation calculation for obtaining the reception time. As shown in FIG. A demodulated signal range corresponding to the time existence range can be determined. By performing the reception time measurement process only on the demodulated signal within the demodulated signal range in FIG. 16, the reception time measurement process can be reduced.
[0090]
FIG. 17 is a main part configuration diagram of the fifth embodiment of the present invention in which the reception time measurement process is performed only in a predetermined demodulated signal range.
In FIG. 17, the reception time measuring device of the present embodiment has a configuration in which a signal storage unit 60 is provided before the singular part detection unit 53 of FIG.
[0091]
The signal storage unit 60 stores an A / D converter 61 that converts a demodulated signal into a discretized demodulated signal, a discretized demodulated signal and time information in association with each other, and a reference reception time that is input thereafter. And a storage device 62 that defines a demodulated signal range for use in signal processing.
[0092]
The operation of this embodiment will be described.
When the demodulated signal is input to the signal storage unit 60, the demodulated signal is converted to a discretized demodulated signal by the A / D converter 61 and sequentially input to the storage device 62. The storage device 62 stores time information associated with each time a discretized demodulated signal is input. When the reference reception time is input from the second time measuring unit 56, a demodulated signal range is determined based on the reference reception time, and stored demodulated demodulated signals corresponding to the demodulated signal range are input in the input order. In addition to sequentially outputting to the singular part detection unit 53 as a read demodulated signal, the time information stored in association therewith is sequentially output to the first time measurement unit 54 as read time information. The singular part detection unit 53 outputs a singular part detection signal based on the input discrete demodulated signal to the first time measurement unit 54. The first time measurement unit 54 measures the reception time from the correlation signal based on the singular part detection signal and the readout time information.
[0093]
According to such a configuration, only signal processing within a predetermined range of the demodulated signal is required, so that signal processing for reception time measurement can be reduced.
In the case of the configuration of FIG. 17, the A / D converter in FIG. Further, the signal storage unit 60 may be provided after the singular part detection unit 53 so as to determine and store the singular part detection signal range. In this case, the singular part detection unit 53 has the configuration of FIG. Therefore, the A / D converter of the signal storage unit 60 is not necessary.
[0094]
Next, FIG. 18 shows a configuration of an embodiment of a distance measuring device according to the present invention to which the above-described reception time measuring device of the present invention is applied.
In FIG. 18, the distance measuring device 70 according to the present embodiment includes a transmitting device 80 and a receiving device 90.
[0095]
The transmitter 80 includes a modulation signal generator 81, a modulator 82, and a timer 83 as a transmitter-side timer. The modulation signal generation unit 81 and the modulation unit 82 have the same configuration as that of the reception time measuring device described above. The timer 83 is synchronized with the timer 92 on the receiving device 90 side with sufficient accuracy.
[0096]
The receiving device 90 includes the above-described receiving time measuring device 91 of the present invention, a time measuring unit 92 as a receiving device side time measuring unit, and a distance calculating unit 93. The distance calculation unit 93 calculates the distance between the transmission device 80 and the reception device 90 from the reception time information from the reception time measuring device 91 and the time information from the time measuring unit 92.
[0097]
Next, the distance measurement operation of the distance measurement device of this embodiment will be described.
The transmission device 80 generates a modulation signal by the modulation signal generation unit 81 and transmits a transmission signal at a predetermined transmission time based on the time information of the time measuring unit 83. In the reception device 90 that has received the transmission signal, the reception time measurement device 91 measures the reception time and notifies the distance calculation unit 93 of the measurement result. The distance calculating unit 93 obtains transmission time information determined in advance based on the time information of the time measuring unit 92 that is synchronized with the time measuring unit 83 on the transmission side. Thereby, the distance calculation part 93 calculates a distance by the arithmetic expression of (distance) = (light speed) × ((reception time) − (transmission time)).
[0098]
It should be noted that the transmission time is preferably set with a sufficient interval from the time calculated by (maximum distance to be measured) / (light speed) so that the distance calculation unit 93 can associate the reception time with the transmission time. .
[0099]
【The invention's effect】
As described above, according to the reception time measuring device of the present invention, the configuration is such that the reception time is measured by detecting the singular part of the demodulated signal in which the singular part is locally present on the time axis and the signal level is continuous. Therefore, it is possible to measure the reception time with high accuracy by using a demodulated signal having a continuous signal level. Therefore, it is not necessary to expand the occupied frequency bandwidth of the radio signal, the finite frequency resource can be effectively used, and the information transmission speed in the radio communication can be saved.
[0100]
Further, according to the distance measuring device of the present invention, since the distance between the transmitting device and the receiving device is measured using the receiving time measuring device of the present invention, the distance between the transmitting device and the receiving device can be measured with high accuracy.
[Brief description of the drawings]
FIG. 1 is a configuration diagram showing a first embodiment of a reception time measuring apparatus according to the present invention.
FIG. 2 is a configuration diagram of a singular part detection unit.
FIG. 3 is a block diagram of the time measuring unit.
FIG. 4 is a waveform example of a signal level changing portion where a discontinuity appears in the first-order differentiation.
FIG. 5 is a waveform example of a signal level changing portion where a discontinuity appears in second-order differentiation.
FIG. 6 is a waveform example of a signal level changing portion where a discontinuity appears in the third-order differentiation.
7A and 7B show configuration examples of a modulation unit and a demodulation unit, where FIG. 7A shows a modulation unit and FIG. 7B shows a demodulation unit.
FIGS. 8A and 8B show another configuration example of a modulation unit and a demodulation unit. FIG. 8A shows a modulation unit, and FIG. 8B shows a demodulation unit.
FIG. 9 is a configuration diagram of a correlation calculation unit in the second embodiment of the reception time measuring device according to the present invention.
FIG. 10 is a diagram illustrating the operation of the correlation calculation unit.
FIG. 11 is a configuration diagram of a noise suppression unit.
FIG. 12 is a block diagram of a main part of a third embodiment of a reception time measuring apparatus according to the present invention.
FIG. 13 is a block diagram of the sum of squares calculation unit.
FIG. 14 is a configuration diagram showing a fourth embodiment of a reception time measuring apparatus according to the present invention.
FIG. 15 is a diagram illustrating a relationship on a time axis between a despread signal and a correlation signal.
FIG. 16 is a diagram schematically showing the relationship of signal data ranges used for reception time measurement.
FIG. 17 is a block diagram showing a fifth embodiment of a reception time measuring apparatus according to the present invention.
FIG. 18 is a block diagram showing an embodiment of a distance measuring device according to the present invention.
19A and 19B are explanatory diagrams of a conventional transmission signal generation example in wireless communication, where FIG. 19A is a discontinuous data signal level signal, and FIG. 19B is a signal obtained by cutting high-frequency components with a low-pass filter.
[Explanation of symbols]
1,80 Transmitter
10,90 receiver
11, 91 Reception time measuring device
12, 42, 52 Demodulator
13, 43, 53 Specific part detector
14, 44, 54 Time measurement unit
20, 45 Correlation calculator
30 Noise suppressionPart
46    Square sum operation unit
55 Despreading processing unit
56 Second time measurement unit
57 Measurement value judgment unit
60 Signal storage unit
70 Distance measuring device
92 Timekeeping Department
93 Distance calculator

Claims (14)

A reception time measurement device that measures the reception time of a transmission signal from a transmission device,
A demodulator for outputting a demodulation signal by demodulating the transmission signal inputted,
And specificity portion detection unit for outputting a specific portion detection signal by detecting a singular part that is part of the demodulated signal output from the demodulation unit,
A first time measuring unit that measures a reception time of the transmission signal based on a singular part detection signal output by the singular part detection unit;
Equipped with a,
The transmission signal is a signal that generates a demodulated signal in which the singular part is locally present on the time axis and the signal level is continuous, and the singular part is a discontinuity that appears as an M-th derivative of the waveform of the demodulated signal. A reception time measuring device characterized by being a point . The demodulated signal has a signal level change portion configured by connecting a plurality of waveforms represented by different continuous functions, and the discontinuity point is a signal generated at a waveform connection point in the signal level change portion. The reception time measuring device according to claim 1 . The reception time measuring apparatus according to claim 2 , wherein the waveform obtained by first-order differentiation of the waveform of the demodulated signal is continuous at least at the start point and the end point of the signal level changing portion. The reception time measuring device according to any one of claims 1 to 3 , wherein the singular part detection unit is configured to detect a discontinuous point that appears in an M-th derivative of a waveform of the demodulated signal. The reception time measuring apparatus according to claim 4 , wherein the singular part detection unit is configured to generate a singular part detection signal by wavelet transform using wavelets capable of detecting discontinuous points appearing in M-order differentiation. The reception time measuring device according to any one of claims 1 to 5 , wherein an erroneous measurement preventing unit that prevents erroneous measurement of the reception time due to noise present in a demodulated signal is provided. The erroneous measurement prevention unit is a correlation calculation unit that performs a correlation calculation between a singular part detection signal and a predetermined reference signal for correlation and outputs a higher level correlation signal as the similarity between the two signals is higher. The reception time measuring device according to claim 6 , wherein the one time measuring unit is configured to measure a reception time based on a correlation signal from the correlation calculating unit. When the demodulator is configured to output two first and second demodulated signals that are orthogonal to each other, the singular part detector detects the singular part included in the first and second demodulated signals, respectively. And first and second detection units that output first and second singular part detection signals, and the first time measurement unit outputs first and second outputs from the first and second detection units, respectively. A configuration is provided that includes a square sum calculation unit that outputs a square sum signal obtained by adding the square calculation results of the singular part detection signals, and measures the reception time of the received signal based on the square sum signal from the square sum calculation unit. The reception time measuring device according to any one of claims 1 to 5 . The first time measurement unit performs a correlation calculation with a predetermined correlation reference signal for the first and second singular part detection signals before the square sum calculation unit, and the similarity between both signals is high. A square operation based on the first and second correlation signals output from the square sum calculation unit, comprising a correlation calculation unit having first and second calculation units for outputting first and second correlation signals at higher levels. The reception time measuring apparatus according to claim 8 , wherein the reception time of the reception signal is measured by a sum signal. The reception time measuring device according to any one of claims 1 to 9 , wherein the demodulated signal is a PN sequence code signal. A despreading unit that performs despreading processing on the demodulated signal and outputs a despread signal; and a second time measuring unit that measures the reception time of the received signal based on the despread signal from the despreading unit. The reception time measuring device according to claim 10 , which is configured. Using the reception time measured by the second time measurement unit as a reference reception time, a range of the reception time measured by the first time measurement unit is determined in advance based on the reference reception time, and measured by the first time measurement unit The reception time measuring device according to claim 11 , further comprising a measurement value determining unit that notifies the measurement time normality when the received reception time is within the existence range and reports the measurement time abnormality when the reception time is outside the existence range. Using the reception time measured by the second time measurement unit as a reference reception time, a demodulation signal range or a singular part detection signal range for performing time measurement processing by the first time measurement unit is set based on the reference reception time, and the demodulation The reception time measuring device according to claim 12 , wherein only the signal range or the singular part detection signal range is configured to perform time measurement processing in the first time measurement unit. A distance measuring device for measuring a distance between a transmitting device and a receiving device,
The transmission device includes a transmission device side timing unit synchronized with the reception device side timing unit, and is configured to transmit a transmission signal at a predetermined transmission time based on time information of the transmission device side timing unit,
The reception device includes: the reception time measurement device according to any one of claims 1 to 13 , the reception device side timing unit, transmission time information obtained from time information of the reception device side timing unit, and the reception A distance measurement device comprising: a distance calculation unit that calculates a distance between a transmission device and a reception device from a reception time measured by the time measurement device.

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