JP3980324B2 - Motor driving current control device and method thereof - Google Patents

Description

とする。
【００４１】
つづいて、本システムの電池電圧Ｅとコンデンサ電圧Ｖｃの関係から電圧Ｖｗを定義する。これは、相電流の振幅Ｉａｃは、コンデンサ電圧Ｖｃから電池電圧Ｅを減算した電圧が最大値になるからである。また、同時に各コイルに印加される電圧ｖｖは上述の電流ｉａｃと一定の位相差（力率ｃｏｓφ）で推移すると仮定する。
【００４２】
すなわち、
【数２】
Ｖｗ＝Ｖｃ−Ｅ （６）
ｖｖ＝Ｖｗｇ（θ＋φ） （７）
とする。
【００４３】
また、モータ出力Ｗｏと各コイルがする仕事との関係は、コイルが６本あるので、次式のように整理できる。
【００４４】
【数３】

また、モータ出力は損失が十分に小さいとしてＷｏ＝ｉｅＥと近似できる。これの関係より次式を得る。
【００４５】
【数４】
ｉｅ＝Ｗｏ／Ｅ （１３）
以上より、各相コイルを流れる電流は式（１２）、（１３）で求められるＩａｃ，ｉｅを利用し次式で求められる。ただし、ｉｅのリップル分は考慮していない。
【００４６】
【数５】

次に、解析に用いる条件を示す。電池電圧Ｅ＝４２Ｖまたは１０５Ｖ、コンデンサ電圧Ｖｃ＝２１０Ｖ（昇圧率Ｖｃ／Ｅ＝５，または２）、力率ｃｏｓθ＝０．８で、モータ出力Ｗｏに対する交流電流振幅の大きさの最大値の通電方法による違いを示す。
【００４７】
この結果を、図３〜図５に示す。これらの図は、昇圧率の違いによる相電流最大値の違いを示しており、横軸がモータ出力、縦軸が相電流最大値（ｉｍａｘ）、実線が相電流最大値、破線が相電流最大値のうちの直流成分（ｉｅ／３）を示している。
【００４８】
図３は従来の通電時の相電流最大値、図４は零相リップル非許容条件での最大抑制通電時の相電流最大値、図５は零相リップル許容条件での最大抑制通電方法（４．２．２ 節）時の、相電流最大値を示している。
【００４９】
これらの図より以下のことがわかる。
【００５０】
・いずれの場合にも、相電流の大きさは昇圧率により大きく変化し、昇圧率が高いほうが相電流に占める直流成分の比率が大きい。
【００５１】
・また、通電法の違いによる相電流の大きさの抑制効果が確認できる。
【００５２】
・Ｗｏ＝４０ｋＷ、昇圧比５倍で相電圧の最大値（交流成分、直流成分）を比較すると、図３の従来通電では、最大値４７７Ａ（１５９，３１７Ａ）、図４では、４５４Ａ（１３６，３１７Ａ）、図５では、４０２Ａ（８５，３１７Ａ）である。
【００５３】
「本発明の基本となる従来の通電方法の説明」
図３に示す２ＹＤＣ可変型インバータの従来の通電方法について、説明する。図１に示す２ＹＤＣ可変型インバータに、通常流される相電流ｉｕ１，ｉｖ１，ｉｗ１，ｉｕ２，ｉｖ２，ｉｗ２は、零相電流をｉｅ、交流電流振幅をＡ、ロータ回転数、回転角をそれぞれω，θ（θ＝ωｔ）とすれば、次式で表される。
【００５４】
【数６】

ここで、Ａ＝１（Ａ），ｉｅ＝３（Ａ）とすると、式（１５）〜（２０）は、図６の関係が有る。ただし、図６は上段から、ｉｕ１，ｉｕ２各々の電流が１段目、ｉｕ１とｉｕ２との電流の和が２段目、相電流をｄｑ軸変換した後のｄ軸電流ｉｄ１，ｉｄ２が３段目、ｑ軸電流ｉｑ１，ｉｑ２が４段目、最終段が零相電流ｉｅの３分の１（１相分）を示してある。
【００５５】
ここで、リラクタンストルクを考えなければｉｑ１＋ｉｑ２がモータトルクに寄与する電流成分（今回のケースでは、磁石位置を解析に入れていないので、ｉｕ１＋ｉｕ２がモータトルクに寄与する電流成分ともいえる）、ｉｅが電池とコンデンサ間を流れる電流である。そして、この時の相電流の大きさの最大値は２．００（Ａ）である。図６の関係のうち、モータ駆動トルクを発生するための電流と電池・コンデンサ間の電流の条件は、式（２１）で書くことができる。
【００５６】
【数７】

さらに、式（２２）、（２３）を導入することにより、式（２１）は以下のようにも書きかえられる。なお、式（２４）において、ｉｄ，ｉｑは、ｄｑ軸で表される電流成分で、ここではコンスタントとなる。
【００５７】
【数８】

モータの巻線間に位相差がある場合（あるスター結線のコイル位置と、他のスター結線のコイル位置とが角度ξでずれている場合）には、通電される電流は式（２５）〜（３０）の様に書きかえられ、式（２４）は式（３１）となる。
【００５８】
【数９】

Ａ＝１（Ａ）、ｉｅ＝３（Ａ）、ξ＝３０°とすると、式（２５）〜（３０）は、図７の関係が有る。このように、コイルの位相差を考慮した場合にも図６と同様の関係があることが分かる。
【００５９】
「実施形態の２ＹＤＣ可変型インバータの説明」
図４の実施形態では、零相電流におけるリップルの発生を許容せずに相電流の最大値を抑制する。
【００６０】
すなわち、本実施形態では、図１の２ＹＤＣ可変型インバータにおいて、相電流ｉｕ１，ｉｖ１，ｉｗ１に対し所定の関数を加算することで、最大振幅を抑制する。そして、加算した関数を相電流ｉｕ２，ｉｖ２，ｉｗ２から減算することで、モータの出力トルクを変動させることなく、電流の最大振幅を抑制する。また、本実施形態では、零相電流のリップルを許容しない。
【００６１】
モータ出力トルクおよび零相電流の大きさを変えずに、電流振幅を減少させるためには、相電流ｉｕ１，ｉｖ１，ｉｗ１，ｉｕ２，ｉｖ２，ｉｗ２は、式（２１）の関係を満足する必要がある。すなわち、次式（３２）を満足する必要がある。この式は、各スター結線の対応する相の電流の和が正弦波であり、かつ各スター結線内の各相の電流の総和が零相電流の値、若しくは零相電流の値の符号を変えたものに等しいことを意味している。
【００６２】
【数１０】

ここで式（３２）の左辺の行列のランクが４であり、２つのフリーパラメータｆｕ（θ），ｆｖ（θ）を導入し、式（３２）を満足するように、以下のような十分条件に書きかえることができる。
【００６３】
【数１１】

ここで、ｆｕ（θ），ｆｖ（θ），ｆｗ（θ）が、設計に利用できるパラメータ（自由度は２）である。
【００６４】
従って、式（３３）〜（４２）を満たすｆｕ（θ），ｆｖ（θ），ｆｗ（θ）（自由度は２）を与えることで、出力トルクおよび零相電流を変動させることなく、相電流ｉｕ１，ｉｖ１，ｉｗ１，ｉｕ２，ｉｖ２，ｉｗ２を変動させることができる。そして、ｆｕ（θ），ｆｖ（θ），ｆｗ（θ）を相電流ｉｕ１，ｉｖ１，ｉｗ１，ｉｕ２，ｉｖ２，ｉｗ２の最大振幅を減少するように選択することで、所期の目的を達成することができる。
【００６５】
図５の実施形態では、条件を緩和し、零相電流におけるリップルの発生を許容して、相電流の最大値を抑制する。この場合には、式（３９）の条件をはずすことができる。従って、ｆｕ（θ），ｆｖ（θ），ｆｗ（θ）を選択する場合の自由度が広がる。そして、相電流の最大値をより小さくすることが可能となる。
【００６６】
また、上述の説明では、２つのモータコイルＭ１，Ｍ２間に位相差がないことを前提とした。実際には、コイル間に位相差を持たせて配置する場合も多い。この場合には、コイル電流に対応した位相差を持たせることで、位相差を持たせたことの影響を排除する。
【００６７】
このような各スター結線のコイル間に位相差ξを持つ場合には、式（３１）が、式（２１）に変わる条件となる。すなわち、モータ発生トルクや零相電流の大きさを変えずに電流振幅を減少するためには、相電流ｉｕ１，ｉｖ１，ｉｗ１，ｉｕ２，ｉｖ２，ｉｗ２が、式（４３）を満足する必要がある。この式は、各スター結線の対応するｄｑ軸電流の和が一定であり、かつ各スター結線内の各相の電流の総和が零相電流の値、若しくは零相電流の値の符号を変えたものに等しいことを意味している。
【００６８】
【数１２】

ここで、式（４３）を満たす解の１つとして、前述の場合と同様に以下の結果が導かれる。
【００６９】
【数１３】

ここで、ｆｕ（θ），ｆｖ（θ），ｆｗ（θ），ｈｕ（θ），ｈｖ（θ），ｈｗ（θ）が、設計に利用できるパラメータである。さらに、ξ＝０°の時、式（３３）、（４２）の関数は、式（４４）、（５９）を満たす。
【００７０】
そして、式（４４）〜（５９）を満たすｆｕ（θ），ｆｖ（θ），ｆｗ（θ），ｈｕ（θ），ｈｖ（θ），ｈｗ（θ）を与えることで、出力トルクおよび零相電流を変動させることなく、相電流ｉｕ１，ｉｖ１，ｉｗ１，ｉｕ２，ｉｖ２，ｉｗ２を変動させることができる。さらに、ｆｕ（θ），ｆｖ（θ），ｆｗ（θ），ｈｕ（θ），ｈｖ（θ），ｈｗ（θ）を相電流ｉｕ１，ｉｖ１，ｉｗ１，ｉｕ２，ｉｖ２，ｉｗ２の最大値を抑制できる形にすることで、所期の目的を達成することができる。
【００７１】
また、条件を緩和し、零相電流にリップル電流を許せば、式（５０）、（５１）に代り、ｆｕ（θ）＋ｆｖ（θ）＋ｆｗ（θ）＋ｈｕ（θ）＋ｈｖ（θ）＋ｈｗ（θ）＝０が条件となる。
【００７２】
「零相電流にリップルを許さない場合の具体例」
コイル間位相差ξ＝０°で、上述の条件を満足する通電方法は、ｆｕ（θ），ｆｖ（θ），ｆｗ（θ）を式（６０）〜（６２）のように設定することにより得られる。なお、式のｇ１は、式（４０）〜（４２）の条件を満たすために入れた定数で、この場合はｇ１＝０．８６７である。
【００７３】
【数１４】

Ａ＝１（Ａ），ｉｅ＝３（Ａ）の場合について、ｆｕ（θ）の波形をｉｕ１と比較し、図８、図９に示す。図８は、ｆｕ（θ）の図とｉｕ１の図の縦軸のスケールをあわせたもの、図９は、ｆｕ（θ）の波形を見やすくするために拡大したものである。図より、ｆｕ（θ）の波形は正弦波のピーク部分を６０度幅で切り出し、それを正側負側正側の順にならべ、負側の大きさを正側の２倍に設定した波形となっている。すなわち、ｉｕ１の最大ピークのところを最も抑制する波形となっている。
【００７４】
従って、このような形のｆｕ（θ），ｆｖ（θ），ｆｗ（θ）をサインカーブに加算することによって相電流を最大電流を抑制することができ、かつこれに基づく出力トルクの変化はない。さらに、この例では、零相電流を発生しないという条件も満たしている。
【００７５】
さらに、式（６０）〜（６２）の条件を用いた結果を図１０に示す。図より、以下のことがわかる。
【００７６】
・零相電流
零相電流はｉｅ＝３（Ａ）であり、リップル成分は含まれない。
【００７７】
・トルク
モータトルクを発生する電流（ｉｕ１＋ｉｕ２）は図６と同等で、意図どおりのトルクを発生している。
【００７８】
・相電流の大きさ
相電流の大きさの最大値は、１．８６６（Ａ）である。大きさの内訳は、交流による成分が０．８６６Ａ、直流による成分が１Ａである。
【００７９】
このように、式（６０）〜（６２）に示すｆｕ（θ），ｆｖ（θ），ｆｗ（θ）を利用することによって、零相電流、モータ出力トルクに影響を与えることなく、相電流の最大値を抑制することができる。
【００８０】
また、ξ＝０°で、零相電流にリップルを許さない場合の他の例として、３倍の高調波で交流振幅を変調する場合を示す。
【００８１】
すなわち、Ａ＝１（Ａ），ｉｅ＝３（Ａ）の場合について、ｆｕ（θ）の波形をｉｕ１と比較し、図１１に示す。この波形は、元々の交流波形の振幅を３倍の周波数を持つ正弦波で、次式のように変調した波形になっている。
【００８２】
【数１５】

さらに、式（６３）と（６４）はつぎのように整理できる。
【００８３】
【数１６】

ここで、ｆｕ（θ）＝αｓｉｎ（３θ）Ａｓｉｎ（θ）とおけば、式（３３）〜（４２）の条件を満足する。すなわち、ｆｕ（θ）＝αｓｉｎ（３θ）Ａｓｉｎ（θ）に設定することで、下記のような結果が得られる。
【００８４】
図１１は、このｆｕ（θ）を示したものである。さらに、このｆｕ（θ）を用いた結果を図１２に示す。図より、以下のことがわかる。
【００８５】
・零相電流
零相電流は平均値はｉｅ＝３（Ａ）である。その大きさは、加えたｆｕ（θ）の３倍の振幅である。
【００８６】
・トルク
モータトルクを発生する電流（ｉｕ１＋ｉｕ２）は図６と同等であり、意図どおりのトルクを発生している。
【００８７】
・相電流の大きさ
相電流の大きさの最大値は１．８７２（Ａ）である。大きさの内訳は、交流による成分が０．８７２Ａ、直流による成分が１Ａである。
【００８８】
次に、各スター結線間のコイル位置に位相差が３０°ずれた場合（ξ＝３０°）の結果を図１３に示す。この図より、以下のことがわかる。
【００８９】
・零相電流
零相電流は、ｉｅ＝３（Ａ）であり、リップル成分は含まれない。
【００９０】
・トルク
モータトルクを発生する電流（ｉｄとｉｑ）は図６と同等で、意図どおりのトルクを発生している。
【００９１】
・相電流の大きさ
相電流の大きさの最大値は、１．８６６（Ａ）である。大きさの内訳は、交流による成分が０．８６６Ａ、直流による成分が１Ａである。
【００９２】
・相電流の波形
ここで用いた指令値は、電流の大きさを抑制するために、急峻に変化する波形である。しかし、実際の場合には、これをフィルタリングし高周波成分を除くことにより実現する。ただし、その場合は若干電流の抑制効果は悪くなる。
【００９３】
「零相電流にリップルを許す場合の具体例」
ξ＝０°で、零相電流リップルを許す条件で、相電流の大きさを抑制できる通電方法の１つは、ｆｕ（θ）、ｆｖ（θ），ｆｗ（θ）を式（６７）〜（６９）のように流すことである。なお、式のｇ２は、式（４０）〜（４２）の条件を満たすために入れた定数で、この場合はｇ２＝−０．６３７である。
【００９４】
【数１７】

Ａ＝１（Ａ）、ｉｅ＝３（Ａ）の場合について、ｆｕ（θ）の波形をｉｕ１と比較し、図１４に示す。さらに、式（６７）〜（６９）の条件を用いた結果を図１５に示す。図より、以下のことがわかる。
【００９５】
・零相電流
零相電流の平均値はｉｅ＝３（Ａ）であるが、リップル成分が含まれその大きさは０．４６Ａである。
【００９６】
・トルク
モータトルクを発生する電流（ｉｕ１＋ｉｕ２）は図６と同等で、意図どおりのトルクを発生している。
【００９７】
・相電流の大きさ
相電流の大きさの最大値は、１．６３（Ａ）である。大きさの内訳は、交流による成分が０．６３Ａ、直流による成分が１Ａである。
【００９８】
次に、ξ＝０°で、零相電流にリップルを許す場合のその他の方法の一例として、６倍の高調波を加える場合を示す。Ａ＝１（Ａ）、ｉｅ＝３（Ａ）の場合について、ｆｕ（θ）の波形をｉｕ１と比較し、図１６に示す。なお高調波の振幅は、相電流が最小になるように最適化した値を用いている。
【００９９】
さらに、このｆｕ（θ）を用いた結果を図１７に示す。図より、以下のことがわかる。
【０１００】
・零相電流
零相電流は平均値はｉｅ＝３（Ａ）であるが、リップル成分が含まれる。その大きさは、加えたｆｕ（θ）の３倍の振幅である。
【０１０１】
・トルク
モータトルクを発生する電流（ｉｕ１＋ｉｕ２）は図６と同等で、意図どおりのトルクを発生している。
【０１０２】
・相電流の大きさ
相電流の大きさの最大値は１．９６（Ａ）である。大きさの内訳は、交流による成分が０．９６Ａ、直流による成分が１Ａである。
【０１０３】
「１ＹＤＣ電圧可変型インバータでの通電」
モータコイルを１つとし、その中性点にバッテリを接続した１ＹＤＣ電圧可変型インバータにおける振幅最大値の低減について、説明する。
【０１０４】
図１８には、１ＹＤＣ電圧可変型インバータを利用したシステムの構成例が示されている。このように、基本的に、図１の２ＹＤＣ型のシステムにおいてインバータおよびモータを１つにした構成である。コンデンサＣは、インバータＩＮＶ１に電源として供給されており、このインバータＩＮＶ１の３相出力がモータコイルＭ１のＵ，Ｖ，Ｗ相のコイル端に接続されている。そして、バッテリＢは、負極がコンデンサの一端に接続され、正極がモータコイルＭ１の中性点に接続されている。
【０１０５】
従って、この構成によれば、バッテリ電圧Ｅに対するインバータＩＮＶ１の入力電圧Ｖｃは、インバータＩＮＶ１の変調率ｄによって、Ｖｃ＝Ｅ／ｄで決定される。
【０１０６】
「具体例」
このシステムにおいて、零相電流を制御することによって、出力トルクを変動させることなく、相電流の振幅の最大値を抑制する。すなわち、１ＹＤＣ電圧可変型インバータの電流低減方法の一例について説明する。ここでは、零相電流に回転周期の３倍の周期を持つ高調波を加える。
【０１０７】
Ａ＝１（Ａ）、ｉｅ＝３（Ａ）の場合について、３倍の高調波を加えない場合（従来）と加えた場合との比較を、図１９、２０に示す。なお高調波の振幅は、相電流が最小になるように最適化した値を用いている。図より、以下のことがわかる。
【０１０８】
・零相電流
零相電流は平均値はｉｅ＝３（Ａ）であるが、リップル成分が含まれる。
【０１０９】
・トルク
モータトルクを発生する電流（ｉｑ１とｉｄ１）は図１９と図２０と同等で、意図どおりのトルクを発生している。
【０１１０】
・相電流の大きさ
相電流の大きさの最大値は、１．８９（Ａ）であり、図１８の２Ａの最大値を、０．１１Ａ低減している。大きさの内訳は、交流による成分が０．８９Ａ、直流による成分が１Ａである。
【０１１１】
このように、零相電流に、回転周期の３倍の周期を持つ高周波を加えることによって、出力トルクに影響を与えることなく、相電流の振幅の最大値を抑制することができる。
【０１１２】
なお、ここでは、回転周期の３倍の周期を持つ高周波を加えたが、上述のように６倍の周期のものでも、他の関数でもよい。すなわち、モータトルクに影響を与えないという条件を維持でき、かつ相電流の最大振幅に同期して値が下がるような関数であれば、どのような関数を利用することもできる。
【０１１３】
【発明の効果】
以上説明したように、本発明によれば、トルクの増減を伴うことなく、相電流の最大電流値を抑制することができ、モータとしての機能を損なうことなく、デバイスの電流容量を下げられるので，同等性能を保ちつつ、システムの低コスト化を実現できる。トルクリップルを抑制することで、モータの機能を十分なものにできる。
【０１１４】
また、電流抑制には、高周波成分が電流に重畳する必要がある。このため、高周波域まで電流を制御することが必要になる。しかし、回転数により制御を切り替えることにより、より効果的な制御が可能になる。
【０１１５】
すなわち、電流値が大きい低回転域で振幅最大値抑制を行うため、元々の制御周波数帯域が低いので，高調波を重畳しても制御が極端に難しくはならない。一方、高回転域では従来法を用いるため前述の高周波重畳時の制御問題は発生しない。さらに、中回転領域において、零相電流のリップルを抑制することで、適切な制御が行える。
【０１１６】
このような制御の切り替えにより、電流抑制による制御上の問題を回避しつつ，電流抑制を実現できる。
【図面の簡単な説明】
【図１】 ２ＹＤＣの装置構成を示す図である。
【図２】 電圧指令値と、インバータ搬送波の関係を示す図である。
【図３】 従来の通電方法における電流最大振幅値を示す図である。
【図４】 零相リップル非許容時における電流低減の場合の電流最大振幅値を示す図である。
【図５】 零相リップル許容時における電流低減の場合の電流最大振幅値を示す図である。
【図６】 従来の相電流と零相電流などを示す図である。
【図７】 従来の相電流と零相電流など（コイル間位相差作がある場合）を示す図である。
【図８】 リップル電流を抑制する場合における相電流と関数ｆを示す図である。
【図９】 図８の拡大図である。
【図１０】 リップル電流を抑制する場合における相電流などを示す図である。
【図１１】 ３倍の高調波で交流振幅を変調する場合における相電流およびその振幅最大値を示す図である。
【図１２】 ３倍の高調波で交流振幅を変調する場合における相電流などを示す図である。
【図１３】 リップルを抑制する場合（位相差あり）における相電流などを示す図である。
【図１４】 リップル電流を許容する場合における相電流および関数ｆを示す図である。
【図１５】 リップル電流を許容する場合における相電流などを示す図である。
【図１６】 ６倍の高調波で変調する場合における相電流およびその振幅最大値を示す図である。
【図１７】 ６倍の高調波で交流振幅を変調する場合における相電流などを示す図である。
【図１８】 １ＹＤＣの装置構成を示す図である。
【図１９】 １ＹＤＣの従来法における相電流などを示す図である。
【図２０】 １ＹＤＣにおいて３倍の高調波で交流振幅を変調する場合における相電流などを示す図である。
【符号の説明】
Ｂ バッテリ、Ｃ コンデンサ、ＩＮＶ１，ＩＮＶ２ インバータ、Ｍ１，Ｍ２ モータコイル。[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a motor drive current control apparatus and method for controlling a motor drive current supplied to a plurality of coils whose one ends are commonly connected to a neutral point, and more particularly to suppression of a maximum amplitude value of the motor drive current.
[0002]
[Prior art]
Conventionally, a system has been proposed in which a battery is connected to a neutral point in a star (star-shaped) motor coil and a capacitor is connected to a power line of an inverter that controls each phase current of the motor. For example, it is shown by Unexamined-Japanese-Patent No. 10-337047 etc.
[0003]
In this system, by superimposing an alternating current on a zero-phase current that is a current other than each phase current flowing through a neutral point, the battery voltage can be used for charging a capacitor by using a motor coil, and the motor Can be driven as usual.
[0004]
Furthermore, Japanese Patent Application No. 2000-346967 shows that a battery is arranged between the neutral points of two motor coils and the power supply potential (capacitor voltage) of an inverter connected to the two motor coils is controlled. ing.
[0005]
[Problems to be solved by the invention]
However, in these conventional techniques, an alternating current is superimposed on the zero-phase current, and this is added to the phase current of each phase coil. For this reason, the magnitude | size of the electric current which flows into each phase coil becomes large, and there exists a problem that it is necessary to enlarge the current capacity of the device in an inverter inevitably.
[0006]
[Means for Solving the Problems]
The present invention relates to a motor drive current control apparatus or method for controlling a motor drive current supplied to a plurality of coils whose one ends are commonly connected to a neutral point. In the motor drive current control apparatus or method, the neutral point holds a neutral point potential. A power supply is connected, and by controlling the current supplied to each phase coil, the zero-phase current, which is a current other than the phase current entering and exiting the neutral point, is fluctuated, whereby the output torque of the motor is set to a predetermined value. The maximum value of the amplitude of the phase current is suppressed while maintaining the same.
[0007]
Thus, by devising a method for energizing the phase current of the motor, the flow of current other than the motor phase current at the neutral point of the motor is controlled. Thereby, the maximum current value of the phase current can be suppressed without increasing or decreasing the motor torque.
[0008]
In other words, in a motor inverter system where current other than the motor phase current flows in and out at the neutral point, the motor torque affects the phase current other than the zero phase current, and changing the zero phase current does not affect the torque. There is no. Therefore, by adjusting the zero-phase current, the magnitude of the phase current can be suppressed without affecting the motor torque.
[0009]
Also ,one In a motor drive current control device for controlling a motor drive current to be supplied to a coil of each phase of two motor coils, having two or more motor coils composed of coils of a plurality of phases whose ends are commonly connected to a neutral point, A power supply that holds these potential differences is connected between the neutral points of the two motor coils, and the phase current supplied to each phase coil of the two motor coils is the same as the phase current of one motor coil. Reduce the phase current of the portion where the amplitude is maximum, adjust the phase current of the other motor coil accordingly, and maintain the motor output torque at a predetermined value, while increasing the maximum value of the phase current amplitude. Suppression Can .
[0010]
In this way, in a system in which a power source is connected between the neutral points of two motor coils, the maximum current value of the phase current is suppressed without increasing or decreasing the motor torque by devising a method for energizing the phase current. be able to.
[0011]
In a system having two motor coils, a current that generates torque in a coordinated manner flows through the two motor coils. That is, for each phase current of the motor coil, the sum of the two motor coils affects the torque, but its distribution does not affect the output torque. Therefore, in the present invention, by changing this distribution, the maximum amplitude of the motor current is suppressed without affecting the motor output torque.
[0012]
Furthermore, it is preferable to adjust each phase current under the condition that the zero-phase current, which is a current other than the phase current entering and exiting the neutral point, is not changed.
[0013]
By devising the method of energizing the phase current, the maximum current value of the phase current can be suppressed without increasing the torque ripple, without increasing or decreasing the torque, and without changing the zero-phase current. As a result, it is possible to eliminate an adverse effect on the control of the zero-phase current.
[0014]
Further, a rotation speed detecting means for detecting the rotation speed of the motor is provided, and when the motor rotation speed is relatively high, control for suppressing the maximum amplitude value is not performed, and when the motor rotation speed is relatively low, the maximum amplitude value is set. It is preferable to perform switching so as to perform control for suppressing the above.
[0015]
Since the current value is low and the current frequency is high in the high rotation region, suitable current control can be performed by the conventional method, and the maximum amplitude value can be improved in the low rotation point region.
[0016]
Further, a rotational speed detection means for detecting the rotational speed of the motor is provided. When the motor rotational speed is relatively high, control for suppressing the maximum value of the amplitude is not performed. The maximum amplitude control is performed under the condition that the zero-phase current, which is a current other than the phase current entering and exiting the point, is not changed, and the control is performed so as to suppress the maximum amplitude when the motor speed is relatively low. Is preferred.
[0017]
In the middle rotation region, the maximum amplitude value need not be suppressed so much, so the ripple of the zero phase current is suppressed, and in the low rotation region, the ripple of the zero phase current is allowed and the maximum amplitude maximum value is suppressed. It can be carried out.
[0018]
Furthermore, it is preferable to adjust each phase current under the condition that the torque ripple is not increased. For example, by using a harmonic having a period three times that of the phase current, the change in the motor current becomes gentle and the occurrence of torque ripple can be reduced.
[0019]
Thus, according to the present invention, the maximum current value of the phase current can be suppressed without increasing or decreasing the torque, and the current capacity of the device can be reduced without impairing the function as a motor. For this reason, the cost reduction of a system is realizable, maintaining equivalent performance. Further, by suppressing the torque ripple, the function of the motor can be maintained sufficiently.
[0020]
Further, for current suppression, a high frequency component needs to be superimposed on the current. For this reason, it is necessary to control the current up to a high frequency range. However, more effective control is possible by switching the control depending on the rotation speed.
[0021]
That is, since the maximum amplitude value is suppressed in a low rotation region where the current value is large, the original control frequency band is low, so that control is not extremely difficult even if harmonics are superimposed. On the other hand, since the conventional method is used in the high rotation range, the above-described control problem at the time of high frequency superposition does not occur. Furthermore, appropriate control can be performed by suppressing the ripple of the zero-phase current in the middle rotation region.
[0022]
By such control switching, effective current suppression can be realized while avoiding control problems due to current suppression.
[0023]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, embodiments of the present invention will be described with reference to the drawings.
[0024]
"System configuration (2YDC)"
FIG. 1 shows a 2YDC voltage variable inverter having two motor coils and a battery B arranged between neutral points of these motor coils. One end of the capacitor C is connected to a constant voltage power source (for example, ground). Then, both ends of the capacitor C are connected to the inverter INV1 and the inverter INV2, respectively. That is, the output of the capacitor C is input to the inverters INV1 and INV2 as a power source.
[0025]
The inverter INV1 has three-phase outputs U1, V1, and W1, and the three-phase coils U, V, and W of the motor coil M1 are connected to the inverter INV1, respectively. The inverter INV2 has three-phase outputs U2, V2, and W2, to which the three-phase coils U, V, and W of the motor coil M2 are connected, respectively.
[0026]
Here, although the motor coils M1 and M2 are shown separately, they are coils of one motor, and are usually arranged to be different from each other by a predetermined angle to the motor, and currents having phases different from each other by the predetermined angle are supplied. The As a result, both currents supplied to both motor coils M1, M2 function as motor drive currents.
[0027]
Each phase motor coil of the motor coils M1 and M2 is commonly connected at a neutral point, and the neutral points of the motor coils M1 and M2 are connected via a battery B. In this example, the positive electrode of the battery B is connected to the neutral point of the motor coil M1, and the negative electrode of the battery B is connected to the neutral point of the motor coil M2.
[0028]
Although not shown, the inverters INV1 and INV2 are each composed of two switching transistors connected in series between the first power source p and the second power source m (the first power source p is grounded in the illustrated example). There are three arms, and the midpoint of these arms is connected to the end of each phase coil.
[0029]
Therefore, by controlling on / off of the switching transistors in the inverters INV1 and INV2, a desired current can be supplied from the capacitor C to the motor coils M1 and M2 to drive them. Further, by adding a difference in the length of the ON period of the upper transistor and the ON period of the lower transistor in the inverters INV1 and INV2, other than the motor driving phase current entering and exiting from the neutral point in the motor coils M1 and M2 The current (zero phase current) is controlled.
[0030]
Here, in the present embodiment, the inverters INV1 and INV2 are driven using the voltage (output voltage) Vc across one capacitor C as a power source. And the both-ends voltage (output voltage) E of the battery B does not fluctuate fundamentally. Therefore, by controlling the zero-phase current, the midpoint potential of the motor coils M1 and M2 can be arbitrarily set while maintaining a difference corresponding to the voltage of the battery B.
[0031]
As shown in FIG. 1, the voltage of the first power source p is Vp, the voltage of the second power source m is Vm, the output current of the capacitor C is ic, and the voltage across the capacitor C is Vc (= | Vm−Vp |). The current from the first power source p of the inverter INV1 is ip1, the current from the second power source m of the inverter INV1 is im1, the current from the first power source p of the inverter INV2 is ip2, and the current from the second power source m of the inverter INV2 Is im2. For the motor coil M1, the u-phase current iu1, the v-phase current iv1, the w-phase current iw1, the u-phase end voltage Vu1, the v-phase end voltage Vv1, the w-phase end voltage Vw1, and the motor coil M2 for the u-phase current iu2. The v-phase current iv2, the w-phase current iw2, the u-phase end voltage Vu2, the v-phase end voltage Vv2, and the w-phase end voltage Vw2. The neutral point voltage of the motor coil M1 is Vz1, the neutral point voltage of the motor coil M2 is Vz2, the battery B voltage is E, and the zero-phase current is ie.
[0032]
"Capacitor voltage control in 2YDC"
In particular, in the present system, the relationship between the neutral point potentials Vz1 and Vz2 of the motor coils M1 and M2 and the power supply voltage of the inverters INV1 and INV2, that is, the output voltage Vc of the capacitor C is such that the upper transistors in the inverters INV1 and INV2 The potential difference between the neutral points of the two motor coils M1 and M2 is determined by the ratio of the ON period of the side transistor, and is the voltage E (= | Vz1-Vz2 |) of the battery B. Therefore, the voltage across the capacitor C is determined by the ratio (modulation rate) of the ON periods of the upper transistors and the lower transistors of the inverters INV1 and INV2.
[0033]
The inverters INV1 and INV2 control the neutral point potentials Vz1 and Vz2 of the motor coils M1 and M2 by PWM control of the internal switching transistors. Here, the ratio (modulation rate) between the ON period of the upper transistor and the ON period of the lower transistor is the ratio of the amplitude of the voltage command value to one cycle of the carrier wave, which is a triangular wave, as shown in FIG. That is, when the voltage command value is increased, the period during which the triangular wave exceeds the command value is reduced accordingly. Then, by setting the period in which the triangular wave exceeds the command value as the ON period of the upper transistor of each phase and the OFF period of the lower transistor, the ratio of the ON period of the upper and lower transistors (that is, the modulation factor) is determined. FIG. 2 (a) shows the modulation factor d1 of the inverter INV1, and FIG. 2 (b) shows the modulation factor d2 of the inverter INV2.
[0034]
Thus, the neutral point potential is determined by the modulation rate, and the ratio between the neutral point potential and the capacitor voltage is determined by the modulation rate. Further, the potential difference between the two neutral point potentials is the voltage E of the battery B. Therefore, there is the following relationship between the modulation factor and the capacitor voltage Vc.
[0035]
Vc = E / (d1-d2)
Therefore, the capacitor voltage Vc can be determined by controlling the modulation rate of both inverters INV1, INV2.
[0036]
In the above example, the switching transistor is turned on / off without dead time with respect to the carrier wave period Ts of the inverter. That is, when the duty ratio is 50%, both the upper and lower transistors are turned on for a period of 50%. However, in order to completely eliminate the through current during the switching period, a dead time Td for turning off both the upper and lower transistors is often provided. In this case, the above equation is rewritten and applied as follows.
[0037]
Vc = E / {(d1-Td / Ts)-(d2 + Td / Ts)}
As described above, even when the dead time is provided, the capacitor voltage Vc can be determined by controlling the modulation factors d1 and d2.
[0038]
"The invention's effect"
Before describing a specific example of control according to the embodiment, the relationship between the motor output and the phase current is shown by simulation, and the current reduction effect of the present invention is shown.
[0039]
This simulation was performed according to the following procedure. First, the phase current iu1 of one phase (here, u phase) is divided into an average value (DC component) idc and one component (AC component) iac in one rotation. Further, for the AC component iac, a function g (θ) normalized by the amplitude Iac is introduced.
[0040]
That is,
[Expression 1]

And
[0041]
Subsequently, the voltage Vw is defined from the relationship between the battery voltage E and the capacitor voltage Vc of this system. This is because the phase current amplitude Iac has a maximum value obtained by subtracting the battery voltage E from the capacitor voltage Vc. Further, it is assumed that the voltage vv applied to each coil at the same time changes with the above-described current iac with a constant phase difference (power factor cos φ).
[0042]
That is,
[Expression 2]
Vw = Vc-E (6)
vv = Vwg (θ + φ) (7)
And
[0043]
Further, since there are six coils, the relationship between the motor output Wo and the work performed by each coil can be arranged as follows.
[0044]
[Equation 3]

The motor output can be approximated to Wo = ieE, assuming that the loss is sufficiently small. From this relationship, the following equation is obtained.
[0045]
[Expression 4]
ie = Wo / E (13)
From the above, the current flowing through each phase coil is obtained by the following equation using Iac, ie obtained by equations (12) and (13). However, the ripple of ie is not taken into consideration.
[0046]
[Equation 5]

Next, conditions used for the analysis are shown. The battery voltage E = 42V or 105V, the capacitor voltage Vc = 210V (step-up rate Vc / E = 5 or 2), the power factor cos θ = 0.8, and the energization of the maximum value of the AC current amplitude with respect to the motor output Wo The difference by the method is shown.
[0047]
The results are shown in FIGS. These figures show the difference in the maximum phase current value due to the difference in step-up rate. The horizontal axis shows the motor output, the vertical axis shows the maximum phase current value (imax), the solid line shows the maximum phase current value, and the broken line shows the maximum phase current value. The DC component (ie / 3) of the values is shown.
[0048]
3 shows a conventional maximum phase current value during energization, FIG. 4 shows a maximum phase current value during maximum suppression energization under zero-phase ripple non-permissible conditions, and FIG. 5 shows a maximum suppression energization method under zero-phase ripple permissible conditions (4 The maximum value of the phase current at the time of Section 2.2) is shown.
[0049]
From these figures, the following can be understood.
[0050]
In any case, the magnitude of the phase current varies greatly depending on the boosting rate, and the higher the boosting rate, the larger the proportion of the DC component in the phase current.
[0051]
・ Also, the effect of suppressing the magnitude of the phase current due to the difference in energization method can be confirmed.
[0052]
When comparing the maximum value of the phase voltage (AC component, DC component) at Wo = 40 kW and a step-up ratio of 5 times, the maximum value 477A (159, 317A) in the conventional energization of FIG. 3 and 454A (136, 136) in FIG. 317A) and 402A (85, 317A) in FIG.
[0053]
"Description of the conventional energization method as the basis of the present invention"
A conventional energization method for the 2YDC variable inverter shown in FIG. 3 will be described. The phase currents iu1, iv1, iw1, iu2, iv2, and iw2 that normally flow through the 2YDC variable inverter shown in FIG. 1 are the zero-phase current ie, the alternating current amplitude A, the rotor rotation speed, and the rotation angle ω, If θ (θ = ωt), it is expressed by the following equation.
[0054]
[Formula 6]

Here, assuming that A = 1 (A) and ie = 3 (A), equations (15) to (20) have the relationship shown in FIG. However, in FIG. 6, the current of each of iu1 and iu2 is the first stage, the sum of the currents of iu1 and iu2 is the second stage, and the d-axis currents id1 and id2 after the phase current is dq-axis converted are the third stage. The q-axis currents iq1 and iq2 are the fourth stage, and the last stage is one third (one phase) of the zero-phase current ie.
[0055]
Here, if reluctance torque is not considered, iq1 + iq2 contributes to the motor torque (in this case, the magnet position is not included in the analysis, so iu1 + iu2 can also be said to be a current component that contributes to the motor torque), ie is the battery And the current flowing between the capacitors. The maximum value of the phase current at this time is 2.00 (A). In the relationship of FIG. 6, the condition for the current for generating the motor driving torque and the current between the battery and the capacitor can be expressed by Equation (21).
[0056]
[Expression 7]

Furthermore, by introducing equations (22) and (23), equation (21) can be rewritten as follows. In equation (24), id and iq are current components represented by the dq axis, and are constant here.
[0057]
[Equation 8]

When there is a phase difference between the windings of the motor (when the coil position of one star connection and the coil position of another star connection are deviated by an angle ξ), the energized current is expressed by equations (25) to (25) (30) is rewritten, and equation (24) becomes equation (31).
[0058]
[Equation 9]

When A = 1 (A), ie = 3 (A), and ξ = 30 °, the equations (25) to (30) have the relationship shown in FIG. Thus, it can be seen that there is the same relationship as in FIG. 6 even when the phase difference of the coil is taken into consideration.
[0059]
“Description of 2YDC Variable Type Inverter of Embodiment”
In the embodiment of FIG. 4, the maximum value of the phase current is suppressed without allowing the generation of ripples in the zero-phase current.
[0060]
That is, in this embodiment, the maximum amplitude is suppressed by adding a predetermined function to the phase currents iu1, iv1, and iw1 in the 2YDC variable inverter of FIG. Then, by subtracting the added function from the phase currents iu2, iv2, and iw2, the maximum amplitude of the current is suppressed without changing the output torque of the motor. Further, in the present embodiment, zero-phase current ripple is not allowed.
[0061]
In order to reduce the current amplitude without changing the motor output torque and the magnitude of the zero-phase current, the phase currents iu1, iv1, iw1, iu2, iv2, and iw2 need to satisfy the relationship of Expression (21). is there. That is, it is necessary to satisfy the following expression (32). In this equation, the sum of the currents of the corresponding phases of each star connection is a sine wave, and the sum of the currents of each phase in each star connection changes the zero-phase current value or the sign of the zero-phase current value. Is equal to
[0062]
[Expression 10]

Here, the rank of the matrix on the left side of Equation (32) is 4, and two free parameters fu (θ) and fv (θ) are introduced, and the following sufficient condition is satisfied so as to satisfy Equation (32): Can be rewritten.
[0063]
[Expression 11]

Here, fu (θ), fv (θ), and fw (θ) are parameters (degree of freedom is 2) that can be used for design.
[0064]
Therefore, by giving fu (θ), fv (θ), fw (θ) (degrees of freedom of 2) satisfying the equations (33) to (42), the phase without changing the output torque and the zero-phase current. The currents iu1, iv1, iw1, iu2, iv2, and iw2 can be varied. Then, fu (θ), fv (θ), and fw (θ) are selected so as to reduce the maximum amplitude of the phase currents iu1, iv1, iw1, iu2, iv2, and iw2, thereby achieving the intended purpose. be able to.
[0065]
In the embodiment of FIG. 5, the conditions are relaxed, the generation of ripples in the zero-phase current is allowed, and the maximum value of the phase current is suppressed. In this case, the condition of Expression (39) can be removed. Accordingly, the degree of freedom in selecting fu (θ), fv (θ), and fw (θ) is increased. And it becomes possible to make the maximum value of a phase current smaller.
[0066]
In the above description, it is assumed that there is no phase difference between the two motor coils M1 and M2. In practice, the coils are often arranged with a phase difference between them. In this case, by giving a phase difference corresponding to the coil current, the influence of having the phase difference is eliminated.
[0067]
When there is a phase difference ξ between the coils of each star connection, Equation (31) is a condition that changes to Equation (21). That is, in order to reduce the current amplitude without changing the motor generation torque and the magnitude of the zero-phase current, the phase currents iu1, iv1, iw1, iu2, iv2, and iw2 need to satisfy Expression (43). . In this equation, the sum of the dq-axis currents corresponding to each star connection is constant, and the sum of the currents of each phase in each star connection changes the value of the zero-phase current or the value of the zero-phase current. Means equal to things.
[0068]
[Expression 12]

Here, as one of the solutions satisfying the equation (43), the following result is derived as in the case described above.
[0069]
[Formula 13]

Here, fu (θ), fv (θ), fw (θ), hu (θ), hv (θ), and hw (θ) are parameters that can be used for the design. Further, when ξ = 0 °, the functions of the equations (33) and (42) satisfy the equations (44) and (59).
[0070]
Then, by giving fu (θ), fv (θ), fw (θ), hu (θ), hv (θ), hw (θ) satisfying the equations (44) to (59), the output torque and zero The phase currents iu1, iv1, iw1, iu2, iv2, and iw2 can be changed without changing the phase current. Further, fu (θ), fv (θ), fw (θ), hu (θ), hv (θ), hw (θ) are suppressed to the maximum value of the phase currents iu1, iv1, iw1, iu2, iv2, iw2. By making it possible, the intended purpose can be achieved.
[0071]
If the condition is relaxed and the ripple current is allowed for the zero-phase current, fu (θ) + fv (θ) + fw (θ) + hu (θ) + hv (θ) + hw ( θ) = 0 is a condition.
[0072]
"Specific example when zero-phase current does not allow ripple"
The energization method satisfying the above condition with the inter-coil phase difference ξ = 0 ° can be obtained by setting fu (θ), fv (θ), and fw (θ) as shown in equations (60) to (62). can get. In addition, g1 of a formula is a constant put in order to satisfy the conditions of formulas (40) to (42), and in this case, g1 = 0.867.
[0073]
[Expression 14]

In the case of A = 1 (A) and ie = 3 (A), the waveform of fu (θ) is compared with iu1 and shown in FIGS. FIG. 8 is a diagram in which the scale of the vertical axis of the diagram of fu (θ) and the diagram of iu1 are combined, and FIG. 9 is an enlarged view for easy viewing of the waveform of fu (θ). From the figure, the waveform of fu (θ) is a waveform in which the peak portion of the sine wave is cut out with a width of 60 degrees, arranged in the order of positive side, negative side, and positive side, and the negative side is set to double the positive side. It has become. That is, it has a waveform that most suppresses the maximum peak of iu1.
[0074]
Therefore, by adding fu (θ), fv (θ), fw (θ) of such a form to the sine curve, the phase current can be suppressed to the maximum current, and the change in the output torque based on this can be Absent. Furthermore, in this example, the condition that no zero-phase current is generated is also satisfied.
[0075]
Furthermore, the result using the conditions of Formula (60)-(62) is shown in FIG. The figure shows the following.
[0076]
・ Zero phase current
The zero-phase current is ie = 3 (A) and does not include a ripple component.
[0077]
·torque
The current (iu1 + iu2) for generating the motor torque is the same as that in FIG. 6, and the torque is generated as intended.
[0078]
-Phase current magnitude
The maximum value of the magnitude of the phase current is 1.866 (A). The breakdown of the size is 0.866A for the AC component and 1A for the DC component.
[0079]
Thus, by using fu (θ), fv (θ), and fw (θ) shown in equations (60) to (62), the phase current can be obtained without affecting the zero-phase current and the motor output torque. The maximum value of can be suppressed.
[0080]
In addition, as another example in the case where ξ = 0 ° and no ripple is allowed in the zero-phase current, a case where the AC amplitude is modulated by a triple harmonic is shown.
[0081]
That is, in the case of A = 1 (A) and ie = 3 (A), the waveform of fu (θ) is compared with iu1, and is shown in FIG. This waveform is a sine wave having a frequency that is three times the amplitude of the original AC waveform, and is modulated by the following equation.
[0082]
[Expression 15]

Furthermore, equations (63) and (64) can be organized as follows.
[0083]
[Expression 16]

Here, if fu (θ) = αsin (3θ) Asin (θ), the conditions of the equations (33) to (42) are satisfied. That is, by setting fu (θ) = αsin (3θ) Asin (θ), the following result is obtained.
[0084]
FIG. 11 shows this fu (θ). Furthermore, the result using this fu (θ) is shown in FIG. The figure shows the following.
[0085]
・ Zero phase current
The average value of the zero-phase current is ie = 3 (A). Its magnitude is three times the amplitude of added fu (θ).
[0086]
·torque
The current (iu1 + iu2) for generating the motor torque is equivalent to that in FIG. 6, and the torque is generated as intended.
[0087]
-Phase current magnitude
The maximum value of the phase current is 1.872 (A). The breakdown of the size is 0.872A for the AC component and 1A for the DC component.
[0088]
Next, FIG. 13 shows the result when the phase difference is shifted by 30 ° in the coil position between the star connections (ξ = 30 °). This figure shows the following.
[0089]
・ Zero phase current
The zero-phase current is ie = 3 (A) and does not include a ripple component.
[0090]
·torque
The currents (id and iq) for generating the motor torque are the same as those in FIG. 6, and the torque is generated as intended.
[0091]
-Phase current magnitude
The maximum value of the magnitude of the phase current is 1.866 (A). The breakdown of the size is 0.866A for the AC component and 1A for the DC component.
[0092]
・ Phase current waveform
The command value used here is a waveform that changes sharply in order to suppress the magnitude of the current. However, in the actual case, this is realized by filtering to remove high frequency components. In this case, however, the current suppressing effect is slightly worse.
[0093]
"Specific example of allowing zero-phase current ripple"
One of the energization methods that can suppress the magnitude of the phase current under the condition that ξ = 0 ° and the zero-phase current ripple is allowed is that fu (θ), fv (θ), and fw (θ) are expressed by the equations (67) to (69). In addition, g2 of a formula is a constant put in order to satisfy the conditions of formulas (40) to (42), and in this case, g2 = −0.637.
[0094]
[Expression 17]

In the case of A = 1 (A) and ie = 3 (A), the waveform of fu (θ) is compared with iu1, and is shown in FIG. Furthermore, the result using the conditions of Formula (67)-(69) is shown in FIG. The figure shows the following.
[0095]
・ Zero phase current
The average value of the zero-phase current is ie = 3 (A), but the ripple component is included and the magnitude thereof is 0.46A.
[0096]
·torque
The current (iu1 + iu2) for generating the motor torque is the same as that in FIG. 6, and the torque is generated as intended.
[0097]
-Phase current magnitude
The maximum value of the phase current is 1.63 (A). The breakdown of the size is 0.63A for the AC component and 1A for the DC component.
[0098]
Next, as an example of another method for allowing ripples in the zero-phase current at ξ = 0 °, a case where a 6-fold higher harmonic is added is shown. In the case of A = 1 (A) and ie = 3 (A), the waveform of fu (θ) is compared with iu1, and is shown in FIG. For the amplitude of the harmonic, a value optimized so as to minimize the phase current is used.
[0099]
Furthermore, the result using this fu (θ) is shown in FIG. The figure shows the following.
[0100]
・ Zero phase current
The average value of the zero-phase current is ie = 3 (A), but includes a ripple component. Its magnitude is three times the amplitude of added fu (θ).
[0101]
·torque
The current (iu1 + iu2) for generating the motor torque is the same as that in FIG. 6, and the torque is generated as intended.
[0102]
-Phase current magnitude
The maximum value of the phase current is 1.96 (A). The breakdown of the size is 0.96A for the AC component and 1A for the DC component.
[0103]
"Energization with 1YDC voltage variable inverter"
The reduction of the maximum amplitude value in a 1YDC variable voltage inverter having one motor coil and a battery connected to the neutral point will be described.
[0104]
FIG. 18 shows a configuration example of a system using a 1YDC voltage variable inverter. Thus, basically, the inverter and the motor are combined into one in the 2YDC type system of FIG. The capacitor C is supplied as a power source to the inverter INV1, and the three-phase output of the inverter INV1 is connected to the U, V, and W phase coil ends of the motor coil M1. The battery B has a negative electrode connected to one end of the capacitor and a positive electrode connected to the neutral point of the motor coil M1.
[0105]
Therefore, according to this configuration, the input voltage Vc of the inverter INV1 with respect to the battery voltage E is determined by Vc = E / d by the modulation factor d of the inverter INV1.
[0106]
"Concrete example"
In this system, by controlling the zero phase current, the maximum value of the amplitude of the phase current is suppressed without changing the output torque. That is, an example of a current reduction method for the 1YDC voltage variable inverter will be described. Here, a harmonic having a period three times the rotation period is added to the zero-phase current.
[0107]
19 and 20 show a comparison between the case where A = 1 (A) and ie = 3 (A) and the case where three times higher harmonics are not added (conventional) and the case where they are added. For the amplitude of the harmonic, a value optimized so as to minimize the phase current is used. The figure shows the following.
[0108]
・ Zero phase current
The average value of the zero-phase current is ie = 3 (A), but includes a ripple component.
[0109]
·torque
The currents (iq1 and id1) for generating the motor torque are equivalent to those in FIGS. 19 and 20, and generate the torque as intended.
[0110]
-Phase current magnitude
The maximum value of the phase current is 1.89 (A), and the maximum value of 2A in FIG. 18 is reduced by 0.11 A. The breakdown of the size is 0.89A for the AC component and 1A for the DC component.
[0111]
Thus, by adding a high frequency having a period three times the rotation period to the zero-phase current, the maximum value of the amplitude of the phase current can be suppressed without affecting the output torque.
[0112]
Here, a high frequency having a period that is three times the rotation period is added, but it may have a period that is six times as described above or another function. That is, any function can be used as long as it can maintain the condition that the motor torque is not affected and the value decreases in synchronization with the maximum amplitude of the phase current.
[0113]
【The invention's effect】
As described above, according to the present invention, the maximum current value of the phase current can be suppressed without increasing or decreasing the torque, and the current capacity of the device can be reduced without impairing the function as a motor. , It is possible to reduce the cost of the system while maintaining equivalent performance. By suppressing the torque ripple, the function of the motor can be made sufficient.
[0114]
Further, for current suppression, a high frequency component needs to be superimposed on the current. For this reason, it is necessary to control the current up to a high frequency range. However, more effective control is possible by switching the control depending on the rotation speed.
[0115]
That is, since the maximum amplitude value is suppressed in a low rotation region where the current value is large, the original control frequency band is low, so that control is not extremely difficult even if harmonics are superimposed. On the other hand, since the conventional method is used in the high rotation range, the above-described control problem at the time of high frequency superposition does not occur. Furthermore, appropriate control can be performed by suppressing the ripple of the zero-phase current in the middle rotation region.
[0116]
By such control switching, current suppression can be realized while avoiding control problems due to current suppression.
[Brief description of the drawings]
FIG. 1 is a diagram illustrating a device configuration of 2YDC.
FIG. 2 is a diagram showing a relationship between a voltage command value and an inverter carrier wave.
FIG. 3 is a diagram showing a maximum current amplitude value in a conventional energization method.
FIG. 4 is a diagram showing a current maximum amplitude value in the case of current reduction when zero-phase ripple is not allowed;
FIG. 5 is a diagram showing a current maximum amplitude value in the case of current reduction when zero-phase ripple is allowed.
FIG. 6 is a diagram illustrating a conventional phase current, a zero-phase current, and the like.
FIG. 7 is a diagram showing a conventional phase current and a zero-phase current (when there is a phase difference between coils).
FIG. 8 is a diagram showing a phase current and a function f when suppressing a ripple current.
FIG. 9 is an enlarged view of FIG.
FIG. 10 is a diagram showing a phase current and the like when suppressing a ripple current.
FIG. 11 is a diagram showing a phase current and its maximum amplitude value when the AC amplitude is modulated with a triple harmonic.
FIG. 12 is a diagram showing a phase current and the like in the case of modulating the AC amplitude with a triple harmonic.
FIG. 13 is a diagram illustrating a phase current and the like when ripples are suppressed (with phase difference).
FIG. 14 is a diagram showing a phase current and a function f when a ripple current is allowed.
FIG. 15 is a diagram showing a phase current and the like when a ripple current is allowed.
FIG. 16 is a diagram showing a phase current and a maximum amplitude value in the case of modulation with a harmonic of 6 times.
FIG. 17 is a diagram showing a phase current and the like in the case of modulating an AC amplitude with a 6-fold higher harmonic wave.
FIG. 18 is a diagram illustrating a device configuration of 1YDC.
FIG. 19 is a diagram showing phase currents and the like in the conventional method of 1YDC.
FIG. 20 is a diagram showing a phase current and the like in the case where the AC amplitude is modulated with a triple harmonic in 1YDC.
[Explanation of symbols]
B battery, C capacitor, INV1, INV2 inverter, M1, M2 motor coil.

Claims (5)

In a motor drive current control device for controlling a motor drive current supplied to a coil of a plurality of phases, one end of which is commonly connected to a neutral point,
A power source that holds a neutral point potential is connected to the neutral point,
By controlling the current supplied to each phase coil, the zero phase current, which is a current other than the phase current entering and exiting the neutral point, is varied, thereby maintaining the output torque of the motor at a predetermined level. Motor drive current control device which suppresses the maximum value of the amplitude of the motor. The apparatus of claim 1, comprising:
further,
A rotation speed detection means for detecting the rotation speed of the motor is provided. When the motor rotation speed is relatively high, the control to suppress the maximum amplitude value is not performed, and the maximum amplitude value is suppressed when the motor rotation speed is relatively low. A motor drive current control device that switches to perform control. In a motor drive current control device for controlling a motor drive current to be supplied to a coil of each phase of two motor coils having two or more motor coils composed of a plurality of phase coils commonly connected at one end to a neutral point,
A power source that holds these potential differences is connected between the neutral points of the two motor coils, and the phase current supplied to each phase coil of the two motor coils is equal to the phase current of one motor coil. Reduce the phase current of the part where the amplitude is maximum, adjust the phase current of the other motor coil correspondingly, and maintain the motor output torque at a predetermined value, while increasing the maximum value of the phase current amplitude. While controlling and adjusting each phase current under the condition that the zero phase current which is a current other than the phase current entering and exiting the neutral point is not changed ,
Furthermore, a rotational speed detection means for detecting the rotational speed of the motor is provided,
When the motor speed is relatively high, control to suppress the maximum value of the amplitude is not performed, and when the motor speed is medium, the zero-phase current, which is a current other than the phase current entering and exiting the neutral point in the previous period, fluctuates. A motor drive current control device that performs control so that the maximum amplitude value is controlled under the condition that the maximum amplitude value is not controlled, and the maximum amplitude value is controlled when the motor rotation speed is relatively low . The device according to any one of claims 1 to 3,
further,
A motor drive current control device that adjusts each phase current under the condition that torque ripple is not increased . In a motor drive current control method for controlling a motor drive current to be supplied to a plurality of coils whose one end is commonly connected to a neutral point,
A power source that holds a neutral point potential is connected to the neutral point, and a zero phase that is a current other than the phase current that enters and exits the neutral point by controlling the current supplied to each phase coil. A motor drive current control method that suppresses the maximum value of the amplitude of the phase current while maintaining the output torque of the motor at a predetermined value by changing the current .

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