JP3851208B2 - Inverter integrated drive - Google Patents

Description

とする。
【００４７】
つづいて、本システムの電池電圧Ｅとコンデンサ電圧Ｖｃの関係から電圧Ｖｗを定義する。これは、相電流の振幅Ｉａｃは、コンデンサ電圧Ｖｃから電池電圧Ｅを減算した電圧が最大値になるからである。また、同時に各コイルに印加される電圧ｖｖは上述の電流ｉａｃと一定の位相差（力率ｃｏｓφ）で推移すると仮定する。
【００４８】
すなわち、
【数２】
Ｖｗ＝Ｖｃ−Ｅ （６）
ｖｖ＝Ｖｗｇ（θ＋φ） （７）
とする。
【００４９】
また、モータ出力Ｗｏと各コイルがする仕事との関係は、コイルが６本あるので、次式のように整理できる。
【００５０】
【数３】

また、モータ出力は損失が十分に小さいとしてＷｏ＝ｉｅＥと近似できる。この関係より次式を得る。
【００５１】
【数４】
ｉｅ＝Ｗｏ／Ｅ （１３）
以上より、各相コイルを流れる電流は式（１２）、（１３）で求められるＩａｃ，ｉｅを利用し次式で求められる。ただし、ｉｅのリップル分は考慮していない。
【００５２】
【数５】

次に、解析に用いる条件を示す。電池電圧Ｅ＝４２Ｖ または１０５Ｖ、コンデンサ電圧Ｖｃ＝２１０Ｖ（昇圧率Ｖｃ／Ｅ＝５，または２）、力率ｃｏｓθ＝０．８で、モータ出力Ｗｏに対する交流電流振幅の大きさの最大値の通電方法による違いを示す。
【００５３】
この結果を、図６〜図８に示す。これらの図は、昇圧率の違いによる相電流最大値の違いを示しており、横軸がモータ出力、縦軸が相電流最大値（ｉｍａｘ）、実線が相電流最大値、波線が相電流最大値のうちの直流成分（ｉｅ／３）を示している。
【００５４】
図６は従来の通電時の相電流最大値、図７は零相リップル非許容条件での最大抑制通電時の相電流最大値、図８は零相リップル許容条件での最大抑制通電方法（４．２．２ 節）時の、相電流最大値を示している。
【００５５】
これらの図より以下のことがわかる。
【００５６】
・いずれの場合にも、相電流の大きさは昇圧率により大きく変化し、昇圧率が高いほうが相電流に占める直流成分の比率が大きい。
【００５７】
・また、通電法の違いによる相電流の大きさの抑制効果が確認できる。
【００５８】
・Ｗｏ＝４０ｋＷ、昇圧比５倍で相電圧の最大値（交流成分、直流成分）を比較すると、図６の従来通電では、最大値４７７Ａ（１５９，３１７Ａ）、図７では、４５４Ａ（１３６，３１７Ａ）、図８では、４０２Ａ（８５，３１７Ａ）である。
【００５９】
「本発明の基本となる従来の通電方法の説明」
図６に示す２ＹＤＣ可変型インバータの従来の通電方法について、説明する。図３に示す２ＹＤＣ可変型インバータに、通常流される相電流ｉｕ１，ｉｖ１，ｉｗ１，ｉｕ２，ｉｖ２，ｉｗ２は、零相電流をｉｅ、交流電流振幅をＡ、ロータ回転数、回転角をそれぞれω，θ（θ＝ωｔ）とすれば、次式で表される。
【００６０】
【数６】

ここで、Ａ＝１（Ａ），ｉｅ＝３（Ａ）とすると、式（１５）〜（２０）は、図９の関係が有る。ただし、図９は上段から、ｉｕ１，ｉｕ２各々の電流が１段目、ｉｕ１とｉｕとの電流の和が２段目、相電流をｄｑ軸変換した後のｄ軸電流ｉｄ１，ｉｄ２が３段目、ｑ軸電流ｉｑ１，ｉｑ２が４段目、最終段が零相電流ｉｅの３分の１（１相分）を示してある。
【００６１】
ここで、リラクタンストルクを考えなければｉｑ１＋ｉｑ２がモータトルクに寄与する電流成分（今回のケースでは、磁石位置を解析に入れていないので、ｉｕ１＋ｉｕ２がモータトルクに寄与する電流成分ともいえる）、ｉｅが電池とコンデンサ間を流れる電流である。そして、この時の相電流の大きさの最大値は２．００（Ａ）である。図９の関係のうち、モータ駆動トルクを発生するための電流と電池・コンデンサ間の電流の条件は、式（２１）で書くことができる。
【００６２】
【数７】

さらに、式（２２）、（２３）を導入することにより、式（２１）は以下のようにも書きかえられる。なお、式（２４）において、ｉｄ，ｉｑは、ｄｑ軸で表される電流成分で、ここではコンスタントとなる。
【００６３】
【数８】

モータのコイル間に位相差がある場合（あるスター結線のコイル位置と、他のスター結線のコイル位置とが角度ξでずれている場合）には、通電される電流は式（２５）〜（３０）の様に書きかえられ、式（２４）は式（３１）となる。
【００６４】
【数９】

Ａ＝１（Ａ）、ｉｅ＝３（Ａ）、ξ＝３０°とすると、式（２５）〜（３０）は、図１０の関係が有る。このように、コイルの位相差を考慮した場合にも図９と同様の関係があることが分かる。
【００６５】
「実施形態の２ＹＤＣ可変型インバータの説明」
図７の実施形態では、零相電流におけるリップルの発生を許容せずに相電流の最大値を抑制する。
【００６６】
すなわち、本実施形態では、図３の２ＹＤＣ可変型インバータにおいて、相電流ｉｕ１，ｉｖ１，ｉｗ１に対し所定の関数を加算することで、最大振幅を抑制する。そして、加算した関数を相電流ｉｕ２，ｉｖ２，ｉｗ２から減算することで、モータの出力トルクを変動させることなく、電流の最大振幅を抑制する。また、本実施形態では、零相電流のリップルを許容しない。
【００６７】
モータ出力トルクおよび零相電流の大きさを変えずに、電流振幅を減少させるためには、相電流ｉｕ１，ｉｖ１，ｉｗ１，ｉｕ２，ｉｖ２，ｉｗ２は、式（２１）の関係を満足する必要がある。すなわち、次式（３２）を満足する必要がある。この式は、各スター結線の対応する相の電流の和が正弦波であり、かつ各スター結線内の各相の電流の総和が零相電流の値、若しくは零相電流の値の符号を変えたものに等しいことを意味している。
【００６８】
【数１０】

ここで式（３２）の左辺の行列のランクが４だから、２つのフリーパラメータｆｕ（θ），ｆｖ（θ）を導入し、式（３２）を満足するように、以下のような十分条件に書きかえることができる。
【００６９】
【数１１】

ここで、ｆｕ（θ），ｆｖ（θ），ｆｗ（θ）が、設計に利用できるパラメータ（自由度は２）である。
【００７０】
従って、式（３３）〜（４２）を満たすｆｕ（θ），ｆｖ（θ），ｆｗ（θ）（自由度は２）を与えることで、出力トルクおよび零相電流を変動させることなく、相電流ｉｕ１，ｉｖ１，ｉｗ１，ｉｕ２，ｉｖ２，ｉｗ２を変動させることができる。そして、ｆｕ（θ），ｆｖ（θ），ｆｗ（θ）を相電流ｉｕ１，ｉｖ１，ｉｗ１，ｉｕ２，ｉｖ２，ｉｗ２の最大振幅を減少するように選択することで、所期の目的を達成することができる。
【００７１】
図８の実施形態では、条件を緩和し、零相電流におけるリップルの発生を許容して、相電流の最大値を抑制する。この場合には、式（３９）の条件をはずすことができる。従って、ｆｕ（θ），ｆｖ（θ），ｆｗ（θ）を選択する場合の自由度が広がる。そして、相電流の最大値をより小さくすることが可能となる。
【００７２】
また、上述の説明では、２つのモータコイルＭ１，Ｍ２間に位相差がないことを前提とした。実際には、コイル間に位相差を持たせて配置する場合も多い。この場合には、コイル電流に対応した位相差を持たせることで、位相差を持たせたことの影響を排除する。
【００７３】
このような各スター結線のコイル間に位相差ξを持つ場合には、式（３１）が、式（２１）に変わる条件となる。すなわち、モータ発生トルクや零相電流の大きさを変えずに電流振幅を減少するためには、相電流ｉｕ１，ｉｖ１，ｉｗ１，ｉｕ２，ｉｖ２，ｉｗ２が、式（４３）を満足する必要がある。この式は、各スター結線の対応するｄｑ軸電流の和が一定であり、かつ各スター結線内の各相の電流の総和が零相電流の値、若しくは零相電流の値の符号を変えたものに等しいことを意味している。
【００７４】
【数１２】

ここで、式（４３）を満たす解の１つとして、前述の場合と同様に以下の結果が導かれる。
【００７５】
【数１３】

ここで、ｆｕ（θ），ｆｖ（θ），ｆｗ（θ），ｈｕ（θ），ｈｖ（θ），ｈｗ（θ）が、設計に利用できるパラメータである。さらに、ξ＝０°の時、式（３３）、（４２）の関数は、式（４４）、（５９）を満たす。
【００７６】
そして、式（４４）〜（５９）を満たすｆｕ（θ），ｆｖ（θ），ｆｗ（θ），ｈｕ（θ），ｈｖ（θ），ｈｗ（θ）を与えることで、出力トルクおよび零相電流を変動させることなく、相電流ｉｕ１，ｉｖ１，ｉｗ１，ｉｕ２，ｉｖ２，ｉｗ２を変動させることができる。さらに、ｆｕ（θ），ｆｖ（θ），ｆｗ（θ），ｈｕ（θ），ｈｖ（θ），ｈｗ（θ）を相電流ｉｕ１，ｉｖ１，ｉｗ１，ｉｕ２，ｉｖ２，ｉｗ２の最大値を抑制できる形にすることで、所期の目的を達成することができる。
【００７７】
また、条件を緩和し、零相電流にリップル電流を許せば、式（５０）、（５１）の条件に代り、ｆｕ＋ｆｖ＋ｆｗ＋ｈｕ＋ｈｖ＋ｈｗ＝０が条件となる。
【００７８】
「零相電流にリップルを許さない場合の具体例」
コイル間位相差ξ＝０°で、上述の条件を満足する通電方法は、ｆｕ（θ），ｆｖ（θ），ｆｗ（θ）を式（６０）〜（６２）のように設定することにより得られる。なお、式のｇ１は、式（４０）〜（４２）の条件を満たすために入れた定数で、この場合はｇ１＝０．８６７である。
【００７９】
【数１４】

Ａ＝１（Ａ），ｉｅ＝３（Ａ）の場合について、ｆｕ（θ）の波形をｉｕ１と比較し、図１１、図１２に示す。図１１は、ｆｕ（θ）の図とｉｕ１の図の縦軸のスケールをあわせたもの、図１２は、ｆｕ（θ）の波形を見やすくするために拡大したものである。図より、ｆｕ（θ）の波形は正弦波のピーク部分を６０度幅で切り出し、それを正側負側正側の順にならべ、負側の大きさを正側の２倍に設定した波形となっている。すなわち、ｉｕ１の最大ピークのところを最も抑制する波形となっている。
【００８０】
従って、このような形のｆｕ（θ），ｆｖ（θ），ｆｗ（θ）をサインカーブに加算することによって相電流を最大電流を抑制することができ、かつこれに基づく出力トルクの変化はない。さらに、この例では、零相電流を発生しないという条件も満たしている。
【００８１】
さらに、式（６０）〜（６２）の条件を用いた結果を図１３に示す。図より、以下のことがわかる。
【００８２】
・零相電流
零相電流はｉｅ＝３（Ａ）であり、リップル成分は含まれない。
【００８３】
・トルク
モータトルクを発生する電流（ｉｕ１＋ｉｕ２）は図９と同等で、意図どおりのトルクを発生している。
【００８４】
・相電流の大きさ
相電流の大きさの最大値は、１．８６６（Ａ）である。大きさの内訳は、交流による成分が０．８６６Ａ、直流による成分が１Ａである。
【００８５】
このように、式（６０）〜（６２）に示すｆｕ（θ），ｆｖ（θ），ｆｗ（θ）を利用することによって、零相電流、モータ出力トルクに影響を与えることなく、相電流の最大値を抑制することができる。
【００８６】
また、ξ＝０°で、零相電流にリップルを許さない場合の他の例として、３倍の高調波で交流振幅を変調する場合を示す。
【００８７】
すなわち、Ａ＝１（Ａ），ｉｅ＝３（Ａ）の場合について、ｆｕ（θ）の波形をｉｕ１と比較し、図１４に示す。この波形は、元々の交流波形の振幅を３倍の周波数を持つ正弦波で、次式のように変調した波形になっている。
【００８８】
【数１５】

さらに、式（６３）と（６４）はつぎのように整理できる。
【００８９】
【数１６】

ここで、ｆｕ（θ）＝αｓｉｎ（３θ）Ａｓｉｎ（θ）とおけば、式（３３）〜（４２）の条件を満足する。すなわち、ｆｕ（θ）＝αｓｉｎ（３θ）Ａｓｉｎ（θ）に設定することで、下記のような結果が得られる。
【００９０】
図１４は、このｆｕ（θ）を示したものである。さらに、このｆｕ（θ）を用いた結果を図１５に示す。図より、以下のことがわかる。
【００９１】
・零相電流
零相電流は平均値はｉｅ＝３（Ａ）である。その大きさは、加えたｆｕ（θ）の３倍の振幅である。
【００９２】
・トルク
モータトルクを発生する電流（ｉｕ１＋ｉｕ２）は図９と同等であり、意図どおりのトルクを発生している。
【００９３】
・相電流の大きさ
相電流の大きさの最大値は１．８７２（Ａ）である。大きさの内訳は、交流による成分が０．８７２Ａ、直流による成分が１Ａである。
【００９４】
次に、各スター結線間のコイル位置に位相差が３０°ずれた場合（ξ＝３０°）の結果を図１６に示す。この図より、以下のことがわかる。
【００９５】
・零相電流
零相電流は、ｉｅ＝３（Ａ）であり、リップル成分は含まれない。
【００９６】
・トルク
モータトルクを発生する電流（ｉｄとｉｑ）は図９と同等で、意図どおりのトルクを発生している。
【００９７】
・相電流の大きさ
相電流の大きさの最大値は、１．８６６（Ａ）である。大きさの内訳は、交流による成分が０．８６６Ａ、直流による成分が１Ａである。
【００９８】
・相電流の波形
ここで用いた指令値は、電流の大きさを抑制するために、急峻に変化する波形である。しかし、実際の場合には、これをフィルタリングし高周波成分を除くことにより実現する。ただし、その場合は若干電流の抑制効果は悪くなる。
【００９９】
「零相電流にリップルを許す場合の具体例」
ξ＝０°で、零相電流リップルを許す条件で、相電流の大きさを抑制できる通電方法の１つは、ｆｕ（θ）、ｆｖ（θ），ｆｗ（θ）を式（６７）〜（６９）のように流すことである。なお、式のｇ２は、式（４０）〜（４２）の条件を満たすために入れた定数で、この場合はｇ２＝−０．６３７である。
【０１００】
【数１７】

Ａ＝１（Ａ）、ｉｅ＝３（Ａ）の場合について、ｆｕ（θ）の波形をｉｕ１と比較し、図１７に示す。さらに、式（６７）〜（６９）の条件を用いた結果を図１８に示す。図より、以下のことがわかる。
【０１０１】
・零相電流
零相電流の平均値はｉｅ＝３（Ａ）であるが、リップル成分が含まれその大きさは０．４６Ａである。
【０１０２】
・トルク
モータトルクを発生する電流（ｉｕ１＋ｉｕ２）は図９と同等で、意図どおりのトルクを発生している。
【０１０３】
・相電流の大きさ
相電流の大きさの最大値は、１．６３（Ａ）である。大きさの内訳は、交流による成分が０．６３Ａ、直流による成分が１Ａである。
【０１０４】
次に、ξ＝０°で、零相電流にリップルを許す場合のその他の方法の一例として、６倍の高調波を加える場合を示す。Ａ＝１（Ａ）、ｉｅ＝３（Ａ）の場合について、ｆｕ（θ）の波形をｉｕ１と比較し、図１９に示す。なお高調波の振幅は、相電流が最小になるように最適化した値を用いている。
【０１０５】
さらに、このｆｕ（θ）を用いた結果を図２０に示す。図より、以下のことがわかる。
【０１０６】
・零相電流
零相電流は平均値はｉｅ＝３（Ａ）であるが、リップル成分が含まれる。その大きさは、加えたｆｕ（θ）の３倍の振幅である。
【０１０７】
・トルク
モータトルクを発生する電流（ｉｕ１＋ｉｕ２）は図９と同等で、意図どおりのトルクを発生している。
【０１０８】
・相電流の大きさ
相電流の大きさの最大値は１．９６（Ａ）である。大きさの内訳は、交流による成分が０．９６Ａ、直流による成分が１Ａである。
【０１０９】
このように、上記実施形態にによれば、トルクの増減を伴うことなく、相電流の最大電流値を抑制することができ、モータとしての機能を損なうことなく、デバイスの電流容量を下げられるので，同等性能を保ちつつ、システムの低コスト化を実現できる。トルクリップルを抑制することで、モータの機能を十分なものにできる。
【０１１０】
また、電流抑制には、高周波成分が電流に重畳する必要がある。このため、高周波域まで電流を制御することが必要になる。しかし、回転数により制御を切り替えることにより、より効果的な制御が可能になる。
【０１１１】
すなわち、電流値が大きい低回転域で振幅最大値抑制を行うため、元々の制御周波数帯域が低いので，高調波を重畳しても制御が極端に難しくはならない。一方、高回転域では従来法を用いるため前述の高周波重畳時の制御問題は発生しない。さらに、中回転領域において、零相電流のリップルを抑制することで、適切な制御が行える。
【０１１２】
このような制御の切り替えにより、電流抑制による制御上の問題を回避しつつ，電流抑制を実現できる。
【０１１３】
「一体型駆動装置」
上述のように、本実施形態によれば、２つの三相コイル２４、２６を有し、これによって１つのモータを構成するシステムが示されている。そして、このようなモータをインバータと一体形成することが好適である。
【０１１４】
図２１には、全体的な構成が示されている。筐体１００内の一方側には、インバータ１１０が配置される。例えば、このインバータ１１０は、２つのインバータ回路３０、３２を含んでおり、１つの基板１１２上に形成される。インバータ１１０は、複数パワートランジスタなどの素子１１４から構成され、このインバータ１１０と、ステータ（ステータコイル）１５０が配線１１６で接続されている。なお、図においては、配線１１６を１つだけ示したが、インバータ１１０の出力端のそれぞれがステータ１５０の各入力端に接続されている。また、仕切り１２０と基板１１２の間には、内部に冷却水路を含む冷却板１１８が配置されており、これによって素子１１４を冷却している。
【０１１５】
筐体１００の仕切り１２０には、その中心部に軸受け（ベアリング）１２２が設けられ、ここにシャフト１３０が軸支される。筐体１００のインバータ１１０の反対側には、シャフト１３０が貫通される位置に軸受け（ベアリング）１２４が設けられ、この軸受け１２４を介し、シャフト１３０が筐体１００から突出されている。
【０１１６】
シャフト１３０の筐体内の中間部には、ロータ１４０が配置され、このロータ１４０を取り囲むように、ステータ１５０が配置されている。このロータ１４０とステータ１５０は、図２１の右図に示すように、円筒形のロータ１４０を同心状に中空円筒状のステータ１５０が取り囲む形になっている。
【０１１７】
この例では、ステータ１５０は、周方向に４つの部分ステータｐ１，ｐ２，ｐ３，ｐ４に分割されており、４極対（ポール）のモータである。そして、この部分ステータｐ１，ｐ２，ｐ３，ｐ４は、それぞれが三相コイル２４、２６を含んでいる。従って、部分ステータｐ１において、２つの三相コイル２４、２６が設けられている。そして、部分ステータは４つあるため、２つの三相コイル２４、２６は、各相のコイルがそれぞれ直列接続され、全体として６相のコイルとなっている。
【０１１８】
そして、この２つの三相コイル２４、２６の６つのコイル端部にインバータ回路３０、３２の出力端がそれぞれ接続される。この接続は、筐体１００内で行われる。一方、直流電源４０は、外付けであり、筐体１００に設けられた一対の電池用端子（それぞれが三相コイル２４、２６の中性点に接続されている）に接続される。
【０１１９】
また、この例では、コンデンサ３８も外付けで形成されている。そこで、インバータ回路３０、３２の正極母線３４と、負極母線３６に接続される一対の端子が筐体１００に設けられ、ここに外付けのコンデンサ３８が接続される。なお、コンデンサ３８を、筐体１００内に収容することも好適である。これによって、外部配線をさらに減少することができる。
【０１２０】
また、すべての部分ステータｐ１〜ｐ４に、それぞれ三相コイル２４、２６の両方を配置する必要はない。例えば、部分ステータｐ１、ｐ３に三相コイル２４を配置し、部分ステータｐ２、ｐ４に三相コイル２６を配置してもよい。この場合、部分ステータｐ１、ｐ３の各相コイルがそれぞれ直列接続され、部分ステータＰ２、ｐ４の各相コイルがそれぞれ直列接続される。
【０１２１】
この構成であっても、三相コイル２４、２６の端部は６つであり、三相コイル２４の３つの端部にインバータ回路３０の３つの出力端が接続され、三相コイル２６の３つの出力端にインバータ回路３２の３つの出力端が接続される。
【０１２２】
さらに、図２２には、他の例が示されている。この例では、三相コイル（ステータコイル）２４、２６をロータ１４０のシャフト１３０方向において、分割して形成されている。図において、左側が三相コイル２４、右側が三相コイル２６である。４極対のモータであれば、各三相コイル２４、２６共、４つの部分ステータに分割され、各部分ステータの各相のコイルが直列接続される。従って、３相のコイルがシャフト１３０の長さ方向に２つ配置されることになる。この構成においても、三相コイル２４、２６のコイル端は３つずつであり、インバータ回路３０、３２との接続に変更はない。
【０１２３】
なお、上述の例では、４極対のモータについて説明したが、２極対以上であれば、何極対でもよい。
【０１２４】
【発明の効果】
以上説明したように、本発明によれば、コイルの中性点に電池を接続することで、電池の電力をコンデンサに輸送して昇圧することができる。そして、このような昇圧機能を有するにも拘わらず、別体の昇圧コンバータは不要であり、インバータ一体型駆動装置を得ることができる。
【０１２５】
また、本発明よれば、一対のスター結線コイルの中性点間に電池を接続する装置では、２つのスター結線コイルの中性点間の電位差が規定されるだけであって、それぞれのスター結線コイルの平均電位（中性点電位）自体は任意に設定することができる。そこで、コンデンサ電圧（インバータ入力電圧）について、幅広い制御を行うことができる。
【０１２６】
また、第１および第２のスター結線のコイルのうちの一方のコイルに供給する電流の最大振幅値を減少させ、その減少分に対応する電流を他方のコイルに供給する電流に加算することで、最大電流を低減することができ、インバータやコイルも小さくすることができ、好適な一体型駆動装置を得ることができる。
【図面の簡単な説明】
【図１】 本発明の一実施例である動力出力装置２０の構成の概略を示す構成図である。
【図２】 ２Ｙモータ２２の三相コイル２４と三相コイル２６との関係を説明する説明図である。
【図３】 ２ＹＤＣの装置構成を示す図である。
【図４】 電圧指令値と、インバータ搬送波の関係を示す図である。
【図５】 モータコイルを３つとした例を示す図である。
【図６】 従来の通電方法における電流最大振幅値を示す図である。
【図７】 零相リップル非許容時における電流低減の場合の電流最大振幅値を示す図である。
【図８】 零相リップル許容時における電流低減の場合の電流最大振幅値を示す図である。
【図９】 従来の相電流と零相電流などを示す図である。
【図１０】 従来の相電流と零相電流など（コイル間位相差作がある場合）を示す図である。
【図１１】 リップル電流を抑制する場合における相電流と関数ｆを示す図である。
【図１２】 図１１の拡大図である。
【図１３】 リップル電流を抑制する場合における相電流などを示す図である。
【図１４】 ３倍の高調波で交流振幅を変調する場合における相電流およびその振幅最大値を示す図である。
【図１５】 ３倍の高調波で交流振幅を変調する場合における相電流などを示す図である。
【図１６】 リップルを抑制する場合（位相差あり）における相電流などを示す図である。
【図１７】 リップル電流を許容する場合における相電流および関数ｆを示す図である。
【図１８】 リップル電流を許容する場合における相電流などを示す図である。
【図１９】 ６倍の高調波で変調する場合における相電流およびその振幅最大値を示す図である。
【図２０】 ６倍の高調波で交流振幅を変調する場合における相電流などを示す図である。
【図２１】 一体型駆動装置の概略図である。
【図２２】 一体型駆動装置の他の例を示す概略図である。
【符号の説明】
２０ 動力出力装置、２２ ２Ｙモータ、２４，２６ 三相コイル、３０，３２ インバータ回路、３４ 正極母線、３６ 負極母線、３８ コンデンサ、４０ 直流電源、５０ 電子制御ユニット、５２ ＣＰＵ、５４ ＲＯＭ、５６ ＲＡＭ、６１〜６７ 電流センサ、６８ 回転角センサ、Ｔ１１〜Ｔ１６，Ｔ２１〜Ｔ２６ トランジスタ、Ｄ１１〜Ｄ１６，Ｄ２１〜Ｄ２６ ダイオード、Ｂバッテリ、Ｃ コンデンサ、ＩＮＶ１，ＩＮＶ２ インバータ、Ｍ１，Ｍ２ モータコイル。[0001]
BACKGROUND OF THE INVENTION
The present invention relates to an inverter and an inverter-integrated drive device having a built-in coil and a boosting function.
[0002]
[Prior art]
2. Description of the Related Art Conventionally, systems that convert DC power into AC using an inverter and drive an AC motor have been widespread. In this system, the current supply to the AC motor can be easily controlled over a wide range by controlling the inverter, and is widely used in various applications. For example, in an electric vehicle or a hybrid vehicle, direct current power from a battery mounted therein is converted into a desired alternating current (according to an output torque command) by an inverter and supplied to a drive motor. In this system, regenerative power can be used for charging the battery, which is very advantageous.
[0003]
Here, a wiring through which a large current flows is required between the inverter and the motor, and it is required to arrange them as close as possible. In particular, in the case of a motor mounted on a vehicle, it is required to be compact.
[0004]
Therefore, an inverter-integrated motor in which an inverter and a motor are housed in a single housing is conventionally known. According to this configuration, wiring can be easily routed, and the overall configuration can be made compact.
[0005]
[Problems to be solved by the invention]
Here, a DC voltage from the battery is supplied to the inverter, and the inverter converts this into AC. In a motor for a vehicle, it is necessary to generate a large torque at the time of starting, and the output of the motor becomes very large. In such a case, if the battery voltage is low, the current value for obtaining the output becomes very large and the energy loss increases. Therefore, I want to increase the battery voltage sufficiently. If the battery capacity is sufficiently increased, fluctuations in battery voltage can be suppressed, but it is difficult to increase the battery capacity very much, and fluctuations in battery voltage are inevitable.
[0006]
On the other hand, a system is also known in which the battery voltage is set to a relatively low voltage and the voltage is increased using a boost converter. According to this system, the battery voltage can be made relatively low, and the inverter input voltage and the motor input voltage can be maintained at a high voltage.
[0007]
However, the boost converter requires a coil in addition to the switching transistor. Therefore, integrating such a boost converter with a motor has a problem in that it takes up space and installation becomes difficult.
[0008]
The present invention has been made in view of the above problems, and an object of the present invention is to provide an inverter-integrated drive device that can be compactly formed while having a boosting function.
[0009]
[Means for Solving the Problems]
  The present inventionsoIs connected to the neutral point of the star connection coil, the battery terminal for connection to the battery, the output side is connected to the end of the coil, and the DC power on the input side is changed to polyphase AC An inverter that converts and supplies the coil, and a capacitor terminal that is connected to the input side of the inverter and connects to a capacitor that supplies DC power to the inverter. It is possible to transport battery power to the capacitor via a coil or inverter.The
[0010]
In this way, by connecting the battery to the neutral point of the coil, the power of the battery can be transported to the capacitor and boosted. In spite of having such a boost function, a separate boost converter is not required, and an inverter-integrated drive device can be obtained.
[0011]
Further, the present invention is connected to the first coil of the star connection, the second coil of the star connection, and the neutral points of the first and second coils, respectively, and to connect the battery between the pair of neutral points. A pair of battery terminals, an output side connected to an end of the first coil, a DC inverter on the input side is converted into a polyphase AC and supplied to the first coil, and the second coil An output side is connected to the end, and a second inverter that converts the DC power on the input side to a multiphase AC and supplies the second coil to the second coil, and is connected to the commonly connected input side of the first and second inverters, A second terminal for connecting to the capacitor, housing these in one housing, and transporting the power of the battery to the capacitor via the first and second coils and the first and second inverters. It is possible.
[0012]
In this way, in the device for connecting a battery between the neutral points of a pair of star connection coils, only the potential difference between the neutral points of the two star connection coils is defined, and the average of the respective star connection coils is determined. The potential (neutral point potential) itself can be set arbitrarily. Thus, a wide range of control can be performed on the capacitor voltage (inverter input voltage).
[0013]
In addition, it is preferable that one rotor is disposed in the vicinity of the first coil and the second coil, and the one rotor is rotationally driven by the first and second coils.
[0014]
In addition, the maximum amplitude value of the current supplied to one of the first and second star connection coils may be reduced, and the current corresponding to the decrease may be added to the current supplied to the other coil. Is preferred.
[0015]
With this configuration, the maximum current can be reduced, the inverter and the coil can be made small, and a suitable integrated drive device can be obtained.
[0016]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, embodiments of the present invention will be described with reference to the drawings.
[0017]
FIG. 1 is a diagram showing a circuit configuration of an entire system including an inverter-integrated motor according to the present invention. This configuration is the same as that proposed in Japanese Patent Application No. 2001-331175.
[0018]
That is, the power output apparatus 20 shown in the figure includes a two-coil motor (hereinafter referred to as a 2Y motor) 22 having two three-phase coils 24 and 26 that are Y-connected (star-connected), and two three-phase coils 24, 26, two inverter circuits 30 and 32 that share the positive electrode bus 34 and the negative electrode bus 36, a capacitor 38 connected to the positive electrode bus 34 and the negative electrode bus 36, and the two three-phase coils 24 of the 2Y motor 22. , 26 is provided with a DC power supply 40 provided between neutral points, and an electronic control unit 50 for controlling the entire apparatus.
[0019]
FIG. 2 is an explanatory diagram illustrating the relationship between the two three-phase coils 24 and 26 of the 2Y motor 22. The 2Y motor 22 includes, for example, a rotor having a permanent magnet attached to the outer surface, and a stator wound by shifting two three-phase coils 24 and 26 in the rotational direction by an angle α as illustrated in FIG. The configuration is the same as that of an ordinary synchronous generator motor capable of generating power except that two three-phase coils 24 and 26 are wound. Since the three-phase coils 24 and 26 are shifted in the rotational direction by an angle α, the 2Y motor 22 can be considered as a six-phase motor. In order to drive the 2Y motor 22, a three-phase alternating current having a phase difference by a coil deviation angle α with respect to the three-phase alternating current applied to the three-phase coil 24 by the inverter circuit 30 is applied to the three-phase coil 26. Thus, the inverter circuit 32 may be controlled. The rotating shaft of the 2Y motor 22 is an output shaft of the power output device 20 of the embodiment, and power is output from this rotating shaft. Since the 2Y motor 22 of the embodiment is configured as a generator motor as described above, power can be generated by the 2Y motor 22 if power is input to the rotating shaft of the 2Y motor 22.
[0020]
The inverter circuits 30 and 32 are each composed of six transistors T11 to T16 and T21 to T26 and six diodes D11 to D16 and D21 to D26. The six transistors T11 to T16 and T21 to T26 are arranged in pairs so as to be on the source side and the sink side with respect to the positive electrode bus 34 and the negative electrode bus 36, respectively. Each of phase coils 24 and 26 (UVW) is connected. Accordingly, if the on-time ratios of the transistors T11 to T16 and T21 to T26 that make a pair in a state where the voltage is applied to the positive electrode bus 34 and the negative electrode bus 36 are controlled by the phase difference of the coil deviation angle α, the 2Y motor A rotating magnetic field is formed by the 22 three-phase coils 24 and 26, and the 2Y motor 22 can be rotationally driven.
[0021]
Further, in the present embodiment, the two three-phase coils 24 and 26 may be separately disposed, and a corresponding rotor may be provided to constitute two completely different motors.
[0022]
The electronic control unit 50 is configured as a microprocessor centered on a CPU 52, and includes a ROM 54 that stores a processing program, a RAM 56 that temporarily stores data, and an input / output port (not shown). The electronic control unit 50 includes phase currents Iu1, Iv1, Iw1, Iu2, Iv2, Iw2 and 2Y from current sensors 61 to 66 attached to uvw phases of the three-phase coils 24 and 26 of the 2Y motor 22. The neutral point current Io from the current sensor 67 attached to the neutral point of the motor 22, the rotational angle θ of the rotor of the 2Y motor 22 from the rotational angle sensor 68 attached to the rotational shaft of the 2Y motor 22, and the capacitor 38. The voltage Vc between the terminals of the capacitor 38 from the voltage sensor 70 attached to the A and the command value related to the driving of the 2Y motor 22 are input via the input port. Here, any one of the current sensors 61 to 63 and the current sensors 64 to 66 can be omitted, and any one of them may be used as a sensor dedicated to abnormality detection. The electronic control unit 50 outputs a control signal for performing switching control of the transistors T11 to T16 and T21 to T26 of the inverter circuits 30 and 32 through an output port.
[0023]
"Capacitor voltage control in 2YDC"
As described above, in the present embodiment, a DC power source is disposed between the neutral points of the two multiphase coils, and the switching of the inverter circuits 30 and 32 that control the power supply to the two multiphase coils is controlled. Thus, the voltage of the capacitor 38 which is the power source of the two inverter circuits 30 and 32 was controlled.
[0024]
Here, when the 2YDC system of this embodiment is rewritten with the inside of the inverter omitted, it can be expressed as shown in FIG.
[0025]
That is, one end of the capacitor C is connected to a constant voltage power source (for example, ground). Then, both ends of the capacitor C are connected to the inverter INV1 and the inverter INV2, respectively. That is, the output of the capacitor C is input to the inverters INV1 and INV2 as a power source.
[0026]
The inverter INV1 has three-phase outputs U1, V1, and W1, and the three-phase coils U, V, and W of the motor coil M1 are connected to the inverter INV1, respectively. The inverter INV2 has three-phase outputs U2, V2, and W2, to which the three-phase coils U, V, and W of the motor coil M2 are connected, respectively.
[0027]
Here, although the motor coils M1 and M2 are shown separately, they are coils of one motor as described above, and are normally arranged so as to be different from each other by a predetermined angle with respect to the motor, and have phases different from each other by the predetermined angle. Current is supplied. As a result, both currents supplied to both motor coils M1, M2 function as motor drive currents.
[0028]
Each phase motor coil of the motor coils M1 and M2 is commonly connected at a neutral point, and the neutral points of the motor coils M1 and M2 are connected via a battery B. In this example, the positive electrode of the battery B is connected to the neutral point of the motor coil M1, and the negative electrode of the battery B is connected to the neutral point of the motor coil M2.
[0029]
Although not shown, the inverters INV1 and INV2 are each composed of two switching transistors connected in series between the first power source p and the second power source m (the first power source p is grounded in the illustrated example). There are three arms, and the midpoint of these arms is connected to the end of each phase coil.
[0030]
Therefore, by controlling on / off of the switching transistors in the inverters INV1 and INV2, a desired current can be supplied from the capacitor C to the motor coils M1 and M2 to drive them. Further, by adding a difference in the length of the ON period of the upper transistor and the ON period of the lower transistor in the inverters INV1 and INV2, other than the motor driving phase current entering and exiting from the neutral point in the motor coils M1 and M2 The current (zero phase current) is controlled.
[0031]
Here, in the present embodiment, the inverters INV1 and INV2 are driven using the voltage (output voltage) Vc across one capacitor C as a power source. And the both-ends voltage (output voltage) E of the battery B does not fluctuate fundamentally. Therefore, by controlling the zero-phase current, the midpoint potential of the motor coils M1 and M2 can be arbitrarily set while maintaining a difference corresponding to the voltage of the battery B.
[0032]
As shown in FIG. 3, the voltage of the first power source p is Vp, the voltage of the second power source m is Vm, the output current of the capacitor C is ic, and the voltage across the capacitor C is Vc (= | Vm−Vp |). The current from the first power source p of the inverter INV1 is ip1, the current from the second power source m of the inverter INV1 is im1, the current from the first power source p of the inverter INV2 is ip2, and the current from the second power source m of the inverter INV2 Is im2. For the motor coil M1, u-phase current iu1, v-phase current iv1, w-phase current iw1, u-phase end voltage Vu1, v-phase end voltage Vv1, w-phase end voltage Vw1, and u-phase current iu2 for motor coil M2. The v-phase current iv2, the w-phase current iw2, the u-phase end voltage Vu2, the v-phase end voltage Vv2, and the w-phase end voltage Vw2. The neutral point voltage of the motor coil M1 is Vz1, the neutral point voltage of the motor coil M2 is Vz2, the battery B voltage is E, and the zero-phase current is ie.
[0033]
In particular, in the present system, the relationship between the neutral point potentials Vz1 and Vz2 of the motor coils M1 and M2 and the power supply voltage of the inverters INV1 and INV2, that is, the output voltage Vc of the capacitor C is such that the upper transistors in the inverters INV1 and INV2 The potential difference between the neutral points of the two motor coils M1 and M2 is determined by the ratio of the ON period of the side transistor, and is the battery B voltage E (= | Vz1-Vz2 |). Therefore, the voltage across the capacitor C is determined by the ratio (modulation rate) of the ON periods of the upper transistors and the lower transistors of the inverters INV1 and INV2.
[0034]
The inverters INV1 and INV2 control the neutral point potentials Vz1 and Vz2 of the motor coils M1 and M2 by PWM control of the internal switching transistors. Here, the ratio (modulation rate) between the ON period of the upper transistor and the ON period of the lower transistor is, as shown in FIGS. 4 (a) and 4 (b), the voltage command value for one cycle of the carrier wave that is a triangular wave. It is the ratio of amplitude. That is, when the voltage command value is increased, the period during which the triangular wave exceeds the command value is reduced accordingly. Then, by setting the period in which the triangular wave exceeds the command value as the ON period of the upper transistor of each phase and the OFF period of the lower transistor, the ratio of the ON period of the upper and lower transistors (that is, the modulation factor) is determined. 4A shows the modulation factor d1 of the inverter INV1, and FIG. 4B shows the modulation factor d2 of the inverter INV2.
[0035]
Thus, the neutral point potential is determined by the modulation rate, and the ratio between the neutral point potential and the capacitor voltage is determined by the modulation rate. Further, the potential difference between the two neutral point potentials is the voltage E of the battery B. Therefore, there is the following relationship between the modulation factor and the capacitor voltage Vc.
[0036]
Vc = E / (d1-d2)
Therefore, the capacitor voltage Vc can be determined by controlling the modulation rate of both inverters INV1, INV2.
[0037]
In the above example, the switching transistor is turned on / off without dead time with respect to the carrier wave period Ts of the inverter. That is, when the duty ratio is 50%, both the upper and lower transistors are turned on for a period of 50%. However, in order to completely eliminate the through current during the switching period, a dead time Td for turning off both the upper and lower transistors is often provided. In this case, the above equation is rewritten and applied as follows.
[0038]
Vc = E / {(d1-Td / Ts)-(d2 + Td / Ts)}
As described above, even when the dead time is provided, the capacitor voltage Vc can be determined by controlling the modulation factors d1 and d2.
[0039]
Further, FIG. 5 shows still another modification. In this example, there are three motor coils M1, M2 and M3. The neutral points of the motor coils M1, M2 are connected by the battery B1, and the neutral points of the motor coils M2, M3 are connected by the battery B2. Further, the output of the inverter INV1 is connected to the motor coil M1, the output of the inverter INV2 is connected to the motor coil M2, and the output of the inverter INV3 is connected to the motor coil M3. Then, both ends of the capacitor C are connected to the inputs of the inverters INV1, INV2, and INV3.
[0040]
In such a system, the output voltage of the capacitor C is Vc, the output voltage of the battery B1 is E1, the output voltage of the battery B2 is E2, the modulation rate of the inverter INV1 is d1, the modulation rate of the inverter INV2 is d2, and the modulation of the inverter INV3 If the rate is d3, these have the following relationship.
[0041]
Vc = E1 / (d1-d2) = E2 / (d2-d3)
Therefore, the desired capacitor voltage Vc can be obtained by controlling the modulation factors d1, d2, and d3 so as to satisfy this equation. Moreover, the electric charge between batteries B1 and B2 can be transported by making the values of E1 / (d1-d2) and E2 / (d2-d3) different.
[0042]
Although three motor coils M1, M2, and M3 are used, the same control can be performed with four or more motor coils. Further, the plurality of motor coils may constitute one electric motor or a plurality of electric motors.
[0043]
"Suppression of maximum amplitude"
Next, suppression of the maximum current amplitude value in this system will be described. This is achieved by changing the current distribution to the two motor coils M1, M2.
[0044]
"Effect of the embodiment"
Before explaining a specific example of control according to the embodiment, the relationship between the motor output and the phase current is shown by simulation, and the current reduction effect of the present invention is shown.
[0045]
This simulation was performed according to the following procedure. First, the phase current iu1 of one phase (here, u phase) is divided into an average value (DC component) idc and one component (AC component) iac in one rotation. Further, for the AC component iac, a function g (θ) normalized by the amplitude Iac is introduced.
[0046]
That is,
[Expression 1]

And
[0047]
Subsequently, the voltage Vw is defined from the relationship between the battery voltage E and the capacitor voltage Vc of this system. This is because the phase current amplitude Iac has a maximum value obtained by subtracting the battery voltage E from the capacitor voltage Vc. Further, it is assumed that the voltage vv applied to each coil at the same time changes with the above-described current iac with a constant phase difference (power factor cos φ).
[0048]
That is,
[Expression 2]
Vw = Vc-E (6)
vv = Vwg (θ + φ) (7)
And
[0049]
Further, since there are six coils, the relationship between the motor output Wo and the work performed by each coil can be arranged as follows.
[0050]
[Equation 3]

The motor output can be approximated to Wo = ieE, assuming that the loss is sufficiently small. From this relationship, the following equation is obtained.
[0051]
[Expression 4]
ie = Wo / E (13)
From the above, the current flowing through each phase coil is obtained by the following equation using Iac, ie obtained by equations (12) and (13). However, the ripple of ie is not taken into consideration.
[0052]
[Equation 5]

Next, conditions used for the analysis are shown. The battery voltage E = 42V or 105V, the capacitor voltage Vc = 210V (step-up rate Vc / E = 5 or 2), the power factor cos θ = 0.8, and the energization of the maximum value of the AC current amplitude with respect to the motor output Wo The difference by the method is shown.
[0053]
The results are shown in FIGS. These figures show the difference in the maximum phase current value due to the difference in step-up rate. The horizontal axis represents the motor output, the vertical axis represents the maximum phase current value (imax), the solid line represents the maximum phase current value, and the broken line represents the maximum phase current value. The DC component (ie / 3) of the values is shown.
[0054]
6 shows a conventional maximum phase current value during energization, FIG. 7 shows a maximum phase current value during maximum suppression energization under zero-phase ripple non-permissible conditions, and FIG. 8 shows a maximum suppression energization method under zero-phase ripple permissible conditions (4 The maximum value of the phase current at the time of Section 2.2) is shown.
[0055]
From these figures, the following can be understood.
[0056]
In any case, the magnitude of the phase current varies greatly depending on the boosting rate, and the higher the boosting rate, the larger the proportion of the DC component in the phase current.
[0057]
・ Also, the effect of suppressing the magnitude of the phase current due to the difference in energization method can be confirmed.
[0058]
When comparing the maximum value of the phase voltage (AC component, DC component) at Wo = 40 kW and a step-up ratio of 5 times, the maximum value 477A (159, 317A) in the conventional energization of FIG. 6 and 454A (136, 136) in FIG. 317A) and 402A (85, 317A) in FIG.
[0059]
"Description of the conventional energization method as the basis of the present invention"
A conventional energization method for the 2YDC variable inverter shown in FIG. 6 will be described. The phase currents iu1, iv1, iw1, iu2, iv2, and iw2 that normally flow through the 2YDC variable inverter shown in FIG. 3 are zero phase current ie, alternating current amplitude A, rotor rotational speed, and rotational angle ω, If θ (θ = ωt), it is expressed by the following equation.
[0060]
[Formula 6]

Here, assuming that A = 1 (A) and ie = 3 (A), equations (15) to (20) have the relationship shown in FIG. However, in FIG. 9, the current of each of iu1 and iu2 is the first stage, the sum of the currents of iu1 and iu is the second stage, and the d-axis currents id1 and id2 after the phase current is converted to the dq axis are the third stage. The q-axis currents iq1 and iq2 are the fourth stage, and the last stage is one third (one phase) of the zero-phase current ie.
[0061]
Here, if reluctance torque is not considered, iq1 + iq2 contributes to the motor torque (in this case, the magnet position is not included in the analysis, so iu1 + iu2 can be said to be a current component that contributes to the motor torque), ie is the battery And the current flowing between the capacitors. The maximum value of the phase current at this time is 2.00 (A). In the relationship of FIG. 9, the condition for the current for generating the motor driving torque and the current between the battery and the capacitor can be expressed by Equation (21).
[0062]
[Expression 7]

Furthermore, by introducing equations (22) and (23), equation (21) can be rewritten as follows. In equation (24), id and iq are current components represented by the dq axis, and are constant here.
[0063]
[Equation 8]

When there is a phase difference between the motor coils (when the coil position of one star connection and the coil position of another star connection are deviated by an angle ξ), the energized current is expressed by equations (25) to (25) 30), and equation (24) becomes equation (31).
[0064]
[Equation 9]

When A = 1 (A), ie = 3 (A), and ξ = 30 °, the equations (25) to (30) have the relationship shown in FIG. Thus, it can be seen that there is the same relationship as in FIG. 9 when the phase difference of the coil is taken into consideration.
[0065]
“Description of 2YDC Variable Type Inverter of Embodiment”
In the embodiment of FIG. 7, the maximum value of the phase current is suppressed without allowing the occurrence of ripples in the zero-phase current.
[0066]
That is, in this embodiment, the maximum amplitude is suppressed by adding a predetermined function to the phase currents iu1, iv1, and iw1 in the 2YDC variable inverter of FIG. Then, by subtracting the added function from the phase currents iu2, iv2, and iw2, the maximum amplitude of the current is suppressed without changing the output torque of the motor. Further, in the present embodiment, zero-phase current ripple is not allowed.
[0067]
In order to reduce the current amplitude without changing the magnitude of the motor output torque and the zero-phase current, the phase currents iu1, iv1, iw1, iu2, iv2, and iw2 need to satisfy the relationship of Expression (21). is there. That is, it is necessary to satisfy the following expression (32). In this equation, the sum of the currents of the corresponding phases of each star connection is a sine wave, and the sum of the currents of each phase in each star connection changes the zero-phase current value or the sign of the zero-phase current value. Is equal to
[0068]
[Expression 10]

Here, since the rank of the matrix on the left side of Expression (32) is 4, two free parameters fu (θ) and fv (θ) are introduced, and the following sufficient conditions are satisfied so as to satisfy Expression (32). Can be rewritten.
[0069]
## EQU11 ##

Here, fu (θ), fv (θ), and fw (θ) are parameters (degree of freedom is 2) that can be used for design.
[0070]
Therefore, by giving fu (θ), fv (θ), fw (θ) (degrees of freedom of 2) satisfying the equations (33) to (42), the phase without changing the output torque and the zero-phase current. The currents iu1, iv1, iw1, iu2, iv2, and iw2 can be varied. Then, fu (θ), fv (θ), and fw (θ) are selected so as to reduce the maximum amplitude of the phase currents iu1, iv1, iw1, iu2, iv2, and iw2, thereby achieving the intended purpose. be able to.
[0071]
In the embodiment of FIG. 8, the conditions are relaxed, the generation of ripples in the zero-phase current is allowed, and the maximum value of the phase current is suppressed. In this case, the condition of Expression (39) can be removed. Accordingly, the degree of freedom in selecting fu (θ), fv (θ), and fw (θ) is increased. And it becomes possible to make the maximum value of a phase current smaller.
[0072]
In the above description, it is assumed that there is no phase difference between the two motor coils M1 and M2. In practice, the coils are often arranged with a phase difference. In this case, by giving a phase difference corresponding to the coil current, the influence of having the phase difference is eliminated.
[0073]
When there is a phase difference ξ between the coils of each star connection, Equation (31) is a condition that changes to Equation (21). That is, in order to reduce the current amplitude without changing the motor generation torque and the magnitude of the zero-phase current, the phase currents iu1, iv1, iw1, iu2, iv2, and iw2 need to satisfy Expression (43). . In this equation, the sum of the dq-axis currents corresponding to each star connection is constant, and the sum of the currents of each phase in each star connection changes the value of the zero-phase current or the value of the zero-phase current. Means equal to things.
[0074]
[Expression 12]

Here, as one of the solutions satisfying the equation (43), the following result is derived as in the case described above.
[0075]
[Formula 13]

Here, fu (θ), fv (θ), fw (θ), hu (θ), hv (θ), and hw (θ) are parameters that can be used for the design. Further, when ξ = 0 °, the functions of the equations (33) and (42) satisfy the equations (44) and (59).
[0076]
Then, by giving fu (θ), fv (θ), fw (θ), hu (θ), hv (θ), hw (θ) satisfying the equations (44) to (59), the output torque and zero The phase currents iu1, iv1, iw1, iu2, iv2, and iw2 can be changed without changing the phase current. Further, fu (θ), fv (θ), fw (θ), hu (θ), hv (θ), hw (θ) are suppressed to the maximum value of the phase currents iu1, iv1, iw1, iu2, iv2, iw2. By making it possible, the intended purpose can be achieved.
[0077]
Further, if the condition is relaxed and the ripple current is allowed for the zero-phase current, fu + fv + fw + hu + hv + hw = 0 is a condition instead of the conditions of equations (50) and (51).
[0078]
"Specific example when zero-phase current does not allow ripple"
The energization method satisfying the above condition with the inter-coil phase difference ξ = 0 ° can be obtained by setting fu (θ), fv (θ), and fw (θ) as shown in equations (60) to (62). can get. In addition, g1 of a formula is a constant put in order to satisfy the conditions of formulas (40) to (42), and in this case, g1 = 0.867.
[0079]
[Expression 14]

In the case of A = 1 (A) and ie = 3 (A), the waveform of fu (θ) is compared with iu1 and shown in FIGS. FIG. 11 is a graph in which the scale of the vertical axis of the graph of fu (θ) and the graph of iu1 are combined, and FIG. 12 is an enlarged view for easy viewing of the waveform of fu (θ). From the figure, the waveform of fu (θ) is a waveform in which the peak portion of a sine wave is cut out by a width of 60 degrees, arranged in the order of positive side, negative side, and positive side, and the negative side is set to double the positive side. It has become. That is, it has a waveform that most suppresses the maximum peak of iu1.
[0080]
Therefore, by adding fu (θ), fv (θ), fw (θ) of such a form to the sine curve, the phase current can be suppressed to the maximum current, and the change in the output torque based on this can be Absent. Furthermore, in this example, the condition that no zero-phase current is generated is also satisfied.
[0081]
Furthermore, the result using the conditions of Formula (60)-(62) is shown in FIG. The figure shows the following.
[0082]
・ Zero phase current
The zero-phase current is ie = 3 (A) and does not include a ripple component.
[0083]
·torque
The current (iu1 + iu2) for generating the motor torque is equivalent to that in FIG. 9, and the torque is generated as intended.
[0084]
-Phase current magnitude
The maximum value of the magnitude of the phase current is 1.866 (A). The breakdown of the size is 0.866A for the AC component and 1A for the DC component.
[0085]
Thus, by using fu (θ), fv (θ), and fw (θ) shown in equations (60) to (62), the phase current can be obtained without affecting the zero-phase current and the motor output torque. The maximum value of can be suppressed.
[0086]
As another example of the case where ξ = 0 ° and no ripple is allowed in the zero-phase current, a case where the AC amplitude is modulated by three times higher harmonics is shown.
[0087]
That is, in the case of A = 1 (A) and ie = 3 (A), the waveform of fu (θ) is compared with iu1, and is shown in FIG. This waveform is a sine wave having a frequency that is three times the amplitude of the original AC waveform, and is modulated by the following equation.
[0088]
[Expression 15]

Further, equations (63) and (64) can be arranged as follows.
[0089]
[Expression 16]

Here, if fu (θ) = αsin (3θ) Asin (θ), the conditions of the equations (33) to (42) are satisfied. That is, by setting fu (θ) = αsin (3θ) Asin (θ), the following result is obtained.
[0090]
FIG. 14 shows this fu (θ). Furthermore, the result using this fu (θ) is shown in FIG. The figure shows the following.
[0091]
・ Zero phase current
The average value of the zero-phase current is ie = 3 (A). Its magnitude is three times the amplitude of added fu (θ).
[0092]
·torque
The current (iu1 + iu2) for generating the motor torque is equivalent to that in FIG. 9, and the torque is generated as intended.
[0093]
-Phase current magnitude
The maximum value of the phase current is 1.872 (A). The breakdown of the size is 0.872A for the AC component and 1A for the DC component.
[0094]
Next, FIG. 16 shows the result when the phase difference is shifted by 30 ° in the coil position between the star connections (ξ = 30 °). This figure shows the following.
[0095]
・ Zero phase current
The zero-phase current is ie = 3 (A) and does not include a ripple component.
[0096]
·torque
The currents (id and iq) for generating the motor torque are the same as in FIG. 9, and the torque is generated as intended.
[0097]
-Phase current magnitude
The maximum value of the magnitude of the phase current is 1.866 (A). The breakdown of the size is 0.866A for the AC component and 1A for the DC component.
[0098]
・ Phase current waveform
The command value used here is a waveform that changes sharply in order to suppress the magnitude of the current. However, in the actual case, this is realized by filtering to remove high frequency components. In this case, however, the current suppressing effect is slightly worse.
[0099]
"Specific example of allowing zero-phase current ripple"
One of the energization methods that can suppress the magnitude of the phase current under the condition that ξ = 0 ° and the zero-phase current ripple is allowed is that fu (θ), fv (θ), and fw (θ) are expressed by the equations (67) to (69). In addition, g2 of a formula is a constant put in order to satisfy the conditions of formulas (40) to (42), and in this case, g2 = −0.637.
[0100]
[Expression 17]

In the case of A = 1 (A) and ie = 3 (A), the waveform of fu (θ) is compared with iu1, and is shown in FIG. Furthermore, the result using the conditions of Formula (67)-(69) is shown in FIG. The figure shows the following.
[0101]
・ Zero phase current
The average value of the zero-phase current is ie = 3 (A), but the ripple component is included and the magnitude thereof is 0.46A.
[0102]
·torque
The current (iu1 + iu2) for generating the motor torque is equivalent to that in FIG. 9, and the torque is generated as intended.
[0103]
-Phase current magnitude
The maximum value of the phase current is 1.63 (A). The breakdown of the size is 0.63A for the AC component and 1A for the DC component.
[0104]
Next, as an example of another method for allowing ripples in the zero-phase current at ξ = 0 °, a case of adding 6 times higher harmonics is shown. In the case of A = 1 (A) and ie = 3 (A), the waveform of fu (θ) is compared with iu1, and is shown in FIG. For the amplitude of the harmonic, a value optimized so as to minimize the phase current is used.
[0105]
Furthermore, the result of using this fu (θ) is shown in FIG. The figure shows the following.
[0106]
・ Zero phase current
The average value of the zero-phase current is ie = 3 (A), but includes a ripple component. Its magnitude is three times the amplitude of added fu (θ).
[0107]
·torque
The current (iu1 + iu2) for generating the motor torque is equivalent to that in FIG. 9, and the torque is generated as intended.
[0108]
-Phase current magnitude
The maximum value of the phase current is 1.96 (A). The breakdown of the size is 0.96A for the AC component and 1A for the DC component.
[0109]
As described above, according to the above embodiment, the maximum current value of the phase current can be suppressed without increasing or decreasing the torque, and the current capacity of the device can be reduced without impairing the function as a motor. , It is possible to reduce the cost of the system while maintaining equivalent performance. By suppressing the torque ripple, the function of the motor can be made sufficient.
[0110]
Further, for current suppression, a high frequency component needs to be superimposed on the current. For this reason, it is necessary to control the current up to a high frequency range. However, more effective control is possible by switching the control depending on the rotation speed.
[0111]
That is, since the maximum amplitude value is suppressed in a low rotation region where the current value is large, the original control frequency band is low, so that control is not extremely difficult even if harmonics are superimposed. On the other hand, since the conventional method is used in the high rotation range, the above-described control problem at the time of high frequency superposition does not occur. Furthermore, appropriate control can be performed by suppressing the ripple of the zero-phase current in the middle rotation region.
[0112]
By such control switching, current suppression can be realized while avoiding control problems due to current suppression.
[0113]
"Integrated drive"
As described above, according to the present embodiment, there is shown a system that includes two three-phase coils 24 and 26 and thereby constitutes one motor. And it is suitable to integrally form such a motor with an inverter.
[0114]
FIG. 21 shows the overall configuration. An inverter 110 is arranged on one side in the housing 100. For example, the inverter 110 includes two inverter circuits 30 and 32 and is formed on one substrate 112. The inverter 110 includes an element 114 such as a plurality of power transistors, and the inverter 110 and a stator (stator coil) 150 are connected by a wiring 116. In the figure, only one wiring 116 is shown, but each output terminal of the inverter 110 is connected to each input terminal of the stator 150. In addition, a cooling plate 118 including a cooling water passage is disposed between the partition 120 and the substrate 112, thereby cooling the element 114.
[0115]
The partition 120 of the housing 100 is provided with a bearing (bearing) 122 at the center thereof, and a shaft 130 is supported by the bearing. A bearing (bearing) 124 is provided at a position where the shaft 130 penetrates on the opposite side of the inverter 110 of the housing 100, and the shaft 130 protrudes from the housing 100 through the bearing 124.
[0116]
A rotor 140 is disposed at an intermediate portion in the housing of the shaft 130, and a stator 150 is disposed so as to surround the rotor 140. As shown in the right view of FIG. 21, the rotor 140 and the stator 150 are formed so that the cylindrical rotor 140 is concentrically surrounded by the hollow cylindrical stator 150.
[0117]
In this example, the stator 150 is divided into four partial stators p1, p2, p3, and p4 in the circumferential direction, and is a four-pole pair (pole) motor. The partial stators p1, p2, p3, and p4 include three-phase coils 24 and 26, respectively. Therefore, two three-phase coils 24 and 26 are provided in the partial stator p1. Since there are four partial stators, the two three-phase coils 24 and 26 are each connected in series to each other, and are formed into six-phase coils as a whole.
[0118]
The output ends of the inverter circuits 30 and 32 are connected to the six coil ends of the two three-phase coils 24 and 26, respectively. This connection is made within the housing 100. On the other hand, the DC power supply 40 is externally connected and is connected to a pair of battery terminals (each connected to a neutral point of the three-phase coils 24 and 26) provided in the housing 100.
[0119]
In this example, the capacitor 38 is also formed externally. Therefore, a pair of terminals connected to the positive electrode bus 34 and the negative electrode bus 36 of the inverter circuits 30 and 32 are provided in the housing 100, and an external capacitor 38 is connected thereto. It is also preferable to accommodate the capacitor 38 in the housing 100. Thereby, the external wiring can be further reduced.
[0120]
Moreover, it is not necessary to arrange both the three-phase coils 24 and 26 in all the partial stators p1 to p4. For example, the three-phase coil 24 may be disposed in the partial stators p1 and p3, and the three-phase coil 26 may be disposed in the partial stators p2 and p4. In this case, the phase coils of the partial stators p1 and p3 are connected in series, and the phase coils of the partial stators P2 and p4 are connected in series.
[0121]
Even in this configuration, there are six end portions of the three-phase coils 24 and 26, and the three output ends of the inverter circuit 30 are connected to the three end portions of the three-phase coil 24, so Three output terminals of the inverter circuit 32 are connected to one output terminal.
[0122]
Further, another example is shown in FIG. In this example, the three-phase coils (stator coils) 24 and 26 are divided and formed in the direction of the shaft 130 of the rotor 140. In the figure, the left side is the three-phase coil 24, and the right side is the three-phase coil 26. In the case of a four-pole motor, each of the three-phase coils 24 and 26 is divided into four partial stators, and the coils of each phase of each partial stator are connected in series. Therefore, two three-phase coils are arranged in the length direction of the shaft 130. Also in this configuration, there are three coil ends of the three-phase coils 24 and 26, and the connection with the inverter circuits 30 and 32 is not changed.
[0123]
In the above example, a 4-pole motor has been described. However, any number of pole pairs may be used as long as it is 2 pole pairs or more.
[0124]
【The invention's effect】
As described above, according to the present invention, the battery power can be transported to the capacitor and boosted by connecting the battery to the neutral point of the coil. In spite of having such a boost function, a separate boost converter is not required, and an inverter-integrated drive device can be obtained.
[0125]
Further, according to the present invention, in the apparatus for connecting a battery between the neutral points of a pair of star connection coils, only the potential difference between the neutral points of the two star connection coils is defined, The average potential (neutral point potential) of the coil itself can be set arbitrarily. Thus, a wide range of control can be performed on the capacitor voltage (inverter input voltage).
[0126]
Further, the maximum amplitude value of the current supplied to one of the first and second star connection coils is reduced, and the current corresponding to the reduced amount is added to the current supplied to the other coil. The maximum current can be reduced, the inverter and the coil can be reduced, and a suitable integrated drive device can be obtained.
[Brief description of the drawings]
FIG. 1 is a configuration diagram showing an outline of a configuration of a power output apparatus 20 according to an embodiment of the present invention.
FIG. 2 is an explanatory diagram for explaining a relationship between a three-phase coil 24 and a three-phase coil 26 of a 2Y motor 22;
FIG. 3 is a diagram illustrating a device configuration of 2YDC.
FIG. 4 is a diagram illustrating a relationship between a voltage command value and an inverter carrier wave.
FIG. 5 is a diagram showing an example in which three motor coils are used.
FIG. 6 is a diagram showing a maximum current amplitude value in a conventional energization method.
FIG. 7 is a diagram showing a current maximum amplitude value in the case of current reduction when zero-phase ripple is not allowed;
FIG. 8 is a diagram showing a current maximum amplitude value in the case of current reduction when zero-phase ripple is allowed.
FIG. 9 is a diagram showing a conventional phase current and a zero-phase current.
FIG. 10 is a diagram showing a conventional phase current and a zero-phase current (when there is a phase difference between coils).
FIG. 11 is a diagram showing a phase current and a function f when suppressing a ripple current.
12 is an enlarged view of FIG.
FIG. 13 is a diagram showing a phase current and the like when suppressing a ripple current.
FIG. 14 is a diagram showing a phase current and its maximum amplitude value when the AC amplitude is modulated by a triple harmonic.
FIG. 15 is a diagram showing a phase current and the like in the case of modulating an AC amplitude with a triple harmonic.
FIG. 16 is a diagram showing a phase current and the like when ripples are suppressed (with phase difference);
FIG. 17 is a diagram showing a phase current and a function f when a ripple current is allowed.
FIG. 18 is a diagram showing a phase current and the like when a ripple current is allowed.
FIG. 19 is a diagram showing a phase current and a maximum amplitude value in the case of modulation with 6 times higher harmonics.
FIG. 20 is a diagram showing a phase current and the like in the case where the AC amplitude is modulated by a 6-fold higher harmonic wave.
FIG. 21 is a schematic view of an integrated drive device.
FIG. 22 is a schematic view showing another example of the integrated drive device.
[Explanation of symbols]
20 Power output device, 22 2Y motor, 24, 26 Three-phase coil, 30, 32 Inverter circuit, 34 Positive bus, 36 Negative bus, 38 Capacitor, 40 DC power supply, 50 Electronic control unit, 52 CPU, 54 ROM, 56 RAM , 61-67 Current sensor, 68 Rotation angle sensor, T11-T16, T21-T26 Transistor, D11-D16, D21-D26 Diode, B battery, C capacitor, INV1, INV2 inverter, M1, M2 Motor coil.

Claims (3)

A first coil of star connection;
A second coil of star connection;
A pair of battery terminals connected to neutral points of the first and second coils, respectively, for connecting a battery between the pair of neutral points;
An output side connected to an end of the first coil, a first inverter for converting the DC power on the input side into a polyphase AC and supplying the first coil;
An output side connected to an end of the second coil, a DC inverter on the input side is converted into a polyphase AC and supplied to the second coil; and
A second terminal connected to the commonly connected input side of the first and second inverters and connected to a capacitor;
Including
An inverter-integrated drive apparatus in which these are housed in one housing and the electric power of the battery can be transported to the capacitor via the first and second coils and the first and second inverters. The apparatus of claim 1 .
An inverter-integrated drive apparatus in which one rotor is disposed in the vicinity of the first coil and the second coil, and one rotor is rotationally driven by the first and second coils. The apparatus according to claim 1 or 2 ,
Inverter-integrated drive for reducing the maximum amplitude value of the current supplied to one of the first and second star-connected coils and adding the current corresponding to the reduced amount to the current supplied to the other coil apparatus.

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