JP3643908B2 - Brushless DC motor control method and apparatus - Google Patents

Description

【０１０１】
【数３】

【０１０２】
【数４】

【０１０３】
【数５】

【０１０４】
【数６】

【０１０５】
【数７】

なお、図８中において、θは電気角、ωは電気角速度、ｖ_γ、ｖ_δはγδ軸電圧、ｉ_γ、ｉ_δはγδ電流、ε_γ、ε_δはγδ軸誘起電圧、φは永久磁石による電機子鎖交磁束、Ｒは電機子抵抗、Ｌ_d、Ｌ_qはｄｑ軸インダクタンス、ｓは微分演算子、＾は推定値を、それぞれ示している。また、図８中のＩ、ｖ_γδ、ｉ_γδ、ε_γδ、Ｌ_I、Ｌ_J、θ_e、Ｔ（θ_e）、α_I、β_Iは数８に示すとおりである。
【０１０６】
【数８】

さらに、フィードバックループ中の非線形要素を線形化すると（ｃｏｓθ_e→１、ｓｉｎθ_e→θ_e、＾ω→ωとすると）、図９に示すブロック構成が得られる。
【０１０７】
図９のブロック構成を用い、ｉ_dqが周期的な場合の収束性を考える。線形近似ブロック線図よりｘ_γは数９で与えられる。
【０１０８】
【数９】

ここで、数９の｛｝内をａ₀＋ａ_1sｓｉｎθ₁＋ａ_1cｃｏｓθ₁、
θ_e＝ｅ₀＋ｅ_1sｓｉｎθ₁＋ｅ_1cｃｏｓθ₁＋ｅ_2sｓｉｎ２θ₁＋ｅ_2cｃｏｓ２θ₁の場合を考える。ｘ_γは数１０となる。
【０１０９】
【数１０】

θ_eは検出できないので、代わりにｘ_γのＤＣ成分と１次成分とを０に制御（ｋθ→∞が必要）すると、数９中の点線部が０になるようにｅ₀、ｅ_1s、ｅ_1cが調整される。数１０より目的（ｅ₀＝ｅ_1s＝ｅ_1c＝０）を達成するためには、ａ₀≠０、かつａ_1s＝ａ_1c＝０であればよい。そして、前者は、｛φ＋（Ｌ_d−Ｌ_q）ｉ_d｝ωの平均値が０でないことと等価であり、後者は、｛φω＋（Ｌ_d−Ｌ_q）［ω_s −ｓ］ｉ_dq｝の１次成分、２次成分が０であることと等価である。これは、ωが一定、かつＬ_d＝Ｌ_qまたはｉ_d、ｉ_qが一定であることと等価である。
【０１１０】
この結果、＾θ→θとなるために十分な条件は、安定、かつ（１）θ_eは小さい（ｃｏｓθ_e→１、ｓｉｎθ_e→θ_e）、（２）φ以外の機器定数（Ｒ、Ｌ_d、Ｌ_q）のずれなし、（３）ｋ_θ＝∞、（４）＾ω＝ω＝一定、（５）Ｌ_d＝Ｌ_qまたはｉ_d、ｉ_qが一定、（６）｛φ＋（Ｌ_d−Ｌ_q）ｉ_d｝ωの平均値が０でない、を満たすことである。 この際、特に（Ｌ_d−Ｌ_q）ｉ_d＋φ≠０、および安定性確保は位置検出のための絶対条件であるが、他は誤差縮小のための条件である。
【０１１１】
また、モータ逆モデルを用い、かつセンサを用いることなく位置検出を行うための他のブロック構成を図１０に示す。
【０１１２】
なお、図１０において、θは電気角、ωは電気角速度、γδ座標は＾θ回転座標、αβ座標は２相直交固定子座標、ｖ_γ、ｖ_δはγδ軸電圧、ｖ_α、ｖ_βはαβ軸電圧、ｉ_α、ｉ_βはαβ軸電流、φは電機子鎖交磁束、Ｒは電機子抵抗、Ｌ_d、Ｌ_qはｄｑ軸インダクタンス、ｓは微分演算子、＾は推定値、￣はセンサ値を、それぞれ示している。また、図１０中のＬ_p、Ｌ_m、、ｖ_γδ、ｖ_αβ、ｉ_αβ、Φ_αβ、ｆ（θ）は数１１に示すとおりである。
【０１１３】
【数１１】

そして、このブロック構成を採用した場合にも、｛（Ｌ_d−Ｌ_q）Φ_αβ ^Tｉ_αβ＋φ｝＝（Ｌ_d−Ｌ_q）ｉ_d＋φ≠０が位置検出のための条件であることが分かる。
【０１１４】
図１１にｄ軸を基準とした電流位相と２相変換後の電流値Ｉ_aに対するΨ＝（Ｌ_d−Ｌ_q）ｉ_d＋φのグラフを示す。
【０１１５】
電流位相が−５０°から−１３０°、Ｉ_a＞７Ａの領域でΨ＝０が認められ、この領域では位置検出が不可能になる可能性があることが分かる。なお、Ｉ_aは３相モータにおける相電流のピーク値の（３／２）^1/2である。
【０１１６】
【発明の実施の形態】
以下、添付図面を参照して、この発明のブラシレスＤＣモータ制御方法およびその装置の実施の態様を詳細に説明する。
【０１１７】
図１はこの発明のブラシレスＤＣモータ制御装置の一実施態様を示すブロック図である。
【０１１８】
このブラシレスＤＣモータ制御装置は、直流電力を入力とするインバータ（電圧形インバータであることが好ましい）１からの出力を、回転子の内部に永久磁石を装着してなるブラシレスＤＣモータ２に供給している。そして、位置センサを用いることなくブラシレスＤＣモータ２の回転子の回転位置θを検出するとともに、回転速度ωを検出する回転子回転位置検出部３と、外部から与えられる速度指令および検出された回転速度ωを入力として速度制御演算を行って電流位相指令および電流振幅指令を出力する速度制御部４と、回転位置θ、電流位相指令および電流振幅指令を入力として電流制御演算を行ってＰＷＭ（パルス幅変調）指令を出力し、インバータ１に供給する電流制御部５とを有している。
【０１１９】
前記回転子回転位置検出部３では、位置検出部において、ブラシレスＤＣモータ１２の固定子巻線の中性点の電圧と固定子巻線の端子電圧の平均値との差を算出し、差電圧のゼロクロスを検出し、検出されたゼロクロスから回転子の予測位置を得て出力し、位置補正部において、差電圧の振幅と差電圧の位相誤差との関係を予め求めておき、それを打ち消すように、位置検出部から出力される予測位置を補正する。これにより、回転子の回転位置を正確に検出する。
【０１２０】
ただし、上記位置補正部に代えて、差電圧の振幅、差電圧の積分信号の振幅、ブラシレスＤＣモータの相電流、ブラシレスＤＣモータの相電圧、インバータの入力電流（ＤＣ電流）、インバータの入力電圧（ＤＣ電圧）、ブラシレスＤＣモータの回転数、インバータ内部情報｛パルス幅変調（ＰＷＭ）指令、回転速度など｝から選択された少なくとも１つの値を入力として補正量を算出し、算出された補正量に基づいて位置検出部から出力される仮の予測位置を補正して、補正後の予測位置として出力する位置補正部を採用することが可能である。
【０１２１】
また、上記位置補正部に代えて、差電圧の振幅およびブラシレスＤＣモータの相電流の振幅を入力として補正量（補正角度）を出力する補正マップと、位置検出部から出力される回転子の仮の予測位置に対して補正量を加算して補正後の予測位置を得て出力する加算部とを採用することも可能である。
【０１２２】
さらに、差電圧を入力として回転子の第１の仮の予測位置を出力する第１位置検出部と、相電圧ＰＷＭ指令および回転子位置を入力としてモデル化された差電圧を出力するモータモデルと、モデル化された差電圧を入力として回転子の第２の仮の予測位置を出力する第２位置検出部と、第１の仮の予測位置と第２の仮の予測位置との差を算出する差算出部と、予測位置どうしの差を入力としてこれを０にする補正量（補正角度）を得て出力する補正量出力部と、第１の仮の予測位置から補正量を減算して補正後の予測位置（上記回転子位置）を得て出力する減算部とを有する構成を採用することも可能である。この場合において、差電圧の振幅を検出する第１振幅検出部と、モデル化された差電圧の振幅を検出する第２振幅検出部と、両振幅の差を算出する差振幅算出部とをさらに有するとともに、補正量出力部に代えて、予測位置どうしの差および両振幅の差を入力としてこれらを０にする補正量（補正角度）を得て出力する補正量出力部を採用することも可能である。
【０１２３】
さらにまた、差電圧を入力として、回転子の第１の仮の予測位置を出力する第１位置検出部と、ブラシレスＤＣモータの相電圧および相電流を入力として回転子の第２の仮の予測位置を出力する第２位置検出部と、第１の仮の予測位置と第２の仮の予測位置との差を算出する差算出部と、算出された差を入力として記憶する誤差記憶部と、第１の仮の予測位置に対して誤差記憶部に記憶されている誤差を加算して補正後の予測位置として出力する加算部とを有する構成を採用することも可能である（特願平１０−３１５０３５号参照）。
【０１２４】
また、前記回転子回転位置検出部３においては、指令電圧とモータ電流とから速度起電力を推定し、推定速度起電力に対応してγ−δ軸を定義し、推定速度起電力を、所定の大きさのｑ軸方向ベクトルとして表現される速度起電力に一致させる制御を行って、回転子の回転位置と速度とを推定する（具体的には、推定速度起電力の方向により推定回転位置を得、推定速度起電力の大きさにより推定速度を得る）（「速度起電力推定に基づくセンサレス突極形ブラシレスＤＣモータ制御」、竹下隆晴他、Ｔ．ＩＥＥ Ｊａｐａｎ，Ｖｏｌ．１１７−Ｄ，Ｎｏ．１，’９７参照）。
【０１２５】
また、前記回転子回転位置検出部３においては、モータ端子電圧と電流に含まれる高調波成分を抽出し（インバータの平均出力電圧ベクトルと各インバータ出力電圧ベクトルとの差を算出することによりモータ端子電圧に含まれる高調波成分を抽出するとともに、変調期間内で用いない電圧ベクトルによる電流ベクトルの変化量から、変調期間に対する所定期間における変調周期の最初と最後の電流ベクトルの差を算出することによりモータ電流ベクトルの高調波成分を抽出する）、回転子の回転位置を推定することもできる（１変調周期の高調波成分に対する方程式に対して左側疑似逆行列を用いてインダクタンス行列を求め、このインダクタンス行列から回転子の回転位置を推定する）（「突極性に基づく位置推定法を用いた位置センサレスＩＰＭモータ駆動システム」、小笠原悟司他、Ｔ．ＩＥＥ Ｊａｐａｎ，Ｖｏｌ．１１８−Ｄ，Ｎｏ．５，’９８参照）。
【０１２６】
前記速度制御部４においては、速度指令と回転速度とから発生すべきトルクを修正し、新たな電流位相指令および電流振幅指令を出力する。具体的には、（１）速度指令と回転速度とから電流振幅指令を算出し（ＰＩ制御などで、一般には回転速度が遅ければ電流指令増加方向に動かし）、（２）修正された電流振幅指令、修正前の電流位相指令から次式の演算を行ってトルクＴを算出して、ブラシレスＤＣモータ２を最大効率にするための電流位相を算出・出力し、（３）修正された電流振幅指令、電流位相指令から次式の演算を行ってトルクＴを算出して、ブラシレスＤＣモータ２を最大効率にするための電流位相を算出・出力し、（４）（３）の処理を適宜繰り返して行う。
Ｔ＝Ｐ_n｛φ＋（Ｌ_d−Ｌ_q）（−Ｉｓｉｎβ）｝Ｉｃｏｓβ
ここで、Ｐ_nは極対数、Ｉは電流振幅、βは電流位相である。
【０１２７】
もちろん、正確かつ高速にトルクを制御する必要がある場合には、電流位相を組み込んだトルク式から、逆問題を解いて電流振幅指令、電流位相指令を直接算出することもできる。
【０１２８】
前記電流制御部５においては、電流振幅指令、電流位相指令、回転子の回転位置（又は回転速度）から、次の演算を行って直接ＰＷＭ指令を生成することができる。
【０１２９】
回転座標系（ｄ、ｑ軸）で表したＩＰＭの電圧方程式は数１２で示される。
【０１３０】
【数１２】

ここで、Ｒ_aは巻線抵抗、ωは回転速度、Ｌ_dはｄ軸インダクタンス、Ｌ_qはｑ軸インダクタンス、Ｉ_d、Ｉ_qはそれぞれｄ軸、ｑ軸電流、Ｖ_d、Ｖ_qはそれぞれｄ軸、ｑ軸電圧、φは鎖交磁束数を表している。
【０１３１】
したがって、電流指令と回転速度とを用いてｄ軸、ｑ軸電圧Ｖ_d、Ｖ_qを算出できることが分かる。
【０１３２】
この時、電圧振幅Ｖ、および電圧位相δは、
Ｖ＝（Ｖ_d ²＋Ｖ_q ²）^1/2
δ＝ｔａｎ^-1（−Ｖ_d／Ｖ_q）
と表され、相電圧Ｖ_u（Ｖ_v、Ｖ_wも位相が１２０°ずつシフトしている他は同様）は、
Ｖ_u＝（２／３）^1/2Ｖｓｉｎ（θ＋δ）
と表される。
【０１３３】
上記の構成のブラシレスＤＣモータ制御装置を採用した場合には、回転子位置に従って最大効率になる電流位相でブラシレスＤＣモータにモータ電流を供給するため、負荷が変動する環境にあっても最大効率制御を行うことができるだけでなく、安定に最大効率制御を行うことができる。
【０１３４】
また、電流制御部５として、電流位相指令、電流振幅指令、回転子回転位置（回転速度）から直接ＰＷＭ信号を作成するものを採用し、回転子回転位置検出部として、固定子巻線の中性点の電圧と、固定子巻線の端子電圧の平均値とに基づいて回転しの回転位置を検出するものを採用し、検出された回転位置信号をブラシレスＤＣモータの運転条件に基づいて補正する（特願平１０−３１５０３５号参照）ことで回転子回転位置、回転速度を算出して制御を行えば、電流センサなどが不要となり、低コストに位置センサレスでの最大効率制御が実現できる。
【０１３５】
図２はこの発明のブラシレスＤＣモータ制御装置の他の実施態様を示すブロック図である。
【０１３６】
このブラシレスＤＣモータ制御装置は、直流電力を入力とするインバータ（電圧形インバータであることが好ましい）１１からの出力をブラシレスＤＣモータ１２に供給している。そして、回転子の回転位置θおよび回転速度ωを回転子回転位置検出部１３によって検出し、駆動波形生成部１４に供給している。また、負荷情報を入力とするトルク算出部１５によって算出されたトルクＴを電流位相決定部１６に供給して電流位相βを表す近似式の演算を行って電流位相βを決定し、駆動波形生成部１４に供給している。さらに、速度制御部１７から出力される速度指令値ω＊、およびインバータ１１における運転電流Ｉを駆動波形生成部１４に供給している。駆動波形生成部１４においては、これらの入力信号に基づいてＰＷＭ指令（ＰＷＭパターン信号）を生成し、インバータ１１に供給する。
【０１３７】
上記の構成のブラシレスＤＣモータ制御装置の作用は次のとおりである。
【０１３８】
先ず、負荷情報（例えば、空気調和機の場合には、室内外の現在温度や温度設定値など）からブラシレスＤＣモータ１２にかかるトルクＴをトルク算出部１５によって算出する。そして、算出されたトルクＴを電流位相決定部１６に供給して電流位相βを表す近似式の演算を行い、制御目標の電流位相βを決定する。
【０１３９】
また、ブラシレスＤＣモータ１２からの信号を用いて（例えば、固定子巻線の中性点の３次高調波を負荷条件に応じて補正して）ブラシレスＤＣモータ１２の回転子の現在の回転位置θを算出する。
【０１４０】
そして、決定された電流位相βと現在の回転位置θに加え、速度制御部１７からの速度指令値ω＊、およびインバータ１２における運転電流Ｉを駆動波形生成部１４に供給してブラシレスＤＣモータ１２を駆動するためのＰＷＭパターン信号を生成してインバータ１１に供給する。インバータ１１においては、ＰＷＭパターン信号に基づいてＩＧＢＴなどのパワーデバイスを制御し、ブラシレスＤＣモータ１２を駆動する。
【０１４１】
ここで、ある回転数（回転速度）でのＩＰＭのトルク毎の最大効率となる電流位相は、例えば、図３に示すとおりである。また、回転数毎に最大効率になる電流位相特性は、例えば、図４に示すとおりである。図３、図４から、電流位相が２０°から４０°付近で最大効率になることが分かる。そこで、この最大効率位相βをトルクＴの関数｛例えば、近似式β＝Ａ・Ｔ＋Ｂ（Ａ、Ｂは定数）｝で表し、この関数を電流位相決定部１６に組み込んでおくことにより、簡単に最大効率位相βを得て、駆動波形の生成に反映させることができる。
【０１４２】
したがって、図２のブラシレスＤＣモータ制御装置を採用すれば、最大効率位相βを簡単に算出することができ、この最大効率位相βを制御目標としてブラシレスＤＣモータを制御することによって、ブラシレスＤＣモータの最大効率制御を達成することができる。
【０１４３】
図５はこの発明のブラシレスＤＣモータ制御装置のさらに他の実施態様を示すブロック図である。
【０１４４】
このブラシレスＤＣモータ制御装置が図２のブラシレスＤＣモータ制御装置と異なる点は、電流位相決定部１６に代えて、トルクＴに対応させて電流位相βを格納した最大効率位相テーブル１８を採用した点のみである。
【０１４５】
この最大効率位相テーブル１８の内容の一例を表１に示す。
【０１４６】
【表１】

もちろん、この最大効率位相テーブル１８の内容は、図３に示すように予め求められた最大効率位相特性に基づいて設定される。ここで、最大効率位相テーブル１８の分解能はメモリの容量を考慮して設定すればよく、メモリの容量が少ない場合はトルクＴの分解能を粗くし、それぞれの点の間を補間すればよい。
【０１４７】
上記の構成のブラシレスＤＣモータ制御装置の作用は次のとおりである。
【０１４８】
先ず、負荷情報（例えば、空気調和機の場合には、室内外の現在温度や温度設定値など）からブラシレスＤＣモータ１２にかかるトルクＴをトルク算出部１５によって算出する。そして、算出されたトルクＴを最大効率位相テーブル１８に供給して、制御目標の電流位相βを決定する。
【０１４９】
また、ブラシレスＤＣモータ１２からの信号を用いて（例えば、固定子巻線の中性点の３次高調波を負荷条件に応じて補正して）ブラシレスＤＣモータ１２の回転子の現在の回転位置θを算出する。
【０１５０】
そして、決定された電流位相βと現在の回転位置θに加え、速度制御部１７からの速度指令値ω＊、およびインバータ１２における運転電流Ｉを駆動波形生成部１４に供給してブラシレスＤＣモータ１２を駆動するためのＰＷＭパターン信号を生成してインバータ１１に供給する。インバータ１１においては、ＰＷＭパターン信号に基づいてＩＧＢＴなどのパワーデバイスを制御し、ブラシレスＤＣモータ１２を駆動する。
【０１５１】
図６はこの発明のブラシレスＤＣモータ制御装置のさらに他の実施態様を示すブロック図である。
【０１５２】
このブラシレスＤＣモータ制御装置が図２のブラシレスＤＣモータ制御装置と異なる点は、トルク算出部１５および電流位相決定部１６に代えて、電流位相設定部１９を設けた点のみである。
【０１５３】
前記電流位相設定部１９においては、ブラシレスＤＣモータ１２の運転状態に拘わらず予め設定された所定の電流位相βを出力するものである。ここで、電流位相βを一定に保持すれば、トルクの変動、回転数の変動などが生じた場合に、必ずしも最大効率位相を実現できなくなってしまう可能性がある。しかし、図３、図４を参照すれば分かるように、トルクの変動、回転数の変動などが生じても最大効率位相の変動は余り大きくないのであるから、所定の電流位相βを図３、図４などを参照して設定することにより、最大効率位相を実現した場合と比較してほぼ同程度の効率でブラシレスＤＣモータ１２を運転することができる（図７参照）。
【０１５４】
上記の各実施態様を採用した場合には、モータ効率を検出する手段、例えば、電流センサや電圧センサなどの部品を追加することなく低コストで、ブラシレスＤＣモータを最大効率制御することができる。特に、図５、図６の実施態様を採用した場合には、リアルタイムに最大効率位相を算出する必要がなく、低コストなマイコンで制御を実現することができる。
【０１５５】
図１２はこの発明のブラシレスＤＣモータ制御装置のさらに他の実施態様を示すブロック図である。
【０１５６】
このブラシレスＤＣモータ制御装置は、交流電源２１を入力とするコンバータ２２と、コンバータ２２の出力電圧を平滑化する平滑コンデンサ２３と、平滑化された直流電圧を入力とするインバータ２４と、インバータ２４から出力される交流電圧が印加されるブラシレスＤＣモータ２５と、ブラシレスＤＣモータ２５への供給電流を検出する電流検出部２６と、ブラシレスＤＣモータ２５に印加される電圧を検出する電圧検出部２７と、検出電流および検出電圧を入力として磁束Ψ＝（Ｌ_ｄ−Ｌ_ｑ）ｉ_ｄ＋φ、回転子位置および速度を検出する位置・速度検出部２８と、検出された磁束Ψを入力として磁束Ψ≒０であることを検出して進相指令を出力するΨ≒０検出部２９と、検出された速度および外部から与えられる速度指令を入力として速度制御演算を行って電流指令を出力する速度制御部３０と、電流指令、外部から与えられる位相指令、および進相指令を入力として位相制御演算を行って位相を出力する位相制御部３１と、位相制御部３１から出力される位相、検出電流および回転子位置を入力として電流制御演算を行って電圧指令を出力し、インバータ２４に供給する電圧制御部３２とを有している。
【０１５７】
前記位置・速度検出部２８にはモータの機器定数が予め設定されており、この機器定数とモータ電流（供給電流）とモータ電圧（印加電圧）とを用いて磁束情報を算出し、算出された磁束情報から回転子位置を推定して回転子位置を出力するとともに、モータ電流の周期などから速度を検出するものである。
【０１５８】
上記の構成のブラシレスＤＣモータ制御装置を採用した場合には、Ψ≒０検出部２９が磁束Ψ≒０であることを検出していないことを条件として、位相制御部３１が位相指令および電流指令に基づいて位相制御演算を行うので、モータ電流とモータ電圧と機器定数とを用いて回転子位置を推定し、推定された回転子位置に基づいてブラシレスＤＣモータ２５を制御することができる。
【０１５９】
逆に、Ψ≒０検出部２９が磁束Ψ≒０であることを検出した場合には、進相指令を出力するのであるから、位相指令および電流指令に基づいて位相制御演算を行って得られる位相よりも進められた位相を位相制御部３１から出力し、磁束Ψ＝０を避けることによって、回転子位置の正確な推定を可能とし、ひいてはブラシレスＤＣモータ２５を安定に動作させることができる（脱調を防止することができる）。
【０１６０】
なお、図１２の実施態様においては、位置・速度検出部２８において磁束Ψを算出するようにしているが、別途磁束Ψを算出するΨ算出部を設けてもよく、Ψ＝０と等価なモータ電流値、モータ電流位相などを算出して用いてもよい（図１１参照）。また、電流制御系を設けることなく、直接電圧制御を行ってもよい。さらに、モータ電圧を検出する代わりにインバータ指令からモータ電圧を算出するようにしてもよい。
【０１６１】
図１３はこの発明のブラシレスＤＣモータ制御装置のさらに他の実施態様を示すブロック図である。
【０１６２】
このブラシレスＤＣモータ制御装置が図１２のブラシレスＤＣモータ制御装置と異なる点は、進相指令を出力するΨ≒０検出部２９に代えて、電流垂下指令を出力して電流制御部３２に供給するΨ≒０検出部２９’を採用した点のみである。
【０１６３】
したがって、この実施態様においては、Ψ≒０検出部２９が磁束Ψ≒０であることを検出した場合に、電流垂下指令に基づいてインバータ出力電流を減少させ、磁束Ψ＝０を避けることによって、回転子位置の正確な推定を可能とし、ひいてはブラシレスＤＣモータ２５を安定に動作させることができる（脱調を防止することができる）。
【０１６４】
図１４はこの発明のブラシレスＤＣモータ制御装置のさらに他の実施態様を示すブロック図である。
【０１６５】
このブラシレスＤＣモータ制御装置が図１２のブラシレスＤＣモータ制御装置と異なる点は、進相指令を出力するΨ≒０検出部２９に代えて、Ψ＞０の電流値、電流位相を予め格納したΨ＞０テーブル２９’’を設け、速度制御部３０から出力される電流指令および外部から与えられる位相指令を読み出し指示信号として電流位相を出力し、位相制御部３１に供給した点のみである。
【０１６６】
したがって、この実施態様においては、Ψ＞０となるような電流位相をΨ＞０テーブル２９’’から読み出して位相制御部３１に供給することによって、磁束Ψ＝０を避け、ひいては、回転子位置の正確な推定を可能とし、ひいてはブラシレスＤＣモータ２５を安定に動作させることができる（脱調を防止することができる）。
【０１６７】
なお、Ψ＞０テーブル２９’’を設ける代わりに、Ψ＞０となるような位相リミット部を設けるようにしてもよい。
【０１６８】
図１５はこの発明のブラシレスＤＣモータ制御装置のさらに他の実施態様を示すブロック図である。
【０１６９】
このブラシレスＤＣモータ制御装置は、直流電力を入力とするインバータ４１と、インバータ４１から出力される交流電圧が印加されるブラシレスＤＣモータ４２と、３相分のモータ電流を検出する電流検出部４３と、３相分のモータ電流を入力として２相電流（αβ軸電流）への変換を行う３相→２相変換部４４と、変換された２相電流（αβ軸電流）および回転子角度（θ）を入力としてｄｑ軸電流への変換を行うαβ軸→ｄｑ軸変換部４５と、外部から与えられる位相指令および電流指令を入力として−ｓｉｎ処理を施してｄ軸電流指令を出力する−ｓｉｎ処理部４６と、外部から与えられる位相指令および電流指令を入力としてｃｏｓ処理を施してｑ軸電流指令を出力するｃｏｓ処理部４７と、−ｓｉｎ処理部４６から出力されるｄ軸電流とαβ軸→ｄｑ軸変換部４５から出力されるｄ軸電流との差分を算出する第１差算出部４８と、ｃｏｓ処理部４７から出力されるｑ軸電流とαβ軸→ｄｑ軸変換部４５から出力されるｑ軸電流との差分を算出する第２差算出部４９と、第１差算出部４８から出力されるｄ軸差電流に対して比例ゲインＫ_ｄｐを乗算する比例処理部４８ａと、、第１差算出部４８から出力されるｄ軸差電流に対して積分ゲインＫ_ｄｉを乗算する積分乗算部４８ｂと、積分乗算部４８ｂからの出力に対して積分処理（１／ｓ処理）を施す積分処理部４８ｃと、比例処理部４８ａからの出力と積分処理部４８ｃからの出力とを加算してｄ軸電圧指令として出力する加算部４８ｄと、第２差算出部４９から出力されるｑ軸差電流に対して比例ゲインＫ_ｑｐを乗算する比例処理部４９ａと、、第２差算出部４９から出力されるｑ軸差電流に対して積分ゲインＫ_ｑｉを乗算する積分乗算部４９ｂと、積分乗算部４９ｂからの出力に対して積分処理（１／ｓ処理）を施す積分処理部４９ｃと、比例処理部４９ａからの出力と積分処理部４９ｃからの出力とを加算してｑ軸電圧指令として出力する加算部４９ｄと、ｄ軸電圧、ｑ軸電圧および回転子角度（θ）を入力としてαβ軸電圧への変換を行うｄｑ軸→αβ軸変換部５０と、αβ軸電圧を入力として３相電圧指令に変換してインバータ４１に供給する２相→３相変換部５１とを有している。
【０１７０】
なお、前記回転子角度（θ）を推定する部分については図示を省略しているが、前述のように、モータ電流、モータ電圧および機器定数から推定するようにしている。
【０１７１】
この実施態様のブラシレスＤＣモータ制御装置を採用した場合には、目標とする電流位相になるように電流位相を制御してブラシレスＤＣモータを制御することができる。したがって、目標とする電流位相を磁束Ψ＝０とならないように設定することによって、ブラシレスＤＣモータを常に安定に制御することができる。
【０１７２】
図１６はこの発明のブラシレスＤＣモータ制御装置のさらに他の実施態様を示すブロック図である。
【０１７３】
このブラシレスＤＣモータ制御装置は、直流電力を入力とするインバータ６１と、インバータ６１から出力される交流電圧が印加されるブラシレスＤＣモータ６２と、３相分のモータ電流を検出する電流検出部６３と、３相分のモータ電流を入力としてｒ、θ変換を行って電流ベクトル長および電流位相を出力する３相→ｒ、θ変換部６４と、電流位相と回転子角度（θ）との差を算出する第１位相差算出部６５と、外部から与えられる電流指令と電流ベクトル長との差を算出する電流差算出部６６と、外部から与えられる位相指令と第１位相差算出部６５から出力される差との差を算出する第２位相差算出部６７と、電流差算出部６６からの出力に対して比例ゲインＫ_ｌｐを乗算する比例処理部６６ａと、電流差算出部６６からの出力に対して積分ゲインＫ_ｌｉを乗算する積分乗算部６６ｂと、積分乗算部６６ｂからの出力に対して積分処理を施す積分処理部６６ｃと、比例処理部６６ａからの出力と積分処理部６６ｃからの出力とを加算して電流ベクトル長として出力する加算部６６ｄと、第２位相差算出部６７からの出力に対して比例ゲインＫ_θｐを乗算する比例処理部６７ａと、第２位相差算出部６７からの出力に対して積分ゲインＫ_θｉを乗算する積分乗算部６７ｂと、積分乗算部６７ｂからの出力に対して積分処理を施す積分処理部６７ｃと、比例処理部６７ａからの出力と積分処理部６７ｃからの出力とを加算して出力する加算部６７ｄと、加算部６７ｄからの出力と回転子角度（θ）とを加算して電圧位相として出力する加算部６８と、加算部６６ｄから出力される電圧ベクトル長および加算部６８から出力される電圧位相を入力として３相変換処理を行って３相分の電圧指令に変換してインバータ６１に供給するｒ、θ→３相変換部６９とを有している。
【０１７４】
なお、前記回転子角度（θ）を推定する部分については図示を省略しているが、前述のように、モータ電流、モータ電圧および機器定数から推定するようにしている。
【０１７５】
この実施態様のブラシレスＤＣモータ制御装置を採用した場合にも、目標とする電流位相になるように電流位相を制御してブラシレスＤＣモータを制御することができる。したがって、目標とする電流位相を磁束Ψ＝０とならないように設定することによって、ブラシレスＤＣモータを常に安定に制御することができる。
【０１７６】
図１７は前記位置・速度検出部２８の構成の一例を示す示すブロック図である。
【０１７７】
この位置・速度検出部２８は、３相電圧を入力としてγδ変換を行ってγδ電圧ベクトルを出力する３相→γδ変換部２８ａと、３相電流を入力としてγδ変換を行ってγδ電流ベクトルを出力する３相→γδ変換部２８ｂと、γδ電流ベクトルを入力として電圧ベクトルを出力するモータ逆モデル部２８ｃと、モータ逆モデル部２８ｃから出力される電圧ベクトルと３相→γδ変換部２８ａから出力される電圧ベクトルとの差を算出する差算出部２８ｄと、差算出部２８ｄから出力される差を入力とするフィルタ２８ｅと、フィルタ２８ｅからの出力を入力として位置・速度の推定を行う位置・速度推定部２８ｆとを有している。
【０１７８】
したがって、この場合には、回転座標モータモデルを用いて回転子位置を検出することができる。ただし、電流、角速度の変動に対し、角度誤差が打ち消される可能性があるので、変動の大きな用途には不向きであるが、変動の少ない用途に好適に適用することができる。
【０１７９】
また、図８の座標変換部をテイラー１次線形近似して次式を得る。
Ｔ＝ｉ＋Ｊθ_ｅ、ただし、数１３である。
【０１８０】
【数１３】

上式を図８のブロック線図に代入して図９のブロック線図を得、これから数１４の閉ループ系の一巡伝達関数（一巡ループ伝達関数）Ｇ（ｓ）が得られる。
【０１８１】
【数１４】

そして、Ｇ／（１＋Ｇ）の分母多項式のｓの実根が全て負となるように各部のゲインＫ_θとＦ（ｓ）を設定することにより安定性を確保することができる。
【０１８２】
次いで、ゲイン設定の手順を説明する。
（１）安定に運転したい電流位相、電流振幅、速度の範囲を設定する。
【０１８３】
ゲインＫ_θとＦ（ｓ）、ループ中のゲイン初期値を設定する。
（２）（１）で設定される各種諸量の全組み合わせにおいて、前記有理関数Ｇを用いて、Ｇ／（１＋Ｇ）の分母多項式のｓの実根が全て負がどうかチェックする。
（３）正の実根が現れる場合にはゲインを変化させて（２）を実行する。
【０１８４】
逆に、正の実根が現れない場合は、その値を採用する。
【０１８５】
図９の線形近似ブロック線図より、（２）の結果、実根を全て負とするゲインの組み合わせは、安定なゲインである。
【０１８６】
具体的には、「速度起電力推定に基づくセンサレス突極形ブラシレスＤＣモータ制御」、竹下他、Ｔ．ＩＥＥ Ｊａｐａｎ，Ｖｏｌ．１１７−Ｄ，Ｎｏ．１，’９７の構成の伝達関数Ｇは数１５となる。
【０１８７】
【数１５】

そして、上記の文献と同じモータ機器定数を採用し、例えば、速度１５ｒｐｓ、電流振幅０．８４Ａ、電流位相０〜３６０°を運転範囲としてゲインを設定する。この時、Ｋ_θ＝１０７７、Ｋ_e＝５２０２０００を設定すると、従来の安定性判定では安定として求まるが、上記の判定法を用いると、図１８中（Ａ）に示すように、電流位相がおよそ１３０°〜２３０°で不安定となることが分かる。
【０１８８】
ここで、Ｋ_θ＝１０７．７、Ｋ_e＝５２０２００を設定すると、全ての電流位相で安定となることがわかり｛図１８中（Ｂ）参照｝、この運転範囲を満足するためにはこのようなゲイン設定が必要であることが分かる。
【０１８９】
また、前記位置・速度検出部２８として、図１０に示すような固定座標方式位置検出器を採用することもできる。
【０１９０】
この場合には、電流や角速度の変化による位置推定誤差がなく、電流や角速度の変化が大きい場合であっても好適なブラシレスＤＣモータの制御を達成することができる。
【０１９１】
さらに、上記の実施態様において採用されるモータ機器定数は、予め設定されていてもよいが、運転開始時に同定処理を行ってモータ機器定数を設定し、その後にブラシレスＤＣモータの制御を行うことが好ましい。なお、同定処理は、例えば、次のようにして行う。
ステップ１
強制動機運転または位置センサを用いない運転などを行い、速度がほぼ一定、電圧と電流がほぼ一定の条件におけるｖ_αβ、ｉ_αβを検出し、または推定し、数１６の演算を行う。
【０１９２】
【数１６】

ただし、ｎは１以上の整数である。
ステップ２
次いで、速度がほぼ一定、電圧と電流がほぼ一定の条件下において、速度、電圧、電流の１つ以上が前記と異なる条件におけるｖ_αβ、ｉ_αβを検出し、または推定し、数１７の演算を行う。
【０１９３】
【数１７】

ただし、ｎは１以上の整数である。
【０１９４】
電圧と電流の基本波成分以外の成分が無視できると仮定し、数１８を得る。
【０１９５】
【数１８】

ステップ３
そして、同期モータの固定子座標上の電圧方程式
ｖ_αβ＝Ｒｉ_αβ＋ｓ｛Ｌ_pｉ_αβ＋Ｌ_mｆ（２θ）ｉ_αβ｝＋ｓ｛φΘ｝
に数１８を代入し、互いに直交する変数である数１９、数２０のそれぞれに関する係数を抜き出し、数２１を得る。
【０１９６】
【数１９】

【０１９７】
【数２０】

【０１９８】
【数２１】

ステップ４
前記ステップ１、ステップ２におけるθ_eをそれぞれθ_e1、θ_e2とおき、数１６、数１７を数２１に代入し、８つの未知定数をもつ８連立方程式を得る。機器定数の初期値として予め粗設定しておいた値、θ_e1、θ_e2の初期値として０を設定し、得られた方程式をニュートン法などにより解き、機器定数を得ることができる（同定することができる）。
【０１９９】
さらにまた、前記実施態様における目標電流位相をブラシレスＤＣモータの最高効率位相または最大トルク位相以上に進めることが好ましい。
【０２００】
最大効率電流位相−トルク特性を示す図１９、電流位相−トルクに対するモータ端子電圧を示す図２０、および最大トルク電流位相を示す図２１を参照すれば、電流位相を進めるほどモータ端子電圧が下がることが分かる。したがって、インバータ出力電圧がリミットに達した場合にも、電流位相を進めることによってさらに高速回転を行わせることができる。換言すれば、運転範囲を高速側に拡大することができる。
【０２０１】
次いで、ブラシレスＤＣモータによって圧縮機を駆動する場合を考える。
【０２０２】
１シリンダロータリー圧縮機における回転角度に対する圧縮トルクを示す図２２を参照すれば、空調用圧縮機においては負荷変動が大きく、かつ高速であるから、（Ｌ_ｄ−Ｌ_ｑ）ｉ_ｄ＋φ＞０を満足させることは容易でないが、高速な電流位相制御により安定に制御することが可能である。
【０２０３】
したがって、上記の実施態様のブラシレスＤＣモータ制御装置を採用することによって、モータモデルを用いる方法による安定、かつ緻密な制御を圧縮機においても実現することができ、省エネルギー、高効率化などの効果を達成することができる。
【０２０６】
【発明の効果】
請求項１の発明は、位置センサを用いることなく回転子の回転位置を検出することができるとともに、目標電流位相を測定する必要がなく、処理を簡単化することができ、しかも最大効率制御を達成することができるという特有の効果を奏する。
【０２０８】
請求項２の発明は、ブラシレスＤＣモータの固定子の中性点の電圧と、固定子巻線の端子電圧の平均値とに基づいて回転子の回転位置を検出する場合に適用することができ、最大効率制御を達成することができるという特有の効果を奏する。
【０２０９】
請求項３の発明は、固定子巻線の印加電圧、モータ電流、およびブラシレスＤＣモータの機器定数を用いて所定の演算を行って回転子の回転位置を算出する場合に適用することができ、最大効率制御を達成することができるという特有の効果を奏する。
【０２１０】
請求項４の発明は、電圧型インバータが発生する高調波電流から求められたインダクタンスおよび回転子の突極性から回転子の回転位置を算出する場合に適用することができ、最大効率制御を達成することができるという特有の効果を奏する。
【０２１４】
請求項５の発明は、位置センサを用いることなく回転子の回転位置を検出することができるとともに、目標電流位相を測定する必要がなく、処理を簡単化することができ、しかも最大効率制御を達成することができるという特有の効果を奏する。
【０２１６】
請求項６の発明は、ブラシレスＤＣモータの固定子の中性点の電圧と、固定子巻線の端子電圧の平均値とに基づいて回転子の回転位置を検出する場合に適用することができ、最大効率制御を達成することができるという特有の効果を奏する。
【０２１７】
請求項７の発明は、固定子巻線の印加電圧、モータ電流、およびブラシレスＤＣモータの機器定数を用いて所定の演算を行って回転子の回転位置を算出する場合に適用することができ、最大効率制御を達成することができるという特有の効果を奏する。
【０２１８】
請求項８の発明は、電圧型インバータが発生する高調波電流から求められたインダクタンスおよび回転子の突極性から回転子の回転位置を算出する場合に適用することができ、最大効率制御を達成することができるという特有の効果を奏する。
【０２１９】
請求項９の発明は、モータの構造に左右されることなく適用範囲を拡大することができ、しかも、回転子の回転位置を常に正確に推定してブラシレスＤＣモータを安定に制御することができ、さらに、位置推定不能による脱調を避けることができるという特有の効果を奏する。
【０２２１】
請求項１０の発明は、請求項９と同様の効果を奏する。
【０２２２】
請求項１１の発明は、請求項９と同様の効果を奏する。
【０２２４】
請求項１２の発明は、請求項９から請求項１１の何れかの効果に加え、電流、角速度の変動が小さい用途に好適に適用することができるという特有の効果を奏する。
【０２２５】
請求項１３の発明は、回転子の回転位置を常に正確に推定してブラシレスＤＣモータを安定に制御することができるという特有の効果を奏する。
【０２２６】
請求項１４の発明は、請求項１３と同様の効果を奏する。
【０２２７】
請求項１５の発明は、請求項９から請求項１１の何れかの効果に加え、電流や角速度変化に拘わらず正確に回転子の回転位置の推定を行うことができるという特有の効果を奏する。
【０２２８】
請求項１６の発明は、請求項９から請求項１５の何れかの効果に加え、正確に回転子の回転位置の推定を行うことができるという特有の効果を奏する。
【０２２９】
請求項１７の発明は、請求項９から請求項１６の何れかの効果に加え、ブラシレスＤＣモータの運転範囲を高速側に拡大することができるという特有の効果を奏する。
【０２３０】
請求項１８の発明は、請求項９から請求項１７の何れかの効果に加え、負荷変動が大きく、かつ高速動作させる必要がある圧縮機を安定に駆動することができるという特有の効果を奏する。
【０２３１】
請求項１９の発明は、モータの構造に左右されることなく適用範囲を拡大することができ、しかも、回転子の回転位置を常に正確に推定してブラシレスＤＣモータを安定に制御することができ、さらに、位置推定不能による脱調を避けることができるという特有の効果を奏する。
【０２３３】
請求項２０の発明は、請求項１９と同様の効果を奏する。
【０２３４】
請求項２１の発明は、請求項１９と同様の効果を奏する。
【０２３６】
請求項２２の発明は、請求項１９から請求項２１の何れかの効果に加え、電流、角速度の変動が小さい用途に好適に適用することができるという特有の効果を奏する。
【０２３７】
請求項２３の発明は、回転子の回転位置を常に正確に推定してブラシレスＤＣモータを安定に制御することができるという特有の効果を奏する。
【０２３８】
請求項２４の発明は、請求項２３と同様の効果を奏する。
【０２３９】
請求項２５の発明は、請求項１９から請求項２１の何れかの効果に加え、電流や角速度変化に拘わらず正確に回転子の回転位置の推定を行うことができるという特有の効果を奏する。
【０２４０】
請求項２６の発明は、請求項１９から請求項２５の何れかの効果に加え、正確に回転子の回転位置の推定を行うことができるという特有の効果を奏する。
【０２４１】
請求項２７の発明は、請求項１９から請求項２６の何れかの効果に加え、ブラシレスＤＣモータの運転範囲を高速側に拡大することができるという特有の効果を奏する。
【０２４２】
請求項２８の発明は、請求項１９から請求項２７の何れかの効果に加え、負荷変動が大きく、かつ高速動作させる必要がある圧縮機を安定に駆動することができるという特有の効果を奏する。
【図面の簡単な説明】
【図１】この発明のブラシレスＤＣモータ制御装置の一実施態様を示すブロック図である。
【図２】この発明のブラシレスＤＣモータ制御装置の他の実施態様を示すブロック図である。
【図３】埋込磁石モータの電流位相特性の一例を示す図である。
【図４】埋込磁石モータの回転数毎の最大効率電流位相を示す図である。
【図５】この発明のブラシレスＤＣモータ制御装置のさらに他の実施態様を示すブロック図である。
【図６】この発明のブラシレスＤＣモータ制御装置のさらに他の実施態様を示すブロック図である。
【図７】電流位相を一定に保持した場合における効率の変化を示す図である。
【図８】回転座標モデルに基づく位置検出部を示すブロック図である。
【図９】図８を線形近似したブロック図である。
【図１０】固定座標モデルに基づく位置検出部を含むモータ制御装置を示すブロック図である。
【図１１】ｄ軸を基準とした電流位相と２相変換後の電流値Ｉ_ａに対するΨ＝（Ｌ_ｄ−Ｌ_ｑ）ｉ_ｄ＋φのグラフである。
【図１２】この発明のブラシレスＤＣモータ制御装置のさらに他の実施態様を示すブロック図である。
【図１３】この発明のブラシレスＤＣモータ制御装置のさらに他の実施態様を示すブロック図である。
【図１４】この発明のブラシレスＤＣモータ制御装置のさらに他の実施態様を示すブロック図である。
【図１５】この発明のブラシレスＤＣモータ制御装置のさらに他の実施態様を示すブロック図である。
【図１６】この発明のブラシレスＤＣモータ制御装置のさらに他の実施態様を示すブロック図である。
【図１７】位置・速度検出部の構成を示すブロック図である。
【図１８】閉ループ極の実部最大値−電流位相特性を示す図である。
【図１９】最大効率電流位相−トルク特性を示す図である。
【図２０】電流位相−トルクに対するモータ端子電圧を示す図である。
【図２１】最大トルク電流位相を示す図である。
【図２２】１シリンダロータリー圧縮機のトルクパターンを示す図である。
【符号の説明】
１、１１、２４、４１、６１ インバータ
２、１２、２５、４２、６２ ブラシレスＤＣモータ
３、１３ 回転子回転位置検出部 ４ 速度制御部
１４ 駆動波形生成部 ２９、２９’ Ψ≒０検出部
２９’’ Ψ＞０テーブル[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a brushless DC motor control method and apparatus, and more particularly to a method and apparatus for operating a brushless DC motor with maximum efficiency without using a position sensor such as an encoder.
[0002]
[Prior art]
Conventionally, it is known that it is essential to detect the rotational position of a rotor in controlling a brushless DC motor. As a method for detecting the rotational position of the rotor, a method using a rotary encoder, a method using a Hall element, etc., and a method for detecting a zero cross of an induced voltage while employing a 120 ° energization waveform are known. .
[0003]
Among these methods, the method using a rotary encoder causes a significant increase in the cost of the entire apparatus. Therefore, the method using a rotary encoder is applied to a device with a strong demand for cost reduction such as a household electric appliance using a brushless DC motor as a power source. I can't. In addition, a method using a Hall element or the like is extremely difficult or impossible to incorporate into a device that generates a severe environment such as a high temperature and high pressure such as a compressor.
[0004]
Unlike these, the 120 ° energization waveform is adopted, and the method of detecting the zero cross of the induced voltage is easily applied even for applications where it is difficult to apply a method using a rotary encoder or a method using a Hall element. can do.
[0005]
Further, conventionally, the following control method has been employed for controlling the brushless DC motor based on the rotational position detected as described above.
[0006]
When controlling a brushless DC motor (hereinafter abbreviated as a surface magnet motor) that has a permanent magnet mounted on the surface of the rotor, a reluctance torque is not generated, so a control method that sets the d-axis current id to 0 is adopted. Has been.
[0007]
In controlling a brushless DC motor (hereinafter abbreviated as an embedded magnet motor or IPM) having a permanent magnet mounted inside the rotor, reluctance torque is generated._dA control method for advancing the current phase larger than = 0 is employed.
[0008]
[Problems to be solved by the invention]
However, when controlling the embedded magnet motor by detecting the rotational position without using a sensor, in general, 120 ° energization is adopted, and the zero position of the induced voltage is detected to detect the rotational position. Therefore, the desired current phase β cannot be obtained.
[0009]
In addition, as a conventional control method, a method for detecting the rotational position using the third harmonic, and controlling the maximum efficiency of the embedded magnet motor while monitoring the supplied power, and the amplitude of the third harmonic are monitored. However, a method for controlling the maximum efficiency of the embedded magnet motor has been proposed.
[0010]
However, when the former method is adopted, maximum efficiency control cannot be performed in an environment where the load fluctuates, and when the latter method is adopted, it is caused by inaccuracy of the third harmonic. Thus, accurate phase control cannot be performed, and thus maximum efficiency control cannot be achieved.
[0011]
Further, in any of the above-described methods, the detection of the rotational position is discrete and precise control cannot be performed, and the detected rotational position is likely to deviate from the actual rotational position depending on the operating state of the motor. is there.
[0012]
OBJECT OF THE INVENTION
The present invention has been made in view of the above-described problems, and an object thereof is to provide a brushless DC motor control method and apparatus capable of controlling the maximum efficiency of an embedded magnet motor without using a rotational position sensor. Yes.
[0015]
[Means for Solving the Problems]
  In the brushless DC motor control method according to claim 1, in detecting and controlling the rotational position of the rotor without using a rotational position sensor, the brushless DC motor having a permanent magnet mounted inside the rotor is controlled.
  In order to achieve maximum efficiency control, the current phase is controlled, and the current phase is calculated based on an equation that approximates the current phase that can be achieved for maximum efficiency, obtained in advance for each operating state of the brushless DC motor. This is a method of setting the target current phase.
[0017]
  The brushless DC motor control method according to claim 2 detects the rotational position of the rotor based on the voltage at the neutral point of the stator of the brushless DC motor and the average value of the terminal voltages of the stator windings. The position signal is corrected based on the operating condition of the brushless DC motor, and the current phase is controlled based on the corrected position signal.
[0018]
  The brushless DC motor control method according to claim 3 calculates the rotational position of the rotor by performing a predetermined calculation using the applied voltage of the stator winding, the motor current, and the device constant of the brushless DC motor. In this method, the current phase is controlled based on the rotational position.
[0019]
  The brushless DC motor control method according to claim 4 calculates the rotational position of the rotor from the inductance obtained from the harmonic current generated by the voltage-type inverter and the saliency of the rotor, and the current based on the calculated rotational position. This is a method for controlling the phase.
[0023]
  The brushless DC motor control device according to claim 5 detects and controls the rotational position of the rotor without using a rotational position sensor for a brushless DC motor in which a permanent magnet is mounted inside the rotor.
  In order to achieve maximum efficiency control, the current phase is controlled, and the current phase is calculated based on an equation that approximates the current phase that can be achieved for maximum efficiency, obtained in advance for each operating state of the brushless DC motor. Current phase control means for setting the target current phase is included.
[0025]
The brushless DC motor control device according to claim 6 detects a rotational position of the rotor based on a neutral point voltage of the stator of the brushless DC motor and an average value of the terminal voltages of the stator windings. And a correction means for correcting the detected position signal based on the operating condition of the brushless DC motor, and the current phase control means adopts a means for controlling the current phase on the basis of the corrected position signal. To do.
[0026]
  The brushless DC motor control device according to claim 7 is a rotational position calculating means for performing a predetermined calculation using the applied voltage of the stator winding, the motor current, and the device constant of the brushless DC motor to calculate the rotational position of the rotor. Further, the current phase control means that controls the current phase based on the calculated rotational position is adopted.
[0027]
  The brushless DC motor control device according to claim 8 further includes rotational position calculation means for calculating a rotor rotational position from an inductance obtained from a harmonic current generated by the voltage type inverter and a saliency of the rotor, and the current phase As the control means, one that controls the current phase based on the calculated rotational position is adopted.
[0030]
  The brushless DC motor control method according to claim 10 is a region in which the magnetic flux is almost zero and the phase advance control is performed in response to detecting that the magnetic flux is almost zero, so that the magnetic flux becomes almost zero. This is how to avoid it.
[0031]
  The brushless DC motor control method according to claim 11 is a region in which the magnetic flux is almost zero, and the current drooping control is performed in response to detecting that the magnetic flux is almost zero, so that the magnetic flux becomes almost zero. This is how to avoid it.
[0033]
  A brushless DC motor control method according to a twelfth aspect of the present invention is a method for estimating the rotational position of the rotor using a rotational coordinate motor model.
[0034]
  The brushless DC motor control method according to claim 13 estimates the rotational position of the rotor by using the motor voltage, the motor current, and the device constants of the brushless DC motor, controls the inverter using the estimation result, and the brushless DC motor. In controlling
  This is a method of setting the gain of the loop transfer function relating to the rotational position estimation to a value that satisfies the stability condition in the target phase and the torque range.
[0035]
  In the brushless DC motor control method according to claim 14, the transfer function G is expressed by Formula 1, and the gains Kθ and F (s of the transfer function G are set so that all real roots of s of the denominator polynomial of G / (1 + G) are all negative. ).
[0036]
  A brushless DC motor control method according to a fifteenth aspect of the present invention is a method for estimating the rotational position of the rotor using a fixed coordinate motor model.
[0037]
  A brushless DC motor control method according to a sixteenth aspect is a method of identifying motor device constants before starting operation.
[0038]
  The brushless DC motor control method according to claim 17 is a method of advancing the target current phase to a maximum efficiency phase or a maximum torque phase of the motor.
[0039]
  The brushless DC motor control method according to claim 18 is a method that employs a brushless DC motor that drives a compressor.
[0040]
  The brushless DC motor control device according to claim 19 estimates the rotational position of the rotor from the magnetic flux information using the motor voltage, the motor current, and the device constant of the brushless DC motor, and controls the inverter using the estimation result, It controls a brushless DC motor with saliency,
  Inverter control means for controlling the inverter to change the operating point in response to the magnetic flux being substantially zero is included.
[0042]
  The brushless DC motor control device according to claim 20 detects, as the inverter control means, that the magnetic flux is substantially zero, and performs phase advance control in response to detecting that the magnetic flux is substantially zero. The one that avoids the region where the value of N is almost 0 is adopted.
[0043]
  The brushless DC motor control device according to claim 21 detects, as the inverter control means, that the magnetic flux is substantially zero, and performs current droop control in response to detecting that the magnetic flux is substantially zero, whereby the magnetic flux The one that avoids the region where the value of N is almost 0 is adopted.
[0045]
  A brushless DC motor control device according to a twenty-second aspect employs, as the inverter control means, one that estimates a rotational position of a rotor using a rotational coordinate motor model.
[0046]
  The brushless DC motor control device according to claim 23 estimates the rotational position of the rotor using the motor voltage, the motor current, and the device constants of the brushless DC motor, controls the inverter using the estimation result, and the brushless DC motor. Which controls
  In addition to performing rotational position estimation using a transfer function, inverter control means for setting the gain of the round transfer function related to rotational position estimation to a value that satisfies the stability condition in the target phase and torque range is included.
[0047]
  The brushless DC motor control device according to claim 24 employs a transfer function G expressed by Equation 1 as the inverter control means, and transmits so that all real roots of the denominator polynomial of G / (1 + G) are negative. What sets the gain Kθ and F (s) of the function G is adopted.
[0048]
  The brushless DC motor control device according to claim 25 employs, as the inverter control means, one that estimates the rotational position of the rotor using a fixed coordinate motor model.
[0049]
  A brushless DC motor control device according to a twenty-sixth aspect employs, as the inverter control means, one that identifies a motor device constant before starting operation.
[0050]
  A brushless DC motor control device according to a twenty-seventh aspect employs, as the inverter control means, one that advances a target current phase to a maximum efficiency phase or a maximum torque phase of a motor.
[0051]
  The brushless DC motor control apparatus according to claim 28 employs a brushless DC motor that drives a compressor.
[0054]
[Action]
  In the brushless DC motor control method according to claim 1, in detecting and controlling the rotational position of the rotor without using the rotational position sensor, the brushless DC motor having a permanent magnet mounted inside the rotor is controlled.
  In order to achieve maximum efficiency control, the current phase is controlled, and the current phase is calculated based on an equation that approximates the current phase that can be achieved for maximum efficiency, obtained in advance for each operating state of the brushless DC motor. Since the target current phase is set, the rotational position of the rotor can be detected without using a position sensor, the target current phase need not be measured, the processing can be simplified, and the maximum efficiency can be achieved. Control can be achieved.
[0056]
  According to the brushless DC motor control method of claim 2, the rotational position of the rotor is detected based on the voltage at the neutral point of the stator of the brushless DC motor and the average value of the terminal voltages of the stator windings. The detected position signal is corrected based on the operating condition of the brushless DC motor, and the current phase is controlled based on the corrected position signal, so that the voltage at the neutral point of the stator of the brushless DC motor is fixed. This can be applied to the case where the rotational position of the rotor is detected based on the average value of the terminal voltages of the child windings, and maximum efficiency control can be achieved.
[0057]
  According to the brushless DC motor control method of claim 3, the rotation position of the rotor is calculated by performing a predetermined calculation using the applied voltage of the stator winding, the motor current, and the device constant of the brushless DC motor. Since the current phase is controlled based on the rotation position thus determined, the rotation position of the rotor is calculated by performing predetermined calculations using the applied voltage of the stator winding, the motor current, and the device constants of the brushless DC motor. Can be applied to achieve maximum efficiency control.
[0058]
  According to the brushless DC motor control method of claim 4, the rotational position of the rotor is calculated from the inductance obtained from the harmonic current generated by the voltage type inverter and the saliency of the rotor, and based on the calculated rotational position. Therefore, it can be applied to calculate the rotational position of the rotor from the inductance obtained from the harmonic current generated by the voltage-type inverter and the saliency of the rotor. Can be achieved.
[0063]
  In the brushless DC motor control device according to claim 5, in detecting and controlling the rotational position of the rotor without using the rotational position sensor, the brushless DC motor having a permanent magnet mounted inside the rotor is controlled.
  An equation that approximates the current phase that can be achieved by the current phase control means to obtain the maximum efficiency by controlling the current phase so as to achieve the maximum efficiency control and obtaining the current phase for each operation state of the brushless DC motor. The target current phase calculated based on can be made.
  Accordingly, the rotational position of the rotor can be detected without using a position sensor, the target current phase need not be measured, the processing can be simplified, and the maximum efficiency control can be achieved. .
[0065]
  In the brushless DC motor control device according to claim 6, the rotational position detecting means detects the rotor voltage based on the neutral point voltage of the stator of the brushless DC motor and the average value of the terminal voltages of the stator windings. The rotational position is detected, the correction means corrects the detected position signal based on the operating condition of the brushless DC motor, and the current phase control means can control the current phase based on the corrected position signal. .
[0066]
Therefore, it can be applied when detecting the rotational position of the rotor based on the neutral point voltage of the stator of the brushless DC motor and the average value of the terminal voltages of the stator windings, and maximum efficiency control can be performed. Can be achieved.
[0067]
  In the brushless DC motor control device according to claim 7, the rotation position calculation means performs a predetermined calculation using the applied voltage of the stator winding, the motor current, and the device constants of the brushless DC motor to rotate the rotor. The position can be calculated, and the current phase can be controlled by the current phase control means based on the calculated rotational position.
[0068]
Therefore, it can be applied when calculating the rotational position of the rotor by performing a predetermined calculation using the applied voltage of the stator winding, the motor current, and the device constants of the brushless DC motor, thereby achieving maximum efficiency control. can do.
[0069]
  According to the brushless DC motor control device of claim 8, the rotational position calculating means calculates the rotor rotational position from the inductance obtained from the harmonic current generated by the voltage type inverter and the saliency of the rotor, and the current phase. The current phase can be controlled by the control means based on the calculated rotational position.
[0070]
Therefore, the present invention can be applied to the case where the rotational position of the rotor is calculated from the inductance obtained from the harmonic current generated by the voltage type inverter and the saliency of the rotor, and maximum efficiency control can be achieved.
[0071]
  According to the brushless DC motor control method of claim 9, the rotational position of the rotor is estimated from the magnetic flux information using the motor voltage, the motor current, and the device constant of the brushless DC motor, and the inverter is controlled using the estimation result. In controlling a brushless DC motor with saliency,
  Since the inverter is controlled to change the operating point in response to the magnetic flux being almost zero, the applicable range can be expanded without being influenced by the motor structure, and the rotational position of the rotor. Can always be accurately estimated and the brushless DC motor can be stably controlled, and step-out due to inability to estimate the position can be avoided.
[0073]
  According to the brushless DC motor control method of claim 10, it is detected that the magnetic flux is almost zero, and in response to detecting that the magnetic flux is almost zero, the phase advance control is performed and the magnetic flux becomes almost zero. Therefore, the same effect as that of the ninth aspect can be achieved.
[0074]
  According to the brushless DC motor control method of claim 11, it is detected that the magnetic flux is substantially zero, and the current drooping control is performed in response to detecting that the magnetic flux is substantially zero, so that the magnetic flux becomes substantially zero. Therefore, the same effect as that of the ninth aspect can be achieved.
[0076]
  In the brushless DC motor control method according to claim 12, since the rotational position of the rotor is estimated using the rotary coordinate motor model, in addition to the effects of any one of claims 9 to 11, It can be suitably applied to applications where the fluctuation of angular velocity is small.
[0077]
  According to the brushless DC motor control method of claim 13, the rotational position of the rotor is estimated using the motor voltage, the motor current, and the device constants of the brushless DC motor, and the inverter is controlled using this estimation result. In controlling the DC motor,
  Since the gain of the loop transfer function relating to the rotational position estimation is set to a value that satisfies the stability condition in the target phase and torque range, the rotational position of the rotor is always accurately estimated to stably control the brushless DC motor. Can do.
[0078]
  According to the brushless DC motor control method of claim 14, the transfer function G is expressed by Equation 1, and the gains Kθ and F of the transfer function G are set so that all real roots of s in the denominator polynomial of G / (1 + G) are all negative. Since (s) is set, the same effect as that of the thirteenth aspect can be achieved.
[0079]
  According to the brushless DC motor control method of claim 15, since the rotational position of the rotor is estimated using a fixed coordinate motor model, in addition to the effects of any of claims 9 to 11, in addition to the effects of The rotational position of the rotor can be accurately estimated regardless of the change in angular velocity.
[0080]
  According to the brushless DC motor control method of claim 16, since the motor device constant is identified before the start of operation, in addition to the effect of any of claims 9 to 15, the rotational position of the rotor can be accurately determined. Estimation can be performed.
[0081]
  According to the brushless DC motor control method of claim 17, the target current phase is advanced beyond the maximum efficiency phase or the maximum torque phase of the motor. Therefore, in addition to the effect of any of claims 9 to 16, the brushless The operating range of the DC motor can be expanded to the high speed side.
[0082]
  Since the brushless DC motor control method according to claim 18 employs a brushless DC motor that drives a compressor, in addition to the effect of any one of claims 9 to 17, the load fluctuation is large. In addition, it is possible to stably drive a compressor that needs to be operated at high speed.
[0083]
  According to the brushless DC motor control device of claim 19, the rotational position of the rotor is estimated from the magnetic flux information using the motor voltage, the motor current, and the device constants of the brushless DC motor, and the inverter is controlled using this estimation result. In controlling a brushless DC motor with saliency,
  The inverter control means can control the inverter to change the operating point in response to the magnetic flux being almost zero.
[0084]
  Therefore, the applicable range can be expanded without being influenced by the structure of the motor, and the brushless DC motor can be stably controlled by always accurately estimating the rotational position of the rotor. Step-out due to inability can be avoided.
[0086]
  In the brushless DC motor control device according to claim 20, the inverter control means detects that the magnetic flux is almost zero and performs phase advance control in response to detecting that the magnetic flux is almost zero. Therefore, the same action as that of the nineteenth aspect can be achieved.
[0087]
  The brushless DC motor control device according to claim 21, wherein the inverter control means detects that the magnetic flux is almost zero and performs current droop control in response to detecting that the magnetic flux is almost zero. Therefore, the same action as that of the nineteenth aspect can be achieved.
[0090]
  In the brushless DC motor control device according to claim 22, since the inverter control means employs a device that estimates the rotational position of the rotor using a rotational coordinate motor model. In addition to any of the above effects, the present invention can be suitably applied to applications in which fluctuations in current and angular velocity are small.
[0091]
  According to the brushless DC motor control device of claim 23, the rotational position of the rotor is estimated using the motor voltage, the motor current, and the device constants of the brushless DC motor, and the inverter is controlled using the estimation result, and the brushless In controlling the DC motor,
  The inverter control means can estimate the rotational position using the transfer function and set the gain of the round transfer function related to the rotational position estimation to a value that satisfies the stability condition in the target phase and torque range.
[0092]
Therefore, the brushless DC motor can be stably controlled by always accurately estimating the rotational position of the rotor.
[0093]
  In the brushless DC motor control device according to claim 24, as the inverter control means, a transfer function G expressed by Equation 1 is adopted so that all real roots of s of the denominator polynomial of G / (1 + G) are negative. Since the one that sets the gains Kθ and F (s) of the transfer function G is adopted, the same effect as in the twenty-third aspect can be achieved.
[0094]
  In the brushless DC motor control device according to claim 25, the inverter control means employs a device that estimates the rotational position of the rotor using a fixed coordinate motor model. In addition to any of the above effects, the rotational position of the rotor can be accurately estimated regardless of changes in current and angular velocity.
[0095]
  In the brushless DC motor control device according to claim 26, the inverter control means employs a device that identifies a motor device constant before starting operation. Therefore, the effect of any one of claims 19 to 25 is achieved. In addition, the rotational position of the rotor can be accurately estimated.
[0096]
  In the brushless DC motor control device according to claim 27, the inverter control means employs a device that advances the target current phase to the maximum efficiency phase or the maximum torque phase of the motor. In addition to any of the effects of No. 26, the operating range of the brushless DC motor can be expanded to the high speed side.
[0097]
  Since the brushless DC motor control device according to claim 28 employs a brushless DC motor that drives a compressor, in addition to the effects of any one of claims 19 to 27, the load fluctuation is large. In addition, it is possible to stably drive a compressor that needs to be operated at high speed.
[0098]
Further explanation will be given.
[0099]
Substituting Formula 7 of coordinate transformation into Formula 2, Formula 3, Formula 4, Formula 5, and Formula 6 described in “Sensorless Control of IPM Motor”, Motor Technology Symposium B-5, 99 / March, dq Voltage on axis V_dq, Current i_dqWhen the true rotor angle θ is used to rewrite the system using the motor inverse model and feedback, the result is as shown in FIG.
[0100]
[Expression 2]

[0101]
[Equation 3]

[0102]
[Expression 4]

[0103]
[Equation 5]

[0104]
[Formula 6]

[0105]
[Expression 7]

In FIG. 8, θ is an electrical angle, ω is an electrical angular velocity, and v_γ, V_δIs the γδ axis voltage, i_γ, I_δIs γδ current, ε_γ, Ε_δIs the γδ-axis induced voltage, φ is the armature flux linkage by the permanent magnet, R is the armature resistance, L_d, L_qIs a dq axis inductance, s is a differential operator, and ^ is an estimated value. Also, I and v in FIG._γδ, I_γδ, Ε_γδ, L_I, L_J, Θ_e, T (θ_e), Α_I, Β_IIs as shown in Eq.
[0106]
[Equation 8]

Furthermore, when the nonlinear element in the feedback loop is linearized (cos θ_e→ 1, sin θ_e→ θ_e, ^ Ω → ω), the block configuration shown in FIG. 9 is obtained.
[0107]
Using the block configuration of FIG._dqConsider the convergence when is periodic. X from linear approximation block diagram_γIs given by equation (9).
[0108]
[Equation 9]

Here, the inside of {} in Equation 9 is a₀+ A_1ssinθ₁+ A_1ccosθ₁,
θ_e= E₀+ E_1ssinθ₁+ E_1ccosθ₁+ E_2ssin2θ₁+ E_2ccos2θ₁Consider the case. x_γIs given by Equation 10.
[0109]
[Expression 10]

θ_eCannot be detected, so instead x_γWhen the DC component and the primary component of are controlled to 0 (kθ → ∞ is required), e is so that the dotted line in Equation 9 becomes 0₀, E_1s, E_1cIs adjusted. The purpose (e₀= E_1s= E_1c= 0) to achieve a₀≠ 0 and a_1s= A_1c= 0. And the former is {φ + (L_d-L_q) I_d} The average value of ω is equivalent to non-zero, the latter being {φω + (L_d-L_q) [Ω_s  -S] i_dq} Is equivalent to the primary component and the secondary component of. This is because ω is constant and L_d= L_qOr i_d, I_qIs equivalent to being constant.
[0110]
As a result, a sufficient condition for ^ θ → θ is stable and (1) θ_eIs small (cos θ_e→ 1, sin θ_e→ θ_e), (2) Equipment constants other than φ (R, L_d, L_q) No deviation, (3) k_θ= ∞, (4) ^ ω = ω = constant, (5) L_d= L_qOr i_d, I_qIs constant, (6) {φ + (L_d-L_q) I_d} The average value of ω is not 0. At this time, in particular (L_d-L_q) I_d+ Φ ≠ 0 and ensuring stability are absolute conditions for position detection, while others are conditions for error reduction.
[0111]
FIG. 10 shows another block configuration for performing position detection using a motor inverse model and without using a sensor.
[0112]
In FIG. 10, θ is an electrical angle, ω is an electrical angular velocity, γδ coordinates are ^ θ rotation coordinates, αβ coordinates are two-phase orthogonal stator coordinates, v_γ, V_δIs the γδ axis voltage, v_α, V_βIs the αβ axis voltage, i_α, I_βIs the αβ axis current, φ is the armature flux linkage, R is the armature resistance, L_d, L_qIs a dq axis inductance, s is a differential operator, ^ is an estimated value, and ￣ is a sensor value. In addition, L in FIG._p, L_m,, V_γδ, V_αβ, I_αβ, Φ_αβ, F (θ) are as shown in Equation 11.
[0113]
## EQU11 ##

Even when this block configuration is adopted, {(L_d-L_q) Φ_αβ ^Ti_αβ+ Φ} = (L_d-L_q) I_dIt can be seen that + φ ≠ 0 is a condition for position detection.
[0114]
FIG. 11 shows the current phase based on the d-axis and the current value I after two-phase conversion._aΨ = (L_d-L_q) I_dA graph of + φ is shown.
[0115]
Current phase is -50 ° to -130 °, I_aΨ = 0 is recognized in the region of> 7A, and it can be seen that position detection may be impossible in this region. I_aIs (3/2) of the peak value of the phase current in the three-phase motor^1/2It is.
[0116]
DETAILED DESCRIPTION OF THE INVENTION
Embodiments of a brushless DC motor control method and apparatus according to the present invention will be described below in detail with reference to the accompanying drawings.
[0117]
FIG. 1 is a block diagram showing an embodiment of the brushless DC motor control device of the present invention.
[0118]
This brushless DC motor control device supplies the output from an inverter (preferably a voltage source inverter) 1 with DC power as an input to a brushless DC motor 2 in which a permanent magnet is mounted inside the rotor. ing. Then, the rotational position θ of the rotor of the brushless DC motor 2 is detected without using a position sensor, and the rotor rotational position detector 3 that detects the rotational speed ω, the speed command given from the outside, and the detected rotation Speed control unit 4 that performs speed control calculation with speed ω as input and outputs current phase command and current amplitude command, and performs current control calculation with rotation position θ, current phase command and current amplitude command as input, and performs PWM (pulse (Width modulation) command and a current control unit 5 for supplying the command to the inverter 1.
[0119]
In the rotor rotation position detector 3, the position detector calculates the difference between the neutral point voltage of the stator winding of the brushless DC motor 12 and the average value of the terminal voltage of the stator winding, and the difference voltage The detected zero cross is detected, the predicted position of the rotor is obtained from the detected zero cross and output, and the position correction unit obtains the relationship between the amplitude of the difference voltage and the phase error of the difference voltage in advance and cancels it. In addition, the predicted position output from the position detection unit is corrected. Thereby, the rotational position of the rotor is accurately detected.
[0120]
However, instead of the position correction unit, the amplitude of the differential voltage, the amplitude of the integrated signal of the differential voltage, the phase current of the brushless DC motor, the phase voltage of the brushless DC motor, the input current (DC current) of the inverter, the input voltage of the inverter (DC voltage), brushless DC motor rotation speed, inverter internal information {pulse width modulation (PWM) command, rotation speed, etc.} is used as an input to calculate a correction amount, and the calculated correction amount It is possible to employ a position correction unit that corrects the provisional predicted position output from the position detection unit based on the above and outputs the corrected predicted position as a corrected predicted position.
[0121]
Further, instead of the position correction unit, a correction map for outputting a correction amount (correction angle) with the amplitude of the difference voltage and the amplitude of the phase current of the brushless DC motor as inputs, and a temporary rotor output from the position detection unit. It is also possible to employ an adding unit that adds a correction amount to the predicted position and obtains and outputs a corrected predicted position.
[0122]
Furthermore, a first position detection unit that outputs the first temporary predicted position of the rotor with the difference voltage as an input, and a motor model that outputs the difference voltage modeled with the phase voltage PWM command and the rotor position as inputs. A second position detection unit that outputs the second temporary predicted position of the rotor with the modeled difference voltage as an input, and calculates a difference between the first temporary predicted position and the second temporary predicted position A difference calculation unit that performs the calculation, a correction amount output unit that obtains and outputs a correction amount (correction angle) that sets the difference between the predicted positions as 0, and subtracts the correction amount from the first temporary predicted position. It is also possible to adopt a configuration including a subtracting unit that obtains and outputs a corrected predicted position (the rotor position). In this case, a first amplitude detector that detects the amplitude of the difference voltage, a second amplitude detector that detects the amplitude of the modeled difference voltage, and a difference amplitude calculator that calculates the difference between the two amplitudes are further provided. In addition to the correction amount output unit, it is also possible to adopt a correction amount output unit that obtains and outputs a correction amount (correction angle) that makes the difference between the predicted positions and the difference between both amplitudes as input and sets these to zero. It is.
[0123]
Furthermore, a first position detector that outputs the first tentative predicted position of the rotor with the difference voltage as an input, and a second tentative prediction of the rotor with the phase voltage and phase current of the brushless DC motor as inputs. A second position detection unit that outputs a position; a difference calculation unit that calculates a difference between the first temporary predicted position and the second temporary predicted position; and an error storage unit that stores the calculated difference as an input It is also possible to adopt a configuration having an addition unit that adds the error stored in the error storage unit to the first provisional predicted position and outputs it as a corrected predicted position (Japanese Patent Application No. Hei. 10-315035).
[0124]
The rotor rotational position detector 3 estimates a speed electromotive force from the command voltage and the motor current, defines a γ-δ axis corresponding to the estimated speed electromotive force, and calculates the estimated speed electromotive force to a predetermined value. The rotational position and speed of the rotor are estimated by matching the speed electromotive force expressed as a q-axis direction vector with a magnitude of (specifically, the estimated rotational position based on the direction of the estimated speed electromotive force) And obtain the estimated speed by the magnitude of the estimated speed electromotive force) ("Sensorless salient pole type brushless DC motor control based on the speed electromotive force estimation", Takaharu Takeshita et al., T. IEEE Japan, Vol. 117-D, No. .1, '97).
[0125]
Further, the rotor rotational position detector 3 extracts harmonic components contained in the motor terminal voltage and current (by calculating the difference between the average output voltage vector of the inverter and each inverter output voltage vector, By extracting the harmonic component contained in the voltage and calculating the difference between the first and last current vectors of the modulation period in the predetermined period with respect to the modulation period from the amount of change in the current vector due to the voltage vector not used in the modulation period The harmonic component of the motor current vector is extracted), and the rotational position of the rotor can also be estimated (the inductance matrix is obtained using the left pseudo inverse matrix for the equation for the harmonic component of one modulation period, and this inductance is calculated. Estimate the rotational position of the rotor from the matrix) ("Position sensor using position estimation method based on saliency Scan IPM motor drive system ", Satoshi Ogasawara other, T.IEE Japan, Vol.118-D, No.5, see '98).
[0126]
The speed control unit 4 corrects the torque to be generated from the speed command and the rotation speed, and outputs a new current phase command and current amplitude command. Specifically, (1) a current amplitude command is calculated from the speed command and the rotation speed (in PI control or the like, generally, if the rotation speed is slow, the current command is increased), and (2) the corrected current amplitude The torque T is calculated from the command and the current phase command before correction, the torque T is calculated, and the current phase for making the brushless DC motor 2 maximum efficiency is calculated and output. (3) The corrected current amplitude The torque T is calculated by calculating the following equation from the command and current phase command, the current phase for making the brushless DC motor 2 maximum efficiency is calculated and output, and the processes of (4) and (3) are repeated as appropriate. Do it.
T = P_n{Φ + (L_d-L_q) (− Isinβ)} Icosβ
Where P_nIs the number of pole pairs, I is the current amplitude, and β is the current phase.
[0127]
Of course, when it is necessary to control the torque accurately and at high speed, the current amplitude command and the current phase command can be directly calculated by solving the inverse problem from the torque equation incorporating the current phase.
[0128]
The current controller 5 can directly generate a PWM command by performing the following calculation from the current amplitude command, current phase command, and rotational position (or rotational speed) of the rotor.
[0129]
The voltage equation of the IPM expressed in the rotating coordinate system (d, q axis) is expressed by Equation 12.
[0130]
[Expression 12]

Where R_aIs winding resistance, ω is rotation speed, L_dIs d-axis inductance, L_qIs the q-axis inductance, I_d, I_qAre the d-axis, q-axis current, and V, respectively._d, V_qRepresents the d-axis and q-axis voltages, and φ represents the number of flux linkages.
[0131]
Therefore, using the current command and the rotation speed, the d-axis and q-axis voltage V_d, V_qIt can be seen that can be calculated.
[0132]
At this time, the voltage amplitude V and the voltage phase δ are
V = (V_d ²+ V_q ²)^1/2
δ = tan^-1(-V_d/ V_q)
And the phase voltage V_u(V_v, V_wIs the same except that the phase is shifted by 120 °)
V_u= (2/3)^1/2Vsin (θ + δ)
It is expressed.
[0133]
When the brushless DC motor control device having the above configuration is adopted, the motor current is supplied to the brushless DC motor with a current phase that achieves the maximum efficiency according to the rotor position, so that the maximum efficiency control is performed even in an environment where the load fluctuates. In addition, the maximum efficiency control can be stably performed.
[0134]
Further, as the current control unit 5, a unit that directly generates a PWM signal from the current phase command, current amplitude command, and rotor rotation position (rotation speed) is adopted. A device that detects the rotational position of the rotation based on the voltage of the sex point and the average value of the terminal voltage of the stator winding is adopted, and the detected rotational position signal is corrected based on the operating conditions of the brushless DC motor. By performing the control (see Japanese Patent Application No. 10-315035), the rotor rotational position and the rotational speed are calculated and controlled, so that a current sensor or the like is not required, and maximum efficiency control without a position sensor can be realized at low cost.
[0135]
FIG. 2 is a block diagram showing another embodiment of the brushless DC motor control device of the present invention.
[0136]
This brushless DC motor control device supplies an output from an inverter 11 (preferably a voltage source inverter) 11 having DC power as an input to a brushless DC motor 12. Then, the rotational position θ and the rotational speed ω of the rotor are detected by the rotor rotational position detector 13 and supplied to the drive waveform generator 14. Also, the torque T calculated by the torque calculation unit 15 that receives the load information is supplied to the current phase determination unit 16 to calculate an approximate expression representing the current phase β to determine the current phase β, and generate a drive waveform It supplies to the part 14. Further, the speed command value ω * output from the speed control unit 17 and the operation current I in the inverter 11 are supplied to the drive waveform generation unit 14. The drive waveform generator 14 generates a PWM command (PWM pattern signal) based on these input signals and supplies the PWM command to the inverter 11.
[0137]
The operation of the brushless DC motor control device having the above-described configuration is as follows.
[0138]
First, the torque calculator 15 calculates the torque T applied to the brushless DC motor 12 from the load information (for example, in the case of an air conditioner, the current temperature inside and outside of the room, the temperature setting value, etc.). Then, the calculated torque T is supplied to the current phase determination unit 16 to calculate an approximate expression representing the current phase β, and the control target current phase β is determined.
[0139]
Further, the current rotational position of the rotor of the brushless DC motor 12 is obtained using a signal from the brushless DC motor 12 (for example, correcting the third harmonic of the neutral point of the stator winding according to the load condition). θ is calculated.
[0140]
Then, in addition to the determined current phase β and the current rotational position θ, the speed command value ω * from the speed control unit 17 and the operation current I in the inverter 12 are supplied to the drive waveform generation unit 14 to supply the brushless DC motor 12. The PWM pattern signal for driving is generated and supplied to the inverter 11. The inverter 11 controls a power device such as an IGBT based on the PWM pattern signal and drives the brushless DC motor 12.
[0141]
Here, the current phase which becomes the maximum efficiency for each torque of the IPM at a certain number of rotations (rotation speed) is as shown in FIG. 3, for example. Moreover, the current phase characteristic which becomes the maximum efficiency for each rotation speed is as shown in FIG. 4, for example. 3 and 4 that the maximum efficiency is obtained when the current phase is in the vicinity of 20 ° to 40 °. Therefore, the maximum efficiency phase β is expressed by a function of the torque T {for example, an approximate expression β = A · T + B (A and B are constants)}, and this function is incorporated in the current phase determination unit 16 to simplify the operation. The maximum efficiency phase β can be obtained and reflected in the generation of the drive waveform.
[0142]
Therefore, if the brushless DC motor control device of FIG. 2 is adopted, the maximum efficiency phase β can be easily calculated, and the brushless DC motor is controlled by controlling the brushless DC motor using the maximum efficiency phase β as a control target. Maximum efficiency control can be achieved.
[0143]
FIG. 5 is a block diagram showing still another embodiment of the brushless DC motor control apparatus of the present invention.
[0144]
The brushless DC motor control device is different from the brushless DC motor control device of FIG. 2 in that a maximum efficiency phase table 18 storing the current phase β corresponding to the torque T is employed instead of the current phase determining unit 16. Only.
[0145]
An example of the contents of the maximum efficiency phase table 18 is shown in Table 1.
[0146]
[Table 1]

Of course, the content of the maximum efficiency phase table 18 is set based on the maximum efficiency phase characteristic obtained in advance as shown in FIG. Here, the resolution of the maximum efficiency phase table 18 may be set in consideration of the capacity of the memory, and when the capacity of the memory is small, the resolution of the torque T may be coarsened and interpolation between each point may be performed.
[0147]
The operation of the brushless DC motor control device having the above-described configuration is as follows.
[0148]
First, the torque calculator 15 calculates the torque T applied to the brushless DC motor 12 from the load information (for example, in the case of an air conditioner, the current temperature inside and outside of the room, the temperature setting value, etc.). Then, the calculated torque T is supplied to the maximum efficiency phase table 18 to determine the current phase β of the control target.
[0149]
Further, the current rotational position of the rotor of the brushless DC motor 12 is obtained using a signal from the brushless DC motor 12 (for example, correcting the third harmonic of the neutral point of the stator winding according to the load condition). θ is calculated.
[0150]
Then, in addition to the determined current phase β and the current rotational position θ, the speed command value ω * from the speed control unit 17 and the operation current I in the inverter 12 are supplied to the drive waveform generation unit 14 to supply the brushless DC motor 12. The PWM pattern signal for driving is generated and supplied to the inverter 11. The inverter 11 controls a power device such as an IGBT based on the PWM pattern signal and drives the brushless DC motor 12.
[0151]
FIG. 6 is a block diagram showing still another embodiment of the brushless DC motor control apparatus of the present invention.
[0152]
The brushless DC motor control device is different from the brushless DC motor control device of FIG. 2 only in that a current phase setting unit 19 is provided in place of the torque calculation unit 15 and the current phase determination unit 16.
[0153]
The current phase setting unit 19 outputs a predetermined current phase β set in advance regardless of the operating state of the brushless DC motor 12. Here, if the current phase β is kept constant, there is a possibility that the maximum efficiency phase cannot always be realized when torque fluctuation, rotation speed fluctuation, or the like occurs. However, as can be seen from FIG. 3 and FIG. 4, even if torque fluctuation, rotation speed fluctuation, etc. occur, the maximum efficiency phase fluctuation is not so large. By setting with reference to FIG. 4 and the like, the brushless DC motor 12 can be operated with substantially the same efficiency as compared with the case where the maximum efficiency phase is realized (see FIG. 7).
[0154]
When each of the above embodiments is adopted, the brushless DC motor can be controlled at maximum efficiency at a low cost without adding means for detecting motor efficiency, for example, parts such as a current sensor and a voltage sensor. In particular, when the embodiments of FIGS. 5 and 6 are employed, it is not necessary to calculate the maximum efficiency phase in real time, and control can be realized with a low-cost microcomputer.
[0155]
FIG. 12 is a block diagram showing still another embodiment of the brushless DC motor control apparatus of the present invention.
[0156]
This brushless DC motor control device includes a converter 22 that receives an AC power supply 21, a smoothing capacitor 23 that smoothes the output voltage of the converter 22, an inverter 24 that receives the smoothed DC voltage, and an inverter 24. A brushless DC motor 25 to which an output AC voltage is applied; a current detection unit 26 that detects a supply current to the brushless DC motor 25; a voltage detection unit 27 that detects a voltage applied to the brushless DC motor 25; Magnetic flux Ψ = (L with detection current and detection voltage as inputs_d-L_q) I_d+ Φ, a position / speed detector 28 for detecting the rotor position and speed, and a Ψ≈0 detector 29 for detecting that the magnetic flux Ψ≈0 by inputting the detected magnetic flux Ψ and outputting a phase advance command; A speed control unit 30 that outputs a current command by performing a speed control calculation with the detected speed and a speed command given from the outside as input, and a phase with the current command, a phase command given from the outside, and a phase advance command as inputs A phase control unit 31 that outputs a phase by performing a control calculation, and a voltage command is output by performing a current control calculation with the phase, detection current, and rotor position output from the phase control unit 31 as inputs, and is supplied to the inverter 24 And a voltage control unit 32.
[0157]
The device constant of the motor is preset in the position / speed detection unit 28, and the magnetic flux information is calculated by using the device constant, the motor current (supply current), and the motor voltage (applied voltage). The rotor position is estimated from the magnetic flux information and the rotor position is output, and the speed is detected from the cycle of the motor current.
[0158]
When the brushless DC motor control device having the above-described configuration is adopted, the phase control unit 31 detects the phase command and the current command on the condition that the Ψ≈0 detection unit 29 does not detect that the magnetic flux Ψ≈0. Therefore, the rotor position can be estimated using the motor current, the motor voltage, and the equipment constant, and the brushless DC motor 25 can be controlled based on the estimated rotor position.
[0159]
Conversely, when the Ψ≈0 detection unit 29 detects that the magnetic flux Ψ≈0, the phase advance command is output, so that the phase control calculation is performed based on the phase command and the current command. By outputting the phase advanced from the phase from the phase control unit 31 and avoiding the magnetic flux Ψ = 0, the rotor position can be accurately estimated, and the brushless DC motor 25 can be stably operated. Step-out can be prevented).
[0160]
In the embodiment shown in FIG. 12, the position / velocity detection unit 28 calculates the magnetic flux Ψ. However, a Ψ calculation unit for calculating the magnetic flux Ψ may be provided separately, and a motor equivalent to Ψ = 0. You may calculate and use an electric current value, a motor electric current phase, etc. (refer FIG. 11). Further, direct voltage control may be performed without providing a current control system. Further, the motor voltage may be calculated from the inverter command instead of detecting the motor voltage.
[0161]
FIG. 13 is a block diagram showing still another embodiment of the brushless DC motor control apparatus of the present invention.
[0162]
This brushless DC motor control device is different from the brushless DC motor control device of FIG. 12 in that a current drooping command is output and supplied to the current control unit 32 instead of the Ψ≈0 detection unit 29 that outputs a phase advance command. It is only the point which employ | adopted (PSI) = 0 detecting part 29 '.
[0163]
Therefore, in this embodiment, when the Ψ≈0 detector 29 detects that the magnetic flux Ψ≈0, the inverter output current is decreased based on the current droop command and the magnetic flux Ψ = 0 is avoided, The rotor position can be accurately estimated, and the brushless DC motor 25 can be stably operated (a step-out can be prevented).
[0164]
FIG. 14 is a block diagram showing still another embodiment of the brushless DC motor control apparatus of the present invention.
[0165]
This brushless DC motor control device is different from the brushless DC motor control device of FIG. 12 in that a current value and a current phase of Ψ> 0 are stored in advance instead of the Ψ≈0 detection unit 29 that outputs a phase advance command. The only difference is that a> 0 table 29 ″ is provided, the current command output from the speed control unit 30 and the phase command given from the outside are output as read instruction signals, and the current phase is output to the phase control unit 31.
[0166]
Therefore, in this embodiment, the current phase that satisfies Ψ> 0 is read from the Ψ> 0 table 29 ″ and supplied to the phase control unit 31, thereby avoiding the magnetic flux Ψ = 0 and thus the rotor position. Thus, the brushless DC motor 25 can be stably operated (step-out can be prevented).
[0167]
Instead of providing the Ψ> 0 table 29 ″, a phase limit unit that satisfies Ψ> 0 may be provided.
[0168]
FIG. 15 is a block diagram showing still another embodiment of the brushless DC motor control apparatus of the present invention.
[0169]
This brushless DC motor control device includes an inverter 41 having DC power as an input, a brushless DC motor 42 to which an AC voltage output from the inverter 41 is applied, and a current detection unit 43 that detects motor currents for three phases. A three-phase-to-two-phase conversion unit 44 that converts a motor current for three phases into a two-phase current (αβ-axis current), and a converted two-phase current (αβ-axis current) and rotor angle (θ ) As input, αβ axis → dq axis conversion unit 45 that performs conversion to dq axis current, and -sin processing that outputs phase command and current command given from outside as input and outputs d-axis current command Unit 46, a cosine processing unit 47 that outputs a q-axis current command by performing a cosine process with a phase command and a current command given from the outside as inputs, and a d-axis power output from -sin processing unit 46 And the first difference calculation unit 48 that calculates the difference between the d axis current output from the αβ axis → dq axis conversion unit 45 and the q axis current output from the cos processing unit 47 and the αβ axis → dq axis conversion unit 45. A second difference calculation unit 49 that calculates a difference from the q-axis current output from the first and second differential calculation units 48, and a proportional gain K for the d-axis difference current output from the first difference calculation unit 48._dpAnd the integral gain K with respect to the d-axis difference current output from the first difference calculation unit 48._di, An integration processing unit 48c that performs integration processing (1 / s processing) on the output from the integration multiplication unit 48b, an output from the proportional processing unit 48a, and an output from the integration processing unit 48c. And an adder 48d that outputs a d-axis voltage command and a proportional gain K with respect to the q-axis difference current output from the second difference calculator 49._qpAnd the integral gain K with respect to the q-axis difference current output from the second difference calculation unit 49._qi, An integration processing unit 49c that performs integration processing (1 / s processing) on the output from the integration multiplication unit 49b, an output from the proportional processing unit 49a, and an output from the integration processing unit 49c And an adder 49d that outputs a q-axis voltage command, and a dq-axis → αβ-axis conversion unit 50 that converts the d-axis voltage, the q-axis voltage, and the rotor angle (θ) into an αβ-axis voltage. And a two-phase → three-phase conversion unit 51 that converts the αβ-axis voltage into a three-phase voltage command and supplies it to the inverter 41.
[0170]
Although the illustration of the portion for estimating the rotor angle (θ) is omitted, as described above, it is estimated from the motor current, the motor voltage and the equipment constant.
[0171]
When the brushless DC motor control device of this embodiment is employed, the brushless DC motor can be controlled by controlling the current phase so that the target current phase is obtained. Therefore, the brushless DC motor can always be controlled stably by setting the target current phase so that the magnetic flux Ψ = 0 does not occur.
[0172]
FIG. 16 is a block diagram showing still another embodiment of the brushless DC motor control apparatus of the present invention.
[0173]
This brushless DC motor control device includes an inverter 61 that receives DC power, a brushless DC motor 62 to which an AC voltage output from the inverter 61 is applied, and a current detector 63 that detects motor currents for three phases. Three-phase → r, θ conversion unit 64 that outputs r and θ conversion by inputting motor current for three phases and outputs current vector length and current phase, and the difference between current phase and rotor angle (θ) The first phase difference calculation unit 65 to be calculated, the current difference calculation unit 66 to calculate the difference between the current command given from outside and the current vector length, and the phase command given from the outside and output from the first phase difference calculation unit 65 A second phase difference calculation unit 67 for calculating a difference from the difference to be generated, and a proportional gain K for the output from the current difference calculation unit 66_lpAnd an integral gain K for the output from the proportional processing unit 66a and the current difference calculation unit 66._li, An integration multiplication unit 66b that multiplies the output from the integration multiplication unit 66b, an integration processing unit 66c that performs an integration process on the output from the integration multiplication unit 66b, and an output from the proportional processing unit 66a and an output from the integration processing unit 66c. A proportional gain K with respect to the output from the adder 66d that outputs the vector length and the second phase difference calculator 67_θpAnd an integral gain K with respect to the output from the proportional processing unit 67a and the second phase difference calculating unit 67._θi, An integration multiplication unit 67b for multiplying the output from the integration multiplication unit 67b, an integration processing unit 67c for performing an integration process on the output from the integration multiplication unit 67b, an output from the proportional processing unit 67a and an output from the integration processing unit 67c. An adder 67d for adding, an adder 68 for adding the output from the adder 67d and the rotor angle (θ) and outputting it as a voltage phase, a voltage vector length output from the adder 66d and an output from the adder 68 The three-phase conversion processing is performed by performing a three-phase conversion process using the voltage phase to be input, converting the voltage phase into a voltage command for three phases, and supplying the voltage command to the inverter 61.
[0174]
Although the illustration of the portion for estimating the rotor angle (θ) is omitted, as described above, it is estimated from the motor current, the motor voltage and the equipment constant.
[0175]
Even when the brushless DC motor control device of this embodiment is employed, the brushless DC motor can be controlled by controlling the current phase so that the target current phase is obtained. Therefore, the brushless DC motor can always be controlled stably by setting the target current phase so that the magnetic flux Ψ = 0 does not occur.
[0176]
FIG. 17 is a block diagram showing an example of the configuration of the position / velocity detection unit 28.
[0177]
The position / velocity detection unit 28 performs a γδ conversion by inputting a three-phase voltage and outputs a γδ voltage vector, and outputs a γδ voltage vector. The position / speed detection unit 28 performs a γδ conversion by inputting a three-phase current and a γδ current vector. A three-phase to γδ conversion unit 28b to output, a motor inverse model unit 28c to output a voltage vector with the γδ current vector as an input, a voltage vector output from the motor inverse model unit 28c and a three-phase to γδ conversion unit 28a A difference calculating unit 28d that calculates a difference from the voltage vector to be output, a filter 28e that receives the difference output from the difference calculating unit 28d, and a position / speed that estimates the position / velocity using the output from the filter 28e as an input. And a speed estimation unit 28f.
[0178]
Therefore, in this case, the rotor position can be detected using the rotary coordinate motor model. However, since the angular error may be canceled with respect to fluctuations in current and angular velocity, it is unsuitable for uses with large fluctuations, but can be suitably applied to uses with little fluctuations.
[0179]
Further, the following equation is obtained by Taylor linear approximation of the coordinate conversion unit of FIG.
T = i + Jθ_eHowever, it is Formula 13.
[0180]
[Formula 13]

The above equation is substituted into the block diagram of FIG. 8, and the block diagram of FIG. 9 is obtained. From this, the loop transfer function (round loop transfer function) G (s) of the closed loop system of Equation 14 is obtained.
[0181]
[Expression 14]

The gain K of each part is set so that the real root of s of the denominator polynomial of G / (1 + G) is all negative._θAnd F (s) can be set to ensure stability.
[0182]
Next, a procedure for setting the gain will be described.
(1) Set the range of current phase, current amplitude, and speed to be operated stably.
[0183]
Gain K_θAnd F (s), the initial gain value in the loop is set.
(2) In all combinations of various quantities set in (1), the rational function G is used to check whether the real roots of s of the denominator polynomial of G / (1 + G) are all negative.
(3) If a positive real root appears, change the gain and execute (2).
[0184]
Conversely, if a positive real root does not appear, that value is adopted.
[0185]
From the linear approximation block diagram of FIG. 9, as a result of (2), the combination of gains whose real roots are all negative is a stable gain.
[0186]
Specifically, “Sensorless salient pole type brushless DC motor control based on speed electromotive force estimation”, Takeshita et al. IEEE Japan, Vol. 117-D, no. The transfer function G having the configuration of 1, '97 is expressed by Equation 15.
[0187]
[Expression 15]

Then, the same motor equipment constants as in the above document are adopted, and for example, the gain is set with a speed of 15 rps, a current amplitude of 0.84 A, and a current phase of 0 to 360 ° as an operating range. At this time, K_θ= 1077, K_eWhen 5202000 is set, it is obtained as stable in the conventional stability determination. However, when the above determination method is used, the current phase becomes unstable at about 130 ° to 230 ° as shown in FIG. I understand that.
[0188]
Where K_θ= 107.7, K_eWhen setting = 520200, it can be seen that all current phases are stable {see (B) in FIG. 18}, and it is understood that such a gain setting is necessary to satisfy this operating range.
[0189]
Further, as the position / velocity detection unit 28, a fixed coordinate system position detector as shown in FIG. 10 may be employed.
[0190]
In this case, there is no position estimation error due to changes in current and angular velocity, and suitable brushless DC motor control can be achieved even when the changes in current and angular velocity are large.
[0191]
Furthermore, the motor device constants employed in the above embodiment may be set in advance, but the identification processing is performed at the start of operation, the motor device constants are set, and then the brushless DC motor is controlled. preferable. The identification process is performed as follows, for example.
Step 1
Forced motivation operation or operation that does not use a position sensor is performed, and the speed is almost constant and the voltage and current are almost constant._αβ, I_αβIs detected or estimated, and the calculation of Expression 16 is performed.
[0192]
[Expression 16]

However, n is an integer of 1 or more.
Step 2
Then, under conditions where the speed is substantially constant and the voltage and current are substantially constant, one or more of the speed, voltage, and current are_αβ, I_αβIs detected or estimated, and the calculation of Expression 17 is performed.
[0193]
[Expression 17]

However, n is an integer of 1 or more.
[0194]
Assuming that components other than the fundamental components of voltage and current are negligible, Equation 18 is obtained.
[0195]
[Expression 18]

Step 3
And the voltage equation on the stator coordinates of the synchronous motor
v_αβ= Ri_αβ+ S {L_pi_αβ+ L_mf (2θ) i_αβ} + S {φΘ}
Substituting Equation 18 into the above, the coefficients relating to Equations 19 and 20 that are mutually orthogonal variables are extracted, and Equation 21 is obtained.
[0196]
[Equation 19]

[0197]
[Expression 20]

[0198]
[Expression 21]

Step 4
Θ in Step 1 and Step 2_eΘ_e1, Θ_e2Then, Equations 16 and 17 are substituted into Equation 21 to obtain 8 simultaneous equations having 8 unknown constants. A value that is roughly set in advance as the initial value of the device constant, θ_e1, Θ_e2Is set to 0 as an initial value, and the obtained equation can be solved by Newton's method or the like to obtain an instrument constant (can be identified).
[0199]
Furthermore, it is preferable to advance the target current phase in the above-described embodiment to the maximum efficiency phase or maximum torque phase of the brushless DC motor.
[0200]
Referring to FIG. 19 showing the maximum efficiency current phase-torque characteristics, FIG. 20 showing the motor terminal voltage with respect to the current phase-torque, and FIG. 21 showing the maximum torque current phase, the motor terminal voltage decreases as the current phase is advanced. I understand. Therefore, even when the inverter output voltage reaches the limit, it is possible to perform further high-speed rotation by advancing the current phase. In other words, the operating range can be expanded to the high speed side.
[0201]
Next, consider the case where the compressor is driven by a brushless DC motor.
[0202]
Referring to FIG. 22 showing the compression torque with respect to the rotation angle in the one-cylinder rotary compressor, since the load fluctuation is large and the speed is high in the air conditioning compressor, (L_d-L_q) I_dIt is not easy to satisfy + φ> 0, but stable control is possible by high-speed current phase control.
[0203]
Therefore, by adopting the brushless DC motor control device of the above embodiment, stable and precise control by the method using the motor model can be realized even in the compressor, and effects such as energy saving and high efficiency can be achieved. Can be achieved.
[0206]
【The invention's effect】
  The invention of claim 1 can detect the rotational position of the rotor without using a position sensor, eliminates the need to measure the target current phase, simplifies the process, and provides maximum efficiency control. There is a unique effect that can be achieved.
[0208]
  The invention of claim 2 can be applied when detecting the rotational position of the rotor based on the neutral point voltage of the stator of the brushless DC motor and the average value of the terminal voltages of the stator windings. Thus, there is a specific effect that maximum efficiency control can be achieved.
[0209]
  The invention of claim 3 can be applied to the case where the rotation position of the rotor is calculated by performing a predetermined calculation using the applied voltage of the stator winding, the motor current, and the device constant of the brushless DC motor, There is a specific effect that maximum efficiency control can be achieved.
[0210]
  The invention of claim 4 can be applied to the case where the rotational position of the rotor is calculated from the inductance obtained from the harmonic current generated by the voltage type inverter and the saliency of the rotor, thereby achieving maximum efficiency control. There is a unique effect that can be.
[0214]
  The invention of claim 5 can detect the rotational position of the rotor without using a position sensor, eliminates the need to measure the target current phase, simplifies the process, and provides maximum efficiency control. There is a unique effect that can be achieved.
[0216]
  The invention of claim 6 can be applied when detecting the rotational position of the rotor based on the voltage at the neutral point of the stator of the brushless DC motor and the average value of the terminal voltages of the stator windings. Thus, there is a specific effect that maximum efficiency control can be achieved.
[0217]
  The invention of claim 7 can be applied to the case where the rotation position of the rotor is calculated by performing a predetermined calculation using the applied voltage of the stator winding, the motor current, and the device constant of the brushless DC motor, There is a specific effect that maximum efficiency control can be achieved.
[0218]
  The invention according to claim 8 can be applied to the case where the rotational position of the rotor is calculated from the inductance obtained from the harmonic current generated by the voltage type inverter and the saliency of the rotor, thereby achieving maximum efficiency control. There is a unique effect that can be.
[0219]
  According to the ninth aspect of the present invention, the range of application can be expanded without being influenced by the structure of the motor, and the brushless DC motor can be stably controlled by always accurately estimating the rotational position of the rotor. Furthermore, there is a specific effect that step-out due to inability to estimate the position can be avoided.
[0221]
  The invention of claim 10 has the same effect as that of claim 9.
[0222]
  The invention of claim 11 has the same effect as that of claim 9.
[0224]
  In addition to the effects of any one of claims 9 to 11, the invention of claim 12 has a specific effect that it can be suitably applied to applications in which fluctuations in current and angular velocity are small.
[0225]
  The invention according to claim 13 has a specific effect that the brushless DC motor can be stably controlled by always accurately estimating the rotational position of the rotor.
[0226]
  The invention of claim 14 has the same effect as that of claim 13.
[0227]
  In addition to the effects of any one of claims 9 to 11, the invention of claim 15 has a specific effect that the rotational position of the rotor can be accurately estimated regardless of changes in current and angular velocity.
[0228]
  The invention of claim 16 has the specific effect that the rotational position of the rotor can be accurately estimated in addition to the effect of any of claims 9 to 15.
[0229]
  In addition to the effects of any one of claims 9 to 16, the invention of claim 17 has a specific effect that the operating range of the brushless DC motor can be expanded to the high speed side.
[0230]
  In addition to the effect of any one of claims 9 to 17, the invention of claim 18 has a specific effect that a compressor that needs to be operated at a high speed with a large load fluctuation can be stably driven. .
[0231]
  The invention of claim 19 can expand the application range without being influenced by the structure of the motor, and can stably control the brushless DC motor by always accurately estimating the rotational position of the rotor. Furthermore, there is a specific effect that step-out due to inability to estimate the position can be avoided.
[0233]
  The invention of claim 20 has the same effect as that of claim 19.
[0234]
  The invention of claim 21 has the same effect as that of claim 19.
[0236]
  In addition to the effect of any one of claims 19 to 21, the invention of claim 22 has a specific effect that it can be suitably applied to applications in which fluctuations in current and angular velocity are small.
[0237]
The invention according to claim 23 has a specific effect that the brushless DC motor can be stably controlled by always accurately estimating the rotational position of the rotor.
[0238]
  The invention of claim 24 has the same effect as that of claim 23.
[0239]
  In addition to the effects of any one of claims 19 to 21, the invention of claim 25 has a specific effect that the rotational position of the rotor can be accurately estimated regardless of changes in current and angular velocity.
[0240]
  The invention of claim 26 has the specific effect that the rotational position of the rotor can be accurately estimated in addition to the effect of any of claims 19 to 25.
[0241]
  The invention of claim 27 has the specific effect that, in addition to the effect of any of claims 19 to 26, the operating range of the brushless DC motor can be expanded to the high speed side.
[0242]
  In addition to the effect of any one of claims 19 to 27, the invention of claim 28 has a specific effect that it is possible to stably drive a compressor having a large load fluctuation and requiring high speed operation. .
[Brief description of the drawings]
FIG. 1 is a block diagram showing an embodiment of a brushless DC motor control device of the present invention.
FIG. 2 is a block diagram showing another embodiment of the brushless DC motor control device of the present invention.
FIG. 3 is a diagram illustrating an example of a current phase characteristic of an embedded magnet motor.
FIG. 4 is a diagram showing a maximum efficiency current phase for each rotation speed of an embedded magnet motor.
FIG. 5 is a block diagram showing still another embodiment of the brushless DC motor control device of the present invention.
FIG. 6 is a block diagram showing still another embodiment of the brushless DC motor control device of the present invention.
FIG. 7 is a diagram showing a change in efficiency when the current phase is kept constant.
FIG. 8 is a block diagram showing a position detection unit based on a rotating coordinate model.
FIG. 9 is a block diagram obtained by linearly approximating FIG.
FIG. 10 is a block diagram showing a motor control device including a position detection unit based on a fixed coordinate model.
FIG. 11 shows a current phase based on the d-axis and a current value I after two-phase conversion._aΨ = (L_d-L_q) I_dIt is a graph of + φ.
FIG. 12 is a block diagram showing still another embodiment of the brushless DC motor control device of the present invention.
FIG. 13 is a block diagram showing still another embodiment of the brushless DC motor control device of the present invention.
FIG. 14 is a block diagram showing still another embodiment of the brushless DC motor control device of the present invention.
FIG. 15 is a block diagram showing still another embodiment of the brushless DC motor control device of the present invention.
FIG. 16 is a block diagram showing still another embodiment of the brushless DC motor control device of the present invention.
FIG. 17 is a block diagram illustrating a configuration of a position / speed detection unit.
FIG. 18 is a diagram showing a real part maximum value-current phase characteristic of a closed loop pole;
FIG. 19 is a diagram showing maximum efficiency current phase-torque characteristics.
FIG. 20 is a diagram showing a motor terminal voltage with respect to current phase-torque.
FIG. 21 is a diagram showing a maximum torque current phase.
FIG. 22 is a diagram showing a torque pattern of a one-cylinder rotary compressor.
[Explanation of symbols]
1, 11, 24, 41, 61 Inverter
2, 12, 25, 42, 62 Brushless DC motor
3, 13 Rotor rotational position detector 4 Speed controller
14 Drive waveform generator 29, 29 'Ψ≈0 detector
29 ″ Ψ> 0 table

Claims (28)

A brushless DC motor control method for detecting and controlling a rotational position of a rotor without using a rotational position sensor for a brushless DC motor (2) (12) having a permanent magnet mounted inside the rotor,
An equation that approximates the current phase that can be achieved for maximum efficiency, which is obtained by measuring the current phase in advance for each operating state of the brushless DC motors (2) and (12) in order to control the current phase to achieve maximum efficiency control. A method for controlling a brushless DC motor, wherein the target current phase is calculated based on   The rotational position of the rotor is detected based on the neutral point voltage of the stator of the brushless DC motor (2) (12) and the average value of the terminal voltage of the stator winding, and the detected position signal is brushless. The brushless DC motor control method according to claim 1, wherein the brushless DC motor is controlled based on the operating conditions of the DC motors (2) and (12), and the current phase is controlled based on the corrected position signal.   A predetermined calculation is performed using the applied voltage of the stator winding, the motor current, and the device constants of the brushless DC motors (2) and (12) to calculate the rotational position of the rotor, and based on the calculated rotational position The brushless DC motor control method according to claim 1, wherein the current phase is controlled.   Claims: The rotational position of the rotor is calculated from the inductance obtained from the harmonic current generated by the voltage type inverter (1) (11) and the saliency of the rotor, and the current phase is controlled based on the calculated rotational position. Item 2. The brushless DC motor control method according to Item 1. A brushless DC motor control device for detecting and controlling the rotational position of a rotor without using a rotational position sensor for a brushless DC motor (2) (12) having a permanent magnet mounted inside the rotor,
An equation that approximates the current phase that can be achieved for maximum efficiency, which is obtained by measuring the current phase in advance for each operating state of the brushless DC motors (2) and (12) in order to control the current phase to achieve maximum efficiency control. A brushless DC motor control device comprising current phase control means (4) (14) for setting a target current phase calculated based on   Rotational position detecting means (3) for detecting the rotational position of the rotor based on the neutral voltage of the stator of the brushless DC motor (2) (12) and the average value of the terminal voltage of the stator winding ( 13) and correction means (3) and (13) for correcting the detected position signal based on the operating condition of the brushless DC motor, and the current phase control means (4) and (14) are corrected. 6. The brushless DC motor control device according to claim 5, wherein the current phase is controlled based on the position signal.   Rotation position calculation means (3) (13) for calculating the rotation position of the rotor by performing a predetermined calculation using the applied voltage of the stator winding, the motor current, and the device constants of the brushless DC motor (2) (12). The brushless DC motor control device according to claim 5, wherein the current phase control means (4) (14) controls the current phase based on the calculated rotational position.   Rotation position calculation means (3) (13) for calculating the rotor rotation position from the inductance obtained from the harmonic current generated by the voltage type inverter (1) (11) and the saliency of the rotor, The brushless DC motor control device according to claim 5, wherein the phase control means (4) (14) controls the current phase based on the calculated rotational position. The rotational position of the rotor is estimated from the magnetic flux information using the motor voltage, the motor current, and the device constants of the brushless DC motors (25), (42), and (62), and the inverter (24) (41) is estimated using this estimation result. (61) is a method for controlling brushless DC motors (25), (42) and (62) having saliency,
A method of controlling a brushless DC motor, wherein the inverter (24) (41) (61) is controlled to change the operating point in response to the magnetic flux being substantially zero.   10. The brushless DC motor according to claim 9, wherein the brushless DC motor is detected by detecting that the magnetic flux is substantially zero and performing a phase advance control in response to detecting that the magnetic flux is substantially zero to avoid the region where the magnetic flux is substantially zero. Control method.   10. The brushless DC motor according to claim 9, wherein the brushless DC motor is detected by detecting that the magnetic flux is substantially zero and performing a current droop control in response to detecting that the magnetic flux is substantially zero to avoid a region where the magnetic flux is substantially zero. Control method.   The brushless DC motor control method according to claim 9, wherein the rotational position of the rotor is estimated using a rotational coordinate motor model. The rotational position of the rotor is estimated using the motor voltage, the motor current, and the device constants of the brushless DC motors (25), (42), and (62), and the inverters (24), (41), and (61) are estimated using the estimation results. And controlling the brushless DC motor (25) (42) (62),
A brushless DC motor control method, characterized in that a gain of a loop transfer function relating to rotational position estimation is set to a value that satisfies a stability condition in a target phase and a torque range. 14. The brushless according to claim 13, wherein the transfer function G is expressed by Formula 1 and the gains Kθ and F (s) of the transfer function G are set so that all real roots of s of the denominator polynomial of G / (1 + G) are negative. DC motor control method.

@ 0001   The brushless DC motor control method according to claim 9, wherein the rotational position of the rotor is estimated using a fixed coordinate motor model.   The brushless DC motor control method according to any one of claims 9 to 15, wherein a motor device constant is identified before starting operation.   The brushless DC motor control method according to any one of claims 9 to 16, wherein the target current phase is advanced to a maximum efficiency phase or a maximum torque phase of the motor.   The brushless DC motor control method according to any one of claims 9 to 17, wherein the brushless DC motor (25) (42) (62) drives a compressor. The rotational position of the rotor is estimated from the magnetic flux information using the motor voltage, the motor current, and the device constants of the brushless DC motors (25), (42), and (62), and the inverter (24) (41) is estimated using this estimation result. (61) is a device for controlling brushless DC motors (25) (42) (62) having saliency,
Inverter control means (29) (29 ') (29'') for controlling the inverter (24) (41) (61) to change the operating point in response to the magnetic flux being substantially zero Brushless DC motor control device.   The inverter control means (29) detects that the magnetic flux is almost zero and performs phase advance control in response to detecting that the magnetic flux is almost zero to avoid a region where the magnetic flux is almost zero. The brushless DC motor control device according to claim 19, wherein   The inverter control means (29 ′) detects that the magnetic flux is almost zero, and performs a current droop control in response to detecting that the magnetic flux is almost zero, so that a region where the magnetic flux becomes almost zero is obtained. The brushless DC motor control device according to claim 19, which is to be avoided.   The brushless DC motor control device according to any one of claims 19 to 21, wherein the inverter control means estimates a rotational position of a rotor using a rotary coordinate motor model. The rotational position of the rotor is estimated using the motor voltage, the motor current, and the device constants of the brushless DC motors (25), (42), and (62), and the inverters (24), (41), and (61) are estimated using the estimation results. Which controls the brushless DC motors (25), (42) and (62),
A brushless DC motor comprising: inverter control means for performing rotational position estimation using a transfer function and setting a gain of a round transfer function relating to rotational position estimation to a value that satisfies a stability condition in a target phase and a torque range Control device.   The inverter control means adopts the transfer function G expressed by Equation 1, and sets the gains Kθ and F (s) of the transfer function G so that all real roots of s in the denominator polynomial of G / (1 + G) are negative. The brushless DC motor control device according to claim 23, which is set.   The brushless DC motor control device according to any one of claims 19 to 21, wherein the inverter control means estimates a rotational position of a rotor using a fixed coordinate motor model.   The brushless DC motor control device according to any one of claims 19 to 25, wherein the inverter control means identifies a motor device constant before starting operation.   The brushless DC motor control device according to any one of claims 19 to 26, wherein the inverter control means advances the target current phase to a maximum efficiency phase or a maximum torque phase of the motor.   The brushless DC motor control device according to any one of claims 19 to 27, wherein the brushless DC motor drives a compressor.

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