JP3636340B2 - Power converter for AC rotating machine - Google Patents

Description

【００２８】
α−β座標系における電流ベクトルの方向は数式２となる。なお、θ_cは電流ベクトルのα軸に対する角度である。
【００２９】
【数２】
θ_c＝ｔａｎ^-1（ｉβ／ｉα）
【００３０】
一方、永久磁石同期電動機２の電動機方程式は、数式３で与えられる。
なお、数式３において、ｖは電動機端子電圧、ｉは電機子電流、Ｐは微分演算子、ωは回転子の回転角速度（電気角）、ψ_fは逆起電力定数、Ｒは電動機の巻線抵抗、Ｌは電動機の巻線インダクタンス、添字ｄ，ｑは各量のｄ軸、ｑ軸成分である。
【００３１】
【数３】

【００３２】
数式３を変形すると、数式４に示すように電流に関する状態方程式を得る。
【００３３】
【数４】

【００３４】
数式４の状態方程式を解くと、数式５を得る。
但し、条件としてωは一定値、Ｒ＝０、ｖとｉの初期値は何れも０とする。実際には、回転子の空転状態で巻線短絡により電流が流れるとトルクが発生し、これがωを変化させるが、短絡期間が短い場合にはωの変化を無視することができるので、ωは一定値として扱って良い。また、短絡期間が短い場合、巻線抵抗による波形の減衰も小さいため、Ｒ＝０と近似することができる。更に、短絡開始時のｖ，ｉの初期値は０となる。
【００３５】
【数５】

【００３６】
数式５によれば、回転子の回転角速度がωであるときに巻線を短絡した場合のｉ_d，ｉ_qを求めることができる。すなわち、図３に示したように巻線を短絡してから一定期間ｔ₀〔ｓ〕経過後に電流を検出する場合、数式５にｔ＝ｔ₀を代入することでｉ_d，ｉ_qを演算することができる。
ここで、回転子の磁極位置をθとすると、数式５からα−β座標系における電流ベクトルの方向は数式６となる。
【００３７】
【数６】
θ_c＝θ＋ｔａｎ^-1（ｉ_q／ｉ_d）
【００３８】
従って、数式２及び数式６により、磁極位置θは数式７から求められることが判る。
【００３９】
【数７】
θ＝ｔａｎ^-1（ｉβ／ｉα）−ｔａｎ^-1（ｉ_q／ｉ_d）
【００４０】
次に、数式７により磁極位置θを求めるにはωを知る必要があるので、これについて説明する。
短絡電流の絶対値Ｉ_sは数式８によって得られる。
【００４１】
【数８】

【００４２】
上記数式８は、０＜ωｔ＜π／２〔ｒａｄ〕の範囲で単調増加関数である。従って、電動機の巻線を短絡してから０＜ωｔ＜π／２〔ｒａｄ〕を満たすような時間後のＩ_sを検出することにより、ωを逆算することができる。
ここで、数式８からωに関して解いた関数は複雑な式になるため、あらかじめωとＩ_sとの関係を調べてテーブルを作成し、制御装置に記憶させておけば、複雑な演算を行わなくても直ちにωを求めることができる。
【００４３】
以上のように、電動機の回転方向が既知であると仮定した場合には、上述した原理で同期電動機２の磁極位置と回転角速度とを知ることができ、これらに基づいて逆起電力の周波数、振幅、位相を求めることができる。従って、インバータ１が発生するべき出力電圧の周波数、振幅、位相がわかり、これらに従った電流制御部７からの指令によりインバータ１を運転すれば、空転中の同期電動機２を再起動することができる。再起動後は、従来から知られている同期電動機２の逆起電力に基づく制御方法に則って駆動すれば良い。
【００４４】
次に、請求項３に記載した発明の実施形態を説明する。
上述した原理を用いて実際に空転状態の電動機２を再起動する場合、巻線を短絡して一定期間後の電流値を検出し、逆起電力の位相と速度とを演算してインバータ１の出力電圧を求めるという一連の動作を短期間で行う必要がある。この期間が長くなると、短絡電流Ｉ_sが増大して過電流状態になり、素子を破壊してしまう。
【００４５】
そこで、巻線を短絡して一定期間ｔ₀後に電流を検出した後、インバータ１の半導体スイッチング素子をすべてターンオフすること（以後、これら一連の動作をゼロ電圧パルス（ＺＶＰ）の印加という）により電流を一旦ゼロとし、その状態で電流検出時の逆起電力の位相及び速度を推定演算して全スイッチング素子がターンオフしてから一定期間ｔ₁後に再起動を行う。
この再起動までの期間ｔ₁が例えば数〔ｍｓ〕と短ければ、回転子の回転速度はその間、一定であると考えられる。
よって、再起動を行う時の磁極位置θ_sは、巻線短絡による電流検出時の磁極位置θ、角速度ω及び期間ｔ₁から、数式９により求めることができる。
【００４６】
【数９】
θ_s＝θ＋ωｔ₁
【００４７】
次いで、請求項４に記載した発明の実施形態を説明する。回転子の回転速度を電流の振幅から求める場合、前述のように複雑な演算を行うか、回転速度と電流振幅との関係を制御回路が記憶しておくかの何れかの方法をとる必要があり、制御装置にかかる負担が大きいという問題がある。
このため、一層容易に回転子の回転速度を推定する方法として、前述したＺＶＰの印加を２回行い、各回で得られた電流ベクトルの方向の差から回転子の回転速度を求めることも可能である。
すなわち、２回のＺＶＰの印加の間隔をｔ₂とし、各回において検出された電流ベクトルの方向の差をδ〔ｒａｄ〕とすると、回転子の回転角速度ωは数式１０によって求めることができる。但し、期間ｔ₂は数〔ｍｓ〕という短い期間であるとする。
【００４８】
【数１０】
ω＝δ／ｔ₂
【００４９】
次に、請求項５に記載した発明の実施形態を説明する。回転子が両方向に回転し得る場合には、回転方向を知る必要がある。これは、上記の実施形態において、ＺＶＰを２回印加する間隔ｔ₂を、回転子が最大回転速度（規定値）にて回転する際に電気角で１８０°進む時間よりも短くすることによって求めることができる。
すなわち、回転子の最大回転角速度をω_maxとすると、間隔ｔ₂の満たすべき条件は数式１１となる。
【００５０】
【数１１】
ｔ₂＜π／ω_max
【００５１】
もし、数式１１の条件が満足されない場合には、例えば回転子が正方向に１６０°回転したときと、逆方向に２００°回転したときとでδの演算結果が等しくなるため、回転方向を判別することはできない。
【００５２】
次いで、請求項６に記載した発明の実施形態を説明する。図４は、巻線短絡時に検出した電流の瞬時ベクトルｉが、３６０°を３０°ごとに分割した扇形領域のうち何れの領域に存在するかを検出する動作を説明するためのものである。
電流ベクトルｉがどの領域に存在するかを検出する手順は以下のとおりである。
（１）手順１
｜ｉα|，｜ｉβ|を演算する。また、ベクトルｉ’を数式１２のように新たに定義する。
【００５３】
【数１２】
｜ｉ’｜＝√（ｉα²＋ｉβ²），
∠ｉ’＝θ_c’＝ｔａｎ^-1｜ｉβ／ｉα｜
【００５４】
（２）手順２
ベクトルｉ’の位相角θ_c’は、次のように判定することができる。
２×｜ｉβ｜＜｜ｉα｜ならば、０°≦θ_c’＜３０°
２×｜ｉα｜＜｜ｉβ｜ならば、６０°＜θ_c’≦９０°
上記２つの条件の何れにも該当しなければ、３０°≦θ_c’≦６０°
【００５５】
（３）手順３
手順２により判定したθ_c’と、ｉα，ｉβの極性とに基づいて、ベクトルｉの位相角θ_cとθ_c’とを次のように関係付けることができる。
ｉα＞０，ｉβ＞０ならば、θ_c＝θ_c’
ｉα＞０，ｉβ＜０ならば、θ_c＝−θ_c’
ｉα＜０，ｉβ＜０ならば、θ_c＝１８０°＋θ_c’
ｉα＜０，ｉβ＞０ならば、θ_c＝１８０°−θ_c’
【００５６】
以上の手順１〜手順３により、図４においてベクトルｉが存在する領域を特定することができる。再起動のための磁極位置の初期値θ_sを演算する場合、θ_cが存在する扇形領域内の適当な値、例えば扇形領域の中心となる角度（例：０°〜３０°の領域では１５°）をθ_cとして設定すればよい。
その場合の実際値との誤差は、上記の例では±１５°となる。なお、ｉα,ｉβ及びそれらの絶対値に関する条件判定において等号が成り立つ場合には、ベクトルｉは各扇形領域の境界線上にあることは言うまでもない。
【００５７】
次いで、請求項７に記載した発明の実施形態を説明する。
上述した請求項６記載の発明の実施形態のように、磁極位置を正確な値でなくベクトルｉの存在する領域内のある角度として決める場合、誤差が発生する。これは、再起動時の磁極位置の初期値（推定値）θ_sの誤差要因となるため、再起動時の動作の乱れを誘発し、最悪の場合には再起動に失敗して回転子の回転が制御不能となる。
特に、請求項４記載の発明の実施形態で説明したように、いわゆるＺＶＰの印加を２回行って回転子の回転角速度ωと磁極位置の初期値θ_sとを算出する場合、その両者に誤差が生じるため影響が非常に大きくなる。
【００５８】
そこで、請求項７の発明では、ＺＶＰの印加を２回ではなく３回以上行うことにより、ω及びθ_sの推定誤差を低減するようにした。なお、ＺＶＰの複数回の印加は、例えば数〔ｍｓ〕といった短い期間に行われるものとし、その間、回転子の回転速度は一定値と見なせるものとする。
以下、本発明の実施形態を図５を参照しつつ説明する。第ｎ回目のＺＶＰ印加による磁極位置推定値をθ_en、その時の磁極位置の実際値をθ_n、磁極位置を決めるために３６０°を等間隔で複数に分割した扇形領域の中心角を２δとし、その中心に磁極位置を決めるものとすると、数式１３が成り立つ。
【００５９】
【数１３】
θ_n−δ＜θ_en＜θ_n＋δ
【００６０】
ここで、ＺＶＰを印加する間隔を一定値ｔ₂とする。第ｎ回目のＺＶＰ印加時の回転子の角速度推定値をω_enとすると、数式１４が成り立つ。但し、数式１４において、ｎ≧３である。
【００６１】
【数１４】
ω_en＝θ_en−θ_e(n-1)／ｔ₂
【００６２】
ｎ回のＺＶＰ印加によって得られたｎ−１個のωの推定値の平均値をω_avrとし、これを再起動時の回転子の回転角速度推定値として用いるものとする。ω_avrに関しては、数式１５が成り立つ。
【００６３】
【数１５】

【００６４】
数式１３から、θ_enの最大誤差は±δであるので、数式１５から、ω_avrのとり得る最大値及び最小値は数式１６のようになる。
【００６５】
【数１６】
（θ_n−θ₂）／｛（ｎ−１）ｔ₂｝±２δ／｛（ｎ−１）ｔ₂｝
【００６６】
数式１６の第１項がωの真値、第２項が誤差である。数式１６から、ＺＶＰ印加の回数ｎを増加させることによってω_avrの誤差を低減できることが判る。
第ｎ回目のＺＶＰ印加後、ｔ₁を経過してから再起動を行うものとすれば、前述した請求項３の実施形態の数式９に準じて、再起動時の磁極位置の初期値は数式１７のように導出される。
【００６７】
【数１７】
θ_s＝θ_en＋ω_avrｔ₁
【００６８】
従って、ωの推定誤差を低減することにより、再起動時の磁極位置θ_sの誤差をも低減することができる。
【００６９】
請求項８に記載した発明の実施形態を説明する。この実施形態は、ωの推定誤差を低減する別の方法に関する。
前述の数式１３、数式１４から、ω_enのとり得る最大値、最小値は、数式１８のようになる。
【００７０】
【数１８】
（θ_n−θ_n-1）／ｔ₂±２δ／ｔ₂
【００７１】
数式１８の第１項がωの真値、第２項が誤差である。δは一定なので、ＺＶＰの印加間隔ｔ₂を大きくすることによってωの推定誤差を小さくすることができる。
但し、請求項５の発明の実施形態で説明したように、回転子が両方向に回転し得る場合には回転方向を知る必要があり、そのためには、ＺＶＰ印加の間隔ｔ₂は前述した数式１１を満たす必要がある。
【００７２】
従って、３回以上のＺＶＰ印加のうち、少なくとも１回は、ＺＶＰ印加の間隔ｔ₂を数式１１を満足するような値にして回転方向を判定する必要がある。そして、他の少なくとも１回は、ＺＶＰ印加の間隔ｔ₂を回転子が最大回転数で回転している時に電気角で１８０°移動する期間以上に設定することにより、ωの推定誤差を低減させればよい。
【００７３】
最後に、請求項９に記載した発明の実施形態を説明する。
巻線の短絡期間ｔ₀が一定である場合、回転速度が大きいほどＺＶＰ印加時の電流値は大きくなる。従って、ｔ₀は最大回転速度で回転子が回転している場合においても巻線電流が過電流とならないような値（ｔ₀₀）に決定する。これにより、回転速度が小さい場合にはＺＶＰ印加時に流れる電流が小さくなる。
【００７４】
一方、電流検出値にはノイズが重畳しているので、電流検出値が小さくなると相対的にノイズの影響が大きくなり、その結果、磁極位置及び回転速度の推定誤差が大きくなるという問題を生じる。
これを解決するために、まず過電流を防止するように設定した短絡期間ｔ₀₀によってＺＶＰ印加を行い、このときに検出された電流ベクトルの振幅が設定値よりも小さい場合には短絡期間をｔ₀₀よりも長くしてＺＶＰ印加を再度行う。
これにより、低速回転時でも電流検出値の振幅を十分に大きくすることができ、磁極位置及び回転速度の高精度な推定が可能になる。
【００７５】
なお、上記実施形態では本発明をもっぱら永久磁石同期電動機の駆動装置に適用した場合につき説明したが、先に述べたように、本発明は、インバータが停止していて端子電圧が印加されていない状態で回転子が空転することにより電動機の固定子巻線に逆起電力（誘導起電力）が生じているような電動機一般の駆動装置に適用可能である。
【００７６】
更に、本発明は、例えば風力発電機等の交流発電機が発生する交流電力を、電力変換器により直流電力に変換して直流電源に回生するシステムをも含むものである。その場合の回路構成は、例えば図１における同期電動機２を発電機に置き換えれば良い。
【００７７】
【発明の効果】
以上のように請求項１，２記載の発明によれば、例えばインバータにより駆動される位置・速度センサレス方式の電動機において、空転状態から再起動する際に、インバータの半導体スイッチング素子の少なくとも一つ、または全相のスイッチング素子をオンさせて巻線を短絡し、その時に流れる電流値から回転子の位置を検出するため、位置・速度センサレスの駆動方式においても空転状態での再起動が可能になる。また、電動機の端子電圧を検出して回転子の位置を推定する必要がないから、高価な電圧検出器が不要になり、コストの上昇を招くこともない。
【００７８】
更に、請求項３記載の発明では、巻線短絡後にゼロ電圧パルスを印加して電流を一旦ゼロにした状態で回転子の位置を推定演算するため、短絡による過電流が長期にわたって流れるのを回避して素子を保護することができる。
【００７９】
請求項４記載の発明では、ゼロ電圧パルスを２回印加し、各回の電流位相差から回転速度及び回転子位置を推定するようにしたので、回転速度の推定演算が一層容易になる。
【００８０】
請求項５記載の発明では、請求項４の発明における２回のゼロ電圧パルスの印加間隔を、回転子が最大回転数で回転しているときに電気角で１８０°移動する期間よりも短くすることにより、回転子位置の最大変化量が１８０°以下になるので、回転方向を容易に特定することができる。
【００８１】
請求項６記載の発明によれば、巻線電流の瞬時ベクトルが複数の扇形領域のうちの何れの領域に存在するかを判定して再起動のための磁極位置の初期値を決定することができる。
【００８２】
請求項７または請求項８記載の発明では、請求項６の発明に起因する磁極位置の誤差や回転角周波数の誤差を低減することにより、再起動の確実性が向上する。
【００８３】
請求項９記載の発明によれば、低速回転時においても、過電流にならない範囲で電流検出値の大きさをノイズの影響を受けない程度に大きくでき、これによって磁極位置及び回転速度を高精度に推定することが可能になる。
【図面の簡単な説明】
【図１】請求項１に記載した発明の構成を例示的に示すブロック図である。
【図２】本発明の実施形態を示すブロック図である。
【図３】図２の実施形態における再起動手順を示すフローチャートである。
【図４】巻線短絡時に検出した電流の瞬時ベクトルの存在領域を説明するための図である。
【図５】請求項７に記載した発明の実施形態を説明するための図である。
【図６】従来の磁極位置センサ付き永久磁石同期電動機の駆動装置を示すブロック図である。
【図７】永久磁石回転子及びα−β座標系、ｄ−ｑ座標系の説明図である。
【図８】従来の磁極位置・速度センサレス方式の永久磁石同期電動機の駆動装置を示すブロック図である。
【符号の説明】
１ インバータ
２ 永久磁石同期電動機
３ｕ，３ｖ，３ｗ 電流検出器
４ｕ，４ｖ，４ｗ 電流検出器ゲイン
５ 相数変換部
６ 座標変換部
７ 電流制御部
８ 速度演算部
９ 磁極位置推定部（定常時）
１０ 磁極位置推定部（起動時）
１１ 状態制御部
１２ 巻線短絡部
１３ ゲート信号切替部
２０ 直流電源
３０ 永久磁石回転子
Ａ 空転再起動制御部[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a position / speed sensorless type power converter for an AC rotating machine that operates an AC rotating machine without using a rotor position sensor, a speed sensor, and a terminal voltage sensor. For example, the inverter is stopped and the terminal is stopped. It may be a drive device for a permanent magnet synchronous motor in which a counter electromotive force (inductive electromotive force) is generated in a stator winding of an electric motor due to idling of a rotor in a state where no voltage is applied.
[0002]
[Prior art]
In the following conventional technology, a drive device for a permanent magnet synchronous motor is exemplified.
FIG. 6 shows a drive device for a permanent magnet synchronous motor with a magnetic pole position sensor. In the figure, 20 is a DC power source, 1 is an inverter having a parallel circuit composed of a semiconductor switching element and an anti-parallel diode in upper and lower arms for three phases, 2 is a permanent magnet synchronous motor (PM motor) having a permanent magnet rotor, 3u, 3v and 3w are current detectors for detecting the stator winding current of each phase, 4u, 4v and 4w are current detector gains, and 5 is a detected phase current i._u, I_v, I_wIs converted to two-phase components iα and iβ, and 6 is a magnetic flux direction component of the two-phase components iα and iβ based on the rotor magnetic pole position (magnetic flux direction produced by the magnetic pole) θ detected by the magnetic pole position sensor 14. i_dAnd a directional component i orthogonal thereto_qThe coordinate conversion unit 7 for converting to the rotation angle velocity ω calculated by inputting the magnetic pole position θ to the velocity calculation unit 8 and the magnetic flux direction component i_dAnd its orthogonal component i_qIs a current control unit that generates a gate signal of the semiconductor switching element of the inverter 1 and controls the current of the synchronous motor 2 to a predetermined value.
Here, FIG. 7 is an explanatory diagram of the permanent magnet rotor and the α-β coordinate system which is a stationary coordinate system and the dq coordinate system which is a rotary coordinate system, and θ is the magnetic pole position of the permanent magnet rotor 30. .
[0003]
In this drive device, since the magnetic pole position θ of the rotor can always be detected by the magnetic pole position sensor 14, even if the rotor 30 is idling while the inverter 1 is stopped, the inverter 1 and the rotor 1 and the rotor 1 according to the detected magnetic pole position θ. It is possible to restart the synchronous motor 2.
[0004]
On the other hand, FIG. 8 shows a drive device for a permanent magnet synchronous motor of a magnetic pole position / speed sensorless system. In addition, the same code | symbol is attached | subjected to the component same as FIG.
This sensorless drive device is described in, for example, a paper “Sensorless salient pole type brushless DC motor control based on speed electromotive force estimation” published in the January 1997 issue of the IEEJ Transaction D “Industrial Applications”. ing.
[0005]
The drive apparatus of FIG. 8 is provided with a magnetic pole position estimation unit 9, and the magnetic flux direction component v of the terminal voltage of the synchronous motor 2 output from the current control unit 7._dAnd orthogonal component v_qAnd the magnetic flux direction component i of the armature current output from the coordinate conversion unit 6_dAnd orthogonal component i_qBased on the above, the magnetic pole position estimation unit 9 estimates the magnetic pole position θ of the rotor.
[0006]
Here, the terminal voltage of the synchronous motor 2 can be detected by using a voltage detector, but in many cases, the synchronous motor is directly connected to the inverter, so that the terminal voltage is equal to the output voltage of the inverter. You can think about it. Therefore, when the actual output voltage value of the inverter can be regarded as the command value, the output voltage command value of the inverter can be replaced with the terminal voltage of the synchronous motor and used to estimate the magnetic pole position θ. Therefore, the method of FIG. 8 does not require a voltage detector for detecting the terminal voltage, and the necessary detection amount is only the armature current of the synchronous motor 2.
[0007]
[Problems to be solved by the invention]
However, in the method of FIG. 8, no armature current flows when the inverter 1 is stopped, that is, when all the semiconductor switching elements are off, so that no information on the magnetic pole position θ is obtained. For this reason, when the inverter 1 is stopped and the rotor 30 is idling, the magnetic pole position θ cannot be known, so that the synchronous motor 2 cannot be restarted via the inverter 1.
If a voltage detector that detects the terminal voltage of the synchronous motor 2 is used, the phase and frequency of the back electromotive force of the motor can be known, and hence the magnetic pole position can be estimated. However, this voltage detector is generally expensive. It is.
[0008]
Therefore, in the sensorless type driving apparatus as shown in FIG. 8, in order to perform restart without the voltage detector, for example, restart in a state where the inverter is stopped and the rotor is idle, It was very inconvenient because I had to wait until the rotor stopped.
Therefore, the present invention intends to provide a power converter for an AC rotating machine such as a motor driving device, which can be restarted in a state where the rotor is idling without using an expensive voltage detector or the like. Is.
[0009]
[Means for Solving the Problems]
In order to solve the above-mentioned problems, a first aspect of the present invention provides a power converter that operates an AC rotating machine, and a semiconductor switching element that constitutes the power converter to control the winding current of the rotating machine to a predetermined value. And a control device for generating and outputting an on / off signal for the AC rotating machine, and at least one of the semiconductor switching elements by the control device when the rotor of the rotating machine idles Is turned on, the winding of the rotating machine is short-circuited, the position of the rotor is estimated based on the winding current flowing at that time, and the power converter is restarted.
[0010]
FIG. 1 is a block diagram exemplarily showing the configuration of the invention according to claim 1 described above, and the same components as those in FIG. 8 are denoted by the same reference numerals.
In the following description, a drive device for a permanent magnet synchronous motor is illustrated. However, as described above, in the present invention, the rotor rotates idly when the power converter is stopped and no terminal voltage is applied. Applicable to AC motor drives where back electromotive force (inductive electromotive force) is generated in the stator windings, and power converters that convert the generated power of the AC generator into direct current and regenerate it into a direct current power supply It is.
In FIG. 1, an idling restart control unit A and a gate signal switching unit 13 are provided in the control device of the inverter 1. The idling restart control unit A receives the two-phase components iα and iβ of the winding current (armature current) output from the phase number conversion unit 5 and outputs the switching elements of the inverter 1 at the idling restart. Is a gate signal. Then, the gate signal switching unit 13 operates to switch between the steady gate signal from the current control unit 7 and the gate signal at the idling restart by the switching control signal s from the idling restart control unit A.
[0011]
The operation of the present invention will be described below. Since the two-phase components iα and iβ of the winding current are input to the idling restart control unit A, the idling restart control unit A is in a state where the inverter 1 is stopped and the current of the synchronous motor 2 is zero. Can be detected.
At this time, the idling restart control unit A generates and outputs a gate signal that turns on at least one of the semiconductor switching elements of the inverter 1, and simultaneously the idling restart of the gate signal switching unit 13 by the switching control signal s. When switching to the control unit A side, at least one of the semiconductor switching elements is turned on, and at least one phase of the stator winding of the electric motor 2 is short-circuited.
At this time, if the rotor 30 is idling, a current flows through the winding by the action of the counter electromotive force. Since this current depends on the magnetic pole position and the rotational speed, it is possible to determine the magnetic pole position and the rotational speed from the flowing current value.
[0012]
As described above, the idling restart control unit A turns on at least one of the semiconductor switching elements of the inverter 1 to turn on the stator winding when the rotor is idling with the winding current being zero. Is operated in a short-circuit state so that a winding current caused by a back electromotive force flows. Then, the current at this time is detected, and the magnetic pole position and the rotation speed are calculated from the current value. Furthermore, the initial value of a current command or the like for performing normal control is set from the obtained calculation result, and this is transferred to the current control unit 7 or the like to control the inverter 1, thereby restarting the apparatus. Executed.
[0013]
The invention described in claim 1 is further embodied by the following claims 2-9.
That is, the invention according to claim 2 is the power converter for an AC rotating machine according to claim 1, wherein when the rotor of the rotating machine is idling, all the phase windings of the rotating machine are short-circuited by the power converter. The position of the rotor is estimated based on the winding current flowing through the rotor.
[0014]
According to a third aspect of the present invention, in the power converter for an AC rotating machine according to the first or second aspect, the winding current is detected after a certain period of time has elapsed since the winding current is detected by short-circuiting the winding of the rotating machine. The power converter is switched so as to be zero. Thereby, it is possible to prevent overcurrent from flowing for a long period of time.
[0015]
According to a fourth aspect of the present invention, in the power converter for an AC rotating machine according to the third aspect, the operation of making the winding current zero by switching the power converter is performed twice, and a short circuit detected at each time The rotational speed of the rotor is calculated on the basis of the phase difference at which the winding current becomes maximum, that is, the phase difference of the current vectors. As a result, the rotational speed of the rotor can be estimated and calculated without requiring complicated calculations or storing a large amount of data.
[0016]
According to a fifth aspect of the present invention, there is provided the power converter for an AC rotating machine according to the fourth aspect of the present invention, wherein the rotor is set at the maximum number of rotations so that the winding current is zero by switching the power converter. This is set to be shorter than the period of 180 ° movement in electrical angle when rotating. Thereby, it becomes easy to specify the rotation direction of the rotor.
[0017]
According to a sixth aspect of the present invention, in the power converter for an AC rotating machine according to the third aspect, the phase of the instantaneous vector of the detected winding current is expressed by a sector region formed by dividing the vector into 360 °. It is estimated according to which region it is in.
[0018]
According to a seventh aspect of the present invention, in the power converter for an AC rotating machine according to the sixth aspect, the operation of making the winding current zero by switching the power converter is performed three times or more, and a short circuit detected at each time The position, rotation direction, and rotation speed of the rotor are estimated using the winding current at the time.
[0019]
According to an eighth aspect of the present invention, in the power converter for an AC rotating machine according to the seventh aspect, at least one of the intervals of the three or more operations in which the winding current is zero is set at the maximum rotation of the rotor. Set shorter than the period of 180 ° electrical angle when rotating at a number, and at least one other interval is moved 180 ° at electrical angle when the rotor is rotating at maximum speed It is set to be longer than the period.
[0020]
According to a ninth aspect of the present invention, in the power converter for an AC rotating machine according to the third aspect, when the detected winding current is smaller than a predetermined specified value within a range that does not become an overcurrent, the winding The winding current exceeding the specified value is detected again by setting the short-circuit period for a long time, and after that, after a certain period of time, the power converter is switched so that the winding current becomes zero.
[0021]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, embodiments of the present invention will be described with reference to the drawings.
FIG. 2 is a block diagram showing an embodiment of the present invention. The same components as those in FIG. 1 are denoted by the same reference numerals, and different portions will be mainly described below.
In FIG. 2, the magnetic pole position estimation unit (at startup) 10, the state control unit 11, and the winding short-circuit unit 12 correspond to the idling restart control unit A in FIG. 1. Among them, the magnetic pole position estimation unit (at startup) 10 receives the two-phase components iα and iβ of the current output from the phase number conversion unit 5 and the control signal a from the state control unit 11. Magnetic pole position estimate θ_iIs output to the magnetic pole position estimator 9 (steady state), and the rotational angular velocity estimated value ω_iIs output to the current control unit 7.
The state control unit 11 outputs a switching control signal s and a control signal b. The switching control signal s is applied to the gate signal switching unit 13 and the control signal b is applied to the winding short-circuit unit 12 as described above. .
[0022]
The state control unit 11 detects the stop state of the inverter 1 and sends a control signal b to the winding short-circuit unit 12 to turn on at least one of the semiconductor switching elements of the inverter 1 to short-circuit the stator winding. In addition, the control signal a is sent to the magnetic pole position estimation unit (at startup) 10 to restart the magnetic pole position θ at the time of restart._iAnd rotational angular velocity ω_iThe estimation calculation is executed.
[0023]
Next, the restart procedure according to the present embodiment will be described with reference to the flowchart of FIG. This procedure is executed centering on the state control unit 11 and corresponds to an embodiment of the invention described in claim 2.
When the inverter 1 is stopped and the rotor is idling (S1), when the switching control signal s is sent from the state control unit 11 to the gate signal switching unit 13, the gate signal line to the inverter 1 is wound. It switches to the short circuit part 12 side.
Next, when all the semiconductor switching elements of the upper arm or the lower arm of each phase of the inverter 1 are turned on by the winding short-circuit unit 12, the stator windings of all the phases of the motor 2 are short-circuited, and the counter electromotive force (inductive electromotive force) ) Causes a current to flow through the winding (S2).
The magnetic pole position estimation unit (at startup) 10₀[S] The winding current (two-phase components iα, iβ) after elapse is detected (S3), and the magnetic pole position and the rotational angular velocity are calculated based on this (S4)._iAnd rotational angular velocity ω_iAre estimated and calculated (S5). Of these, the estimated magnetic pole position θ_iIs the estimated rotational angular velocity value ω_iAre sent to the current control unit 7 as initial values, and the gate signal from the current control unit 7 switched by the gate signal switching unit 13 by the switching control signal s is sent to the inverter 1 and restarted (S6). .
[0024]
Now, in the present embodiment, a method for estimating the magnetic pole position and the rotational angular velocity based on the current when the winding is short-circuited will be described.
In order to restart from the idling state, it is necessary to make the amplitude, phase and frequency of the output voltage of the inverter 1 substantially equal to those of the counter electromotive force of the synchronous motor 2 at the time of restart. Otherwise, a large potential difference is generated between the synchronous motor 2 and the inverter 1, and an overcurrent state occurs and cannot be started correctly. Also, if the rotor can rotate in both directions, it is necessary to know the direction of rotation. Here, since the amplitude of the counter electromotive force is proportional to the frequency of the counter electromotive force, that is, the rotational speed of the rotor, it can be obtained immediately if the amplitude of the counter electromotive force can be known.
Therefore, what is necessary for restarting is the frequency and phase of the counter electromotive force, that is, the magnetic pole position and rotational speed of the rotor, and the rotational direction.
[0025]
First, a method for estimating the magnetic pole position will be described. Here, the case where the permanent magnet synchronous motor 2 is a three-phase two-pole motor will be described. However, the following description is similarly applied to the case where the number of phases and the number of poles of the synchronous motor are other than the three-phase two-pole motor.
Now, the coordinate system related to the synchronous motor is defined as shown in FIG. That is,
Dq coordinate system: a coordinate system in which the direction of the N pole of the rotor 30 is the d axis, and the direction orthogonal to the d axis is the q axis.
.Alpha .-. Beta. Coordinate system: a coordinate system in which the direction in which magnetic flux is generated when a positive current is passed through the u-phase winding is the .alpha. Axis and the direction orthogonal to the .alpha. Axis is the .beta. Axis.
[0026]
The current that flows by short-circuiting the windings during idling of the rotor is detected as a three-phase current. u, v, w phase current i_u, I_v, I_wThen, iα and iβ, which are two-phase components, can be obtained by Equation 1.
[0027]
[Expression 1]

[0028]
The direction of the current vector in the α-β coordinate system is expressed by Equation 2. Θ_cIs the angle of the current vector with respect to the α-axis.
[0029]
[Expression 2]
θ_c= Tan^-1(Iβ / iα)
[0030]
On the other hand, the motor equation of the permanent magnet synchronous motor 2 is given by Equation 3.
In Equation 3, v is the motor terminal voltage, i is the armature current, P is the differential operator, ω is the rotational angular velocity (electrical angle) of the rotor, ψ_fIs the back electromotive force constant, R is the winding resistance of the motor, L is the winding inductance of the motor, and the subscripts d and q are the d-axis and q-axis components of each quantity.
[0031]
[Equation 3]

[0032]
When Equation 3 is transformed, a state equation relating to current is obtained as shown in Equation 4.
[0033]
[Expression 4]

[0034]
When the state equation of Equation 4 is solved, Equation 5 is obtained.
However, as a condition, ω is a constant value, R = 0, and the initial values of v and i are both 0. Actually, torque is generated when a current flows due to a short circuit of the winding in the idling state of the rotor, and this changes ω. It can be treated as a constant value. Further, when the short-circuit period is short, the waveform attenuation due to the winding resistance is also small, so that R = 0 can be approximated. Furthermore, the initial values of v and i at the start of the short circuit are zero.
[0035]
[Equation 5]

[0036]
According to Equation 5, i when the winding is short-circuited when the rotational angular velocity of the rotor is ω._d, I_qCan be requested. That is, as shown in FIG.₀[S] When current is detected after elapse, t = t₀By substituting i_d, I_qCan be calculated.
Here, assuming that the magnetic pole position of the rotor is θ, the direction of the current vector in the α-β coordinate system from Equation 5 is Equation 6.
[0037]
[Formula 6]
θ_c= Θ + tan^-1(I_q/ I_d)
[0038]
Therefore, it can be seen from Equation 2 and Equation 6 that the magnetic pole position θ is obtained from Equation 7.
[0039]
[Expression 7]
θ = tan^-1(Iβ / iα) -tan^-1(I_q/ I_d)
[0040]
Next, since it is necessary to know ω in order to obtain the magnetic pole position θ using Equation 7, this will be described.
Absolute value of short circuit current I_sIs obtained by Equation 8.
[0041]
[Equation 8]

[0042]
Equation 8 is a monotonically increasing function in the range of 0 <ωt <π / 2 [rad]. Therefore, after short-circuiting the windings of the motor, I after a time such that 0 <ωt <π / 2 [rad] is satisfied._sΩ can be calculated backward by detecting.
Here, since the function solved for ω from Equation 8 is a complicated equation, ω and I_sIf a table is created by checking the relationship between and stored in the control device, ω can be obtained immediately without performing complicated calculations.
[0043]
As described above, when it is assumed that the rotation direction of the motor is known, the magnetic pole position and the rotation angular velocity of the synchronous motor 2 can be known based on the above-described principle, and based on these, the frequency of the counter electromotive force, Amplitude and phase can be obtained. Therefore, the frequency, amplitude, and phase of the output voltage that should be generated by the inverter 1 can be known, and if the inverter 1 is operated in accordance with a command from the current control unit 7 according to these, the idling synchronous motor 2 can be restarted. it can. After restarting, it may be driven in accordance with a conventionally known control method based on the counter electromotive force of the synchronous motor 2.
[0044]
Next, an embodiment of the invention described in claim 3 will be described.
When actually restarting the idling motor 2 using the above-described principle, the winding is short-circuited, the current value after a certain period is detected, the phase and speed of the counter electromotive force are calculated, and the inverter 1 It is necessary to perform a series of operations for obtaining the output voltage in a short period of time. As this period increases, the short-circuit current I_sIncreases to an overcurrent state, destroying the device.
[0045]
Therefore, the winding is short-circuited for a certain period t.₀After detecting the current later, all the semiconductor switching elements of the inverter 1 are turned off (hereinafter, a series of operations is referred to as application of a zero voltage pulse (ZVP)), and the current is once set to zero. The period and time after all the switching elements are turned off by estimating and calculating the phase and speed of the back electromotive force₁Reboot later.
Period t until this restart₁Is as short as several [ms], for example, it is considered that the rotational speed of the rotor is constant during that time.
Therefore, the magnetic pole position θ when restarting_sIs the magnetic pole position θ, the angular velocity ω, and the period t at the time of current detection due to a winding short circuit₁From the above, it can be obtained by Equation 9.
[0046]
[Equation 9]
θ_s= Θ + ωt₁
[0047]
Next, an embodiment of the invention described in claim 4 will be described. When calculating the rotation speed of the rotor from the amplitude of the current, it is necessary to take either the complicated calculation as described above or the control circuit to store the relationship between the rotation speed and the current amplitude. There is a problem that the burden placed on the control device is large.
Therefore, as a method of estimating the rotation speed of the rotor more easily, it is possible to apply the ZVP twice as described above and obtain the rotation speed of the rotor from the difference in the direction of the current vector obtained each time. is there.
That is, the interval between two ZVP applications is t₂Assuming that the difference in the direction of the current vector detected at each time is δ [rad], the rotational angular velocity ω of the rotor can be obtained by Equation 10. However, period t₂Is a short period of several [ms].
[0048]
[Expression 10]
ω = δ / t₂
[0049]
Next, an embodiment of the invention described in claim 5 will be described. If the rotor can rotate in both directions, it is necessary to know the direction of rotation. In the above embodiment, this is the interval t at which ZVP is applied twice.₂Can be obtained by making it shorter than the time to advance 180 ° in electrical angle when the rotor rotates at the maximum rotation speed (specified value).
That is, the maximum rotational angular velocity of the rotor is_maxThen the interval t₂The condition to be satisfied is expressed by Equation 11.
[0050]
## EQU11 ##
t₂<Π / ω_max
[0051]
If the condition of Expression 11 is not satisfied, for example, the calculation result of δ is equal when the rotor is rotated 160 ° in the forward direction and when it is rotated 200 ° in the reverse direction. I can't do it.
[0052]
Next, an embodiment of the invention described in claim 6 will be described. FIG. 4 is a diagram for explaining an operation for detecting in which of the fan-shaped regions obtained by dividing 360 ° every 30 ° the instantaneous vector i of the current detected when the winding is short-circuited.
The procedure for detecting in which region the current vector i exists is as follows.
(1) Procedure 1
| Iα | and | iβ | are calculated. Also, the vector i 'is newly defined as in Expression 12.
[0053]
[Expression 12]
| I ’| = √ (iα²+ Iβ²),
∠i ’= θ_c‘= Tan^-1| Iβ / iα |
[0054]
(2) Procedure 2
Phase angle θ of vector i '_c'Can be determined as follows.
If 2 × | iβ | <| iα |, 0 ° ≦ θ_c‘<30 °
If 2 × | iα | <| iβ |, then 60 ° <θ_c′ ≦ 90 °
If neither of the above two conditions is met, 30 ° ≦ θ_c′ ≦ 60 °
[0055]
(3) Procedure 3
Θ determined by procedure 2_c′ And the polarities of iα and iβ, the phase angle θ of the vector i_cAnd θ_cCan be related as follows.
If iα> 0 and iβ> 0, θ_c= Θ_c’
If iα> 0 and iβ <0, θ_c= −θ_c’
If iα <0, iβ <0, θ_c= 180 ° + θ_c’
If iα <0, iβ> 0, θ_c= 180 ° -θ_c’
[0056]
By the above procedure 1 to procedure 3, the region where the vector i exists in FIG. 4 can be specified. Initial value θ of magnetic pole position for restart_sWhen calculating_cAn appropriate value in the sector area where the angle exists, for example, an angle that becomes the center of the sector area (e.g., 15 degrees in an area of 0 ° to 30 °) is θ_cCan be set as
The error from the actual value in that case is ± 15 ° in the above example. Needless to say, the vector i is on the boundary line of each sector area when the equal sign holds in the condition determination regarding iα, iβ and their absolute values.
[0057]
Next, an embodiment of the invention described in claim 7 will be described.
As in the embodiment of the invention described in claim 6 described above, when the magnetic pole position is determined not as an accurate value but as an angle in the region where the vector i exists, an error occurs. This is the initial value (estimated value) θ of the magnetic pole position at restart_sTherefore, the disturbance of the operation at the time of restart is induced, and in the worst case, the restart fails and the rotation of the rotor becomes uncontrollable.
In particular, as described in the embodiment of the invention described in claim 4, the so-called ZVP is applied twice, and the rotational angular velocity ω of the rotor and the initial value θ of the magnetic pole position_sWhen calculating the above, the effect is very large because an error occurs in both of them.
[0058]
Therefore, in the invention of claim 7, by applying ZVP three times or more instead of twice, ω and θ_sThe estimation error was reduced. It is assumed that the ZVP is applied a plurality of times in a short period, for example, a few [ms], and the rotational speed of the rotor can be regarded as a constant value during that period.
Hereinafter, an embodiment of the present invention will be described with reference to FIG. The estimated magnetic pole position by the nth ZVP application is θ_en, The actual value of the magnetic pole position at that time_nAssuming that the central angle of a sector region obtained by dividing 360 ° into a plurality of equal intervals in order to determine the magnetic pole position is 2δ and the magnetic pole position is determined at the center, Equation 13 is established.
[0059]
[Formula 13]
θ_n−δ <θ_en<Θ_n+ Δ
[0060]
Here, the interval at which ZVP is applied is set to a constant value t.₂And The estimated angular velocity of the rotor when the nth ZVP is applied is ω_enThen, Formula 14 is established. However, in Expression 14, n ≧ 3.
[0061]
[Expression 14]
ω_en= Θ_en−θ_{e (n-1)}/ T₂
[0062]
The average value of n-1 estimated values of ω obtained by applying ZVP n times is expressed as ω_avrAnd this is used as the estimated rotational angular velocity of the rotor at the time of restart. ω_avrWith respect to, Equation 15 holds.
[0063]
[Expression 15]

[0064]
From Equation 13, θ_enSince the maximum error of δ is ± δ, from Equation 15, ω_avrThe maximum value and the minimum value that can be taken are given by Equation 16.
[0065]
[Expression 16]
(Θ_n−θ₂) / {(N-1) t₂} ± 2δ / {(n−1) t₂}
[0066]
The first term of Equation 16 is the true value of ω, and the second term is the error. From Equation 16, ω is increased by increasing the number n of ZVP applications._avrIt can be seen that the error can be reduced.
After nth ZVP application, t₁If the restart is performed after elapse of, the initial value of the magnetic pole position at the time of restart is derived as shown in Expression 17 in accordance with Expression 9 of the embodiment of claim 3 described above.
[0067]
[Expression 17]
θ_s= Θ_en+ Ω_avrt₁
[0068]
Therefore, by reducing the estimation error of ω, the magnetic pole position θ_sThis error can also be reduced.
[0069]
An embodiment of the invention described in claim 8 will be described. This embodiment relates to another method for reducing the estimation error of ω.
From Equation 13 and Equation 14 above, ω_enThe maximum value and the minimum value that can be taken are as shown in Equation 18.
[0070]
[Expression 18]
(Θ_n−θ_n-1) / T₂± 2δ / t₂
[0071]
The first term of Equation 18 is the true value of ω, and the second term is the error. Since δ is constant, the ZVP application interval t₂The estimation error of ω can be reduced by increasing.
However, as described in the embodiment of the invention of claim 5, when the rotor can rotate in both directions, it is necessary to know the rotation direction, and for that purpose, the ZVP application interval t is required.₂Needs to satisfy Expression 11 described above.
[0072]
Accordingly, at least one of the three or more ZVP applications is at least one ZVP application interval t.₂It is necessary to determine the rotation direction with a value satisfying Equation 11. And at least once, the ZVP application interval t₂Is set to be equal to or longer than a period in which the rotor moves 180 ° in electrical angle when the rotor is rotating at the maximum rotation speed, the estimation error of ω may be reduced.
[0073]
Finally, an embodiment of the invention described in claim 9 will be described.
Winding short-circuit period t₀Is constant, the current value when ZVP is applied increases as the rotational speed increases. Therefore, t₀Is a value such that the winding current does not become an overcurrent even when the rotor is rotating at the maximum rotational speed (t₀₀). As a result, when the rotational speed is low, the current that flows when ZVP is applied becomes small.
[0074]
On the other hand, since noise is superimposed on the current detection value, when the current detection value becomes small, the influence of noise becomes relatively large, resulting in a problem that the estimation error of the magnetic pole position and the rotation speed becomes large.
In order to solve this, first, the short-circuit period t set so as to prevent overcurrent.₀₀When ZVP application is performed by the above, and the amplitude of the current vector detected at this time is smaller than the set value, the short-circuit period is set to t₀₀The ZVP application is performed again after a longer time.
As a result, the amplitude of the current detection value can be sufficiently increased even during low-speed rotation, and the magnetic pole position and rotation speed can be estimated with high accuracy.
[0075]
In the above embodiment, the case where the present invention is applied exclusively to the drive device for a permanent magnet synchronous motor has been described. However, as described above, the present invention is such that the inverter is stopped and no terminal voltage is applied. The present invention can be applied to a general drive device of an electric motor in which a counter electromotive force (inductive electromotive force) is generated in a stator winding of the electric motor due to idling of the rotor in the state.
[0076]
Furthermore, the present invention includes a system that converts AC power generated by an AC generator such as a wind power generator into DC power by a power converter and regenerates the DC power. The circuit configuration in that case may be replaced with, for example, the synchronous motor 2 in FIG.
[0077]
【The invention's effect】
As described above, according to the first and second aspects of the invention, for example, in the position / speed sensorless type motor driven by the inverter, when restarting from the idling state, at least one of the semiconductor switching elements of the inverter, Alternatively, the switching elements of all phases are turned on to short-circuit the windings, and the rotor position is detected from the current value flowing at that time, so even in the position / speed sensorless drive system, it is possible to restart in the idle state. . Further, since it is not necessary to detect the terminal voltage of the electric motor to estimate the position of the rotor, an expensive voltage detector is not required and the cost is not increased.
[0078]
Furthermore, in the invention described in claim 3, since the rotor position is estimated and calculated with a zero voltage pulse applied after the winding is short-circuited and the current is once zero, it is avoided that an overcurrent due to a short-circuit flows for a long period of time. Thus, the element can be protected.
[0079]
In the invention of claim 4, since the zero voltage pulse is applied twice and the rotational speed and the rotor position are estimated from the current phase difference of each time, the rotational speed estimation calculation is further facilitated.
[0080]
In the fifth aspect of the invention, the application interval of the two zero voltage pulses in the fourth aspect of the invention is made shorter than the period in which the rotor moves 180 degrees in electrical angle when the rotor rotates at the maximum rotational speed. As a result, the maximum amount of change in the rotor position is 180 ° or less, so the rotation direction can be easily specified.
[0081]
According to the sixth aspect of the present invention, the initial value of the magnetic pole position for restart can be determined by determining in which of the plurality of sector regions the instantaneous vector of the winding current exists. it can.
[0082]
In the invention according to claim 7 or claim 8, the reliability of restart is improved by reducing the error of the magnetic pole position and the error of the rotational angular frequency caused by the invention of claim 6.
[0083]
According to the ninth aspect of the present invention, the magnitude of the detected current value can be increased to the extent that it is not affected by noise even in the case of low-speed rotation without causing an overcurrent. Can be estimated.
[Brief description of the drawings]
FIG. 1 is a block diagram exemplarily showing a configuration of an invention described in claim 1;
FIG. 2 is a block diagram showing an embodiment of the present invention.
FIG. 3 is a flowchart showing a restart procedure in the embodiment of FIG. 2;
FIG. 4 is a diagram for explaining a region where an instantaneous vector of current detected when a winding is short-circuited;
FIG. 5 is a view for explaining an embodiment of the invention as set forth in claim 7;
FIG. 6 is a block diagram showing a driving device of a conventional permanent magnet synchronous motor with a magnetic pole position sensor.
FIG. 7 is an explanatory diagram of a permanent magnet rotor, an α-β coordinate system, and a dq coordinate system.
FIG. 8 is a block diagram showing a conventional magnetic pole position / speed sensorless type permanent magnet synchronous motor driving apparatus;
[Explanation of symbols]
1 Inverter
2 Permanent magnet synchronous motor
3u, 3v, 3w current detector
4u, 4v, 4w Current detector gain
5 Phase number converter
6 Coordinate converter
7 Current controller
8 Speed calculator
9 Magnetic pole position estimation unit (steady state)
10 Magnetic pole position estimation unit (at startup)
11 State controller
12 Winding short circuit
13 Gate signal switching part
20 DC power supply
30 Permanent magnet rotor
A idling restart control unit

Claims (9)

A power converter that operates an AC rotating machine, and a control device that generates and outputs an on / off signal for a semiconductor switching element that constitutes the power converter in order to control the winding current of the rotating machine to a predetermined value; In the power converter for an AC rotary machine with
When the rotor of the rotating machine idles, the control device turns on at least one of the semiconductor switching elements to short-circuit the winding of the rotating machine, and the position of the rotor is determined based on the winding current flowing at that time. A power conversion device for an AC rotary machine, wherein the power converter is estimated and restarted. In the power converter for AC rotating machines according to claim 1,
When rotating the rotor of a rotating machine, the power for an AC rotating machine is characterized by short-circuiting all phase windings of the rotating machine by a power converter and estimating the position of the rotor based on the winding current flowing at that time. Conversion device. In the AC converter for AC rotating machines according to claim 1 or 2,
A power converter for an AC rotating machine, wherein the power converter is switched so that the winding current becomes zero after a lapse of a certain period of time after the winding of the rotating machine is short-circuited to detect the winding current. In the power converter for AC rotating machines according to claim 3,
The operation of zeroing the winding current by switching the power converter is performed twice, and the rotation speed of the rotor is calculated from the phase difference at which the winding current at the short-circuit detected at each time becomes the maximum. A power converter for an AC rotating machine, which is characterized. In the AC converter for AC rotating machines according to claim 4,
The electric power for an AC rotating machine is characterized in that the interval between two operations for setting the winding current to zero is set to be shorter than a period in which the rotor is moved at an electrical angle of 180 ° when the rotor is rotating at the maximum rotational speed. Conversion device. In the power converter for AC rotating machines according to claim 3,
The phase conversion of the instantaneous vector of the detected winding current is estimated according to which one of the fan-shaped regions formed by dividing 360 ° into a plurality of segments. apparatus. In the power converter for AC rotating machines according to claim 6,
Perform the operation of zeroing the winding current by switching the power converter three times or more, and estimate the rotor position, rotation direction and rotation speed using the winding current detected at each short-circuit. A power converter for an AC rotating machine. In the power converter for AC rotating machines according to claim 7,
At least one of the three or more operation intervals in which the winding current is zero is set shorter than a period in which the rotor moves 180 degrees in electrical angle when the rotor rotates at the maximum rotation speed; and The power converter for an AC rotating machine is characterized in that at least one other interval is set to be longer than a period of 180 ° in electrical angle when the rotor rotates at the maximum rotational speed. In the power converter for AC rotating machines according to claim 3,
When the magnitude of the detected winding current is smaller than the predetermined specified value in the range where no overcurrent occurs, the winding short-circuit period is set longer and the winding current exceeding the specified value is detected again, and then An AC rotary machine power converter characterized by performing an operation of switching a power converter so that a winding current becomes zero after a certain period of time.

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