JP3544147B2 - OFDM signal receiving apparatus, OFDM signal communication system and communication control method therefor - Google Patents

Description

（１）式において、ＮｓはＯＦＤＭ信号を構成するサブキャリアの数、ＮはＦＦＴ のポイント数、ｕはシンボル番号、ｖはサブキャリア数である。また、ｘ_ｕｖは送信する符号であり、ＱＰＳＫの場合には、（２）式で表される。
【００５０】
【数２】

（２）式において、Ａ_ｕｖは振幅であり、ヌルサブキャリアの場合には、Ａ_ｕｖ＝０である。また、（１）式において、ｇ_ｕｖ（ｔ）は、第ｕシンボルおよび第ｖシンボルにおけるＯＦＤＭの孤立パルス応答であり、（３）式で表される。
【００５１】
【数３】

（３）式において、ｆｖは第ｖサブキャリア周波数、Ｔｇはガードインターバル長、Ｔｓは有効シンボル長、Ｔ＝Ｔｇ＋Ｔｓであり、ＩＩ（ｔ）は（４）式で表さ れる。
【００５２】
【数４】

（３）式中の受信信号ｒ（ｔ）は、無線伝搬路が無ひずみであるとすると、（５）式で表される。
【００５３】
【数５】

（５）式において、Δｆは周波数オフセット、Δθ_０は位相オフセット、ｎ（ ｔ）は複素の白色ガウス雑音である。
【００５４】
受信信号ｒ（ｔ）を、シンボル当たりＮサンプルで標本化した場合には、標本化出力系列は、ｒ_ｋ＝ｒ（ｔ_ｋ）となる。ただし、ｔ_ｋ＝（ｋ／Ｔｓ）＋Ｔｇ＋Δτ 、ｋ＝０，１，…，Ｎ−１であり、Δτはタイミングオフセットである。
【００５５】
パイロットシンボルにおいて伝送される信号は既知信号系列であるため、ＯＦＤＭ受信機１１で受信されるパイロットシンボルの時間波形は既知である。したがって、ＯＦＤＭ受信機１１は、予め用意しておいた時間波形と受信信号との相関演算を行うことによって、相関値列を得ることができる。例えば図３において、複素相関器３１〜３３から出力される信号列がこの相関値列に当る。
【００５６】
図４は、複素相関器３１から出力される相関値列の例を示したものである。この相関値列は、無線伝搬路のインパルス応答に相当する。つまり、遅延プロファイルを表している。伝搬路が２パスでモデル化できる場合には、２つの相関ピークを観測することができる。
【００５７】
パイロットシンボルにおいて伝送される信号を受信した信号と、受信機で予め用意しておいた信号との相関演算は、自己相関演算に相当する。図５に、この自己相関演算によって得られる系列を示す。自己相関演算によって得られる系列には、図５に示すような相関ピークが出現する。相関ピーク以外の部分は、相関サイドローブとなる。相関サイドローブの特性は、ＯＦＤＭ信号の時間波形の自己相関特性に依存する。
【００５８】
図５の相関ピークは、受信信号に含まれる自己相関特性の良好な既知信号の位置を示しており、この位置を検出することで、バーストフレームの同期を確立することができる。
【００５９】
なお、受信信号に含まれる自己相関特性の良好な既知信号を利用したピークサーチでは、周波数誤差があると、複素相関器３１〜３３の出力する信号レベルが低下し、同期特性が劣化してしまう。しかしながら、本実施形態におけるＯＦＤＭ信号通信システムでは、図３に示すように、パイロットシンボルにおいて伝送される信号に対して、それぞれ異なる周波数オフセットを与えた信号を参照系列として複数の複素相関器３１〜３３に設定しているため、複数の周波数オフセットに対応した相関をとることで、周波数誤差の影響による相関出力レベルの低下を防止し、かつ同期特性の劣化を抑えることが出来る。
【００６０】
パイロットシンボルにおいて伝送される信号に対して周波数オフセットｆ_ｋを 与えた参照系列ｄ_ｋは、（６）式で表される。
【００６１】
【数６】

（６）式において、ｕ_ｋは送信信号の離散値であり、ｕ_ｋ＝ｓ（ｋ／Ｎ）である。た だし、ＮはＦＦＴのポイント数である。
【００６２】
パイロットシンボルにおいて伝送される信号に対して周波数オフセットｆ_ｋを 与えた参照系列と受信信号の相関出力との周波数応答ａ（Δｆ）は、（７）式で表される。
【００６３】
【数７】

（７）式において、ｄｉ^＊はｄｉの複素共役である。ここで、雑音、サンプリン グのタイミングずれ、および位相オフセットの影響が無視できると仮定すると、（７）式は（８）式のように変形される。
【００６４】
【数８】

（８）式より、相関出力の周波数応答は、パイロットシンボルにおいて伝送される信号系列に依存することがわかる。
【００６５】
（８）式において、｜ｕ（ｉ）｜^２は送信信号系列の大きさを表し、ｅｘｐ（ｊ２π（Δｆ−ｆ_ｋ）・ｉ／Ｎ）は参照系列ｄ_ｋとの周波数のずれを表している。周波数のずれを表すベクトルは、系列番号に応じて回転し、回転量は周波数のずれに依存する。すなわち、ｅｘｐ（ｊ２π（Δｆ−ｆ_ｋ）・ｉ／Ｎ）が表現する系列は、向きが系列番号に応じて回転し、長さが送信信号系列の大きさであるベクトルである。
【００６６】
図６は、（８）式に示す周波数応答、すなわち、パイロットシンボルにおいて伝送される信号に対して周波数オフセットΔｆを与えた参照系列と受信信号の相関出力の大きさ｜ａ（Δｆ）｜^２の周波数応答の例を示す図である。ただし、図６に おいて、ｆ_ｋは０である。図６より、周波数オフセットが０の場合に、相関値は 最大になることがわかる。
【００６７】
すなわち、図３に示す複数の複素相関器３１〜３３に、それぞれ異なる周波数オフセットを有する参照系列を設定したときに、各複素相関器３１〜３３から出力される、参照系列と受信信号との相関出力絶対値の最大値を検索し、最大値を出力した複素相関器に対応する周波数オフセットにより、ＯＦＤＭ送信信号のキャリア周波数とＯＦＤＭ受信信号のキャリア周波数とのずれを検出することができる。
【００６８】
図３における最大値検出部３４は、前述したサーチ手順を実行し、キャリア周波数の誤差信号と相関出力の最大値信号とを出力する。
【００６９】
なお、図６では、パイロットシンボルにおいて伝送される既知信号系列が異なる２つの参照系列Ａ，Ｂと受信信号との相関出力の大きさ｜ａ（Δｆ）｜^２の周波数 応答の例を示している。図６より、参照系列と受信信号の相関出力の大きさ｜ａ（Δｆ）｜^２の周波数応答は、既知信号系列に依存することがわかる。
【００７０】
図３において、最大値検出部３４に接続されたシフトレジスタ３５は、最大値検出部３４が出力した相関出力の最大値信号を順にシフトし、最大値検出部３６は、シフトレジスタ３５の各段の出力の中から最大値を検出する。この最大値は、前述した自己相関演算によって得られる系列の自己相関ピークを示しており、このピーク位置は、無線伝搬路を経て受信される信号のバーストフレームの基準タイミングを示している。
【００７１】
最大値検出部３６は、検出されたバーストフレームの基準タイミングと、ＯＦＤＭ受信機１１のタイミングとの誤差であるタイミング誤差信号を出力する。また、最大値検出部３６は、最大値検出部３４で検出された相関出力の最大値を出力した複素相関器が出力した相関出力の最大値を検出し、相関出力の最大値が出力されるタイミングに対応する周波数オフセットを、周波数誤差信号として出力する。
【００７２】
なお、ここで得られる周波数オフセット量は、複素相関器の個数に依存した離散値である。したがって、補間演算を行うことにより、周波数誤差信号の精度を向上させることができる。
【００７３】
以上のような構成により、サンプリングタイミング誤差とキャリア周波数の誤差の検出が可能となり、信頼性の高い信号伝送が可能なＯＦＤＭ信号通信システムを実現できる。
【００７４】
前述したように、マルチメディア情報等を扱うＯＦＤＭ信号通信システムでは、伝送される信号がバースト的に発生するため、バーストフレーム毎に短時間で同期を確立する必要がある。高速な初期同期のためには、ある程度の回路規模の増大は避けられないが、回路規模の増大は、製造コストの上昇につながるため、回路規模は、できるだけ小さくするのが望ましい。つまり、全体の回路負担に占める割合が比較的大きい複素のスライディング相関器を、如何にして簡易な構成とするかが重要な技術課題となる。
【００７５】
そこで、以下に、図３に示す複素相関器の回路規模を低減する手法について説明する。以下に説明する手法は、パイロットシンボルにおいて伝送される信号に対して異なる周波数オフセットを与えた参照系列との相関演算に着目したものである。
【００７６】
前述したように、パイロットシンボルにおいて伝送される信号に対して周波数オフセットΔｆを与えた参照系列と受信信号との相関出力ａ（Δｆ）は、前述した（８）式により表現できる。なお、説明を簡略化するため、（８）式では、雑音、サンプリングタイミングのずれ、および位相オフセットの影響が無視できると仮定している。
【００７７】
周波数のずれを表すベクトルは、図７に示すように、パイロット信号の信号系列の系列番号に応じて回転し、回転量は周波数のずれに依存する。Δｆ＝１のとき、このベクトルは一回転する。すなわち、ｅｘｐ（ｊ２π（Δｆ−ｆｋ）ｉ／Ｎ）が表 現する系列は、向きが系列番号に応じて回転し、長さが送信信号系列の大きさであるベクトルである。
【００７８】
ここで、｜ｕ（ｉ）｜^２・ｅｘｐ（−ｊ２π（Δｆ−ｆｋ）ｉ／Ｎ）の特徴について考察す る。ベクトル軌跡の変化を理解しやすいように、送信信号系列の大きさ｜ｕ（ｉ）｜^２の変化が小さい例を用いて説明する。
【００７９】
図８および図９にベクトル軌跡の一例を示す。ＦＦＴのポイント数Ｎ＝６４、ＯＦＤＭ信号を構成するサブキャリアの数Ｎｓ＝４であり、図８はΔｆ＝０．２５、図９ はΔｆ＝０．５である。
【００８０】
ベクトル軌跡が図８および図９のようになることは、送信信号系列の大きさ｜ ｕ（ｉ）｜^２を観測すると理解できる。
【００８１】
図１０は図８および図９に対応する送信信号系列の大きさ｜ｕ（ｉ）｜^２を示す図 である。図１０の横軸はサンプル番号ｉであり、縦軸は｜ｕ（ｉ）｜^２である。
【００８２】
図８〜図１０より、｜ｕ（ｉ）｜^２・ｅｘｐ（−ｊ２π（Δｆ−ｆｋ）ｉ／Ｎ）が表現する系列は、向きが系列番号に応じて回転し、長さが送信信号系列の大きさであるベクトルであることが確認できる。また、（８）式で示される相関出力ａ（Δｆ）は、これらの軌跡のベクトル和から算出される。
【００８３】
ここで、図８および図９に示したベクトル軌跡の分布に着目する。図８および図９より、原点付近に多数の分布が存在することがわかる。このことは、送信信号系列の大きさ｜ｕ（ｉ）｜^２の分布は、０付近が高く、｜ｕ（ｉ）｜^２が大きい値をとる分布はわずかであることを示している。（８）式で示される相関出力ａ（Δｆ）は、これらの軌跡のベクトル和であるため、送信信号系列の大きさ｜ｕ（ｉ）｜^２が大 きな値をとるときの影響が支配的となる。
【００８４】
このため、本実施形態では、参照系列をいくつかの代表的なベクトルのみで表現する。
【００８５】
図８および図９では、ベクトル軌跡の変化を理解しやすいように、送信信号系列の大きさ｜ｕ（ｉ）｜^２の変化が小さい場合の例を示した。しかしながら、ＯＦＤＭの 時間波形は、比較的変動が大きいという特徴を有している。逆に言うと、変化の小さいケースはまれである。
【００８６】
図１１はＱＰＳＫマッピングされたデータを伝送した場合のベクトル軌跡の例を示す図である。図１１では、ＦＦＴのポイント数Ｎ＝３２、ＯＦＤＭ信号を構成するサブキャリアの数Ｎｓ＝３２である。図１１（ａ）はパイロットシンボルにおいて 伝送される信号に対して与えられる周波数オフセットΔｆ＝０．４の場合、図１１（ｂ）はΔｆ＝０．９の場合である。
【００８７】
図１１より、送信信号系列の大きさ｜ｕ（ｉ）｜^２の分布は、０付近が高く、｜ｕ（ ｉ）｜^２が大きい値をとる分布はわずかであることがわかる。
【００８８】
本実施形態では、参照系列をいくつかの代表的なベクトルのみで表現する。以下、代表ベクトルの例を図１２および図１３を用いて説明する。
【００８９】
図１２はランダムデータを伝送した場合のベクトル軌跡を示す図である。図１２は、ＦＦＴのポイント数Ｎ＝６４、ＯＦＤＭ信号を構成するサブキャリアの数Ｎｓ ＝６４、周波数オフセットΔｆ＝１．０の例を示している。図１２からも、送信信 号系列の大きさ｜ｕ（ｉ）｜^２が大きな値となるベクトルはわずかであることがわか る。
【００９０】
このため、本実施形態では、図１２のベクトルを図１３に示すような代表的なベクトルで表現する。すなわち、参照系列を図１３に示すような代表ベクトル以外を０としたベクトルで表現する。
【００９１】
（複素相関器の第１の構成例）
次に、図３に示す複素相関器３１〜３３の具体例について説明する。図１４は複素相関器の第１の構成例を示すブロック図である。図１４の複素相関器５０は、段数がＬのシフトレジスタ５１と、参照系列５２と、Ｍ個の複素乗算器５３と、各複素乗算器５３の出力を加算する加算器５４とを有する。
【００９２】
参照系列５２は、パイロットシンボルにおいて伝送される信号の時間領域における大きさが大きな値をとる系列番号に対応した代表ベクトルである。乗算器５３は、前述した代表ベクトルの番号に対応する位置に配置する。図１４は、Ｌ＝６、代表ベクトルの番号が１、４、６である場合の例を示している。
【００９３】
（複素相関器の第２の構成例）
図１５は複素相関器の第２の構成例を示すブロック図である。図１５の複素相関器６０は、段数がともにＬであるシフトレジスタ（第２および第３の信号シフト手段）６１，６２と、Ｍ個の複素乗算器６３と、各複素乗算器６３の出力を加算する加算器６４とを有する。乗算器６３は、前述した代表ベクトルの番号に対応する位置に配置される。図１５は、Ｌ＝６、代表ベクトルの番号が１、４、６の場合の例を示している。
【００９４】
図１５の複素相関器６０においても、乗算器６３の数を減らすことができるため、回路規模の増大を抑制でき、製造コストの削減が図れる。
【００９５】
図１６は図１５の複素相関器６０を有するＯＦＤＭ受信機１１の構成の一部を示すブロック図である。図１６のＯＦＤＭ受信機１１は、ＯＦＤＭ受信信号を標本化する標本化器１９と、１シンボル遅延器（遅延手段）６５と、図１５の複素相関器６０とを有する。
【００９６】
図１６のＯＦＤＭ受信機１１では、複素相関器６０により、２つの連続するパイロットシンボルが配置されたバーストフレームデータの相関ピークを得ることができ、複素相関器６０から出力される信号の最大値をサーチすることで、バーストフレームを検出することができる。
【００９７】
（複素相関器の第３の構成例）
上述した図１４および図１５では、パイロットシンボルにおいて伝送される信号の時間領域における大きさが大きな値をとる代表ベクトルに対応して複素乗算器を配置する例を説明したが、パイロットシンボルにおいて伝送される信号として、巡回系列の信号を伝送すれば、複素乗算器を等間隔に配置することができる。
【００９８】
以下、複素乗算器を等間隔に配置できるような既知信号系列について説明する。この既知信号系列は、パイロット信号が伝送されるタイミングで伝送されるものである。
【００９９】
参照系列ｄ_ｋをいくつかの代表ベクトルで表現するために、既知信号系列とし て、周波数領域における位相配置がサブキャリアに対して巡回するように設定した系列を採用する。ここでは、このような系列を巡回系列と呼ぶ。
【０１００】
巡回系列では、繰り返し周期をＩｇとすると、既知信号系列｛Ｘｋ｝をＩＦＦＴした時間応答の系列｛ｘｋ｝は、Ｎ／Ｉｇごとに値をもつようになる。
【０１０１】
すなわち、パイロットシンボルに巡回系列を用いることにより、複素乗算器の数を減らすことができ、回路規模の増加を極力抑えることができる。
【０１０２】
巡回系列は（９）式で表される。
Ｘ_ｋ＋１・Ｘ_ｋ ^＊＝ｅｘｐ（ｊφ_ｋ ｍｏｄｕｌｏ Ｉ_ｇ） …（９）
ただし、ｋ（ｋ＝０，１，…，Ｎ−１）はサブキャリア番号、φ_ｉはサブキャリア間の位相変化である。
【０１０３】
（９）式を満足する系列は複数存在する。しかしながら、巡回系列では、特定サンプルに対して電力が集中するため、ＰＡＰＲ（Ｐｅａｋ ｔｏ Ａｖｅｒａｇｅ Ｐｏｗｅｒ Ｒａｔｉｏ）の 点からは不利である。ＰＡＰＲが大きいということは、ＯＦＤＭ変調方式の欠点の一つである。ＰＡＰＲは、電力効率の観点からはできる限り小さい方が望ましい。
【０１０４】
このため、各代表ベクトルに電力が等分配される巡回系列、すなわち、｜ｘｉ｜^２＝Ｎ／Ｉｇとなるような巡回系列を選ぶ必要がある。ここで、ｉは代表ベクトル の番号を、ＮはＦＦＴのポイント数を示す。巡回周期Ｉｇは、代表ベクトルの数 に一致する。
｜ｘｉ｜^２＝Ｎ／Ｉｇとなる系列の一例として、（１０）式がある。
（φ０，φ１，φ２，…，φ１５）
＝（０，π／２，−π／２，π，π／２，−π／２，π，０，π，π／２，π／２，π，−π／２，−π／２，０，０）
…（１０）
ここで、Ｉｇ＝１６、ＸｋはＱＰＳＫマッピングとしている。
【０１０５】
この系列では、代表ベクトルの間隔は、等間隔（ｉ＝０，Ｎ／Ｉｇ，２・Ｎ／Ｉｇ，…，（Ｉｇ−１）・Ｎ／Ｉｇ）となる。
【０１０６】
図１７は巡回系列のパイロット信号を用いた場合の複素相関器の第３の構成例を示すブロック図である。既知信号系列に巡回系列を用いると、その時間応答は、等間隔ごとに値を有するため、乗算器１０２は等間隔に配置される。図１７は、乗算器の数が４の場合の例を示している。なお、乗算器１０２の数は巡回周期Ｉｇに一致する。すなわち、パイロットシンボルに巡回系列を用いることにより 、スライディング相関器における複素乗算器の数を減らすことができ、回路規模の増加を抑制できる。
【０１０７】
次に、巡回系列のパイロット信号を送信する場合の複素相関器の構成について説明する。
【０１０８】
（複素相関器の第４の構成例）
図１８は複素相関器の第４の構成例を示すブロック図である。図１８の複素相関器７０は、段数がともにＬのシフトレジスタ７１，７２と、Ｍ個の複素乗算器７３と、各複素乗算器７３の出力を加算する加算器７４とを有する。乗算器７３は、代表ベクトルの番号に対応する位置に配置される。
【０１０９】
パイロット信号は巡回系列であるため、図１８に示すように乗算器７３を等間隔に配置することができる。
【０１１０】
図１８の複素相関器は乗算器７３，８５を等間隔に配置する以外は、図１５と同様に構成される。したがって、図１５と同様の作用・効果が得られる。
【０１１１】
（複素相関器の第５の構成例）
図１９は複素相関器の第５の構成例を示すブロック図である。図１９の複素相関器９０は、巡回系列のパイロット信号を伝送する場合の構成であり、入力系列を１：Ｋの率でパラレルに変換するシリアル／パラレル変換器９１と、段数がＭのシフトレジスタ９２と、Ｍ個の複素乗算器９３と、各複素乗算器９３の出力を加算する加算器９４と、参照系列９５とを有する。参照系列９５は、パイロットシンボルにおいて伝送される信号の時間領域における大きさが大きな値をとる系列番号に対応した代表ベクトルである。
【０１１２】
図１９の複素相関器２０は、受信されたパイロット信号と参照系列との間で複素演算を行う点で、図１８の複素相関器２０と異なる。図１９の複素相関器９０の場合も、乗算器の数が少なくて済むため、回路規模の増大を抑制できるとともに、製造コストを削減できる。
【０１１３】
（周波数誤差＆タイミング誤差推定器の他の構成例）
図２０は周波数誤差＆タイミング誤差推定器の他の構成例を示すブロック図である。図２０の周波数誤差＆タイミング誤差推定器２０ａには、図１に示した標本化器１９から出力された信号が入力される。周波数誤差＆タイミング誤差推定器２０ａからは、局部発振器１８に供給される周波数誤差信号と、ＦＦＴ２１に供給されるタイミング誤差信号とが出力される。
【０１１４】
図２０の周波数誤差＆タイミング誤差推定器２０ａは、シフトレジスタ３０と、複数の複素相関器３１〜３３と、最大値検出部３４と、比較器１１１と、平均受信電力測定部１１２とを有する。各複素相関器３１〜３３には、それぞれ異なる参照系列３７〜３９が設定される。各複素相関器３１〜３３は、シフトレジスタ３０から入力される信号と参照系列とを掛け合わせる複素乗算器４１と、複素乗算器４１の出力を加算する加算器４２とを有する。複素相関器３１〜３３に設定される参照系列３７〜３９は、パイロットシンボルにおいて伝送される信号に対して異なる周波数オフセットを与えた信号である。
【０１１５】
最大値検出部３４は、各複素相関器３１〜３３から出力される相関値の最大値を検出し、相関値の最大値と、そのときの周波数オフセット量とを出力する。キャリア周波数の誤差は、最大値を出力した複素相関器に対応する周波数から検出できる。ここで得られる周波数オフセット量は、複素相関器の個数に依存した離散値である。このため、最大値を出力した複素相関器と、その複素相関器に隣接する複素相関器からの出力を用いた補間により、周波数誤差を得る。
【０１１６】
複数の複素相関器の各出力信号の最大値を検出することにより、受信信号に含まれる自己相関特性の良好な既知信号の位置を検出でき、これにより、バーストフレームの同期を確立することができる。
【０１１７】
最大値検出部３４から出力された相関値および周波数オフセット量は、比較器１１１に入力される。比較器１１１には、平均受信電力測定部１１２の出力信号が入力される。平均受信電力測定部１１２には、図１に示した標本化器１９が出力する信号が入力され、平均受信電力値が出力される。平均受信電力測定部１１２は、受信信号の平均電力を測定する。比較器１１１は、受信信号の平均電力の情報に基づいて比較値を設定し、この比較値と、最大値検出部３４から出力された相関値の最大値との比較演算を行う。
【０１１８】
この比較演算により、比較器１１１は、受信信号に含まれる自己相関特性の良好な既知信号の位置と内部クロックとの誤差を求め、その誤差値をタイミングオフセットとして出力する。複素相関器３１〜３３から出力される相関出力の絶対値が大きいほど、平均電力は大きくなるため、図示の比較器１１１は、平均電力測定部１１２で測定された電力が予め設定された比較値以上になったときを基準として周波数誤差信号と相関出力最大値信号を出力する。
【０１１９】
図２０のような構成により、サンプリングタイミング誤差とキャリア周波数誤差の検出が可能になり、信頼性の高い信号伝送が可能なＯＦＤＭ信号受信装置を実現することができる。
【０１２０】
上述した実施形態では、最大値検出部３６から周波数誤差信号とタイミング誤差信号を出力する例を説明したが、周波数誤差信号のみを出力してキャリア周波数の誤差調整のみを行ってもよい。
【０１２１】
【発明の効果】
以上詳細に説明したように、本発明によれば、受信したパイロット信号と、パイロット信号に対してそれぞれ異なる周波数オフセットが設定された参照信号との間で複素相関演算を行った結果に基づいて、再生キャリア信号の周波数を制御するため、受信フレームの同期確立を迅速に行うことができ、安定した信頼性の高い信号伝送およびその受信が可能となる。また、複素相関演算手段の内部の構成を簡略化できるため、回路規模の増大を抑制でき、それに伴って製造コストを削減できる。
【図面の簡単な説明】
【図１】本発明に係るＯＦＤＭ信号受信装置を備えたＯＦＤＭ信号通信システムの一実施形態のブロック図。
【図２】ＯＦＤＭ送信機から送信されるＯＦＤＭ信号のバーストフレームのデータ構成を示す図。
【図３】
図１に示した周波数誤差＆タイミング誤差推定器の内部構成を示すブロック図。
【図４】
複素相関器から出力される相関値列の例を示す図。
【図５】
自己相関演算によって得られる系列を示す図。
【図６】
パイロットシンボルにおいて伝送される信号に対して周波数オフセットΔｆを与えた参照系列と受信信号の相関出力の大きさ｜ａ（Δｆ）｜^２の周波数応答の例を 示す図。
【図７】
周波数のずれを表すベクトルを示す図。
【図８】
Δｆ＝０．２５の場合のベクトル軌跡の一例を示す図。
【図９】
Δｆ＝０．５の場合のベクトル軌跡の一例を示す図。
【図１０】
図８および図９に対応する送信信号系列の大きさ｜ｕ（ｉ）｜^２を示す図。
【図１１】ＱＰＳＫマッピングされたデータを伝送した場合のベクトル軌跡の例を示す図。
【図１２】
ランダムデータを伝送した場合のベクトル軌跡を示す図。
【図１３】
代表ベクトル以外を０としたベクトル軌跡を示す図。
【図１４】複素相関器の第１の構成例を示すブロック図。
【図１５】複素相関器の第２の構成例を示すブロック図。
【図１６】図１５の複素相関器を有するＯＦＤＭ受信機の構成の一部を示すブロック図。
【図１７】パイロット信号に巡回系列を用いた場合の複素相関器の第３の構成例を示すブロック図。
【図１８】複素相関器の第４の構成例を示すブロック図。
【図１９】複素相関器の第５の構成例を示すブロック図。
【図２０】周波数誤差＆タイミング誤差推定器の他の構成例を示すブロック図。
【図２１】送信側に用いられるＯＦＤＭ変調装置の構成を示すブロック図。
【図２２】ガードインターバルを説明する図。
【図２３】受信側に用いられるＯＦＤＭ復調装置の構成を示すブロック図。
【図２４】文献に記載された周波数オフセット検出・除去方式を示す図。
【符号の説明】
１０ ＯＦＤＭ送信機
１１ ＯＦＤＭ受信機
１２ 逆高速フーリエ変換器（ＩＦＦＴ）
１３，１７ 周波数変換器
１４，１８ 局部発振器
１５，１６ アンテナ
１９ 標本化器
２０ タイミング誤差推定器
２１ 高速フーリエ変換器（ＦＦＴ）
３０，３５ シフトレジスタ
３１〜３３ 複素相関器
３４，３６ 最大値検出部
３７〜３９ 参照系列
４０ 複素乗算器
４１ 加算器[0001]
TECHNICAL FIELD OF THE INVENTION
The present invention relates to an OFDM signal receiving apparatus that transmits and receives data using an orthogonal frequency division multiplexing (OFDM) signal including a plurality of orthogonally arranged carrier signals.
[0002]
[Prior art]
In recent years, an OFDM signal communication system for transmitting digital information such as voice, video or data has been developed, and digital communication is becoming mainstream even in mobiles such as mobile phones.
[0003]
In mobile communication, it is necessary to consider the effects of a plurality of reflected waves (multipath) due to buildings such as buildings and other reflecting objects. That is, radio waves from a plurality of transmitting stations reach the receiving point. Such a multipath phenomenon causes distortion in a signal and is a major cause of deterioration in reception quality.
[0004]
In addition, an OFDM signal communication system that handles multimedia information and the like is required to support various required qualities. For example, multimedia digital communication using a small portable information terminal requires highly reliable signal transmission while having the convenience of mobile communication for connecting to a network or the like from an arbitrary point.
[0005]
In digital communication as well as in mobile communication, it is necessary to establish frequency synchronization and timing synchronization in order to restore information transmitted from a transmitter. In particular, in mobile communication, a reception process fluctuates, so that synchronization processing is indispensable, but it takes a certain amount of time to achieve synchronization. In a state where synchronization is lost, information cannot be restored. Therefore, high-speed frequency synchronization and timing synchronization are required for recovery in the case where synchronization is lost.
[0006]
An OFDM signal communication system that handles multimedia information and the like is suitable for packet communication because a signal to be transmitted occurs in a burst. In packet communication, packets are transmitted at random instead of at a fixed cycle as in time division multiple access (TDMA). Therefore, synchronization must be established for each packet, and synchronization must be established in a short time. Furthermore, in a portable information terminal that handles multimedia information, it is difficult to use a high-precision oscillator from the viewpoint of miniaturization, so it is necessary to apply a high-performance carrier frequency synchronization method.
[0007]
By the way, as an effective method of reducing the influence of a delayed wave on a multipath propagation path, there is OFDM (Orthogonal Frequency Division Multiplexing). OFDM is a scheme in which transmission information is divided to generate a plurality of low-speed digital signals, and the plurality of signals are used to independently modulate orthogonal subcarriers. By parallel transmission using multicarriers, the signal transmission speed can be reduced, and by providing a guard section unique to OFDM, the effect of a delayed wave can be reduced as compared with the single carrier modulation method.
[0008]
The outline of the OFDM method will be described below.
[0009]
FIG. 21 is a block diagram showing a configuration of an OFDM modulator used on the transmission side. Transmission data is input to the OFDM modulator. This transmission data is supplied to the serial / parallel conversion unit 201 and is converted into data composed of a plurality of low-speed transmission symbols. That is, the transmission information is divided to generate a plurality of low-speed digital signals. The parallel data is supplied to an inverse fast Fourier transform (IFFT) unit 202.
[0010]
Parallel data is allocated to each subcarrier constituting OFDM, and is mapped in the frequency domain. Here, modulation such as BPSK, QPSK, 16QAM, and 64QAM is performed on each subcarrier. The mapping data is converted from transmission data in the frequency domain to transmission data in the time domain by performing an IFFT operation. As a result, a multicarrier modulated signal is generated in which a plurality of subcarriers that are orthogonal to each other are independently modulated. The output of IFFT section 202 is supplied to guard interval adding section 203.
[0011]
As shown in FIG. 22, guard interval adding section 203 adds a copy to the front of the effective symbol period for each transmission symbol, using the rear of the effective symbol of the transmission data as a guard interval. The baseband signal obtained by the guard interval addition unit is supplied to the quadrature modulation unit 204.
[0012]
The quadrature modulation section 204 performs quadrature modulation on the baseband OFDM signal supplied from the guard interval addition section 203 using the carrier signal supplied from the local oscillator 205 of the OFDM modulation apparatus, and performs an intermediate frequency (IF). Frequency conversion to a signal or radio frequency (RF) signal. That is, the quadrature modulation unit frequency-converts the baseband signal into a desired transmission frequency band and then outputs the signal to the transmission path.
[0013]
FIG. 23 is a block diagram showing a configuration of an OFDM demodulator used on the receiving side. An OFDM signal generated by the OFDM modulator of FIG. 21 is input to the OFDM demodulator via a predetermined transmission path.
[0014]
The OFDM reception signal input to the OFDM demodulation device is supplied to the quadrature demodulation unit 211. The quadrature demodulation unit 211 performs quadrature demodulation on the OFDM reception signal using the carrier signal supplied from the local oscillator 212 of the OFDM demodulation apparatus, frequency-converts the RF signal or the IF signal into a baseband signal, Obtain an OFDM signal. This OFDM signal is supplied to guard interval removing section 213.
[0015]
The guard interval removing section 213 removes the signal added by the guard interval adding section 203 of the OFDM modulator according to a timing signal supplied from a symbol timing synchronization section (not shown). The signal obtained by the guard interval removing unit 203 is supplied to a fast Fourier transform (FFT) unit 214.
[0016]
The FFT unit 214 converts the input reception data in the time domain into reception data in the frequency domain by performing FFT. Furthermore, demapping is performed in the frequency domain, and parallel data is generated for each subcarrier. Here, the demodulation for the modulation such as BPSK, QPSK, 16QAM, or 64QAM applied to each subcarrier is performed. The parallel data obtained by the FFT unit 214 is supplied to a parallel / serial conversion unit 215 and output as reception data.
[0017]
As described above, the OFDM demodulator needs to establish carrier frequency synchronization and timing frequency synchronization in order to restore information transmitted from the OFDM modulator.
[0018]
In OFDM, since the subcarrier spacing is narrow and each subcarrier is orthogonally arranged, the carrier frequency supplied from the local oscillator 212 of the OFDM demodulator and the carrier frequency of the OFDM modulator differ from each other, ie, the frequency offset Is present, the orthogonality between the subcarriers is broken and the reception characteristics are significantly degraded. Therefore, in OFDM, establishment of carrier frequency synchronization is extremely important.
[0019]
The frequency synchronization method of OFDM is described in "Study on Synchronous System of OFDM Modulation System for High-Speed Wireless LAN" in IEICE Technical Report RCS97-210 (1998-01).
[0020]
OFDM synchronization methods are classified into those based on processing in the frequency domain and those based on processing in the time domain. The synchronization method described in the above report is based on processing in the time domain. In this method, two OFDM symbols are used, the same signal is allocated to the two pilot symbols, and a carrier frequency shift and a timing shift are estimated by a correlation operation between the two.
[0021]
In addition, there is also known a method of separately transmitting a start symbol for determining an initial phase before transmitting a data symbol to estimate a frequency error. However, in this type of frequency error estimation method, it is necessary to transmit two or more pilot symbols, so that there is a problem that transmission efficiency is reduced.
[0022]
In order to improve the transmission efficiency, it is possible to use both the start symbol and the pilot symbol. However, in this case, two or more pilot symbols are required for synchronization. There is a problem that the efficiency is reduced.
[0023]
On the other hand, IEICE Transactions B-II Vol. J75-B-II, No. 12, p884-895 (December 1992) discloses a carrier frequency offset estimation method using a plurality of complex correlators. This document proposes a frequency offset detection / removal method shown in FIG. This document is directed to a single-carrier modulation scheme using a training signal added to a TDMA slot, and a plurality of correlators are provided with the same reference sequence and receive signals input to the correlator. Frequency is changed.
[0024]
To speed up the initial synchronization, it is inevitable to increase the circuit scale to some extent. However, since an increase in the circuit scale leads to an increase in manufacturing cost, it is desirable to keep the circuit scale as low as possible. However, in the case of the above document, a plurality of complex correlators are required, so that there is a problem that the circuit scale becomes large.
[0025]
The present invention has been made in view of such a point, and an object of the present invention is to provide an OFDM signal receiving apparatus and an OFDM signal communication system capable of performing highly reliable data transmission without increasing the circuit scale. To provide.
[0026]
In order to solve the above-mentioned problem, the present invention provides a receiving means for receiving an OFDM (orthogonal frequency division multiplexing) signal in which a pilot signal composed of a predetermined known signal sequence is inserted by an antenna, and for demodulating the OFDM signal. A local oscillator for generating a reproduced carrier signal as a reference signal, a frequency converter for converting the OFDM signal to a baseband signal based on the reproduced carrier signal, and performing an FFT operation based on the baseband signal to obtain a frequency. An OFDM signal receiving apparatus comprising: an FFT operation unit that generates reception data in a region; a sampling unit that samples the baseband signal; and different frequency offsets are set for the pilot signal. A plurality of complex correlation operation means for performing a complex correlation operation between a reference signal and the pilot signal to obtain a correlation output; The frequency error between the reproduced carrier signal and the carrier signal of the OFDM signal, based on the correlation output obtained by each of the plurality of complex correlation operation means, and the timing error when the FFT operation means performs the operation. Error estimation means for estimating, the local oscillator controls the frequency of the reproduced carrier signal based on the frequency error, the FFT calculation means determines an FFT window based on the timing error, Each of the plurality of complex correlation operation means includes a first signal shift means having L (L is an integer of 2 or more) shift stages to which the baseband signal sampled by the sampling means is input; M complex multiplying means for multiplying the output of M stages (M is an integer equal to or greater than 2 and M <L) selected from the respective stages of the 1 signal shifting unit and the reference signal, respectively. And an adder for calculating the correlation output by adding the multiplication results of the M complex multiplyers.
[0036]
BEST MODE FOR CARRYING OUT THE INVENTION
Hereinafter, an OFDM signal receiving apparatus and an OFDM signal communication system according to the present invention will be specifically described with reference to the drawings.
[0037]
FIG. 1 is a block diagram of an embodiment of an OFDM signal communication system including an OFDM signal receiving apparatus according to the present invention. The illustrated OFDM signal communication system transmits and receives the above-described OFDM signal, and includes an OFDM transmitter 10 and an OFDM receiver 11. FIG. 1 shows only some of the components of the OFDM signaling system.
[0038]
An OFDM transmitter (transmission means) 10 includes an inverse fast Fourier transformer (IFFT) 12 for converting a transmission signal mapped in the frequency domain into a signal in the time domain, and a frequency conversion for frequency-converting the signal in the time domain into an RF signal. A local oscillator 14 for supplying a sinusoidal carrier signal (local oscillation signal) to the frequency converter 13, and an antenna 15 for radiating an RF signal as a radio wave to a propagation path.
[0039]
The OFDM receiver 11 receives an OFDM signal transmitted from the transmitter 10 via a radio channel, and receives an OFDM signal. The frequency converter converts the RF signal received by the antenna 16 into a baseband signal. , A local oscillator 18 for supplying a sinusoidal local oscillation signal to the frequency converter 17, a sampler (sampling means) 19 for sampling the received baseband signal, and an output signal of the sampler 19 , A frequency error & timing error estimator (error estimating means) 20 for detecting a frequency offset and a timing offset, and a fast Fourier transformer (FFT) for converting a time domain signal output from the sampler 19 into a frequency domain signal. , FFT operation means) 21.
[0040]
In FIG. 1, the guard interval adding unit and the guard interval removing unit described in FIG. 22 are omitted for simplification.
[0041]
FIG. 2 is a diagram showing a data structure of a burst frame of the OFDM signal transmitted from the OFDM transmitter 10. The burst frame includes a pilot symbol (pilot signal) for transmitting a known signal sequence. In an OFDM signal communication system that handles multimedia information, a signal to be transmitted occurs in a burst. Therefore, it is necessary to establish synchronization for each burst frame. FIG. 2 shows one pilot symbol, but one or more pilot symbols (pilot signals) may be added.
[0042]
In order to perform synchronous detection on the OFDM signal, it is necessary to transmit a known signal so that the receiver can grasp the absolute phase. In order to perform differential detection, a start symbol must be transmitted. In the present embodiment, one or more pilot symbols are added as a burst frame configuration of the OFDM signal. For this reason, it becomes possible to serve both as a pilot symbol for frequency synchronization and as a known signal for synchronous detection and a start symbol for differential detection, thereby preventing deterioration of transmission efficiency.
[0043]
The frequency error & timing error estimator 20 shown in FIG. 1 uses a signal output from the sampler 19 to generate a frequency error and a timing by a signal obtained by a sliding correlation operation between a signal transmitted in a pilot symbol and a received signal. Detect errors.
[0044]
The carrier frequency error information output from the frequency error & timing error estimator 20 is supplied to the local oscillator 18. The local oscillator 18 changes the oscillation frequency of the reproduced carrier signal (local oscillation signal) based on the input carrier frequency error information. The timing error information output from the frequency error & timing error estimator 20 is supplied to the FFT 21. The FFT 21 determines an FFT window based on the timing error information.
[0045]
As described above, in the present embodiment, the estimation of the carrier frequency shift and the sampling timing shift are performed by using the time domain signal in the pilot symbol in the OFDM reception signal, and the carrier frequency synchronization acquisition and Synchronous acquisition of sampling timing is performed. The operation of this embodiment will be described later in detail.
[0046]
FIG. 3 is a block diagram showing an internal configuration of the frequency error & timing error estimator 20 shown in FIG. As shown in FIG. 3, the frequency error & timing error estimator 20 shifts a signal output from the sampler 19 by a sampling clock input to the sampler 19 (a first signal shifter). ) 30, a plurality of complex correlators (complex correlation operation means) 31 to 33, and a maximum value of signals output by the complex correlators 31 to 33 (a maximum value signal of a frequency error and a maximum value signal of a correlation output) are detected. A maximum value detecting section (first maximum value searching means) 34, a shift register 35 for shifting a signal output from the maximum value detecting section 34 by a sampling clock input to the sampler 19, and a shift register 35. (A maximum value detecting section) for detecting the maximum value of the output (the signal obtained by shifting the output of the maximum value detecting section 34).
[0047]
The frequency error & timing error estimator 20 has a plurality of complex correlators 31 to 33, and different reference sequences 37 to 39 are set in each of the complex correlators 31 to 33. Each of the complex correlators 31 to 33 includes a complex multiplier (complex multiplier) 40 for multiplying the output signal of each stage of the shift register 30 by the reference sequence, and an adder 41 for adding the output of the complex multiplier 40. Have. Reference sequences 37 to 39 set in complex correlators 31 to 33 are signals obtained by giving different frequency offsets to signals transmitted in pilot symbols. Details of the reference sequence will be described later.
[0048]
Next, a method for detecting a shift in sampling timing and a shift in carrier frequency will be described. To simplify the description, the OFDM signal in the equivalent reduction with the carrier signal component removed is handled. The OFDM transmission signal s (t) in the equivalent reduction can be expressed by equation (1).
[0049]
(Equation 1)

In equation (1), Ns is the number of subcarriers constituting the OFDM signal, N is the number of points of FFT, u is the symbol number, and v is the number of subcarriers. Also, x_uvIs a code to be transmitted, and in the case of QPSK, is represented by equation (2).
[0050]
(Equation 2)

In equation (2), A_uvIs the amplitude, and for the null subcarrier, A_uv= 0. In the equation (1), g_uv(T) is an isolated pulse response of OFDM at the u-th symbol and the v-th symbol, and is represented by Expression (3).
[0051]
(Equation 3)

In equation (3), fv is the v-th subcarrier frequency, Tg is the guard interval length, Ts is the effective symbol length, T = Tg + Ts, and II (t) is represented by equation (4).
[0052]
(Equation 4)

The received signal r (t) in the expression (3) is represented by the expression (5), assuming that the wireless propagation path has no distortion.
[0053]
(Equation 5)

In the equation (5), Δf is a frequency offset, Δθ₀Is the phase offset and n (t) is the complex white Gaussian noise.
[0054]
When the received signal r (t) is sampled at N samples per symbol, the sampled output sequence is r_k= R (t_k). Where t_k= (K / Ts) + Tg + Δτ, k = 0, 1,..., N−1, and Δτ is a timing offset.
[0055]
Since the signal transmitted in the pilot symbol is a known signal sequence, the time waveform of the pilot symbol received by the OFDM receiver 11 is known. Therefore, the OFDM receiver 11 can obtain a correlation value sequence by performing a correlation operation between a time waveform prepared in advance and a received signal. For example, in FIG. 3, a signal sequence output from the complex correlators 31 to 33 corresponds to the correlation value sequence.
[0056]
FIG. 4 shows an example of a correlation value sequence output from the complex correlator 31. This correlation value sequence corresponds to the impulse response of the radio channel. That is, it represents the delay profile. If the propagation path can be modeled in two passes, two correlation peaks can be observed.
[0057]
The correlation operation between the signal that has received the signal transmitted in the pilot symbol and the signal prepared in advance by the receiver corresponds to an autocorrelation operation. FIG. 5 shows a sequence obtained by this autocorrelation calculation. A correlation peak as shown in FIG. 5 appears in the series obtained by the autocorrelation calculation. The portion other than the correlation peak becomes a correlation side lobe. The characteristics of the correlation side lobe depend on the autocorrelation characteristics of the time waveform of the OFDM signal.
[0058]
The correlation peak in FIG. 5 indicates the position of a known signal having good autocorrelation characteristics included in the received signal. By detecting this position, burst frame synchronization can be established.
[0059]
In a peak search using a known signal having good autocorrelation characteristics included in a received signal, if there is a frequency error, the signal level output from the complex correlators 31 to 33 decreases, and the synchronization characteristics deteriorate. . However, in the OFDM signal communication system according to the present embodiment, as shown in FIG. 3, a plurality of complex correlators 31 to 33 are used as a reference sequence with a signal given a different frequency offset for a signal transmitted in a pilot symbol. Thus, by taking a correlation corresponding to a plurality of frequency offsets, it is possible to prevent a decrease in the correlation output level due to the influence of the frequency error and to suppress the deterioration of the synchronization characteristics.
[0060]
Frequency offset f for the signal transmitted in the pilot symbol_kThe reference sequence d given_kIs represented by equation (6).
[0061]
(Equation 6)

In equation (6), u_kIs the discrete value of the transmitted signal and u_k= S (k / N). However, N is the number of FFT points.
[0062]
Frequency offset f for the signal transmitted in the pilot symbol_kThe frequency response a (Δf) between the reference sequence given by and the correlation output of the received signal is expressed by equation (7).
[0063]
(Equation 7)

In equation (7), di^*Is the complex conjugate of di. Here, assuming that the effects of noise, sampling timing shift, and phase offset can be ignored, Equation (7) is transformed into Equation (8).
[0064]
(Equation 8)

Equation (8) shows that the frequency response of the correlation output depends on the signal sequence transmitted in the pilot symbol.
[0065]
In equation (8), | u (i) |²Represents the size of the transmission signal sequence, and exp (j2π (Δf−f_k) · I / N) is the reference sequence d_kRepresents the frequency shift with respect to. The vector representing the frequency shift rotates according to the sequence number, and the amount of rotation depends on the frequency shift. That is, exp (j2π (Δf−f_kThe sequence represented by) .i / N) is a vector whose direction is rotated according to the sequence number and whose length is the size of the transmission signal sequence.
[0066]
FIG. 6 shows the frequency response shown in equation (8), that is, the magnitude | a (Δf) | of the correlation output between the reference sequence in which the frequency offset Δf is given to the signal transmitted in the pilot symbol and the received signal.²FIG. 5 is a diagram showing an example of the frequency response of FIG. However, in FIG._kIs 0. FIG. 6 shows that the correlation value becomes maximum when the frequency offset is 0.
[0067]
That is, when reference sequences having different frequency offsets are respectively set in the plurality of complex correlators 31 to 33 shown in FIG. 3, the correlation between the reference sequence and the received signal output from each of the complex correlators 31 to 33 is set. The maximum value of the output absolute value is searched, and the difference between the carrier frequency of the OFDM transmission signal and the carrier frequency of the OFDM reception signal can be detected by the frequency offset corresponding to the complex correlator that has output the maximum value.
[0068]
3 executes the above-described search procedure, and outputs a carrier frequency error signal and a correlation output maximum signal.
[0069]
In FIG. 6, the magnitude | a (Δf) | of the correlation output between two reference sequences A and B having different known signal sequences transmitted in pilot symbols and a received signal.²An example of the frequency response is shown. From FIG. 6, the magnitude of the correlation output between the reference sequence and the received signal | a (Δf) |²It can be seen that the frequency response of is dependent on the known signal sequence.
[0070]
3, the shift register 35 connected to the maximum value detection unit 34 sequentially shifts the maximum value signal of the correlation output output from the maximum value detection unit 34, and the maximum value detection unit 36 The maximum value is detected from the output of. This maximum value indicates the autocorrelation peak of the sequence obtained by the above-described autocorrelation calculation, and the peak position indicates the reference timing of the burst frame of the signal received via the radio channel.
[0071]
The maximum value detection unit 36 outputs a timing error signal that is an error between the reference timing of the detected burst frame and the timing of the OFDM receiver 11. Further, the maximum value detection unit 36 detects the maximum value of the correlation output output by the complex correlator that outputs the maximum value of the correlation output detected by the maximum value detection unit 34, and outputs the maximum value of the correlation output. A frequency offset corresponding to the timing is output as a frequency error signal.
[0072]
Note that the frequency offset amount obtained here is a discrete value depending on the number of complex correlators. Therefore, the accuracy of the frequency error signal can be improved by performing the interpolation calculation.
[0073]
With the above configuration, it is possible to detect a sampling timing error and a carrier frequency error, and to realize an OFDM signal communication system capable of highly reliable signal transmission.
[0074]
As described above, in an OFDM signal communication system that handles multimedia information and the like, since a signal to be transmitted occurs in a burst, it is necessary to establish synchronization in a short time for each burst frame. For high-speed initial synchronization, an increase in the circuit scale to some extent is inevitable. However, since an increase in the circuit scale leads to an increase in manufacturing cost, it is desirable to reduce the circuit scale as much as possible. In other words, how to make a complex sliding correlator having a relatively large proportion of the entire circuit load a simple configuration is an important technical problem.
[0075]
Therefore, a method for reducing the circuit scale of the complex correlator shown in FIG. 3 will be described below. The method described below focuses on a correlation operation with a reference sequence in which a different frequency offset is given to a signal transmitted in a pilot symbol.
[0076]
As described above, the correlation output a (Δf) between the received signal and the reference sequence to which the frequency offset Δf is applied to the signal transmitted in the pilot symbol can be expressed by the above-described equation (8). For simplicity of description, equation (8) assumes that the effects of noise, sampling timing shift, and phase offset can be ignored.
[0077]
As shown in FIG. 7, the vector representing the frequency shift rotates according to the sequence number of the signal sequence of the pilot signal, and the amount of rotation depends on the frequency shift. When Δf = 1, this vector makes one rotation. That is, the sequence represented by exp (j2π (Δf−fk) i / N) is a vector whose direction rotates according to the sequence number and whose length is the size of the transmission signal sequence.
[0078]
Where | u (i) |²-Consider the features of exp (-j2π (Δf-fk) i / N). The size of the transmission signal sequence | u (i) |²This will be described with reference to an example in which the change is small.
[0079]
8 and 9 show examples of the vector locus. The number of FFT points N = 64, the number of subcarriers constituting the OFDM signal Ns = 4, FIG. 8 shows Δf = 0.25, and FIG. 9 shows Δf = 0.5.
[0080]
The vector trajectory becomes as shown in FIGS. 8 and 9 because the magnitude of the transmission signal sequence | u (i) |²Can be understood by observing
[0081]
FIG. 10 shows the magnitude | u (i) | of the transmission signal sequence corresponding to FIGS. 8 and 9.²FIG. The horizontal axis in FIG. 10 is the sample number i, and the vertical axis is | u (i) |²It is.
[0082]
From FIG. 8 to FIG. 10, | u (i) |²It can be confirmed that the sequence represented by exp (-j2π (Δf-fk) i / N) is a vector whose direction rotates in accordance with the sequence number and whose length is the size of the transmission signal sequence. Further, the correlation output a (Δf) represented by the equation (8) is calculated from the vector sum of these trajectories.
[0083]
Here, attention is paid to the distribution of the vector trajectories shown in FIGS. 8 and 9 that there are many distributions near the origin. This means that the size of the transmission signal sequence | u (i) |²Distribution is high near 0, and | u (i) |²Indicates that the distribution with large values is small. Since the correlation output a (Δf) represented by the equation (8) is the vector sum of these trajectories, the magnitude | u (i) | of the transmission signal sequence²The effect when takes a large value becomes dominant.
[0084]
For this reason, in the present embodiment, the reference sequence is represented by only some representative vectors.
[0085]
In FIGS. 8 and 9, the size of the transmission signal sequence | u (i) |²The example in the case where the change of the value is small was shown. However, the time waveform of OFDM is characterized by relatively large fluctuations. Conversely, small changes are rare.
[0086]
FIG. 11 is a diagram illustrating an example of a vector trajectory when QPSK-mapped data is transmitted. In FIG. 11, the number of FFT points N = 32 and the number of subcarriers constituting the OFDM signal Ns = 32. FIG. 11A shows the case where the frequency offset Δf = 0.4 given to the signal transmitted in the pilot symbol, and FIG. 11B shows the case where Δf = 0.9.
[0087]
From FIG. 11, the size of the transmission signal sequence | u (i) |²Distribution is high near 0, and | u (i) |²It can be seen that the distribution in which is large is slight.
[0088]
In the present embodiment, the reference sequence is represented by only some representative vectors. Hereinafter, an example of the representative vector will be described with reference to FIGS.
[0089]
FIG. 12 is a diagram showing a vector locus when random data is transmitted. FIG. 12 shows an example in which the number of FFT points N = 64, the number of subcarriers constituting the OFDM signal Ns = 64, and the frequency offset Δf = 1.0. From FIG. 12, the size of the transmission signal sequence | u (i) |²It can be seen that there are only a few vectors with large values.
[0090]
For this reason, in the present embodiment, the vector in FIG. 12 is represented by a representative vector as shown in FIG. That is, the reference sequence is represented by a vector in which a vector other than the representative vector as shown in FIG.
[0091]
(First Configuration Example of Complex Correlator)
Next, a specific example of the complex correlators 31 to 33 shown in FIG. 3 will be described. FIG. 14 is a block diagram showing a first configuration example of the complex correlator. The complex correlator 50 in FIG. 14 includes a shift register 51 having L stages, a reference sequence 52, M complex multipliers 53, and an adder 54 for adding the output of each complex multiplier 53.
[0092]
The reference sequence 52 is a representative vector corresponding to a sequence number having a large value in a time domain of a signal transmitted in a pilot symbol. The multiplier 53 is arranged at a position corresponding to the representative vector number described above. FIG. 14 shows an example where L = 6 and the numbers of the representative vectors are 1, 4, and 6.
[0093]
(Second configuration example of complex correlator)
FIG. 15 is a block diagram showing a second configuration example of the complex correlator. The complex correlator 60 in FIG. 15 includes shift registers (second and third signal shift means) 61 and 62 having L stages, M complex multipliers 63, and outputs of the complex multipliers 63. And an adder 64 for adding. The multiplier 63 is arranged at a position corresponding to the representative vector number described above. FIG. 15 shows an example where L = 6 and the numbers of the representative vectors are 1, 4, and 6.
[0094]
Also in the complex correlator 60 of FIG. 15, the number of the multipliers 63 can be reduced, so that an increase in the circuit scale can be suppressed and the manufacturing cost can be reduced.
[0095]
FIG. 16 is a block diagram showing a part of the configuration of the OFDM receiver 11 having the complex correlator 60 of FIG. The OFDM receiver 11 in FIG. 16 includes a sampler 19 for sampling an OFDM reception signal, a one-symbol delay unit (delay means) 65, and a complex correlator 60 in FIG.
[0096]
In the OFDM receiver 11 of FIG. 16, the complex correlator 60 can obtain the correlation peak of the burst frame data in which two consecutive pilot symbols are arranged, and the maximum value of the signal output from the complex correlator 60 By performing a search, a burst frame can be detected.
[0097]
(Third Configuration Example of Complex Correlator)
In FIG. 14 and FIG. 15 described above, an example is described in which a complex multiplier is arranged corresponding to a representative vector having a large value in the time domain of a signal transmitted in a pilot symbol. If a cyclic sequence signal is transmitted as a signal to be transmitted, complex multipliers can be arranged at equal intervals.
[0098]
Hereinafter, a known signal sequence in which complex multipliers can be arranged at equal intervals will be described. This known signal sequence is transmitted at the timing of transmitting the pilot signal.
[0099]
Reference sequence d_kIs represented by several representative vectors, a sequence in which the phase arrangement in the frequency domain is set so as to circulate around the subcarriers is adopted as the known signal sequence. Here, such a sequence is called a cyclic sequence.
[0100]
In the cyclic sequence, assuming that the repetition period is Ig, a sequence {xk} of a time response obtained by IFFT of a known signal sequence {Xk} has a value for each N / Ig.
[0101]
That is, by using a cyclic sequence for pilot symbols, the number of complex multipliers can be reduced, and an increase in circuit scale can be suppressed as much as possible.
[0102]
The cyclic sequence is represented by equation (9).
X_{k + 1}・ X_k ^*= Exp (jφ_k  modulo I_g…… (9)
Here, k (k = 0, 1,..., N−1) is a subcarrier number, φ_iIs a phase change between subcarriers.
[0103]
There are a plurality of sequences satisfying the expression (9). However, in the cyclic sequence, since power is concentrated on a specific sample, it is disadvantageous from the point of PAPR (Peak to Average Power Ratio). The large PAPR is one of the disadvantages of the OFDM modulation method. PAPR is preferably as small as possible from the viewpoint of power efficiency.
[0104]
Therefore, a cyclic sequence in which power is equally distributed to each representative vector, that is, | xi |²= N / Ig needs to be selected. Here, i represents the number of the representative vector and N represents the number of points of the FFT. The cyclic period Ig matches the number of representative vectors.
| Xi |²Equation (10) is an example of a series where = N / Ig.
(Φ0, φ1, φ2, ..., φ15)
= (0, π / 2, -π / 2, π, π / 2, -π / 2, π, 0, π, π / 2, π / 2, π, -π / 2, -π / 2, 0,0)
… (10)
Here, Ig = 16 and Xk are QPSK mappings.
[0105]
In this sequence, the intervals between the representative vectors are equal intervals (i = 0, N / Ig, 2 · N / Ig,..., (Ig−1) · N / Ig).
[0106]
FIG. 17 is a block diagram showing a third configuration example of a complex correlator when a cyclic sequence pilot signal is used. When a cyclic sequence is used as a known signal sequence, its time response has a value at regular intervals, and therefore, the multipliers 102 are arranged at regular intervals. FIG. 17 shows an example where the number of multipliers is four. Note that the number of multipliers 102 matches the cyclic period Ig. That is, by using a cyclic sequence for pilot symbols, the number of complex multipliers in a sliding correlator can be reduced, and an increase in circuit size can be suppressed.
[0107]
Next, the configuration of a complex correlator when transmitting a cyclic sequence pilot signal will be described.
[0108]
(Fourth Configuration Example of Complex Correlator)
FIG. 18 is a block diagram showing a fourth configuration example of the complex correlator. The complex correlator 70 of FIG. 18 includes shift registers 71 and 72 each having L stages, M complex multipliers 73, and an adder 74 for adding the output of each complex multiplier 73. The multiplier 73 is arranged at a position corresponding to the number of the representative vector.
[0109]
Since the pilot signal is a cyclic sequence, multipliers 73 can be arranged at equal intervals as shown in FIG.
[0110]
The complex correlator of FIG. 18 has the same configuration as that of FIG. 15 except that the multipliers 73 and 85 are arranged at equal intervals. Therefore, the same operation and effect as those in FIG. 15 can be obtained.
[0111]
(Fifth Configuration Example of Complex Correlator)
FIG. 19 is a block diagram showing a fifth configuration example of the complex correlator. A complex correlator 90 shown in FIG. 19 is a configuration for transmitting a pilot signal of a cyclic sequence, a serial / parallel converter 91 for converting an input sequence into parallel at a rate of 1: K, and a shift register having M stages. 92, M complex multipliers 93, an adder 94 for adding the output of each complex multiplier 93, and a reference sequence 95. The reference sequence 95 is a representative vector corresponding to a sequence number having a large value in a time domain of a signal transmitted in a pilot symbol.
[0112]
The complex correlator 20 of FIG. 19 differs from the complex correlator 20 of FIG. 18 in performing a complex operation between a received pilot signal and a reference sequence. Also in the case of the complex correlator 90 of FIG. 19, the number of multipliers is small, so that an increase in the circuit scale can be suppressed and the manufacturing cost can be reduced.
[0113]
(Another configuration example of the frequency error & timing error estimator)
FIG. 20 is a block diagram showing another configuration example of the frequency error & timing error estimator. The signal output from the sampler 19 shown in FIG. 1 is input to the frequency error & timing error estimator 20a in FIG. The frequency error & timing error estimator 20a outputs a frequency error signal supplied to the local oscillator 18 and a timing error signal supplied to the FFT 21.
[0114]
The frequency error & timing error estimator 20a of FIG. 20 includes a shift register 30, a plurality of complex correlators 31 to 33, a maximum value detecting unit 34, a comparator 111, and an average received power measuring unit 112. Different reference sequences 37 to 39 are set in the complex correlators 31 to 33, respectively. Each of the complex correlators 31 to 33 includes a complex multiplier 41 that multiplies a signal input from the shift register 30 by a reference sequence, and an adder 42 that adds an output of the complex multiplier 41. Reference sequences 37 to 39 set in complex correlators 31 to 33 are signals obtained by giving different frequency offsets to signals transmitted in pilot symbols.
[0115]
The maximum value detector 34 detects the maximum value of the correlation value output from each of the complex correlators 31 to 33, and outputs the maximum value of the correlation value and the frequency offset at that time. The carrier frequency error can be detected from the frequency corresponding to the complex correlator that has output the maximum value. The frequency offset amount obtained here is a discrete value depending on the number of complex correlators. Therefore, a frequency error is obtained by interpolation using a complex correlator that outputs the maximum value and an output from a complex correlator adjacent to the complex correlator.
[0116]
By detecting the maximum value of each output signal of the plurality of complex correlators, it is possible to detect the position of a known signal having a good autocorrelation characteristic included in the received signal, thereby establishing synchronization of burst frames. .
[0117]
The correlation value and the frequency offset amount output from the maximum value detection unit 34 are input to the comparator 111. The output signal of the average received power measuring unit 112 is input to the comparator 111. The signal output from the sampler 19 shown in FIG. 1 is input to the average received power measuring unit 112, and the average received power value is output. Average received power measuring section 112 measures the average power of the received signal. The comparator 111 sets a comparison value based on the information on the average power of the received signal, and performs a comparison operation between the comparison value and the maximum value of the correlation value output from the maximum value detection unit 34.
[0118]
By this comparison operation, the comparator 111 finds an error between the position of the known signal having good autocorrelation characteristics included in the received signal and the internal clock, and outputs the error value as a timing offset. Since the average power increases as the absolute value of the correlation output output from the complex correlators 31 to 33 increases, the comparator 111 shown in the figure compares the power measured by the average power measurement unit 112 with a preset comparison value. The frequency error signal and the correlation output maximum value signal are output based on the above.
[0119]
With the configuration as shown in FIG. 20, it is possible to detect a sampling timing error and a carrier frequency error, and it is possible to realize an OFDM signal receiving device capable of highly reliable signal transmission.
[0120]
In the above-described embodiment, an example in which the frequency error signal and the timing error signal are output from the maximum value detection unit 36 has been described. However, only the frequency error signal may be output and only the carrier frequency error adjustment may be performed.
[0121]
【The invention's effect】
As described in detail above, according to the present invention, based on the result of performing a complex correlation operation between a received pilot signal and a reference signal having different frequency offsets set for the pilot signal, Since the frequency of the reproduction carrier signal is controlled, synchronization of the received frame can be quickly established, and stable and reliable signal transmission and reception can be performed. Further, since the internal configuration of the complex correlation operation means can be simplified, the increase in circuit scale can be suppressed, and the manufacturing cost can be reduced accordingly.
[Brief description of the drawings]
FIG. 1 is a block diagram of an embodiment of an OFDM signal communication system including an OFDM signal receiving apparatus according to the present invention.
FIG. 2 is a diagram showing a data configuration of a burst frame of an OFDM signal transmitted from an OFDM transmitter.
FIG. 3
FIG. 2 is a block diagram showing an internal configuration of the frequency error & timing error estimator shown in FIG. 1.
FIG. 4
The figure which shows the example of the correlation value sequence output from a complex correlator.
FIG. 5
The figure which shows the series obtained by an autocorrelation calculation.
FIG. 6
Magnitude of correlation output | a (Δf) | between a reference sequence having a frequency offset Δf given to a signal transmitted in a pilot symbol and a received signal²The figure which shows the example of the frequency response of FIG.
FIG. 7
FIG. 4 is a diagram showing a vector representing a frequency shift.
FIG. 8
The figure which shows an example of the vector locus in case of (DELTA) f = 0.25.
FIG. 9
The figure which shows an example of the vector locus in case of (DELTA) f = 0.5.
FIG. 10
Size of transmission signal sequence corresponding to FIGS. 8 and 9 | u (i) |²FIG.
FIG. 11 is a diagram showing an example of a vector locus when QPSK-mapped data is transmitted.
FIG.
The figure which shows the vector locus at the time of transmitting random data.
FIG. 13
The figure which shows the vector locus which set other than the representative vector to 0.
FIG. 14 is a block diagram showing a first configuration example of a complex correlator.
FIG. 15 is a block diagram showing a second configuration example of the complex correlator.
FIG. 16 is a block diagram showing a part of the configuration of an OFDM receiver having the complex correlator of FIG. 15;
FIG. 17 is a block diagram showing a third configuration example of a complex correlator when a cyclic sequence is used for a pilot signal.
FIG. 18 is a block diagram showing a fourth configuration example of the complex correlator.
FIG. 19 is a block diagram showing a fifth configuration example of the complex correlator.
FIG. 20 is a block diagram showing another configuration example of the frequency error & timing error estimator.
FIG. 21 is a block diagram showing the configuration of an OFDM modulator used on the transmission side.
FIG. 22 is a diagram illustrating a guard interval.
FIG. 23 is a block diagram showing a configuration of an OFDM demodulator used on the receiving side.
FIG. 24 is a diagram showing a frequency offset detection / removal method described in the literature.
[Explanation of symbols]
10 OFDM transmitter
11 OFDM receiver
12 Inverse Fast Fourier Transformer (IFFT)
13,17 Frequency converter
14,18 Local oscillator
15, 16 antenna
19 Sampler
20 Timing error estimator
21 Fast Fourier Transformer (FFT)
30,35 shift register
31-33 complex correlator
34, 36 Maximum value detector
37-39 reference series
40 complex multiplier
41 Adder

Claims (2)

Receiving means for receiving by an antenna an OFDM (orthogonal frequency division multiplex) signal into which a pilot signal composed of a predetermined known signal sequence is inserted;
A local oscillator that generates a reproduced carrier signal that is a reference signal for demodulating the OFDM signal,
A frequency converter that converts the OFDM signal to a baseband signal based on the reproduced carrier signal,
An FFT operation unit that performs an FFT operation based on the baseband signal to generate reception data in a frequency domain, comprising:
Sampling means for sampling the baseband signal;
A plurality of complex correlation operation means for performing a complex correlation operation between the reference signal and the pilot signal each having a different frequency offset set for the pilot signal to obtain a correlation output,
Based on the correlation output obtained by each of the plurality of complex correlation operation means, a frequency error between the reproduced carrier signal and the carrier signal of the OFDM signal, and a timing error when the FFT operation means performs the operation. Error estimating means for estimating,
The local oscillator controls the frequency of the reproduced carrier signal based on the frequency error,
The FFT operation means determines an FFT window based on the timing error,
Each of the plurality of complex correlation operation means,
A first signal shift means having L (L is an integer of 2 or more) shift stages to which the baseband signal sampled by the sampling means is input;
M complex multiplying means for multiplying the output of M stages (M is an integer of 2 or more and M <L) selected from the stages of the first signal shifting unit and the reference signal, respectively;
An adding means for adding the results of the multiplication by the M complex multiplication means to calculate the correlation output. The OFDM signal includes at least two of the pilot signals,
Delay means for delaying the baseband signal sampled by the sampling means;
A second signal shift unit having L (L is an integer of 2 or more) shift stages to which the baseband signal sampled by the sampling unit is input;
Third signal shift means having L (L is an integer of 2 or more) shift stages to which the baseband signal delayed by the delay means is input;
The output of M stages (M is an integer of 2 or more and M <L) selected from the respective stages of the second signal shifter and the output of the respective stages of the third signal shifter are selected. M multiplying means for multiplying the output of the M stages,
The OFDM signal receiving apparatus according to claim 1, further comprising: an adding unit that adds each multiplication result obtained by the M multiplying units.

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