JP3518997B2 - Power converter - Google Patents

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power converter for converting power between alternating current and direct current.

[0002]

2. Description of the Related Art FIG. 16 shows, for example, Japanese Patent Laid-Open No. 8-15292.
It is a block diagram which shows the structure of the conventional power converter device which prevents the DC bias magnetism of the self-excited converter connected to an alternating current power system through the several transformer shown by the 7th publication. Figure 1
In 6, the reference numeral 1 is an AC power system, 2A to 2D are transformers,
3A to 3D are GTO thyristor, GCT thyristor, I
A self-exciting converter composed of a semiconductor element such as a GBT and a transistor capable of self-extinguishing, capable of AC / DC conversion, 4 a DC voltage source such as a capacitor, 5 a current detector, 6 a voltage detector, 7 an output current Reference, 8 is a system voltage reference, 9 is a voltage / current control circuit, 10A to 10D are transformer bias magnetic amount calculation units, 1
1A to 11D are output voltage DC component correction control units, 12 are DC component correction decoupling units, 13A to 13D are adders, and 14A to
Reference numeral 14D is a PWM control circuit, and 15A to 15D are gate pulse amplification circuits.

Next, the operation of the conventional power converter shown in FIG. 16 will be described. In the power converter shown in FIG. 16, the voltage of the AC power system 1 or the self-excited converter 3
When the AC side output voltage of A to 3D includes a DC component, an exciting current including the DC component flows in the transformers 2A to 2D, and the DC component of the exciting current is the transformer 2
A-2D is demagnetized, and as a result, the iron cores of the transformers 2A-2D are saturated. When the iron cores of the transformers 2A to 2D are saturated, the transformer iron cores have the excitation characteristics shown in FIG. 17, so that a slight voltage change causes a rapid increase in the current, which causes a deviation in the self-excited converters 3A to 3D. Overcurrent occurs in the converter connected to the transformer iron core that has caused magnetic saturation, and protection stops.

The amount of eccentricity that causes the above-mentioned transformers 2A to 2D to reach the eccentricity saturation is determined by the transformer eccentricity amount calculators 10A to 10D, which are connected to the self-excited converters 3A to 3D.
Electricity P required to calculate the deviation of the magnetic flux of the iron core of D
11 to P4n are detected, and the deviation of the magnetic flux of the iron cores of the transformers 2A to 2D is calculated based on the detected amount of electricity. Based on the result, the output voltage DC component correction control units 11A to 11D calculate temporary correction values for the AC side output voltage control of the self-excited converters 3A to 3D necessary to eliminate the bias of the magnetic flux. . In the DC component correction decoupling unit 12, each self-excited converter 3A to 3D.
In the above, the temporary correction values of the self-excited converters other than the self-excited converter are added based on the decoupling principle, and the final correction value for the AC side output voltage control of the self-excited converters 3A to 3D becomes It is calculated.

This final correction value is obtained from the output current reference 7, the current feedback value detected by the current detector 5, the system voltage reference 8, and the voltage feedback value detected by the voltage detector 6, based on the voltage-current control circuit. The self-excited converter for eliminating the bias of the magnetic flux of the iron cores of the transformers 2A to 2D by adding the voltage command values of the self-excited converters 3A to 3D created in 9 with the adders 13A to 13D. The AC side output voltage of 3A to 3D is corrected. In the PWM control circuits 14A to 14D, the adder 13 is used.
The ignition pulse width signals of the semiconductor elements of the self-excited converters 3A to 3D are created based on the corrected voltage command values output from A to 13D, and the gate pulse amplifier circuits 15A to 15D generate self-excited converters 3A to 3D. Amplification of the gate pulse signal for firing the semiconductor element is performed.

[0006]

Since the conventional power converter is configured as described above, the DC component correction decoupling unit causes the mutual interference of the bias magnetic control by the output of the output voltage DC component correction control unit. However, it is possible to suppress
In the non-interference principle disclosed in Japanese Patent No. 52927, FIG.
Only valid for independent transformer configurations of the magnetic circuit with individual cores as shown in 6.

On the other hand, in a large-capacity self-exciting converter, in order to keep the transformer volume small, a multiple transformer called an outer iron type multiple transformer shown in FIG. 18 or an inner iron type multiple transformer shown in FIG. In many cases, since the magnetomotive force created by the windings of each stage has a common magnetic path, it affects the magnetic flux of the iron cores of other stages via the common magnetic path. For example, in the case of the outer iron type multiple transformer shown in FIG. 18, when the leakage magnetic flux is ignored and only one side of the iron core is considered as shown in the figure, the magnetomotive force F1 generated in the first-stage winding passes through the magnetic path M11. A magnetic flux Φ11 is created by shunting the magnetic flux Φ11 in a magnetic path M12 and a magnetic path M13 to create a magnetic flux Φ12 and a magnetic flux Φ13. Similarly, the magnetic flux Φ13 that has passed through the magnetic path M13 is shunted into the magnetic path M14 and the magnetic path M15 to form the magnetic flux Φ14 and the magnetic flux Φ15, and the magnetic flux 15 and the magnetic flux 14 merge to form the magnetic flux 16. Magnetic flux Φ11
Therefore, the relationship of φ16 is as shown in the following equations (Equation 1) to (Equation 3). Φ13 = Φ11−Φ12 (Equation 1) Φ15 = Φ13−Φ14 (Equation 2) Φ16 = Φ14 + Φ15 (Equation 3)

A magnetic flux path passing through the magnetic path M11, magnetic path M13, magnetic path M15, and magnetic path M16 is called a main iron core. When there is no magnetic flux Φ12 passing through the magnetic path M12 and magnetic flux Φ14 passing through the magnetic path M14, From 1) to (Equation 3) and Φ12 = Φ14 = 0, Φ11 = Φ16 = Φ15, and it is understood that all magnetic flux passes through the main iron core and the magnetic flux states of the iron core are the same in all 1 to 3 stages. However, in reality, the magnetic path M12 and the magnetic path M1
Since there are magnetic fluxes Φ12 and Φ14 passing through 4, (Equation 1)
~ As seen from (Equation 3), the magnetic flux state Φ of the iron core at each stage
11, φ16, and φ15 are not always the same. In this case, for example, the magnetomotive force F1 generated in the first-stage winding affects the magnetic fluxes Φ16 and Φ15 of the other stages through the magnetic paths M11 to M16 according to (Equation 1) to (Equation 3). The magnetic fluxes Φ11, Φ16, and Φ15 of the iron core of each stage cannot be accurately calculated unless the magnetomotive force generated by the winding of the stage is taken into consideration.

It should be noted that this phenomenon occurs similarly in the case of a transformer structure having a common magnetic path between the windings of each stage. When the inner iron type multiple transformer shown in FIG. 19 is used. However, for example, the magnetomotive force F1 generated by the first winding is the magnetic path M
1. The magnetic flux Φ5 of the iron core of the magnetic path M5 is affected via the magnetic path M2, the magnetic path M3, and the magnetic path M4. Therefore, if the magnetomotive force generated by the windings of the other steps is not taken into consideration Cannot accurately calculate the magnetic flux state of.

However, in the conventional method for obtaining the amount of bias magnetization correction by using the non-interference principle disclosed in Japanese Patent Laid-Open No. 8-152927, the transformer cores are independent transformers that do not have a common magnetic path. It is considered only in the case of multiple connection using a device. Transformer bias magnetic amount calculation unit 10A
-10D, output voltage DC component correction control units 11A to 11D,
Neither of the DC component correction decoupling units 12 accurately considers the effect of the magnetomotive force generated by the windings of the other stages, so that the transformer core has the outer iron type multiple transformer shown in FIG. In the case where the inner iron type multiple transformer shown in FIG. 19 has a common magnetic path and is magnetically influenced, the bias correction amount may be erroneously output.

As described above, in the conventional power converter, for example, the self-excited converters are connected in multiple via the outer iron type multiple transformer shown in FIG. 18 or the inner iron type multiple transformer shown in FIG. Since it is not possible to consider the effect of the magnetomotive force created by the windings of the other stages in this case, it is not possible to calculate the accurate transformer bias amount and the bias bias correction amount calculated from the transformer bias amount, There is a problem that it is difficult to obtain a sufficient DC bias magnetic suppression effect. The present invention has been made to solve the above problems, and even when a multiple transformer having a common magnetic path is used, the magnetic flux of the transformer iron core is correctly calculated and the direct current of the transformer is calculated. An object of the present invention is to obtain a power conversion device capable of suppressing magnetic bias.

[0012]

According to a first aspect of the present invention, there is provided a power converter including a plurality of power converters having switching elements, an iron core having a plurality of leg irons magnetically coupled to each other, and a leg iron. A multiple transformer having a plurality of DC side windings wound around and connected to the AC sides of a plurality of power converters, and a plurality of AC side windings wound around a leg iron and connected to an AC power source, and a power converter Voltage command creating means for calculating the AC side output voltage command, and voltage command correction calculating means for calculating the voltage command correction for correcting the AC side output voltage command according to the magnetic flux states of the plurality of leg irons of the multiple transformer. And the AC side output voltage of the power converter is controlled based on the output voltage command and the voltage command correction.

A power conversion device according to a second aspect of the present invention is the power conversion device of the first aspect.
In the above, the voltage command correction calculation means calculates the temporary voltage command correction for each leg iron individually from the electric quantities of the DC side winding and the AC side winding wound on the leg irons of the multiple transformer. And a correction value correction means for calculating the voltage command correction from the provisional voltage command correction using the relationship between the amount of electricity of the DC side winding and the AC side winding between the leg irons and the magnetic flux state. is there. According to a third aspect of the present invention, in the power conversion device according to the second aspect, the relationship between the amount of electricity and the magnetic flux state is a relationship expressed by using a magnetic resistance. A power conversion device according to a fourth aspect is the power conversion device according to the first aspect , wherein the voltage command correction calculation means is a plurality of direct currents of a multiple transformer.
Magnetic flux of multiple leg irons from the electric quantities of the side winding and AC side winding
Magnetic flux state calculation means for calculating the state and this magnetic flux state calculation
Calculate the voltage command correction from the magnetic flux state calculated by
And a correction value calculation means for outputting the magnetic flux state calculation means.
The step has a non-linear characteristic that changes according to the amount of electricity.
To calculate the magnetic flux state. A power conversion device according to a fifth aspect is the power conversion device according to the second aspect,
The voltage command correction is calculated as a non-linear characteristic that changes the relationship between the electric quantity and the magnetic flux state according to the input electric quantity.

A power conversion device according to a sixth aspect of the present invention is the power conversion device of the first aspect.
In the above, a plurality of magnetic detectors are provided in the vicinity of the iron core of the multiple transformer, and the voltage command correction calculation means calculates the voltage command correction from the outputs of these magnetic detectors. Power converting apparatus according to claim 7, claim
6 , the voltage command correction calculation means calculates the magnetic flux status calculation means for calculating the magnetic flux status of the plurality of leg irons from the outputs of the plurality of magnetic detectors, and the voltage command correction calculation from the outputs of the magnetic flux status calculation means. It comprises a correction value calculation means.

An electric power converter according to an eighth aspect is the seventh aspect.
, The magnetic flux state calculation means includes a plurality of magnetic detection
Using the magnetic resistance of the output of the vessel and the magnetic flux state of multiple leg irons
From the output of the magnetic detector using the relationship
The magnetic flux state is calculated. Claim 9
The power converter according to claim 7, wherein the magnetic flux is
The state calculation means is composed of outputs of a plurality of magnetic detectors and a plurality of leg bars.
The magnetic flux state of the
Calculate the magnetic flux state as a nonlinear characteristic that changes
It is something that is done. According to a tenth aspect of the power conversion device of the third or eighth aspect , the magnetic resistance is changed by the magnetomotive force.

[0016]

BEST MODE FOR CARRYING OUT THE INVENTION Embodiment 1. 1 is a block diagram showing a configuration of a power conversion device according to a first embodiment of the present invention. In FIG. 1, 1 is an AC power system that is an AC power source, 2 is a multiple transformer, and three leg irons 201, 202, 203 of the first to third stages and these leg irons 201 to 203 are magnetically connected to each other. Of the iron core 200 having the yoke 204 and the inter-stage magnetic path 205 coupled to each other, and the three DC sides respectively wound around the leg irons 201 to 203 and connected to the AC sides of the three power converters 3A to 3C described later. Windings (hereinafter referred to as secondary windings) 211, 212, 2
13 and three AC side windings (hereinafter referred to as primary windings) 221, 222, 223 that are respectively wound on the leg irons 201 to 203 and connected in series to each other to the AC power system 1.
And have. 3A to 3C are GTO thyristors, GC
A self-exciting power converter that is composed of a semiconductor switching element such as a T thyristor, an IGBT, and a transistor that can be self-extinguished, and is capable of AC / DC conversion, 4 is a capacitor, a DC voltage source such as a DC power supply, 51, 52A to 52C are currents The current detector 51 detects a current (hereinafter, referred to as a primary current) with the AC power system 1, and the current detectors 52A to 52C respectively detect the AC side currents of the power converters 3A to 3C. Hereinafter, it will be referred to as a secondary current). 6 is a voltage detector that detects the voltage of the AC power system 1, 7 is an output current reference, 8 is a system voltage reference, and 9 is a voltage / current control circuit that is a voltage command creating means. Voltage detector 6 and current detector 5
In addition to using the output of 1, the set values of the output current reference 7 and the system voltage reference 8 are used as the setting conditions to calculate the AC side output voltage command of the power converters 3A to 3C. 10A to 10C are transformer bias magnetic amount calculation units, 11A
11C are output voltage DC component correction control units, 13A to 13C.
Is an adder, 14A to 14C are PWM control circuits, and 15A to
Reference numeral 15C is a gate pulse amplifier circuit, 16 is a transformer magnetic flux state calculation unit as magnetic flux state calculation means for calculating the magnetic flux state of the iron core 200, and transformer bias magnetic field amount calculation units 10A to 10C.
The output voltage DC component correction control units 11A to 11C constitute a correction value calculating unit, and the correction value calculating unit and the transformer magnetic flux state calculating unit 16 constitute a voltage command correction calculating unit.

FIG. 2 is a block diagram showing a configuration example of the transformer magnetic flux state calculation unit 16 of this embodiment, and FIG. 3 is an explanatory diagram showing a magnetic circuit model of the multiple transformer 2. In FIG. 2, 171, 172A to 172C are gains that multiply the number of turns of the primary and secondary windings of the transformer, 181A to 181C are subtractors that perform subtraction calculation, and 19 is gains 171, 172A to 1
72C and a subtraction unit 181A to 181C are made into a block, and a magnetomotive force calculation part, 173AA to 173AC, 1
73BA to 173BC and 173CA to 173CC are gains that are multiplied by a proportional constant according to magnetic resistance, 131A to 131A.
C is an adder for performing a three-input addition operation, 20A to 20C
Are gains 173AA to 173AC, 173BA to 17
3BC, 173CA to 173CC and adder 131A
To 131C in a block form, the magnetic flux calculation unit, and the magnetomotive force calculation unit 19 and the magnetic flux calculation units 20A to 20C.
And constitutes the transformer magnetic flux state calculation unit 16.

Next, the operation will be described. Current detector 5
1 and the output of the voltage detector 6 are sent to the voltage / current control circuit 9. The set values of the output current reference 7 and the system voltage reference 8 are input to the voltage / current control circuit 9, and the AC side output voltage commands of the power converters 3A to 3C are calculated and output based on these values. . For this AC side output voltage command,
In order to prevent DC bias magnetization from occurring in the leg irons 201 to 203 of the iron core 200 of the multiple transformer 2, correction is performed as described below.

The primary current I1 and the secondary currents I2A to I2C are detected by the current detectors 51 and 52A to 52C as the amounts of electricity necessary to detect the magnetic flux state of the iron core 200 of the multiple transformer 2. The transformer magnetic flux state calculation unit 16 calculates the magnetic flux state of each step leg of the transformer 2 from the current detected by the current detectors 51, 52A to 52C. The detailed operation will be described with reference to FIGS. The magnetomotive forces F1 to F3 of the first to third stages of the transformer 2 are calculated by the following (Equation 4). Magnetomotive force = (number of turns of secondary winding x secondary current)-(number of turns of primary winding x primary current) (Equation 4) Generally, the relationship between magnetomotive force F and magnetic flux Φ is calculated by using magnetic resistance R F = R It can be written as × Φ (Equation 5). However, the magnetic resistance R is the magnetic path length L, the magnetic permeability μ,
It is represented by R = L / (μ × s) (Equation 6) using the magnetic circuit cross-sectional area s. In this embodiment, the magnetic permeability μ is constant for simplicity.

If the iron core 200 of the multiple transformer 2 is left-right symmetric in FIG. 1 and the magnetic circuit on one side of the iron core 200 is considered and leakage is neglected, it can be modeled as shown in FIG. In FIG. 3, the magnetic resistances R1 to R10 correspond to those in the iron core 200 of the multiple transformer 2. Regarding the relationship between the magnetomotive forces F1 to F3 and the magnetic fluxes Φ1 to Φ3 in each of the first to third stages, the magnetic resistances R1 to R10 in FIG. The following (formula 7) to (formula 9) are established. F1 = R1 × Φ1 ＋ R5 × Φ1 ＋ R2 × (Φ1-Φ2) ＋ R8 × Φ1 ・ ・ ・ (Formula 7) F2 = -R2 × (Φ1-Φ2) ＋ R6 × Φ2 + R3 × (Φ2-Φ3) + R9 × Φ2 ・ ・ ・ (Formula 7) 8) F3 = -R3 x (Φ2-Φ3) + R7 x Φ3 + R4 x Φ3 + R10 x Φ3 (Equation 9) From (Equation 7) to (Equation 9), the magnetic fluxes Φ1 to Φ3 of the leg irons 201 to 203 of each stage are calculated. Φ1 = K11 × F1 + K12 × F2 + K13 × F3 (Equation 10) Φ2 = K21 × F1 + K22 × F2 + K23 × F3 (Equation 11) Φ3 = K31 × F1 + K32 × F2 + K33 × F3 ・ ・ ・ ( Equation 12) is obtained. However, K11 to K33 are (formula 7) to (formula 9)
It is a coefficient when the magnetic fluxes Φ1 to Φ3 are obtained.

Therefore, since the magnetic flux state of the iron core of the transformer 2 can be correctly estimated by using (Equation 10) to (Equation 12), the primary current I1 and the secondary current I2A in FIG.
.About.I2C is multiplied by the number of turns N1 of the primary winding and the number of turns N2A to N2C of the secondary winding with a gain 171, gains 172A to 172C, and subtracted by subtracters 181A to 181C to obtain magnetomotive forces F1 to F3. 173AA to 173AC multiply the coefficients K1 to K13 in (Equation 10) and adder 131
A is added to obtain the magnetic flux Φ1, and the gains 173BA to 173BC are used to calculate the coefficients K21 to K23 in (Equation 11).
And the magnetic flux Φ2 is added by the adder 131B and the gain 173CA to the gain 173CC (Equation 12).
The magnetic flux Φ3 can be correctly obtained by multiplying the coefficients K31 to K33 in the inside and adding them by the adder 131C. These magnetic fluxes Φ1 to Φ3 are output as transformer magnetic flux states.

Here, the transformer magnetic flux state calculation unit 16 may have the configuration shown in FIG. 4 for the purpose of reducing the number of addition and subtraction calculations. FIG. 4 is a block diagram showing another configuration example of the transformer magnetic flux state calculation unit 16, in which 174AA to 17A are shown.
4AD, 174BA to 174BD, 174CA to 174
CD is a gain that is multiplied by a proportional constant, 132A to 132C is 4
It is an adder that performs an addition operation of inputs. Gain 174AA
~ 174AD, 174BA ~ 174BD, 174CA ~
174CD has a number of turns N1, N2A to N2C and a coefficient K11.
To K13, K21 to K23, and K31 to K33, the secondary currents I2A to I2C and the primary current I are obtained.
1 for gain 174AA to 174AD, 174BA to 17
4BD, multiply by 174CA to 174CD, adder 1
By adding 32A to 132C, the same calculation result as that of the transformer magnetic flux state calculation unit 16 shown in FIG. 2 can be obtained. By configuring the transformer magnetic flux state calculation unit 16 of FIG. 4, the transformer magnetic flux state calculation unit 16 illustrated in FIG.
Compared with, there is an advantage that the number of addition and subtraction calculations can be reduced and the calculation time can be shortened.

In this way, the transformer magnetic flux state calculation unit 16
Since it is possible to accurately calculate the magnetic flux states Φ1 to Φ3 of the iron cores of the respective stages, from the calculated magnetic flux states Φ1 to Φ3 of the iron core of the multiple transformer 2, the transformer bias magnetic amount calculating units 10A to 1 are calculated.
The bias amount is correctly calculated at 0C. Since the magnetic flux waveform when the iron core is saturated becomes positive and negative asymmetrical, the transformer bias magnetic amount calculation units 10A to 10C are means for extracting only the DC component corresponding to the bias amount from the AC signal including the DC component, for example,
It may be composed of a low-pass filter, an integrator, a moving average filter, or a means for extracting the asymmetry of the positive and negative waveforms, for example, a comparator of maximum and minimum values, an average value calculator of maximum and minimum values, Fourier It can be composed of an even harmonic extractor by series expansion.

The output voltage DC component correction control units 11A to 11C calculate correction values for the output voltage DC component based on the transformer bias magnetic components calculated by the transformer bias magnetic component calculation units 10A to 10C. The voltage command is corrected for the AC output voltage command.
The output voltage DC component correction control units 11A to 11C use the transformer bias amount calculated by the transformer bias amount calculation units 10A to 10C as the transformer bias amount feedback value and the command value of the transformer bias amount. Since the feedback control system can be regarded as zero, it can be configured by, for example, proportional control, integral control, differential control, or a combination thereof. The voltage command correction calculated by the output voltage DC component correction control units 11A to 11C is performed by the system voltage reference 8, the system voltage feedback value detected by the voltage detector 6, the output current reference 7, and the current detector 51. The AC side output voltage command calculated by the voltage / current control circuit 9 from the detected output current feedback value is added by the adders 13A to 13C, and is input to the PWM control circuits 14A to 14C.

In the PWM control circuits 14A to 14C, the GTO thyristors of the self-exciting power converters 3A to 3D, based on the voltage command values of the respective stages output from the adders 13A to 13C,
An ignition pulse width signal of a semiconductor switching device capable of self-extinguishing such as a GCT thyristor, an IGBT and a transistor is created. In the gate pulse amplification circuits 15A to 15D, P
The ignition pulse width signal generated by the WM control circuits 14A to 14C is amplified to generate a gate pulse drive signal for igniting and extinguishing the semiconductor switching elements of the self-excited converters 3A to 3D. This gate drive signal is a self-excited power converter 3A
To 3C, power converters 3A to 3C
Is a GTO thyristor, GC according to the voltage of the DC voltage source 4.
An adder 13A for igniting and extinguishing a self-extinguishing type semiconductor switching element such as a T thyristor, an IGBT, and a transistor
Generate a voltage corresponding to the output of ~ 13C.

As described above, in the power converter of the first embodiment shown in FIG. 1, the magnetomotive force generated by the windings of other stages is calculated by the calculation method shown in (Equation 10) to (Equation 12). By correctly calculating the transformer magnetic flux state in consideration of the influence, correct output voltage DC component correction control can be performed without miscalculating the transformer magnetic flux state. Even in a self-excited converter connected via a multi-transformer that has the effect of suppressing the DC bias magnetism of the multi-transformer, it is possible to avoid the protection stop due to overcurrent and to have a high reliability with high operation continuity. A power converter can be provided. Although the magnetic permeability is not actually constant and changes depending on the magnetic field, there is no problem even if the magnetic permeability is constant within the linear region of the excitation characteristic shown in FIG. By controlling as described above, since it is possible to suppress the bias in the initial stage of the bias where the amount of the bias is small and the change in the permeability is small, the permeability may be regarded as constant.

Embodiment 2. FIG. 5 is a block diagram showing the configuration of the transformer magnetic flux state calculation unit 16 in the power conversion device according to the second embodiment of the present invention, and FIG.
6 is a flowchart showing an operation procedure of a transformer magnetic flux state calculation unit 16 in the power conversion device according to the second embodiment of the present invention. In FIG. 5, 21AA to 21AC, 21BA to
21BC and 21CA to 21CC are variable gains whose gains change according to the magnetomotive forces F1 to F3.
A-21AC, 21BA-21BC, 21CA-21C
The magnetic flux calculation unit 20A includes the C and the adders 131A to 131C.
.About.20C, the transformer magnetic flux state calculation unit 16 is configured by the magnetomotive force calculation unit 19 and the magnetic flux calculation units 20A to 20C. The other parts except the transformer magnetic flux state calculation unit 16 are similar to those of the first embodiment shown in FIGS.

The difference from the first embodiment is that in the transformer magnetic flux state calculation unit 16, the constant K1 is calculated from the magnetomotive forces F1 to F3.
1 to K13 are multiplied and added by the adder 131A to obtain the magnetic flux Φ
1 is calculated and multiplied by constants K21 to K23 to adder 131B
To obtain the magnetic flux Φ2, and the constants K31 to K3
3, and the magnetic flux Φ3 is obtained by adding with the adder 131C, and at that time, the variable gains 21AA to 21AC, 21
It is a point to calculate the transformer magnetic flux state according to the non-linear relationship between the magnetic field of the transformer core and the magnetic flux using BA to 21BC and 21CA to 21CC.

In the background of this embodiment, the transformer core is shown in FIG.
As shown in FIG. 7, it has a non-linear characteristic between the exciting current and the magnetic flux as the quantity of electricity. FIG. 17 shows the relationship between the exciting current and the magnetic flux of the transformer core, but the relationship between the exciting current I and the magnetic field H is as follows (Formula 13) using the number of turns N and the magnetic path length L, and Since the relationship between the magnetic flux Φ and the magnetic field H is as shown in (Equation 14) using the magnetic permeability μ and the cross-sectional area s, the magnetic permeability μ of the transformer core also depends on the exciting current according to the nonlinear relationship between the exciting current and the magnetic flux. Change. H = N × I / L (Equation 13) Φ = s × μ × H (Equation 14) Therefore, the magnetic resistance R of the iron core changes from (Equation 5), and (Equation 10) to (Equation 10) Coefficients K11 to K13 and K21 to in Expression 12)
K23 and K31 to K33 may also change, and in particular in a region close to saturation, there may be a large difference in estimation of the transformer magnetic flux state. However, since the exciting current I and the magnetomotive force F are expressed by the following (Equation 15) using the number of turns N, the variable gains 21AA to 21AC, 21BA to 21BC, and 21C are shown.
A to 21 CC are provided to provide coefficients K11 to K13 and K21 to K
By making 23 and K31 to K33 variable according to the magnetomotive force F, it is possible to compensate for the non-linear relationship of the transformer core and prevent a deviation in the estimation of the transformer magnetic flux state. I = F / N (Equation 15)

Next, the operation will be described. In FIG. 5, the magnetomotive force calculator 19 calculates the magnetomotive forces F1 to F3.
Variable gains 21AA to 21AC, 21BA to 21BC,
In 21CA to 21CC, the magnetic permeability μ is calculated from the characteristics of the transformer core obtained by actual measurement or theoretical calculation and the magnetomotive forces F1 to F3 calculated by the magnetomotive force calculation unit 19.
The magnetic resistances R1 to R10 are calculated from the relationship shown in (Expression 6), and the magnetic fluxes Φ1 to Φ3 are calculated from (Expression 1) to (Expression 9).
0) to (Equation 12) in the process of obtaining coefficients K11 to K
13, K21 to K23, K31 to K33 can be obtained, and the coefficients K11 to K13 and K are calculated according to the result.
21 to K23 and K31 to K33 are changed. Magnetomotive force F
1 to F3 are variable gains 21AA as in the first embodiment.
.About.21AC to multiply the coefficients K11 to K13 and adder 13
The magnetic flux Φ1 and the variable gains 21BA to 21 are added by adding at 1A.
The coefficient K21 to K23 is multiplied by BC, the adder 131B adds the magnetic flux Φ2, the variable gains 21CA to 21CC multiply the coefficient K31 to K33, and the adder 131C adds the magnetic flux Φ3 to calculate the magnetic flux Φ3. Φ1 to Φ3 are output as transformer magnetic flux states.

FIG. 6 is a flowchart for explaining the operation procedure of the magnetomotive force calculating section 16 shown in FIG. Since the magnetic permeability μ of the transformer core depends on the magnetic flux, the magnetic fluxes Φ1 to Φ3 calculated by the method shown in FIG. 6 include an error.
Usually, the error is small, so Φ1 to Φ3 obtained in the previous calculation
Is used to calculate the magnetic permeability μ, and variable gains 21AA to 21
The coefficients K11 to K13, K21 to K23, and K31 to K33 of AC, 21BA to 21BC, and 21CA to 21CC may be obtained. Alternatively, in order to correct this error, Φ1
.About..PHI.3, the magnetic permeability .mu. Is recalculated, and steps 3 to 5 of FIG. 6 are repeated to converge the error, and then the variable gains 21AA to 21AC, 21BA to 21BC, 2
Coefficients K11 to K13 and K21 to K2 of 1CA to 21CC
3, K31 to K33 may be obtained.

In the transformer magnetic flux state calculator 16 shown in FIG. 5, the coefficients K11 to K13 are calculated from the magnetomotive forces F1 to F3.
Although K21 to K23 and K31 to K33 are obtained, the coefficients K11 to K13, K21 to K23, and K31 to K are calculated from the exciting current that is the difference between the primary current and the secondary current in consideration of the turns ratio.
You may ask for 33. Further, the magnetomotive forces F1 to F3 are obtained from the primary current I1 and the secondary currents I2A to I2C, and the variable gains 21AA to 21AC, 21BA to 21BC, 21C are obtained.
The magnetic flux densities Φ1 to Φ3 are obtained by using the A to 21 CC and the adders 131A to 131C, but the primary current I1 and the secondary current I2 are similar to the transformer magnetic flux state calculation unit 16 shown in FIG.
The same effect can be obtained by using a method of multiplying A to I2C by a coefficient and adding them. Further, in this embodiment, the variable gain is used to compensate for the non-linear relationship of the transformer core. However, the non-linear relationship between the magnetomotive forces F1 to F3 and the magnetic resistances R1 to R10 or the magnetic fluxes Φ1 to Φ3, or the excitation The same effect can be obtained by a method in which a non-linear relationship between the current and the magnetic resistances R1 to R10 or the magnetic fluxes Φ1 to Φ3 is obtained in advance and tabulated and appropriately read.

As described above, the power converter of the second embodiment provided with the transformer magnetic flux calculating means 16 shown in FIG.
Coefficients K11 to K13, K21 according to the magnetomotive forces F1 to F3
~ K23, K31 to K33 variable gain 21AA
-21AC, 21BA-21BC, 21CA-21CC
By providing, the coefficient K11 to K13, in accordance with the magnetomotive forces F1 to F3, even in the region where the transformer core has a large non-linearity,
By making K21 to K23 and K31 to K33 variable, a correct transformer magnetic flux state can be calculated. Even in such an area, it is possible to perform correct output voltage DC component correction control by correctly calculating the transformer magnetic flux state according to the effect of the magnetomotive force generated by the windings of other stages, which is difficult to achieve in the past. It is possible to obtain the effect of suppressing the DC bias magnetism of the multiple transformer having the common magnetic path, and it is possible to provide a highly reliable power conversion device with high continuity of operation while avoiding protection stop due to overcurrent.

Embodiment 3. 7 is a block diagram showing a configuration of a power conversion device according to a third embodiment of the present invention, and FIG. 8 is a block diagram showing a configuration of another stage generated magnetomotive force effect correction section 22 of the power conversion device shown in FIG. is there. In FIG. 7, reference numerals 182A to 182C are subtractors for obtaining an exciting current by taking a difference in consideration of the winding ratio of the secondary current and the primary current detected by the current detectors 52A to 52C and 51, and 22 is an output voltage DC component. Correction control unit 11A-
11C is another stage generated magnetomotive force influence correction unit as a correction value correction unit that corrects the output of 11C according to the magnetomotive force generated in another stage. In FIG. 8, 175AA to 175AC, 175
BA to 175BC, 175CA to 175CC use the outputs of the output voltage DC component correction control units 11A to 11C as inputs, and multiply the coefficients K14 to K16, K24 to K26, and K34 to K36, and 132A to 132C are gains 175A.
A ~ 175AC, 175BA ~ 175BC, 175CA
It is an adder that adds the outputs of ˜175 CC and corrects the effect of the magnetomotive force generated in the other stage. 23A to 23C have a gain of 175AA
~ 175AC, adder 132A, gain 175BA ~ 1
75BC, adder 132B, gain 175CA to 175
A second stage generated magnetomotive force effect correction calculation unit in which the CC and the adder 132C are blocked, and the other stage generated magnetomotive force effect correction unit 22 is provided.
Are configured. The transformer bias magnetic amount calculation units 10A to 10C and the output voltage DC component correction control units 11A to 11C constitute a temporary voltage command correction unit, and the temporary voltage command correction unit and the other stage generated magnetomotive force influence correction unit 22 generate a voltage. It constitutes a command correction means. 1 and 2 except the subtracters 182A to 182C and the other-stage generated magnetomotive force effect correction unit 22.
The description is omitted because it is the same as the first embodiment shown in FIG.

Next, the operation will be described. Embodiment 1
The difference is that the transformer magnetic flux state calculation unit 16 does not calculate the transformer magnetic flux state in consideration of the influence of the magnetomotive force generated by the windings of other stages, but the other stage generated magnetomotive force correction unit 22 The same effect was obtained by correcting the effect of the magnetomotive force generated by the windings of the other stages. In FIG. 7, the current detector 52A
The secondary current of the multi-transformer 2 detected by -52C is subtracted by the subtracters 182A-182C in consideration of the primary current detected by the current detector 51 and the turns ratio, and the exciting current is calculated. The exciting current calculated by the subtracters 182A to 182C is applied to the leg iron 20 by the transformer bias magnetic amount calculating units 10A to 10C.
The transformer magnetic bias amount is calculated individually for each of 1 to 203, and the output voltage DC component correction value is calculated by the output voltage DC component correction control units 11A to 11C, and is output as the provisional voltage command correction. Although the effect of the magnetomotive force generated by the other-stage winding shown in (Expression 10) to (Expression 12) is not corrected in this temporary voltage command correction, the other-stage generated magnetomotive force effect correction unit 22 does The influence of the magnetomotive force generated by the winding is corrected, and the adders 13A to 13C are added.
The error due to the influence of the magnetomotive force generated in the other-stage winding can be eliminated by outputting the final voltage command correction to the.

Here, the operation of the other stage generated magnetomotive force effect correction unit 22 will be described with reference to FIG. In the transformer magnetic flux calculating means 16 in which the influence of the magnetomotive force generated in the winding of the other stage is corrected in the first embodiment, the magnetomotive force F is obtained from the secondary current and the primary current, and the magnetic flux calculating units 20A to 20C. By calculating the magnetic fluxes Φ1 to Φ3, the error due to the effect of the magnetomotive force generated by the winding of the other stage is eliminated, but in the present embodiment, the subtractor 1
By obtaining the exciting current at 82A to 182C, a signal proportional to the magnetomotive force at each stage can be obtained, so that the transformer bias magnetic amount calculation units 10A to 10C and the output voltage DC component correction control units 11A to 11C are used. The output signal is also a signal proportional to the magnetomotive force of each stage. Therefore, the first embodiment
Coefficients K11 to K13, K21 to K23, K31 to K33 multiplied by the magnetic flux calculators 20A to 20C used in Step S11 or coefficients K11 to K in consideration of the relationship between the exciting current and the magnetomotive force.
Coefficients K14 to K16, K24 to K26, K34 to K obtained by multiplying 13, K21 to K23, K31 to K33 by the number of turns N
36 to gain 175AA to gain 175AC, 175B
Multiply by A ~ 175BC, 175CA ~ 175CC,
The same effect as that of the first embodiment can be obtained by adding in the adders 132A to 132C.

Incidentally, the transformer bias magnetic amount calculation units 10A to 10C.
And the output voltage DC component correction control units 11A to 11C both have signals proportional to the magnetomotive force of each stage. Therefore, the other stage generated magnetomotive force effect compensation unit 22 uses the transformer bias magnetic amount calculation unit 10A. -10C and output voltage DC component correction controller 11A
11C, or between the adders 13A to 13C and the PWM control circuits 14A to 14C, the same effect can be obtained.

As described above, in the power converter of the third embodiment shown in FIG. 7, the magnetomotive force generated by the windings of other stages is calculated by the calculation method shown in (Equation 10) to (Equation 12). By correcting the influence, correct output voltage DC component correction control can be performed according to the transformer magnetic flux state, so self-excited type connected via a multiple transformer with a common magnetic path, which was difficult in the past. Also in the converter, the effect of suppressing the DC bias magnetism of the multiple transformer can be obtained, and it is possible to provide a highly reliable power conversion device with high operation continuity while avoiding protection stop due to overcurrent.

Fourth Embodiment 9 is a block diagram showing a configuration of a power conversion device according to a fourth embodiment of the present invention, and FIG. 10 is a block diagram showing a configuration of another stage generated magnetomotive force effect correction section 22 of the power conversion device shown in FIG. is there. In FIG. 9, the input of the other stage generated magnetomotive force effect correction unit 22 is the excitation current and output voltage DC component correction control unit 11A to which the subtractors 182A to 182C obtain.
This is the output of 11C. In FIG. 10, 21AA-21
AC, 21BA to 21BC, 21CA to 21CC subtract the output of the output voltage DC component correction control units 11A to 11C from the subtracter 1
The coefficients K14 to K16 according to the outputs of 82A to 182C,
Variable gains for multiplying K24 to K26 and K34 to K36 as variable coefficients, 132A to 132C are variable gains 21
AA-21AC, 21BA-21BC, 21CA-21
It is an adder that adds the output of CC and outputs the voltage command correction in which the effect of the magnetomotive force generated in the other stage is corrected. The other components except the other-stage generated magnetomotive force effect correction unit 22 are the same as those in the third embodiment shown in FIGS.

Next, the operation will be described. Embodiment 3
Is different from the gain 175 in the magnetomotive force correction unit 22 generated in the other stage.
AA ~ 175AC, 175BA ~ 175BC, 175C
A to 175CC are variable gains 21AA to 21AC, 21
BA to 21BC and 21CA to 21CC are set, and it is the point that the nonlinear relation of the transformer core is compensated so as to be folded as in the second embodiment. In FIG. 9, as in the third embodiment,
The outputs of the output voltage DC component correction control units 11A to 11C are input to the other-stage generated magnetomotive force effect correction unit 22, but in FIG. 10, variable gains 21AA to 21 that multiply this input by a factor.
The coefficients K14 to K16, K24 to K26, and K34 to K36 of AC, 21BA to 21BC, and 21CA to 21CC are variable according to the outputs of the subtracters 182A to 182C. By obtaining the exciting current by the subtractors 182A to 182C. Since a signal proportional to the magnetomotive force of each stage is obtained, the signals output through the transformer bias magnetic amount calculation units 10A to 10C and the output voltage DC component correction control units 11A to 11C are also proportional to the magnetomotive force. And the coefficients K11 to K13, K21 to K23, and K31 by the same method as in the second embodiment.
.About.K33 or coefficients K14 to K16, K24 to K2 obtained by multiplying the coefficients K11 to K13, K21 to K23, K31 to K33 by the number of turns N in consideration of the relationship between the exciting current and the magnetomotive force.
6, K34 to K36 are variable, and adders 132A to 132A
The same effect as in the second embodiment can be obtained by adding 2C.

Since the outputs of the transformer bias magnetic amount calculation units 10A to 10C and the output voltage DC component correction control units 11A to 11C are signals proportional to the magnetomotive force of each stage,
As in the third embodiment, the other stage generated magnetomotive force effect correction unit 22 is
The adders 13A to 13 are also provided between the transformer bias magnetic amount calculation units 10A to 10C and the output voltage DC component correction control units 11A to 11C.
Even if it is provided between C and the PWM control circuits 14A to 14C, the same effect can be obtained. Further, as in the second embodiment, instead of using the variable gain, the magnetomotive forces F1 to F3 and the magnetic resistances R1 to R10 or the magnetic fluxes Φ1 to Φ3 are in a non-linear relationship, or the exciting current and the magnetic resistances R1 to R10 or the magnetic fluxes Φ1 to Φ1. The same effect can be obtained by a method in which a non-linear relationship of Φ3 is obtained in advance and tabulated and read out appropriately.

As described above, the power converter of the fourth embodiment shown in FIG. 9 has the coefficients K14 to K depending on the exciting current.
Variable gains 21AA to 21AC, 21BA to 21BC, and 21C are obtained by calculating 16, K24 to K26, and K34 to K36.
By providing A to 21 CC, the coefficients K14 to K1 can be adjusted according to the exciting current even in the region where the transformer core has a large non-linearity.
By making 6, K24 to K26 and K34 to K36 variable, the correct transformer magnetic flux state can be calculated. Even in such an area, it is possible to perform correct output voltage DC component correction control by correctly calculating the transformer magnetic flux state according to the effect of the magnetomotive force generated by the windings of other stages, which is difficult to achieve in the past. It is possible to obtain the effect of suppressing the DC bias magnetism of the multiple transformer having the common magnetic path, and it is possible to provide a highly reliable power conversion device with high continuity of operation while avoiding protection stop due to overcurrent.

Fifth Embodiment 11 is a block diagram showing a configuration of a power conversion device according to a fifth embodiment of the present invention, and FIG. 12 is a block showing a configuration of transformer magnetic flux state calculation unit 16 of the power conversion device according to the fifth embodiment shown in FIG. It is a figure. In FIG. 11, 24A to
24C is leg irons 201 to 20 of the iron core 200 of the multiple transformer 2.
A search coil provided in the vicinity of 3 for detecting a magnetic field as a magnetic detector, and a transformer magnetic flux state calculation unit 16 for calculating a transformer magnetic flux state from the outputs of the search coils 24A to 24C. Further, the transformer bias magnetic amount calculation units 10A to 10C and the output voltage DC component correction control units 11A to 11C constitute a correction value calculation means.

In FIG. 12, 25A to 25C are integrators for integrating the outputs of the search coils 24A to 24C to obtain a magnetic field, and 26 is a magnetic field calculation unit in which the integrators 25A to 25C are blocked. 176AA-176AC, 176BA
˜176BC, 176CA to 176CC are gains that are multiplied by a proportional constant, 133A to 133C are adders that perform addition operation of three inputs, and 20A to 20C are gains 176AA to 176.
AC, 176BA to 176BC, 176CA to 176C
This is a magnetic flux calculation unit configured by blocking C and the adders 133A to 133C, and the magnetic field calculation unit 26 and the magnetic flux calculation units 20A to 20C configure the transformer magnetic flux state calculation unit 16. Other parts are the same as the first embodiment shown in FIGS. 1 and 2.
The description is omitted because it is the same as the case.

Next, the operation will be described. In FIG. 11, the output voltage V of the search coils 24A to 24C is the magnetic field H of the space where the search coils 24A to 24C are installed.
_Since it is the time derivative of _S , the number of turns of the search coil is N _S ,
Assuming that the cross-sectional area of the search coil is S _S , the magnetic permeability of the search coil is μ _S , and the time is t, V = (N _S × S _S × μ _S ) dH _S / dt (Expression 16) To be done. Therefore, as shown in FIG. 12, in the transformer magnetic flux density state calculation unit 16, by integrating the outputs of the search coils 24A to 24C by the integrators 25A to 25C, the search coils 24A to 24C are calculated by the following (Equation 17). The magnetic field H _{S of the} space where is installed is obtained.

[0046]

[Equation 1]

In equation (17), H0 is the initial value of integration and represents the initial magnetic field. The initial value H0 may be 0 when there is no residual magnetic flux at startup. If the residual magnetic flux exists, the initial value cannot be known unless the residual magnetic flux is directly detected.
Further, when the offset remains due to the start phase, the initial magnetic field H0 cannot be distinguished from the offset, but the initial value H0
Even if the initial value H0 is unknown, it is possible to determine the bias by a method such as a method of determining the even harmonics included in the magnetic field H _{S in} the space obtained by (Equation 17) and determining the progress of the bias from the increase or decrease thereof. As a method of removing the offset, there are a method of removing with a low-pass filter, and a method of determining and removing the offset amount by monitoring the magnetic field H _{S of} the space obtained from (Equation 17) for a certain period of time. Alternatively, if the positive and negative maximums in the instantaneous waveform of the magnetic field H _{S in} the space obtained from (Equation 17) are compared, it is not necessary to consider the offset.

Since the search coils 24A to 24C are installed near the leg irons 201 to 203, that is, near the windings of each stage, the magnetic field around the search coils 24A to 24C is the same as the magnetic field generated by the windings of each stage. However, since the primary winding and the secondary winding are densely installed, a combined magnetic field of the magnetic field generated by the primary winding and the magnetic field generated by the secondary winding, that is, an exciting magnetic field, becomes the search coils 24A to 24C. Integrators 25A-2
An exciting magnetic field can be obtained by integrating at 5C. As described above, the search coil operates as a magnetic detector. As can be seen from (Equation 13) and (Equation 15), the exciting magnetic field obtained by the integrators 25A to 25C is (Equation 1
0) to (Equation 12), since the amount is proportional to the magnetomotive forces F1 to F3, constants K11 to K13 are calculated in the same manner as in the first embodiment.
K21 to K23 and K31 to K33 can be obtained, and gains of 176AA to 176AC and 176BA to 176B are obtained.
C, 176CA to 176CC, coefficients K11 to K13,
K21 to K23 and K31 to K33 are multiplied to adder 133A
The magnetic fluxes Φ1 to Φ3 can be obtained by adding at ˜133C. The magnetic flux states Φ1 to Φ3 of the transformer 2 obtained in this way are calculated from the transformer bias magnetic amount calculation units 10A to 10
10C, and operates similarly to the first embodiment.

As described above, the power converter of the fifth embodiment shown in FIG. 11 uses the calculation method shown in (Equation 10) to (Equation 12) to calculate the exciting magnetic field generated by the windings of other stages. By correctly calculating the transformer magnetic flux state in consideration of the influence, correct output voltage DC component correction control can be performed without miscalculating the transformer magnetic flux state. Even in a self-excited converter connected via a multi-transformer that has the effect of suppressing the DC bias magnetism of the multi-transformer, it is possible to avoid the protection stop due to overcurrent and to have a high reliability with high operation continuity. A power converter can be provided.

Sixth Embodiment FIG. 13 is a block diagram showing the configuration of the transformer magnetic flux state calculation unit 16 according to the sixth embodiment of the present invention. In FIG.
21AA-21AC, 21BA-21BC, 21CA-
21 CC is a coefficient K17 to K19, K27 to K29, K37 to K3 for the output of the integrators 25A to 25C according to the input value.
9 is a variable gain that is multiplied by a variable coefficient, and is variable gains 21AA to 21AC, 21BA to 21BC, and 21C.
Variable gain 2 that constitutes magnetic flux calculation units 133A to 133C with A to 21CC and adders 133A to 133C.
1AA to 21AC, 21BA to 21BC, 21CA to 2
The other components except 1CC are the same as those in the fifth embodiment, and the description thereof will be omitted.

Next, the operation will be described. Embodiment 5
Is different from the gain 176 in the transformer magnetic flux state calculation unit 16
AA ~ 176AC, 176BA ~ 176BC, 176C
A to 176CC are variable gains 21AA to 21AC, 21
BA to 21BC and 21CA to 21CC are set, and the non-linear relation of the transformer core is compensated as in the second and fourth embodiments. In FIG. 13, the outputs of the search coils 24A to 24C are input to the transformer magnetic flux state calculation unit 16 as in the case of the fifth embodiment.
By integrating the output of 4C by the integrators 25A to 25C, the exciting magnetic field at each stage can be obtained.
Since the exciting magnetic field obtained at ˜25 C is an amount proportional to the magnetomotive force as in the fifth embodiment, the coefficients K11 to K13, K21 to K23, K31 to K33 are the same as in the second embodiment.
Corresponding to K17 to K19, K27 to K29, K3
7 to K39 with variable gains 21AA to 21AC, 21BA
~ 21BC, 21CA ~ 21CC multiply, adder 1
It is possible to obtain the same effect as that of the second embodiment by adding magnetic fluxes Φ1 to Φ3 by adding 33A to 133C.

As described above, the power converter of the sixth embodiment has the coefficients K17 to K19 and K27 depending on the exciting magnetic field.
Variable gain 21 for varying K29 to K37 to K39
AA-21AC, 21BA-21BC, 21CA-21
By providing CC, the coefficients K17 to K19 and K2 can be adjusted according to the exciting current even in the region where the transformer core has a large non-linearity.
By making 7 to K29 and K37 to K39 variable, the correct transformer magnetic flux state can be calculated. Even in such an area, it is possible to perform correct output voltage DC component correction control by correctly calculating the transformer magnetic flux state according to the effect of the magnetomotive force generated by the windings of other stages, which is difficult to achieve in the past. It is possible to obtain the effect of suppressing the DC bias magnetism of the multiple transformer having the common magnetic path, and it is possible to provide a highly reliable power conversion device with high continuity of operation while avoiding protection stop due to overcurrent.

Embodiment 7. 14 is a block diagram showing a configuration of a power conversion device according to a seventh embodiment of the present invention, and FIG. 15 is a block showing a configuration of transformer magnetic flux state calculation unit 16 of the power conversion device according to the seventh embodiment shown in FIG. It is a figure. In FIG. 15, reference numerals 25A to 25C denote integrators, which form the transformer magnetic flux state calculation unit 16. The difference between the fifth embodiment and the fifth embodiment is that the search coils 24A to 24C are not located near the transformer windings on the leg irons 201 to 203 but are directly affected by the magnetic field generated by the windings 204. The configuration is different from that of the transformer magnetic flux state calculation unit 16 in the vicinity of the above, and the other portions are the same as those of the fifth embodiment, and therefore the description thereof will be omitted.

Next, the operation will be described. It has been described in the fifth embodiment that the magnetic fields at the search coil installation points can be detected by the search coils 24A to 24C, but FIG.
When installed on the side of the yoke 204 outside the transformer core that is not directly affected by the magnetic field generated by the winding as shown in, the magnetic field at the installation point is the magnetic field generated by the magnetic flux of the yoke 204 of the transformer. , Which is an amount proportional to the magnetic flux of the yoke 204. Here, considering the magnetic flux of the yoke 204 of the transformer with reference to FIG. 3, when the influence of leakage is ignored, the magnetic flux path is closed in the transformer core, so the magnetic flux of the yoke 204 is Φ1.
.Apprxeq..PHI.3, which coincides with the magnetic fluxes .PHI.1 to .PHI.3 of the leg irons 201 to 203 wound with the respective windings. Therefore, by detecting the magnetic field near the side surface of the yoke outside the transformer core, the same amount as the magnetic flux states Φ1 to Φ3 of the leg iron of each stage can be detected. It is not necessary to correct the influence of the magnetomotive force generated in the other stage calculated by Therefore, FIG.
As shown in FIG. 4, the magnetic flux states Φ1 to Φ3 of the iron core of the transformer 2 can be calculated by integrating the signals detected by the magnetic field monitoring windings 24A to 24C by the integrators 25A to 25C. By setting the outputs of the transformers 25A to 25C as the inputs of the respective transformer bias magnetic field amount calculating units 10A to 10C, the same effect as that of the first embodiment can be obtained.

As described above, the power converter of the seventh embodiment shown in FIG. 14 uses the search coil in the vicinity of the yoke, which is not directly affected by the magnetic field generated by the windings, to correctly perform the transformer magnetic flux state. By calculating, the correct output voltage DC component correction control can be performed without erroneously calculating the transformer magnetic flux state, so it was connected via a multiple transformer with a common magnetic path, which was difficult in the past. Even in the self-excited converter, the effect of suppressing the DC bias magnetism of the multiple transformer can be obtained, and it is possible to provide a highly reliable power conversion device with high continuity of operation while avoiding protection stop due to overcurrent.

Although the search coil is used as the magnetic detector in the fifth to seventh embodiments, any detector capable of detecting the magnetic amount such as a Hall element may be used. Further, although the output of the search coil is integrated to obtain the magnetic field, the output of the search coil may be directly used when the influence of noise is small. In this case, the noise component is not attenuated, but the attenuation of the even-numbered harmonic component can be prevented when the bias amount is calculated by extracting the even-numbered harmonic component. Further, although the outer iron type three-stage multiple transformer has been described as an example in the first to seventh embodiments, a transformer having a common magnetic path between stages and having magnetic interaction is shown in FIG. A transformer of other configuration such as the inner iron type multi-transformer as shown may be used, and if the same method as that for obtaining (Equation 10) to (Equation 12) is used, two-stage or four-stage or more multi-transformer may be used. It is also applicable to vessels. Further, the magnetic resistance R is not limited to the iron core, and when a structure having a gap is adopted, the magnetic resistance of air or the magnetic resistance of the insulating material is obtained from (Equation 6) as the magnetic resistance R, and the magnetic resistance R is connected in series or according to the configuration. It can be applied to any transformer with an iron core structure by calculating the magnetic resistance by the method of connecting in parallel.

Further, the transformer magnetic flux state calculation unit 16 shown in the first to seventh embodiments is replaced by a microcomputer,
It may be realized by using a computing means such as a DSP. Further, in the first to fourth embodiments, the current is detected as the electric quantity of the transformer, but the same effect can be obtained by other electric quantities such as voltage. Further, in the first to seventh embodiments, a single-phase circuit has been described for simplification of description, but a multi-phase configuration of two or more phases is also possible, and in the case of a three-phase configuration, the transformer wiring is The same effect can be obtained by using the same method for star / star connection, star / delta connection, and delta / delta connection. In addition, the first to the first embodiments
Although the primary windings of the transformer are connected in series in the seventh embodiment, the transformer windings may be wired in parallel, and the same effect can be obtained by using the same method.

Further, in the first to seventh embodiments,
As a converter that generates a voltage according to the output of the adder 13A, a self-exciting converter that uses a semiconductor switching element such as GTO, GCT, IGBT, and transistor that can self-extinguish has been described. Any converter such as a thyristor converter or a cycloconverter may be used as long as it is a converter that generates a signal. Further, instead of the variable gain in the second, fourth, and sixth embodiments, a storage unit such as a memory table prepared in advance is used to output a necessary value according to the input. I don't mind.

[0059]

The power conversion according to the present invention is configured as described above, and is corrected according to the magnetic flux states of the plurality of leg bars of the multiple transformer by the voltage command correction calculation means, and the AC output voltage of the power converter is corrected. Since it is possible to correct the output voltage DC component correction control by compensating the influence of the magnetomotive force generated in the windings of other stages without miscalculating the transformer magnetic flux state, which was difficult in the past. The effect of suppressing the DC bias magnetization of the multiple transformer having the common magnetic path can be obtained, and it is possible to provide a highly reliable power converter with high continuity of operation while avoiding protection stop due to overcurrent.

[Brief description of drawings]

FIG. 1 is a block diagram showing a configuration of a power conversion device according to a first embodiment of the present invention.

FIG. 2 is a block diagram showing a configuration of a transformer magnetic flux state calculation unit of the power conversion device according to the first embodiment of the present invention.

FIG. 3 is an explanatory diagram showing a model of a magnetic circuit of a multiple transformer of the power conversion device according to the first embodiment of the present invention.

FIG. 4 is a block diagram showing another configuration of the transformer magnetic flux state calculation unit of the power conversion device according to the first embodiment of the present invention.

FIG. 5 is a block diagram showing a configuration of a transformer magnetic flux state calculation unit of the power conversion device according to the second embodiment of the present invention.

FIG. 6 is a flowchart showing an operation algorithm of a transformer magnetic flux state calculation unit of the power conversion device according to the second embodiment of the present invention.

FIG. 7 is a block diagram showing a configuration of a power conversion device according to a third embodiment of the present invention.

FIG. 8 is a block diagram showing a configuration of another stage generated magnetomotive force effect correction unit of the power conversion device according to the third embodiment of the present invention.

FIG. 9 is a block diagram showing a configuration of a power conversion device according to a fourth embodiment of the present invention.

FIG. 10 is a block diagram showing a configuration of another stage generated magnetomotive force effect correction unit of the power conversion device according to the fourth embodiment of the present invention.

FIG. 11 is a block diagram showing a configuration of a power conversion device according to a fifth embodiment of the present invention.

FIG. 12 is a block diagram showing a configuration of a transformer magnetic flux state calculation unit of a power conversion device according to a fifth embodiment of the present invention.

FIG. 13 is a block diagram showing a configuration of a transformer magnetic flux state calculation unit of a power conversion device according to a sixth embodiment of the present invention.

FIG. 14 is a block diagram showing a configuration of a power conversion device according to a seventh embodiment of the present invention.

FIG. 15 is a block diagram showing a configuration of a transformer magnetic flux state calculation unit of a power conversion device according to a seventh embodiment of the present invention.

FIG. 16 is a block diagram showing a configuration of a conventional power conversion device.

FIG. 17 is a characteristic diagram showing a relationship between an exciting current and a magnetic flux of a transformer core.

FIG. 18 is a perspective view showing an outer iron type multiple transformer.

FIG. 19 is a perspective view showing an inner iron type multiple transformer.

[Explanation of symbols]

1 AC power system, 2 multiple transformers, 3A to 3C power converter, 9 voltage and current control circuit, 10A to 10C transformer bias magnetic amount calculation unit, 11A to 11C output voltage DC component correction control unit, 16 transformer magnetic flux state Calculation unit, 22 Other stage generated magnetomotive force effect correction unit, 24A to 24C Search coil, 2
00 iron core, 201-203 leg iron, 204 yoke iron, 2
11-213 DC side winding, 221-223 AC side winding.

─────────────────────────────────────────────────── ─── Continuation of the front page (51) Int.Cl. ⁷ Identification code FI H02M 7/219 H02M 7/219 (56) Reference JP-A-10-56739 (JP, A) (58) Fields investigated (Int .Cl. ⁷ , DB name) H02M 7/48 H01F 19/00 H01F 27/42 H02J 3/38 H02M 7/219

Claims (10)

(57) [Claims] 1. A plurality of power converters each having a switching element, an iron core having a plurality of leg irons magnetically coupled to each other, and an AC core of each of the plurality of power converters wound around the leg irons. A multiple transformer having a plurality of connected DC side windings and a plurality of AC side windings respectively wound on the leg irons and connected to an AC power source, and a voltage for calculating an AC side output voltage command of the power converter Command output means, and a voltage command correction calculation means for calculating a voltage command correction for correcting the AC side output voltage command according to the magnetic flux states of a plurality of leg irons of the multiple transformer, the output voltage command And an AC side output voltage of the power converter is controlled based on the voltage command correction. 2. The voltage command correction calculation means temporarily calculates the temporary voltage command correction for each leg iron from the electric quantities of the DC side winding and the AC side winding wound on the leg iron of the multiple transformer. Voltage command correction calculation means and correction value correction for calculating voltage command correction using the relationship between the amount of electricity and the magnetic flux state of the DC side winding and the AC side winding between the leg irons from the temporary voltage command correction The power conversion device according to claim 1, comprising a means. 3. The power converter according to claim 2, wherein the relationship between the quantity of electricity and the magnetic flux state is a relationship expressed by using a magnetic resistance. 4. The voltage command correction calculation means is a multi-transformer.
From the electric quantities of multiple DC side windings and AC side windings,
A magnetic flux state calculating means for calculating the magnetic flux state of the leg iron and this magnetic
From the magnetic flux state calculated by the flux state calculation means,
Correction value calculating means for calculating the correction
The state calculation means is a non-linear device that changes according to the amount of electricity.
That the magnetic flux state is calculated as a characteristic
The power converter according to claim 1, wherein the power converter is a power converter. 5. The power conversion according to claim 2, wherein the voltage command correction is calculated as a non-linear characteristic that changes the relationship between the electric quantity and the magnetic flux state according to the input electric quantity. apparatus. 6. A plurality of magnetic detectors are provided in the vicinity of the iron core of the multiple transformer, and the voltage command correction calculating means calculates the voltage command correction from the outputs of these magnetic detectors. The power converter according to claim 1. 7. The voltage command correction calculation means calculates the magnetic flux status calculation means for calculating the magnetic flux status of the plurality of leg irons from the outputs of the plurality of magnetic detectors, and the voltage command correction calculation from the output of the magnetic flux status calculation means. 7. The power conversion device according to claim 6, comprising a correction value calculation means. 8. The magnetic flux state calculation means comprises a plurality of magnetic detectors.
Output and the magnetic flux state of multiple leg irons, using reluctance
From the output of the magnetic detector,
The magnetic flux state is calculated so that it can be calculated.
7. The power conversion device according to 7 . 9. The magnetic flux state calculation means comprises a plurality of magnetic detectors.
Of the magnetic field of multiple leg irons,
As a non-linear characteristic that changes according to the output of the air detector
A contract characterized in that the magnetic flux state is calculated.
The power conversion device according to claim 7 . 10. The magnetic resistance changes when the magnetomotive force changes.
The power conversion device according to claim 3 or 8, wherein