JP3435392B2 - Wireless communication method and wireless communication device - Google Patents

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a wireless communication method and a wireless communication device under a fading environment. For example, TDMA (Time Division Multiple Acces
In the digital mobile communication system for public service of the s) system, multi-valued QAM (Quadrature Amplitude Modulation:
The present invention relates to a fading distortion compensation technique when performing modulation by a quadrature amplitude modulation method.

[0002]

2. Description of the Related Art In mobile radio communication, the amplitude and phase of a received signal fluctuate due to fading. “Pilot signal insertion method” is a technique for compensating for fluctuations due to this fading. Seiichi Sampei, “16QAM for land mobile communication”
"Fading distortion compensation method" of the Institute of Electronics, Information and Communication Engineers, B-II, Vol.J72-B-II, No.1, pp.7-15, 1989-1. This method is known. A pilot symbol having a value of is inserted into the information symbol section and transmitted, and the receiving side estimates the fading fluctuation at the information symbol position from the received signal of the pilot symbol to perform fading distortion compensation.

FIG. 6 is a block diagram of a conventional radio communication apparatus using the "pilot symbol insertion method". 6A is a block diagram of the transmitter side, and FIG. 6B is a block diagram of the receiver side. In FIG. 6A, 1 is transmission data,
A serial-parallel converter (S / P) 2 converts the transmission data into parallel data every 4 bits. A baseband signal generator (BSG) 3 converts 4-bit parallel data into a baseband signal corresponding to one symbol of 16QAM modulation. A frame signal generation unit 41 periodically inserts pilot symbols at equal intervals in the information symbol section. As the pilot symbols, among the four symbols having the maximum amplitude in the 16QAM signal space diagram, the symbols are appropriately switched and used. A low-pass filter (LPF) 5 limits the band of the baseband signal. 6 is a quadrature modulator, and 7 is a local oscillator. The reference frequency signal and the quadrature reference frequency signal output from the local oscillator 7 are 16QAM-modulated with the band-limited baseband signal. Reference numeral 8 is an amplifier, and 9 is a transmission antenna, which amplifies and transmits the modulated signal.

On the other hand, in FIG. 6B, 11 is a receiving antenna, and 12 is a bandpass filter (BPF), which band-limits the received signal in order to normally operate the AGC 13 and AFC 14 which will be described later. An automatic gain controller (AGC) 13 keeps the received signal level constant.
An automatic frequency controller (AFC) 14 roughly adjusts the frequency offset between the transmitter side and the receiver side. 1
5 is a quadrature demodulator, 16 is a local oscillator, and 17 is a low-pass filter (LPF). 16QA is obtained by multiplying the received signal whose frequency offset is roughly adjusted by the reference frequency signal and the quadrature reference frequency signal output from the local oscillator 16.
The quasi-coherent detection of M is performed and the band is limited by the LPF 17 to output a reception signal (baseband signal of two channels of I phase and Q phase). The frequency of the reference frequency signal is orthogonally demodulated in a state where it does not completely match the frequency of the reference frequency signal of FIG.

Reference numeral 18 denotes a fading distortion estimation / compensation unit, which uses a pilot symbol to perform fading distortion estimation and fading distortion compensation, which will be described later with reference to FIG. 8, and also finely adjust the offset frequency. A symbol determination unit 19 outputs 4-bit output data 23 per symbol by determining the received signal in which the fading distortion is compensated for at the symbol timing.

FIG. 7 is an explanatory diagram showing a frame structure of transmission data in the "pilot symbol insertion method".
In the illustrated example, a known 1-symbol pilot symbol is periodically inserted for each (N-1) symbol of the information symbol portion. One pilot symbol and the (N-1) information symbol portion form one frame. On the receiver side, amplitude / phase fluctuation compensation is performed with reference to this pilot symbol. The received signal of the pilot symbol is used to estimate the fading fluctuation which fluctuates with time. Next, for information symbols between pilot symbols, fading fluctuations are estimated using an interpolation method.
A Gaussian interpolation formula or the like is generally used for the interpolation method.

FIG. 8 is a block diagram for explaining the fading distortion estimation / compensation unit 18 shown in FIG. 6 (b). In the figure, 18a is a fading distortion estimator, and 18b is a fading distortion compensator. A sampling switch 51 samples the reception signal U (t) from the LPF 17 in FIG. 6B. A frame synchronization unit 52 reproduces the frame timing of the cycle T _f from the received signal U (t). Sampling switch 51
Is extracted by the frame synchronization unit 52 and outputs only the pilot symbols in the input signal U (t) to the delay unit 54. Reference numeral 53 is a clock recovery unit, which is an input signal U
From (t), the clock timing (symbol timing) of the cycle T _S is reproduced. When the frame length is N,
There is a relationship of T _f = NT _S.

A delay unit 54 receives the pilot symbol output from the sampling switch 51,
An estimated value of the fading fluctuation obtained by dividing this by a predetermined pilot symbol value (for example, 3 + j · 3 having the maximum symbol amplitude) is delayed by one symbol timing and two symbol timings and output.
In FIG. 8, the estimated value of the fading fluctuation obtained by dividing the output of the sampling switch 51 as it is by the value of the pilot symbol has a delay amount of zero, but the delay unit 54
Is shown as the first output of The delay unit 54 is
It is realized by a shift register by converting an analog delay element such as a CCD (Charge Coupled Device) or an input signal into a digital value by using an A / D converter (not shown).

Reference numerals 55, 56 and 57 denote coefficient multipliers, which are pilot symbols output by the sampling switch 51, outputs obtained by delaying the pilot symbols by one clock timing,
Coefficients Q ₁ (m / N), Q ₀ (m / N), and Q, which will be described later, are respectively applied to the outputs delayed by two clock timings.
Multiply by _-1 (m / N). Here, N is the number of symbols in one frame, m is a symbol position in one frame, and a pilot symbol is inserted at a position of m = 0. The adder 58 adds the outputs of the coefficient multipliers 55, 56, 57. The reciprocal calculator 59 calculates the reciprocal of the output of the adder 58 and outputs it to the multiplier 62.

Reference numeral 60 is also a sampling switch, and the received signal U (t) from the LPF 17 in FIG.
Is sampled by the output of the clock reproducing unit 53, and the delay unit 6 of the fading distortion compensating unit 18b is sampled.
Output to 1. A delay unit 61 is used for timing adjustment of two inputs in the multiplier 62, and outputs the output of the sampling switch 60 to one frame timing T.
_{It is} delayed by _f and output to the multiplier 62.

The operation of estimating and compensating for fading distortion will be described using mathematical expressions with reference to FIG. Play the clock,
When one pilot symbol is inserted in one frame (N symbols) while maintaining frame synchronization and symbol synchronization, the pilot symbol at the m = 0th position of the kth frame (t = kT _f , k = 0. 1, 2, ..., Estimated values of the fading fluctuation in the symbol value 3 + j3) are as follows.

[Equation 1] However, the noise component included in the sample value is regarded as zero. Similarly, the estimated value of the fading fluctuation in the time before and after one symbol of t = kT _f is as follows.

[Equation 2]

The m-th information symbol of the k-th frame (t = kT _f + (m / N) T _f , k = 0.1, 2, ..., M =
The estimated value of the fading fluctuation in 1, 2, ..., N−1) is (k−1) th frame, kth frame,
By using a quadratic Gaussian interpolation formula based on the fading variation at each pilot symbol (m = 0) in the (k + 1) th frame, it is shown by the following equation.

[Equation 3] However,

[Equation 4]

[Equation 5]

[Equation 6]

The value of the above equation (1) is output from the adder 58 shown in FIG. The signal that compensates each information symbol of the received signal U (k + (m / N)) is the complex baseband signal U
Dividing (k + (m / N)) by the value of the equation (1) gives the following equation.

[Equation 7] The output of the equation (5) is obtained from the multiplier 62 shown in FIG. The above-mentioned “pilot symbol insertion method” assumes uniform Rayleigh fading (when there is no delayed wave after the direct wave or it can be ignored). If a delay wave is present, the influence of delay distortion increases as the delay time increases, and fading distortion compensation fails.

On the other hand, when there is a delayed wave and the influence of delay distortion is great, frequency selective fading occurs. As a delay distortion countermeasure at this time, conventionally, a method using an adaptive equalizer is described in, for example, “Mobile Communication” by Sasaoka, Ohmsha,
(Heisei 10-5-25), p. It is known as 256-282. The adaptive equalizer includes a decision feedback equalizer (DFE) and a maximum likelihood sequence estimation (MLSE).
n). However, the maximum likelihood sequence estimation is 16QA
When the modulation multi-value number becomes large like M, the amount of calculation increases. For the initial pull-in of the adaptive equalizer,
A training symbol sequence described later with reference to FIG. 10 is added before the information symbol section. Further, in the case where the time fluctuation is severe like in mobile communication, the decision feedback equalizer has an RLS (Recursive Least Squares) with good followability.
An algorithm is used.

FIG. 9 is a block diagram of a conventional radio communication apparatus using an adaptive equalizer. 9A is a block diagram showing the configuration of the transmitter side, and FIG. 9B is a block diagram showing the configuration of the receiver side. In the figure, the same parts as those in FIG. 6 are designated by the same reference numerals and the description thereof will be omitted. In FIG. 9A, reference numeral 71 is a frame signal generation unit that adds a training symbol sequence to an information symbol section. In FIG. 9B, 20 is RL
Decision feedback equalizer (RLS) using the S algorithm
An adaptive equalizer such as -DFE). FIG. 10 is a frame configuration diagram when an adaptive equalizer is used. In the adaptive equalizer, the information symbol section (N
A training sequence (Nt symbol) composed of a known signal is added before the d symbol).

FIG. 11 shows a pilot symbol insertion method (P
SAM) and RLS decision feedback equalizer (RLS-DFE)
FIG. 6 is a diagram showing a simulation result of a fading distortion compensation method using the. In the figure, the horizontal axis represents the delay time of the delayed wave with respect to the direct wave, and the maximum is one symbol period. The vertical axis represents BER (bit error rate). The characteristic is shown with the energy-to-noise power spectral density ratio (Eb / No) of the received modulated wave per bit as a parameter.

The simulation results of the public mobile digital mobile communication system in the 150 MHz band and 400 MHz band are shown below. Bandwidth 12.5kHz / ch, 9.6ksymb
It is 16QAM of ol / sec (104 μsec / symbol).
The maximum Doppler frequency fd = 20 Hz. The power of the direct wave and the power of the delayed wave are made equal. As the frame format, the frame format shown in FIG. 3, which will be described later as an embodiment of the present invention, is adopted. However, unlike the embodiment of the present invention, the modified M series is not used for either the synchronization word SW or the unique word UW. The synchronization word SW and the unique word UW are 16 symbols. R
The feed forward (FF) taps of the LS decision feedback equalizer are 4 (tap interval 1/2 symbol), and the feedback (FB) taps are 2 (tap interval 1 symbol).

As can be seen from FIG. 11, the effect of delay distortion can be reduced by using the adaptive equalizer. However, even when the delay time is short, the BER does not decrease so much,
A sufficient equalization effect cannot be obtained. On the other hand, the pilot symbol insertion method is effective in a uniform fading environment where the influence of delayed waves can be ignored, but if the delay time becomes long, the BER rapidly deteriorates and the fading distortion compensation effect deteriorates. The environment for use in a large zone digital mobile communication system for public works is not limited to an area where the delay time of the delayed wave is short, such as an urban area. There are also long regions. In mobile terminals such as mobile phones, the area used is not constant, and
Since the use area may change depending on the time of use, if one of the above fading distortion compensation methods is fixedly used, it is not possible to effectively use the fading distortion compensation method to reduce the bit error rate. There was a problem that I could not.

[0019]

SUMMARY OF THE INVENTION The present invention has been made to solve the above-mentioned problems, and the bit error rate can be lowered by using a fading distortion compensation method suitable for the propagation environment. An object of the present invention is to provide a wireless communication method and a wireless communication device. Further, it is another object of the present invention to provide a wireless communication method and a wireless communication device that automatically switch to a fading distortion compensation method suitable for the environment used by estimating a fading environment.

[0020]

[0021]

Means for Solving the Problems The present invention, in the invention according to claim 1, in a wireless communication method, is inserted a pilot symbol periodically in the information symbol section, Ri mower in preparative rate training symbol sequence, and it generates a frame signal series, which is also the symbol sequence has been added to the information symbol section autocorrelation function has a peak at phase coincidence timing, and transmits the modulated the frame signal, demodulates the received signal, demodulates A first determination for fading distortion compensation of the demodulated information symbols by estimating fading distortion using the demodulated pilot symbols; and the demodulated training
A second determination is performed to determine the demodulated information symbol by performing adaptive equalization using a symbol sequence, and the autocorrelation function generates a symbol sequence having a peak at a phase matching timing, A correlation function between the symbol sequence and the demodulated received signal is calculated, and a delay spread is output based on the correlation function. When the delay spread is smaller than a threshold, the first
When the delay spread is larger than the threshold value, the information symbol output according to the above determination is selected and output at a predetermined switching timing.
Therefore, the autocorrelation function adds and transmits a symbol sequence having a peak at the phase coincidence timing, and estimates the delay spread on the receiving side, thereby fading distortion compensation by the pilot symbol insertion method and fading by the adaptive equalizer. Among the distortion compensation methods, the bit error rate can be lowered by automatically switching to the fading distortion compensation method suitable for the environment in which it is used at each predetermined switching timing.

[0022]

In the invention according to claim 2 , the pyro
Fading distortion compensation method using
Phasing by adaptive equalization using a training symbol sequence
One of the distortion compensation methods depending on the fading condition.
Radio communication device for transmitting side selectively to receiving side
Met with, while periodically inserted into information symbol section the pilot symbols for performing the full <br/> Ejingu distortion compensation method that by the estimating the fading distortion at the receiving side, according to the applicable応等of Ri More training symbol sequence for carrying out the fading distortion compensation method on the receiving side, or
One was added for outputting delay spread at the reception side, a sequence in the autocorrelation function even symbol sequence having a peak at phase match timing to the information symbol interval,
It has a frame signal generating means for generating a frame signal, a modulating means for modulating the frame signal, and a means for transmitting the modulated frame signal. Thus, at the receiving side, the fading distortion compensation method according to a pilot symbol insertion method, among the fading distortion compensation method using an adaptive equalizer, the fading distortion compensation method suitable for the environment used, Ki automatically switch it out It
As a result, it is possible to obtain a transmission signal that can reduce the bit error rate.

[0024] In the invention of claim 3, claim
2. The wireless communication device according to 2 , wherein the symbol sequence having the peak at the phase matching timing by the quadrature amplitude modulation method is at least one of the binary data sequences having the peak at the auto matching function by the phase matching timing.
The series is assigned to two points of maximum amplitude for each of the I phase and the Q phase. Therefore, it is possible to easily realize a symbol sequence in which the autocorrelation function has a peak at the phase matching timing. Moreover, since the amplitude of this symbol sequence can be set to the maximum value, the estimation accuracy of the delay spread can be improved on the receiving side.

[0025]

[0026] In the invention according to claim 4, in the wireless communication device, with pilot symbols are periodically inserted into information symbol interval, Ri mower in preparative rate training symbol sequence, and the autocorrelation function phase matched Symbol system that is also a symbol sequence that has a peak in timing
Receiving means for receiving a transmission signal whose sequence is modulated by the frame signal added to the information symbol section, demodulation means for demodulating the reception signal, and estimating fading distortion using the demodulated pilot symbol According to the above, the first demodulation means determines the demodulated information symbol after fading distortion compensation, and adaptive demodulation using the demodulated training signal to determine the demodulated information symbol. Second determining means, a symbol sequence generating means for generating a symbol sequence having a peak in the autocorrelation function at a phase coincidence timing, a symbol sequence generated by the symbol sequence generating means, and the demodulated received signal Calculate the correlation function with
The delay spread output means for outputting a delay spread based on the correlation function, and the information symbol output from the first determination means when the delay spread is smaller than a threshold, the delay spread being larger than the threshold. Sometimes, it has an output selecting means for selecting and outputting the information symbol output from the second determining means at a predetermined switching timing. . Therefore, of the fading distortion compensation method by the pilot symbol insertion method and the fading distortion compensation method by the adaptive equalizer, the fading distortion compensation method suitable for the environment used is automatically switched at each predetermined switching timing. , The bit error rate can be reduced.

In the invention described in claim 5 ,
5. In the wireless communication device according to 4 , the quadrature amplitude modulation method is used, and the symbol sequence in which the autocorrelation function has a peak at a phase coincidence timing is at least one of binary data sequences having an autocorrelation function at a peak at a phase coincidence timing.
A sequence is assigned to two points of maximum amplitude for each of the I-phase and the Q-phase, and the symbol sequence generation means responds to at least one sequence of the binary data sequence in which the autocorrelation function has a peak at the phase coincidence timing. And a delay spread output means calculates a correlation function between the symbol series generated by the symbol series generation means and the demodulated received signal as the I-phase component and the Q-phase component.
It is calculated based on the correlation output of the phase component. Therefore, the correlation function can be easily calculated, and the amplitude of this symbol sequence is set to the maximum, so that the estimation accuracy of the delay spread can be improved.

According to the invention described in claim 6 ,
In the wireless communication device according to 4 or 5 , the receiving means receives the frame signal in frame units, and the first and second determining means each temporarily output the determined information symbol. The output selection means has a frame buffer for storing, and when the delay spread is smaller than the threshold value on a frame-by-frame basis, with the predetermined switching timing as the frame timing, the output selection means of the first determination means. The information symbol is output from the frame buffer, and when the delay spread is larger than the threshold value, the information symbol is output from the frame buffer of the second determining means. Therefore, it is possible to reduce the bit error rate by switching to a fading distortion compensation method suitable for the environment used for each frame.

[0029] In the invention of claim 7, claim
In the wireless communication device according to 4 or 5 , the output selection unit operates the first determination unit at the predetermined switching timing when the delay spread is smaller than a predetermined threshold, and the output determination unit operates the first determination unit. When the delay spread is larger than a predetermined threshold value, the first determination means is stopped and the second determination means is controlled to stop the second determination means. It is controlled to operate. Therefore, at a predetermined switching timing, the load of software and the power consumption of hardware can be reduced by stopping the determination means on the side where the information symbol is not output from the output selection means.

[0030]

DESCRIPTION OF THE PREFERRED EMBODIMENTS FIG. 1 is a block diagram showing an embodiment of the present invention. FIG. 1A is a block diagram of a transmitter, and FIG. 1B is a block diagram of a receiver. In the figure, the same parts as those in FIGS. 6 and 9 are designated by the same reference numerals, and the description thereof will be omitted. In the transmitter configuration of FIG. 1A, reference numeral 4 is a frame signal generation unit that inserts a pilot symbol, a synchronization word, and a unique word into information symbols. The synchronization word and the unique word form a training sequence of the adaptive equalizer. At least one of the synchronization word and the unique word is a symbol sequence created by a modified M sequence, which will be described later, and is used to output the delay spread. In the receiver configuration of FIG. 1B, 21 is a delay spread estimation unit, 22 is an output selection unit, and according to the output of the delay spread estimation unit 21, the output of the symbol determination unit 19 and the adaptive equalizer 20. Selectively outputs one of the outputs.

FIG. 2 is a block diagram for explaining the internal processing of the delay spread estimation unit 21 and the output selection unit 22 shown in FIG. 1 (b). In the figure, the same parts as those in FIG. 1 are designated by the same reference numerals. Reference numeral 31 denotes a modified M sequence generation unit, which generates the same modified M sequence as the transmitted modified M sequence on the receiving side. Reference numeral 32 denotes a correlation function calculation unit, which is shown in FIG.
The received signal (I phase, Q output from the LPF 17 of (b)
The correlation function between the two-phase baseband signal of the phase) and the symbol sequence created by the modified M sequence generator 31 is calculated. A delay spread calculation unit 33 estimates the delay spread σ from the correlation function. The method of estimating the delay spread σ will be described later. A comparator 34 compares the delay spread σ with a predetermined threshold σ _th . Three
5 is a switching unit, when the delay spread sigma is smaller than a predetermined threshold value sigma _th is the output of the symbol decision unit 19 as the output data 23, when the delay spread sigma is greater than a predetermined threshold value sigma _th is adapted such The output of the digitizer 20 is output data 23.

FIG. 3 is an explanatory diagram showing a frame format according to an embodiment of the present invention. In the figure, SW (Sync Wo
rd) is a synchronization word, and UW (Unique Word) is a unique word. The number of symbols is appropriately determined. p _{1 to} p ₁₀ are pilot symbols of 1 symbol each, and Data is an information symbol of 15 symbols each. The first and last 1 symbols of the unique word UW can also be used as pilot symbols. The sync word SW and unique word UW are
It is a known symbol series.

The synchronization word SW is used not only for frame synchronization, but also as a training signal for initial pull-in of the adaptive equalizer 20. The unique word UW is used as a training signal for retraining the adaptive equalizer 20 in the middle of the information symbol section. Also, the unique word UW, and
At least one of the synchronization words SW is used as a known symbol sequence for outputting the delay spread. Techniques for estimating delayed waves with delay spread are Kei Suzuki, Seiichi Sampei, Norihiko Morinaga, "Channel characteristics estimation method in adaptive modulation method" IEICE Technical Report RCS94-65.
(1994-09) and the like.

In the above-mentioned paper, the modified M sequence is used as the symbol sequence for estimating the delay spread. The modified M sequence is an M sequence (Maximum-length linear shift-register sequenc
The side lobe of the autocorrelation function is made zero by adding an appropriate direct current component (DC offset) to e). The autocorrelation function at the phase where the timings do not match is completely zero. The autocorrelation function when the timing coincides with the period N _est of the modified M sequence is (N _est +
It becomes 1). Further, the value (D) of the above-mentioned DC component is given by the following equation.

[Equation 8] Delay wave that delays the direct wave within 1 symbol time
The delay spread of this delayed wave, assuming that
To output N as the symbol sequence _estof
(Modified) Add 1 symbol before and after the M-sequence code
(N_est+2) Use the symbol.

More generally, a modified PN sequence may be formed by adding a DC offset to a PN sequence (pseudo noise sequence), similar to the modified M sequence described above. Further, as the above-mentioned symbol series, M series, more generally PN series may be used. Since even the M series and the PN series have sharp peaks of autocorrelation when the timings match, they can be used for estimating the delay spread. However, when the period of this symbol sequence cannot be taken long, an estimation error occurs because the value of the autocorrelation function when the timings do not match is not completely zero.

Next, the delay spread output method will be described in accordance with the above-mentioned paper. The delay profile of the reception slot is obtained by calculating the correlation function of the received signal and the modified M sequence, and the delay spread is output from this delay profile. The delay spread is the standard deviation (square root of variance) centered on the average received power delay time with respect to the average power delay profile.

First, the propagation path model will be described. Land movement
In communication, the maximum delay time is about 20 μsec.
Therefore, in the case of transmitting several tens of k symbols / sec,
It is sufficient to consider the delay within 1 symbol. Also,
Received signal is 1 symbol length T _SSampling only at intervals
Is not done, 1 symbol length T_SInterval discrete channel model
Written in Dell. Therefore, as a propagation path model,
Delay time T_S2 wave Rayleigh model is used. Impal
The response h (t) is approximately given by the following equation. h (t) ＝ h₀(t) δ (t) ＋ h₁(t) δ (t-T_s) (7) Here, δ (t) is a delta function. Fading fluctuation
Is gradual and is constant during measurement (within one frame)
In the receiver, the received signal after sampling and transmission
Correlate with the modified M series.

This correlation is performed in two orthogonal I-phase and Q-phase channels, and the complex delay profile h _j (k) shown in the following equation is measured in the k-th frame. However, the correlation output due to the noise component is ignored as a measurement error.

[Equation 9] Where w _i is the i-th symbol based on the modified M sequence, y
(i) indicates the received signal (complex envelope) sampled at t = iT _s , and * indicates that w is a complex conjugate number. As already mentioned, a maximum delay of 1 symbol is assumed. When measuring delayed and direct waves, a dummy signal of 1 symbol is inserted before and after the modified M sequence in order to remove the mutual correlation. Word SW is N
_est +2 symbols.

The power delay profile p (k) of the modified M sequence corresponding to the impulse response h (t) of the above-mentioned two-wave model is given by the following equation. p (k) ＝ p ₀ (k) δ (t) ＋ p ₁ (k) δ (tT _s ) (9) where p _i (k) (i = 0,1) is the complex of equation (8). It is given by the following equation using the delay profile h _j (k). p _i (k) = | h _i (k) | ² (10)

As already explained, the delay spread is
Since it is the standard deviation of the power delay profile centered on the average delay time, the delay spread is estimated from the above equations (9) and (10). In the case of a two-wave model in which the delay time is T _s (one symbol time) and the amplitude ratio of the delayed wave and the direct wave is D / U = 1 / α, the fading fluctuation is gradual and p _i ( When the value of k) does not change, p _i (k) = p _i (i = 0, 1) and the delay spread σ is as follows.

[Equation 10]

Here, in one embodiment of the present invention,
A method of creating a symbol sequence according to the 16QAM modulation rule based on the modified M sequence will be described. Figure 4 shows 16QA
It is a figure which shows the signal space diagram of M. 16QAM
, The binary 4-bit transmission data is the I-phase component,
The value of the Q-phase component is assigned to one of 16 symbol points of -3, -1, 1, and 3, respectively. In contrast,
Similar to the training sequence of the adaptive equalizer, the symbol sequence for calculating the correlation function for estimating the delay spread has the maximum symbol amplitude.
Use the D symbol. By using the one having the largest symbol amplitude, the signal-to-noise ratio at the time of estimating the delay spread on the receiving side is improved, so that the estimation accuracy is improved. At least two symbols are selected from the above-mentioned four symbols in accordance with the modified M-sequence to be the above-mentioned symbol sequence. If the period of the modified M series is 15,
1 by adding dummy symbols at the beginning and end
It becomes a symbol sequence of 7 symbols.

There are a plurality of methods for assigning the modified M sequence to symbols. The modified M series is a binary 0, 1 data series and has a DC offset added thereto. Firstly, there is a method of assigning two symbol points to binary 0 and 1 of one series of modified M series. For example, A, C or C, A or B, D or D, B are assigned to binary 0 and 1 of one modified M sequence, respectively. In this case, the phase difference can also be increased. other than this,
Assignments such as A, D, A and B can also be made.
Secondly, there is a method using two modified M sequences having the same period. Binary combination of two series (0,0),
4 for each of (0,1), (1,0), and (1,1)
One of the symbols A, B, C, D is assigned. In this case, the randomness of the phase of the modulated signal increases.

However, when the modified M-sequence is used, the modified M-sequence is the M-sequence added with the DC offset. Therefore, in both the first and second methods, the symbol points A, B, and At points C and D, the values of the I-phase component and the Q-phase component are (3, 3), (-3, 3), (-3, -3),
From (3, -3), DC for I-phase component and Q-phase component
It is necessary to shift the position by the offset and perform quadrature modulation. On the reception side, binary data of the modified M series for I phase and Q phase is generated in correspondence with one series or two series on the transmission side. The binary value 0 is set to -1 for calculating the correlation function. For the symbol correlation function, a correlation function is calculated for the same phase between the I-phase baseband signal and the Q-phase baseband signal, which are received signals after quadrature demodulation, and the generated binary data, The delay profile is calculated by adding the calculation results of the respective phases to obtain the delay profile represented by the above-mentioned equation (8).

The modified M sequence or M sequence does not necessarily have to correspond to the symbol having the maximum amplitude. Further, it is not always necessary to assign it to one of the 16QAM symbol points, but one of the 16QAM symbol points
If assigned to one, the symbol sequence for delay spread estimation can also be used as the training sequence of the adaptive equalizer. Even if it is not assigned to one of the symbol points, it is desirable to increase the amplitude.

Returning to FIG. 2, the description will be continued. The correlation function calculation unit 32 measures the complex delay profile by calculating the correlation function of the received signal and the symbol sequence based on the transmission modified M sequence. The delay spread calculator 33
The delay spread σ is output from the complex delay profile. The comparison unit 34 compares the estimated value σ with a predetermined delay spread threshold σ _th .

The fading distortion estimation / compensation unit 18 estimates the fading distortion from the received complex baseband signal with the periodically inserted known pilot symbols by the interpolation method, and based on this estimated value, performs the fading distortion compensation. Then, the symbol determination unit 19 determines the symbol of the multi-level QAM and outputs the output signal Out1. On the other hand, the adaptive equalizer 20 estimates the fading gain using the training sequence, updates the tap coefficient based on the RLS algorithm or the like, compensates the delay distortion and the fading distortion, and determines the multi-level QAM symbol. Then, the output signal Out2 is output.

In the above-mentioned configuration, the fading distortion compensation mode by the "pilot symbol insertion method" and the fading distortion compensation mode by adaptive equalization using the synchronization word and the unique word as the training sequence are simultaneously performed in parallel. The switching unit 35 selects Out1 that has been subjected to fading distortion compensation by the pilot symbol insertion method as output data 23 when σ <σ _{th in} the comparison result of the comparator 34. On the other hand, when σ ≧ σ _th , Out2 whose fading distortion is compensated by the adaptive equalizer is selected and used as the output data 23.

FIG. 5 is a diagram schematically showing the relationship between delay time and BER (bit error rate) for the pilot symbol insertion method and the two fading distortion compensation methods using the adaptive equalizer. In the fading distortion compensation method by inserting pilot symbols, BE is used when the delay time is small.
Although R is low, the BER rapidly increases as the delay time increases. On the other hand, in the fading distortion compensation method using the adaptive equalizer, when the delay time is small, the BER
Is not so low, but BE is
The increase rate of R is small.

Yoshio Karasawa, "Analysis of Wideband Digital Transmission Characteristics by SB-1-4 Equivalent Transmission Line Model", Proc. Of the 2000 IEICE General Conference, p. 709-710, (2000
-3-7), Yoshio Karasawa, Masato Iwai, "Nakagami-Rice Fading Equivalent Transmission Line Model-Regarding Expansion of Its Function-", IEICE Technical Report RCS98-24, pp.1-6
(1998-5), in the range where the average delay amount or delay spread σ is 30% or less of the symbol length T _s , the two-wave model of the equivalent propagation path model is effective, and the average delay amount is And the delay spread are equal, the delay times of the direct wave and the delayed wave have a relation of twice the delay spread σ (the delay time is within the range of 60% or less of the symbol length T _s , the model is effective). Therefore, the delay time τ of the delayed wave with respect to the direct wave is estimated by doubling the measured value σ of the delay spread output from the delay spread estimation unit 21 shown in FIGS.

Therefore, as a method for setting σ _th , for example, based on the simulation result as shown in FIG. 11, the characteristic curve of the pilot symbol insertion method (PSAM) and the decision feedback equalizer (RLS- The value of 1/2 of the delay time at the intersection with the characteristic curve of DFE) may be set as the threshold σ _th for the measured delay spread σ.
As a result, when the measured delay spread σ is small with _respect to the threshold σ _th , the pilot symbol insertion method (PSAM) is used. When the delay spread is large, the fading distortion compensation method by the adaptive equalizer is used. Therefore, fading distortion compensation can be efficiently performed regardless of the difference in the propagation path characteristics.

In the above description, the synchronization word SW and / or the unique word UW is the symbol series based on the modified M series or the like. The unique word UW is also used for training the adaptive equalizer, and the synchronization word is
It is also used for frame synchronization and training of adaptive equalizers. Therefore, the number of symbols that need to be transmitted other than the information symbols is not increased, so that the transmission efficiency does not decrease. However, in addition to the synchronization word SW and the unique word UW, a dedicated symbol sequence for delay spread estimation may be added to the information symbol section. In addition, TD
In a public mobile digital mobile communication system of MA (Time Division Multiple Access) system, the frame format shown in FIG. 3 corresponds to one slot (one burst) of one mobile terminal. In this case, when transmitted from the mobile terminal to the base station, a guard symbol, a ramp symbol, etc. are added to the beginning of the frame.

In the above-described embodiment, the symbol for estimating the delay spread is inserted for each frame. Therefore, it is possible to select the output on a frame-by-frame basis. In this case, the symbol determination unit 19 and the adaptive equalizer 20 temporarily store the determined information symbol in the frame buffer. The output selection unit 22 outputs the information symbol determined by the symbol determination unit 19 from the frame buffer when the delay spread is smaller than the threshold value, and outputs the information symbol determined by the adaptive equalizer 20 when the delay spread is larger than the threshold value. Output from the buffer.

In the above description, the output selection unit 22 can switch to one of the information symbols determined by the symbol determination unit 19 and the information symbols determined by the adaptive equalizer 20, for each frame. However, any predetermined timing other than the frame timing may be used as the switching timing. Even if the mobile terminal moves, the fading environment does not change rapidly, so depending on the delay spread,
The symbol determination unit 19 and the adaptive equalizer 20 may be switched at predetermined plural frame intervals or at regular time intervals. In the call setting sequence at the start of communication or at the time of handoff for switching the zone, the symbol determination unit 19, depending on the delay spread,
The adaptive equalizer 20 may be switchable.

In the above description, fading distortion estimation
The compensating unit 18, the symbol determining unit 19, and the adaptive equalizer 20 are operated simultaneously. Instead of this, when the delay spread is smaller than a predetermined threshold value, the output selection unit 22 causes the fading distortion estimation / compensation unit 18 and the symbol determination unit 19 to operate at a predetermined switching timing, and the adaptive equalizer. Control to stop 20
When the delay spread is larger than a predetermined threshold, fading distortion estimation /
The compensation unit 18 and the symbol determination unit 19 may be stopped and the adaptive equalizer 20 may be controlled to operate.

On the side where the operation is stopped, the processing load of software can be reduced when the signal processing is performed by software. When the signal processing is performed by hardware, power supply can be cut off to reduce power consumption. At the next predetermined switching timing, the fading distortion estimation / compensation unit 18, the symbol determination unit 19, and the
Among the adaptive equalizers 20, the side to operate and the side to stop the operation are determined.

In the above description, the delay spread estimation is performed for each frame. However, if the switching is not performed on a frame-by-frame basis, statistically performing delay spread calculation for a plurality of frames improves the accuracy.
In addition, the spread calculation may be performed to reduce the processing only at the time of call setup sequence or handoff when it is necessary to determine whether to switch. Furthermore, the symbol sequence for delay spread estimation may not be added for each frame, but may be transmitted only when it is necessary to determine whether to switch.

In the above description, the estimation of the delay spread automatically switches to the fading distortion compensation suitable for the environment in which the receiver of the mobile terminal is used. But like a base station receiver,
Once installed, the usage environment may not change. In such a case, it is sufficient to first select one of the fading distortion compensation methods, and it is not necessary to estimate the delay spread. Therefore, the output mode selection unit 22 shown in FIGS. 1 and 2 selects one of the fading distortion estimation / compensation unit 18 and the symbol determination unit 19 or the adaptive equalizer 20 by the hand of the user. To do. In this case, signal processing and power supply on the unused side are stopped. The signal processing and power supply of the delay spread estimation unit 21 are also stopped.

[0058]

As is apparent from the above description, according to the present invention, it is possible to reduce the bit error rate by using the fading distortion compensation method suitable for the propagation environment. Furthermore, by estimating the fading environment, it is possible to automatically switch to a fading distortion compensation method suitable for the environment used.

[Brief description of drawings]

FIG. 1 is a block diagram showing an embodiment of the present invention.

FIG. 2 is a delay spread estimation unit 21 shown in FIG.
3 is a block diagram illustrating an internal process of an output selection unit 22. FIG.

FIG. 3 is an explanatory diagram showing a frame format according to an embodiment of the present invention.

FIG. 4 is a diagram showing a signal space diagram of 16QAM.

FIG. 5 is a diagram schematically showing the relationship between delay time and BER (bit error rate) for two fading distortion compensation methods.

FIG. 6 is a block configuration diagram of a conventional wireless communication device using the “pilot symbol insertion method”.

FIG. 7 is an explanatory diagram showing a frame structure of transmission data in the “pilot symbol insertion method”.

8 is a block configuration diagram for explaining a fading distortion estimation / compensation unit 18 shown in FIG. 6 (b).

FIG. 9 is a block configuration diagram of a conventional wireless communication device using an adaptive equalizer.

FIG. 10 is a frame configuration diagram when an adaptive equalizer is used.

FIG. 11: Pilot symbol insertion method (PSAM) and R
It is a diagram which shows the simulation result of the fading distortion compensation method which uses an LS decision feedback equalizer (RLS-DFE).

[Explanation of symbols]

1 ... Data signal, 2 ... Serial / parallel converter, 3 ...
Baseband signal generator 4, frame signal generator 5,
... low-pass filter, 6 ... quadrature modulator, 7 ... local oscillator, 8 ... amplifier, 9 ... transmission antenna, 11 ... receiving antenna, 12 ... bandpass filter, 13 ... automatic gain control section, 14 ... automatic frequency control section, 15 ... Quadrature demodulator, 16
... local oscillator, 17 ... low-pass filter, 18 ... fading distortion estimation / compensation unit, 19: symbol determination unit, 20:
Adaptive equalizer, 21 ... Delay spread estimation section, 22 ... Output selection section, 23 ... Output data

Continuation of the front page (56) Reference JP-A-11-112591 (JP, A) JP-A-10-41876 (JP, A) JP-A-6-232939 (JP, A) JP-A-2000-59269 (JP, A) JP 2000-216843 (JP, A) Takashi Suzuki, Propagation path characteristic estimation method in adaptive modulation method, IEICE Technical Report RCS94-65, Japan, September 1994, Vol. 94, No. 238, pages 37-42 (58) Fields investigated (Int. Cl. ⁷ , DB name) H04L 27/36 H04B 7/005 H04J 3/00 H04L 27/38

Claims (7)

(57) [Claims] With periodically inserted 1. A pilot symbol information symbol period, Ri mower in preparative rate training symbol sequence and the information symbol sequence, which is also a symbol sequence autocorrelation function has a peak at phase coincidence timing By generating a frame signal added to the section, modulating the frame signal and transmitting it, demodulating the received signal, and estimating fading distortion using the demodulated pilot symbol, the demodulated information symbol is A first determination is performed after fading distortion compensation is performed, and a second determination is performed to perform demodulation of the information symbols by performing adaptive equalization using the demodulated training symbol sequence . The autocorrelation function generates a symbol sequence having a peak at phase matching timing, A correlation function between the symbol sequence and the demodulated received signal is calculated, a delay spread is output based on the correlation function, and the information symbol output by the first determination when the delay spread is smaller than a threshold value. When the delay spread is greater than the threshold, the second
The wireless communication method, characterized in that the information symbol output according to the determination is selected and output at a predetermined switching timing. 2. Fadin using pilot symbols
Distortion compensation method and adaptation using training symbol sequence
One of the fading distortion compensation methods by equalization is
To the receiver, which executes selectively according to the state of
A wireless communication apparatus on the transmission side, while periodically inserted into information symbol section the pilot symbols for performing the fading distortion compensation method that by the estimating the fading distortion at the receiving side, the suitable応等of More training symbol sequence for performing the fading distortion compensation method on the receiving side according to is, and delay spray
For outputting red on the receiving side, the autocorrelation function is even symbol sequence having a peak at phase coincidence timing
The sequence was added to the information symbol interval, radio, wherein the frame signal generating means for generating a frame signal, and modulating means for modulating the frame signal, means for transmitting the modulated the frame signal, to have a Communication device. 3. A quadrature amplitude modulation method is used, and the symbol sequence having a peak in the autocorrelation function at the phase coincidence timing is at least one sequence of binary data sequences having an autocorrelation function at the phase coincidence timing in the I phase. , Q phase is assigned to two points of maximum amplitude respectively, The wireless communication device according to claim 2 , characterized in that. Together are inserted periodically to 4. A pilot symbol is an information symbol period, Ri mower in preparative rate training symbol sequence, and the symbol sequence autocorrelation function is even symbol sequence having a peak at phase coincidence timing <br /> Receiving means for receiving the transmission signal modulated by the frame signal added to the information symbol section, demodulating means for demodulating the received signal, and estimating fading distortion using the demodulated pilot symbol Thus, the demodulated information symbol is subjected to fading distortion compensation and then is determined, and the demodulated training symbol is used to perform adaptive equalization to determine the demodulated information symbol. Second determining means for determining, and a symbol having a peak in the autocorrelation function at a phase coincidence timing. And a symbol sequence generated by the symbol sequence generation means,
A delay spread output unit that calculates a correlation function with the demodulated received signal and outputs a delay spread based on the correlation function; and a delay spread output unit that outputs the delay spread when the delay spread is smaller than a threshold value. Output selection means for selecting and outputting the information symbol at a predetermined switching timing, the information symbol output from the second determination means when the delay spread is larger than the threshold value. Wireless communication device. 5. A quadrature amplitude modulation method is used, and the symbol sequence having a peak in the autocorrelation function at a phase coincidence timing is at least one sequence of a binary data sequence in which the autocorrelation function has a peak at a phase coincidence timing. , Q phases are assigned to two points of maximum amplitude, and the symbol sequence generation means generates a symbol sequence corresponding to at least one binary data sequence having a peak in the autocorrelation function at a phase coincidence timing. The delay spread output means generates a correlation function between the symbol sequence generated by the symbol sequence generation means and the demodulated received signal based on a correlation output of the I-phase component and the Q-phase component. It calculates, The wireless-communications device of Claim 4 characterized by the above-mentioned. 6. The receiving means receives the frame signal on a frame-by-frame basis, and the first and second determining means each include a frame buffer for temporarily storing the determined information symbol. The output selection means uses the predetermined switching timing as the frame timing, and the information from the frame buffer of the first determination means when the delay spread is smaller than the threshold value in units of the frame. The radio communication device according to claim 4 or 5 , wherein a symbol is output, and when the delay spread is larger than the threshold value, the information symbol is output from the frame buffer of the second determining unit. 7. The output selecting means operates the first determining means and stops the second determining means at the predetermined switching timing when the delay spread is smaller than a predetermined threshold value. When the delay spread is larger than a predetermined threshold value, the first determination means is stopped and the second determination means is operated at the predetermined switching timing. The wireless communication device according to claim 4 or 5 , characterized in that.