JP3430194B2 - Pulse width modulation power converter - Google Patents

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power converter for converting AC power into DC power by pulse width modulation of a switching circuit, and more particularly to a power converter suitable as a converter in an inverter device.

[0002]

2. Description of the Related Art A well-known diode bridge circuit is a typical example of a converter device for converting AC power into DC power. However, this diode bridge circuit has a problem that the power supply current contains many harmonics, and further,
Since it does not have an inverse conversion function, there is a problem that power regeneration cannot be performed.

On the other hand, as a converter circuit that can cope with the problems of the diode bridge circuit, for example, a PWM
(Pulse width modulation) There is a power converter. But this PW
The M power conversion device requires a control device which is not necessary in the diode bridge circuit, and similarly, there is a problem in that a switching loss occurs due to a switching element which does not exist in principle in the diode bridge circuit.

Therefore, as one method for reducing the problems of the PWM power converter, a simple control method in which the instantaneous value control of the current is omitted is disclosed in, for example, the following document. 1990, IEEJ Transactions, Volume 110, 7
No., “Power Factor Control Method Three-Phase Voltage PWM Control Power Converter”

The conventional technique disclosed in this document will be described below. This conventional technique relates to an inverter device for converting DC power converted by a converter device into AC power of variable frequency, and FIG. 1 shows a circuit configuration, 1 is a three-phase AC power supply, 2 is a converter unit for converting three-phase AC into DC, 3 is a capacitor for smoothing DC voltage, and 4 is an inverter unit.

Here, the converter section 2 is composed of self-extinguishing type switching elements (or switching circuits) 2a to 2f, which are connected by means of the Grenz wire, and diodes 2g to 2l (ell), which are connected in antiparallel to these switching elements. Has been done. Similarly, the inverter unit 4 is also composed of self-extinguishing type switching elements (or switching circuits) 4a to 4f and diodes 4g to 4l connected in antiparallel to the respective switching elements. The induction motor 5 is connected to the AC output of the inverter unit 4.

Next, 6 is a reactor, which acts to suppress the PWM frequency component in the harmonics contained in the alternating current, and 7 is an alternating voltage / current detector, which is a current transformer 7a and an instrument transformer. The device 7b serves to detect an AC voltage and an AC current, respectively, 8 is a DC voltage detector, and 9 is a power factor detector. From the AC voltage and the AC current detected by the AC voltage / current detector 7, It works to detect the power factor.

Further, 10 is a PLL control circuit which outputs a phase command α for keeping the power factor constant, and 11 is a DC voltage control circuit which sets the DC voltage detected by the DC voltage detector 8 to a set value. A device for outputting a modulation degree command M for adjustment, 12 is a sine wave PWM control signal generator, and is an ignition (ON) / extinguishment (OFF) command 12 for the switching elements 2a to 2f.
a to 12f are output, 13 is a sine wave PWM control signal generator on the side of the inverter, and switching elements 4a to 4
It outputs the firing (ON) / extinguishing (OFF) commands 13a to 13f of f.

FIG. 23 shows a sine wave PWM control signal generator 1
2 is a diagram for explaining the operation of FIG. 2, in which a modulated wave having an amplitude modulation rate M is combined in a phase that leads the AC voltage signal ER by α.
The ignition / extinguishing commands 12a and 12d for the switching elements 2a and 2d are output according to the magnitude of the triangular waveform carrier.

FIG. 24 shows an AC power supply voltage ER, a current IR,
In the vector diagram for one phase showing the relationship between the AC reactor voltage drop EX and the converter voltage EC, if the modulation factor M is equal to or higher than the critical modulation factor MC determined by the load state, the phase command α is adjusted to constantly adjust the power factor. It is said that control to 1 can be performed. If the load fluctuation on the inverter side is small, the modulation factor M may be fixed to a value close to 1. In this case, the DC voltage control circuit 11 for adjusting the modulation factor M is not always necessary. It is said that.

Therefore, according to the converter section 2 of the inverter device shown in FIG. 22, the harmonic current can be suppressed more than that of the diode bridge circuit, and the AC current control required in the usual PWM power converter is omitted. can do.

[0012]

The above-mentioned prior art cannot be said to be sufficiently careful in maintaining the stability of the operation and improving the efficiency, and as will be described below, when the operating state changes, There are problems in terms of continuity of operation and reduction of loss in the switching element or switching circuit. According to the above-mentioned conventional technique, the harmonic current can be suppressed more than that of the diode bridge circuit, and the AC current control required in a normal PWM power converter can be omitted.

However, when the operation needs to be continued even when the AC power supply voltage drops or is unbalanced, the inductance value of the AC reactor 6 is increased to suppress the overcurrent, or the converter unit 2 is configured. There is a problem that it is necessary to increase the capacity of existing elements and parts so that they can withstand overcurrent sufficiently.

Further, although the DC voltage control circuit can be omitted in the above-mentioned prior art, in this case, even if the load fluctuation on the inverter side is small at normal times, it is transiently connected directly to, for example, the AC motor of the inverter load. When the load torque of the machine is suddenly increased, there is a problem that the DC voltage drop becomes large and the operation cannot be continued.

It is an object of the present invention to maintain a continuous operation when the power source or the load fluctuates without controlling the AC current and the DC voltage, and to improve the efficiency by reducing the loss by the switching element or the switching circuit. It is to provide a PWM power conversion device that can be sufficiently obtained.

[0016]

The above object is to provide a pulse width with a self-extinguishing type switching circuit and a diode circuit, which are connected in antiparallel with each other, between a three-phase AC power supply and a DC smoothing circuit having a capacitor. In the modulation power conversion device, for the arm connected to the phase of the three-phase AC power supply in which the amplitude of the phase voltage is minimized, the switching circuit is subjected to pulse width modulation control, and the arm connected to the remaining two phases. The above can be achieved by controlling so that the diode circuit is made to flow during the forward conversion operation and the switching circuit is made to flow during the reverse conversion operation. Further, at this time, an abnormality on the AC power supply side may be detected to block the ignition command to all switching circuits, and the block may be released after the abnormality detection is completed.

Next, the reason why the object of the present invention is achieved by the above-mentioned means will be described below in comparison with the known converter circuit using a diode bridge shown in FIG. First, in a diode bridge converter circuit as shown in FIG. 25, this is also well known, but as shown in FIG. 26, the waveform of the AC power supply current IR becomes a distorted waveform that is far away from the sine waveform. As a result, a large harmonic current is included, which causes various troubles.

Therefore, as a countermeasure against this, there is a method of suppressing the harmonic current by increasing the inductance value of the AC reactor. The operation of the diode bridge converter circuit in this case is expressed by the following formula (1), that is, ,

[0019]

[Equation 1]

When the coordinates are converted to the αβ axis by and displayed on the plane, the locus of the alternating current corresponding to the current waveform of FIG. 26 is as shown in FIG. 27, and the locus of the converter voltage is shown in FIG.
As shown in 8. In this equation (1), f represents a general variable.

The current and voltage at points A to I in FIG. 26 correspond to points A to I in FIGS. 28 and 28, respectively. At this time, the AC power supply voltage is rotating at a constant angular velocity on the circumference of a constant radius, so the difference between this and the converter voltage locus becomes the AC reactor voltage, and the locus of AC current and the locus of voltage are both It is symmetrical at a 60 ° cycle. In the following description, the R-phase current waveform shown in FIG. 26 will be described, but the other S-phase and T-phase are also symmetrical and the same.

First, in the current locus of FIG. 27, between BC and EF
As shown in FIG.
The waveform of 6 corresponds to the period in which the current flows in two of the three phases. Next, on the converter voltage locus in FIG. 28,
During the two-phase flow period, the rotational angular velocity is low and the commutation period is A
The rotational angular velocity is high between B and between GH. Then, when the inductance of the AC reactor is increased, the AC current is suppressed because the commutation period becomes longer and the angular velocity becomes smaller, and as a result, the rotation of the voltage locus approaches the uniform angular velocity. Even if the voltage locus remains hexagonal, if it can be rotated at a constant angular velocity, the harmonic current can be further reduced.

Here, in the present invention, the means "with PWM control of the switching element of the arm connected to the phase having the smallest absolute value among the three-phase AC power supplies," is a means for calculating the voltage locus at a constant angular velocity. It acts to rotate.

On the other hand, in the converter circuit using the diode bridge, the DC voltage is automatically adjusted to be substantially constant even when the load on the DC side suddenly changes, so that even if the AC reactor is not changed, the converter circuit does not control anything. However, it has the advantage that operation can be continued stably. This is because the size of the hexagon of the converter voltage locus shown in FIG. 28 is proportional to the DC voltage of the smoothing capacitor, so that even if the DC load fluctuates, it is automatically adjusted accordingly, and the PWM voltage This is because, unlike the converter, the modulation factor is not suppressed to 1 or less, so that the fluctuation rate of the DC voltage is suppressed to be small.

Therefore, in the present invention, two phases having a large absolute value among the three-phase AC power supplies are continuously ignited.
The magnitude of the voltage locus is proportional to the DC voltage, and the DC voltage characteristic close to that of the diode bridge converter circuit can be obtained.

As a result, when the ignition command to the self-extinguishing type switching element is completely blocked to shift to the operation equivalent to that of the diode bridge converter circuit, or when the ignition command is restarted, the capacitor voltage changes. In most cases, the operation on the DC load side can be continued even when the AC power supply is abnormal.

Further, according to the present invention, the switching elements of the three-phase AC power supply, which have a large absolute value, are continuously ignited, so that the frequency of the switching operation is reduced, and the switching loss is accordingly reduced. The efficiency is reduced.

[0028]

BEST MODE FOR CARRYING OUT THE INVENTION A pulse width modulation power converter according to the present invention will be described in detail below with reference to the embodiments shown in the drawings. First, the operation mode of the switching element in each embodiment will be defined by the equivalent circuit shown in FIG. 29 and then described. In FIG. 29, three switches SR, SS and ST correspond to R, S and T phases,
The +1 represents the conduction of the upper arm, and the -1 represents the conduction of the lower arm. Then, the operation modes of these switches are eight kinds as shown in the following (Table 1).

[0029]

[Table 1]

Here, assuming that the DC voltage VDC is constant, the voltage generated by the converter in each mode is
FIG. 30 shows the coordinates converted to the αβ axis by the equation (1). Therefore, in FIG. 30, only the switch SS has a different switch state between the point A in mode 1 and the point B in mode 2, and the switch SR remains the +1 side and the switch ST
Remains at the −1 side, and the switch state does not change in either case.

Therefore, now, the switch SR and the switch S
If T is fixed and only the switch SS is PWM-modulated, and the ratio of the time connected to the −1 side is γ and the ratio of the time connected to the +1 side is (1−γ), the converter is As shown in FIG. 30, the locus of the voltage is CB / AB
= Γ, therefore, when γ is continuously changed from 1 to 0, the locus moves from point A to point B. Further, the relationship between the phases θ and γ of the locus is given by the following expression (2), and the condition for the phase θ to have a constant angular velocity can be obtained from this.

[0032]

[Equation 2]

The above is the movement between the points AB. Similarly, when the conditions for rotating the voltage locus at a constant angular velocity are obtained, the following equation (3) is obtained and PW is obtained.
The M-modulated switch is as shown in FIG.

[0034]

[Equation 3]

For example, in FIG. 31, the phase is θ ₁
At this time, the switch SR is switched, and the conduction ratio on the −1 side is the length AB and the conduction ratio on the +1 side is the length BC.
It turns out that Therefore, assuming that one of the three switches is PWM-controlled in order according to FIG. 31, the converter voltage (R phase) at this time is as shown in FIG. Note that, in FIG. 32, the average current value excluding the pulsation due to PWM control is shown.

As a result, among the three-phase AC power supplies, the switching element of the arm connected to the phase having the smallest absolute value is PWM-controlled, and the arm connected to the remaining two phases is subjected to the forward conversion operation. Sometimes it goes through a diode,
During the inverse conversion operation, the switching element is controlled so as to flow continuously, and in the case of the voltage waveform of FIG. 32, the fifth and seventh harmonics are about 3 with respect to the fundamental wave.
%, The 11th and 13th harmonics are each about 0.8%.
This is based on the point that the locus of the converter voltage is controlled to be hexagonal as shown in FIG. 30 in the first embodiment of the present invention. Mold embodiment.

As described above, the magnitude of the harmonic current is determined by the inductance value of the AC reactor,
In the case of this hexagonal basic type embodiment, as described above, it is understood that the harmonic current can be suppressed more than the diode converter circuit even with the same inductance value.

Next, FIG. 33 shows a case where the locus of the converter voltage is a dodecagon so that the fifth and seventh harmonic components are theoretically eliminated. The embodiment is referred to as a dodecagonal basic embodiment.
In FIG. 33, points E and F are angles AOE,
At a position where the angle AOF is 15 °, the point C is at a position on the line segment AB where the angle AOC becomes the phase θ. Then, C in FIG.
The point is the same as the point C in FIG.
The conduction rate γ ₁ during the time of connection to the +1 side of S is calculated by the following (4)
The control according to FIG. 33 can be realized by setting the value determined by the formula.

[0039]

[Equation 4]

Here, in order to make the voltage locus have a dodecagonal shape, it is necessary to further move the voltage vector to point D. For this purpose, the conduction period from mode 7 to mode 8 shown in Table 1 is set. Should be provided. In this case, since the period of the mode 1 is longer than the period of the mode 2 on the locus from the point A to the point E, the voltage vector at the point O is changed from the mode A to the mode SR by changing the switch SR to -1. However, if the ratio of the period of mode 7 is γ ₂ here, OD / OC
= 1-γ ₂ .

On the other hand, the line segment OD and the line segment OC have the following (5)
There is an expression relationship.

[0042]

[Equation 5]

Therefore, the phases θ and γ ₂ may be made to have a relationship represented by the following equation (6).

[0044]

[Equation 6]

As a result, at the voltage vector D point, the ratio of the period of mode 1 is (1-γ ₂ ) × γ ₁ and the ratio of mode 2 is
(1-γ ₁ ) × (1-γ ₂ ), and the ratio of mode 7 is γ ₂
It can be realized by By the way, the above is the case where the point C is on the line segment GE, but when the point C is on the line segment EH, γ ₂ = 0. Therefore, 6 in FIG.
The flow rate may be controlled as in the case of the prism.

Next, the above is the case of the movement between the line segments AH. Similarly, the conditions for rotating the voltage locus at a constant angular velocity are obtained as shown in FIG. In FIG. 32, for example, when the phase is θ ₂ , it is the switches SS and SR that are switched. The -1 side conduction ratio of the switch SS is the line segment AB, and the +1 side conduction ratio is the line segment B.
D, the flow rate on the -1 side of the switch SR is the line segment AC, and the +1 side is the line segment CD.

When the three switches are PWM-controlled according to FIG. 34, the switching element of the arm connected to the phase having the smallest absolute value in the three-phase AC power supply is PWM-controlled and the remaining ones are PWM-controlled. The arms connected to the two phases are controlled so that they flow through the diode during the forward conversion operation and the switching element continuously flows during the reverse conversion operation. The converter voltage (R phase) at this time is As shown in FIG.

In the case of the voltage waveform of FIG. 35, the fifth harmonic and the seventh harmonic are theoretically eliminated from the fundamental wave, and the 11th 1st
The third harmonics are each about 0.8%, so it can be seen that the dodecagonal basic embodiment can further suppress harmonic currents than the hexagonal basic embodiment of FIG. .

Next, specific examples of each embodiment of the present invention will be described. First, FIG. 1 is a specific example of the above-described hexagonal basic type embodiment. Here, the same elements as those in the conventional technique shown in FIG. 22 are denoted by the same reference numerals, and detailed description thereof will be omitted. To do. In this embodiment also, the inverter section 4 is shown as an example of the load on the DC side.
Embodiments of the present invention are not limited to inverters, and can be implemented with any DC load.

Reference numeral 14 denotes a voltage detecting device, which is composed of photo couplers 14a to 14f which are connected by means of Glens and a current suppressing resistor 14.
and the voltage ER, E of each phase of the three-phase AC power supply 1
It functions to detect the maximum voltage detection signals 15a to 15f from S and ET. Here, the maximum voltage detection signals 15a to 15
f becomes level 1 when the photocouplers 14a to 14f are conductive, and level 0 when they are opened. Reference numeral 16 is a minimum voltage detection circuit, which is composed of a combination of the logical product circuits as shown in the drawing, and serves to output the minimum voltage detection signals 17a to 17f.

Reference numeral 18 denotes an AC voltage control circuit, which functions to output PWM ignition commands 19a to 19f, which will be described in detail later. Reference numeral 20 denotes a driver circuit, which is a PWM ignition command 19a-
19f and an OR circuit and an amplifier to which the maximum voltage detection signals 15a to 15f are input, and an ignition command 21a for each switching element 2a to 2f of the converter unit 2.
It works to output ~ 21f.

Here, as an embodiment of the present invention, self-extinguishing type switching elements (circuits) 2a to 2f and diodes
(Although it may be a diode circuit) 2g to 2l are shown as examples in which independent elements or circuits are connected in antiparallel, but there are various other examples, such as a self-extinguishing type switching circuit and an antiparallel diode. The circuit is provided on the same wafer,
For example, a GTO thyristor with an anti-parallel diode connected by a Glenz connection has the same operation. In addition, this is the same in the following embodiments.

Next, referring to FIG. 2, the AC voltage control circuit 18
Will be described in detail. First, the minimum voltage detection signal 17
a to 17f are input to the rising edge detection circuits 18a to 18f, pulse output signals representing the rising edges of the respective pulses are generated, and they are supplied to the logical sum circuit 18g. Next, the output of the logical sum circuit 18a is input to the counter 18h, the phase signal θ is output, and the function generators 18i and 18 are output.
supply to j.

FIG. 3 shows the operation of the counter 18h. As shown in the figure, the counter 18h is reset each time an input pulse appears, and at the same time, the counter 18h is restarted to start counting up, and the phase signal .theta. To occur. At this time, as described above, in this embodiment, θmax
Is set to a phase equivalent to 60 °, so the counter 1
The output of 8h is limited so as not to exceed the set value θmax.

Therefore, these function generators 18i, 18
h outputs the conduction ratio according to this phase signal θ, but at this time, as shown in the figure, in the function generator 18i, the conduction ratio increases from 0 according to the phase signal θ, and the input phase 60 When the phase signal θ is 0, the conduction ratio of 1 is generated in the function generator 18j.
Then, a conduction ratio having a characteristic that decreases from that and saturates to 0 at a phase of 60 ° is generated.

For this reason, these two function generators 18
The sum of the outputs of i and 18j is almost 1 at all times, so that the conduction ratio approaches the value of equation (2), and the line approximation is set in consideration of the dwell period required for the switching element. It will be.

On the other hand, two carrier wave generators 18m and 18n which operate in synchronization with the output of the clock generator 18k are provided, and a triangular wave carrier wave signal is generated from these, and these carrier wave signals and functions are generated. Generator 18i, 1
The output of 8j is input to the comparators 18p and 18q.

Therefore, these comparators 18p, 18q
Is the ignition command 19g, 19h as a result of comparison between them.
Are output, and these are input to the analog switch 18r which is opened / closed by the minimum voltage detection signals 17a to 17f.
FIG. 4 shows the operation of these comparators 18p and 18q. As shown in the figure, the ignition command is output by comparing the conduction ratio command and the carrier wave.

The analog switch 18r is composed of a plurality of switching contact type switching elements, and the minimum voltage detection signal 1
When 7a to 17f are at level 1, they switch as shown to output the ignition command signals 19g and 19h, and at level 0, they switch to the opposite of those shown and output a 0 signal (common potential signal). As a result, the output from the OR circuit 18s that receives the output signal of the analog switch 18r is
PWM ignition command 19a to each switching element (circuit)
19f, and the control shown in FIG. 32 is obtained. Here, the polygonal line approximation in the function generator may be omitted and only the limiter with the upper limit 1 and the lower limit 0 may be omitted. However, according to this embodiment, the effect of suppressing harmonics is large. There is an advantage.

Next, FIG. 5 shows another embodiment of the present invention.
In this figure, 22 is a voltage phase detection device, 23 is an external setting device, and the voltage phase detection device 22 functions to output the voltage phase signal θ corrected by the external setting device 23. Reference numeral 24 is a function generator for calculating the conduction ratio, which functions to output the conduction ratio commands 25a to 25c for each phase of the RST.

Reference numerals 26a to 26d are comparators, which compare the magnitude of the carrier wave signal from the carrier wave generator 27 and the duty ratio command,
Each of the RSTs functions to generate an ignition command to a switching element (circuit) connected to each phase. The outputs of these comparators 26a to 26d are supplied to an ignition command 2 via an amplifier 28.
9a to 29f, each switching element (circuit) 2a
~ 2f.

The other structure is the same as that of the embodiment shown in FIG.

An alternating current ammeter 30 is used when the phase correction value α is adjusted by the external setting device 23. At this time, the phase correction value α is a force with which the current amplitude at the converter rated load is minimum. The rate is set to 1.

FIG. 6 shows an example of the configuration of the voltage phase detector 22. In FIG. 6, a low-pass filter (low-frequency filter) 22a is provided for the purpose of removing harmonic components of a voltage signal. Further, in this embodiment, the low-pass filter 22a is set so that the output signal is delayed by about 30 ° when the power supply frequency signal is input, whereby the output signal becomes substantially in phase with the ER phase voltage. Reference numeral 22b is a comparator, which inputs the output signal of the low-pass filter 22a and outputs a signal which becomes level 1 when the signal is positive and level 0 when the signal is negative.

22c is a rising edge detection pulse generator,
22d is a pulse shift circuit, and in this embodiment,
The output pulse of the rising edge detection pulse generator 22c is 3
Assuming that the phase order of the phase AC power supply 1 is ER → ES → ET, the three phases are balanced, and the frequency is constant, the phase at the time of the positive peak of the phase voltage of the ER phase is θ = 0, and further external It is output via the pulse shift circuit 22d that shifts the pulse by the phase correction value α manually set by the setter 23.

Reference numeral 22e is a counter, which is a pulse shift circuit 2
It operates by using the pulse output from 2d as an input and outputs the voltage phase signal θ. Therefore, the operation of the counter 22e is the same as that of the counter 18h in the embodiment of FIG. Then, θmax is the phase 36
It is set to 0 °.

FIG. 7 shows each part of the voltage phase detector 22.
FIG. 8 shows an example of setting of the function generator 24. In this embodiment, the characteristic of FIG. 31 is approximated by a polygonal line, and the S-phase conduction coefficient command is approximated by a polygonal line passing through points A and B. As a result, at θ = 30 °, a C point with a conduction rate of 0.5 is obtained. Pass through.

FIG. 9 shows the operation of the comparators 24d to 24f. The ignition command is output by comparing the conduction ratio command and the carrier wave. Then, a rest period TD for preventing an arm short circuit is set between the ignition command and the ignition command.

According to this embodiment, since the harmonic components are removed by the low-pass filter 22a, the detection of the rising edge of the pulse is disturbed by the voltage pulsation appearing due to the influence of the power source impedance of the three-phase AC power source 1, and the counter is detected. Although the malfunction appears, it is suppressed so that stable control can be obtained. Further, since the impedance drop caused by the AC reactor is phase-compensated and the driving power factor can be maintained at 1 near the rated output, the input current can be suppressed.

Next, FIG. 10 is also an embodiment of the present invention, in which 31
Is a voltage phase detector using a digital arithmetic unit, and has a function of outputting the voltage phases θR, θS, and θT of each phase. As a concrete calculation method at this time, a method using a moving Fourier transform is known, for example, Japanese Patent Laid-Open No. 1-231.
No. 682 and Japanese Patent Application Laid-Open No. 1-283015 disclose voltage phase calculation methods that take into account malfunctions inside the calculation device, and these may be used.

The phase detection signals representing the voltage phases θR, θS, θT of the respective phases are corrected by the adders 32a to 32c by the correction value α output from the external setter 23. 33
a to 33c are function generators for calculating the conduction ratio, and the function generator 33a for the R phase has the same configuration as the function generator 24 in the embodiment of FIG. 5, that is, the R phase conduction ratio setting function of FIG. It has been used. As for the S phase and the T phase, the phase signal input to the function generators 33b and 33c is a signal at which θ = 0 at a positive peak value as in the R phase, so the function generator has the same setting as the R phase. good.

The other structure is the same as that of the embodiment shown in FIG.

According to this embodiment, since the conduction ratio is controlled in each phase according to the individual voltage phase, the converter input current is disturbed even when a reverse phase voltage is generated due to an abnormality on the AC power supply side. Can be suppressed.

Next, FIG. 11 is a specific example of the above-described dodecagonal basic type embodiment. In the example of FIG. 11, in FIG.
Similar to the embodiment of FIG. 0, the case where the digital arithmetic unit 31 is used is shown. The difference from the embodiment of FIG.
The other parts are the same except that the function generators 33a to 33c for calculating the conduction ratio are changed to the function generators 34a to 34c.

FIG. 12 shows details of setting characteristics in the function generators 34a to 34c, which are the same for each phase. Here, the difference between this embodiment and the setting of the R phase in FIG. 8 is that the phase θ is 0.
The settings are based on the same polygonal line approximation in the other sections only in the sections of 30 degrees centering on ° and 180 degrees.

In this embodiment, from the value obtained by substituting θ = 0 ° into the above equation (6), first, the value at point A is reduced to 0.866, and it is linearly changed to 1 before and after. Then, at point B, the value is increased to 0.134, and the value also linearly changes to 1 before and after. Therefore, according to the embodiment of FIG. 11, it is possible to suppress the fifth and seventh harmonics while obtaining the same effect as that of the embodiment of FIG.

In the embodiment shown in FIG. 11, FIG.
Similar to the above, the case where the digital arithmetic unit 31 is used is shown. However, like the embodiment of FIG.
In this case, the function setters 34a to 34c output the pattern of each phase by 120 °.

Next, FIG. 13 shows an embodiment of the present invention in which the driving power factor is automatically controlled. In the figure, reference numeral 35 is a voltage-current transformer, which serves to detect an AC waveform signal. Next, 36 is a reactive power detector, which detects reactive power from the AC waveform signal input from the voltage-current transformer 35 and generates a detection value Qm. 37
Is the reactive power control circuit, and the reactive power detection value Qm is the command value Q
The phase correction command α is output so as to match ref.

Then, this phase correction command α is supplied to the adders 32a to 32c to correct the phase signal θ of each phase, as in the embodiment of FIG. , Which is the same as the embodiment of FIG.

FIG. 14 shows an example of the reactive power control circuit 37.
The difference between the command value Qref and the reactive power detection value Qm is calculated by the amplifier 38
And the upper and lower limit output is input to a proportional integration circuit including an integrator 39 with a limiter, and a phase correction command α is output via a limiter circuit 40. Where usually
The power factor is set to 1 by setting the command value Qref to 0 and operating.
Therefore, according to the embodiment of FIG. 13, control is always performed so that the driving power factor becomes the command value Qref, and as a result, when the DC load 4 suddenly changes, and further, the converter unit 2 The driving power factor can be maintained at 1 even when the regenerative driving is performed.

Next, FIG. 15 shows a concrete example of the control circuit of the inverter section 4 in the embodiment of FIG. 13, and the configuration of the inverter section 4 is the same as that of the prior art shown in FIG. Is. A current transformer 41 is provided on the AC side of the inverter unit 4, and the measured current signal of the AC motor 5 is supplied to the current control circuit 42. Further, a speed detector 43 directly connected to the AC motor 5 is provided, and the speed control circuit 4
At 4, the current command 45 is output and is also supplied to the current control circuit 42. As a result, the current control circuit 42 causes the current command 4
Modulation rate commands γU, γV,
Output γW to the sine wave PWM control signal generator.

In this case, since the current is controlled on the side of the inverter unit 4, the converter unit 2 and the smoothing capacitor 3 have variable DC current source characteristics on the side of the inverter.
In this embodiment, however, the converter unit 2 does not fluctuate its output DC voltage even when the DC current on the inverter side fluctuates, so that the output of the AC motor 5 has good responsiveness. Can be controlled.

Next, FIG. 16 shows an embodiment of the present invention having a voltage protection function. In the figure, reference numeral 46 is an AC voltage detector, which, like the voltage phase detector 31, uses a moving Fourier transform. It functions to output the positive phase voltage amplitude VP and the negative phase voltage amplitude VN of the AC power supply 1. Reference numeral 47 is an AC voltage protection circuit that controls the control signal C according to the voltage amplitude signals VP and VN.
It functions to output an opening / closing command Mg to TR and the electromagnetic switch 49. The electromagnetic switch 49 is provided with an auxiliary contact so that a signal for confirming the opening / closing operation can be obtained. Reference numeral 48 is an OR circuit, which functions to prevent the outputs of the comparators 26a to 26d from being supplied to the amplifier 28 when the control command signal CTR is at level 0. The rest of the configuration is the same as that of the embodiment shown in FIG.

FIG. 17 shows a voltage comparison characteristic in the AC voltage protection circuit 47. When the positive phase voltage amplitude VP becomes equal to or lower than the set value VP1, the low voltage detection signal VP59 becomes level 1. In this embodiment, as shown in the figure, the hysteresis characteristic is given, and the low voltage detection signal VP59 does not return to level 0 unless the positive phase voltage amplitude VP becomes larger than the set value VP2. ing. Further, the anti-phase voltage detection signal VN59 becomes level 1 when the anti-phase voltage amplitude VN becomes larger than the set value VN1,
When it becomes smaller than the set value VN2, the level is returned to 0.

FIG. 18 shows a concrete example of the AC voltage protection circuit 47, which is composed of a relay protection interlocking circuit. Then, first, the relay V30 detects the voltage amplitude signals VP59 and VN.
It is excited when 59 is at level 1. In addition, relay R1
Is the start command STR and stop command ST from the manual switch.
It is controlled by P, and its contact controls the opening / closing command Mg to the electromagnetic switch 49 and the timer T1. Therefore, the relay R1 closes the electromagnetic switch 49, and after the timer T1 secures the time for the smoothing capacitor 3 to be charged by the diode circuit of the converter unit 2, the control command signal CTR becomes level 1 by the closing operation of the contact T1. As a result, the blocking of the ignition command to the switching element (circuit) is released, the converter operation is started, and the normal operation state is entered.

When the voltage is abnormal, the control command signal CTR becomes level 0 by the b contact operation (opening operation of the normally closed contact) of the relay V30, and the ignition command is stopped, but at the same time when the V30 returns to level 0, the CTR is restarted. Level 1 On the other hand, the timer T2 is configured to excite the relay V86 and activate the emergency stop signal X86 to stop the operation when the voltage abnormality detected by the relay V30 continues longer than the set time value. Here, the contact Z86 is controlled to open / close by another emergency stop protection device (not shown), for example, an overcurrent protection relay.

Next, the operation of the AC voltage protection circuit 47 will be described with reference to the timing chart of FIG. Now
If the start command STR is energized at time t1, the timer T1 starts to operate, and the electromagnetic switch 49 closes after an operation delay time. When the timer T1 is closed at time t2, the operation command signal CTR becomes level 1 and the ignition control of the switching element (circuit) is started. If the stop command STP is activated at time t3,
Here, it returns to the stopped state. The above is the normal operation stop operation.

Next, at this time, after the STP is energized again at time t4 and the CTR becomes level 1 at time t5 and the operation is started, the abnormal voltage signal V30 is detected during the period from time t6 to time t7. To do. Then, this time t6
During a period from t7 to t7, the CTR becomes level 0 and the ignition of the switching element (circuit) is stopped, but the electromagnetic switch 49 remains closed, so the diode 2 of the converter unit 2
The operation is continued by the rectifying operation by 2a (2l).

If the voltage abnormality is detected at time t8 and the abnormality continues until the set time value of the timer T2, V86 is excited at time t9, which further excites X86 and the timer T1. It is opened. As a result, this time the electromagnetic switch 49 is opened with a delay time, the auxiliary contact x Mg is opened, the excitation of the timer T2 is released, and the emergency stop operation is started. During this time, the excitation of V86 is maintained, but after that, since RST is activated at time t10, it is reset.

According to this embodiment, it is possible to obtain the operation of shifting to the operation by the diode bridge before the overcurrent flows through the converter section 2 due to the abnormality of the AC voltage, and as a result, the switching that constitutes the converter section 2 is performed. element
(Circuit) The capacities of 2a to 2f can be reduced.

Next, FIG. 20 shows an embodiment of the present invention having a current protection function. In the figure, reference numerals 50 and 51 denote AC overcurrent detectors, which are output signals CR5 that become level 1 when the absolute values of the AC currents IR and IS exceed set values.
1. It works to generate CS51. Next, 51 is an alternating current protection circuit, which is an overcurrent detection output signal CR51, CS5.
In response to 1, the control command signal CTR and the switching command Mg to the electromagnetic switch 49 are output. The rest of the configuration is the same as that of the embodiment shown in FIG.

FIG. 21 shows a concrete example of the AC current protection circuit 52, which is a VP in the AC voltage protection circuit 47 shown in FIG.
59 and VN59 are replaced with CR51 and CS51, respectively. Here, when CR51 and CS51 are at level 1, the relay C30 is excited. When the current is abnormal, the control command signal CTR becomes level 0 and the ignition command is stopped by the contact operation of the relay C30, but the relay C30 returns to level 0 and the CTR becomes level 1 again.

On the other hand, when the current abnormality detected by C30 continues beyond the set time value, the timer T2 relays C
86 is excited, the emergency stop signal X86 is operated, and the operation is stopped. Here, Z86 is controlled to be opened / closed by another emergency stop protection device (not shown). Therefore, according to the embodiment of FIG. 20, even when the alternating current is in the overcurrent state, it is detected and the switching operation of the converter unit 2 is automatically stopped. Therefore, the switching element (circuit) 2a It is possible to reliably protect ~ 2f.

In the above embodiments, the self-extinguishing type switching elements (circuits) 2a to 2f and the diodes (or diode circuits may be used) 2g to 2l are shown as examples in which independent elements or circuits are connected in antiparallel. However, as described above, there are various other examples. For example, a GTO thyristor with an anti-parallel diode, in which a self-extinguishing type switching circuit and an anti-parallel diode circuit are provided on the same wafer, is used. Therefore, it goes without saying that the present invention may also be implemented using these GTO thyristors with anti-parallel diodes, for example.

[0095]

According to the present invention, continuous operation can be maintained even when the power source or load fluctuates without controlling the AC current and the DC voltage. Therefore, the structure can be simplified and the pulse width is highly reliable. A modulation type power conversion device can be easily provided. Further, according to the present invention, since the switching frequency of the switching element (circuit) can be reduced, the switching loss can be reduced, and a highly efficient pulse width modulation power conversion device can be easily provided.

[Brief description of drawings]

FIG. 1 shows a first pulse width modulation power converter according to the present invention.
3 is a block circuit diagram showing an embodiment of FIG.

FIG. 2 is a circuit diagram showing a specific example of an AC voltage control circuit according to the first embodiment.

FIG. 3 is an explanatory diagram of an operation of a counter in the first embodiment.

FIG. 4 is an operation explanatory diagram of the comparator in the first embodiment.

FIG. 5 is a second pulse width modulation power converter according to the present invention.
3 is a block circuit diagram showing an embodiment of FIG.

FIG. 6 is a block diagram showing a specific example of a voltage phase detection device according to a second embodiment.

FIG. 7 is an operation explanatory diagram of the voltage phase detection device according to the second embodiment.

FIG. 8 is an explanatory diagram of setting characteristics of the function generator according to the second embodiment.

FIG. 9 is an operation explanatory diagram of a comparator in the second embodiment.

FIG. 10 is a block circuit diagram showing a third embodiment of the pulse width modulation power converter according to the present invention.

FIG. 11 is a block circuit diagram showing a fourth embodiment of the pulse width modulation power converter according to the present invention.

FIG. 12 is an explanatory diagram of setting characteristics of the function generator according to the fourth embodiment.

FIG. 13 is a block circuit diagram showing a fifth embodiment of the pulse width modulation power converter according to the present invention.

FIG. 14 is a block diagram showing a specific example of a reactive power control circuit according to a fifth embodiment.

FIG. 15 is a block circuit diagram showing a sixth embodiment of the pulse width modulation power converter according to the present invention.

FIG. 16 is a block circuit diagram showing a seventh embodiment of the pulse width modulation power converter according to the present invention.

FIG. 17 is an explanatory diagram showing a characteristic example of the voltage comparator according to the seventh embodiment.

FIG. 18 is a block diagram showing a specific example of an AC voltage protection circuit according to a seventh embodiment.

FIG. 19 is an operation explanatory diagram of the AC voltage protection circuit according to the seventh embodiment.

FIG. 20 is a block circuit diagram showing an eighth embodiment of the pulse width modulation power conversion device according to the present invention.

FIG. 21 is an operation explanatory diagram of the AC voltage protection circuit according to the eighth embodiment.

FIG. 22 is a block circuit diagram showing an example of a pulse width modulation power conversion device according to a conventional technique.

FIG. 23 is an operation control diagram of a sine wave pulse width modulation control signal generator in the related art.

FIG. 24 is a vector diagram of an alternating voltage and current in the related art.

FIG. 25 is a circuit diagram showing an example of a diode bridge circuit.

FIG. 26 is a waveform diagram of a diode bridge circuit.

FIG. 27 is a locus diagram of an alternating current of a diode bridge circuit.

FIG. 28 is a locus diagram of an AC voltage of a diode bridge circuit.

FIG. 29 is a circuit diagram defining an operation mode of a switching element (circuit) in the pulse width modulation power conversion device.

FIG. 30 is a coordinate conversion diagram for explaining the operation of the pulse width modulation power conversion device according to the present invention.

FIG. 31 is a characteristic diagram showing an example of switching control in the pulse width modulation power converter according to the present invention.

FIG. 32 is a characteristic diagram showing an example of voltage control in the pulse width modulation power converter of the present invention.

FIG. 33 is a coordinate conversion diagram for explaining the operation of the pulse width modulation power conversion device according to the present invention.

FIG. 34 is a characteristic diagram showing another example of switching control by the pulse width modulation power converter of the present invention.

FIG. 35 is a characteristic diagram showing another example of voltage control in the pulse width modulation power converter of the present invention.

[Explanation of symbols]

1 3-phase AC power supply 2 Converter section 3 smoothing capacitors 4 Inverter section 5 AC motor 6 AC reactor 14 Voltage detector 16 Minimum voltage detection circuit 18 AC voltage control circuit 20 driver circuit

─────────────────────────────────────────────────── ─── Continuation of the front page (72) Inventor Tsunehiro Endo 7-1 Higashi Narashino, Narashino City, Chiba Prefecture Hitachi Industrial Co., Ltd. Industrial Equipment Division (72) Seiji Ishida 7-1 Higashi Narashino City, Narashino City, Chiba Prefecture No. 1 Hitachi Industrial Co., Ltd. Industrial Equipment Division (72) Inventor Hiroshi Fujii 7-1-1 Higashi Narashino, Narashino City, Chiba Prefecture Hitachi Industrial Co., Ltd. Industrial Equipment Division (72) Inventor Kazuhiro Imaya Saiwaicho, Hitachi City, Ibaraki Prefecture 3-1-1, Hitachi Ltd., Hitachi factory (72) Inventor, Yasunori Nomoto 5-2-1 Omika-cho, Hitachi-shi, Ibaraki Hitachi Ltd. Omika factory, Hitachi (72) Inventor Kiichi Tokunaga Hitachi, Ibaraki prefecture 7-1-1 Omika-cho, Oita-shi Hitachi Ltd. Hitachi Research Laboratory (72) Inventor Tanaka Main tax 1-1-1 Kokubun-cho, Hitachi-shi, Ibaraki Company, Kokubun factory, Hitachi, Ltd. (72) Inventor Katsuya Furukawa 3-1-1, Saicho-cho, Hitachi, Hitachi, Ibaraki (72) Invention, Hitachi factory Person Takashi Hotta 7-1, 1-1 Omika-cho, Hitachi-shi, Ibaraki Hitachi, Ltd. Hitachi Research Laboratory (72) Inventor Keijiro Sakai 7-1-1, Omika-cho, Hitachi-shi, Ibaraki Hitachi Ltd. (72) ) Inventor Katsuhiro Okusawa 7-1, 1-1 Omika-cho, Hitachi-shi, Ibaraki, Hitachi Research Laboratory, Hitachi, Ltd. (56) Reference JP-A-60-9384 (JP, A) JP-A-62-104481 (JP, A) ) JP-A-8-251947 (JP, A) (58) Fields investigated (Int.Cl. ⁷ , DB name) H02M 7/219 H02M 7/06 H02M 7/155 H02M 7/48 H02P 7/63

Claims (7)

(57) [Claims] 1. A pulse width modulation power conversion device having a self-extinguishing type switching circuit and a diode circuit connected in anti-parallel between a three-phase AC power supply and a DC smoothing circuit having a capacitor as an arm. For the arm connected to the phase of the phase AC power supply in which the amplitude of the phase voltage is minimized, the switching circuit is subjected to pulse width modulation control. A pulse width modulation power conversion device characterized in that the switching circuit is controlled to flow through a diode circuit and during reverse conversion operation. 2. A pulse width modulation power converter using a self-extinguishing type switching circuit and a diode circuit connected in anti-parallel with each other between a three-phase AC power supply and a DC smoothing circuit having a capacitor. The phase with the smallest amplitude of the phase voltage of the phase AC power supply is the first
Minimum voltage detecting device for detecting the phase voltage phase, and the arm connected to the positive side of the capacitor and the negative side of the two arms connected to the phase selected by the minimum voltage detecting device. The self-extinguishing type switching circuit of the arm connected to is alternately ignited and extinguished to adjust the conduction ratio of the arm according to the command value, and the remaining two phases are connected to the positive voltage phase.
Keeping the conduction ratio of the positive arm of the arms to 1 and the conduction ratio of the negative arm of the two arms connected to the negative voltage phase to 1
And a gate control device for keeping the phase voltage phase of the first phase as an input, and during the period in which the voltage of the first phase changes from negative to positive, the positive arm current flows according to the detected phase voltage phase. An AC voltage control device is provided that continuously changes the rate command value from 0 to 1 and continuously changes the current flow rate command value of the negative arm from 0 to 1 during the period from positive to negative. And a pulse width modulation power converter. 3. A pulse width modulation power conversion device having a self-extinguishing type switching circuit and a diode circuit connected in anti-parallel between a three-phase AC power supply and a DC smoothing circuit having a capacitor as an arm. The phase with the smallest amplitude of the phase voltage of the phase AC power supply is the first
Minimum voltage detecting device for detecting the phase voltage phase, and the arm connected to the positive side of the capacitor and the negative side of the two arms connected to the phase selected by the minimum voltage detecting device. The self-extinguishing type switching circuit of the arm connected to is alternately ignited and extinguished, and in the period in which the phase voltage of the first phase changes from negative to positive, the positive side is detected in accordance with the detected phase voltage phase. The flow rate of the arm is continuously changed from 0 to 1,
During the period in which the phase voltage of the first phase changes from positive to negative, the conduction ratio of the negative arm is 0 according to the detected phase voltage phase.
AC voltage control device for continuously variably adjusting from 1 to 1 and a positive voltage phase of the remaining two phases of the three-phase AC power supply is selected as the second phase, and a positive phase voltage phase is detected. A phase in which the voltage of the second phase is maximized by alternately igniting and extinguishing the voltage detection device and the self-extinguishing type switching circuit of the positive arm and the negative arm connected to the second phase. During a predetermined positive side partial flow period centered on θ2, the negative side arm is partially ignited to continuously variably adjust the flow rate of the positive side arm from 1 to the minimum set value at θ2. A second AC voltage control device that keeps the conduction ratio of the positive arm at 1 except the positive side partial flow period, and a negative voltage phase that is the remaining phase of the three-phase AC power supply as a third phase, A negative voltage detecting device for detecting the phase voltage phase, and a positive arm and a negative arm connected to the third phase. The self-extinguishing type switching circuit is alternately ignited and extinguished, and the positive side arm is partially operated during a predetermined negative side partial conduction period centered on the phase θ3 at which the voltage of the third phase becomes maximum on the negative side. And then continuously variably adjust the duty ratio of the negative arm from 1 to the minimum set value in θ3, and keep the duty ratio of the negative arm at 1 except during the partial partial conduction period. A pulse width modulation power conversion device comprising a third AC voltage control device. 4. The power factor detection device for detecting the power factor of the three-phase AC power supply, the power factor detection value by the power factor detection device, and a predetermined power factor setting according to claim 2 or claim 3. And a power factor adjusting device for adjusting the voltage phase input to the AC voltage control device according to the deviation from the value. 5. The invention according to any one of claims 1 to 3, wherein a voltage type inverter circuit connected to the DC smoothing circuit and an AC output of the inverter circuit are matched with a variable frequency / variable amplitude command. A pulse width modulation power conversion device comprising: a pulse width modulation control device. 6. The AC voltage detecting device according to claim 2 or 3, wherein the voltage abnormal signal is detected by detecting that the voltage of the three-phase AC power supply exceeds a set range, and the voltage abnormal signal. According to the above, an overvoltage protection device is provided which blocks the ignition command from the AC voltage control device to the self-extinguishing type switching circuits of all arms and releases the ignition command block after the abnormal voltage output is stopped. And a pulse width modulation power converter. 7. The invention according to claim 1 or 3, wherein an alternating current detection device for detecting that the current of the three-phase alternating current power supply exceeds a set range and generating an abnormal current signal; and the abnormal current signal. According to the above, there is provided an overcurrent protection device for blocking an ignition command from the alternating current control device to the self-extinguishing type switching circuits of all arms, and for canceling the ignition command blocking after the abnormal voltage output is stopped. And a pulse width modulation power converter.