JP3387606B2 - Propagation path estimation device and mobile communication receiving device - Google Patents

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a propagation path estimating device and a receiving device for a mobile communication system.

[0002]

2. Description of the Related Art A mobile communication system has been studied in which transmission data is differentially encoded on the transmitting side, and then directly spread with a predetermined spreading code sequence (for example, PN sequence), PSK-modulated and transmitted. As a receiving device for such a mobile communication system, there has been proposed a device for demodulating transmission data by applying adaptive RAKE (Kumate) combining described in the following document.

Reference “Veli-Pekka Kaasila and Aarne Mam
mela, ”THE ADAPTIVE RAKE MATCHED FILTER IN A TIME
-VARIANT TWO-PATH CHANNEL ”, IEEE PIMRC'92, pp.4
41-445, October 1992] Here, the adaptive RAKE combining means a plurality of receptions by despreading a baseband reception signal that has undergone frequency selective fading due to movement of the reception device or the transmission device. Wave diversity (generally, the earliest one is called the leading wave, and the others are all called the delayed waves), and the leading and delayed waves are weighted and combined according to their reliability to realize path diversity. To do. When the weighting parameter can be accurately estimated, the maximum ratio combining diversity is obtained, and the S / N ratio can be maximized.

The above-mentioned document describes that the complex conjugate of the fading propagation path characteristic β · exp (jφ) having the amplitude characteristic β and the phase characteristic φ is used as the weighting parameter. It is described that the fading propagation path characteristics can be estimated by time averaging processing (moving averaging processing) of the signal by removing the data modulation component by multiplying the determination data d by the complex correlation signal R obtained by despreading. Has been done.

[0005]

However, in the conventional mobile communication receiver, in estimating the channel characteristics,
It introduces the assumptions that "the amplitude characteristic and the phase characteristic of the fading propagation path change slowly and are constant when a short time section is considered", and "the noise is a random value with an average value of 0". That is, as described above, the channel characteristic is estimated by the time averaging process of the signal including the fading component from which the data modulation component is removed and the noise component.

Therefore, if the assumption is not valid because the change of the propagation path characteristics is rapid such as when the moving body moves at high speed in the city, the propagation path characteristics cannot be accurately estimated and the data It may happen that the judgment is not accurately made. In other words, the time period in which the above assumption is considered to work effectively differs depending on the speed of change in the channel characteristics (fading speed), but since the processing is not switched, the channel characteristics are not changed. May not be accurately estimated, and the data may not be accurately determined. as a result,
There is a demand for a mobile communication receiver capable of accurately determining data.

By the way, various adaptive channel estimators with high estimation accuracy have been proposed. However, in a mobile communication receiver having the characteristics that the number of received waves involved in propagation path estimation (the number of propagation path estimators) is large, there is a strong demand for small size, light weight, low power consumption, etc. It is preferable to apply an adaptive channel estimator if the channel estimator can obtain sufficient estimation accuracy.

Therefore, there is a demand for a channel estimation apparatus with a fixed coefficient that can accurately estimate channel characteristics.

[0009]

In order to solve the above problems, according to the first aspect of the present invention, (1) a data series that reflects only the propagation path characteristics from which the data modulation component has been removed is input, and the data A Doppler frequency estimator that detects the Doppler frequency that the sequence is receiving, and (2) the above-mentioned data sequence is input to obtain an estimated value of its channel characteristics, and depending on the detected Doppler frequency, Change and use the number of data used to obtain the estimated value of the channel characteristics.
Is this the reciprocal of the number of data for each of the data series?
A propagation path estimator is configured by a propagation path estimator that multiplies the coefficients of ## EQU1 ## and sums the multiplied values .

A second aspect of the present invention relates to a mobile communication receiving apparatus in which a transmission side directly spreads transmission data with a spreading code sequence, digital phase modulates and gives a transmitted signal.

The mobile communication receiving apparatus according to the second aspect of the present invention is
(1) Correlation calculation between the spread code sequence synchronized with any of the received waves and the baseband received signal, which is provided corresponding to each of the plurality of received waves received through the plurality of radio propagation paths A plurality of correlation calculation means for outputting the correlation calculation signal, and (2) a correlation calculation signal provided from the corresponding correlation calculation means based on the judgment data, which is provided corresponding to each received wave. A plurality of data modulation component removing means for removing the data modulation component from (3) each corresponding to each received wave, the data modulation component output from the corresponding data modulation component removing means is removed. A plurality of Doppler frequency estimating means for detecting the Doppler frequency based on the signal obtained, and (4) the data provided from the corresponding data modulation component removing means provided corresponding to each received wave. Strange Based on the signal from which the tonal component has been removed, the propagation path characteristics of the radio propagation path are estimated, and a signal for weighting is output, and the propagation is performed according to the detected Doppler frequency from the corresponding Doppler frequency estimation means. change the number of data signals the data modulation components have been removed for use in estimating the road, this for each of the data sequences used
Multiply the coefficient consisting of the reciprocal value of the number of
A plurality of radio channel estimation means for summing each value ,
(5) Weighting means provided corresponding to each received wave, and weighting the correlation calculation signal from the corresponding correlation calculation means based on the signal from the corresponding radio channel estimation means, and ( 6) Combining means for adding and combining the outputs of all the weighting means, and (7) judging means for judging data based on the outputs of the combining means.

[0012]

The channel estimating apparatus of the first aspect of the present invention does not adaptively change the parameters, but the parameters are fixed.

Generally, in a channel estimation device with fixed parameters, the parameters are fixed, so some assumption is introduced to determine the parameters so as to best meet various channel characteristics. There is a limit. Especially, when the fading speed is high, the estimation tends to be bad. Therefore, the accuracy can be improved if the estimation according to the fading speed can be performed even if it is not a completely adaptive type. Although the fading speed and the Doppler frequency are closely related to each other, the Doppler frequency is easier than the fading speed in the detection configuration and can be detected at high speed.

Further, if the number of data used for estimating the propagation path characteristics is large, the time variation of fading cannot be followed, and if the number is small, the noise component removing ability is deteriorated.
The optimum number depends on the fading speed. Therefore, if the estimation is performed with a uniform number of data, the accuracy may be poor.

The first aspect of the present invention is based on the above idea and attempts to improve the estimation accuracy by detecting the Doppler frequency and switching the number of data used for estimation of the propagation path characteristic in accordance with the detection. Is.

The second aspect of the present invention has the features of the first aspect of the present invention.
The present invention is applied to a mobile communication receiving device applying adaptive RAKE combining in which the estimation accuracy of channel characteristics affects the accuracy of decision data, and thereby the decision data of decision data can be obtained without complicating the configuration. This is an improvement in accuracy.

[0017]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS An embodiment of the present invention will be described in detail below with reference to the drawings. In the mobile communication receiving apparatus of this embodiment, the transmission side differentially encodes the transmission data, and then directly spreads it with a predetermined spreading code series (for example, PN series) to perform two-phase PSK.
The modulated and transmitted signal is received and data demodulated, and the data is demodulated from a total of two waves, a preceding wave and a delayed wave.

(A) Data Demodulation Principle Applied to the Embodiment In mobile communication, fading occurs with the movement of the receiving apparatus or the partner transmitting apparatus. Therefore, a complex correlation signal (a signal after despreading processing) in which the correlation between the in-phase component of the baseband received signal and the spreading code sequence is the real part and the correlation between the orthogonal component of the baseband received signal and the spreading code sequence is the imaginary part Considering the above, the signal R1 for the preceding wave and the signal R2 for the delayed wave are those in which the fading channel characteristics are affected as shown in equations (1) and (2), respectively.

R1 = β1dexp (jφ1) + N1 (1) R2 = β2dexp (jφ2) + N2 (2) where β1, φ1 and N1 are fading of the preceding wave propagation path, respectively. Amplitude characteristics, fading phase characteristics, noise, and β2, φ2, and N2 are fading amplitude characteristics, fading phase characteristics, and fading amplitude characteristics of the delayed wave propagation path, respectively.
It is noise. Also, d is transmission data that takes +1 or -1. Furthermore, exp (jφ) is, as is well known,
It is an expression of cos φ + j sin φ in exponential notation.

By multiplying each of the complex correlation signals R1 and R2 as described above by the judgment data d, the data modulation component can be removed. That is, the component of the determination data d that takes -1 or 1 is squared to become 1, and the data modulation component is removed. Fading characteristics β 1 · exp (jφ in each propagation path from the signal from which the data modulation component is removed
1), β 2 · exp (jφ 2) is estimated based on the following assumptions.

That is, "fading channel fading amplitude characteristic β (β1 or β2) and fading phase characteristic φ (φ
1 or φ 2) changes over time, considering a short time interval,
It is based on the assumption that it is “constant” and “noise is a random value with an average value of 0”.

Then, the fading channel characteristic β1.exp (j.phi.1) + d.N1 from which the data modulation component is removed
, Β2 · exp (jφ2) + d · N2 are time averaged (moving average) to remove noise components and fading channel characteristics β1 · exp (jφ1), β2 · exp (jφ2
) Is estimated.

As described above, fading characteristics β 1 .exp (jφ 1) and β 2 .exp (jφ) in each propagation path
2) is obtained, each complex conjugate β 1 · exp (-
j.phi.1), .beta.2 .multidot.exp (-j.phi.2) and the corresponding complex correlation signals R1 and R2 are multiplied, and the two multiplication results are added to obtain the weighted combined value C shown in equation (3).

C = R1 .beta.1 .exp (-j.phi.1) + R2 .beta.2 .exp (-j.phi.2) = .beta.1 {.beta.1 .d + N1 .exp (-j.phi.1)} + .beta.2 {.beta.2 .d + N2 .exp (-j.phi.2)} ... 3) As described above, the combining method shown in the equation (3) is called maximum ratio combining diversity and has been conventionally performed. Fading propagation path characteristics β1 · exp (jφ1), β2 ·
This is a synthesizing method that maximizes the S / N ratio when the estimation accuracy of exp (jφ 2) is high.

Then, the data value is determined based on the positive / negative of the real part of the value C, and the determined data is differentially decoded to demodulate the transmission data.

(B) Configuration of the Embodiment Next, the overall configuration of the mobile communication receiving apparatus and the detailed configuration of the radio channel estimator of this embodiment will be described. here,
FIG. 1 is a block diagram showing the overall configuration of the mobile communication receiving apparatus of this embodiment, FIG. 2 is a block diagram showing the detailed configuration of the radio propagation path estimator, and FIG. 3 is a diagram showing the Doppler frequency / It shows an example of the configuration of the data number conversion unit.
It should be noted that in FIG. 1, the dedicated configuration for either the preceding wave or the delayed wave at that time is indicated by the symbol “a” at the end of the symbol.
Is added to the other dedicated structure, and the code “b” is added to the end of the code.

In FIG. 1, a reception signal input terminal 1 is a terminal for inputting a baseband reception signal (multipath wave signal: here, two waves), and the reception signal input from this input terminal 1 is 2 It is provided to the correlators 2a and 2b and the two synchronization acquisition devices 3a and 3b. In addition,
The correlators and the synchronization acquisition devices 2a and 3a, 2b and 3b constitute a correlation calculation means.

Each of the synchronization capturing devices 3a and 3b captures the synchronization timing of the preceding wave and the delayed wave from the received signal,
The timing signal is given to the corresponding correlators 2a and 2b. Although illustration of the signal line is omitted, a signal is transmitted and received between the synchronous capture devices 3a and 3b so that the synchronous capture device 3a functions as a preceding wave or a delayed wave at that time, and the other synchronous capture device. 3b functions as a delayed wave or a preceding wave. That is, the received waves that have passed through different propagation paths are targeted.

Each of the correlators 2a and 2b generates a spread code sequence (for example, a PN sequence) in synchronization with the preceding wave or the delayed wave on the basis of the timing signal from the corresponding synchronization acquisition device 3a, 3b, and outputs the base code. A complex correlation calculation between the band reception signal (complex signal) and the spread code sequence is performed, and the calculation result is given to the corresponding data modulation component removers and weighters 4a and 5a, 4b and 5b. Here, the complex correlation calculation is a process of integrating the product of the baseband received signal and the spread code sequence in a 1-bit section.

In the data demodulation mode, both data modulation component removers 4a and 4b are provided with the decision data output by the decision device 9 described later via the switch 10. In the case of this embodiment, since two-phase PSK modulation is performed on the transmission side, each data modulation component remover 4a,
4b removes the data modulation component included in the complex correlation signal by multiplying the determination data by the complex correlation signal from the corresponding correlators 2a and 2b, respectively. The signal from which the data modulation component has been removed is provided to the corresponding radio propagation path estimators 6a and 6b and Doppler frequency estimators 7a and 7b.

In the preamble mode, the preamble data (all 1's data) input from the preamble data input terminal 11 is applied to both data modulation component removers 4a and 4b via the switch 10. ing.

Each of the Doppler frequency estimators 7a and 7b accumulates a predetermined number of outputs of the corresponding data modulation component removers 4a and 4b, and uses them to perform frequency analysis (for example, Fourier transform is performed). ), And outputs the frequency having the peak value to the corresponding radio channel estimators 6a and 6b as the Doppler frequency.

This processing corresponds to the moving speed of the receiving apparatus or the partner transmitting apparatus receiver in the signal obtained by despreading the received signal that has undergone frequency selective fading due to the movement of the receiving apparatus or the partner transmitting apparatus. The Doppler frequency is estimated by analyzing the frequency of the received signal by utilizing the characteristic that the Doppler frequency component is included.

Each of the radio propagation path estimators 6a and 6b estimates the characteristics of the radio propagation path relating to the preceding wave or the delayed wave as a target based on the signals from the corresponding data modulation component removers 4a and 4b, The complex conjugate of the propagation path characteristic (fading characteristic) is given to the corresponding weighter 5a, 5b. At this time, each radio channel estimator 6a, 6b switches the number of data used for channel estimation based on the Doppler frequency signal from the corresponding Doppler frequency estimator 7a, 7b.

Both radio channel estimators 6a and 6b have the same detailed configuration. FIG. 2 shows a detailed configuration of the radio propagation path estimator 6a or 6b in this embodiment, that is, a configuration mainly including a non-recursive filter (transversal filter).

In FIG. 2, the radio propagation path estimator 6a or 6b comprises an M-stage shift register 20, M coefficient multipliers 211 to 21M, a summation unit 22, a complex conjugate calculation circuit 23, and a Doppler. It is composed of a frequency / data number converter 24. Here, M is the number of data corresponding to the Doppler frequency that requires the most number of data.

, 21M, the tap outputs P (1), ..., P of the shift register 20 are supplied to the coefficient multipliers 211 ,.
(M) is given, and each coefficient multiplier 211, ..., 21M is provided with a corresponding tap output P.
(1), ..., P (M), the coefficient A set by the Doppler frequency / data number conversion unit 24 as described later.
(1), ..., A (M) are multiplied. The sum total of the respective multiplication results is obtained by the totalizer 22. The output from the adder 22 is the value of the propagation path characteristic at the present time, and the complex conjugate is obtained by the complex conjugate calculating circuit 23, and the corresponding weighting is performed as the output of the radio propagation path estimator 6a or 6b. It is given to the vessels 5a and 5b. The complex conjugate calculation circuit 23 may be provided on the weighter 5a, 5b side.

The Doppler frequency / data number converter 24
For example, it is mainly composed of a conversion table as shown in FIG. Doppler frequency / data number converter 24
Extracts the number m of data to be used and the setting coefficient 1 / m based on the Doppler frequency f given from the corresponding Doppler frequency estimators 7a and 7b. Then, the tap output P (1) from the first stage to the m-th stage of the shift register 20
The setting coefficient 1 / m extracted as the coefficients A (1) to A (m) for P (m) is set, and the coefficient A (for the tap outputs P (m + 1) to P (M) from the (m + 1) th to Mth stages is set. m
0 is set as +1) to A (M). Thereby, the moving average of the period corresponding to the number of data m, that is, the tap outputs P (1) to P (m) and the coefficients A (1) to A (m).
The sum of products with (= 1 / m) is obtained from the summing device 22, and an estimated value of the propagation path characteristic in which the number of data is limited to m is obtained.

Each of the weighters 5a and 5b has a multiplication structure, and the complex correlation signals from the corresponding correlators 2a and 2b are converted into the estimated values of the propagation path characteristics from the corresponding radio propagation path estimators 6a and 6b. The complex conjugate is multiplied, the complex correlation signal is weighted according to the propagation path characteristic, and given to the synthesizer 8.

The combiner 8 has an addition structure, and adds the complex correlation signals weighted according to the propagation path characteristics from both weighters 5a and 5b, that is, obtains the maximum ratio combining diversity value and the judging device Give to 9.

The judging device 9 judges the data value based on the positive / negative of the real part of the maximum ratio combining diversity value, and the judged data is also sent to both the data modulation component removers 4a and 4b via the switching device 10. give. The determination data is also given to a differential decoder (not shown), and the determination data of one bit before and the determination data of this time are multiplied to perform differential decoding.

As described above, the switch 10 supplies the decision data output from the decider 9 to both the data modulation component removers 4a and 4b in the data demodulation mode, and from the preamble data input terminal 11 in the preamble mode. The above preamble data is applied to both data modulation component removers 4a and 4b.

(C) Operation of the Embodiment Next, the operation of the mobile communication receiving apparatus of the embodiment constituted by the above-mentioned respective parts will be described.

There are two modes of operation, a preamble mode and a data demodulation mode. The preamble mode operation is performed first, and then the data demodulation mode operation is performed.

In the pull amble mode, all 1's known data from the transmitting side as preamble data are
Data is transmitted without being modulated, while in the data demodulation mode, actual transmission data is transmitted from the transmitting side.

First, the operation of the preamble mode will be described. As described above, in this preamble mode, the switcher 10 has the preamble data input terminal 1
The 1 side is selected, and the preamble data of all 1 is given to both data modulation component removers 4a and 4b. Therefore, in the preamble mode, each data modulation component remover 4a, 4b is in a state of passing the complex correlation signal from the corresponding correlator 2a, 2b as it is, and each complex correlation signal corresponds to the corresponding radio channel estimator 6a, 6b and the corresponding Doppler frequency estimators 7a, 7b.

In each of the radio propagation path estimators 6a and 6b, the corresponding complex correlation signal is input to the shift register 20 (see FIG. 2) and sequentially shifted. That is, the bit period of the complex correlation signal (hence the preamble data)
The number of stages storing the value of the complex correlation signal increases as is increased. Eventually, the values P (1) to P (M) of the complex correlation signal are stored in all the stages of the shift register 20, whereby the initial state for obtaining the estimated value of the propagation path characteristic (fading characteristic) is obtained. , Transition from the preamble mode to the data demodulation mode.

On the other hand, each Doppler frequency estimator 7a, 7b
Also in (1), data is accumulated in the preamble mode, and when the mode is shifted to the data demodulation mode, the Doppler frequency can be estimated immediately.

Next, the operation of the data demodulation mode will be described. As described above, in the data demodulation mode, the switch 10 is connected to the determiner 9 side. That is, the determination data output from the determiner 9 is input to both the data modulation component removers 4a and 4b.

The following operation is repeatedly executed every time one bit of data is received. The operation will be described below assuming that the kth data is received.

One of the correlators 2a outputs a complex correlation signal R1 (k) of the baseband received signal (complex signal) and the spread code sequence synchronized with the preceding wave, and the other correlator 2b. Is the baseband received signal (complex signal)
And a complex correlation signal R2 (k) of the delayed wave and the spread code sequence synchronized with the delayed wave is output.

At this time, each radio propagation path estimator 6a,
The shift register 20 of 6b has k-1 to k-, respectively.
The values P (1) to P (M) of the complex correlation signal after the data modulation component is removed are stored for the M-th bit period, but the radio propagation path estimators 6a and 6b use the Doppler at that time. A value p1 (k) (= β1 · exp (jφ1) obtained by estimating the propagation path characteristic from the number of data corresponding to the frequency f (k).
)), Complex conjugate q1 (k) (= β1 · exp (−jφ1)), p2 (k) (= β2 · exp (jφ2)), q2
(K) (= β2 · exp (-jφ2)) is output.

Therefore, the complex correlation signals R1 from the correlators 2a and 2b corresponding to the respective weighters 5a and 5b are obtained.
(K), R2 (k) and the corresponding radio channel estimator 6
The outputs q1 (k) and q2 (k) of a and 6b are multiplied,
From each weighter 5a, 5b, a complex correlation signal x1 (k) (= β1 {β1.
d + N1.exp (-j.phi.1)}, x2 (k) (= .beta.
2 {β2 · d + N2 · exp (-jφ2)}) is output.

In the combiner 8 to which the two complex correlation signals x1 (k) and x2 (k) weighted by the channel characteristics are given, these signals x1 (k) and x2 (k) are added, and The calculation result is the maximum ratio combining diversity value C
It is output as (k).

Thus, the judging device 9 judges the data based on the maximum ratio combining diversity value C (k) and outputs the judgment data d (k). The determination data d (k) becomes 1 if the real part of the maximum ratio combining diversity value C (k) from the combiner 7 is 0 or positive, and − if the real part is negative.
Becomes 1.

In the differential decoder (not shown) to which the judgment data d (k) from the judgment device 9 is given, the immediately preceding judgment data d (k-1) and the present judgment data d (k) are multiplied. Then, the transmission data is reproduced (differential decoding is performed).

In the respective data modulation component removers 4a and 4b to which the current decision data d (k) is fed back via the switch 10, the corresponding correlator 2 is used.
complex correlation signals R1 (k) and R2 (k) from a and 2b
Is multiplied by the determination data d (k) of this time to remove the data modulation component contained in the complex correlation signals R1 (k) and R2 (k), and the removal results b1 (k) and b2 (k). ) Is output to the corresponding radio channel estimators 6a and 6b and the corresponding Doppler frequency estimators 7a and 7b.

In each of the radio propagation path estimators 6a and 6b, the removal results b1 (k) and b2 (k) are fetched as the first-stage value P (1) of the shift register 20 and the values stored so far. Are shifted by one step (P (i) is P (i + 1)).

Further, each Doppler frequency estimator 7a, 7b
Also in this case, the removal results b1 (k) and b2 (k) are captured and accumulated, and the Doppler frequency f (k + 1) is reviewed.

As a result, a series of data demodulation operations at the time of receiving the kth data are completed, and the operation proceeds to the operation at the time of receiving the k + 1th data.

(D) Effect of Embodiment According to the above embodiment, the Doppler frequency having a close relationship with the fading speed is detected, and the number of data used for estimating the propagation path characteristic is switched. It is possible to accurately estimate the propagation path characteristics even when the change of is rapid or slow.

As a result, even from the viewpoint of the entire mobile communication receiving device, since the characteristic compensation is performed according to the accurately estimated propagation path characteristic, accurate judgment data can be obtained.

In the above-mentioned embodiment, the configuration of compensating two waves is shown for the sake of simplicity. However, in reality, the number of received waves to be compensated is larger than that, and therefore the number of propagation path estimators is also increased. Will increase. In such a situation, it is difficult to apply an adaptive channel estimator with high power consumption and a complicated configuration, and a fixed coefficient channel estimator can estimate channel characteristics with high accuracy. Significant.

(E) Other Embodiments In the above embodiments, the propagation path estimator (excluding the complex conjugate calculating circuit) has the acyclic filter configuration shown in FIG. It is only necessary that the estimation calculation can be executed, and the configuration is not limited to that shown in FIG.

The estimation method is not limited to the moving average. In the invention separately filed by the applicant,
“Time variation of fading amplitude characteristic and fading phase characteristic of propagation path is 1 time when considering a short time section.
In this assumption, the tap coefficient A of the propagation path estimator including an acyclic filter with m stages is introduced.
It is described that (i) may be set to (-6i + 4m + 2) / {m (m-1)}. Even with a channel estimator that employs such a tap coefficient selection method, the number of data used for channel estimation may be changed according to the Doppler frequency.

Further, in the above embodiment, the number of data used for channel estimation is changed by setting some tap coefficients to 0. However, a plurality of shift registers having different stages are prepared. , A method of changing the number of data used for channel estimation by switching the shift register, etc.
Of course, other data number changing methods may be applied.

Furthermore, in the above embodiment, the mobile communication receiving apparatus receives differentially encoded transmission data on the transmitting side, and then directly spreads it with a predetermined spreading code sequence, PSK-modulates it and transmits it. However, the present invention is not limited to this, it is irrelevant to what processing was performed before the direct diffusion, and
The PSK modulation may also be 4-phase or 8-phase PSK modulation.
Depending on such a modification, there is a case where the data modulation component cannot be removed even if the determination data is directly multiplied by the complex correlation signal. In that case, the complex conjugate of the complex correlation signal is multiplied and the transmission side is used. Need to be changed according to the processing of.

The configuration of the propagation path estimator of the above embodiment can be applied not only to the mobile communication receiving device but also to other wireless receiving devices.

In the above embodiment, the processing is performed in the form of a complex signal, but the signal representation format is not limited to this.

Further, in the above embodiment, the mobile communication receiving apparatus for processing two received waves is shown, but the present invention is naturally applicable to an apparatus for processing three or more received waves.

[0071]

As described above, according to the channel estimating apparatus of the first aspect of the present invention, the data sequence reflecting only the channel characteristic with the data modulation component removed is input, and the data sequence is received. The Doppler frequency estimator that detects the Doppler frequency that is present, and the above-mentioned data sequence are input, and the estimation value of the propagation path characteristic is obtained.According to the detected Doppler frequency, the estimation value of the propagation path characteristic is calculated. Change the number of data used to obtain each of the above data series used
Is multiplied by a coefficient consisting of the reciprocal of this number of data
Since it is composed of a propagation path estimator that sums the calculated respective values , the propagation path characteristics can be accurately estimated regardless of the speed of change of the propagation path characteristics.

Further, according to the mobile communication receiving apparatus of the second aspect of the present invention, the characteristic configuration of the first aspect of the present invention is applied to the configuration for estimating the propagation path characteristics for each received wave required for adaptive RAKE combining. Therefore, the accuracy of the determination data can be improved without increasing the size and complexity of the configuration.

[Brief description of drawings]

FIG. 1 is a block diagram showing an overall configuration of a mobile communication receiving apparatus according to an embodiment.

FIG. 2 is a block diagram showing a detailed configuration of a radio channel estimator of the embodiment.

FIG. 3 is an explanatory diagram showing a configuration example of a Doppler frequency / data number conversion unit of the embodiment.

[Explanation of symbols]

1 ... Received signal input terminals, 2a, 2b ... Correlators, 3a, 3
b ... synchronization capture device, 4a, 4b ... data modulation component remover,
5a, 5b ... Weighter, 6a, 6b ... Radio channel estimator, 7a, 7b ... Doppler frequency estimator, 8 ... Combiner,
9 ... Judgment device, 20 ... Shift register, 211-21M ...
Variable coefficient multiplier, 22 ... Totalizer, 24 ... Doppler frequency / data number converter.

Continuation of front page (56) Reference JP-A-60-114780 (JP, A) JP-A-5-37426 (JP, A) JP-A-5-268108 (JP, A) JP-A-7-115373 (JP , A) (58) Fields investigated (Int.Cl. ⁷ , DB name) H04B 3/00-3/44 H04B 7/005-7/015 H04B 7/24-7/26 H04Q 7/00-7 / 38

Claims (4)

(57) [Claims] 1. A data sequence reflecting only the channel characteristics from which data modulation components have been removed is input, a Doppler frequency estimator for detecting the Doppler frequency received by the data sequence, and the data sequence are input, For obtaining the estimated value of the channel characteristic, the number of data used to obtain the estimated value of the channel characteristic is changed according to the detected Doppler frequency, and this data sequence is used for each of the above-mentioned data series to be used.
Data multiplied by a coefficient consisting of the reciprocal of the number
And a channel estimator that sums up the respective values . 2. The propagation path estimator receives a shift register having M taps and a corresponding tap output of the shift register, and M variable coefficient multiplications for multiplying the tap output by a variable coefficient. And a totalizer for summing the outputs from all the variable coefficient multipliers. The number of variable coefficients of the variable coefficient multiplier is determined by the number of variable coefficients of the detected Doppler frequency. Reverse
Variable coefficient of other variable coefficient multiplier
The channel estimation apparatus according to claim 1, wherein the number of data and the coefficient used for obtaining the estimated value of the channel characteristic are changed by setting 0 to 0. 3. A mobile communication receiving apparatus, wherein a transmission side directly spreads transmission data with a spreading code sequence, digital phase modulates and gives a transmitted signal, a plurality of receptions received through a plurality of radio propagation paths. A plurality of correlation calculation means provided corresponding to each of the waves, for performing a correlation calculation between the spread code sequence synchronized with any of the reception waves and the baseband received signal, and outputting the correlation calculation signal. And a plurality of data modulation component removing means provided corresponding to each of the received waves, for removing the data modulation component from the correlation calculation signal from the corresponding correlation calculation means based on the determination data. , A signal provided corresponding to each of the received waves, and based on the signal from which the data modulation component has been removed output from the corresponding data modulation component removal means, A plurality of Doppler frequency estimating means for detecting the La frequency and a signal provided corresponding to each of the received waves, in which the data modulation component output from the corresponding data modulation component removing means is removed. Based on this, the channel characteristics of the wireless channel are estimated, and a signal for weighting is output.According to the detected Doppler frequency from the corresponding Doppler frequency estimating means, data modulation used for channel estimation. Change and use the data number of the signal from which the component is removed
For each signal with the data modulation component removed
Multiply the coefficient consisting of the reciprocal of this number of data, and multiply
A plurality of radio channel estimating means for summing the respective values, and those provided corresponding to each of the received waves,
Based on the signal from the corresponding radio channel estimation means,
A weighting means for weighting the correlation calculation signals from the corresponding correlation calculation means, a synthesizing means for adding and synthesizing the outputs of all the weighting means, and a deciding means for making a data decision based on the outputs of the synthesizing means. A mobile communication receiving device, comprising: 4. Each of the radio channel estimating means receives a shift register having M taps and a tap output corresponding to the shift register, and the tap outputs are multiplied by a variable coefficient. a variable coefficient multiplier, and a summer summing the outputs from all of the variable coefficient multiplier, depending on the Doppler frequency detected variable coefficients of the variable coefficient multiplier number corresponding to the detected Doppler frequency The opposite of the number
Variable coefficient of other variable coefficient multiplier
4. The mobile communication receiving apparatus according to claim 3, wherein the number of data and the coefficient used to obtain the estimated value of the propagation path characteristic are changed by setting 0 to 0.