JP3191819B2 - Radio receiver - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a radio receiver and its tracking correction.

[0002]

2. Description of the Related Art For example, AM of a superheterodyne system
In the receiver, fr: tuning frequency of the antenna tuning circuit (frequency desired to receive). fi: Intermediate frequency. For example, 450 kHz. fo: Local oscillation frequency. In the case of upper heterodyne, the tuning frequency f
Regardless of r or the local oscillation frequency fo, fr = fo-fi (i) must always hold.

That is, when the local oscillation frequency is a value fo, the tuning frequency fr of the antenna tuning circuit is a value (fo-
fi). Alternatively, when the tuning frequency of the antenna tuning circuit is the value fr, the local oscillation frequency f
o must be the value (fr + fi).

At this time, fr = 1 / {2π (LrCr) **. 5} fo = 1 / {2π (LoCo) **. 5} Lr: inductance of coil of antenna tuning circuit Cr: antenna Variable condenser for tuning circuit (variable capacitor)
Lo: the inductance of the coil of the local oscillation circuit Co: the capacitance of the variable capacitor of the local oscillation circuit (** indicates a power), the equation (i) is expressed as 1 / ｛2π (Lo Co) **. 5 {−1 / {2π (LrCr) **. 5} = fi (2)

Therefore, when a variable condenser is turned for channel selection, (i) is always set between the capacitance Cr and the capacitance Co regardless of the reception frequency fr or the local oscillation frequency fo.
i) The equation must hold. That is, in the variable condenser for superheterodyne, the capacitances Cr, C
o must always change with respect to the rotation angle of the movable electrode so that equation (ii) is satisfied.

The tuning frequency fr and the local oscillation frequency fo
If an error occurs between Equation 1 and Equation (i) (or Equation (ii)) without being satisfied, this is called a tracking error. If this tracking error is present, the reception sensitivity is reduced.

That is, if the local oscillation frequency is a value fo, the tuning frequency f of the antenna tuning circuit at this time is
r must be the frequency (fo-fi) from equation (i). However, when there is a tracking error, the tuning characteristic of the antenna tuning circuit becomes, for example, as shown by the solid line A in FIG. 3, and the tuning frequency fr is shifted from the original frequency (fo-fi) by the frequency Δf.

When such an error occurs, the gain of the antenna tuning circuit is reduced by ΔG, and as a result, the receiving sensitivity is reduced by ΔG. Therefore, in a superheterodyne receiver, if there is a tracking error, the reception sensitivity is reduced.

For this reason, a variable capacitor having a capacitance change characteristic that always satisfies the equation (ii) is prepared as a standard product. Such a variable condenser is called a trackingless variable condenser.

[0010]

In the case where the AM receiver is formed as a one-chip IC, if the intermediate frequency filter is constituted by a ceramic filter or the like, the intermediate frequency filter cannot be integrated into an IC. , A resistor, a capacitor, and an active filter using an operational amplifier.

However, at this time, if the intermediate frequency fi is set to the standard 450 kHz, the area occupied by the intermediate frequency filter (active filter) in the semiconductor pellet of the IC becomes large, which is not preferable.

Therefore, it is conceivable to further reduce the intermediate frequency fi to, for example, 55 kHz, which is sufficiently lower than the reception band. In other words, with such a frequency, the area occupied by one stage of the intermediate frequency filter in the semiconductor pellet of the IC becomes larger than that in the case where the intermediate frequency is 450 kHz, but the number of stages of the filter can be reduced. The area occupied by the entire frequency filter is reduced, which is advantageous for IC implementation.

However, the standard trackingless variable condenser has a characteristic that no tracking error occurs when the intermediate frequency fi is 450 kHz.
When the frequency is set to 55 kHz, a tracking error occurs. Therefore, a standard trackingless variable condenser cannot be used for a receiver having an intermediate frequency fi of 55 kHz.

For this reason, when the intermediate frequency fi is 55 kHz, the capacitance Cr on the antenna tuning side and the capacitance Co on the local oscillator side are set.
However, it is necessary to use a variable capacitor that changes equally, but such a variable capacitor is out of the standard product, and thus the cost is increased.

The present invention is to solve the above problems.

[0016]

Therefore, in the present invention, 1. The receiving system is a superheterodyne system. 2. The intermediate frequency fi is, for example, 55 kHz. 3. Q-dump the antenna tuning circuit to reduce tracking errors. 4. In this case, when a resistor is connected to the antenna tuning circuit and Q dump is performed, noise is generated by the resistor and the receiving sensitivity is reduced. Therefore, the input impedance of the next stage of the antenna tuning circuit is reduced, and Q dump of the antenna tuning circuit is performed. 5. Connect a capacitor for tracking correction to the antenna tuning circuit and the local oscillation resonance circuit. It is assumed that.

That is, if the reference numerals of the respective parts correspond to the embodiments described later, the antenna tuning circuit 1 having the first coil L1 and the first variable capacitor VC1, the local oscillation circuit 13, and the second coil L2 And second variable capacitor
A resonance circuit 2 having VC2 and determining an oscillation frequency fo of the local oscillation circuit 13 and a broadcast wave signal Sr selected by the antenna tuning circuit 1 are converted into a predetermined intermediate signal by the local oscillation signal So from the local oscillation circuit 13. A mixer circuit 12A for converting the frequency to an intermediate frequency signal Si having a frequency fi, a band-pass filter 17 for extracting the intermediate frequency signal Si from the output signal of the mixer circuit 12A,
And a detection circuit 22 for detecting the intermediate frequency signal Si extracted by the antenna 7, a first variable capacitor of the antenna tuning circuit 1 in the superheterodyne receiver.
A first correction capacitor C11 is connected to VC1, a second correction capacitor C21 is connected to the second variable capacitor VC2 of the resonance circuit 2, and the first and second correction capacitors C11 and C21 First and second variable capacitors VC1
, VC2 in the direction in which the tracking error is eliminated, and a negative feedback circuit 115 is provided in the next stage circuit 11 to which the antenna tuning circuit 1 is connected. Negative feedback is applied to the circuit 11 so that the input impedance Zin becomes small, and the input impedance Zin is reduced by the negative feedback, so that the antenna tuning circuit 1 is Q-dumped to perform tracking correction.

[0018]

[Function] Variable condenser VC by correction capacitors C11 and C21
1. Since the change characteristic of the capacitance of VC2 is corrected and the antenna tuning circuit 1 is Q-dumped by the input impedance Zin of the high-frequency amplifier 11, the intermediate frequency fi can be reduced even if the standard trackingless varicons VC1 and VC2 are used. For example, it can be 55 kHz. Therefore, a one-chip IC including the intermediate frequency filter 17 can be realized.

[0019]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Referring to FIG.
Chip receiver ICs for an AM receiver, T1 to T8 are their terminal pins, pin T3 is a power supply terminal pin, and pin T4 is a ground terminal pin.

The components outside the chain line are externally attached components or circuits, 1 is an antenna tuning circuit, and 2 is a resonance circuit for local oscillation. The tuning circuit 1 includes a bar antenna (antenna tuning coil) L1 and a variable condenser VC.
1 and tracking correction capacitors C11 and C12. The resonance circuit 2 includes a local oscillation coil L2, a variable condenser VC2, and a tracking correction capacitor C21,
C22. In this case, the variable condenser VC1
And VC2 are linked, and the intermediate frequency fi
Is a trackingless variable condenser for 450 kHz,
It is a standard trackingless variable condenser.

Further, SW is a power switch, BATT is a 3V battery for power supply, VR is a variable resistor for volume control, SP
Is a speaker.

Then, the antenna tuning circuit 1 selects and extracts the broadcast wave signal SrSr = Er.sin.omega.rtt.omega.r = 2.pi.fr of the frequency fr. In the following signal processing, only the relative amplitude and phase of each signal are related, so the initial phase of each signal is omitted in the above equation and the following description. At this time, the tuning frequency fr of the tuning circuit 1 is ideally the frequency (fo-fi).

The signal Sr is supplied to the high frequency amplifier 11 through the pin T1 of the IC 10, and the signal Sr from the amplifier 11 is supplied to the first and second mixer circuits 12
A, 12B . Although an example of the high-frequency amplifier 11 will be described later, a resistor R11 for negative feedback is connected to the amplifier 11, and the input impedance Zin is set to a predetermined small value, for example, 20 kΩ by the negative feedback.

Further, the local oscillation circuit 13 has a terminal pin T
2, the resonance circuit 2 is connected to form a local oscillation signal So. In this case, the oscillation frequency of the oscillation signal So is 2fo, and 2fo = (fr + fi) × 2fi is an intermediate frequency, and fi = 55 kHz.

The oscillation signal So is output from the counter 1
4 and is divided into local oscillation signals Soa and Sob having a frequency of ２ and a phase different from each other by 90 °. That is, the frequency is divided into signals Soa and Sob of Soa = Eo.cos.omega.t Sob = Eo.sin.omega.o.t.omega.o = 2.pi.fo.

The signals Soa and Sob are supplied to mixer circuits 12A and 12B and multiplied by the signal Sr, respectively, and the following signals Sia and Sib are extracted from the mixer circuits 12A and 12B. That is, Sia = Sr · Soa = Er · sin ωr t · Eo · cos ωot = α ｛sin (ωr + ωo) t + sin (ωr−ωo) t｝ Sib = Sr · Sob = Er · sin ωrt · Eo · sin ωo t = α ｛-cos (ωr + ωo) t + cos (ωr-ωo) t｝ ωr = 2πfr α = Er · Eo / 2 The signals Sia and Sib are extracted.

Then, as described later, these signals Si
Of the signals a and Sib, the signal component of the angular frequency (ωr−ωo) is used as the intermediate frequency signal, and the angular frequency (ωr + ωo)
Since the signal component of the above formula is eliminated, for simplicity, ignoring the signal component of the angular frequency (ωr + ωo) in the above equation, Sia = α · sin (ωr−ωo) t Sib = α · cos (ωr−ωo) t.

At this time, since the image signal Sm is Sm = Em · sin ωmt tωm = ωo + ωiωi = 2πfi, the broadcast wave signal Sr from the tuning circuit 1 includes the image signal Sm. Then, the signals Sia and Sib at this time are as follows: Sia = α · sin (ωr−ωo) t + β · sin (ωm−ωo) Sib = α · cos (ωr−ωo) t + β · cos (ωm−ωo) β = Em · Eo / 2. Further, since ωr <ωo <ωm, the above equation is given by: Sia = α · sin (ωr−ωo) t + β · sin (ωm−ωo) = − α · sin (ωo−ωr) t + β · sin (ωm −ωo) Sib = α · cos (ωr−ωo) t + β · cos (ωm−ωo) = α · cos (ωo−ωr) t + β · cos (ωm−ωo)

Then, these signals Sia and Sib are supplied to the phase shift circuits 15A and 15B. This phase shift circuit 15A,
15B is constituted by, for example, an active filter using a capacitor, a resistor, and an operational amplifier. The phase shift circuit 15A shifts the phase of the signal Sia by the value φ, and the phase shift circuit 15B shifts the signal Sib to the value (φ + 90 °). )
, The phase difference between the two input signals Sia and Sib is 90 ° ± 1 ° in the band of 55 kHz ± 10 kHz.
The phase shifts to the following relationship.

Thus, the signal Sib is advanced by 90 ° with respect to the signal Sia by the phase shift circuits 15A and 15B, and Sia = −α · sin (ωo−ωr) t + β · sin (ωm−ωo) Sib = α Cos (ωo−ωr + 90 °) t + β · cos (ωm−ωo + 90 °) = α · sin (ωo−ωr) t + β · sin (ωm−ωo), and these signals Sia and Sib are supplied to the subtraction circuit 16. From the subtraction circuit 16, Si = Sib−Sia = α · sin (ωo−ωr) t + β · sin (ωm−ωo) − ｛− α · sin (ωo−ωr) t + β · sin (ωm− ωo)｝ = 2α · sin (ωo−ωr) t A signal Si is extracted.

Here, since ωo−ωr = 2π (fo−fr) = 2πfi, the signal Si is a target intermediate frequency signal. Even if the broadcast wave signal Sr from the tuning circuit 1 includes the image signal Sm, the signal component due to the image signal Sm is not included in the intermediate frequency signal Si because it is canceled.

Thus, the intermediate frequency signal Si (and the signal component of the angular frequency (ωr + ωo)) converted from the broadcast wave signal Sr is extracted from the subtraction circuit 16.

The intermediate frequency signal Si is supplied to a band-pass filter 17 for an intermediate frequency filter.
The band-pass filter 17 includes, for example, a capacitor,
It consists of a biquad-type active filter using a resistor and an operational amplifier, and its pass band is 55k.
Hz ± 3 kHz. Thus, the bandpass filter 1
At 7, the unnecessary signal components are removed and only the intermediate frequency signal Si is extracted.

Then, the extracted intermediate frequency signal S
i is supplied to an AM detection circuit 22 through an amplifier 21 to extract an audio signal Ss (and a DC component V22 having a level corresponding to the level of the intermediate frequency signal Si), and the audio signal Ss is converted to a differential input audio amplifier. 23
The signal Ss from the amplifier 23 is supplied to the pin T
8 and to the speaker SP through the capacitor C5.

Further, the signal Sib from the mixer circuit 12B
Is supplied to the AGC voltage forming circuit 18 to form an AGC voltage. The AGC voltage is supplied to the amplifier 11 as a gain control signal, and the AGC voltage is applied to the signals Sia and Sib.
Is performed. In this case, a capacitor C3 for smoothing the AGC voltage is connected to the forming circuit 18 through the pin T5. Also, this AGC voltage is applied to the phase shift circuit 15A,
15B and the respective operational amplifiers constituting the bandpass filter 17 are supplied as reference voltages.

The detection output of the detection circuit 22 is AGC
The AGC voltage is supplied to the voltage forming circuit 24 to form the AGC voltage.
The AGC voltage is supplied to the amplifiers 11 and 21 as a gain control signal, and AGC is performed on the intermediate frequency signals Sia, Sib and Si.

In this case, a capacitor C4 is connected to the forming circuit 24 through the pin T6, and a low-pass filter is formed by the capacitor C4 to take out the DC voltage V22 from the detection output. A voltage is formed. Also, this DC voltage V22 is
The DC component V22 supplied to the differential input of the amplifier 23 and supplied from the detection circuit 22 to the amplifier 23 together with the audio signal Ss is equivalently canceled.

Further, a variable resistor VR is connected to the amplifier 23 through a pin T7, and the gain of the amplifier 23 is controlled in accordance with the resistance value of the variable resistor VR.
The volume is adjusted by the variable resistor VR.

The capacitor C6 is for bypassing signal components other than the audio signal Ss.

As described above, the AM broadcast is received. In this case, according to the present invention, the variable condensers VC1,
Since the correction capacitors C11 and C21 are connected to VC2, the intermediate frequency fi is 55 kHz, while the
1, Even if VC2 is a standard trackingless variable capacitor, the variable capacitor VC when capacitors C11 and C21 are included
1. The change characteristic of the capacitance of VC2 can be corrected in a direction satisfying the expression (ii).

In contrast to fi = 55 kHz, even if the variable condensers VC1 and VC2 are standard trackingless variable condensers, since the amplifier 11 is subjected to negative feedback, there is little reduction in receiving sensitivity due to tracking errors.

That is, as shown in FIG. 2, Av: the voltage gain of the amplifier 11 when no negative feedback is applied. Vin: the signal voltage of the signal Sr supplied to the amplifier 11. ir: signal current flowing through the resistor R11. Zin: input impedance of the amplifier 11 when negative feedback is applied. Then, since (Av.Vin-Vin) / R11 = ir, Vin / ir = R11 / (Av-1). If the gain Av is sufficiently large, the above equation becomes Vin / ir = R11 / Av. Further, since Zin = Vin / ir, the above equation becomes Zin = R11 / Av.

Therefore, for example, if R11 = 400 kΩ and Av = 20 times, the input impedance Zin of the amplifier 11 becomes 20 k
Ω, and the antenna tuning circuit 1 is Q-dumped by the input impedance Zin of 20 kΩ.

Therefore, as a result of the Q dump, the tuning characteristic of the tuning circuit 1 becomes broad as shown by a broken line B in FIG. 3, and the tuning frequency fr of the tuning circuit 1 is changed from the original frequency (fo-fi). Even if it is shifted, the decrease in gain at that frequency (fo-fi) is smaller than the amount of decrease ΔG in the tuning characteristic of the solid line A. That is, even if there is a tracking error, a decrease in the receiving sensitivity due to the tracking error becomes small.

In addition, when the Q dump is performed, if a resistor is connected to the antenna tuning circuit 1 and the Q dump is performed, noise increases due to the resistor, and as a result, the receiving sensitivity decreases. According to this, since Q-dump is performed by the input impedance of the amplifier 11 at the next stage of the tuning circuit 1, there is no increase in noise, and therefore, there is no decrease in reception sensitivity.

Further, since a standard tracking-less variable condenser for an intermediate frequency can be used as the variable condensers VC1 and VC2, the cost is not increased.

FIG. 4 shows a measurement example showing the relationship between the reception frequency fr and the decrease in reception sensitivity due to a tracking error. That is, although the Q of the bar antenna L1 is usually about 100, if the present invention is not applied and Q = 100, the decrease in the receiving sensitivity due to the tracking error is about ± 5 dB. However, when Q dump is performed by applying the present invention and Q = 50, the decrease in the receiving sensitivity due to the tracking error falls within ± 2 dB.

As described above, according to the present invention, even if the intermediate frequency fi is set to 55 kHz using the standard trackingless variable condensers VC1 and VC2, it is possible to suppress the decrease in the receiving sensitivity due to the tracking error.

Further, since the intermediate frequency fi is sufficiently lower than the general intermediate frequency and the reception band, the IC
In this case, the area occupied by the entire band-pass filter 17 can be reduced, which is advantageous for IC implementation.

In general, when the intermediate frequency fi is low,
Although the image characteristics deteriorate, the image characteristics are not deteriorated because the image signals Sm are removed by the circuits 12A to 16A.

Further, since the phase shift circuits 15A, 15B and the band-pass filter 17 are constituted by active filters, these circuits 15A, 15B, 17 have a limit in the signal level that can be handled, but the amplifier 12
AGC is applied to phase shift circuits 15A, 15A,
No excessive input occurs to the 5B and the bandpass filter 17.

FIGS. 5 and 6 show specific examples of the high-frequency amplifier 11. That is, the transistor Q24 for the constant current source is connected to the emitter of the transistor Q11, so that the transistor Q11 functions as an emitter follower and the coil L1
Is connected to the base of transistor Q12 through transistor Q11. And the transistor Q
12. The emitter of Q13 is a transistor Q23 for a constant current source.
And the transistor Q
The collectors of 12 and Q13 are connected to the diode-connected transistors Q15 and Q16 to form a differential amplifier 111. Further, a diode-connected transistor Q14
And the transistor Q22 for the constant current source supplies a base bias to the transistor Q13.

The transistors Q22 to Q24 are connected to the current mirror circuit 112 using the transistor Q21 as an input side.
The transistor Q21 is connected to a constant current source Q25 that determines a current flowing through the current mirror circuit 112.

The transistors Q15 and Q16 form current mirror circuits 113 and 114 with the transistors Q31 and Q32 as output sides, respectively.
4. A current mirror circuit 115 having Q35 is connected. This current mirror circuit 115 includes a transistor Q
The collector of the transistors Q32 and Q34 comprises a negative feedback circuit together with 31, Q32 and the resistor R11.
It is connected to the base of the transistor Q11 through the negative feedback resistor R11.

Further, the current mirror circuits 113 and 11
4 are connected to output transistors Q41 to Q44. The transistors Q41 to Q44 are respectively connected to the mixer circuit 12
It is also an input transistor of the double balance type multiplication circuit constituting A and 12B.

According to such a configuration, the broadcast wave signal Sr is selected by the tuning circuit 1, and this signal Sr is supplied to the transistor Q12 through the transistor Q11, and is converted from a signal voltage into a signal current and amplified. , The signals Sr of opposite phases are output from the collectors of the transistors Q12 and Q13. The signal Sr is supplied to the mixer circuits 12A and 12B through the transistors Q15 and Q16. At this time, the mixer circuits 12A and 12B
B receives local oscillation signals Soa and Sob from the counter 14.

Therefore, the signal Sr is supplied to the mixer circuit 12
In A and 12B, the signals Soa and Sob are frequency-converted into intermediate frequency signals Sia and Sib.
a and Sib are extracted from the mixer circuits 12A and 12B.

Further, in this case, the transistors Q15, Q15
The signal Sr from the transistor 16 is supplied to the current mirror circuit 115 through the transistors Q31 and Q32 and synthesized, and a signal Sr of the opposite phase is taken out from the collectors of the transistors Q32 and Q34. Is fed back to the base.

Therefore, the amplifier 11 is equivalent to FIG.
Therefore, the input impedance Zin of the amplifier 11 becomes Zin = R11 / Av as described above, and the antenna tuning circuit 1 is Q-dumped by the input impedance Zin, for example, Zin = 20 kΩ. .

[0060]

According to the present invention, the variable capacitors VC1, VC2
Are connected to the correction capacitors C11 and C21,
While the intermediate frequency fi is 55 kHz, the variable condenser VC1,
Even if VC2 is a standard trackingless variable capacitor, the variable capacitors VC1 and VC2 when capacitors C11 and C21 are included
Can be corrected in a direction that satisfies the formula (ii).

Further, the input impedance Zin is reduced by applying a negative feedback to the amplifier 11, and the antenna tuning circuit 1 is Q-dumped by the input impedance Zin. Can be suppressed. Therefore, using a standard trackingless variable condenser, the intermediate frequency fi
Can be set to 55 kHz.

Further, when the Q dump is performed, if a resistor is connected to the antenna tuning circuit 1 and the Q dump is performed, noise increases due to the resistor, and as a result, the receiving sensitivity decreases. According to this, since Q-dump is performed by the input impedance of the amplifier 11 at the next stage of the tuning circuit 1, there is no increase in noise, and therefore, there is no decrease in reception sensitivity.

Since standard tracking-less variable capacitors for the intermediate frequency can be used as the variable capacitors VC1 and VC2, there is no increase in cost.

Further, since the intermediate frequency fi is 55 kHz, which is sufficiently lower than the standard intermediate frequency and the reception band, the area occupied by the entire band-pass filter 17 can be reduced in the case of IC. Is advantageous.

Generally, when the intermediate frequency fi is low,
Although the image characteristics deteriorate, the image characteristics are not deteriorated because the image signals Sm are removed by the circuits 12A to 16A.

Further, since the phase shift circuits 15A and 15B and the band pass filter 17 are constituted by active filters, these circuits 15A, 15B and 17 have a limit in the signal level which can be handled, but the amplifier 12
AGC is applied to phase shift circuits 15A, 15A,
No excessive input occurs to the 5B and the bandpass filter 17.

[Brief description of the drawings]

FIG. 1 is a connection diagram showing an example of the present invention.

FIG. 2 is a partial equivalent circuit diagram for explaining the present invention.

FIG. 3 is a characteristic diagram for explaining a tracking error.

FIG. 4 is a characteristic diagram showing a measurement example of a change in gain.

FIG. 5 is a connection diagram showing a part of an example of the present invention.

FIG. 6 is a connection diagram showing a continuation of FIG. 5;

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 Antenna tuning circuit 2 Resonance circuit for local oscillation 10 1 chip IC 11 High frequency amplifier 12A, 12B Mixer circuit 13 Local oscillation circuit 14 Counter 15A, 15B Phase shift circuit 16 Subtraction circuit 17 Band pass filter 18 AGC voltage forming circuit 21 Amplifier 22 AM detection circuit 23 Audio amplifier 24 AGC voltage forming circuit SP Speaker VR Variable resistor C11 to C22 Correction capacitor T1 to T8 Terminal pin VC1 Antenna tuning variable capacitor VC2 Local oscillation variable capacitor

──────────────────────────────────────────────────続 き Continued on the front page (58) Field surveyed (Int.Cl. ⁷ , DB name) H04B 1/26 H03J 3/00-3/32 H03G 1/00-3/34

Claims (2)

(57) [Claims] 1. An antenna tuning circuit having a first coil and a first variable capacitor; a local oscillation circuit; and an oscillation frequency of the local oscillation circuit having a second coil and a second variable capacitor. And a broadcast wave signal selected by the antenna tuning circuit,
A mixer circuit for converting the frequency of the local oscillation signal from the local oscillation circuit into an intermediate frequency signal having a predetermined intermediate frequency, a band-pass filter for extracting the intermediate frequency signal from an output signal of the mixer circuit, and a band-pass filter. A superheterodyne receiver having a detection circuit for detecting the extracted intermediate frequency signal, wherein a first correction capacitor is connected to the first variable capacitor of the antenna tuning circuit; A second correcting capacitor is connected to the second variable capacitor, and the first and second correcting capacitors are connected to the first variable capacitor.
And correcting the change characteristic of the capacitance of the second variable capacitor so as to eliminate the tracking error, and providing a negative feedback circuit in a circuit at the next stage to which the antenna tuning circuit is connected. A radio receiver in which negative feedback is applied to a next stage circuit so as to reduce its input impedance, and the input impedance is reduced by the negative feedback, so that the antenna tuning circuit is Q-dumped to perform tracking correction. 2. An antenna tuning circuit having a first coil and a first variable capacitor; a local oscillation circuit; and a second coil and a second variable capacitor determining an oscillation frequency of the local oscillation circuit. And a broadcast wave signal selected by the antenna tuning circuit,
A mixer circuit for converting the frequency of the local oscillation signal from the local oscillation circuit into an intermediate frequency signal having a predetermined intermediate frequency, a band-pass filter for extracting the intermediate frequency signal from an output signal of the mixer circuit, and a band-pass filter. A superheterodyne receiver having a detection circuit for detecting the extracted intermediate frequency signal, wherein the local oscillation frequency of the local oscillation circuit is sufficient for the intermediate frequency of the intermediate frequency signal to be higher than a reception band. A first correction capacitor is connected to the first variable capacitor of the antenna tuning circuit, and a second correction capacitor is connected to the second variable capacitor of the resonance circuit. With the first and second correction capacitors, the first
And correcting the change characteristic of the capacitance of the second variable capacitor so as to eliminate the tracking error, and providing a negative feedback circuit in a circuit at the next stage to which the antenna tuning circuit is connected. A radio receiver in which negative feedback is applied to a next stage circuit so as to reduce its input impedance, and the input impedance is reduced by the negative feedback, so that the antenna tuning circuit is Q-dumped to perform tracking correction.