JP3087846B1 - Switching power supply - Google Patents

Abstract

A switching power supply device is required to reduce noise. SOLUTION: A first and a second switch Q1, Q2 are provided between a pair of DC output lines 9a, 9b of an input rectifying / smoothing circuit 4.
Connected in series. First and second switches Q1,
Capacitors C1 and C2 are connected in parallel with Q2. A series circuit of a primary winding N1 of a transformer and a capacitor Cr for current resonance is connected in parallel with the second switch Q2. Secondary winding N
2 is connected to an output rectifying / smoothing circuit 5. The first and second switches Q1 and Q2 are alternately turned on with a delay time.
Turn off. The primary winding N1 and the secondary winding N2 of the transformer T are arranged at different positions on the core 10 to reduce the stray capacitance therebetween. Also, the core 10 is connected to a cold end.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply such as a DC-DC converter, and more particularly to a switching power supply capable of reducing noise.

[0002]

2. Description of the Related Art As a switching power supply, DC-DC of a single type, a half bridge type, a full bridge type and the like are used.
Converters are known.

[0003]

The switching power supply generates noise. This noise includes conduction noise and radiation noise, and its main sources are a switching element and a rectifier diode. Noise in the switching element is a transient change in voltage when turning on and off,
Caused by excessive changes in current. The noise in the diode is mainly caused by a transient current change in the recovery time when the diode is off. Recently, voltage-type soft switching, which is one of resonance types, has attracted attention as a low-noise trump card. The voltage type soft switching circuit resonates using LC only for a short period of time when the main switch is on and off, and uses a part of the resonance voltage starting from zero as the switch voltage. It is configured to reduce switching loss. However, it has been difficult to reduce noise and improve efficiency while suppressing an increase in cost.

Accordingly, a first object of the present invention is to provide a switching power supply device capable of suppressing an increase in cost and reducing noise. A second object of the present invention is to provide a switching power supply device capable of suppressing an increase in cost and reducing noise and improving efficiency.

[0005]

To solve the above problems SUMMARY OF THE INVENTION The first invention for achieving the purpose of a DC power supply, pre-Symbol
First and second terminals connected between a pair of DC terminals of a DC power supply.
A series circuit of switches and a core and wound on this core
An output transformer having a primary winding and a secondary winding;
A second switch is connected in parallel via the primary winding.
Connected to the resonance capacitor connected to the secondary winding.
Output rectifying / smoothing circuit, and the first and second switches.
A control circuit for alternately on / off control,
The primary winding and the secondary winding are arranged at different positions of a transformer core so as to reduce stray capacitance therebetween.
The primary winding is arranged such that the end of the primary winding on which high frequency is not superimposed is closer to the secondary winding than the end of the primary winding on which high frequency is superimposed. Wherein the core is connected to a circuit conductor on which a high frequency is not superimposed.

In order to achieve the second object,
As described in claim 2, the first and second switches are connected in parallel.
It is desirable to connect a capacitor or a parasitic capacitance . Further, a circuit in which a high frequency is not superimposed is provided.
The conductor (cold end) turns the second switch into a DC power supply
It is desirable to use a conductor to be connected .

[0007]

According to the invention of each claim, one of the transformers
The secondary winding and the secondary winding are arranged at different positions of the core, and
The end of the primary winding on which the high frequency is not superimposed (cold end side) is closer to the secondary winding than the end of the primary winding on which the high frequency of the primary winding is superimposed (hot end side). 1
Since the secondary winding is provided and the core is connected to a circuit conductor (cold end) on which the high frequency is not superimposed, the stray capacitance between the primary winding and the secondary winding is reduced, so that the secondary winding has It is possible to reduce the transmission of noise in the next winding and in the opposite direction. Further, the leakage current can be reduced. According to the second aspect of the present invention, the first and the second are caused by resonance .
2, it is possible to reduce the switching loss and noise when the switch is turned on and when the switch is turned off.

[0008]

[Embodiment and Examples] Next, with reference to FIGS. 1 to 15 illustrating the examples and embodiments of the present invention.

The switching power supply according to the embodiment shown in FIG. 1 has AC power supply terminals 1a and 1b, a filter 2 for removing high frequency components, and an input rectifying and smoothing circuit 4 as a DC power supply.
, First and second switches Q1, Q2, first and second diodes D1, D2, first and second capacitors C1, C1, an output high-frequency transformer T, and a series resonance capacitor. It has a Cr, an output rectifying and smoothing circuit 5, a pair of output terminals 6a and 6b, and a control circuit 7.

A pair of AC power supply terminals 1a and 1b
Connected to 0Hz commercial AC power supply. The high-frequency component removing filter 3 includes first, second, third and fourth filter capacitors Cf1, Cf2, Cf3 and Cf4, and first, second and third capacitors.
Of the filter reactors Lf1, Lf2, Lf3.
The first and second filter capacitors Cf1 and Cf2 are connected between a pair of lines, and the third and fourth filter capacitors Cf3 and Cf4 are connected between the pair of lines and the ground conductor 8, respectively. . Cf1 and Cf2 are, for example, 0.33
μF, and Cf3 and Cf4 are, for example, 2200 pF.

The input rectifying / smoothing circuit 4 comprises four bridge-connected diodes 4a, 4b, 4c, 4d and a smoothing capacitor 4.
e, and converts the supplied AC voltage into a DC voltage and outputs the DC voltage between a pair of DC lines 9a and 9b.

The first and second switches Q 1 and Q 2 are connected in series with each other, and this series circuit includes a pair of DC lines 9.
a and 9b. First and second diodes
The switches D1 and D2 are connected to the first and second switches Q1 and Q2 in reverse parallel. The first and second capacitors C1, C are also connected in parallel to the first and second switches Q1, Q2. In this embodiment, the source of the first and second switches Q1 and Q2 is an insulated gate field effect transistor having a structure in which the source is connected to the body.
A diode is built in in addition to the T switch. The first and second diodes D1 and D2 are built-in diodes of the FET.
, But are shown here as independent diodes for ease of understanding. Of course, diode D1,
D2 can also be an individual component. The first and second switches Q1 and Q2 have a parasitic capacitance between the drain and the source. The first and second capacitors C1, C2 of FIG.
2 has drains of the first and second switches Q1 and Q2.
The parasitic capacitance between the sources is included. If the parasitic capacitances of the first and second switches Q1 and Q2 are large, individual capacitors for the first and second capacitors C1 and C2 are unnecessary.

The high-frequency transformer T includes magnetic cores 10 and 1
It comprises a secondary winding N1 and a secondary winding N2. The primary winding N1 is connected to a second switch Q2 via a series resonance capacitor Cr.
Are connected in parallel. The primary winding N1 has the leakage inductances Ls1 and Ls2 and the excitation inductance Lp shown in the equivalent circuit of FIG. Secondary winding N2 is center tap 1
1 and is divided into first and second secondary windings N2a and N2b. The primary winding N1 and the secondary winding N2 are wound at different positions on the core 10 as illustrated in FIG. Therefore, the leakage inductances Ls1 and Ls2 are increased, and the stray capacitances Cs7 and Cs8 between the primary winding N1 and the secondary winding N2 are smaller than in the prior art.

A metal fitting is adhered to the iron core, that is, the core 10, and this metal fitting is connected to the cold end side line 9b. The first and second switches Q1 and Q2 are connected to a line 9b as a cold end where no high-frequency component is applied.
And an IC case including the control circuit 7 is also connected.
Further, in this embodiment, in order to surely obtain the effect of reducing the conducted noise, the cold end side of the primary winding N1, ie, FIG.
The lower side close to the resonance capacitor Cr is also disposed on the lower side in FIG. Therefore, in FIG.
1 is closest to the secondary winding N2, and the hot end (the portion where the high frequency is superimposed) is the secondary winding N2.
From far away. The lower DC line 9b shown in FIG. 1 is a cold end, for example, a 2200 pF capacitor Cg.
To the ground conductor 8.

In this embodiment, the primary winding N1 and the secondary winding N
2 and the core 10 is connected to the cold end line 9b, so that the stray capacitance between the primary winding N1 and the secondary winding N2 is extremely small. 5 pF. When the core 10 is not connected to the cold end, the primary winding N1 and the core 10 are connected between the primary winding N1 and the secondary winding N2 in addition to the above-mentioned capacitance of 3.5 pF. (Approximately 8 pF) and the capacitance between the secondary winding N2 and the core 10 (approximately 9 pF), resulting in a combined capacitance of approximately 7.7 pF.

The output rectifying and smoothing circuit 5 includes first and second output rectifying diodes Do1 and Do2 and an output smoothing capacitor Co.
Consists of To form a center tap type full-wave rectifier circuit, the anodes of two diodes Do1 and Do2 are connected to one end and the other end of the secondary winding N2, and the two cathodes are commonly connected. I have. One end of the capacitor Co has two cathodes.
The other end is connected to the center tap 11. A pair of output terminals 6a connected to the capacitor Co,
A load is connected between 6b.

The control circuit 7 includes a first switch Q1 and a second switch Q1.
, And Q2 are alternately turned on and off with a dead time. In addition, in order to control the voltage between the pair of output terminals 6a and 6b to be constant, the control circuit 7 is connected to the pair of output terminals 6a and 6b and detects the voltage there.
The control circuit 7 includes a variable frequency oscillator, and controls the first and second switches Q1, Q2 when controlling the output voltage.
Change the on / off repetition frequency of 2.

FIG. 2 shows only the main part of the switching power supply of FIG. However, in FIG. 2, the input rectifying / smoothing circuit 4 of FIG. 1 is represented by a DC power supply 4 'having a voltage Ei. Also,
The load Ro is connected to the output smoothing capacitor Co.

FIG. 3 shows an equivalent circuit of FIG. Transformer T
Are divided into the leakage inductances Ls1 and Ls2, the excitation inductance Lp, and the ideal transformer. The exciting inductance Lp is connected in parallel with the primary winding N1 of the ideal transformer and the second leakage inductance Ls2. The first leakage inductance Ls1 is the excitation inductance Lp
Are connected in series. Note that an equivalent circuit can be schematically shown ignoring the second leakage inductance Ls2. The capacitor Cq of FIG. 3 is the first and second capacitors of FIG.
Of the capacitors C1 and C2.

When the switching power supply of this embodiment is driven, the first and second switches Q1 and Q2 are switched between the gate and source consisting of a pulse having a duty cycle of about 50% shown in FIG. It is driven by the voltages Vgs1 and Vgs2. Between the pulses of the first and second gate-source voltages Vgs1 and Vgs2, t1 to t1 are used for soft switching.
A dead time is provided as shown in two sections, t6 to t7. The ON / OFF repetition frequency (operating frequency) of the first and second switches Q1 and Q2 is set higher than the series resonance frequency determined by the exciting inductance Lp and the resonance capacitor Cr. For this reason, the load viewed from the switching element becomes inductive impedance, and during the dead time, soft switching is performed using voltage resonance between the inductance Ls1 + Lp, which is the inductive impedance, and the capacitor Cq, which is a parasitic capacitance of the switch. Further, in order to perform soft switching, the gate signal is controlled so as not to input the gate signal until the first and second switches Q1 and Q2 become zero voltage. The control covers the entire area with only the simplest frequency control. The operation is roughly divided into a no-load light-load operation, a heavy-load operation, and a load short-circuit operation. The device of this embodiment can perform ZVS (soft switching) stably in all operations and is efficient over all load regions. It is well known that in a DC-DC converter with current resonance, the output voltage changes in inverse proportion to the switching frequency.

Next, the operation of the switching power supply at no load and light load will be described in detail with reference to FIG. 5 under the following conditions. (A) The capacitors, transformers, and inductances are ideal and the internal resistance is ignored. (B) The switches Q1 and Q2 and the diodes Do1 and Do2 are ideal, and the voltage drop due to the on-resistance and the switching time are zero. (C) The capacitor Cq is connected to the first and second switches Q1
, Q2 in parallel. (D) All stray capacitances and wiring resistances are ignored. FIG. 5 shows the waveform of each part in each period during the no-load light-load operation. There are ten periods during one cycle from t1 to t11, and the first five periods (1 to 5) and the second five periods (6-1)
0) operate symmetrically. As the load approaches, the periods 3, 4, 8, and 9 become shorter, and the output voltage becomes a rectified sinusoidal peak voltage.

[0022]

[Period from t1 to t2] During the period from t1 to t2, a parallel resonance circuit of Lp-Ls1-Cq-Cr is formed in the equivalent circuit of FIG. Before t1, a current flows through the second switch Q2, and when the second switch Q2 is turned off at t1,
Due to the energy stored in the inductance Ls1 + Lp of the transformer, the inductance Ls1 + L of the transformer is obtained.
p and the voltage resonance capacitor Cq resonate and perform soft switching. When the second switch Q2 is turned off, the current flowing there is diverted to the capacitor Cq, and the current load on the second switch Q2 becomes zero. As a result, the switching loss of the second switch Q2 is greatly reduced.
As a result of the flow of energy, the polarity of the voltage resonance capacitor Cp is inverted with the energy of the inductance Ls1 + Lp, and the current resonance capacitor Cr is charged. t
At time 2, the voltage Vq1 of the first switch Q1 becomes zero,
The voltage Vq2 of the second switch Q2 becomes the power supply voltage Ei.
The load Ro is supplied with power from the capacitor Co.

[0023]

[Period from t2 to t3] During the period from t2 to t3, Lp −
A current flows in a closed circuit of Ls1-D1-4'-Cq. During this period, a reverse current flows through the parallel diode D1 of the first switch Q1, which has become zero voltage at time t2. That is, during the period from t2 to t3, the energy stored in the inductance Ls1 + Lp is released through the diode D1. During this period, the gate signal of the first switch Q1 is inputted, and the first switch Q1 is turned on at zero voltage. Due to the energy flow during this period, the inductance Ls1
The energy stored in + Lp and the energy stored in the capacitor Cr are fed back to the input terminal while discharging the capacitor Cr. The period from t2 to t3 ends when the secondary voltage Vt / N of the transformer T becomes equal to the output voltage E0. The load Ro is supplied with electric power from the capacitor Co.

[0024]

[T3 to t4 period] During the t3 to t4 period, the previous t1 to t4
A current path of a closed circuit of Lp-Ls2-N1 and a current path of a closed circuit of N2a-Do1-Co and Ro due to conduction of the diode Do1 are added to the current path for two periods. During this period, power is fed back to the input side and power is also output to the output side. As an energy flow, the energy stored in the inductance Ls1 + Lp and the energy of the capacitor Cr discharge the capacitor Cr while discharging the parallel diode of the first switch Q1.
Both the flow returned to the input side through the node Dq1 and the flow output to the output side occur. The period from t3 to t4 is the first
The process ends when the current of the diode D1 becomes zero.

[0025]

[Period from t4 to t5] During the period from t4 to t5, 4'-Q1-
The circuit of Ls1-Ls2-N1-Cr and 4'-Q1-Ls1-
The circuit of Lp-Cr is formed. The secondary side of the transformer is N22-Do1-Co and Ro which are the same as in the previous section. During this period, the first switch Q1 is turned on, and power is supplied from the power supply 4 'to supply power to the secondary side and charge the resonance capacitor Cr. The period from t4 to t5 ends when the output current Ido1 becomes zero.

[0026]

[Period from t5 to t6] The period from t5 to t6 is a period when the current Ido1 of the first output diode Do1 becomes zero and the secondary winding N2 of the transformer T is in an open state, that is, a no-load state.
During this period, a current flows through the closed circuit of 4'-Q1-Ls1-Lp-Cr, and the capacitor Cr and the inductance Ls1 + L
p is charged. The load Ro is supplied with power from the capacitor Co. This period is ended by turning off the gate signal of the first switch Q1 at t6.

[0027]

[Period t6 to t11] The period t6 to t11 operates symmetrically to the period t1 to t6. Therefore, only the current path in each period is shown, and detailed description is omitted. During the period from t6 to t7, Cq
-Ls1-Lp-Cr voltage resonance circuit is formed, and C
An o-Ro circuit is formed. During the period from t7 to t8, Ls
A closed circuit of 1-Lp-Cr-D2 is formed and Co-R
The circuit of o is formed. During the period from t8 to t9, the previous t7
In addition to the occurrence of the same current path as the period t8,
-Ls2 circuit is formed and N2b-Do2-Co and R
The circuit of o is formed. In the period from t9 to t10, Cr-N
1 -Ls2-Ls1-Q2 closed circuit and Cr-Lp-Ls1-
A closed circuit of Q2 is formed, and the same current path occurs on the secondary side as in the period from t8 to t9. In the period from t10 to t11, C
A closed circuit of r-Lp-Ls1-Q2 and a closed circuit of Co-Ro are generated.

In the switching power supply of this embodiment, both current resonance and voltage resonance are used, and the transformer voltage Vt becomes a trapezoidal wave as a whole as shown in FIG. The harmonic components are reduced as compared with the conventional half-bridge type square wave voltage shown in FIG. Further, zero volt switching of the first and second switches Q1 and Q2 can be achieved, and power loss can be reduced. Further, noise can be reduced.

[0029]

[Mechanism of conducted noise] Conducted noise is noise that is transmitted from a switching power supply to a commercial input line and goes outside, and has a bad influence on other electronic devices. FIG. 7 shows an outline of a measurement circuit for measuring conducted noise and a path of a current flowing between the LISN circuit 20 and the ground G (ground). The noise current flows from the conductive component on which the switching high frequency is superimposed to the ground G through the stray capacitance Cs. The noise current flowing to the ground G returns to the switching circuit through the LISN. At this time, the noise current passing through LISN is 5
Appears at both ends of 0Ω resistors 27-30 and is measured as conducted noise. The LISN circuit 20 includes, for example, two inductors 21 and 22 of 50 μH and a four inductor of 0.1 μF, for example.
It comprises four capacitors 23-26 and four resistors 27, 28, 29, 30 of, for example, 50 ohms. The signal source 31 generates a signal of 60 kHz.

FIG. 8 shows the measurement of stray capacitance and noise of each part of the actual circuit of FIG. The main stray capacitances of the switching power supply include capacitances Cs1 and Cs2 between a pair of DC lines and the ground conductor 8, capacitance Cs3 between the two switches Q1 and Q2 and the ground conductor 8, and secondary transformer. A capacitance Cs4 between the upper end of the winding N2a and the ground conductor 8, a capacitance Cs5 between the pair of DC output lines and the ground conductor 8,
Cs6, the primary winding N1 and the secondary winding N also shown in FIG.
2 between the capacitors Cs7 and Cs8. Cs3, Cs4, and Cs7, which are stray capacitances between the hot end (circuit on which high frequency is superimposed) and the ground G, greatly affect the conduction noise, and are stray between the cold end (circuit on which high frequency is not superimposed) and the ground G. The capacitances Cs1, Cs2, Cs5, and Cs6 have little effect. The noise current transmitted to the ground G is divided by the stray capacitance Cs and the ground capacitor Cg, and after the voltage across the ground capacitor Cg is attenuated through the filter composed of Lf1, Lf2, Lf3, Cf3 and Cf4, the LISN circuit 2
At 0 it is measured as conducted noise. The following methods are conceivable for making this circuit low noise. (1) Reduce the voltage of the noise source. (2) Reduce the number of noise sources. (3) Hot end (conductive parts with high frequency superimposed)
The stray capacitance between the ground and the ground G is minimized. (4) If the stray capacitance cannot be reduced, insert a cold-end conductor between the hot-end conductive component and the ground G to shield it so that no high-frequency current flows to the ground G. (5) Strengthen the line filter.

[0031]

[Noise source] As a noise source, the switching voltage of the main switches Q1 and Q2 is on the primary side, and the switching voltage generated by the transformer T and the rectification diodes Do1 and Do2 are recovered on the secondary side. There is a noise voltage. In addition, there is noise generated by electromagnetic coupling between the transformer T and the line filter. The switching power supply of this embodiment is of a half-bridge type, and its hot-end voltage is about half that of a flyback converter and a forward converter. Therefore, if the stray capacitance is the same as that of the flyback or forward converter, the noise current is reduced to about 1/2. Further, the voltages Vq1 and Vq2 of the main switches Q1 and Q2 of the present embodiment have a waveform like a trapezoidal wave in FIG. For this reason, the harmonic content of the switching frequency is smaller than that of the hard switching which has a waveform like a rectangular wave in FIG. According to the present embodiment, when the fundamental wave is set to 60 kHz, the harmonics become extremely small at a frequency of 1.8 MHz or higher. Also, the secondary side output diode Do1,
Since the current of Do2 flows based on the current resonance between the capacitor Cr and the reactor Ls1 + Ls2, the diodes Do1, D2
The noise generated from the water is relatively small.

[0032]

[Hot end, Cold end] The main circuit of this embodiment is composed of a power IC (STR-Z4304) manufactured by Sanken Electric Co., Ltd., which includes FET switches Q1 and Q2. This IC does not generate noise current due to stray capacitance because the case is at the cold end.
Since the radiator can be formed at the cold end, it is very advantageous for low noise. Normally, in flyback converters and forward converters, the drain of the FET becomes a hot end and the radiator also becomes a hot end, so that a large amount of noise current is generated due to stray capacitance.

[0033]

[Measurement Results] FIG. 9 shows the conduction noise measurement results of the stitching power supply according to the present embodiment having an input voltage of 100 V, an output voltage of 15 V, an output current of 3.3 A, and an output power of 50 W, and FIG. Show. FIG. 11 shows dark conduction noise in an anechoic chamber used for measurement, and FIG. 12 shows dark radiation noise. As is clear from this, according to the present embodiment, an ultra-low noise power supply device can be provided. That is, according to this embodiment, the conducted noise is 4 times higher than that of class A.
The noise level can be 0 dB lower and 30 dB lower than class B. For comparison, FIG.
4 shows conduction noise of a W switching power supply. The conduction noise of this embodiment of FIG. 9 is significantly lower than the conventional conduction noise of FIG. The noise level of the present embodiment was lower than the noise level generated from the commercial rectification diode of the dropper power supply composed of the commercial transformer and the transistor.
Also, the leakage current that tends to increase when the filter is strengthened is A
C was measured at 100 V and 50 Hz, and was able to be suppressed to 0.2 mA.

FIG. 14 shows the relationship between the AC input voltage and the switching frequency of the switching power supply of the present embodiment for no load, light load, and full load. As is apparent from this, the switching frequency of the first and second switches Q1 and Q2 increases as the load decreases, and increases as the input voltage increases.

FIG. 15 shows the relationship between the output current and the efficiency of the switching power supply of the present embodiment in terms of the AC input voltage (INPU).
T) 85V, 100V and 132V are shown separately. In a conventional forward converter or the like, the efficiency decreases when the input voltage increases, but in the present embodiment, the efficiency decreases when the input voltage is low. The reason for this is that, because of the step-up type in which the voltage is stepped up beyond the turns ratio of the transformer, if the input voltage is low, the exciting current of the exciting inductance Lp3 is increased to step up.

The noise reduction means employed in this embodiment will be summarized below. (1) The core 10 of the transformer T was connected to a cold end. (2) The stray capacitance between the primary winding N1 and the secondary winding N2 can be reduced by arranging the primary and secondary windings N1 and N2 of the transformer apart from each other and connecting the core 10 to the cold end. It was about 1/20 of the transformer of the forward converter.
Transformer of forward converter is usually 50-100
In the case of the transformer of the present invention, the core 10 was connected to the cold end in the transformer according to the present invention, and the pF was increased to 3.5 pF. (3) All the switches Q1 and Q2 of the converter were operated by voltage soft switching. (4) The output voltage from the converter is reduced to about 1/2 that of the forward converter. (5) The case and fin of the power IC were used as cold ends. In the forward converter, the drain of the body of the FET becomes a hot end. (6) The body of the diode for output rectification is cold-ended. (7) The current waveforms of the output diodes Io1 and Io2 are changed to current resonance waveforms. (8) Two windings N2a and N2 of the secondary winding N2 of the transformer
By making b a bifilar winding, the leakage inductance between them was reduced. (9) Avoidance of the noise filter and the transformer. (10) The approach of the input terminal and the output terminal was avoided. (11) The noise filter has been strengthened. (12) The capacity between the input and output of the noise filter is reduced by division winding. (13) The distance between the transformer T and the rectifier diodes Do1 and Do2 is minimized. (14) A ferrite bead is inserted in the main switch. That is, a ferrite bead is inserted in the drain current path and the output terminal of the switch Q2. (15) CR (4) is connected in parallel with the output diodes Do1 and Do2.
70 pF + 2.2Ω). Note that only C may be used. (16) 10 parallel to the power supply diode of the control circuit 7
A 0 pF capacitor was connected. Note that CRs may be connected in parallel. The above (13) to (16) are sometimes effective as measures against high frequency noise.

The ultra-low-noise, small-sized, light-weight, and high-efficiency power supply device of the present embodiment is a medical device, a measuring device, a radio communication device, or a device that handles low-level signals, which could only use a dropper power supply due to noise problems. Etc. can also be used. In addition,
The external dimensions of the switching power supply of this embodiment are shown below.
The width is 125 mm, the depth is 80 mm, and the height is 12.5 mm, which is extremely small.

[0038]

[Modifications] The present invention is not limited to the above-described embodiment, and for example, the following modifications are possible. (1) When sufficient leakage inductances Ls1 and Ls2 cannot be obtained, individual inductors can be connected in series to the resonance capacitor Cr. (2) In FIG. 1, the capacitors C1 and C2 are parasitic capacitances of the first and second switches Q1 and Q2, and a separate capacitor can be connected in parallel to the second switch Q2 for the purpose of supplementing the parasitic capacitance. (3) The core 10 can be connected to the DC line 9b in FIG.

[Brief description of the drawings]

FIG. 1 is a circuit diagram illustrating a switching power supply device according to a first embodiment.

FIG. 2 is a circuit diagram showing a main part of the switching power supply device of FIG.

FIG. 3 is an equivalent circuit diagram of FIG.

FIG. 4 is a sectional view schematically showing the high-frequency transformer of FIG.

FIG. 5 is a waveform diagram schematically showing output voltages of the switching circuit of FIG. 1 and a conventional switching circuit.

FIG. 6 is a waveform diagram schematically showing a state of each unit in FIG. 1;

FIG. 7 is a circuit diagram for explaining noise measurement.

FIG. 8 is a circuit diagram showing a flow of stray capacitance and noise of the switching power supply device of FIG. 1;

FIG. 9 is a diagram showing a conduction noise characteristic of the first embodiment.

FIG. 10 is a diagram illustrating radiation noise characteristics of the first embodiment.

FIG. 11 is a diagram showing a conduction noise characteristic of an anechoic chamber.

FIG. 12 is a diagram illustrating radiation noise characteristics of an anechoic chamber.

FIG. 13 is a diagram showing a conduction noise characteristic of a conventional switching power supply device.

FIG. 14 is a diagram illustrating a relationship between an input voltage and a switching frequency according to the first embodiment.

FIG. 15 is a diagram showing the relationship between output current and efficiency in the first embodiment.

[Explanation of symbols]

 Q1, Q2 First and second switches T transformer Cr Current resonance capacitor C1, C2 Voltage resonance capacitor N1, N2 Primary and secondary windings Cs7, Cs8 Floating capacitance 10 core

────────────────────────────────────────────────── ─── Continuation of the front page (56) References JP-A-8-111977 (JP, A) JP-A-9-289774 (JP, A) JP-A-10-327577 (JP, A) Morita, Novel Ultra Low-noise soft switch-mode Power Supply, (58) Fields investigated (Int. Cl. ⁷ , DB name) H02M 3/28 H02M 3/335

Claims (3)

(57) [Claims] A DC power supply and a pair of DC power supplies of the DC power supply.
Series circuit of first and second switches connected between terminals
And a core and a primary winding and a secondary winding wound on the core
And an output transformer having:
Resonant capacitor connected in parallel through the primary winding
Output rectifying / smoothing circuit connected to the secondary winding
And the first and second switches are turned on and off alternately.
A control circuit for controlling the primary winding and the secondary winding are arranged at different positions of a transformer core so as to reduce stray capacitance therebetween.
The primary winding is arranged such that the end of the primary winding on which the high frequency is not superimposed is closer to the secondary winding than the end of the primary winding on which the high frequency is superimposed. Wherein the core is connected to a circuit conductor on which a high frequency is not superimposed. 2. The apparatus according to claim 1, wherein said first and second switches are parallel to each other.
First and second capacitors or parasitic capacitances connected to a column
And the control circuit has a first switch and a second switch.
It is configured to perform on / off control alternately
It is, the first and second switches of data - during turnoff the first
The voltage between both terminals of the first and second switches is gradually increased
Start up each time and turn off the first and second switches
When switching on, during the dead time,
The voltage between both terminals of the first and second switches is set to zero or almost zero.
So that the first and second capacitors can be
Or the parasitic capacitance and the inductance of the primary winding and the
2. The switching power supply according to claim 1, wherein a time width of the dead time is set . 3. The circuit conductor on which the high frequency is not superimposed is
Conductor for connecting the second switch to the DC power supply
3. The switching power supply device according to claim 1, wherein: