JP3049624B2 - Waveform shaping filter and waveform shaping high-pass filter - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a digital subscriber line interface apparatus for transmitting high-speed digital data simultaneously or alternately in both directions by using a metallic pair cable, which is an existing telephone subscriber line. The present invention relates to a waveform shaping filter and a waveform shaping high-pass filter used in processing of a received signal, and in particular, to shape a waveform of a received signal and to shape a waveform of an isolated wave response so that data sampling timing can be controlled. The present invention relates to a waveform shaping filter and a waveform shaping high-pass filter incorporating the function of the high-pass filter.

[0002]

2. Description of the Related Art In order to spread a digital communication network such as ISDN, a digital subscriber line transmission system for transmitting high-speed digital data using an existing metallic pair cable used for transmitting voice band signals has been developed. Is being developed. In this case, inter-symbol interference distortion caused by fluctuation of transmission loss due to a difference in distance between subscriber lines, limitation of a pass band, reflection by an open-ended bridge tap (BT) existing in an existing subscriber line, and the like are given. It needs to be addressed. As a transmission method, a so-called ping-pong transmission method in which transmission and reception are alternately performed in a time-division manner, and an echo canceller method in which transmission and reception are simultaneously performed using a hybrid circuit and an echo canceller have been developed.

FIG. 6 is a block diagram showing a schematic configuration of a conventional digital subscriber line interface device employing the echo canceller system. In FIG. 6, a received signal received via the subscriber line 10 is a two-line signal.
The signal is converted into a digital signal by an oversampling A / D converter 14 through a hybrid 12 for four-wire conversion, and is generated by a echo canceller 22 from transmission data in a subtracter 20 through a waveform shaping filter 16 and a high-pass filter 18. By subtracting the obtained pseudo echo,
The wraparound of the transmission signal in the hybrid circuit 12 is removed, the attenuation characteristic of the transmission path is corrected in the AGC circuit 24, the intersymbol interference distortion is removed in the decision feedback equalizer (DFE) 26, and for example, ± 1. , ± 3 of 4
The value is determined and output. Some of the coefficients calculated by the decision feedback equalizer 26 are used for timing control of the timing control unit 28. As the oversampling A / D converter 14, a delta-sigma converter 27 and a decimation filter 29 cascade connection are used.

The waveform shaping filter 16 will be described later. The high-pass filter 18 is provided to reduce the number of taps of the echo canceller 22 and simplify the circuit. Since the impedance characteristic in the line direction from the hybrid circuit 12 is not ideal, the hybrid circuit 1
In 2, the transmission signal wraps around to the receiving side to generate an echo. An echo canceller 22 is provided to eliminate the echo component. The number of taps serving as an index indicating the size of the echo component is determined by the duration of the solitary wave response of the echo. The digital subscriber line interface device uses a hybrid transformer or a similar two-wire to four-wire converter as the hybrid circuit 12,
Since they are equivalent to a high-pass filter having a cutoff frequency of several hundreds Hz or less, the solitary wave response of the echo has a very slowly attenuating waveform.
A large-scale echo canceller having 100 to 200 taps for a symbol frequency of 0 kHz is required. Therefore, the high-pass filter 18 having a higher cutoff frequency removes low frequency components, thereby speeding up echo attenuation and reducing the number of taps of the echo canceller to 20 to 30.
To the extent that is done. As shown in FIG. 6, the high-pass filter 18 has a transfer function (1−1) that takes the difference between samples separated by one sample period.
The simple one with z ^-1 ) is used.

The decision feedback equalizer 26 has a configuration as shown in FIG. The output of the subtracter 30 is determined by the determiner 32 and ±
The result is output as a determination result of 1, ± 3, and the difference before and after the determination is taken by the subtractor 34 to be used as a determination error. The coefficient calculation circuit 3 determines the determination error and a signal obtained by delaying the determination result by cascade connection of a plurality of delay units 36 having a delay time T of one symbol.
8 by the coefficient _{C -1, C 0, C 1} , ... calculates the C _m, of which C _1, ... multiplied by the delayed signal of the determination result in the multiplier 40 to C _m (post-cursor value) coefficients, summing The feedback signal is subtracted from the input signal by the subtractor 30 and is input to the decision unit 32. Coefficient operation circuit 38
In the calculation of the coefficient in, for example, a well-known LMS (Least
mean square) algorithm is used.

In the coefficient operation circuit 38, the coefficients C _-1 ,
When C ₀ , C ₁ ,..., C _m converge, they correspond to sample values of the solitary wave response of the transmission path up to the input of the decision feedback equalizer 26. The post-cursor values C ₁ ,..., C _m are multiplied by the delayed signals and superimposed to generate a backward intersymbol interference distortion wave, and the subtractor 30
It is eliminated. On the other hand, the coefficients C _-1 and C ₀ are called a precursor value and a main cursor value, respectively.
These times are called a pre-cursor and a main cursor, respectively. The precursor value C _-1 is used for timing control in the timing control unit 28 as described below.

When the waveform shaping filter 16 is not provided, the solitary wave response of the transmission line up to the input of the decision feedback equalizer 26 is as shown in FIG. Therefore, the waveform shaping filter 16
8B, the characteristic changes from negative to positive in the vicinity of the precursor, that is, in the precursor one sample before the main cursor where the solitary wave response amplitude is maximum, as shown in FIG. 8B. In this way, if the precursor value C ₁ calculated by the decision feedback equalizer 26 is negative, the sample timing is delayed, and if the precursor value C ₁ is positive, the control is advanced by the timing control unit 28, and the clock controlled in this manner is controlled. It is used in each unit as an operation clock.

[0008]

Conventionally, a transversal filter having about 7 to 9 taps has been used for waveform shaping. To completely perform the waveform shaping as described above, it is not necessary to operate at the symbol frequency. It was believed that it was necessary to operate at twice or three times that frequency. Normally, there is a limit to complete the waveform shaping as described above by controlling the amplitude of the received signal for each symbol period. The amplitude at the intermediate point or at a point obtained by dividing the symbol period into three equal parts is also controlled. It is considered necessary to do so.

For this reason, conventionally, the frequency of the output of the decimation filter 29 of the A / D converter 14 also needs to be twice or three times the symbol frequency. Here, the decimation filter 29 removes the high-frequency component of the 1-bit quantized signal at a high frequency of nearly 200 times the frequency of the A / D conversion output in the ΔΣ converter 27 to obtain a multivalued digital value. Filter.
It is necessary to operate at a frequency close to 200 times the frequency of the A / D conversion output itself, and the scale of the transversal filter is about 300 taps.
It consisted of dedicated hardware. In general, the hardware scale of a digital transversal filter has a strong correlation with the product of the frequency of input data, the frequency of output data, and the number of taps, and the larger the product, the larger the scale.
Therefore, the fact that the repetition frequency of the output data of the decimation filter 29 must be twice or three times the symbol frequency makes it difficult to reduce the size of the decimation filter. In addition, the fact that the repetition frequency of the output data of the decimation filter is high means that the filter operates at high speed, which makes it difficult to use multiple hardware, which not only increases the actual circuit scale but also increases power consumption. It also leads to the problem of doing.

Conventionally, a decimation filter 2
9 itself has been provided with a waveform shaping function. When the decimation filter is provided with a waveform shaping function, the size of the filter becomes large and a transversal filter having at least about 600 taps is obtained. However, the output cycle is different from the conventional example shown in FIG. In view of the above, the size of the actual hardware can be the same as that of the conventional decimation filter shown in FIG. As a result, the waveform shaping filter 16 required in the example of FIG. 6 becomes unnecessary, and the circuit scale is reduced accordingly. However, in this case, the tap coefficients of the decimation filter become complicated numbers, and it is difficult to generate the tap coefficients using random logic, and a ROM for storing the tap coefficients is required. When a ROM is used, the power consumption is considerably larger than when a random logic is used, and there is a problem that the effects of reducing the circuit scale and reducing the power consumption as expected as a whole cannot be obtained.

Accordingly, it is a first object of the present invention to provide a filter circuit for a digital subscriber line interface device which overcomes the above problems and has a small circuit size and low power consumption. A second object of the present invention is to provide a waveform shaping high-pass filter having a high-pass function for reducing the number of taps of an echo canceller, in addition to the above-described waveform shaping function.

[0012]

According to the present invention, there is provided a waveform shaping filter provided on a receiving side of a transmission line for transmitting a multilevel signal consisting of at least two values at a predetermined symbol period T. A transversal waveform shaping filter for shaping a waveform of a solitary wave response up to an input of an intersymbol interference compensator so that its polarity is inverted near T time before its maximum value, and a transfer function expressed by z-transform. H (z) is characterized in that H (z) = 1−az ⁻¹ , where a> 1.

Further, the waveform shaping high-pass filter of the present invention provides an input to an intersymbol interference compensator provided on a receiving side of a transmission path for transmitting a multilevel signal composed of at least two values at a predetermined symbol period T. A transversal waveform shaping high-pass filter that shapes the solitary wave response waveform up to T time before its maximum value so that the polarity is inverted, and passes only the high-frequency component of the signal. , The transfer function H (z) expressed by z-transform is H (z) = −
1 / (1 + a) + z ⁻¹ -a / (1 + a) · z ⁻² , where a> 1.

[0014]

The fact that the transfer function H (z) is 1-az ^-1 means that the transfer function is input from the viewpoint of 1-az ^-1 = -a (-1 / a + z ^-1 ) in the time domain. Signal (-1 / a)
This corresponds to a process of adding the multiplied waveform to the waveform of the input signal delayed by one cycle T of the signal later and multiplying by (-a). That is, assuming that the solitary wave response before shaping has a waveform shown by a solid line in FIG. 1A, a waveform multiplied by (−1 / a) becomes a waveform shown by a broken line in FIG. When the two are combined, the result is as shown in FIG.
The waveform is shaped so that the polarity is inverted near the period before. Note that for (−1 / a) times, only the overall gain changes and there is no effect on the waveform of the solitary wave response. In addition, the point where the polarity is inverted is slightly closer to the maximum value side (main cursor side) than one cycle before, and this is more remarkable as a is closer to 1, but the value of a is set to a value sufficiently larger than 1. , The deviation is small, and even if the sample timing is controlled based on this point, there is no problem in the processing of waveform equalization and value discrimination.

The filter represented by the above transfer function can be realized by a two-tap filter that moves at the symbol frequency, and can be significantly simplified as compared with the conventional system. Further, as described above, the high-pass filter provided after the waveform shaping filter has a transfer function of 1-z ^-1 . Therefore, the combined function H (z) becomes H (z) = (1-az ^-1 ) (1-z ^-1 ) = 1- (1 + a) z ^-1 + az ^-2 , and H (z) × ( −1 / (1 + a)) is replaced by H
(Z), H (z) = ^-1 / (1 + a) + z ^-1 -a / (1 + a) .z- ² , and the filter having this transfer function has the above-described waveform shaping function and high-pass function. This is a three-tap transversal filter that also has

[0016]

FIG. 2 is a block diagram of an echo canceller type digital subscriber line interface device using a waveform shaping filter 50 according to an embodiment of the present invention. 6 is the same as FIG. 6 except that a transfer function H (z) = 1−az ⁻¹ , where a> 1... (1) is used as the waveform shaping filter 50.

As described above, when the value of a is close to 1, the displacement of the position where the polarity is inverted becomes large, and the post-cursor values C ₁ , C ₂ ... _Cm are the main cursor values. Since it is equal to or more than 1/2 of C ₀ , equalization in the decision feedback equalizer 26 becomes difficult. If a is too large, 1 / a
Becomes small, the effect of polarity reversal is reduced, and the polarity reversal cannot be obtained on a transmission line having a large precursor value C _-1 before waveform shaping (see FIG. 8A), and the timing cannot be extracted. Therefore, a value that is sufficiently larger than 1 and causes a sign inversion near the precursor is appropriately selected according to the characteristics of the transmission path to be applied. For example,
Regarding the state of the 16 transmission lines in the US standard,
Computer simulation including the characteristics of the hybrid transformer 12, the decimation filter 29, etc. of the own apparatus is performed to calculate 16 kinds of solitary wave responses, and it is determined whether or not the timing extraction and the equalization by the decision feedback equalizer 26 are easy as a whole. To determine an appropriate value of a.

As described above, the high-pass filter 18 has a transfer function 1-z ^-1 (2). The waveform shaping filter 50 and the high-pass filter 18 may be used as separate blocks as shown in FIG.
Since these filters are both transversal filters operating at the symbol frequency, it is more efficient to operate them as one filter.

If a filter having a synthesis function is called a waveform shaping high-pass filter, its transfer function H
(Z ^-1 ) is represented by the following equation. Multiplying H (z ^-1 ) = (1-az ^-1 ) (1-z ^-1 ) = 1- (1 + a) z ^-1 + az ^-2 only corresponds to changing the overall gain. putting a -1 / (1 + a) again and multiplied by H (z ^{-1) from, H (z -1) = -} 1 / (1 + a) + z -1 -a (1 + a) z -2 ... (3 ). If this is represented by a block diagram, a three-tap transversal filter is configured as shown in FIG.

If (1 + a) is set to be a power of 4, 8, 16, and 2 as the value of a in equation (3), such as 3, 7, 15,..., -1 / (1 + a) becomes It is a power of two. Since −a / (1 + a) = 1−1 / (1 + a), −a / (1 + a) is also expressed as a difference of a power of 2, so that the operation of this filter is a numerical operation block composed of an adder and a shifter. (ALU) can be processed in several steps.

As a specific example, when a = 7, the transfer function is as follows. H (z ⁻¹ ) = − 0.125 + z ⁻¹ −0.875 z ⁻² (4) = − 2 ⁻³ + z ⁻¹ + (2 ⁻³ −1) z ⁻² (5) FIG. 4 shows an example of a solitary wave response at the output of a filter having a transfer function. Referring to this figure, the amplitude of the signal changes from negative to positive at approximately one cycle before the peak time of the signal, indicating that the characteristic at the precursor point can be used for timing control. I have.

When a = 7, the tap coefficient of the transversal filter is represented by a combination of exponential constants of 2 as shown in equation (5), and the processing of the filter represented by equation (4) is shown in FIG. Two-input adder 52 and shifter 54
When the processing is performed using the ALU (arithmetic unit) 56 composed of the following, the processing ends in the following four steps. The polarity of the latest input data is inverted, shifted right by 3 bits, and added to zero.

The input data one cycle before is added to the above addition result. The input data two cycles before is inverted and added to the above addition result. 3 bits of input data two cycles before to the above addition result
Shift right and add. In the above processing, “1” is used instead of
Invert the latest input data to the input data before the cycle and
shift right and add ".

The ALU 56 shown in FIG. 5 stores the value set in the register A in accordance with the set shift number in the shifter 54.
, And the value set in the register B is added to the adder 52.
And outputs the result to the output register 62, which is also used in other processes such as the echo canceller 22. It is clear that multiplication of -0.875 can be performed in a single processing when a more complicated three-input adder must be used as the ALU for other processing reasons. (3)
This is a filter function in which, for example, the value of a in the expression is 4.33333.

H (z ⁻¹ ) = − 0.1875 + z ⁻¹ −0.8125 z ⁻² = − (0.125 + 0.0625) + z ⁻¹ − (1−0.125−0.0625) z ⁻² . (6) can be used if the number of processes is allowed to increase somewhat. When a = 8 in the equation (3), the transfer function is as follows.

H (z ⁻¹ ) = − 1/9 + z ⁻¹ −8 / 9z ⁻² = −1 + 9z ⁻¹ −8z ⁻² H (z ⁻¹ ) = − 0.125 + 1.125z ⁻¹ −z ⁻² (7) It is clear that equation (7) is also a waveform shaping high-pass filter having the same amount of calculation processing as equation (4).

It is desirable that the amplitude of the solitary wave response at the post-cursor be equal to or less than half the amplitude of the main cursor, and it is necessary to increase the value of a in such a manner. However, depending on the transmission path, the value of a close to 1 must be used for polarity reversal, so that the value of the post cursor becomes half or more of that of the main cursor. In this case, if the filter having the transfer function 1- (1 / a) z ^-1 is further connected in cascade, the value of the post cursor can be reliably reduced to １ ／ or less.

[0028]

By applying the present invention to a digital subscriber line transmission interface device, the output period of the decimation filter can be improved at the symbol frequency.
The circuit scale of the decimation filter can be reduced. Also, since it is not necessary to provide the decimation filter with a waveform shaping function, the tap coefficient generation circuit of the decimation filter can be configured using simple random logic, compared to the case where coefficients are generated using ROM. Power consumption can be significantly reduced.

[Brief description of the drawings]

FIG. 1 is a diagram for explaining the operation of the present invention.

FIG. 2 is a block diagram of a digital subscriber line interface device using a waveform shaping filter according to an embodiment of the present invention.

FIG. 3 is a block diagram showing a configuration of a waveform shaping filter and a waveform shaping high-pass filter of the present invention.

FIG. 4 is a diagram illustrating an example of a solitary wave response at an output of a filter having a transfer function of Expression (4).

FIG. 5 shows A used in the operation of the filter of the present invention.
FIG. 3 is a diagram illustrating a configuration of an LU.

FIG. 6 is a diagram illustrating a conventional digital subscriber line interface device.

FIG. 7 is a block diagram illustrating a configuration of a decision feedback equalizer.

FIG. 8 is a diagram for explaining waveform shaping.

[Explanation of symbols]

 Reference Signs List 12 hybrid 14 oversampling A / D converter 16, 50 waveform shaping filter 18 high-pass filter 22 echo canceller 26 decision feedback equalizer

────────────────────────────────────────────────── ─── Continuation of the front page (72) Inventor Kuroko saki Yuko 1015 Ueodanaka, Nakahara-ku, Kawasaki City, Kanagawa Prefecture Inside Fujitsu Limited (56) References JP-A-64-67764 (JP, A) JP-A Sho 64-4111 (JP, A) JP-A-59-94986 (JP, A) JP-A-4-249429 (JP, A) JP-B-59-3046 (JP, B2) (58) Fields investigated (Int. Cl. ⁷ , DB name) H03H 15/00-17/08 H04B 3/04-3/23

Claims (5)

(57) [Claims] 1. A waveform of a solitary wave response up to an input of an intersymbol interference compensator provided on a receiving side of a transmission path for transmitting a multilevel signal composed of at least two values at a predetermined symbol period T is represented by a maximum value. A transversal waveform shaping filter for shaping the polarity so as to be inverted in the vicinity of the time before T time, wherein a transfer function H (z) expressed by z-transform of the filter is H (z) = 1-az ^-1. Wherein a> 1. 2. A waveform of a solitary wave response up to an input of an intersymbol interference compensator provided on a receiving side of a transmission path for transmitting a multilevel signal composed of at least two values at a predetermined symbol period T is represented by a maximum value. A transversal waveform shaping high-pass filter that is shaped so that the polarity is inverted near T time before and allows only the high-frequency component of the signal to pass, and a transfer function expressed by z-transformation of the filter. H (z) is H (z) = − 1 / (1 + a) + z ⁻¹ −a / (1 + a) ·
A waveform shaping high-pass filter, wherein z ^-2 , where a> 1. 3. A filter according to claim 1, wherein all tap coefficients of the transversal filter are represented as powers of two or a sum or difference of two powers of two. 4. A transfer function H of a transversal filter.
The waveform shaping high-pass filter according to claim ² , wherein (z) is H (z) = -0.125 + z- ¹ -0.875z- ² . 5. A transfer function H of a transversal filter.
The waveform shaping high-pass filter according to claim ² , wherein (z) is H (z) = -0.125 + 1.125z- ¹ -z- ² .