JP2960074B2 - Adaptive half bridge and impedance meter - Google Patents

Description

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a half bridge including a feedback amplifier and an impedance meter using the same, and more particularly, to a technique that enables stable and accurate impedance measurement over a wide band.

<Prior art and its problems> In precision measurement of electric elements, a half-bridge impedance meter using a feedback amplifier has been awarded. For example, there are many commercial products such as 4261A LCR meter, 4275A multi-frequency LCR meter, 4191A RF impedance analyzer, and 4194A impedance / gain phase analyzer which are commercially available from Yokogawa Hewlett-Packard Co., Ltd.

FIG. 3 is a schematic circuit diagram showing the function of an impedance meter controlled by a microprocessor (not shown).
The device under test (hereinafter referred to as DUT) Z is represented as a two-terminal device, but may have other terminals. For example, in a three-terminal device, the third terminal is grounded or biased, and the characteristics (for example, transfer immittance characteristics) for the remaining first and second terminals are measured.

Terminal B of DUTZ (although may have a number of sine waves, considered assuming that the occurring of one of the sine wave) sinusoidal signal source OSC is driven via the cable L ₁ from.

The signal on the terminal B is inputted to the buffer B ₁ via the cable L _2, it is supplied to the switch S _o. The other terminal A of DUTZ is connected to the inverting input terminal of the amplifier A _G via the cable L ₃ as a virtual ground. Terminal A is also connected to one terminal of the reference resistor R _r via the cable L _4, the reference resistance R _r
The other terminal C of which is connected to a switch S _o via the buffer B _3. The amplifier _AG has a non-inverting input terminal grounded, and forms a current-voltage converter (IV converter) having a resistor _Rf connected between the inverting input terminal and the output.

The output of the I-V converter is introduced to the terminal C via a narrow band amplifier NBA and buffer B _2. The cable L ₃ , the amplifier A _G , the narrow-band amplifier NBA, the buffer B ₂ , the reference resistor R _r , and the cable L _{4 form} a negative feedback loop, and are particularly called null loop (NL).

Ideally, the phase transition (or rotation) of NL is 180 °
(Π radians).

FIG. 4 is a detailed block diagram of the narrow band amplifier NBA of FIG. Increase the amplification of the narrow band amplifier NBA (for example, 180
dB), a modulation amplifier is used.

I-V converter output is input to the terminal I ₁ phase detector
Input to PD ₁ and PD ₂ . The detector signals of the phase detectors PD ₁ and PD ₂ are a 0 ° demodulated signal and a 90 ° demodulated signal, respectively, and have a phase difference of 90 ° and have the same frequency as the signal of the signal source OSC.

The outputs of the phase detectors PD ₁ and PD ₂ are integrated and amplified by the respective integrators, and the DC components are input to the modulators VG ₁ and VG ₂ of the respective vector generators. Modulation signals are 90 ° to each other
At 0 ° for modulation signal and 90 ° modulated signal having a phase difference, supplied from the respective 0 ° modulated signal source MO _1, 90 ° modulated signal source MO _2.

In Figure 4, the integration amplifier resistor R ₁ is the phase detector PD _1, the capacitor C _1, consists DC amplifier A _1, the resistor R ₂ is the phase detector PD _2, a capacitor C ₂ DC an amplifier A _2.

The 0 ° demodulated signal and the 90 ° demodulated signal are supplied from the 0 ° demodulated signal source DM ₁ and the 90 ° demodulated signal source DM ₂ and controlled to have a phase difference of θ with the 0 ° modulated signal and the 90 ° modulated signal, respectively. You. Therefore, the output of the modulator VG _1, buffer B ₂ obtained by adding the outputs of VG ₂ appears at the output terminal 0 _1.

Assuming that the 0 ° modulation signal and the 90 ° modulation signal and the 0 ° demodulation signal and the 90 ° demodulation signal have the same magnitude, the input terminal
Input and output signals from the output terminal 0 ₁ to I ₁ is the phase difference θ
Having.

FIG. 3 is referred to again. The outputs of the buffers B ₁ and B ₃ are alternately input to the phase detector PD ₀ by the switch S ₀ , detected by the quadrature signal generator DT, and input to the integrating voltmeter VRD. The integrating voltmeter VRD measures the outputs of the buffers B ₁ and B ₃ by orthogonal decomposition and further measures the immittance of DUTZ. These are commonly performed in the aforementioned devices.

Now, an increase in signal frequency, an increase in cable length, and a change in DUT immittance change the loop phase transition of the null loop (NL), and eventually the null loop becomes unstable. Conventionally, this problem has been solved by changing the aforementioned θ with the signal frequency.

The adjustment of θ is performed by software or hardware, and the value of θ is determined experimentally.

However, the null loop includes cables L ₁ , L ₂ , L ₃ , and L ₄ for external connection as shown in FIG. ₃ , which deviates from the ideal characteristics of the IV converter and the influence of the DUT. Also,
The terminal A is shifted from the virtual ground state. Also the cable
When a cable different from the standard cable is used for L _{1 to} L ₄ , an unstable state may occur similarly.

<Object of the Invention> Accordingly, an object of the present invention is to realize a half bridge and an impedance meter having a function of stabilizing a null loop in accordance with a measurement state.

<Summary of the Invention> In one embodiment of the present invention, a null loop is opened between a modulating unit and a demodulating unit of a narrow band amplifier constituting a null loop, and a loop phase transition of the null loop is measured. From the measured round phase transitions, additional phase transitions (positive and negative) needed to stabilize the null loop are calculated. The calculated additional phase shift is realized by controlling the phase difference between the modulated signal and the demodulated signal.

<Embodiment of the Invention> In one embodiment of the present invention, the narrow band amplifier shown in FIG.
The NBA, that is, the part corresponding to the circuit of FIG. 4 is newly constructed and becomes as shown in FIG. 1 and 4, components having functions and performances equivalent to each other are denoted by the same reference numerals and symbols.

4, the newly added components are switches S ₁ , S ₂ , S ₃ , S ₄ , resistors R ₃ , R ₄ , and DC power supply E. Since an integrating voltmeter VRD is used for measuring the loop phase transition, it is shown in FIG. Integral voltmeter VRD
Is the same as in FIG. 3, and the ratio of the input DC voltage can be measured. It may be possible to measure the input DC voltage value itself. Therefore, when the input DC voltage ratio corresponds to the real part and the imaginary part of the AC, the phase angle of the AC is obtained by a control / calculation device including a microprocessor (not shown).

The above embodiment operates as follows. First, in order to obtain a test state, the switches S ₁ , S ₂ , S ₃ , and S ₄ are closed to the respective terminals T. (It is closed on the terminal N side in the figure). Others are the same as the measurement state, and the DUT is connected. Sinusoidal signal source OSC, 0 ° demodulator signal source DM _1, 90 °
Demodulated signal source DM ₂ , 0 ° modulated signal source MO ₁ , 90 ° modulated signal source MO
Each output of ₂ is a sine wave signal = 0, 0 ° demodulated signal s
in (wt + θ ₁ ), 90 ° demodulated signal cos (wt + θ ₁ ), 0 ° modulated signal sin− (wt + θ ₂ ), 90 ° modulated signal cos (wt + θ ₂ )
It is. Here, it is assumed that the amplitudes are all 1, but slight differences in the amplitude are irrelevant to the gist of the present invention. W is an angular frequency, t is time, θ ₁ and θ ₂ are phase angles, and θ ₁ is usually set to zero.

Further, these modulation / demodulation signals may be non-sinusoidal waves such as rectangular waves. Usually, a rectangular wave signal is used. These square wave signals have the same effect as the sine wave which is the fundamental wave when the modulation product by the harmonic is small. This is the case in the null loop related to the present invention, and it is not hindered to understand the gist of the present invention even if it is considered as a sine wave. Since the input of the modulator VG _1, VG ₂ is a DC voltage from the DC power supply is input, ignoring the absolute magnitude, sin to an output terminal 0 ₁
(Wt + θ ₂ + π / 4) is output. The input I ₁ of the narrow-band amplifier NBA, signal sin (wt have been further phase shift in an external circuit +
θ ₂ + π / 4 + θ _x ) is fed back. theta _x is a phase shift amount of the external circuit. This signal is demodulated by demodulators PD ₁ and PD ₂ , respectively, is subjected to smoothing DC amplification, and is output from amplifiers A ₁ and A ₂ . Output DC voltage V ₁ of the amplifier A ₁ is _{cos (θ 2 + π / 4} + θ x
−θ ₁ ).

For simplicity, choose R ₁ = R ₂ , R ₃ = R ₄ , C ₁ = C ₂ , Becomes

Therefore, δ = θ ₂ + π / 4 + θ _x -θ ₁ = tan ^-1 (V ₁ / V ₂ ) (radian)-(2) is obtained by a calculation device (not shown).

In order for the null loop to operate stably, it suffices that θ ₂ + θ _x −θ ₁ = π (radian) − (3).

Therefore, θ ₂ −θ ₁ may be adjusted so that δ = π / 4 + π− (4).

As can be seen from equation (2), the arc tangent function indicates only the range of the principal value, and is adjusted so that δ = π / 4− (5), as compared with equation (4).

Thus, since generally in the low frequency theta _x is almost [pi theta _x with increasing frequency increases than [pi, theta _x
If there is a possibility that the value of _X will greatly change, δ may be measured at several frequencies from the low frequency, and the true value of θ _x may be approximately obtained.

Alternatively, automatic control may be performed so that θ ₂ −θ ₁ is changed so as to converge to a value of δ = π / 4. In this case, if there is still a large phase shift, θ ₂ −
θ ₁ cannot be uniquely determined. In this case, either measured V ₁ and V ₂ separately, provisionally determined values of theta ₂ - [theta] _1, the switch
Close S ₁ , S ₂ , S ₃ , S ₄ to the respective terminal N side, check the presence or absence of oscillation of the null loop, if there is no oscillation, use the correct answer and use it. the _{2 -θ 1} to add a procedure to reduce only π.

When the above adjustments are completed and the switches S ₁ , S ₂ , S ₃ , S ₄ are closed to the terminals N, respectively, the operation is the same as that of the prior art impedance meter shown in FIG.

Note that the resistances R ₃ and R ₄ limit the amplification degree of the integrator and δ
This is to enable the measurement of

FIG. 2 shows an example of a circuit for providing θ ₂ −θ ₁ . Second
In the figure, the required θ ₂ −θ ₁ is approximated to π / 8 radians (22.5
°) For setting in steps.

In this circuit, when W = 2πfn, a digital signal fm input of frequency fm and a clock signal 16fm clock of frequency 16fm are introduced from input terminals I ₂ and I ₃ , respectively.

Signal input to the I _2, together with it the output of the demodulation signal source DM _1, the input of the 90 ° phase shift circuit IC _2. 90 °
Phase shift circuit IC ₂ includes four beats shifting fm input by 16fm clock provides the output of the demodulated signal source DM _2.

Hexadecimal preset counter IC ₁ loads data from terminal I _{4 at} the rising edge of fm input, counts with 16 fm clock, inputs its 50% duty ratio output to 90 ° phase shifter IC ₃ and modulates give the output of the signal source MO _1. On the other hand, the 90 ° phase shifter IC ₃ shifts the input by 4 beats by the 16 fm clock and gives the output of the modulation signal source MD ₂ .

The input data of the terminal I ₄ is given by 4 bits, for example,
When the value is 4 _16, a _{θ 2 -θ 1 = π / 2} (= 4 × π / 8).

<Effects of the Invention> As described in detail above, the following effects can be obtained by implementing the present invention.

1) In a recent impedance meter that performs automatic ranging, a null loop oscillates due to a difference in a measurement condition or a DUT, and the bridge cannot be balanced.

2) As a result, there is no need to perform calibration at the factory before shipment, determine the amount of phase shift according to each frequency and the side constant range, and write it to the storage device as in the related art.

3) Further, the amount of phase shift can be measured, and the abnormality of the null loop can be automatically detected.

4) If necessary, a special effect can be obtained by setting the phase difference between the modulated signal and the demodulated signal so that the loop phase difference of the null loop is not π.

[Brief description of the drawings]

FIG. 1 is a schematic circuit diagram of a narrow band amplifier which is a part of a null loop for performing a phase shift of a signal used in an embodiment of the present invention. FIG. 2 is a schematic circuit diagram of one embodiment of a circuit for controlling a phase difference between the modulation signal and the demodulation signal of FIG. FIG. 3 is a schematic circuit diagram showing the function of a conventional impedance meter. FIG. 4 is a prior art narrow band amplifier corresponding to the narrow band amplifier of FIG. Z: DUT (device under test) OSC: sinusoidal signal source _{L 1, L 2, L 3} , L 4: cable _{B 1, B 2, B 3} : Buffer S _{0,
S 1, S 2, S 3} , S 4 :( single pole double throw) switches _{A 1, A 2, A 2} : amplifier NBA = narrowband amplifier NL = null loop PD _1, PD ₂ = phase detector VG ₁ , VG ₂ = modulator DM ₁ = 0 ° demodulated signal source DM ₂ = 90 ° demodulated signal source MO ₁ = 0 ° modulated signal source MO ₂ = 90 ° modulated signal source

──────────────────────────────────────────────────続 き Continued on the front page (58) Field surveyed (Int.Cl. ⁶ , DB name) G01R 17/00-17/22 G01R 27/00-27/32

Claims (5)

(57) [Claims] 1. A measurement signal source connected to the first terminal of a device under test (hereinafter referred to as a DUT) having first and second terminals, and a measurement signal source connected to the second terminal of the DUT. A feedback amplifier that converts the current flowing through the DUT into a voltage while virtually grounding the second terminal, and that opens a feedback loop of the feedback amplifier and detects a single-phase rotation of the feedback loop. Control means for controlling the single-phase rotation to a predetermined value; and an adaptive half bridge comprising: 2. The adaptive half bridge according to claim 1, wherein the output of the measurement signal source is set to zero when the round phase rotation is detected. 3. The feedback amplifier according to claim 1, wherein the feedback amplifier is a modulation amplifier.
2. The adaptive half bridge according to claim 1, wherein a phase difference between a modulation signal and a demodulation signal of the modulation type amplifier is controlled by the control means so that the loop phase rotation becomes the predetermined value. . 4. The adaptive half bridge according to claim 3, wherein the detection of the loop phase rotation is performed by opening a gap between a modulator and a demodulator of the modulation type amplifier. 5. A first device to be measured (hereinafter referred to as a DUT).
A measurement signal source connected to a terminal of the DUT, a feedback amplifier connected to a second terminal of the DUT, virtually grounding the second terminal, and converting a current flowing through the DUT into a voltage; And a vector voltmeter for measuring the impedance of the DUT from the first and second inter-terminal voltages and the converted voltage, wherein the vector voltmeter detects a loop phase rotation of the feedback amplifier. An impedance meter, characterized in that: