JP2955974B2 - AC motor rotation speed detector - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an apparatus for detecting the rotational speed of an AC motor driven by a PWM inverter without using a speed sensor.

[0002]

2. Description of the Related Art As a method for estimating a speed of a speed sensorless drive system, a method using the property of vector control,
A model reference adaptation system is known. When the vector control is achieved in the former method, since the torque and the torque current (the torque component on the primary side of the motor) are proportional, the slip angular velocity is obtained using the torque current and the secondary magnetic flux, and the rotation of the induction motor is determined. Speed can be estimated. In the latter model reference adaptive system, the model using the characteristic equation of the induction motor, given the same input and the actual system, assuming error in their output is due to errors in the speed omega _m, rotating The speed is estimated from the output error.

[0003]

A secondary resistance that cannot be directly measured is used in the slip angular velocity calculation formula and the model characteristic equation of these methods, and the measured value of the secondary resistance and the actual system are used. If the values do not match, an error in speed estimation is caused. In addition, since the secondary resistance value changes due to a temperature rise and a skin effect at the time of transition, the speed estimation characteristic deteriorates even during operation in the conventional method.

An object of the present invention is to provide an apparatus which can relatively easily detect the rotational speed of an AC motor every moment.

[0005]

According to the present invention, there is provided an apparatus for detecting a rotational speed of a three-phase AC motor driven by a three-phase PWM inverter. Instantaneous values of the three-phase output voltage (v _1a , v _1b ,
v _1c ), current detecting means for detecting three-phase output current instantaneous values (i _1a , i _1b , i _1c ) of the three-phase PWM inverter, and three-phase output voltage instantaneous values (v _1a , v _1b , v _1c ) are _{converted to} instantaneous two-phase voltage values (v _1d , v _1q )
And the three-phase output current instantaneous values (i _1a , i _1b ,
i _1c ) is converted to a two-phase current instantaneous value (i _1d , i _1q ), and the two-phase output voltage instantaneous value (v _1d , v _1q ) and the two-phase output current instantaneous value (i _1d , i _1q ) are _calculated. The primary magnetic flux (φ _1d , φ _1q ) and the secondary magnetic flux (φ _2d , φ _2q ) of the AC motor are obtained based on the above equation, and the instantaneous two-phase output current values (i _1d , i _1q ) and the temporary magnetic flux are obtained. (Φ _1d , φ _1q ), the secondary magnetic flux (φ _2d , φ _2q ), and the primary, secondary and mutual inductances (L
Those related to the rotational speed detecting device of the motor - AC motor, characterized in that it consists of a rotational speed calculating means for calculating a value corresponding to the rotational speed of the motor - _11, wherein the AC motor based on the L _22, M) .

[0006]

In the present invention, the rotational speed of the motor is estimated without using a secondary resistance. Therefore, the rotation speed can be detected regardless of a change in the secondary resistance value due to a temperature change or the like. Since the rotation speed can be known without using a speed sensor, the cost and size of the device can be reduced.

[0007]

Next, a description will be given of a rotation speed control device for a three-phase AC motor (induction motor) using an impulsive torque drive inverter device according to an embodiment of the present invention. In FIG. 1 showing this speed control device, a three-phase inverter 2 capable of PWM control is connected to a motor 1 composed of a three-phase induction motor. The inverter 2 includes a switch element Q1, Q2, Q3, Q4,
Q5 and Q6 are bridge-connected. The six switch elements Q1 to Q6 perform on / off operations in response to a control signal supplied from the drive circuit 4. The upper three switch elements Q1, Q2, Q3 and the lower three switch elements Q4, Q5, Q6 of the inverter 2 operate in opposite directions, so that if one control is specified, the overall inverter Controls are specified. Here, the inverter control state is specified by the first, second, and third signals A, B, and C read from a ROM (read only memory) 5, and the signals A, B, and C are at a high level, that is, logic "1". ", The switching elements Q1, Q2, Q3 are turned on, and when the logic level is" 0 ", the switching elements Q1, Q2, Q3 are turned off.

[0008]

[Description of ROM address] ROM 5 uses inverter 2 for PW
A PWM switching pattern (unit vector data) for M control is written in advance. This RO
M5 is a normal rotation PWM pattern memory M1, M5, a normal rotation zero vector memory M2, a reverse rotation PWM pattern memory M3, M7, a reverse rotation zero vector memory M4, a normal rotation normal vector M6, and a reverse normal vector M8. Having. Each of the memories M1 to M8 has, for example, one of 0 to 1023.
024 addresses, each addressed by the value of the 10-bit binary output line 6a of the up / down counter 6. However, one of the eight memories M1 to M8 is selected, and only the output of the selected memory is effectively used for controlling the inverter 2. To make this selection, the ROM 5 has a zero vector selection control terminal 7, a forward / reverse rotation selection control signal input terminal 8, and a normal vector selection control terminal 42. First, when the normal vector selection control signal terminal 42 is at logic "0", M1 to M4 including the zero vector memory are selected. When the logic is "1", M5 to M8 including the normal vector memory are selected. When the zero vector selection control signal input terminal 7 is at logic "0", one of the memories M1 and M3 or M5 and M7 is selected, and when it is at logic "1", the memory M2 and M4, or M6 and M8 is selected. Either one is selected. Furthermore,
When the forward / reverse selection control signal input terminal 8 is "0", one of the memories M1 and M2 or M5 and M6 is selected, and when it is "1", the memories M3 and M4 or M
Either 7 or M8 is selected. Now line 6
If the 10 bits of a are represented by 10 bits A0 to A9, the input bit of the input terminal 7 is represented by A10, the input bit of the input terminal 8 is represented by A11, and the input bit of the input terminal 42 is represented by A12, An address is specified by 10 bits A0 to A9. Also, A10, A11, and A12 are replaced with [A12,
A11, A10], the first memory M1 (normal PWM switching pattern) is selected at [000], and the second memory M2 (zero vector for normal rotation) is selected at [001]. In step [111], M8 (normal vector for reverse rotation) is selected.

In order to control the rotation speed of the motor 1, information on the rotation speed is required. Conventionally, the rotation speed is detected by a speed detector, but in this embodiment, a speed estimating means is provided without providing a speed detector in order to achieve a reduction in cost of the apparatus. That is, a rotation speed calculation and magnetic flux control circuit 50 for estimating the rotation speed of the motor 1 based on the output current and output voltage of the inverter 2 and performing magnetic flux control is provided. Details of the rotation speed estimation method will be described later.

[0010] Motor rotation speed omega _m obtained by the rotational speed calculating sought-flux control circuit 50 is input to the subtracter 10 as a difference signal forming means from the line 9a (comparing means), the line 11 as the speed command generating means Is compared with a digital speed command (desired rotation speed), that is, a reference signal, and a difference signal between the two is obtained from the subtractor 10.

[0011] The output of the subtracter 10 is input to a proportional integral compensation circuit 12 represented by _{K (1 + 1 / T is} ), the compensation output or difference signal V _D is obtained in the output line 13. This compensation output indicates the amount of operation in the control, and reading of the vector data from the memory 5 is determined based on this.

[0012] The signal compensation difference signal V _{D of the} compensation output line 13
Is configured to input to the first comparator 14 for determining the sign of the difference signal V _D, the through absolute value circuit 15 2
Is input to the comparator 16. An output terminal of the first comparator 14 for determining forward / reverse rotation (F / B) is connected to an up / down input terminal U / D of the counter 6 and a forward / reverse selection signal input terminal of the ROM 5. 8 is connected.

Reference numeral 17 denotes an oscillator (OSC).
A clock pulse of about 50 kHz is generated. The output terminal of the oscillator 17 is connected to one input terminal of the AND gate 18, and the output terminal of the AND gate 18 is connected to the clock input terminal CL of the counter 6.
The output of the oscillator 17 is input to the counter 6 as a clock pulse only when the other input terminal of the D gate 18 is at a high level.

A triangular wave generator 19 is connected to a non-inverting input terminal of the second comparator 16 for judging drive / stop. Triangular wave generator 19 is, for example, lower than the output frequency of the oscillator 17 2.5 kHz and generates a triangular wave voltage Vc (carrier), and the absolute value of the Vc and the difference signal V _D is compared in a comparator 16. The output terminal of the second comparator 16 is N
It is connected to the input terminal of the AND gate 18 via the OT circuit 20 and to the zero vector selection control signal input terminal 7 of the ROM 5.

The rotation speed calculation and magnetic flux control circuit 50 calculates the slip angular speed ω _s based on the output voltage and current of the inverter 2.
Is calculated and the magnetic flux is detected. Therefore, the three output lines 31a for detecting the instantaneous value of the output voltage of the inverter 2 and the three output lines 34a of the current sensors 43a, 43b, 43c for detecting the instantaneous value of the input current of the motor 1 have the rotational speeds of is connected to the arithmetic and the magnetic flux control circuit 50, the output line 9a of the rotation speed omega _m
The magnetic flux signal output line 55 is connected to the terminal 42 of the ROM 5. Although a circuit for controlling reading of vector data (switching pattern) from the memory 5 is shown in FIG. 1 in an analog manner, actually, a subtractor 10, a proportional-integral compensation circuit 12, a comparator 14, 16, the absolute value circuit 15, the triangular wave generation circuit 19, the rotation speed calculation and magnetic flux control circuit 50
P (digital signal processing device). An A / D converter is connected to the output voltage detection line 31a and the current detection line 34a of the inverter 2, but is omitted in FIG. 1 for simplification of the drawing.

FIG. 2 shows the rotation speed calculation and magnetic flux control circuit 50 of FIG. 1 in detail. Before describing the components of the circuit 50, the principle of the method of detecting the primary flux linkage will be described.

The characteristic equation of the induction motor can be expressed by the following equation (1).

[0018]

(Equation 1)

Here, v _1d and v _1q are two-phase primary voltages indicating the primary voltages v _1a , v _1b and v _1c of the motor 1 on the dq coordinate axis, and i _1d and i _1q are primary currents i of the motor 1. _1a , _i1b , _i1c
The primary current of two phases indicated by d-q coordinate axes, R1, R2 are primary and secondary winding resistance, L _11, L ₁₂ are primary and secondary windings inductance, omega _m is the rotor rotational angular velocity (rotational speed) ,
M is the mutual inductance between the primary and secondary windings, p is d / dt
Is a differential operator.

In the equation (1), undetectable values are the secondary currents i _2d and i _2q . The secondary currents i _2d and i _2q are obtained by the first and second rows of the equation (1), and when these are substituted into the third and fourth rows of the equation (1), the rotational angular velocity ω _m
The following two equations (2) and (3) are obtained.

[0021]

(Equation 2)

By the way, since both the equations (2) and (3) include the undetectable secondary resistor R2, the equation (2)
And Equation (3) are combined to eliminate the secondary resistor R2 to obtain the following Equation (4).

[0023]

(Equation 3)

Here, φ _1d and φ _1q are two-phase primary interlinkage magnetic flux indicating the three-phase primary interlinkage magnetic flux in the motor 1 on the dq coordinate axis, and φ _2d and φ _2q are three-phase primary interlinkage magnetic flux in the motor 2. It is a two-phase secondary flux linkage indicating the secondary flux linkage on the dq coordinate axis.

In the three phases, the primary linkage flux vector φ ₁
The relationship between the primary voltage vector v ₁ , the primary current vector i _1, and the primary resistance R ₁ is given by the following equation (5). In addition, the secondary interlinkage magnetic flux vector φ ₂ and the primary interlinkage magnetic flux vector φ ₁
And the primary and next inductances L ₁ and L ₂ and the mutual inductance M are expressed by the following equation (6).

[0026]

(Equation 4)

Φ _1d and φ _1q are the same as φ ₁ in the following equation (7).
And φ _2d and φ _2q can be expressed by the following equation (8) similarly to φ ₂ .

[0028]

(Equation 5)

In formulas (7) and (8), v _1d ,
v _1q , R ₁ , i _1d , i _1q are values that can be determined based on detection or measurement or calculation. Further, each parameter of Expressions (9) and (10) is a value that can be determined based on detection, measurement, or calculation. Therefore, the rotational angular velocity ω _{m in} equation (4) can be determined by calculation based on the detected value.

It should be noted that the two-phase output voltages v _1d , v _1q and the current i _{1d represented} by the dq coordinate axes included in equation (1)
i _1q is the instantaneous value of the three-phase output voltage of the inverter 2 (primary voltage)
v _1a , v _1b , v _1c and instantaneous value of three-phase output current (primary current)
Based on i _1a , i _1b and i _1c , the following equations (11) and (1)
It is determined in 2).

[0031]

(Equation 6)

Next, the rotation speed calculation and magnetic flux control circuit 50 of FIG. 1 will be described in detail with reference to FIGS. 2, 3 and 4.
FIG. 2 is a functional block diagram of the rotation speed calculation / magnetic flux control circuit 50, that is, an equivalent circuit. In FIG. 2, an output line 31a of the inverter 2 is connected to a three-phase to two-phase conversion circuit 31 composed of an operational amplifier. In the three-phase to two-phase conversion circuit 31, the output voltage v _{1a of} the inverter 2 according to the equation (11),
v _1b and v _1c are converted into two-phase output voltages v _1d and v _1q indicated by dq coordinate axes. The current detection line 34a of the inverter 2 is connected to a three-phase to two-phase conversion circuit 34 composed of an operational amplifier. This conversion circuit 34 converts the inverter output currents i _1a , i _1b , i _1c into two-phase currents i _1d , i _1q indicated by dq coordinate axes according to the equation (12).

The φ _1d and φ _1q operation circuits 60 are connected to voltage and current three-phase to two-phase conversion circuits 31 and 34, respectively.
Based on v _1d , v _1q , i _1d , i _1q , the primary magnetic fluxes φ _1d , φ _1q expressed in rectangular coordinates by executing the operations of equations (7) and (8)
Is output. This arithmetic circuit 60 has a function i as shown in FIG.
Two multipliers 61 and 6 for multiplying _1d and i _1q by R _1.
2, two subtractors 63 and 64 for subtracting v _1d , v _1q and i _1d R ₁ and i _1q R _1, and two integrators 65 and 66 for integrating their outputs. be able to.

As shown in FIG. 2, the φ _2d and φ _2q arithmetic circuit 70
Is a φ _1d and φ _1q arithmetic circuit 60 and a three-phase to two-phase conversion circuit 3 for current.
4 for calculating the secondary magnetic fluxes φ _2d and φ _2q of the motor 1. The details of the φ _2d and φ _2q arithmetic circuits 70 are shown in FIG. 3 as two L ₂₂ / M multipliers 71 and 72 and two (L ₁₁ L ₂₂ −M ² ) / M multipliers 73 and 74. And two subtracters 75 and 76, and execute the operations of equations (9) and (10).

The rotational angular velocity calculating circuit 80 in FIG. 2, phi _1d,
and phi _1q arithmetic circuit 60, φ _2d, φ _2q is connected to an operation circuit 70 and the current three-phase two-phase conversion circuit 34, Equation (4) to calculate the L ₁₁ and L ₂₂ as shown in FIG. 4 Multipliers (coefficient units) 91, 92, two subtractors 93, 94, two differentiators 95, 96, and four multipliers 97, 98, 99,
100, a subtractor 101 and an adder 10 of these output stages.
2 and one divider 103. Considering that the motor is driven by a sinusoidal voltage, the numerator and denominator of equation (4) are both zero in a steady state, making it impossible to use this equation. However, here, since the motor 1 is driven by the PWM inverter, the motor 1 is driven by the waveform in which the high frequency ripple is superimposed, and the steady state disappears due to the effect of the superimposed high frequency ripple, and the numerator and the denominator do not become zero. Instead, the rotation angular velocity ω _m can be obtained by Expression (4).

Next, the fact that the numerator and denominator of equation (4) become zero by the sine wave driving and that they do not become zero by the PWM control will be described below. Equation (4) can be collectively expressed by the following equation (13) in three phases. _{ω m = | (φ 1 -L} 1 i 1) × (d / dt) φ 2 | / (φ 1 -L 1 i 1) · φ 2 = | Mi 2 × (d / dt) φ 2 | / Mi ₂ · φ ₂ (13) where i2 is a secondary current vector. When the motor 1 is driven by a sine wave and the slip S is constant, d / dtφ
₂ and i ₂ are in phase, and φ ₂ and i ₂ are orthogonal. Therefore, in the numerator of the formula (13), since d / dtφ ₂ and i ₂ are in phase, the magnitude of the outer product is | d / dtφ ₂ || i ₂ | sin 0 = 0. Since the denominator of equation (13) is orthogonal to φ ₂ and i ₂ , the value of the inner product is | φ ₂ || i ₂ | cos π / 2 = 0. On the other hand, when the high frequency ripple is superimposed on the drive voltage of the motor 1 based on the PWM control, the rotation of the secondary magnetic flux repeats the forward rotation and the reverse rotation instantaneously, and the high frequency components of φ ₂ and i ₂ are not orthogonal. .

[0037] Figure 10 shows the calculated value waveform of the rotational angular velocity omega _m according to formula (1). As shown in FIG. 10A, the speed calculation value can be obtained stably. On the other hand, the values of the numerator and denominator in the equation (1) fluctuate as shown in FIGS. As shown in FIG. 10C, a period occurs in which the denominator becomes zero instantaneously. Accordingly, at or near the point where the denominator becomes zero, an error occurs in the speed calculation. In order to eliminate the influence of this error, no calculation is performed near the denominator being zero, and L
Speed control equivalent to that smoothed by the PF is executed.

FIG. 11 shows the actual speed, the calculated speed, the load torque, and the calculated value of the secondary resistance R2 of the motor 1 in the apparatus of FIG. In FIG. 11, the motor 1 is moved 180 degrees in the first direction.
After rotating at 0 rpm, the rotor is rotated at 1800 rpm in the second direction.
Although it is rotating at 00 rpm, the calculation speed also corresponds well to this. In addition, even during a period in which the load is in the rated load state and the load torque is generated, the calculation speed substantially matches the actual speed. Further, the fluctuation of the secondary resistance R2 does not affect the calculation speed.

The φ ₁ arithmetic circuit 39 in FIG. 2 is connected to the φ _1d and φ _1q arithmetic circuits 60, and φ ₁ = (φ _1d ² + φ _1q ² ) ^1/2
And outputs the absolute value of the primary magnetic flux φ ₁ . Δ
Hysteresis comparators 40 and 41 having a hysteresis comparison width of | φ ₁ | compare the detected value of primary flux linkage | φ ₁ | with the primary flux linkage command value | φ _1a |. And | φ
₁ | is _{| φ 1a | + Δ | φ} 1 | comparator 41 when further increased beyond selects the block M1 through M4 including zero vectors in ROM5 outputs a logic "0". Also, |
If φ ₁ | further decreases beyond | φ _1a | −Δ | φ ₁ |, a logic “1” is output and blocks M5 to M8 containing the normal vector are selected in the ROM 5. Thereby, the magnitude of the primary magnetic flux φ ₁ is controlled to be constant.

The control of the inverter based on the voltage vector and the zero vector describes an impulse torque drive.

[Contents of ROM] Data is written in the ROM 5 as shown in principle. That is, the ROM has addresses from 0 to 1023, but FIG.
This shows the vector arrangement in the case of an address of 1. Forward rotation PW
For example, data of voltage vectors V6, V2, V6, V2 are sequentially written to addresses 0-3 of the M pattern memories M1, M5, and zero vectors V7, V0, V7 are stored at addresses 0-3 of the zero vector memory M2 for normal rotation. , V0 are sequentially written, and the reverse PWM pattern memories M3 and M7 are written.
Addresses 0 to 3 have voltage vectors V1, V5, V1
, V5 are sequentially written, and the data of the zero vectors V0, V7, V0, V7 are sequentially written in the zero vector memory M4 for reverse rotation, and the normal vector memory M for normal rotation is written.
6 address 0 to 3 is a forward rotation PWM pattern memory M
1, a normal vector V4 corresponding to the vector of addresses 0 to 3 is written, and the normal vector corresponding to the vector of addresses 0 to 3 of the PWM pattern memory M3 for reverse rotation is stored in the addresses 0 to 3 of the normal vector memory M8 for reverse rotation. Vector V4 has been written. The remaining addresses 4 to 511 are also written with vector data according to the same principle as the addresses 0 to 3. The vector data of each address in FIG. 5 is different from actual data because it indicates the principle. Now, addresses 0-84 (0 degrees-
The actual voltage vector data (corresponding to a 60-degree section) is as follows: V6, V6, V6, V6, V2, V2, V2, V2
, V2, V2, V6, V6, V6, V6, V2, V2
, V2, V2, V2, V2, V6, V6, V6, V6
, V2, V2, V2, V2, V2, V2, V2, V2
, V2, V2, V2, V2, V2, V2, V2, V2
, V2, V2, V2, V2, V2, V2, V2, V2
, V2, V2, V2, V2, V2, V2, V2, V2
, V2, V2, V2, V2, V2, V3, V3, V3
, V3, V2, V2, V2, V2, V2, V2, V3
, V3, V3, V3, V2, V2, V2, V2, V2
, V2, V3, V3, V3, V3.

FIG. 6 shows six voltage vectors V1 to V6,
Two zero vectors V0 and V7 are shown. The possible switching states of the switching elements Q1, Q2, Q3 of the inverter 2 are (000), (001), (010), (01)
1), (100), (101), (110), (11)
1), which are V0, V1, V2, V
3, V4, V5, V6, and V7. In the apparatus of this embodiment, the voltage vectors V0 to V7 are written in the ROM 5, and are output as control data (A, B, C). Combining the eight vectors V0-V7 gives
A sine wave output voltage and a rotating magnetic field vector can be obtained.

[0042]

[Vector selection] FIG. 7 shows the selection voltage vectors to obtain a rotating magnetic field vector phi _1. In order to make the locus of the tip (end point) of the rotating magnetic field vector φ ₁ close to a circle, the sixth and second vectors V
6, V2, the second and third vectors V2, V3 in the interval of 30 to 90 degrees, the third and first vectors V3, V1, in the interval of 90 to 150 degrees, and the first and second vectors V3, V1, in the interval of 150 to 210 degrees. Fifth and fourth vectors V5, V4, 270 to 33 in the section from 210 to 270 degrees in the fifth vector V1, V5.
The fourth and sixth vectors V4 and V6 are selected in the 0 degree section. 7, V6 and V2 are selected as significant vectors in the section of 330 to 30 degrees shown in FIG. 7, and the zero vector V7 is selected when the vector rotation is stopped. Also,
The normal vector is a vector that extends in the radial direction from the center of the circular locus of the magnetic flux, and V4 is selected in the 330 to 30 degree section and V6 is selected in the 30 to 90 degree section in FIG. When the motor 1 is rotated forward, the rotating magnetic field vector .phi.1 is rotated in the direction indicated by UP in FIG. 7, and when the motor 1 is rotated or braked, it is rotated in the direction indicated by DOWN.

[0043]

Next, the braking operation of the circuit shown in FIG. 1 will be described with reference to FIGS. When the difference signal VD is obtained based on a comparison between the estimated speed signal obtained on the line 9a and the reference signal (target signal) on the line 11, the first comparator 14 determines whether the signal is positive or negative. If it is a signal, FIG.
The comparison output becomes low level "0" as shown before t4 of (C), and this is input to the counter 6. Therefore, the counter 6 operates up during this period. In the second comparator 16, it is compared as shown in (A) of the absolute value and the triangular wave voltage VC Togazu 8 of the difference signal V _D, the output of FIG. 8 (B) is generated. In other words, generates a high level output "1" when the triangular wave voltage VC is higher than the absolute value of the difference signal V _D (t1 ~t2), a low-level output "0" when the opposite (t2 -t3) Occur. As shown from t1 to t2, the output bit A10 of the second comparator 16 is at high level "1", the output bit A11 of the first comparator 14 is at low level "0". As shown in D), the output of the hysteresis comparators 40 and 41 in FIG. 3 is at the low level (L) t10.
Previously, in ROM 5, [A12 A11 A1
0] = [001], the normal rotation zero vector memory M2 is selected, and [A12 A11 A] as in t2 to t3.
When [10] = [000], the normal rotation PWM pattern M1 is selected. During the period (t1 to t2) when the output of the second comparator 16 is high (H), the output of the NOT circuit 20 becomes low, and the clock pulse of the oscillator 17 may pass through the AND gate 18. Blocked, counter 6
Is not incremented, so the same address is kept specified. On the other hand, during the period when the output of the second comparator 16 is low (t2 to t3), the output of the NOT circuit 20 is high, so that the output clock pulse of the oscillator 17 is AND
The signal passes through the gate 18 and becomes an input pulse of the counter 6.
As a result, the values of the 10 bits A0 to A9 of the counter 6 are increased by the up operation, and the addresses of the memory M1 are sequentially specified. However, when the output of the second comparator 16 becomes high at time t3, the clock input to the counter 6 is inhibited, and the counter 6 retains the address designation at this time. For example, when the memory M2 is selected while the vector V6 of the memory M1 is being read at the address 2 as shown in FIG. 5, the normal rotation zero vector V7 (111) at the same address 2 is selected. The zero vector V7 is the second
When the output of the comparator 16 continues to be high while the comparison output returns to low and the clock pulse of the counter 6 is input again and the output of the counter 6 is incremented by one stage, the normal rotation PWM pattern memory The voltage vector V2 (010) at address 3 of M1 is selected. The zero vector consists of two types, V0 (000) and V7 (111), and the vector that requires less switching of the switch elements Q1 to Q6 is selected. If the counter 6 is decimal 0
After generating the binary number corresponding to 1023, the normal rotation PWM
All the voltage vector data of 0 to 360 degrees of the pattern is read, a three-phase approximate sine wave voltage is generated from the inverter 2, and a rotating magnetic field vector close to a circular locus is generated in the motor 1.

In such control, when the difference between the target rotation speed and the estimated speed becomes small, the period during which the output of the second comparator 16 is at a high level becomes relatively long, and the period during which the zero vector is selected. Becomes longer.

Further, if the low-speed rotation command state by lowering the level of the reference signal on line 11 as T20～t4, also decreases the level of the absolute value of the difference signal V _D, decreases the output frequency f of the inverter 2 At the same time, the output voltage V decreases, and the motor 1 enters a low-speed driving state.

At t4 in FIG. 8, the mode is switched to the reverse rotation command.
When the difference signal V _D is negative, the output of the first comparator 14 goes high, the reverse rotation control. In the above PWM control, when switching of the voltage vector is performed,
A short circuit may occur between the pair of switch elements Q1, Q4, or Q2, Q5, or Q3, Q6 due to storage or the like, which may be destroyed. To prevent this, a non-control period is provided between the vectors. It is desirable.

8 is a period during which the outputs of the hysteresis comparators 40 and 41 in FIG. 3 are at a high level. That is, if the speed reference signal on line 11 in FIG. 3 rapidly increases at t10, abruptly increases as the (A) of the difference signal V _D is 8, thus abruptly period to output a normal rotation vector It operates to increase the rotation speed of the motor rapidly. As a result, the primary current of the motor increases rapidly to obtain the desired acceleration. However, the voltage drop due to the primary winding resistance of the motor also increases, and the primary linkage flux ||
Conversely, ₁ | decreases as shown after t10 in FIG. 8 (G).
When | φ _1a | −Δ | φ ₁ | or less, the hysteresis comparator operates and outputs a high level. At this time, the ROM
In block 5, since the terminal 42 is at a high level, the blocks M5 to M8 including the normal vector are selected. Therefore, t
The period t is from 10 to t20 and the second comparator is in the high level period t.
From 11 to t12, the normal vector memory M6 for normal rotation is selected instead of the zero vector memory M2 for normal rotation. FIG. 9 shows 30
This state is shown in a section between degrees and 90 degrees.
Similarly, when the second comparator is in the low level period t12 to t13.
Is selected as M5. By outputting the normal vector instead of the zero vector in this manner, the magnitude of the primary linkage flux of the motor is increased, and the primary linkage flux | φ ₁ | of the motor is increased to | φ _1a | −Δ | φ ₁ | or more. It becomes possible to. As a result, the motor obtains a desired acceleration, responds to the increase in the speed reference signal with good response, and can reach a desired rotation speed. Next, for example, when the level of the speed reference signal on line 11 decreases and the primary current decreases, the voltage drop due to the primary winding resistance also decreases, and the primary flux linkage increases as a result. At t20, the primary flux linkage becomes | φ _1a | + Δ | φ ₁ |
Exceeding the output of the hysteresis comparator becomes low level, thus the primary interlinkage magnetic flux from block M1 M4 is selected that contains the zero vector for the terminal 42 goes low in ROM5 is reduced | φ _1a | + Δ | φ ₁ | From the above, the primary interlinkage magnetic flux | φ ₁ |
It will be controlled within the range of φ _1a | ± Δ | φ ₁ |.
When the rotation direction is reversed, the same operation is performed after t4.

[0048]

[Modifications] The present invention is not limited to the above-described embodiment, and for example, the following modifications are possible. (1) The proportional integration compensation circuit 12 may be a proportional circuit or an integration circuit. (2) Instead of using only the voltage vector data as the first vector data, the waveform may be improved by using a combination of the voltage vector data and the zero vector data. That is, the zero vector may be arranged in the voltage vector array of the memories M1 and M3. (3) Each circuit in FIGS. 1 to 4 can be an individual circuit or an analog circuit without using a DSP.

[Brief description of the drawings]

FIG. 1 is a block diagram illustrating a motor speed control circuit according to an embodiment.

FIG. 2 is a block diagram showing a rotation angular velocity calculation and magnetic flux control circuit of FIG. 1;

FIG. 3 is a block diagram showing a primary magnetic flux calculation circuit and a secondary magnetic flux calculation circuit of FIG. 2 in detail;

FIG. 4 is a block diagram showing the ω _m operation circuit of FIG. 2 in detail.

FIG. 5 is a diagram showing in principle a part of the contents of the ROM of FIG. 1;

FIG. 6 is a diagram showing a voltage vector.

FIG. 7 is a diagram showing a rotating magnetic field vector.

FIG. 8 is a diagram showing a state of each unit in FIG. 1;

FIG. 9 is a diagram showing a relationship between a change in magnetic flux and a vector.

FIG. 10 is a waveform chart showing a calculated value of Expression (1) and values of its numerator and denominator.

11 is a waveform chart showing an actual speed, a calculation speed, a load torque, and a secondary resistance of the device of FIG.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 Motor 2 Inverter 5 ROM 50 In rotation angular velocity calculation and magnetic flux control circuit

(Equation 6)

──────────────────────────────────────────────────続 き Continued on front page (58) Field surveyed (Int. Cl. ⁶ , DB name) G01P 3/44 H02P 5/41 302

Claims (1)

(57) [Claims] An apparatus for detecting a rotational speed of a three-phase AC motor driven by a three-phase PWM inverter, wherein the three-phase PWM inverter has three-phase instantaneous output voltage values (v ^1a , v _1b , v 1 _1c ), current detecting means for detecting three-phase output current instantaneous values (i _1a , i _1b , i _1c ) of the three-phase PWM inverter, and three-phase output voltage instantaneous values (v _1a). , V _1b , v _1c ) to two-phase voltage instantaneous values (v _1d , v _1q ), and converts the three-phase output current instantaneous values (i _1a , i _1b , i _1c ) to two-phase current instantaneous values (i _1d). , I _1q ), based on the two-phase output voltage instantaneous values (v _1d , v _1q ) and the two-phase output current instantaneous values (i _1d , i _1q ). φ _1d , φ _1q ) and secondary magnetic flux (φ _2d , φ _2q )
The instantaneous values of the two-phase output current (i _1d , i _1q ), the temporary magnetic flux (φ _1d , φ _1q ), the secondary magnetic flux (φ _2d , φ _2q ), and the primary and secondary values of the AC motor Next and mutual inductance (L ₁₁ ,
L22, M), based on _L22 , M), a rotation speed calculating means for obtaining a value corresponding to the rotation speed of the induction motor.