JP2783844B2 - Inverter device - Google Patents

Description

The present invention relates to an inverter device suitable for operating a plurality of loads in parallel.

[Prior Art] FIG. 3 shows a schematic configuration of a conventional lighting load control device. In this device, a dimmer 40 including a phase control element such as a triac is interposed on a power input side of a plurality of lighting devices 20, and is turned on by operating an operation unit (for example, a variable resistor VR) of the dimmer 40. This is a phase control type dimming device for dimming the device 20. Reference numeral ₁ denotes a preheating wire for sending the power supply voltage to the filament preheating circuit section of the lighting device 20 as it is regardless of the dimming level.

This type of phase control type lighting load control device can be configured at a comparatively low cost, but requires a dimming phase control element, and the phase control controls the current supply section and current pause section within a half cycle of the power supply voltage. In this case, there is a problem that the input current waveform is distorted.
Since the input voltage to 20 will be reduced, the lighting device 20
Therefore, it is difficult to secure a stable voltage, and various controls of the lighting device are difficult. For example, in the apparatus shown in Figure 3, in order to obtain a stable pre voltage, provided with a preheating wires l _1, dimmers
The output side of 40 has three wires. In addition, since the rise of the power supply voltage waveform is sharpened by the phase control, the noise (and noise) level increases.

Therefore, recently, as shown in FIG. 4, a high frequency conversion circuit 1 including a transistor inverter and the like is used as a ballast, and the voltage of the AC power supply AC is applied to each lighting device 20 by the power lines l ₀ and l _1.
, The dimming signals l ₂ and l ₃ are separately wired, and the dimming signal is supplied from the dimming circuit 4 to the control circuit 3 of the high-frequency conversion circuit 1. A four-wire dimming system has been proposed in which dimming is arbitrarily performed according to an operation of an operation unit (for example, a variable resistor VR). This system focuses on the fact that the high-frequency conversion circuit 1 serving as a ballast originally has a controllable switching element such as a transistor, and attempts to use this switching element for dimming control. The inconvenience of the above-mentioned phase control type dimming system has been solved at once.

However, in this system, a previously unexpected problem that the output tends to vary between the lighting devices (between the ballasts) has been newly found. Hereinafter, this point will be described in detail.

FIG. 5 is a circuit diagram of the dimming circuit 4 used in the device shown in FIG. 4, and FIG. 6 is an operation waveform diagram thereof. In the dimming circuit 4, the smoothing voltage capacitor C is charged from the AC power supply AC through the step-down transformer Tf, the diode bridge DB for full-wave rectification, and the current limiting resistor R, and the zener diode for voltage regulation is charged. to regulate the charging voltage by ZD, to obtain a DC power source E _3. The DC power supply E ₃ is divided by the variable resistor VR, applied to the non-inverting input terminal of the converter IC ₆ as a reference voltage Vr. The triangular wave oscillator 9 supplied with the DC voltage E ₃ includes a capacitor, a resistor for charging the capacitor with a current from the DC power supply E ₃ , and a switching circuit for discharging the capacitor when the charged voltage of the capacitor reaches a predetermined voltage. more it becomes, and voltage Vc is applied resulting in the capacitor to the inverting input terminal of the comparator IC _6.
The voltage Vc obtained by the triangular wave oscillator 9 is not a strict triangular wave but a voltage that rises non-linearly with respect to the time axis, as shown in FIG. comparator
Output terminals of the IC ₆ is connected to the base of the transistor Q ₁₁ via the resistor R _13. The emitter of the transistor Q ₁₁ is connected to the negative electrode of the DC power source E _3, a collector is connected to the positive electrode of the DC power source E ₃ via a resistor R _14, the resistor R ₁₅
It is connected to the base of the transistor Q ₁₁ via the.
The collector of the transistor Q ₁₂ is connected to the positive electrode of the DC power source E _3, the emitter is connected to the negative electrode of the DC power source E ₃ via a resistor R _16. Both ends of the dimming signal S of the resistor R ₁₆
n is obtained. In other words, the resistance between the transistor Q ₁₁ R _13, R ₁₄
Constitutes a grounded emitter type inverting amplifier circuit.
Those constituting the impedance conversion circuit of the collector grounded (emitter follower) type by transistors Q ₁₂ and the resistor _15, R _16.

Hereinafter, the operation of the dimming circuit 4 will be described with reference to FIG. FIG. 6 (a) shows the relationship between the reference voltage Vr obtained from the variable resistor VR and the voltage Vc obtained from the triangular wave oscillator 9. The reference voltage Vr can be set to an arbitrary voltage. When the voltage Vc obtained from the triangular wave oscillator 9 is less than the reference voltage Vr, the output terminal of the comparator IC ₆ becomes "High" level, the transistor Q ₁₁ is turned on, and drops its collector potential, the dimming signal Sn becomes “Low” level. On the other hand, when the voltage Vc obtained from the triangular wave oscillator 9 becomes higher than the reference voltage Vr, the comparator
Since the output terminal of the IC ₆ becomes "Low" level, the transistor Q ₁₁ is turned off and its collector potential rises, the dimming signal Sn becomes "High" level. Thus, a dimming signal Sn as shown in FIG. 4 (b) is obtained. Here, the voltage level of the dimming signal Sn is, for example, about 10 V, and the frequency is, for example, about 1 KHz. Since the reference voltage Vr can be set to an arbitrary voltage by operating the variable resistor VR,
The on-duty of the light control signal Sn can be set to any magnitude from the minimum value shown in FIG. 7A to the maximum value shown in FIG. 7B. In the figure, S ₁ is turned on and
Duty _{(t 1 / T) × 100} = 10% of the signal, S ₁₀ is on duty _{(t 2 / T) × 100} = 90% of the signal. That is, S
_1, and S _10, when adjusting the variable resistor VR of example dimming circuit 4, on-duty of the dimming signal Sn respectively 10
%, 90%. FIG. 8 shows the dimming signals S _{1 to} S ₁₀ output from the dimming circuit 4 and their ON / OFF signals.
This shows the relationship with the duty. In other words, the dimming signal
When adjusted in a range of S ₁ to S _10, the on-duty of the dimming signal varies linearly in the range of 10% to 90%.

FIG. 9 exemplifies a configuration of a lighting device 20 including a control circuit 3 for performing dimming control in response to a dimming signal whose on-duty is variable as described above. Hereinafter, the circuit configuration will be described. At both ends of the DC power source E _1,
A series circuit of transistors Q ₂ and Q ₃ as main switching elements is connected in parallel, and diodes D ₁ and D ₂ are connected in anti-parallel to the transistors Q ₂ and Q ₃ respectively. Transistor Q
_A load circuit is connected to both ends of the load circuit 2 via a coupling capacitor Cd for cutting a DC component and a current transformer CT for feeding back a load current. The load circuit is an illumination load 2 composed of a discharge lamp, and a current limiting and resonance inductor.
It is composed of an LC resonance circuit including L ₁ , a capacitor C ₂ for resonance, and a capacitor C ₃ for supplying resonance and preheating current, and the load current is a braking current. This oscillating current is
It flows through the primary winding of the CT. Therefore, a voltage whose polarity changes in accordance with the oscillating current flowing through the load circuit is induced in the secondary winding of the current transformer CT.
It is applied between the base and emitter of the transistor Q ₂ through R _2, thereby switching the transistor Q _2. The base of the transistor Q ₃ are the output signal of the control circuit 3 is supplied. In the control circuit 3 detects the voltage across the transistor Q ₃ by the resistor R _3, R _4, is intended to turn on for a predetermined time and the transistor Q ₃ from the fall voltage across the transistor Q ₃ is.

The high frequency conversion circuit 1, when the DC power source E ₁ is turned on, and a starting circuit for starting the self-excited oscillation operation. The starting circuit capacitor C ₁ is charged through the resistor R ₁ when the power source is turned on, the charging voltage 2 reaches the breakover voltage of diode thyristor Q ₁ 2-terminal thyristor Q ₁ is turned on, the base of the transistor Q ₃ first it is turned on the transistor Q ₃ by flowing a base current through a diode thyristor Q ₁ to one in which to start the oscillation operation.

Hereinafter, the operation of the circuit in FIG. 9 will be described. On power up, the transistor Q ₃ is turned on, the voltage across becomes "Low" level by the activation circuit. This allows
The control circuit 3 is triggered, becomes its output is "High" level, the on state of the transistor Q ₃ is maintained. When the transistor Q ₃ is turned on, the diode D ₀ becomes conductive,
The capacitor C ₁ will not be charged, the starting circuit is stopped. In this case, the secondary winding of the current transformer CT, so are wound polarity such as to apply a reverse bias voltage between the base and emitter of the transistor Q _2, the transistor Q ₂
Maintain the off state. Then, after the lapse of a predetermined time set by the dimmer circuit 4, the output of the control circuit 3 becomes the "Low" level, the transistor Q ₃ are turned off. When the transistor Q ₃ is turned off, since the residual inductance of the inductor L ₁ by the collector current of the transistor Q ₃ is reduced to generate a reverse induction voltage, the oscillating current flowing through the inductor L ₁ is going to flow in the same direction, diode D ₁ is conducting. Further, by the secondary winding of the current transformer CT generates a reverse induced voltage, the transistor Q ₂ is forward biased, the transistor Q ₂ is turned on. Diode D ₁
When current is zero, current flows through the transistor Q ₂ charges accumulated in the capacitor Cd as a power source. At this time, the core of the inductor L ₁ is directly magnetized towards the saturation magnetic flux. Eventually, the core reaches saturation flux, inductance rapidly toward the direction of zero, so that the time variation of the collector current of the transistor Q ₂ is infinite. The collector current of the transistor Q ₂ reaches hfe times the base current, the transistor Q ₂ is made unsaturated state, the base current is fed back from the current transformer CT is reduced transistor
Q ₂ turns off. Even after the transistor Q ₂ is turned off, the oscillating current flowing through the inductor L ₁ because the Ni tends to flow in the same direction, the diode D ₂ is conducting, the load circuit, the capacitor C
d, a current flows through a path of the DC power source E _1. When the diode D ₂ is conducting, since the voltage across the transistor Q ₃ are zero, the control circuit 3 is triggered, the output of the control circuit 3 is "High" level, and the transistor Q ₃ are forward biased. After oscillating current flowing through the diode D ₂ becomes zero, from the DC power source E _1, a capacitor Cd, a load circuit, a current flows through a path of the transistor Q _3. Hereinafter, by repeating the above operation, the oscillation operation of the inverter is continued.

Next, the control circuit 3 includes a monostable multivibrator IC ₁ composed of a general-purpose integrated circuit (for example, μPD4538 manufactured by NEC Corporation). This monostable multivibrator IC ₁ changes the falling trigger input terminal B from “High” level to “Lo”.
w ”level, the output terminal Q is“ Hig
h "level, the output terminal" be in. this embodiment the Low "level, detected by dividing the voltage across the transistor Q ₃ by a series circuit of a resistor R _3, R _4, monostable multivibrator and a trigger signal IC ₁ '. monostable multivibrator output terminal Q IC _1' is "High" at a level time (the output terminal is "Low" will level time), the resistor
It is determined by the time constant of R ₅ and capacitor C _4. It is connected to the base of the transistor Q ₄ for driving the output terminal Q, and an output terminal connected to the base of the transistor Q ₅ for driving. The collector of the transistor Q ₄ are the positive electrode of the DC power source E _2,
The negative electrode of the transistor Q emitters of ₅ DC power source E _2, are connected, the collector of the emitter and the transistor Q ₅ of the transistor Q ₄ are, is connected to the base of the transistor Q _3. Therefore, monostable multivibrator IC
₁ operates as a timer circuit for determining the ON period of the transistor Q _3. Monostable multivibrator IC ₁
The connection point of the resistors R ₅ and capacitor C ₄ for constant setting time of,
The output of the operational amplifier IC ₂ is connected through the diode D ₃ and resistor R _6. Operational amplifier IC ₂ is an impedance converter connected to an inverting input terminal to the output terminal and outputs a voltage of the capacitor C _5, which is applied to the non-inverting input terminal and low impedance. The capacitor C ₅ are connected in parallel a resistor R ₇ for charge discharge is charged by the output voltage of the operational amplifier IC _3. The operational amplifier IC ₃ is an impedance converter having an inverting input terminal connected to an output terminal,
The non-inverting voltage of the capacitor C ₆ which is applied to the input terminal and outputs the low impedance. The capacitor C ₆ is charged by a constant current from the current mirror circuit 8 including the transistors Q ₈ and Q ₉ , and the transistor Q ₆ connected in parallel at both ends is charged.
When is turned on, the electric charge is discharged. Constant current supplied from the current mirror circuit 8 to the capacitor C ₆ is the same as the current flowing from the DC power source E ₂ to the resistor R ₈ through the transistor Q _9. To the base of the transistor Q ₆ includes a DC power supply E ₂
By the voltage resistance R _10, dividing the voltage obtained by R ₉ is given a forward bias. At both ends of the resistor R ₉ and the transistor Q ₇ is connected in parallel, when it is turned on by the output of the transistor Q ₇ is dimming circuit 4, the forward bias of the transistor Q ₆ is lost, the transistor Q ₆ is turned off. At this time, the capacitor C ₆ is charged by a constant current from the current mirror circuit 8, the charge voltage V ₁ was linearly increased. The waveform of the charge voltage V ₁ of the capacitor C ₆ is a constant frequency, a triangular wave is equal to the ON time width in the voltage rising period the dimming signal. Therefore, as the on-time width is long in the dimming signal, the peak value of the charging voltage V ₁ of the capacitor C ₆ is high. Operational amplifiers IC ₂ and IC ₃ and capacitor C
₅ and the resistor R ₇ constitute a peak hold circuit of the charging voltage V ₁ of the capacitor C _6, an output voltage V _2, the capacitor
The peak of the DC voltage of the charging voltage V ₁ of the C _6. For this reason,
The output voltage V _2, as shown in FIG. 10, in proportion to the on-duty of the dimming signal of the dimming circuit 4, a linearly varying voltage. In the figure, when _{V 2 = V 2 a, 90} % when the on-duty of the dimming signal is 10% and has a V ₂ = V ₂ b. Further, the resistor R ₆ is a control resistor, by the output voltage V ₂ to form a parallel current path with the resistor R _5, by increasing the charging current of the capacitor C ₄ in response to an increase in the output voltage V _2, This is to control the time constant of the monostable multivibrator IC ₁ to be small. Therefore, when the on-duty of the dimming signal increases, the ON time width of the transistor Q ₃ becomes shorter, the light output of the lamp load 2 is reduced.

[Problems to be Solved by the Invention] The on-duty of the dimming signal from the dimming circuit 4 and the light output (lamp power) when the light output of the lighting load 2 is controlled using the device shown in FIG. Is shown in FIG. As is clear from the figure, the optical output shows a non-linear change with respect to the change of the on-duty of the dimming signal.
Figure 12 shows the waveform of the collector current Ic of the transistor Q _3, a waveform of the base voltage Vb. Thus, the waveform of the collector current Ic of the transistor Q _3, has become non-linear current waveform with respect to the time axis. This is because the collector current Ic of the transistor Q ₃ is or is part of the resonant current waveform including a load. Therefore, change in the current flowing through the load be linearly changing the conduction period of the transistor Q ₃ are no longer a linear. FIG. 13 shows an example of changes in the collector current Ic in the case of changing the base voltage Vb of the transistor Q ₃ each period of 0.1 T. As apparent from FIG. 13, in the range conduction period of the transistor Q ₃ is T～0.8T, the waveform of the collector current Ic of the transistor Q ₃ are not changed much in the range of 0.6～0.4T, the same despite controlled by 0.1T as the waveform of the collector current Ic of the transistor Q ₃ are largely changed.

In other words, the on-duty change in the dimming signal Sn from the dimming circuit 4, by the ON time width of the transistor Q ₃ is changed, the lighting load 2 light output changes, obtained desired dimming state However, it should be noted here that in the conventional dimming system as shown in FIG. 3, the same voltage is applied to each lighting device 20 by controlling the phase of the AC voltage of the commercial power supply AC. and What contrast, in the FIG. 4 later showing such dimming system, the control of switching the time width of the oscillating current determined by the inductor L ₁ and capacitor C ₂ and the like in a high-frequency conversion circuit 1 of the lighting device 20, the Since the high-frequency conversion circuits 1 are individually performed, the output of each high-frequency conversion circuit 1 is more likely to vary than in a phase control type dimming system.

Such a situation is caused not only when a plurality of lighting devices of the same type are connected in parallel to a power supply and the dimming control is performed continuously or stepwise as described above, but also for different types (for example, for a 40 W fluorescent lamp). If lighting devices for 110 W fluorescent lamps are connected in parallel to the same power supply and batch dimming control is performed by the same dimming signal, a more serious problem may occur. In other words, different types of lighting devices have significantly different dimming characteristic curves as shown in FIG. 11, so that the change in brightness is significantly different from the operation of the dimming circuit 4, and the user feels discomfort. There is a fear of giving.

The present invention has been made in view of such a point,
An object of the present invention is to provide an inverter device capable of eliminating variations in outputs of a plurality of loads driven by a high-frequency conversion circuit operated in parallel.

[Means for Solving the Problems] In the present invention, in order to solve the above-mentioned problems, as shown in FIG. 1, a switching element for oscillation and a power supply are turned on and off by the switching element. An inverter device in which a plurality of high-frequency conversion circuits 10A and 10B each having an LC oscillation circuit that supplies an oscillation current to a load and a control unit that controls the ON time width of a switching element are connected in parallel to the same power supply AC. Output correction means 12 comprising a detection unit 12A for detecting the output of at least one high-frequency conversion circuit 10A, and a correction unit 12B for receiving the detection signal of the detection unit 12A and correcting and controlling the output of another high-frequency conversion circuit 10B. Is provided.

[Operation] In the present invention, since it is configured as described above,
The output difference between the output from the high-frequency conversion circuit 10A to the load 11A and the output from the high-frequency conversion circuit 10B to the load 11B is output correction means.
12, the dispersion of the outputs of the loads 11A and 11B is reduced. Accordingly, as indicated by the dashed line in FIG. 1, the radio frequency converting circuit 10A via a dimming circuit 4 dimming signal line l _2, l _3, even when the supply of the same dimming signal 10B, The output of each of the loads 11A and 11B can be controlled at the same rate of change.

Embodiment FIG. 2 is a circuit diagram of one embodiment of the present invention. In the figure, 10
A is a high frequency conversion circuit for lighting one lamp, and 10B is a high frequency conversion circuit for lighting two lights.

Hereinafter, the circuit configuration of the high-frequency conversion circuit 10A will be described. AC input terminal of the diode bridge DB ₁ is connected to a commercial AC power source AC. The DC output terminal of the diode bridge DB ₁ has a transistor as the main switching element.
A series circuit of Q ₂ and Q ₃ is connected in parallel, and each transistor Q ₂ and Q ₃
Have diodes D ₁ and D ₂ connected in anti-parallel, respectively. At both ends of the transistor Q ₂ is, via a coupling capacitor Cd for cutting a DC component, and a current transformer CT ₁ for feeding back the load current, the load circuit is connected. The load circuit is composed of an LC resonance circuit including a discharge lamp l ₁ , an inductor L ₁ for current limiting and resonance, a capacitor C ₂ for resonance, and a capacitor C ₃ for energizing resonance and preheating current,
Load current becomes oscillating current, the oscillating current flows through the primary winding of the current transformer CT _1. Thus, the secondary winding of the current transformer CT _1, the load voltage that changes polarity in response to the oscillating current flowing in the sea is induced, between the base and emitter of the transistor Q ₂ and the induced voltage through the resistor R ₂ It is applied to, thereby switching the transistor Q _2. The base of the transistor Q ₃ are the output signal of the control circuit is supplied. The control circuit is a monostable multivibrator IC composed of a general-purpose integrated circuit (for example, NEC µPD4538) ₁
It has. This monostable multivibrator IC ₁
The falling trigger input terminal B changes from “High” level to “L”.
ow "level, the output terminal Q goes to" Hig
h "level, the output terminal is" In the. present embodiment becomes Low "level, dividing the voltage across the transistor Q ₃ by a series circuit of a resistor R _3, R _4, than the resistance R ₁₇ and capacitor C ₇ Become CR
The signal is used as a trigger signal for the monostable multivibrator IC ₁ via the circuit. The time when the output terminal Q of the monostable multivibrator IC ₁ becomes “High” level (when the output terminal
ow "time to become level) is determined by the time constant of the resistor R ₅ and capacitor C _4. Note that monostable multivibrator IC
₁ operating supply voltage is obtained by charging the capacitor C ₁ by lowering the output voltage of the diode bridge DB ₁ by the resistance R _1. The output terminal Q of the monostable multivibrator IC ₁ is connected to the base of the driving transistor Q ₄ via the resistor R ₁₉ , and the output terminal is connected to the base of the driving transistor Q ₅ via the resistor ₂₀ . . Transistor
The positive electrode of the collector of Q ₄ are capacitors C ₁ through the resistor R _1R, the negative electrode of the emitter capacitor C ₁ of the transistor Q _5, are connected, the collector of the emitter and the transistor Q ₅ of the transistor Q ₄ are, transistor Q Connected to ₃ bases. Thus, the monostable multivibrator IC ₁ ', operates as a timer circuit for determining the ON period of the transistor Q _3.

On power up, the voltage of the capacitor C ₇ is at low level, the monostable multivibrator IC ₁ is triggered, the transistor Q ₃ is turned on. When the transistor Q ₃ is turned on, the secondary winding of the current transformer CT _1, since wound in polarity so as to apply a reverse bias voltage between the base and emitter of the transistor Q _2, the transistor Q ₂ is turned off To maintain. Then, after a predetermined time, Q output of the monostable multivibrator IC ₁ 'becomes "Low" level, the transistor Q ₃ are turned off. Transistor Q ₃
There is turned off, since the residual inductance of the inductor L ₁ by the collector current of the transistor Q ₃ is reduced to generate a reverse induction voltage, the oscillating current flowing through the inductor L ₁ is going to flow in the same direction, the diode D ₁ Becomes conductive. Further, by the secondary winding of the current transformer CT ₁ generates a reverse induced voltage, the transistor Q ₂ is forward biased, the transistor Q ₂ is turned on. Diode D ₁
When current is zero, a current flows through the transistor Q ₂ charges accumulated in the capacitor Cd as a power source. At this time, the core of the inductor L ₁ is linearly magnetized towards the saturation magnetic flux. Eventually, the core reaches saturation flux, inductance rapidly toward the direction of zero, so that the time variation of the collector current of the transistor Q ₂ is infinite. The collector current of the transistor Q ₂ reaches hfe times the base current, the transistor Q ₂ is made an unsaturated state, the transistor Q ₂ based current decreases fed back from the current transformer CT ₁ is turned off. Even after the transistor Q ₂ is turned off, the oscillating current flowing through the inductor L ₁ is going to flow in the same direction, the diode D ₂ is conducting, the load circuit, the capacitor C
d, a current flows through a path of the diode bridge DB _1. When the diode D ₂ is conducting, since the voltage across the transistor Q ₃ are zero, the monostable multivibrator IC ₁ is triggered, the transistor Q ₃ are forward biased. After oscillating current flowing through the diode D ₂ becomes zero, from the diode bridge DB _1, a capacitor Cd, a load circuit, a current flows through a path of the transistor Q _3. Hereinafter, by repeating the above operation, the oscillation operation of the inverter is continued.

Next, the high frequency conversion circuit 10B for lighting two lamps has basically the same configuration as the high frequency conversion circuit 10A for lighting one lamp, but the two fluorescent lamps l ₂ and l ₂ of the load circuit are used. _It differs in that it includes _three series circuits. Resonance and the current flowing through the capacitor C ₃ 'for preheating current supply flows to the primary winding of the preheating transformer T _3, a fluorescent lamp by the secondary winding output of the preheating transformer T ₃
A preheating current is supplied to the common-side filaments l ₂ and l ₃ .
Further, the time constant circuit of the monostable multivibrator IC ₁ ', that the output voltage V _C of the correction unit 12B is supplied are different. Therefore, the ON time width of the transistor Q ₃ 'in the high-frequency converting circuit 10B for 2-lamp is controlled in accordance with the output voltage V _C of the correction unit 12B.

Next, the configuration of the detection unit 12A will be described. First, the load current path and the preheating current path of the fluorescent lamp l ₁ in the high-frequency conversion circuit 10A, 1 primary winding of the current transformer T ₂ is connected with the polarity as shown. Therefore, the secondary winding of the current transformer T _2, the lamp current Il ₁ minus the preheating current from the load current of the fluorescent lamp l ₁ is detected. In response to the lamp current Il _1, capacitor C ₉ a diode D _4, is charged via the resistor R _21. Thus, both ends of the capacitor C _9, the detected voltage V _A corresponding to the lamp current Il ₁ fluorescent lamp l ₁ is obtained.

Next, the configuration of the correction unit 12B will be described. The primary winding of the current transformer T ₂ ′ is connected to the load current path and the preheating current path of the fluorescent lamps l ₂ and l ₃ in the high-frequency conversion circuit 10B with the illustrated polarity. The output of the secondary winding of the current transformer T ₂ ′ is connected to the capacitor C via the diode D ₅ and the resistor R _22.
8 is charged to both ends of the capacitor C _8, the detection voltage V _B according to the lamp current of the fluorescent lamp l _2, l ₃ is obtained. The detected voltage V _A, V _B are differentially amplified by the resistor R ₂₃ to R ₂₆ and differential amplifying circuit IC ₈ consisting of an operational amplifier, the output voltage V _C to the capacitor C ₁₀ is obtained.

The output correction means 12 is provided by these detection unit 12A and correction unit 12B.
Is configured. Hereinafter, the operation will be described. 1 due to fluctuations in power supply voltage, variations in parts, changes in ambient temperature, etc.
When the output of the fluorescent lamp l ₁ in the high-frequency conversion circuit 10A for lamp lighting and increased, the detected voltage V _A by the current transformer T ₂ for detecting the lamp current of the fluorescent lamp l ₁ increases. 2
Detection voltage V _B by current transformer T ₂ ′ for detecting the lamp current of fluorescent lamps l ₂ and l ₃ in high-frequency conversion circuit 10B for lamp lighting.
Wherein comparing the detected voltage V _A, the two detection voltages V _A, the difference V _B output by the differential amplifier circuit IC ₈ and. Differential amplifier circuit IC ₈
Is V _C = (V _B −V _A ) (R ₂₅ / R ₂₃ ). Therefore, when the detection voltage V _A increases, the output voltage V _C decreases, the output pulse width of the monostable multivibrator IC ₁ 'which is determined by the output voltage V _C increases.
As a result, the conduction period of the transistor Q ₃ ′ increases, and the lamp current of the fluorescent lamps l ₂ and l ₃ increases. Phosphor l ₁
If the lamp current is decreased conversely by the operation reverse to the above, also reduces the lamp current of the fluorescent l _2, l _3. Thus, fluorescent lamps l ₁ which is turned on by a high frequency conversion circuit 10A,
Fluorescent l ₂ which is turned on by a high frequency conversion circuit 10B, l ₃ and variations in the output can be reduced, it is possible from the entire fluorescent lamp l ₁ to l ₃ obtain an optical output comparable.

It should be noted that the high frequency conversion circuit 10A, the output pulse width of the monostable multivibrator IC _1, IC ₁ 'in 10B, by adding a circuit as described in the conventional example in that a time constant circuit (Figure 9), a variable The light output of the fluorescent lamps l _{1 to} l ₃ can be made variable by the dimming signal obtained from the same dimming circuit 4. In order to reduce the flicker of the fluorescent lamps l _{1 to} l ₃ , a smoothing capacitor may be connected in parallel to the DC output terminals of the diode bridges DB ₁ and DB ₁ ′.

In the above embodiment, the lamp current of the fluorescent lamp 11 in the high frequency conversion circuit 10A for lighting _one lamp is detected, and the operation of the high frequency conversion circuit 10B for lighting two lights is controlled so that the light output becomes substantially the same. However, conversely, the operation of the high frequency conversion circuit 10A for lighting one lamp may be controlled by detecting the lamp current of the high frequency conversion circuit 10B for lighting two lights.

In the embodiment, the output of each of the high-frequency conversion circuits 10A and 10B is detected by the lamp current via the current transformers T ₂ and T ₂ ′, but the light output of the fluorescent lamps l ₁ , l ₂ and l ₃ is It may be detected by an optical semiconductor element such as a diode.

The application range of the present invention is not limited to the dimmable discharge lamp lighting device as in the embodiment, and a high frequency conversion circuit for controlling the ON width of the oscillation switching element is provided in parallel. In the field of operation, the present invention can be applied to a case where the output of the high-frequency switching circuit is not variably controlled. Also,
The load of the high-frequency conversion circuit is not limited to the lighting load.For example, even when a plurality of high-frequency conversion circuits for driving a motor are connected in parallel to a power supply, by suppressing the output difference between the high-frequency conversion circuits. Therefore, it is possible to suppress the speed difference between the motors.

In addition, the control of the oscillation switching element in the high-frequency conversion knitting circuit also controls the ON width.
It does not ask about the method and the concrete circuit such as the self-excited type and the separately excited type.

[Effects of the Invention] In the present invention, as described above, when a plurality of high-frequency switching circuits including an LC oscillation circuit and including a control unit for controlling the ON time width of the switching element are operated in parallel, at least one Since a detection unit that detects the output of the high-frequency conversion circuit and a correction unit that receives the detection signal of the detection unit and corrects and controls the output of another high-frequency conversion circuit are provided, Even if there are variations in the component constants and the temperature rise, there is an effect that the output from each high-frequency conversion circuit to the load can be prevented from largely fluctuating.

If the present invention is applied to a system that controls the lighting load by each high-frequency conversion circuit, supplies a common dimming signal to each high-frequency conversion circuit, and controls the dimming of each lighting load, the lighting load can be adjusted. When the light is controlled, it is possible to prevent such a problem that the rate of change of the light output is greatly different and the user feels strange.

[Brief description of the drawings]

FIG. 1 is a block circuit diagram showing a basic configuration of the present invention, and FIG.
FIG. 3 is a circuit diagram of one embodiment of the present invention, FIG. 3 is a block diagram of a conventional example, FIG. 4 is a block circuit diagram of another conventional example, FIG. 6 is an operation waveform diagram of the above, FIGS. 7 and 8 are operation explanatory diagrams of the above, FIG. 9 is a circuit diagram of a lighting device used in the above, FIG. 10 and FIG.
FIGS. 12 and 13 are operation explanatory diagrams of the above-mentioned operation, and FIGS. 12 and 13 are operation waveform diagrams of the same. 10A and 10B are high-frequency exchange circuits, 11A and 11B are loads, 12A is a detection unit, 12B is a correction unit, and 12 is an output correction unit.

──────────────────────────────────────────────────の Continued on the front page (51) Int.Cl. ⁶ Identification code FI H05B 41/392 H05B 41/392 G M (72) Inventor Yuji Nakabayashi 1048 Kazuma, Oji, Kadoma, Osaka Prefecture Matsushita Electric Works, Ltd. (58) Field surveyed (Int.Cl. ⁶ , DB name) H02M 7/42-7/98 H05B 41/24-41/42

Claims (1)

(57) [Claims] An RF switching device comprising: a switching element for oscillation; an LC oscillation circuit for supplying an oscillation current from a power supply to a load by an on / off operation of the switching element; and a control unit for controlling an ON time width of the switching element. In an inverter device in which a plurality of conversion circuits are connected in parallel to the same power supply, a detection unit that detects an output of at least one high-frequency conversion circuit, and receives a detection signal from the detection unit and outputs an output of another high-frequency conversion circuit. An output correction means comprising a correction unit for correcting and controlling the output of the inverter.