JP2612761B2 - Control method of synchronous motor - Google Patents

Description

Description: BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control method of an AC synchronous motor used as a servomotor for controlling a feed axis of a machine tool and each axis of a robot.

2. Description of the Related Art FIG. 3 is a block diagram of a speed control loop including a current control loop as a minor loop of a three-phase synchronous motor conventionally used as a servomotor.

In FIG. 3, reference numeral 10 denotes a difference between the speed command Vc and the actual speed V (Vc−
V), that is, a transfer function that inputs a speed deviation and outputs a torque command Tc, where k1v is a speed loop integral gain and k2v is a speed loop proportional gain. 12R, 12S, and 12T are transfer functions that adjust the phase of each phase command with respect to the current position of the rotor so that the combined magnetic field created by the current of each phase in the torque command Tc is in phase with the magnetic field created by the permanent magnet. Torque command
Tc is multiplied by sin θ (θ: phase of the R phase) to generate an R-phase current command Irc. In addition, sin (θ + 2/3)
π) and sin (θ−2 / 3π) to generate S-phase and T-phase current commands Isc and Itc. 16 is a transfer function as an integrator of the current loop, and k1 is an integral gain of the current loop. R
The difference between the phase current command Irc and the R-phase real current Ir is integrated by the integrator 16, and the value obtained by multiplying the R-phase real current Ir by the current loop comparison gain k2 represented by the transfer function 20 is subtracted from the value. , PWM
A command (voltage command) is generated and output to the R phase of the motor. Note that R in the transfer function 18 is the winding resistance of the motor, L
Is the winding inductance.

Further, in FIG. 3, the current feedback section is described in detail for the R phase, but in FIG. 3, a portion shown by a broken line 14R is a similar block diagram for the other S and T phases. Parts 14S and 14T are omitted. Also, transfer function 2
Reference numerals 2 and 24 are transfer functions of the motor mechanical section, Kt is a torque constant, and J is a motor rotor inertia. Further, Td indicates a disturbance torque.

In the above configuration, in the speed control loop, the speed command Vc
In accordance with the difference (Vc−V) between the actual speed V and the actual speed V, a process represented by the transfer function 10 is generated to generate a torque command Tc, and the torque of the torque command is realized by a current control loop which is a minor loop of the speed control loop. The PWM command to be applied to the R-phase, S-phase, and T-phase windings is calculated.

That is, the sine values sinθ and sin of the phases of each phase are set so that the combined magnetic field generated by the current of each phase according to the rotor position in the torque command Tc has the same phase as the magnetic field generated by the permanent magnet.
(Θ + 2π / 3), sin (θ−2π / 3) and multiply by the current commands Irc, Isc, Itc for each phase. From the current commands Irc, Isc, Itc, the actual currents Ir, Is for each phase are respectively obtained. , It is integrated by the integrator, respectively, the value obtained by subtracting It, subtract the value obtained by multiplying the actual speed of each phase by the current loop gain k2 from each integrated value, and calculate the PWM command of each phase,
A PWM command is given to the inverter to drive the motor.

Problems to be Solved by the Invention In the conventional configuration and operation as described above,
As shown in the figure, when the disturbance torque Td is present, its influence first appears as a speed fluctuation, which is reflected in the torque command through the integrator and the proportional unit (the transfer function 10) of the speed control loop by the speed feedback, and the disturbance torque Td Acts to suppress.

Normally, the frequency band of the velocity loop is low, so that it is possible to suppress disturbance components of sufficiently low frequency,
Frequency disturbances of a certain degree or more cannot be suppressed only by the speed control loop due to delay of the phase of the integrator or the like, and if the phase is shifted by 90 degrees or more, it will be amplified.

On the other hand, although the current control loop has a high frequency band, the current is not directly affected by the disturbance torque Td, and when the torque command is changed by the disturbance via the speed control loop, the influence of the disturbance is first canceled out. However, the current control loop having a high frequency band does not directly contribute to disturbance suppression at all.

As a result, the conventional control method has a disadvantage that disturbance torque cannot be suppressed with respect to a frequency disturbance higher than a certain level.

Accordingly, an object of the present invention is to improve disturbance suppression characteristics by performing disturbance suppression by a current control loop having a high frequency band.

Means for Solving the Problems The present invention provides an actual acceleration obtained by differentiating the actual speed of the motor from the torque command value obtained in the speed control loop and the current command value of each phase obtained from the rotor position. The values obtained by subtracting the values of the respective phases determined according to the rotor position after being converted to the dimensions of the respective phases are input to the integrators of the current control loops of the respective phases, or the values are obtained by the speed control loop. The current command value of each phase is obtained from the torque command value obtained by subtracting the value obtained by converting the actual acceleration obtained by differentiating the actual speed of the motor into the current value, and the current command value of each phase from the rotor position. The above problem has been solved by a configuration in which the current command values are respectively input to the integrators of the current control loops of each phase.

The change in speed, or acceleration, is caused by the total torque, which is the sum of the torque generated by the motor itself and the disturbance torque, and the value of each phase corresponding to the rotor position in the actual acceleration converted to current units. Is used instead of the feedback of the actual current in the conventional current control loop, the influence of the disturbance torque occurs as acceleration, and the acceleration is feedback-controlled in the current control loop to act as disturbance suppression. As a result, the current is controlled by the current control loop having a high frequency band, so that the disturbance suppression acts on the high frequency disturbance.

Embodiment FIG. 1 is a block diagram of speed control and current control of a three-phase synchronous motor according to one embodiment of the present invention, and the same components as those in the conventional example shown in FIG. . And
The difference from the conventional example is that the actual current of each phase
Ir, Is, It is fed back and the current command Irc, Is of each phase
c, Itc and the difference (Irc−
Ir), (Isc-Is), and (Itc-It) are input to the integrators of the current control loop (for example, in the R phase, the integrators shown in FIG. 3 at 16). In the embodiment of the present invention, a current obtained by converting the actual acceleration of the motor into a potential of the current is input instead of the feedback current input to the integrator of the current control loop.

That is, a constant ka for differentiating the motor speed V (transfer function 26), obtaining the acceleration α, and converting the acceleration into a current unit.
(Transfer function ka) to obtain a value ka · α corresponding to the actual acceleration α of the motor, and multiply the value ka · α by the sine value of the phase of each phase according to the rotor position. Isf, Itf,
Although the feedback value of the actual current of each phase is changed and input to the integrator (16) of the current control loop, in the present embodiment, the torque command obtained in the speed control loop is used.
Tc is multiplied by the sine values of phases θ ′, (θ ′ + 2π / 3) and (θ′−2π / 3) slightly advanced from the phases of the respective phases corresponding to the rotor position, and a value ka corresponding to the actual acceleration α is obtained. · Α is the phase of each phase corresponding to the rotor position, θ, (θ + 2π / 3), (θ-
2π / 3) (θ ′ = θ−
Δθ). The other structure is the same as the conventional block diagram shown in FIG. In FIG. 1, 14'S and 14'T are 14'R
2 is a block diagram of the same S phase and T phase, and is omitted in FIG.

A speed deviation obtained by subtracting the actual speed V of the motor from the speed command Vc is proportionally / integrally controlled (transfer function 10) to obtain a torque command Tc. In the current control loop, the torque command Tc is set from the phase of each phase with respect to the rotor position. The sine values of each phase advanced by Δθ sin θ ′, sin (θ ′ + 2π / 3), sin (θ′−2π /
3) (Transfer function 12R, 12S, 12T)
Irc, Isc, Itc are obtained. On the other hand, the actual speed V is differentiated and multiplied by a constant ka as described above, and the sine values sinθ, sin (θ + 2π / 3), sin ( θ−
Irf, Isf, Itf multiplied by 2π / 3) are subtracted from the current commands Irc, Isc, Itc of the respective phases, and input to the integrator (16) of the current control loop. Then, the actual current Ir (Is, It) of each phase is subtracted from the output of the integrator (16), and a PWM command PWMr (PWMs, PWMt) to be given to each R-phase, S-phase, and T-phase winding is created. Current will flow through the wire.

In the above operation, if the disturbance torque Td exists, first, the actual speed V of the motor changes due to the disturbance torque Td. That is, acceleration occurs. Therefore, in the current control loop, the actual speed V is differentiated to obtain the acceleration α, multiplied by a constant ka,
Furthermore, the values Irf, Isf, Itf corresponding to the acceleration α are multiplied by a sine value corresponding to the phase of each phase and fed back to the current commands Irc, Isc, Itc of each phase. The feedback control for the disturbance torque Td is performed in the above, and the disturbance torque Td is suppressed.
Disturbance suppression can be performed even for a certain level of frequency disturbance that cannot be suppressed by the speed control loop.

FIG. 2 is a flowchart of a current control loop process in servo control (so-called digital servo control or software servo control) by a processor such as a digital signal processor according to one embodiment of the present invention. Note that the hardware configuration of a processor or the like that performs the servo control is already known, and thus the description is omitted.

In the processor that performs servo control, a position deviation is obtained from a movement command output from a numerical control device or the like and a feedback pulse from a pulse coder as a position detector attached to the motor, and a speed command Vc is calculated from the position deviation. Next, as a speed control loop process, a speed deviation is obtained from the speed command Vc and the actual speed V of the motor obtained from a pulse coder or the like, and a torque command Tc is calculated by performing proportional / integral control as shown in FIG. . Then, based on the torque command Tc obtained in the speed control loop, the processing of the current control loop having a shorter processing cycle than the processing of the speed control loop is started. The processes of the position control loop and the speed control loop are the same as those in the related art, and the process of the current control loop according to the embodiment of the present invention will be described with the flowchart of FIG. .

First, the actual speed V of the motor and the rotor position (the rotor phase in the R phase based on the R phase) are read (step 10).
1) From the read actual speed V of the motor, register R
The actual speed V obtained in the previous cycle of the current control loop process stored in (V) is subtracted to obtain the acceleration α, and the read actual speed V is stored in the register R (V) (steps 102 and 10).
3). Next, the acceleration α obtained in step 102 is multiplied by a constant ka for converting the acceleration into a current unit, and a value α ·
Calculate ka (step 104). Next, from the rotor position read in step 101, the sine values sin θ, sin (θ
+ 2π / 3), sin (θ−2π / 3), and sine values θ ′, sin (θ ′ + 2π / 3), sin (θ ′) for a phase θ ′ (= θ + Δθ) advanced by an amount Δθ smaller than θ. -2π / 3) is read from the ROM storing the sine value corresponding to the phase (step 10).
Five). Then, by multiplying the torque command value Tc obtained in the speed control loop process by the sine value of each phase whose phase has been obtained by the set amount Δθ, the current command values Irc, Isc, Itc of each phase of R, S, and T are obtained. .

Irc = Tc · sin θ ′ (1) Isc = Tc · sin (θ ′ + 2π / 3) (2) Itc = Tc · sin (θ′−2π / 3) (3) Sinθ, sin (θ + 2π / 3), sin (θ-2π /
Multiply by 3) to obtain values Irf, Isf, Itf corresponding to the acceleration of each phase (step 106).

Irf = α · ka sin θ (4) Isf = α · ka sin (θ + 2π / 3) (5) Itf = α · ka sin (θ−2π / 3) (6) Current command values Irc, Isc, It
The current values Irf, Isf, Itf of each phase corresponding to the acceleration are subtracted from c, respectively, and multiplied by the integral gain k1 of the current control loop, and the registers Rr (s), Rs (s), Rt of each phase functioning as integrators
(S). That is, an integration process corresponding to the transfer function 16 in FIG. 1 is performed (step 107).

Rr (s) ← Rr (s) + k1 (Irc−Irf) (7) Rs (s) ← Rs (s) + k1 (Isc−Isf)... (8) Rt (s) ← Rt (s) + k1 (Itc-Itf) (9) Next, the actual currents Ir, Is, It of each phase are read (step 10).
8) Multiply by the control loop proportional gain k2 (step 10)
9), this value is stored in the register Rr (s), Rs
(S) and Rt (s) are subtracted from each other to calculate the PWM signal of each phase.

PWMr = Rr (s) −k2 · Ir (10) PWMs = Rs (s) −k2 · Is (11) PWMt = Rt (s) −k2 · It (12) PWM signal of each phase PWMr, PWMs, PWMt
Is output to the inverter, and the inverter is switched to drive the motor (step 110).

As described above, the processing of the processor of the current control loop in the servo control is performed, but in the present embodiment,
A phase θ ′ (= θ) in which a sine value multiplied by the torque command value Tc is advanced by a set amount Δθ by the phase of each phase at the rotor position.
+ Δθ), the two phases may be the same. In this case, for example, the following (13)
It becomes an expression.

Irc−Irf = Tc sin θ−a · ka sin θ = (Tc−α ·) sin θ (13) As a result, the value obtained by multiplying the acceleration α by a constant ka from the torque command value Tc calculated in the speed control loop processing is calculated as follows. The subtracted value is multiplied by sin θ, sin (θ + 2π / 3) and sin (θ-2π / 3), and is input to the integrator (16) as a current command value for each phase. That is, the processing of equations (7) to (9), which is the processing of step 107, is as follows.

Rr (s) = Rr (s) + k1 · (Tc−α · ka) · sin θ (14) Rs (s) = Rs (s) + k1 · (Tc−α · ka) · sin (θ + 2π / θ) (15) Rt (s) = Rt (s) + k1 · (Tc−α · ka) · sin (θ + 2π / θ) (16) In the above embodiment as well, the current control loop is the same as in the prior art. The proportional term (transfer function 20) uses the actual currents Ir, Is, It to monitor the actual current, prevent oscillation and runaway of the current control loop, and protect the power amplifier (inverter) and motor. ing.

Effect of the Invention Since the influence of the disturbance first appears as a change in speed, that is, acceleration, in the present invention, the current is controlled to suppress the disturbance in the current control loop processing by using the acceleration. Disturbance suppression is performed in a loop, and disturbance suppression can be performed even for a high-frequency disturbance that cannot be suppressed by the control of the speed control loop, so that the disturbance suppression characteristics of the servo system can be improved.

[Brief description of the drawings]

FIG. 1 is a block diagram of a speed control loop and a current control loop of a three-phase synchronous motor in one embodiment of the present invention. FIG. 2 is a flowchart of a current control loop process by a servo control processor in the same embodiment. FIG.
FIG. 5 is a block diagram of a speed control loop and a current control loop of a conventional three-phase synchronous motor. Vc: Speed command, Tc: Torque command, Irc, Isc, Itc ...
R, S, T-phase current command, Irf, Isf, Itf ... Corresponding to acceleration
R, S, and T phase current values, Td ... disturbance torque, V ... actual motor speed, α ... motor acceleration, k1v ... speed loop integral gain, k2v ... speed loop proportional gain, k1 ... Current loop integral gain, k2 ... current loop proportional gain, R ...
... winding resistance, L ... winding inductance, Kt ... torque constant, J ... motor rotor inertia, ka ... constant for converting acceleration into current units.

Claims (3)

(57) [Claims] In a current control loop in controlling an AC synchronous motor used as a servomotor, an actual speed of a motor is determined from a torque command value obtained by a speed control loop and a current command value of each phase obtained by a rotor position. The actual acceleration obtained by differentiation is converted into the dimension of the current command, and the value obtained by subtracting the value of each phase obtained according to the rotor position is used as the input to the integrator of the current control loop of each phase. Characteristic synchronous motor control method. 2. The phase of each phase obtained from the current command value of each phase obtained by multiplying the torque command value by a sine value of a phase advanced by a set amount from the phase of each phase obtained by the rotor position. 2. The synchronous motor control method according to claim 1, wherein the value obtained by multiplying the value obtained from the actual acceleration by the sine value is subtracted and input to the integrators of the respective current control loops. 3. A current control loop for controlling an AC synchronous motor used as a servomotor, wherein an actual acceleration obtained by differentiating an actual speed of the motor from a torque command value obtained by a speed control loop is converted into a current value. The current command value of each phase is obtained based on the value obtained by subtracting the value and the rotor position, and the current command value of each phase is input to the integrator of the current control loop of each phase. Control method for synchronous motor.