JP2024099920A - Inverter control method and inverter control device - Google Patents

Abstract

[Problem] To provide an inverter control method and an inverter control device that suppress erroneous detection by a detector that uses the ground as a reference potential when performing pulse width modulation control of a method of varying the neutral point potential of a rotating electric machine. [Solution] A fluctuation frequency fvn of the neutral point potential of a rotating electric machine 11 is compared with a cutoff frequency fτ of a noise reduction filter of a detector (insulation resistance sensor 21) that uses the ground as a reference potential. Then, when the fluctuation frequency fvn is equal to or lower than the cutoff frequency fτ, the modulation method of the pulse width modulation control is changed from a first modulation method (default two-phase modulation method) selected according to the temperature of an inverter 16 (power element temperature Tpm) to a second modulation method (custom two-phase modulation method) different from the first modulation method for a preset predetermined time (maximum duration T2). [Selected Figure] FIG. 4

Description

The present invention relates to an inverter control method and control device.

Patent Document 1 discloses a rotating electric machine control device that switches inverter control from three-phase modulation to two-phase modulation in order to reduce noise in a resolver that detects the rotation angle of the rotating electric machine.

JP 2009-183092 A

Driving control of a rotating electric machine is performed by various pulse width modulation control methods depending on the state of the rotating electric machine and the inverter that drives it. Specific methods of pulse width modulation control include three-phase modulation, as well as two-phase modulation, for example. Three-phase modulation is a method that does not fluctuate the neutral point potential of the rotating electric machine. In contrast, two-phase modulation can reduce inverter switching more than three-phase modulation, but fluctuates the neutral point potential.

When performing pulse width modulation control using a method that varies the neutral point potential, such as two-phase modulation, if the variation in neutral point potential propagates to the ground via stray capacitance, a detector that uses ground as its reference potential may make a false detection. For example, an insulation resistance sensor injects a signal into a high-voltage system and detects insulation resistance based on the voltage feedback of the ground reference. Generally, such feedback signals use a filter to reduce noise. In this case, if the variation in neutral point potential that is lower than the cutoff frequency of the filter propagates to the ground, a false detection may occur.

Therefore, when performing pulse width modulation control that varies the neutral point potential, it is necessary to prevent erroneous detection by a detector that uses ground as its reference potential.

The present invention aims to provide an inverter control method and an inverter control device that can suppress erroneous detection by a detector that uses ground as a reference potential when performing pulse width modulation control that varies the neutral point potential of a rotating electric machine.

One aspect of the present invention is an inverter control method for controlling an inverter in an electric circuit including a detector including a noise reduction filter, an inverter that converts DC power to AC power by pulse width modulation control, and a rotating electric machine connected to the inverter, with the ground as a reference potential. In this inverter control method, the fluctuation frequency of the neutral point potential of the rotating electric machine is compared with the cutoff frequency of the noise reduction filter. Then, when the fluctuation frequency is equal to or lower than the cutoff frequency, the modulation method of the pulse width modulation control is changed from a first modulation method selected according to the inverter temperature to a second modulation method different from the first modulation method for a predetermined time.

The present invention provides an inverter control method and an inverter control device that suppresses erroneous detection by a detector that uses ground as a reference potential when performing pulse width modulation control that varies the neutral point potential of a rotating electric machine.

FIG. 1 is a block diagram showing a schematic configuration of an electric vehicle. FIG. 2 is a circuit diagram showing the configuration of an electric circuit that drives the rotating electric machine. FIG. 3 is a block diagram showing the configuration of the modulation method setting unit. FIG. 4 is a flowchart relating to switching of the modulation method. FIG. 5 is a graph showing the default two-phase modulation control. FIG. 6 is a graph showing the case where custom two-phase modulation control is inserted. FIG. 7 is a graph showing the spectrum of noise propagating to the ground when the neutral point potential fluctuates due to two-phase modulation control. FIG. 8 is a flowchart of a modified example relating to switching of the modulation method. FIG. 9 is a flowchart of a modified example relating to switching of the modulation method. FIG. 10 is a flowchart of a modified example relating to switching of the modulation method.

The following describes an embodiment of the present invention with reference to the drawings.

[Embodiment]
Fig. 1 is a block diagram showing a schematic configuration of an electric vehicle 100. The electric vehicle 100 is, for example, an electric car or a hybrid vehicle. As shown in Fig. 1, the electric vehicle 100 includes a rotating electric machine 11, a current command calculation unit 12, a current control unit 13, a coordinate conversion unit 14, a modulation method setting unit 15, an inverter 16 (INV), a battery 17, a rotation state detection unit 18, and a coordinate conversion unit 19.

The rotating electric machine 11 is a motor or a generator. In this embodiment, the rotating electric machine 11 is a drive motor that drives the electric vehicle 100, and the electric vehicle 100 is driven by the torque generated by the rotating electric machine 11. The rotating electric machine 11 also functions as a generator, for example, when the electric vehicle 100 performs so-called regenerative control. In this embodiment, the rotating electric machine 11 is a three-phase AC synchronous rotating electric machine (particularly a three-phase synchronous motor), and is driven by adjusting the power supplied to each of the UVW phases.

The current command calculation unit 12 calculates a d-axis current command _Id ^* and a q-axis current command Iq ^* (hereinafter referred to as dq-axis current commands _Id ^* , _Iq ^* ) based on the torque command T ^* and the electrical angular velocity ω of the rotating electric machine 11. The dq-axis current commands _Id ^* , _Iq ^* are command values for currents to be passed through the d-axis and _q -axis of the rotating electric machine 11 in order to cause the rotating electric machine 11 to generate the torque T required by the torque command T ^* . The current command calculation unit 12 has, for example, a current command map that associates the torque command T ^* and the electrical angular velocity ω with the dq-axis current commands _Id ^* , _Iq ^* , and calculates the dq-axis current commands _Id ^* , _Iq ^* corresponding to the torque command T ^* and the electrical angular velocity ω by referring to this. The current command map is determined in advance by an experiment, a simulation, or the like. Further, the torque command T ^* is calculated by a higher-level controller (not shown) based on the amount of operation of the accelerator pedal, etc.

The current control unit 13 calculates a d-axis voltage command _Vd ^* and a q-axis voltage command _Vq ^* (hereinafter referred to as dq-axis voltage commands _Vd ^* , _Vq ^* ). The current control unit 13 acquires the dq-axis current commands _Id ^* , _Iq ^* and the d-axis current _Id and the q-axis current _Iq (hereinafter referred to as dq-axis currents _Id , _Iq ) that actually flow through the rotating electric machine 11. Then, the current control unit 13 determines the dq-axis voltage commands _Vd ^* , _Vq ^* so that the difference between the dq-axis current commands _Id ^* , _Iq ^* and the actual dq-axis currents _Id , _Iq becomes zero.

The current control unit 13 can calculate the d-axis and q-axis voltage commands _Vd ^* and _Vq ^* by, for example, proportional-integral (PI) control shown in the following equation (1). In equation (1), _Ld and _Lq are the inductances of the d-axis and q-axis, _Kpd and _Kpq are the proportional gains of the d-axis and q-axis, _Kid and _Kiq are the integral gains of the d-axis and q-axis, φ is the permanent magnet flux linkage number, and ω is the electrical angular speed of the rotating electric machine 11.

The coordinate conversion unit 14 converts the dq-axis voltage commands V _d ^* , V _q ^* into three-phase AC voltage commands V _U1 ^* , V _V1 ^* , V _W1 ^* by coordinate conversion based on the magnetic pole position θ (detected value) of the rotating electric machine 11 .

The modulation method setting unit 15 calculates the final voltage commands _{VUf*, VVf*, and VWf* to be input to the inverter 16 from the three-phase AC voltage commands VU1} ^* _{, VV1} ^* _, ^and _VW1 ^* based on the battery voltage _Vb and the power element temperature Tpm. The modulation method setting unit 15 may further use the refrigerant temperature _Twtr in the calculation of the final voltage commands _VUf ^{* , VVf*} _, ^{and VWf*. In this embodiment, the modulation method setting unit 15 calculates the final voltage commands VUf*, VVf*} _, ^and _VWf ^* _based ^on _the ^battery _{voltage Vb} and _the power element temperature _Tpm .

The battery voltage _Vb is the output voltage of the battery 17. The power element temperature _Tpm is the temperature of the power elements (switching elements) included in the inverter 16. The coolant temperature _Twtr is the temperature of the inverter 16, particularly the water or other coolant that cools the power elements included in the inverter 16. The final voltage commands _VUf ^* , _VVf ^* , _VWf ^* are final three-phase AC voltage commands to be input to the inverter 16 in order to control the rotating electric machine 11.

In this embodiment, the modulation method setting unit 15 sets or changes the drive method of the inverter 16 according to the final voltage commands _VUf ^* , _VVf ^* , and _VWf ^* . That is, the modulation method setting unit 15 switches the control method of the inverter 16, which is driven by the modulation method, i.e., pulse width modulation control (hereinafter referred to as PWM control), according to the final voltage commands _VUf ^* , _VVf ^* , and _VWf ^* input to the inverter 16. Specifically, the modulation method setting unit 15 can set or change (switch) the modulation method to at least one of a three-phase modulation method or a two-phase modulation method.

The three-phase modulation method is a method of controlling the output power of the inverter 16 by modulating the potentials of all three phases of UVW by PWM control. The sinusoidal three-phase modulation method is the most common driving method for the inverter 16. In the sinusoidal three-phase modulation method, the output power waveforms to the UVW phases are controlled to be sinusoidal. In the three-phase modulation method, the potential at the neutral point 23 (see FIG. 2) of the rotating electric machine 11 (hereinafter referred to as the neutral point potential) is small on average. Hereinafter, the control for driving the inverter 16 by the three-phase modulation method may be referred to as three-phase modulation control or simply as three-phase modulation. In the three-phase modulation control, the three-phase AC voltage commands V _U1 ^* , V _V1 ^* , and V _W1 ^* are output as they are as the final voltage commands V _Uf ^* , V _Vf ^* , and V _Wf ^* .

The two-phase modulation method is a method of controlling the output power of the inverter 16 by fixing the potential of one of the three phases, UVW, at least temporarily, and modulating the potentials of the other two phases by PWM control during that time. In the two-phase modulation method, the output power waveform to each of the UVW phases is also controlled to a sine wave substantially similar to that in the three-phase modulation method. In the two-phase modulation method, the switching operation of the inverter 16 is reduced for one phase whose potential is fixed, so that heat generation (switching loss) of the inverter 16 is suppressed compared to the three-phase modulation method. Therefore, the two-phase modulation method is selected when the temperature of the inverter 16 is determined to be high based on the power element temperature T _pm , the refrigerant temperature T _wtr , etc. For example, the two-phase modulation method is selected when there is a risk of exceeding the heat resistance temperature of the power element. However, in the two-phase modulation method, the neutral point potential of the rotating electric machine 11 fluctuates periodically.

In particular, the modulation method setting unit 15 can drive the inverter 16 using two types of two-phase modulation methods: a default two-phase modulation method and a custom two-phase modulation method.

The default two-phase modulation method is a two-phase modulation method that controls the output power of the inverter 16 by fixing the potential of one of the UVW phases that is the maximum or minimum, and modulating the potentials of the other two phases by PWM control. In other words, the default two-phase modulation method is a standard two-phase modulation method.

The custom two-phase modulation method is a two-phase modulation method that fixes the potential of one of the UVW phases that is different from the default two-phase modulation method, and modulates the potential of the other two phases by PWM control, thereby controlling the output power of the inverter 16. In the custom two-phase modulation method of this embodiment, of the two UVW phases, excluding the one phase with the maximum or minimum potential, the potential of the one phase with the minimum or maximum potential is fixed, and the potentials of the remaining two phases are modulated by PWM control. For example, in a scene where the U phase is at the maximum potential, the V phase is at the minimum potential, and the W phase is at an intermediate potential, the default two-phase modulation method fixes the potential of the U phase, while the custom two-phase modulation method fixes the potential of the V phase.

Hereinafter, the control for driving the inverter 16 using the two-phase modulation method may be referred to as two-phase modulation control or simply as two-phase modulation. The control for driving the inverter 16 using the default two-phase modulation method may be referred to as default two-phase modulation control or simply as default two-phase modulation. Also, the control for driving the inverter 16 using a custom two-phase modulation method may be referred to as custom two-phase modulation control or simply as custom two-phase modulation.

When executing default two-phase modulation control, the modulation method setting unit 15 calculates three-phase AC voltage commands (hereinafter referred to as default two-phase modulation voltage commands _VU2 ^* , _VV2 ^* , _VW2 ^* ) in which the potential of one phase that is maximum or minimum among the UVW phases is fixed, and outputs this as the final voltage commands _VUf ^* , _VVf ^* , _VWf ^* . When executing custom two-phase modulation control, the modulation method setting unit 15 calculates three-phase AC voltage commands (hereinafter referred to as custom two-phase modulation voltage commands _VU3 ^* , _VV3 ^* , _VW3 ^* ) in which the potential of one phase different from that of the default two-phase modulation control is fixed, and outputs this as the final voltage commands _VUf ^* , _VVf ^* , _VWf ^* . In two-phase modulation control, ie, default two-phase modulation control or custom two-phase modulation control, the potential of one phase to be fixed is the maximum positive value (eg, +V _b /2) or the maximum negative value (eg, −V _b /2).

In addition to setting or changing the modulation method as described above, the modulation method setting unit 15 can set or change the frequency of the carrier signal (carrier wave) used in the PWM control. The carrier signal used in the PWM control is, for example, a triangular wave whose frequency (hereinafter referred to as carrier frequency f _c ) is from several kHz to several tens of kHz. In principle, in both the three-phase modulation control and the two-phase modulation control, the modulation method setting unit 15 uses a carrier signal having a predetermined normal carrier frequency (hereinafter referred to as normal carrier frequency f _c0 ) in the two-phase modulation control and the three-phase modulation control. However, the modulation method setting unit 15 may use a carrier signal having a carrier frequency (hereinafter referred to as low carrier frequency f _c1 ) lower than the normal carrier frequency f _c0 in the two-phase modulation control, the three-phase modulation control, or the two-phase modulation control and the three-phase modulation control, as necessary. If the carrier frequency f _c is lowered, the number (frequency) of switching of the power elements in the inverter 16 is reduced. Therefore, when the carrier frequency f _c is changed to a low carrier frequency f _c , the switching loss of the power elements and the heat generated due to the switching loss are reduced.

The inverter 16 converts between DC power and AC power. When the rotating electric machine 11 is driven using the power of the battery 17, the inverter 16 converts the DC power output by the battery 17 into AC power and supplies it to the rotating electric machine 11. On the other hand, when the rotating electric machine 11 generates power by regenerative control, the inverter 16 converts the AC power generated by the rotating electric machine 11 into DC power and inputs it to the battery 17, thereby charging the battery 17. As described above, the inverter 16 is driven by PWM control, which is either three-phase modulation control, default two-phase modulation control, or custom two-phase modulation control.

In this embodiment, for convenience of explanation, the inverter 16 includes sensors (not shown) for measuring the battery voltage _Vb , which is the output power of the battery 17, the power element temperature _Tpm , and the coolant temperature _Twtr . Therefore, the modulation method setting unit 15 can appropriately acquire the battery voltage _Vb , the power element temperature _Tpm , and the coolant temperature _Twtr from the inverter 16.

The battery 17 is a secondary battery that can be discharged and charged, such as a lithium ion battery. In this embodiment, the electric vehicle 100 is driven by the power of the battery 17. The battery 17 is charged by the power generated by the rotating electric machine 11 when braking the electric vehicle 100, and may also be charged by an external power source or a power generation system mounted on the electric vehicle 100.

The rotation state detection unit 18 detects the rotation state of the rotating electric machine 11. In this embodiment, the rotation state detection unit 18 detects the magnetic pole position θ of the rotating electric machine 11 and the electrical angular velocity ω of the rotating electric machine 11 based on a signal output by a rotation detector 20 such as a resolver. As described above, in this embodiment, the electrical angular velocity ω is used in the current command calculation unit 12 and the modulation method setting unit 15. Furthermore, the magnetic pole position θ is used in the coordinate conversion unit 14 and also in the coordinate conversion unit 19.

The coordinate conversion unit 19 converts the three-phase AC currents _IU , _IV , _IW actually flowing through the rotating electric machine 11 into dq-axis currents _Id , _Iq by coordinate conversion based on the magnetic pole position θ. As described above, the dq-axis currents _Id , _Iq are used by the current control unit 13. All or part of the three-phase AC currents _IU , _IV , _IW are detected by a current detector (not shown). In addition, when the currents of two phases of the three-phase AC currents _IU , _IV , _IW are detected, the current of the remaining phase is obtained by calculation.

Of the above components of the electric vehicle 100, all or part of the current command calculation unit 12, current control unit 13, coordinate conversion unit 14, modulation method setting unit 15, rotation state detection unit 18, and coordinate conversion unit 19 are configured by one or more computers (so-called controllers). This controller is programmed to execute the calculations in each of the above units at a predetermined control period. This controller, or a portion of this controller including the modulation method setting unit 15, configures a control device for the inverter 16 (inverter control device).

Figure 2 is a circuit diagram showing the configuration of an electric circuit 101 that drives the rotating electric machine 11. As shown in Figure 2, the electric circuit 101 includes a battery 17, an inverter 16, and the rotating electric machine 11, as well as, for example, an insulation resistance sensor 21.

The inverter 16 is composed of a smoothing capacitor 22 and three-phase legs. The smoothing capacitor 22 smoothes the power input from the battery 17 or the power input to the battery 17. Each phase leg is composed of an upper arm and a lower arm. The stator coils 22u, 22v, and 22w of each phase included in the rotating electric machine 11 are connected between the upper arm and the lower arm of the corresponding leg. Specifically, the U-phase leg is composed of an upper arm UP and a lower arm UN, and one end of the U-phase stator coil 22u is connected between them. The V-phase leg is composed of an upper arm VP and a lower arm VN, and one end of the V-phase stator coil 22v is connected. Similarly, the W-phase leg is composed of an upper arm WP and a lower arm WN, and one end of the W-phase stator coil 22w is connected. The stator coils 22u, 22v, and 22w of each phase are connected at the other end.

The other end where these stator coils 22u, 22v, 22w are connected to each other is the so-called neutral point 23. As mentioned above, the neutral point potential does not fluctuate when three-phase modulation control is performed, but fluctuates when two-phase modulation control is performed. And, the neutral point 23 is apparently electrically connected to the ground (GND) via the stray capacitance Cs. Therefore, when the neutral point potential fluctuates, the fluctuation may propagate to the ground.

Each of the upper arms UP, VP, WP and the lower arms UN, VN, WN of each phase is composed of a power element (switching element) such as an IGBT (insulated gate bipolar transistor) and a free wheel diode. The inverter 16 converts the output power of the battery 17 into AC power or converts the AC power generated by the rotating electric machine 11 into DC power by switching the power elements of these arms. In particular, in the two-phase modulation control, the power element of any one of the upper arms UP, VP, WP and the lower arms UN, VN, WN of each phase is kept on, and the switching is essentially stopped, thereby fixing the potential of that phase. For example, when the potential of the U phase is fixed to the maximum positive value (+V _b /2) in the two-phase modulation control, the upper arm UP of the U phase is fixed to on. Also, when the potential of the U phase is fixed to the maximum negative value (-V _b /2) in the two-phase modulation control, the lower arm UN of the U phase is fixed to on. Hereinafter, an arm that is fixed to ON in two-phase modulation control will be referred to as a fixed arm.

The insulation resistance sensor 21 detects insulation resistance using the ground (GND) as a reference potential. In this embodiment, the insulation resistance sensor 21 detects a ground fault in high-voltage components such as the battery 17, the rotating electric machine 11, and/or the inverter 16 by detecting the insulation resistance to the ground.

For example, the insulation resistance sensor 21 connects a pulse oscillator 33 to the power supply line of the battery 17 via a capacitor 31 and a resistor 32. A low-pass filter 36 consisting of a resistor 34 and a capacitor 35 is connected to the connection point between the capacitor 31 and the resistor 32. The other end of the low-pass filter 36 is connected to the ground (GND). The connection point between the resistor 34 and the capacitor 35 is connected to a comparator 37. Therefore, when the pulse oscillator 33 inputs a predetermined pulse to the power supply line, if the insulation resistance is reduced, the amplitude (voltage) of the response pulse input to the comparator 37 fluctuates accordingly. Therefore, the comparator 37 detects the insulation resistance by comparing the amplitude of the input pulse with a reference voltage 38.

The low-pass filter 36 included in the insulation resistance sensor 21 functions as a noise reduction filter for accurately measuring the insulation resistance by reducing high-frequency noise. Therefore, when the fluctuation of the neutral point potential propagates to the ground via the stray capacitance Cs, it may become noise that may cause erroneous detection in the insulation resistance sensor 21. However, when the fluctuation frequency f _vn of the neutral point potential is sufficiently higher than the cutoff frequency f _τ of the low-pass filter 36, the noise caused by the fluctuation of the neutral point potential is reduced by the low-pass filter 36. On the other hand, when the fluctuation frequency f _vn of the neutral point potential is close to or lower than the cutoff frequency f _τ of the low-pass filter 36, the noise caused by the fluctuation of the neutral point potential passes through the low-pass filter 36, and there is a risk of causing erroneous detection in the insulation resistance sensor 21. In particular, since the fluctuation frequency _fvn of the neutral point potential is proportional to the electrical angular velocity ω of the rotating electric machine 11, when two-phase modulation control is performed in a situation where the electrical angular velocity ω of the rotating electric machine 11 is low, there is a risk that noise generated by fluctuations in the neutral point potential will pass through the low-pass filter 36.

Therefore, in this embodiment, the neutral point potential fluctuation frequency f _vn is compared with the cutoff frequency f _τ of the low-pass filter 36. Then, when the fluctuation frequency f _vn is close to or lower than the cutoff frequency f _τ , the modulation method setting unit 15 is configured to temporarily switch the modulation method of the PWM control for a predetermined time (T ₂ ) that is set in advance. More specifically, for example, when it is necessary to perform two-phase modulation control, the modulation method setting unit 15 performs two-phase modulation control by default two-phase modulation control in principle, but when the neutral point potential fluctuation frequency f _vn is close to or lower than the cutoff frequency f _τ of the low-pass filter 36, the modulation method setting unit 15 temporarily interrupts the custom two-phase modulation control while the default two-phase modulation control is being performed. In this way, the neutral point potential fluctuation frequency f _vn is forcibly made high frequency, and the modulation method of the PWM control is switched so that noise that may be generated by the fluctuation of the neutral point potential is appropriately reduced in the low-pass filter 36.

Figure 3 is a block diagram showing the configuration of the modulation method setting unit 15. As shown in Figure 3, the modulation method setting unit 15 includes a two-phase modulation execution determination unit 41, a carrier frequency setting unit 42, a neutral point potential fluctuation frequency calculation unit 43, a final voltage command calculation unit 44, a first timer 45, and a second timer 46.

The two-phase modulation execution determination unit 41 determines whether or not to execute two-phase modulation control based on the temperature of the inverter 16. In this embodiment, the two-phase modulation execution determination unit 41 uses the power element temperature T _pm as the temperature of the inverter 16. Specifically, the two-phase modulation execution determination unit 41 compares the power element temperature T _pm with a first temperature threshold value _α1 . Then, when the power element temperature T _pm is higher than the first temperature threshold value _α1 (T _pm > _α1 ) and the inverter 16 is in a high temperature state, the two-phase modulation execution determination unit 41 determines that two-phase modulation control should be executed.

As described above, the first temperature threshold _α1 is a criterion for determining whether or not to perform two-phase modulation control instead of three-phase modulation control that should normally be performed, depending on the temperature of the inverter 16. Therefore, when the power element temperature T _pm is equal to or lower than the first temperature threshold _α1 (T _pm ≦ _α1 ), the inverter 16 is within a steady temperature range, and three-phase modulation control can be continued, the two-phase modulation execution determination unit 41 determines that it is not necessary to perform two-phase modulation control and that three-phase modulation control should be performed. The first temperature threshold _α1 is determined in advance by experiment, simulation, or the like.

The carrier frequency setting unit 42 determines the carrier frequency f _c based on the temperature of the inverter 16. In this embodiment, the carrier frequency setting unit 42 uses the power element temperature T _pm as the temperature of the inverter 16. Specifically, the carrier frequency setting unit 42 compares the power element temperature T _pm with a second temperature threshold value _α2 . Then, when the power element temperature T _pm is higher than the second temperature threshold value _α2 (T _pm > _α2 ) and the inverter 16 has reached a particularly high temperature state, the carrier frequency setting unit 42 changes the carrier frequency f _c in various PWM controls from the normal carrier frequency f _c0 to a low carrier frequency f _c1 .

As described above, the second temperature threshold _α2 is a criterion for determining whether or not it is necessary to lower the normal carrier frequency f _c0 to the low carrier frequency f _c1 depending on the temperature of the inverter 16. Therefore, when the power element temperature T _pm is equal to or lower than the second temperature threshold _α2 (T _pm ≦α ₂ ), the carrier frequency setting unit 42 maintains the carrier frequency f _c in various PWM controls at the normal carrier frequency f _c0 . That is, even if the inverter 16 is at a certain high temperature, when the high temperature state has not yet reached a state in which the carrier frequency f _c must be reduced, the carrier frequency f _c is maintained at the normal carrier frequency f _c0 . The second temperature threshold _α2 is determined in advance by experiment, simulation, or the like within a range at least greater than the first temperature threshold _α1 (α ₂ >α ₁ ).

In this embodiment, the carrier frequency setting unit 42 determines the low carrier frequency f _c1 by calculation according to the following formulas (2) and (3). In formula (2), T ₁ is the maximum time for which a specific arm is fixed to on by default two-phase modulation control (hereinafter referred to as the maximum duration T ₁ of default two-phase modulation control). On the other hand, T ₂ is the maximum time for which another arm is fixed to on by custom two-phase modulation control (hereinafter referred to as the maximum duration T ₂ of custom two-phase modulation control). Therefore, T ₁ /(T ₁ +T ₂ ) is the time ratio (hereinafter referred to as the switching ratio κ) for switching between the default two-phase modulation control and the custom two-phase modulation control when performing two-phase modulation control. That is, the carrier frequency setting unit 42 calculates the low carrier frequency f _c1 according to the switching ratio κ by multiplying the normal carrier frequency f _c0 by a coefficient (1.5-κ) according to the switching ratio κ.

When performing two-phase modulation control, the maximum duration _T1 of the default two-phase modulation control and the maximum duration _T2 of the custom two-phase modulation control are determined in advance by experiment, simulation, etc. In other words, the switching ratio κ is determined in advance by experiment, simulation, etc. Specific maximum durations _T1 , _T2 and switching ratio κ of each two-phase modulation control may be determined to change depending on, for example, the temperature of the inverter 16.

In particular, the maximum durations T ₁ and T ₂ of each two-phase modulation control are determined so as to satisfy the following formula (3) in relation to the cutoff frequency f _τ of the low-pass filter 36. That is, in the two-phase modulation control, the maximum durations T ₁ and T ₂ are set so that the change frequency of the fixed arm is at least higher than the cutoff frequency f _τ of the low-pass filter 36. As shown in the following formula (4), the maximum durations T ₁ and T ₂ are determined so that the switching ratio κ is in the range of 0.5 or more and less than 1 (0.5≦κ<1). Therefore, in the calculation of the low carrier frequency f _c1 , the coefficient by which the normal carrier frequency f _c0 is multiplied varies in the range of more than 0.5 and less than 1 (0.5<1.5-κ≦1). Therefore, the low carrier frequency f _c1 calculated by the carrier frequency setting unit 42 is at least equal to or less than the normal carrier frequency f _c0 .

In addition, the carrier frequency setting unit 42 can determine the low carrier frequency f _c1 regardless of the switching ratio κ. In this case, the carrier frequency setting unit 42 can set the low carrier frequency f _c1 to a substantially fixed value, for example, by f _c1 =0.5×f _c0 . That is, the low carrier frequency f _c1 is set to ½ or less of the normal carrier frequency f _c0 . This means that when it is necessary to lower the carrier frequency f _c , the carrier frequency f _c is lowered to the lowest possible frequency according to the case where the temperature of the inverter 16 is under the most severe condition, regardless of the specific temperature of the inverter 16, etc. In this embodiment, in particular, the low carrier frequency f _c1 is set to ½ of the normal carrier frequency f _c0 .

In this embodiment, the modulation method setting unit 15 includes the carrier frequency setting unit 42, but the carrier frequency setting unit 42 may be omitted. That is, when the modulation method setting unit 15 is configured more simply, a carrier signal of the normal carrier frequency f _c0 may be continuously used in three-phase modulation control and two-phase modulation control, regardless of the temperature of the inverter 16, etc.

The neutral point potential fluctuation frequency calculation unit 43 calculates the neutral point potential based on the electrical angular velocity ω of the rotating electric machine 11 at least when two-phase modulation control is executed. In this embodiment, the neutral point potential fluctuation frequency calculation unit 43 particularly calculates the fluctuation frequency _fvn of the neutral point potential. Specifically, the neutral point potential fluctuation frequency calculation unit 43 calculates (estimates) the fluctuation frequency _fvn of the neutral point potential based on the following equation (5). Note that the modulation method setting unit 15 may obtain the fluctuation frequency _fvn of the neutral point potential by measurement or other methods.

The final voltage command calculation unit 44 outputs the final voltage commands ( _VUf ^* , _VVf ^* , _VWf ^* ) according to the determination result of the two-phase modulation execution determination unit 41. When the two-phase modulation execution determination unit 41 determines that it is not necessary to execute two-phase modulation control and that three-phase modulation control should be executed, the final voltage command calculation unit 44 outputs the three-phase AC voltage commands ( _VU1 ^* , _VV1 ^* , _VW1 ^* ) as they are as the final voltage commands ( _VUf ^* , _VVf ^* , _VWf ^* ). As a result, the inverter 16 is driven by three-phase modulation control.

On the other hand, when the two-phase modulation execution determination unit 41 determines that two-phase modulation control should be executed, the final voltage command calculation unit 44 calculates default two-phase modulation voltage commands ( _VU2 ^* , _VV2 ^* , _VW2 ^* ) and/or custom two-phase modulation voltage commands ( _VU3 ^* , _VV3 ^* , _VW3 ^* ) and outputs one of them as the final voltage command ( _VUf ^* , _VVf ^* , _VWf ^* ). As a result, the inverter 16 is driven by the two-phase modulation control. Then, when the final voltage commands ( _VUf ^* , _VVf ^* , _VWf ^* ) are the default two-phase modulation voltage commands ( _VU2 ^* , _VV2 ^* , _VW2 ^* ), the inverter 16 is driven by the default two-phase modulation control. Similarly, when the final voltage commands (V _Uf ^* , V _Vf ^* , V _Wf ^* ) are the custom two-phase modulation voltage commands (V _U3 ^* , V _V3 ^* , V _W3 ^* ), the inverter 16 is driven under the custom two-phase modulation control.

The final voltage command calculation unit 44 calculates the default two-phase modulation voltage commands ( _VU2 ^* , _VV2 ^* , _VW2 ^* ) and the custom two-phase modulation voltage commands ( _VU3 ^* , _VV3 ^* , _VW3 ^* ) using the three-phase AC voltage commands ( _VU1 ^* , _VV1 ^* , _VW1 ^* ) and the battery voltage _Vb .

Specifically, the final voltage command calculation unit 44 first calculates the maximum command _Vp ^* and the minimum command _Vn ^* according to the following equation (6). As a result, the final voltage command calculation unit 44 determines which of the three-phase AC voltage commands ( _VU1 ^* , _VV1 ^* , _VW1 ^* ) is the maximum, and which of the three-phase AC voltage commands ( _VU1 ^* , _VV1 ^* , _VW1 ^* ) is the minimum.

Then, the final voltage command calculation unit 44 calculates the default two-phase modulation voltage commands ( _VU2 ^* , _VV2 *, _VW2 ^* ) and the custom two-phase modulation voltage commands ( _VU3 ^* , _VV3 ^* , _VW3 ^* ) according to the following formula (7) or (8) depending on ^{the magnitude relationship between the maximum command Vp*} _and ^the minimum command _Vn ^* . In formulas (7) and (8), x=2, 3. When x=2, the voltage command calculated by formula (7) or (8) is the default two-phase modulation voltage command ( _VU2 ^* , _VV2 ^* , _VW2 ^* ), and when x=3, the voltage command calculated by formula (7) or (8) is the custom two-phase modulation voltage command ( _VU3 ^* , _VV3 ^* , _VW3 ^* ).

When | _Vp ^* |≧| _Vn ^* |, the final voltage command calculation unit 44 calculates the default two-phase modulation voltage commands ( _VU2 ^* , _VV2 ^* , _VW2 ^* ) according to formula (7). As an example, in a control scene during a period where _VU1 ^* >0> _VV1 ^* > _VW1 ^{*, Vp*=VU1*} _, ^so _the ^default two-phase modulation voltage commands ( _VU2 ^* , _VV2 ^* , _VW2 ^* ) are ( _VU2 ^* , _VV2 ^* , _VW2 ^* )=(+ _Vb /2, _VV1 ^* +( _Vb /2- _VU1 ^* ), _VW1 ^* +( _Vb /2- _VU1 ^* )). That is, in default two-phase modulation control, the potential of the U phase, which is the maximum in the three-phase AC voltage command (V _U1 ^* , V _V1 ^* , V _W1 ^* ), is fixed to +V _b /2 (maximum positive value), and the potentials of the V phase and W phase are controlled.

Furthermore, when | _Vp ^* |≧| _Vn ^* |, the final voltage command calculation unit 44 calculates the customized two-phase modulation voltage commands ( _VU3 ^* , _VV3 ^* , _VW3 ^* ) according to formula (8). As in the above, in the case of a control scene during a period where _VU1 ^* >0> _VV1 ^* > _{VW1*, Vn*=VW1} ^* _, ^so _the ^customized two-phase modulation voltage commands ( _VU3 ^* , _VV3 ^* , _VW3 ^* ) are ( _VU3 ^* , _VV3 ^* , _VW3 ^* )=( _VU1 ^* +( _Vb /2- _VW1 ^* ), VV1 ^* +( _{Vb /} 2- _VW1 ^* ), _-Vb /2). That is, in customized two-phase modulation control, the potential of the W phase, which is the minimum in the three-phase AC voltage commands (V _U1 ^* , V _V1 ^* , V _W1 ^* ), is fixed to −V _b /2 (the maximum negative value), and the potentials of the U phase and V phase are controlled. Therefore, in customized two-phase modulation control, the potentials of the phases different from those in default two-phase modulation control are fixed, and the sign of the fixed potential is also opposite.

On the other hand, when | _Vp ^* |<| _Vn ^* |, the final voltage command calculation unit 44 calculates the default two-phase modulation voltage commands ( _VU2 ^* , _VV2 ^* , _VW2 ^* ) according to formula (8). Taking a control scene during a period where _VU1 ^* > _VV1 ^* >0> _VW1 ^* as an example, _Vn ^* = _VW1 ^* , so the default two-phase modulation voltage commands ( _VU2 ^* , _VV2 ^* , _VW2 ^* ) are ( _VU2 ^* , _VV2 ^* , _VW2 ^* )=( _VU1 ^* +( _Vb /2- _VW1 ^* ), _VV1 ^* +( _Vb /2- _VW1 ^* ), _-Vb /2). That is, in the default two-phase modulation control, the potential of the W phase, which is the minimum in the three-phase AC voltage commands (V _U1 ^* , V _V1 ^* , V _W1 ^* ), is fixed to −V _b /2 (maximum negative value), and the potentials of the U phase and V phase are controlled.

Furthermore, when | _Vp ^* |<| _Vn ^* |, the final voltage command calculation unit 44 calculates the customized two-phase modulation voltage commands ( _VU3 ^* , _VV3 ^* , _VW3 ^* ) according to formula (7). Taking a control scene during a period where _VU1 ^* > _VV1 ^* >0> _VW1 ^* as an example, _Vp ^* = _VU1 ^* , and therefore the customized two-phase modulation voltage commands ( _VU3 ^* , _VV3 ^* , _VW3 ^* ) are ( _VU3 ^* , _VV3 ^* , _VW3 ^* )=(+ _Vb /2, _VV1 ^* +( _Vb /2- _VU1 ^* ), _VW1 ^* +( _Vb /2- _VU1 ^* )). That is, in custom two-phase modulation control, the potential of the U phase, which is maximum in the three-phase AC voltage commands ( _VU1 ^* , _VV1 ^* , _VW1 ^* ), is fixed to + _Vb /2 (maximum positive value), and the potentials of the V phase and W phase are controlled. Therefore, in custom two-phase modulation control, the potentials of the phases different from those in default two-phase modulation control are fixed, and the sign of the fixed potential is also opposite.

The final voltage command calculation unit 44 determines whether to temporarily switch from default two-phase modulation control to custom two-phase modulation control as follows: That is, the final voltage command calculation unit 44 determines which of the default two-phase modulation voltage commands ( _VU2 ^* , _VV2 ^* , _VW2 ^* ) and the custom two-phase modulation voltage commands ( _VU3 ^* , _VV3 ^* , _VW3 ^* ) calculated as described above will be output as the final voltage command ( _VUf ^* , _VVf ^* , _VWf ^* ) as follows.

First, the final voltage command calculation unit 44 determines the need to temporarily switch from the default two-phase modulation control to the custom two-phase modulation control by comparing the fluctuation frequency f _vn of the neutral point potential with the cutoff frequency f _τ of the low-pass filter 36 .

When the fluctuation frequency _fvn of the neutral point potential is close to or lower than the cutoff frequency _fτ of the low-pass filter 36, if the default two-phase modulation control is continued as is, all or part of the noise generated by the fluctuation of the neutral point potential passes through the low-pass filter 36. For this reason, the final voltage command calculation unit 44 determines that it is necessary to temporarily switch to custom two-phase modulation control. If the default two-phase modulation control is temporarily switched to custom two-phase modulation control, the neutral point potential is almost inverted and the fluctuation frequency _fvn becomes higher frequency. As a result, the noise generated by the fluctuation of the neutral point potential is less likely to pass through the low-pass filter 36.

However, when the fixed arm naturally changes in response to the change in the three-phase AC voltage command (V _U1 ^* , V _V1 ^* , V _W1 ^* ) and the phase to which the potential is fixed or the sign of the fixed potential changes, the neutral point potential is almost inverted even if the default two-phase modulation control is continued. That is, this is the timing when it is not necessary to switch to the custom two-phase modulation control. For this reason, the final voltage command calculation unit 44 compares the previous value and the current value of the default two-phase modulation voltage command (V _U2 ^* , V _V2 ^* , V _W2 ^* ), and when the phase with the maximum absolute ^value changes or the sign of the phase changes, ^it exceptionally determines that it is not necessary to switch to the custom two-phase modulation control. That is, when the phase with the maximum absolute value does ^not change and the sign of the _phase does not change in the _{comparison between} the previous value and the current value of the default two-phase modulation voltage command (V U2 *, V V2 *, V W2 *), and the fixed arm is maintained, the determination that it is necessary to switch to the custom two-phase modulation control is maintained.

In this embodiment, the neutral point potential fluctuation frequency f _vn is compared with the cutoff frequency f _τ of the low-pass filter 36 by determining whether or not the absolute value of the difference between them is smaller than a predetermined proximity determination threshold value β (|f _vn -f _τ |<β). This is to determine whether or not the neutral point potential fluctuation frequency f _vn is in the vicinity of the cutoff frequency f _τ of the low-pass filter 36, and is to essentially determine whether or not the neutral point potential fluctuation frequency f _vn and the low-pass filter cutoff frequency f _τ match. Since the neutral point potential fluctuation frequency f _vn is usually larger than the cutoff frequency f _τ of the low-pass filter 36, as described above, by determining whether or not the neutral point potential fluctuation frequency f _vn and the low-pass filter 36 _match (substantially match), it is possible to determine whether or not the default two-phase modulation control should be interrupted by the custom two-phase modulation control.

However, the final voltage command calculation unit 44 may determine whether or not to insert custom two-phase modulation control into the default two-phase modulation control depending on whether the fluctuation frequency f _vn of the neutral point potential is close to or lower than the cutoff frequency f _τ of the low-pass filter 36 (f _vn - _{f τ} < β).

Since the proximity determination threshold value β is set to a substantially small value, it can be said that the final voltage command calculation unit 44 essentially determines that the fluctuation frequency f _vn of the neutral point potential is equal to or lower than the cutoff frequency f _τ of the low-pass filter 36 (f _vn - _{f τ} < 0). In particular, in this embodiment, it can be said that the final voltage command calculation unit 44 simply determines that the fluctuation frequency f _vn of the neutral point potential matches the cutoff frequency f _τ of the low-pass filter 36 (f _vn - f _τ = 0).

Furthermore, the final voltage command calculation unit 44 determines the necessity of temporarily switching from the default two-phase modulation control to the custom two-phase modulation control based on the time during which the same arm is actually fixed on by the default two-phase modulation control (hereinafter referred to as the actual duration _τ1 of the default two-phase modulation control). In this embodiment, the final voltage command calculation unit 44 measures the actual duration _τ1 of the default two-phase modulation control by a first timer 45. Then, the final voltage command calculation unit 44 determines that it is necessary to temporarily switch from the default two-phase modulation control to the custom two-phase modulation control when the actual duration _τ1 of the default two-phase modulation control exceeds a maximum duration _T1 , which is a predetermined time.

The final voltage command calculation unit 44 also determines the timing of returning to the default two-phase modulation control based on the time during which the same arm is actually fixed on by the custom two-phase modulation control (hereinafter referred to as the actual duration τ ₂ of the custom two-phase modulation control). In this embodiment, the final voltage command calculation unit 44 measures the actual duration τ ₂ of the custom two-phase modulation control by the second timer 46. Then, when the actual duration τ ₂ of the custom two-phase modulation control is equal to or less than the maximum duration T ₂ , which is a predetermined time, the final voltage command calculation unit 44 starts or continues the custom two-phase modulation control. On the other hand, when the actual duration τ ₂ of the custom two-phase modulation control exceeds the maximum duration T ₂ , the final voltage command calculation unit 44 switches to the default two-phase modulation control and clears (resets) the first timer 45 and the second timer 46.

As described above, the first timer 45 measures the actual duration _τ1 of the default two-phase modulation control. Therefore, the first timer 45 is set when the default two-phase modulation control is started, and is cleared when the temporarily executed custom two-phase modulation control ends and the default two-phase modulation control is started again. In addition, when the fluctuation frequency _fvn of the neutral point potential is higher than the cutoff frequency _fτ of the low-pass filter 36 and the default two-phase modulation control is not executed, the first timer 45 is maintained in a cleared state.

As described above, the second timer 46 measures the actual duration _τ2 of the custom two-phase modulation control. Therefore, the second timer 46 is set when the custom two-phase modulation control is started, and is cleared when the default two-phase modulation control is restored. In addition, when the fluctuation frequency _fvn of the neutral point potential is greater than the cutoff frequency _fτ of the low-pass filter 36 and the default two-phase modulation control is not executed, the second timer 46 is maintained in a cleared state.

The following describes the operation of switching the modulation method in the electric vehicle 100 configured as described above.

4 is a flow chart relating to the switching of the modulation method. As shown in FIG. 4, in step S110, the two-phase modulation execution determination unit 41 compares the power element temperature T _pm with the first temperature threshold value α _1. When the power element temperature T _pm is equal to or lower than the first temperature threshold value α ₁ and the inverter 16 is within the steady temperature range, the three-phase modulation control can be executed (continued). Therefore, the process proceeds to step S111. In step S111, the carrier frequency setting unit 42 sets the carrier frequency f _c to the normal carrier frequency f _c0 . In step S112, the first timer 45 and the second timer 46 (i.e., the actual durations τ ₁ and τ ₂ ) are cleared. Then, in step S113, final voltage command calculation unit 44 outputs the three-phase AC voltage commands ( _VU1 ^* , _VV1 ^* , _VW1 ^* ) as final voltage commands ( _VUf ^* , _VVf ^* , _VWf ^* ) without modification, and executes three-phase modulation control.

On the other hand, in step S110, when the power element temperature T _pm is greater than the first temperature threshold α ₁ and the inverter 16 is in a high temperature state, the process proceeds to step S114, where the carrier frequency setting unit 42 compares the power element temperature T _pm with the second temperature threshold α _2. Then, when the power element temperature T _pm is equal to or less than the second temperature threshold α ₂ and it is determined that the inverter 16 has not yet reached a high temperature state where the carrier frequency f _c must be reduced, the process proceeds to step S115, where the carrier frequency setting unit 42 sets (maintains) the carrier frequency f _c to the normal carrier frequency f _c0 . On the other hand, when the power element temperature T _pm is greater than the second temperature threshold α ₂ and it is determined that the inverter 16 has reached an especially high temperature state, the process proceeds to step S116, where the carrier frequency setting unit 42 determines the low carrier frequency f _c1 by calculation or the like and sets the carrier frequency f _c to the low carrier frequency f _c1 .

Then, in step S117, the final voltage command calculation unit 44 outputs the default two-phase modulation voltage commands ( _VU2 ^* , _VV2 ^* , _VW2 ^* ) as final voltage commands ( _VUf ^* , _VVf ^* , _VWf ^* ) to execute default two-phase modulation control. Also, in step S118, the neutral point potential fluctuation frequency calculation unit 43 calculates the fluctuation frequency _fvn of the neutral point potential based on the electrical angular velocity ω of the rotating electric machine 11.

In step S119, the final voltage command calculation unit 44 compares the fluctuation frequency f _vn of the neutral point potential with the cutoff frequency f _τ of the low-pass filter 36. When the fluctuation frequency f _vn of the neutral point potential is substantially greater than the cutoff frequency f _τ of the low-pass filter 36, the first timer 45 and the second timer 46 are cleared in step S120, and the default two-phase modulation control is continued. On the other hand, when the fluctuation frequency f _vn of the neutral point potential has decreased to a level substantially matching the cutoff frequency f _τ of the low-pass filter 36 in response to the electrical angular velocity ω of the rotating electric machine 11, the process proceeds to step S121.

In step S121, the final voltage command calculation unit 44 compares the previous and current values of the default two-phase modulation voltage commands ( _VU2 ^* , _VV2 ^* , _VW2 ^* ) to determine whether the fixed arms are maintained. Then, when the fixed arms naturally change in response to changes in the three-phase AC voltage commands ( _VU1 ^* , _VV1 ^* , _VW1 ^* ), the process proceeds to step S120, where the first timer 45 and the second timer 46 are cleared and the default two-phase modulation control is continued.

On the other hand, when the fixed arm is maintained, the process proceeds to step S122, where the final voltage command calculation unit 44 compares the actual duration _τ1 of the default two-phase modulation control with its maximum duration _T1 . Then, when the actual duration _τ1 of the default two-phase modulation control is equal to or shorter than the maximum duration _T1 , the default two-phase modulation control is continued as is. On the other hand, when the actual duration _τ1 of the default two-phase modulation control exceeds the maximum duration _T1 , the process proceeds to step S123, where the final voltage command calculation unit 44 compares the actual duration _τ2 of the custom two-phase modulation control with its maximum duration _T2 .

When the actual duration _τ2 of the customized two-phase modulation control is equal to or shorter than the maximum duration _T2 , the process proceeds to step S124. In step S124, the final voltage command calculation unit 44 executes the customized two-phase modulation control by outputting the customized two-phase modulation voltage commands ( _VU3 ^* , _VV3 ^* , _VW3 ^* ) as the final voltage commands ( _VUf ^* , _VVf ^* , _VWf ^* ). When the actual duration _τ2 of the customized two-phase modulation control exceeds the maximum duration _T2 in step S123, the process proceeds to step 120, the first timer 45 and the second timer 46 are cleared, and the default two-phase modulation control is continued or is restored to the default two-phase modulation control.

FIG. 5 is a graph showing the default two-phase modulation control. In each graph in FIG. 5, the horizontal axis is time. FIG. 5(A) shows examples of three-phase AC voltage commands V _U1 ^* , V _V1 ^* , and V _W1 ^* with a solid line, a dashed line, and a dashed line, respectively. FIG. 5(B) to FIG. 5(D) show the default two-phase modulation voltage commands V _U2 ^* , V _V2 ^* , and V _W2 ^* with a solid line, a dashed line, and a dashed line, respectively, corresponding to the three-phase AC voltage commands V _U1 ^* , V _V1 ^* , and V _W1 ^* . FIG. 5(E) shows the fluctuation of the neutral point potential by the default two-phase modulation control and its fluctuation frequency f _vn (fluctuation period 1/f _vn ). It should be noted that periods P ₁ to P ₈ indicate a time range in which the phase that becomes the maximum command V _p ^* or the minimum command V _n ^* among the three-phase AC voltage commands V _U1 ^* , V _V1 ^* , and V _W1 ^* does not change.

As shown in Fig. 5A, the three-phase AC voltage commands _VU1 ^* , _VV1 ^* , and _VW1 ^* are sine waves with mutually shifted phases. When default two-phase modulation control is performed under these three-phase AC voltage commands, the default two-phase modulation voltage commands _VU2 ^* , _VV2 ^* , and _VW2 ^* change as shown in Fig. 5B to Fig. 5D.

For example, in period _P2 , | _Vp ^* |≧| _Vn ^* |, and among the three-phase AC voltage commands _VU1 ^* , _VV1 ^* , and _VW1 ^* , the U-phase voltage command _VU1 ^* is the maximum command _Vp ^* . Therefore, as shown in Fig. 5B, among the default two-phase modulation voltage commands _VU2 ^* , _VV2 ^* , and _VW2 ^* , the U-phase voltage command _VU2 ^* is fixed to + _Vb /2, which is the maximum positive value, and the V-phase and W-phase voltage commands _VV2 ^* and _VW2 ^* are modulated as shown in Fig. 5C and Fig. 5D.

Also, for example, in period _P3 , | _Vp ^* |<| _Vn ^* |, and among the three-phase AC voltage commands _VU1 ^* , _VV1 ^* , and _VW1 ^* , the W-phase voltage command _VW1 ^* becomes the minimum command _Vn ^* . Therefore, as shown in Fig. 5D, among the default two-phase modulation voltage commands _VU2 ^* , _VV2 ^* , and _VW2 ^* , the W-phase voltage command _VW2 ^* is fixed to the negative maximum value _-Vb /2, and the U-phase and V-phase voltage commands _VU2 ^* and _VV2 ^* are modulated as shown in Fig. 5B and Fig. 5C.

The same is true for period _P1 and periods _P4 to _P8 . That is, in periods _P2 , _P4 , _P6 , and _P8 , | _Vp ^* |≧| _Vn ^* |, and the potentials of the U-phase, V-phase, W-phase, and U-phase are fixed to the maximum positive value. Also, in periods _P1 , _P3 , _P5 , and _P7 , | _Vp ^* |<| _Vn ^* |, and the potentials of the V-phase, W-phase, U-phase, and V-phase are fixed to the maximum negative value.

In the default two-phase modulation control, as described above, the potential of one of the phases fluctuates periodically while being fixed at the positive or negative maximum value, and the neutral point potential also fluctuates periodically accordingly, as shown in Fig. 5(E) In this case, for example, when the electrical angular velocity ω is small and the rotating electric machine 11 is rotating at a low speed, the fluctuating frequency _fvn of the neutral point potential approaches the cutoff frequency _fτ of the low-pass filter 36, which may cause the insulation resistance sensor 21 to make an erroneous detection.

FIG. 6 is a graph showing a case where custom two-phase modulation control is inserted. In each graph of FIG. 6, the horizontal axis is time. FIG. 6(A) shows examples of three-phase AC voltage commands V _U1 ^* , V _V1 ^* , and V _W1 ^* , respectively, with a solid line, a dashed line, and a dashed line. FIG. 6(B) to FIG. 6(D) show final voltage commands V _Uf ^* , V _Vf ^* , and V _Wf ^* , respectively, corresponding to the three-phase AC voltage commands V _U1 ^* , V _V1 ^* , and V _W1 ^* , when custom two-phase modulation control is inserted into the default two-phase modulation control, with a solid line, a dashed line, and a dashed line. FIG. 5(E) shows the fluctuation of the neutral point potential and its fluctuation frequency f _vn (fluctuation period 1/f _vn ). Note that periods P ₁ to P ₈ are the same as in FIG. 5, and indicate a time range in which the phase that becomes the maximum command V _p ^* or the minimum command V _n ^* among the three-phase AC voltage commands V _U1 ^* , V _V1 ^* , V _W1 ^* does not change.

6A, for example, in period _P2 , | _Vp ^* |≧| _Vn ^* | and the U-phase voltage command _VU1 ^* becomes the maximum command _Vp ^* , so if the default two-phase modulation control is continued, this is the period during which the U-phase voltage command _VU2 ^* is maintained at the positive maximum value + _Vb /2. In contrast, when switching between default two-phase modulation control and custom two-phase modulation control is performed by the method of this embodiment, the final voltage commands _VUf ^* , _VVf ^* , and _VWf ^* vary as follows:

For example, in the first quarter _P2a of the period _P2 , the default two-phase modulation voltage commands _VU2 ^* , _VV2 ^* , and _VW2 ^* are output _as the final voltage commands _VUf ^* , _VVf ^* , and _VWf ^* . That is, in the first quarter _P2a , the inverter 16 is driven by the default two-phase modulation control. Therefore, as shown in FIG. 6B, the potential of the U phase is fixed to the positive maximum value, and as shown in FIG. 6C and FIG. 6D, the potentials of the V phase and the W phase are modulated.

Next, in the first quarter P _2a , the actual duration τ ₁ of the default two-phase modulation control exceeds the maximum duration T _1. Therefore, in the second quarter P _2b , the voltage commands V _U3 ^* , V _V3 ^* , and V _W3 ^* for custom two-phase modulation are output as the final voltage commands V _Uf ^* , V _Vf ^* , and V _Wf ^* . That is, in the second quarter P _2b , the inverter 16 is driven by the custom two-phase modulation control. At this time, as shown in FIG. 6(A), the V phase is the minimum command V _n ^* among the three-phase AC voltage commands V _U1 ^* , V _V1 ^* , and V _W1 ^* . Therefore, as shown in FIG. 6(C), in the second quarter P _2b , the potential of the V phase is fixed to the negative maximum value, and the potentials of the U phase and the W phase are modulated as shown in FIG. 6(B) and FIG. 6(D). As a result, in the second quarter _P2b , the final voltage commands _VUf ^* , _VVf ^* , and _VWf ^* are substantially reversed with respect to the first quarter _P2a .

Then, in the second quarter P _2b , the actual duration τ ₂ of the custom two-phase modulation control exceeds the maximum duration T _2. Therefore, in the third quarter P _2c , the default two-phase modulation voltage commands V _U2 ^* , V _V2 ^* , and V _W2 ^* are output again _as the final voltage commands V _Uf ^* , V _Vf ^* , and V _Wf ^* . That is, in the third quarter P _2c , the inverter 16 is driven by the default two-phase modulation control. Therefore, as shown in FIG. 6(B), the potential of the U phase is fixed to the positive maximum value, and as shown in FIG. 6(C) and FIG. 6(D), the potentials of the V phase and the W phase are modulated. As a result, in the third quarter P _2c , the final voltage commands V _Uf ^* , V _Vf ^* , and V _Wf ^* are roughly inverted with respect to the second quarter P _2b .

After that, in the third quarter P _2c , the actual duration τ ₁ of the default two-phase modulation control exceeds the maximum duration T _1. Therefore, in the fourth quarter P _2d , the voltage commands V _U3 ^* , V _V3 ^* , and V _W3 ^* for custom two-phase modulation are output again as the final voltage commands V _Uf ^* , V _Vf ^* , and V _Wf ^* . That is, in the fourth quarter P _2d , the inverter 16 is driven by the custom two-phase modulation control. At this time, as shown in FIG. 6(A), the W phase is the minimum command V _n ^* among the three-phase AC voltage commands V _U1 ^* , V _V1 ^* , and V _W1 ^* . Therefore, as shown in FIG. 6(D), in the fourth quarter P _2d , the potential of the W phase is fixed to the negative maximum value, and the potentials of the U phase and the V phase are modulated as shown in FIG. 6(B) and FIG. 6(C). As a result, in the fourth quarter _P2d , the final voltage commands _VUf ^* , _VVf ^* , and _VWf ^* are substantially reversed with respect to the third quarter _P2c .

As described above, when the default two-phase modulation control and the custom two-phase modulation control are switched during the two-phase modulation control, the neutral point potential also fluctuates accordingly, as shown in FIG. 6E. Therefore, the fluctuation frequency f _vn of the neutral point potential becomes higher than when the default two-phase modulation control is continued (see FIG. 5E). Therefore, even if the fluctuation frequency f _vn of the neutral point potential becomes equal to or lower than the cutoff frequency f _τ of the low-pass filter 36 when the default two-phase modulation control is simply continued, the fluctuation frequency f _vn of the neutral point potential is kept higher than the cutoff frequency f _τ of the low-pass filter 36 by switching between the default two-phase modulation control and the custom two-phase modulation control. As a result, even if noise acts on the insulation resistance sensor 21 through the ground due to the fluctuation of the neutral point potential, this noise is appropriately reduced by the low-pass filter 36, and therefore the insulation resistance sensor 21 does not cause erroneous detection.

Figure 7 is a graph showing the spectrum of noise propagating to the ground when the neutral point potential fluctuates due to two-phase modulation control. Figure 7(A) shows the case where the default two-phase modulation control is continued, and Figure 7(B) shows the case where the default two-phase modulation control and the custom two-phase modulation control are switched.

As shown in Fig. 7A, when performing two-phase modulation control, if default two-phase modulation control is simply continued, the frequency may become the cutoff frequency f _τ of the low-pass filter 36. However, as shown in Fig. 7B, when switching between default two-phase modulation control and custom two-phase modulation control, the fluctuation frequency f _vn of the neutral point potential becomes higher, causing a change in the noise spectrum. As a result, the components of the cutoff frequency f _τ are reduced. Here, the noise components of the cutoff frequency f _τ are reduced to the extent shown by the difference Δ.

Therefore, as in this embodiment, when performing two-phase modulation control, switching between default two-phase modulation control and custom two-phase modulation control can suppress erroneous detection by the insulation resistance sensor 21. Furthermore, two-phase modulation control is performed to suppress heat generation by the inverter 16, but both default two-phase modulation control and custom two-phase modulation control are two-phase modulation controls, and even when switching between them, the switching frequency of the power elements remains reduced compared to three-phase modulation control. Therefore, the switching control between default two-phase modulation control and custom two-phase modulation control in this embodiment satisfies both of these requirements, suppressing heat generation by the inverter 16 while suppressing erroneous detection by the insulation resistance sensor 21. In other words, it is easy to achieve both of these contradictory requirements.

Furthermore, in this embodiment, when it is determined that the temperature of the inverter 16 has reached an especially high temperature, in addition to switching between default two-phase modulation control and custom two-phase modulation control, the carrier frequency f _c is set to a low carrier frequency f _c1 , and furthermore, heat generation of the inverter 16 and sound, vibration, and the like (hereinafter referred to as sound vibration, etc.) generated by the operation of the inverter 16 are suppressed. In this embodiment, the low carrier frequency f _c1 is set according to the switching ratio κ between default two-phase modulation control and custom two-phase modulation control, so that in particular heat generation, sound vibration, and the like of the inverter 16 are suppressed while erroneous detection by the insulation resistance sensor 21 is easily suppressed.

[Modification]
In the above embodiment, two-phase modulation control is taken as an example of PWM control for varying the neutral point potential, but this is not limiting. The neutral point potential also varies when square wave control is performed to control the power input to the rotating electric machine 11 to a square wave, or trapezoidal wave control is performed to control the power input to the rotating electric machine 11 to a trapezoidal waveform by introducing a third harmonic. For this reason, it is preferable to perform the control of the above embodiment even when square wave control or trapezoidal wave control is performed.

In the above embodiment, the insulation resistance sensor 21 is given as an example of a detector (sensor, etc.) that may cause erroneous detection due to fluctuations in the neutral point potential, but this is not limiting. The control of the above embodiment uses the ground as a reference potential and can suppress erroneous detection due to fluctuations in the neutral point potential for other detectors that include a noise reduction filter. Furthermore, when there are multiple detectors that may cause erroneous detection due to fluctuations in the neutral point potential, the control of the above embodiment can be executed in accordance with, for example, the highest cutoff frequency f _τ among the cutoff frequencies f _τ of the noise reduction filters of the respective detectors.

In the above embodiment, the default two-phase modulation control and the custom two-phase modulation control are switched to directly reduce the noise component of the cutoff frequency _fτ , but this is not limiting. For example, the default two-phase modulation control and the custom two-phase modulation control may be switched to suppress noise (fluctuations in the neutral point potential) having a frequency corresponding to the electrical angular velocity ω of the rotating electric machine 11 and/or a multiple of that frequency. In this case, too, it is possible to reduce erroneous detection by the insulation resistance sensor 21, etc., as in the above embodiment.

In the above embodiment, switching between default two-phase modulation control and custom two-phase modulation control is performed, but this is not limited to this. For example, when performing two-phase modulation control, switching between default two-phase modulation control and three-phase modulation control may be performed.

FIG. 8 is a flowchart of a modified example related to switching of the modulation method. Here, an example of switching between the default two-phase modulation control and the three-phase modulation control as described above is shown. As shown in FIG. 8, the three-phase modulation control of step S130 is executed at the timing of executing the custom two-phase modulation control in step S124 of FIG. 4. Since the neutral point potential is small on average in the sinusoidal three-phase modulation control, the fluctuating frequency f _vn of the neutral point potential can be made high frequency, similar to the above embodiment, even when the three-phase modulation control is executed instead of the custom two-phase modulation control. Therefore, the erroneous detection of the insulation resistance sensor 21 can be suppressed, similar to the above embodiment. Furthermore, at least during the period in which the default two-phase modulation control is performed, the switching frequency of the power element is reduced. Therefore, the heat generation and sound vibration of the inverter 16 are suppressed, and the erroneous detection of the insulation resistance sensor 21 is suppressed.

In the above embodiment, the power element temperature T _pm is referred to as the temperature of the inverter 16 to set the carrier frequency f _c , but this is not limiting. For example, the carrier frequency f _c may be set more appropriately by further taking into account the heat dissipation property of the inverter 16.

FIG. 9 is a flowchart of a modified example related to switching of the modulation method. Here, as described above, an example is shown in which the carrier frequency f _c is set by further considering the heat dissipation of the inverter 16 using the refrigerant temperature T _wtr . In this case, for example, as shown in FIG. 9, step S140 can be provided between step S114 and step S115 in FIG. 4. In step S140, the refrigerant temperature T _wtr is compared with a third temperature threshold value γ that is determined in advance by an experiment, a simulation, or the like. Then, when the refrigerant temperature T _wtr is equal to or lower than the third temperature threshold value γ and the refrigerant has sufficient heat dissipation, the process proceeds to step S115, and the carrier frequency setting unit 42 sets (maintains) the carrier frequency f _c to the normal carrier frequency f _c0 . On the other hand, when the refrigerant temperature T _wtr is higher than the third temperature threshold value γ and it is determined that the refrigerant does not have sufficient heat dissipation, the process proceeds to step S116, and the carrier frequency setting unit 42 determines the low carrier frequency f _c1 by calculation, or the like, and sets the carrier frequency f _c to the low carrier frequency f _c1 . In this way, when the carrier frequency f _c is set taking into consideration the heat dissipation property of the inverter 16, it is particularly easy to suppress the heat generation and noise vibration of the inverter 16 while also suppressing erroneous detection by the insulation resistance sensor 21.

In the modified example of Fig. 8 as well, as described above, the carrier frequency f _c can be set more appropriately by taking into consideration the heat dissipation property of the inverter 16 according to the refrigerant temperature T _wtr . Fig. 10 is a flowchart of a modified example relating to switching of the modulation method. Here, step S130 is provided instead of step S124 in Fig. 4, and step S140 is further provided between steps S114 and S115. In this case as well, it is particularly easy to suppress heat generation and noise vibration of the inverter 16 while suppressing erroneous detection of the insulation resistance sensor 21.

As described above, the inverter control method according to the above-mentioned embodiments is an inverter control method for controlling the inverter 16 in an electric circuit 101 including a detector (insulation resistance sensor 21) including a noise reduction filter (low-pass filter 36) with the ground (GND) as a reference potential, an inverter 16 that converts DC power to AC power by pulse width modulation control (PWM control), and a rotating electric machine 11 connected to the inverter 16. In this inverter control method, a fluctuation frequency f _vn of the neutral point potential of the rotating electric machine 11 is compared with a cutoff frequency f _τ of the noise reduction filter. Then, when the fluctuation frequency f _vn is equal to or lower than the cutoff frequency f _τ , the modulation method of the pulse width modulation control is changed from a first modulation method (default two-phase modulation method) selected according to the temperature (power element temperature T _pm ) of the inverter 16 to a second modulation method (custom two-phase modulation method) different from the first modulation method for a predetermined time (maximum duration T ₂ ) that is set in advance.

In this way, when the modulation method of the inverter 16 is switched, the fluctuation frequency _fvn of the neutral point potential becomes high and is kept higher than the cutoff frequency _fτ , so that it is possible to suppress erroneous detection of a detector including a noise reduction filter by using the ground as a reference potential. In addition, since control is performed using the first modulation method (default two-phase modulation control) selected according to at least the temperature of the inverter 16, heat generation of the inverter 16 is also suppressed. In other words, heat generation of the inverter 16 is suppressed while erroneous detection of the insulation resistance sensor 21 is suppressed.

In the inverter control method according to the above embodiment, the rotating electric machine 11 is a three-phase synchronous rotating electric machine, and the first modulation method is a default two-phase modulation method that controls the output power of the inverter 16 by fixing the potential of one phase of the rotating electric machine 11 that is the maximum or minimum, and modulating the other two phases by pulse width modulation control.

In the two-phase modulation method, the switching of the inverter 16 is suppressed compared to the three-phase modulation method. This suppresses heat generation in the inverter 16. Therefore, if the first modulation method is the two-phase modulation method (default two-phase modulation method), heat generation in the inverter 16 is suppressed while erroneous detection by the insulation resistance sensor 21 is suppressed.

In the inverter control method according to the above embodiment, the second modulation method is a custom two-phase modulation method that fixes the potential of a phase different from the phase whose potential is fixed in the first modulation method.

In this way, if the temporarily switched modulation method is a custom modulation method, the heat generation of the inverter 16 is suppressed even in the temporarily switched modulation method. Therefore, if the second modulation method is a custom two-phase modulation method, in particular, the heat generation of the inverter 16 is suppressed while erroneous detection of the insulation resistance sensor 21 is suppressed.

In the inverter control method according to the above embodiment, the second modulation method can also be a three-phase modulation method in which the potentials of the three phases in the rotating electric machine 11 are modulated by pulse width modulation control.

In this way, even when the second modulation method is the three-phase modulation method, the neutral point potential fluctuation frequency _fvn is increased to a high frequency and maintained higher than the cutoff frequency _fτ , so that the ground is used as the reference potential, and erroneous detection of a detector including a noise reduction filter can be suppressed. Furthermore, since control is performed using the first modulation method (default two-phase modulation control) selected according to at least the temperature of the inverter 16, in addition to suppressing heat generation of the inverter 16 by the two-phase modulation method, sound vibration suppression when the three-phase modulation method is applied can also be expected. In other words, heat generation and sound vibration of the inverter 16 are suppressed, and erroneous detection of the insulation resistance sensor 21 is suppressed.

In the inverter control method according to the above embodiment, the carrier frequency f _c , which is the frequency of the carrier signal used in the pulse width modulation control, is changeable, and when the fluctuation frequency f _vn is equal to or lower than the cutoff frequency f _τ , the modulation method of the pulse width modulation control is changed and the carrier frequency f _c is reduced.

Reducing the carrier frequency of the PWM control reduces the number of switching operations of the inverter 16. As a result, heat generation in the inverter 16 is reduced. Therefore, by further reducing the carrier frequency f _c as described above, heat generation in the inverter 16 is particularly reduced, and erroneous detection by the insulation resistance sensor 21 is also suppressed.

In the inverter control method according to the above embodiment, the carrier frequency f _c is set based on the switching ratio κ between the first modulation method (default two-phase modulation) and the second modulation method (custom two-phase modulation).

In this way, by setting the carrier frequency f _c based on the switching ratio κ between the first modulation method (default two-phase modulation) and the second modulation method (custom two-phase modulation), the carrier frequency f _c can be appropriately set according to the temperature status of the inverter 16, etc. This makes it easy to suppress heat generation in the inverter 16 and to suppress erroneous detection by the insulation resistance sensor 21. Furthermore, compared to the case where the carrier frequency f _c is fixed, changing the carrier frequency f _c based on the switching ratio κ as described above may suppress sound vibration of the inverter 16.

In the inverter control method according to the above embodiment, the reduced carrier frequency (low carrier frequency f _c1 ) is set to be equal to or lower than half of the original carrier frequency (normal carrier frequency f _c0 ).

In this way, by setting the reduced carrier frequency (f _c1 ) to half or less of the original carrier frequency (f _c0 ), heat generation in the inverter 16 can be suppressed to the maximum extent in a situation where the carrier frequency f _c should be reduced. In other words, heat generation in the inverter 16 can be easily and reliably suppressed. Therefore, it is easy to suppress erroneous detection by the insulation resistance sensor 21 while suppressing heat generation, noise and vibration, etc., of the inverter 16 in particular.

In the inverter control method according to the above embodiments, the temperature of the inverter 16 (power element temperature T _pm ) is acquired, and the coolant temperature T _wtr , which is the temperature of the coolant that cools the inverter 16, is acquired. Then, at least when the temperature of the inverter 16 exceeds a predetermined threshold (first temperature threshold α ₁ ) and the coolant temperature T _wtr exceeds a predetermined threshold (third temperature threshold γ), the modulation method is changed to the second modulation method (custom two-phase modulation) and the carrier frequency f _c is reduced.

In this way, by setting the carrier frequency f _c in consideration of the heat dissipation property of the inverter 16 according to the refrigerant temperature T _wtr , the carrier frequency f _c can be set more appropriately. Therefore, in particular, heat generation of the inverter 16 is suppressed, and erroneous detection of the insulation resistance sensor 21 is easily suppressed. Furthermore, compared to the case where the carrier frequency f _c is fixed, changing the carrier frequency f _c as described above may suppress sound vibration of the inverter 16.

The inverter control device according to the above embodiment is an inverter control device that controls the inverter 16 in an electric circuit 101 including a detector (insulation resistance sensor 21) including a noise reduction filter (low-pass filter 36) with a ground (GND) as a reference potential, an inverter 16 that converts DC power to AC power by pulse width modulation control (PWM control), and a rotating electric machine 11 connected to the inverter 16. This inverter control device compares a fluctuation frequency f _vn of the neutral point potential of the rotating electric machine 11 with a cutoff frequency f _τ of the noise reduction filter. Then, when the fluctuation frequency f _vn is equal to or lower than the cutoff frequency f _τ , the modulation method of the pulse width modulation control is changed from a first modulation method (default two-phase modulation method) selected according to the temperature of the inverter 16 (power element temperature T _pm ) to a second modulation method (custom two-phase modulation method) different from the first modulation method for a predetermined time (maximum duration T ₂ ) that is set in advance.

In this way, when the modulation method of the inverter 16 is switched, the fluctuation frequency _fvn of the neutral point potential becomes high and is kept higher than the cutoff frequency _fτ , so that it is possible to suppress erroneous detection of a detector including a noise reduction filter by using the ground as a reference potential. In addition, since control is performed using the first modulation method (default two-phase modulation control) selected according to at least the temperature of the inverter 16, heat generation of the inverter 16 is also suppressed. In other words, heat generation of the inverter 16 is suppressed while erroneous detection of the insulation resistance sensor 21 is suppressed.

Although the embodiments of the present invention have been described above, the configurations described in the above embodiments and each modified example merely show some of the application examples of the present invention and are not intended to limit the technical scope of the present invention. For example, the above embodiments and modified examples can be implemented by combining all or some of the elements in any desired manner.

11: rotating electric machine, 12: current command calculation unit, 13: current control unit, 14: coordinate conversion unit, 15: modulation method setting unit, 16: inverter, 17: battery, 18: rotation state detection unit, 19: coordinate conversion unit, 20: rotation detector, 21: insulation resistance sensor, 22: smoothing capacitor, 22u: stator coil, 22v: stator coil, 22w: stator coil, 23: neutral point, 31: capacitor, 32: resistor, 33: pulse oscillator, 34: resistor, 35: capacitor, 36: low-pass filter, 37: comparator, 38: reference voltage, 41: two-phase modulation execution determination unit, 42: carrier frequency setting unit, 43: neutral point potential fluctuation frequency calculation unit, 44: final voltage command calculation unit, 45: first timer, 46: second timer, 100: electric vehicle, 101: electric circuit

Claims (9)

An inverter control method for controlling an inverter in an electric circuit including a detector having a ground as a reference potential and including a noise reduction filter, an inverter that converts DC power into AC power by pulse width modulation control, and a rotating electric machine connected to the inverter, comprising:
comparing a fluctuation frequency of a neutral point potential of the rotating electric machine with a cutoff frequency of the noise reduction filter;
when the fluctuation frequency is equal to or lower than the cutoff frequency, a modulation method of the pulse width modulation control is changed from a first modulation method selected in accordance with a temperature of the inverter to a second modulation method different from the first modulation method for a predetermined time period;
Inverter control method. 2. The inverter control method according to claim 1,
the rotating electric machine is a three-phase synchronous rotating electric machine,
The first modulation method is a default two-phase modulation method in which a potential of one phase, which is maximum or minimum, among the phases of the rotating electric machine is fixed, and the other two phases are modulated by the pulse width modulation control, thereby controlling the output power of the inverter.
Inverter control method. 3. The inverter control method according to claim 2,
the second modulation method is a custom two-phase modulation method in which a potential of a phase different from a phase in which a potential is fixed in the first modulation method is fixed.
Inverter control method. 3. The inverter control method according to claim 2,
The second modulation method is a three-phase modulation method in which each potential of three phases in the rotating electric machine is modulated by the pulse width modulation control.
Inverter control method. The inverter control method according to any one of claims 1 to 4,
A carrier frequency, which is a frequency of a carrier signal used in the pulse width modulation control, is variable,
When the fluctuating frequency is equal to or lower than the cutoff frequency, a modulation method of the pulse width modulation control is changed and the carrier frequency is reduced.
Inverter control method. 6. An inverter control method according to claim 5, comprising:
setting the carrier frequency based on a switching ratio between the first modulation method and the second modulation method;
Inverter control method. 6. An inverter control method according to claim 5, comprising:
The reduced carrier frequency is set to 1/2 or less of the original carrier frequency.
Inverter control method. 6. An inverter control method according to claim 5, comprising:
Acquire the temperature of the inverter;
A refrigerant temperature that is a temperature of a refrigerant that cools the inverter is acquired;
changing the modulation method to the second modulation method and reducing the carrier frequency at least when the temperature of the inverter exceeds a predetermined threshold and the coolant temperature exceeds a predetermined threshold;
Inverter control method. An inverter control device for controlling an inverter in an electric circuit including a detector having a ground as a reference potential and including a noise reduction filter, an inverter for converting DC power into AC power by pulse width modulation control, and a rotating electric machine connected to the inverter,
comparing a fluctuation frequency of a neutral point potential of the rotating electric machine with a cutoff frequency of the noise reduction filter;
when the fluctuation frequency is equal to or lower than the cutoff frequency, a modulation method of the pulse width modulation control is changed from a first modulation method selected in accordance with a temperature of the inverter to a second modulation method different from the first modulation method for a predetermined time period;
Inverter control device.

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