JP2024080883A - Induction Motor Drive - Google Patents

Abstract

[Problem] To accurately identify the electrical constants of an induction motor with the rotor stopped. [Solution] An induction motor drive device according to an embodiment includes a motor constant identification unit that identifies the electrical constants of an induction motor based on a detected value or voltage command value of a phase-to-phase voltage and a current command value or a detected current value. The electrical constants identified by the motor constant identification unit include identified values of a primary resistance, a leakage inductance, a no-load current, and a secondary time constant. A current command value generation unit generates the current command value of a preset magnitude when the rotor of the induction motor is stopped, and after a preset time has elapsed and the output of a power converter is cut off, the motor constant identification unit identifies the no-load current and the secondary time constant based on the detected value of the phase-to-phase voltage. [Selected Figure] Figure 1

Description

An embodiment of the present invention relates to an induction motor drive device.

Variable-speed motor drives that use inverters to drive electric motors at variable speeds are used in a variety of fields. Among electric motors, induction motors are often superior in terms of maintainability and robustness, and also have the advantage of being able to run on commercial power sources for backup purposes. For this reason, induction motors are still used in many applications today. In electric motor drives, high-performance operation can be achieved by detecting the rotation speed and implementing control that accurately grasps slippage, but sensorless vector control has been put into practical use as a method of high-performance driving without using a rotation sensor.

In sensorless vector control, the output voltage and current values are used to obtain electrical information about the motor, such as slippage, during operation, but the calculations required to do so require accurate knowledge of the motor's electrical constants. General-purpose inverters, which do not specify the induction motor used, have an auto-tuning function, or identification function, that automatically measures the electrical constants of the induction motor so that highly accurate sensorless vector control can be performed even when various induction motors are used.

The electrical constants required for sensorless vector control are expressed differently depending on the state variables. For example, in the model of an induction motor described in Non-Patent Document 1, when expressed on the synchronous DQ axes rotating at the primary angular frequency, the left side is expressed as in the following formula (1) with the primary voltage of the motor. However, each variable is expressed as follows. That is, the primary voltage D-axis component is v _sd , the primary voltage Q-axis component is v _sq , the primary current D-axis component is i _sd , the primary current Q-axis component is i _sq , the secondary magnetic flux D-axis component is φ _rd , and the secondary magnetic flux Q-axis component is φ _rq . In addition, the electrical constants are expressed as follows. That is, the primary resistance is R _s , the secondary resistance is R _r , the mutual inductance is M, the primary inductance is L _s , the secondary inductance is L _r , the leakage coefficient is σ, the primary angular frequency is ω _s , the rotor angular frequency is ω _re , and the slip angular frequency is ω _slip .

However, if this equation is used as is, the primary current and secondary current are state variables, and the concept of magnetic flux contributing to torque is difficult to intuitively understand, so a differential equation with the primary current and secondary magnetic flux as state variables is shown below as equation (2). Note that in this case, the approximation " _Lr = _Ls " is used.

When the differential term is placed on the left side of the equation, it becomes the following equation (3).

To calculate the voltage command value required in a steady state, the differential term is set to 0, resulting in the following equation (4).

If we rewrite each component with the left side being the primary voltage, we get the following equations (5) and (6).

If we define it as in equation (7) below, it can be expressed as in equation (8) below.

If the state variables are leakage flux and secondary magnetic flux _φrd , the leakage flux D-axis component _Fsd and the leakage flux Q-axis component _Fsq are steadily expressed by the following equations (9) and (10), respectively. The secondary magnetic flux D-axis component _φrd and the secondary magnetic flux are steadily expressed by the following equations (11) and (12), respectively. Taking these into consideration, equations (5) and (6) become the following equations (13) and (14), respectively.

It can be seen from this that the above-mentioned induction motor can be expressed mathematically if the four electrical constants " _Rs ", " _σLs ", "1/ _Tr (= _Rr / _Lr )" and " _Ls " are clearly defined. Therefore, in this specification, the induction motor model is expressed using four constants: primary resistance _Rs , leakage inductance _Lf ( _Lf = _σLs ), secondary time constant _Tr and primary inductance _Ls .

Specific means for acquiring the electrical constants of an induction motor are explained based on a steady-state equivalent circuit in, for example, JIS C 4210 "General-purpose low-voltage three-phase squirrel-cage induction motor" and JEC-2110 "Induction motor". Hereinafter, such specific means for acquiring the electrical constants of a conventional induction motor will be referred to as the conventional means. When the content of the conventional means is applied to the motor model shown in equation (3), it can be roughly interpreted as follows. In the motor model shown in equation (3), the iron loss resistance is omitted. Identification of the iron loss resistance will not be dealt with in this specification either.

Conventional methods describe a method of applying DC to identify the primary resistance. After DC is applied and a steady state is reached, no electromotive force is generated on the secondary side, i.e., the rotor, so the circuit can be considered to be only the primary resistance. Since no approximation elements are included, no errors due to approximation occur in this equivalent circuit. Figure 13 shows an equivalent circuit for one phase of an induction motor, and Figure 14 shows an equivalent circuit for one phase of an induction motor when DC is applied.

From this equivalent circuit, the following equation (15) is obtained. Note that all state variables are written as values for one phase.

Since the expression on the synchronous coordinate axes is when the motor is stopped, if the current flow is expressed as the D axis only, it becomes the following equation (16). For simplicity, in this explanation and similar explanations that follow, each state quantity is expressed as a state quantity when subjected to relative transformation. Therefore, the primary resistance value is expressed as the resistance value of one phase.

Because voltage is applied to the motor by the power conversion device, unless appropriate corrections are made to voltage values such as the dead time provided to prevent short circuits between the upper and lower arms, the on-resistance, which is the voltage drop in semiconductor devices such as IGBTs, and the collector-emitter saturation voltage, an error will occur between the voltage command value and the actual output voltage, making it impossible to accurately identify the primary resistance value of the motor.

Conventional techniques proposed for correcting values equivalent to these error voltages include a method of detecting the output voltage and feeding back the error voltage so that it matches the voltage command value, and a technique of determining the error voltage from the characteristics of the device measured in advance and compensating in a feedforward manner without providing a configuration for voltage detection. The former is thought to enable accurate correction if there are no voltage detection errors.

However, in the latter case, since there are individual differences in device characteristics, there may be individual differences between phases even when the same device is used, and considering that the on-resistance and collector-emitter saturation voltage also change with temperature, there is a possibility that the identified value of the primary resistance will have some identification error. In other words, in the case of a power conversion device that does not have a configuration for voltage detection such as a voltage detection circuit, it is thought to be unavoidable that there will be some identification error in the identified value of the primary resistance.

The conventional method includes a description of no-load test and a lock test. In a no-load test, the rated voltage at the rated frequency is applied in a no-load state to rotate the motor, and the current value that flows at this time is measured. When there is no load at all, the rotor speed is the synchronous speed, so if we consider the slip to be zero, the term obtained by dividing the secondary resistance by the slip "s" becomes "∞", which leads to the conclusion that current flows only in the excitation circuit. This expansion also does not include any approximation elements, so no errors due to approximation occur. Figure 15 shows the equivalent circuit for one phase of an induction motor during a no-load test.

If the no-load operating point condition is defined as the rated magnetic flux current i _sn , which is the no-load current value that flows when the rated voltage v _sn is applied to the motor at the rated angular frequency ω _sn , then it can be expressed as in the following equations (17) and (18). Here, the rated voltage v _sn is the phase voltage effective value, and the rated magnetic flux current i _sn is the phase current effective value. In addition, in the following equations (17) and (18), the input voltage is the rated voltage, and the torque current = 0 is considered as the condition.

In this case, since it is expressed by relative conversion, there is a relationship expressed by the following formula (19) between the rated magnetic flux current i _sdn value on the DQ synchronous axis and the rated magnetic flux current i _sn .

Taking these factors into consideration, we obtain the following equations (20) and (21).

Since the sum of the squares of each component of the primary voltage expressed by the above equations (20) and (21) coincides with the square value of the magnitude of the rated voltage v _sn , once the no-load current value is obtained, the primary inductance value _Ls can be obtained by the following equation (22).

However, in cases where only the drive unit is replaced while the motor remains connected in an existing device, it can be difficult to mechanically separate the motor from the device at the destination, making it difficult to rotate the motor and measure the no-load current. In such cases, it is desirable to obtain the no-load current or primary inductance without rotating the motor.

In a lock test, the rotor is mechanically locked, so it does not rotate. If it is difficult to actually lock the rotor, a test equivalent to a lock test can be performed by applying a single-phase voltage. In a lock test, the rotor does not rotate, so "slip = 1", but if 1 is substituted for slip s in the equivalent circuit, it becomes as follows. That is, Figures 16 and 17 show the equivalent circuit for one phase of an induction motor during a lock test.

The equivalent circuit shown in Fig. 16 has the same number of components as the equivalent circuit shown in Fig. 13, and is unchanged. Since the current flowing through the excitation inductance is not 0, even if the primary resistance is considered to be known, it is difficult to separate the secondary constants, that is, terms including the secondary inductance _Lr and secondary resistance _Rr , from the equivalent circuit. Therefore, the equivalent circuit shown in Fig. 17 can be obtained by approximately considering that the constrained state is "slip s = 1", that is, the slip is maximum, and the current mainly flows through the secondary circuit. The relationship between the constants from this equivalent circuit is shown in the following equation (23).

Thus, by observing the applied primary voltage and the current flowing, the sum of the primary resistance and the secondary resistance and the leakage inductance can be obtained. The secondary resistance and the leakage inductance are both primary-side converted values. In this case, the primary resistance _Rs is subtracted from the sum of the primary resistance and the secondary resistance " _Rs + _Rreq " to obtain the secondary resistance _Rreq , so it is inevitable that the accuracy of the primary resistance _Rs will affect the secondary resistance identification value _Rreq . As mentioned above, in consideration of individual differences in the characteristics of semiconductor devices and temperature changes, it is desirable to use a method that can accurately identify the secondary resistance _Rreq or the secondary time constant _Lr / _Rr , which are constants on the secondary side, without being affected by the accuracy of the primary resistance identification value _Rs .

Patent Document 1 discloses a technique for driving an induction motor in an unloaded state, observing the time change in induced voltage by utilizing the induced voltage actually generated by the secondary magnetic flux, and identifying the secondary time constant. However, as mentioned above, it may be difficult to mechanically separate the motor from the equipment at the destination, and therefore it may be difficult to drive the motor in an unloaded or lightly loaded state. Therefore, the method described in Patent Document 1 may be difficult to apply.

In addition, Patent Document 2 discloses a conventional technique as a method of not rotating an electric motor. The method described in Patent Document 2 is equivalent to performing a test equivalent to a restraint test by applying a single-phase AC current to a drive device. Therefore, in the method described in Patent Document 2, as in the restraint test described in the standard, the primary resistance _Rs is subtracted from the sum of the primary resistance and the secondary resistance " _Rs + _Rreq " to obtain the secondary resistance _Rreq , and an accurate primary resistance identification value is required. However, as described above, considering individual differences in the characteristics of semiconductor devices and temperature changes, there is a limit to the identification accuracy of the primary resistance _Rs , and therefore it is inevitable that the identification accuracy of the secondary resistance identification value _Rreq will also be affected. For this reason, there is a limit to improving the identification accuracy of the secondary side constants in a test equivalent to a restraint test.

"Theory and Design Practice of AC Servo Systems - From Fundamentals to Software Servo" 1990 Hidehiko Sugimoto (ed.), Masato Koyama/Shinzo Tamai (authors) Patent No. 6591794 Patent No. 6194718

Therefore, we provide an induction motor drive device that can accurately identify the electrical constants of an induction motor with the rotor stopped.

The induction motor drive device of the embodiment includes a power converter that converts a DC voltage into a three-phase AC voltage with variable voltage and variable frequency and supplies the voltage to the induction motor, a phase-to-phase voltage detection unit that detects the phase-to-phase voltage of the induction motor between two or more phases, a current detection unit that detects the current of the induction motor, a current adjustment unit that calculates a voltage adjustment value so that the current detection value, which is a value obtained by converting the detection value of the current onto the same control axis as the current command value, matches the current command value, a current command value generation unit that generates the current command value and provides it to the current adjustment unit, a voltage command generation unit that generates a voltage command value using the voltage adjustment value obtained from the current adjustment unit, and a motor constant identification unit that identifies the electrical constants of the induction motor based on the detection value of the phase-to-phase voltage or the voltage command value and the current command value or the current detection value.

In the above configuration, the electrical constants identified by the motor constant identification unit include the identified values of the primary resistance, leakage inductance, no-load current, and secondary time constant, and the current command value generation unit generates the current command value of a preset magnitude when the rotor of the induction motor is stopped, and after a preset time has elapsed and the output of the power converter is cut off, the motor constant identification unit identifies the no-load current and the secondary time constant based on the detected value of the phase-to-phase voltage.

The induction motor drive device of the embodiment also includes a power converter that converts a DC voltage into a three-phase AC voltage with variable voltage and variable frequency and supplies the voltage to the induction motor, a current detection unit that detects the current of the induction motor, a current adjustment unit that calculates a voltage adjustment value so that the current detection value, which is a value obtained by converting the detection value of the current onto the same control axis as the current command value, matches the current command value, a current command value generation unit that generates the current command value and provides it to the current adjustment unit, a voltage command generation unit that generates a voltage command value using the voltage adjustment value obtained from the current adjustment unit, and a motor constant identification unit that identifies the electrical constants of the induction motor based on the voltage command value, the current command value, or the current detection value.

In the above configuration, the electrical constants identified by the motor constant identification unit include the identified values of the primary resistance, leakage inductance, no-load current, and secondary time constant, the current command value generation unit generates the current command values of two different magnitudes that are preset when the rotor of the induction motor is stopped, and the motor constant identification unit identifies the no-load current and the secondary time constant based on the voltage command value after the current command value generation unit changes the current command value from a high level to a low level.

FIG. 1 is a schematic diagram illustrating an induction motor driving device according to a first configuration example of an embodiment; FIG. 1 is a timing chart for explaining magnetic flux calculation in a first configuration example according to an embodiment; 1 is a timing chart showing waveforms of each part when identifying a no-load current and a secondary time constant in a first configuration example according to an embodiment of the present invention; FIG. 2 is a schematic diagram illustrating an induction motor driving device according to a second configuration example of an embodiment; FIG. 11 is a timing chart for explaining magnetic flux calculation in a second configuration example according to an embodiment; FIG. 11 is a timing chart for explaining a magnetic flux compensation method when identifying a no-load current and a secondary time constant according to a second configuration example of an embodiment; FIG. 1 is a functional block diagram showing a current control system according to an embodiment. FIG. 10 is a timing chart for explaining optimization of identification of a primary resistance and leakage inductance and identification of a no-load current and a secondary time constant according to a second configuration example of an embodiment; FIG. 13 is a diagram showing the relationship between the target magnetic flux and the no-load current identification value when the target magnetic flux is set to a magnetic flux amount that is 0.75 to 1.5 times the rated magnetic flux according to an embodiment; FIG. 1 is a timing chart for explaining an example of identification of a no-load current taking into account magnetic saturation according to an embodiment, showing a period of target magnetic flux 1; FIG. 1 is a timing chart for explaining an example of identification of a no-load current taking into account magnetic saturation according to an embodiment, showing a period of a flux command 2; FIG. 1 is a timing chart for explaining an example of identification of a no-load current taking into account magnetic saturation according to an embodiment, showing a period of a flux command 3; FIG. 1 is a diagram showing an equivalent circuit for one phase of an induction motor according to a conventional technique. FIG. 1 is a diagram showing an equivalent circuit for one phase of an induction motor when DC is applied according to the prior art. FIG. 1 is a diagram showing an equivalent circuit for one phase of an induction motor during a no-load test according to the prior art. FIG. 1 shows an equivalent circuit for one phase of an induction motor during a lock test according to the prior art. FIG. 2 shows an equivalent circuit for one phase of an induction motor during a lock test according to the prior art FIG. 13 is a diagram showing a voltage command value and a current value when identifying a primary resistance and a leakage inductance according to the prior art;

Hereinafter, an embodiment of an induction motor drive device will be described with reference to the drawings.
In this specification, _{i_sdref} represents the D-axis current command, _{i_sqref} represents the Q-axis current command, _{i_sdmeas} represents the D-axis current detection value, _{i_sqmeas} represents the Q-axis current detection value, _{v_sdref} represents the D-axis voltage command value, _{v_sqref} represents the Q-axis voltage command value, _{v_sdadj} represents the D-axis voltage adjustment value, _{v_sqadj} represents the Q-axis voltage adjustment value, _{v_sdmeas} represents the D-axis voltage detection value, _{v_sqmeas} represents the Q-axis voltage detection value, _{θ_re_c} represents the output voltage phase, _{K_d} represents a voltage command coefficient for identifying leakage inductance, _{R_s_calc} represents the primary resistance identification value, _{L_f_calc} represents the leakage inductance identification value, _{i_sdn_calc} represents the no-load current identification value, and _{T_calc} represents the secondary time constant identification value.

<Regarding the induction motor driving device of the first configuration example>
The induction motor drive device 1 of the first configuration example shown in Fig. 1 has a phase-to-phase voltage detection unit which is a voltage detector. The induction motor drive device 1 is a variable speed motor drive device which drives an induction motor 2 at a variable speed. Each phase terminal of a three-phase AC power source 3 is connected to an AC input terminal of a full-wave rectifier circuit 4 which is made up of a diode bridge. A smoothing capacitor 5 and a power converter 6 are connected between the DC output terminals of the full-wave rectifier circuit 4. The power converter 6 which is an inverter converts a DC voltage into a three-phase AC voltage of variable voltage and variable frequency and supplies it to the induction motor 2, and each phase output terminal of the power converter 6 is connected to each phase winding terminal of the induction motor 2.

The control device 7 controls the power converter 6 to convert the DC voltage into a three-phase AC voltage with variable voltage and variable frequency. Current detectors 8U and 8W that detect the current of the induction motor 2 are arranged at the U-phase and W-phase output terminals of the power converter 6. The U-phase and W-phase currents _iu and _iw detected by the current detectors 8U and 8W are input to a UVW/dq converter 9. The UVW/dq converter 9 converts the three-phase currents into dq-axis currents in vector control and inputs them to a current adjustment unit 10.

The UVW/dq converter 9 converts the current detection value onto the same control axis as the current command value. That is, the UVW/dq converter 9 converts the current detection value onto the synchronous coordinate axis with the output voltage phase θ _{re_c} , which is the phase in the D-axis direction of the synchronous coordinate, as a preset arbitrary fixed value. The D-axis current detection value i _sdmeas and the Q-axis current detection value i _sqmeas , which are values converted by the UVW/dq converter 9, are input to a current adjustment unit 10. The D-axis current detection value i _sdmeas is also input to a motor constant identification unit 11.

Each phase output terminal of the power converter 6 is connected to an input terminal of a phase-to-phase voltage detection unit 12. The phase-to-phase voltage detection unit 12 detects two or more phase-to-phase voltages of the induction motor 2. In this case, the phase-to-phase voltage detection unit 12 detects a _UV -to-phase voltage vuv and a WV-to-phase voltage _vwv . The phase-to-phase voltages detected by the phase-to-phase voltage detection unit 12 are input to a UV,WV/UVW converter 13.

The UV,WV/UVW converter 13 converts the UV interphase voltage _vuv and the WV interphase voltage _vwv into three-phase voltages, i.e., a U-phase voltage _vu , a V-phase voltage _vv , and a W-phase voltage _vw . The UVW/dq converter 14 converts the three-phase voltages corresponding to the detected values of the interphase voltages onto the same control axis as the current command value. That is, the UVW/dq converter 14 converts the three-phase voltages corresponding to the detected values of the interphase voltages onto the synchronous coordinate axes with the output voltage phase _{θre_c} , which is the phase in the D-axis direction of the synchronous coordinates, being a preset arbitrary fixed value.

Of the D-axis voltage detection value v _sdmeas and the Q-axis voltage detection value v _sqmeas which are values converted by the UVW/dq converter 14, the D-axis voltage detection value v _sdmeas is input to the motor constant identification unit 11. The current command value generation unit 15 generates a D-axis current command i _sdref and a Q-axis current command i _sqref which are current command values and provides them to the current adjustment unit 10. Of the D-axis current command i _sdref and the Q-axis current command i _sqref generated by the current command value generation unit 15, the D-axis current command i _sdref is also input to the motor constant identification unit 11.

The current adjustment unit 10 includes a proportional-integral compensator that performs PI control. The current adjustment unit 10 calculates a voltage adjustment value so that the current detection value, which is a value obtained by converting the current detection value onto the same control axis as the current command value, that is, the D-axis current detection value _isdmeas and the Q-axis current detection value _isqmeas , coincide with the D-axis current command i _sdref and the Q-axis current command i _sqref , which are the current command values. The D-axis voltage adjustment value v _sdadj and the Q-axis voltage adjustment value v _sqadj , which are values calculated by the current adjustment unit 10, are input to the voltage command generation unit 16.

The voltage command generating unit 16 generates a voltage command value using the D-axis voltage adjustment value v _sdadj and the Q-axis voltage adjustment value v _sqadj , which are voltage adjustment values obtained from the current adjusting unit 10, and the leakage inductance identification voltage command coefficient _Kd . The D-axis voltage command value v _sdref and the Q-axis voltage command value v _sqref generated by the voltage command generating unit 16 are input to the dq/UVW converter 17. The D-axis voltage command value v _sdref is also input to the motor constant identifying unit 11. The dq/UVW converter 17 converts the input D-axis voltage command value v _sdref and the Q-axis voltage command value v _sqref into three-phase voltage commands, and generates a PWM signal to output to the power converter 6. Note that PWM is an abbreviation for Pulse Width Modulation.

Motor constant identification unit 11 identifies the electrical constants of induction motor 2 based on the D-axis voltage detection value v _sdmeas or the D-axis voltage command value v _sdref corresponding to the detection value of the phase-to-phase voltage, and the D-axis current command i _sdref or the D-axis current detection value i _sdmeas . The electrical constants identified by motor constant identification unit 11 include the identified values of the primary resistance, leakage inductance, no-load current, and secondary time constant, that is, the primary resistance identified value R _{s_calc} , the leakage inductance identified value L _{f_calc} , the no-load current identified value i _{sdn_calc} , and the secondary time constant identified value T _{r_calc} .

Details will be described later, but in the above configuration, the electrical constants are identified as follows. That is, the current command value generating unit 15 generates a current command value of a preset magnitude when the rotor of the induction motor 2 is stopped. Then, after a preset time has elapsed and the output of the power converter 6 is cut off, the motor constant identifying unit 11 identifies the no-load current and the secondary time constant based on the detected values of the phase-to-phase voltages.

Next, the principle of identification of each electrical constant implemented in the induction motor driving device 1 of the first configuration example will be described based on equation (3) which is the motor model used in this embodiment.
<Identification of primary resistance>
Identification of the primary resistance is basically the same as the method of applying direct current described in standards, and is performed by induction motor driving device 1 having power converter 6.
When the third line of equation (3) is written, it becomes equation (24) below.

From equation (24), we can see that the behavior of the secondary magnetic flux D-axis component φ _rd has dynamics that change with a secondary time constant so that the magnitude of the secondary magnetic flux D-axis component φ rd becomes " _Mi sd ". If the left side of the first line of equation (3) is changed to the primary voltage v _sd , we obtain the following equation (25).

Here, assuming that there are two conditions, that is, "the rotor is stopped and no frequency component is applied as the output voltage" and "current is applied only to the D-axis side", then " _ωs = 0", "i _sq = 0" and " _φrq = 0". Therefore, the above formula (25) becomes the following formula (26). Note that in this explanation and similar explanations below, the direction of the D-axis is the position _{θre_c} where none of the current values of the U-phase, V-phase and W-phase are close to 0.

In the above formula (26), "i _sd " is the current value of the induction motor 2. Here, it is assumed that the current is applied for a sufficient time T ₀ so that the state can be treated as being in a steady state. Then, when the steady state is reached, taking into consideration that "φ _rd =Mi _sd ", the above formula (26) becomes the following formula (27).

In this case, the applied voltage can be grasped as a voltage command _{v_sdref} , and the current is known from the current detection value _{i_sdmeas} or the current command value _{i_sdref} . Therefore, the primary resistance identification value _{R_s_calc} can be obtained as shown in the following equation (28).

If there are errors in the dead time correction, the on-resistance correction of the semiconductor element, the collector-emitter saturation voltage correction, etc., in the power converter 6 composed of semiconductor elements such as IGBTs, the calculation of only one operating point "v _sdref , i _sdref " as described above may directly affect the primary resistance identified value R _{s_calc} due to the influence of all of the above errors. Therefore, in order to reduce the influence of the error in the dead time correction and the error in the collector-emitter saturation voltage correction, which do not change much depending on the magnitude of the current, as much as possible, the primary resistance identified value R _{s_calc} may be calculated using the difference between the voltage command values and the current command values of two operating points, "v _sdref1 -v _sdref2 , i _sdref1 -i _sdref2 ," as shown in the following formula (29).

Note that even when the primary resistance identified value R _{s_calc} is calculated using the above equation (29), an error in the voltage drop correction for the on-resistance of the semiconductor element, which is a component proportional to the current value, remains as a primary resistance identification error.
The voltage command value and the current value during identification of the primary resistance performed as described above are as shown in the primary resistance identification phase in FIG.

<Identification of leakage inductance>
The leakage inductance value is identified by applying a single-phase AC current after the above-mentioned identification of the primary resistance value is completed. The leakage inductance is identified by a method equivalent to the binding test described in the standard. In this case, the primary voltage D-axis component v _sd and the primary current D-axis component i _sd are expressed by the following formulas (30) and (31), respectively. However, the iron loss resistance is treated as an equivalent circuit that does not take into account the iron loss resistance. The current value when the current i _sd flowing during the primary resistance identification is in a steady state is represented as i _sd0 , the voltage command value at that time is represented as v _sdref0 (=R _s i _sd0 ), and the change in current when Δv _sd is added to the voltage command as a change having a frequency component from the initial state is represented as Δi _sd .

In the identification of the primary resistance, the secondary magnetic flux φ _rd is sufficiently converged to a steady value Mi _sd , and the secondary time constant is sufficiently slow compared to the change in current due to the applied AC voltage. Therefore, if it is considered that the secondary magnetic flux remains almost constant Mi _sd0 during the identification of the leakage inductance, the relational expression between the voltage change Δv _sd and the current change Δi _sd in formula (26) can be expressed as the following formula (32).

If we write the above equation (32) as an equation that only includes the change, we get the following equation (33).

Here, if the leakage inductance _Lf is expressed as in the following formula (34), the secondary resistance _Rreq is expressed as in the following formula (35), and the sum of the primary resistance and the secondary resistance _Rtotal is expressed as in the following formula (36), the relational expression for only the change amount becomes the following formula (37). Note that the secondary resistance in this case is a primary converted value.

The above equation (37) is identical to the equivalent circuit during the restraint test shown in FIG. 16. In other words, the equivalent circuit used in the restraint test is equivalent to the state in which the above approximation is taken into account in the differential equation in equation (3).

As the voltage change amount Δv _sd for generating an AC current, a sine wave is applied whose amplitude is a voltage value obtained by multiplying the voltage value v _sdref0 (=R _{si sd0} ) at the time of identifying the primary resistance by _Kd , which is a coefficient less than 1. The coefficient _Kd can be set to, for example, 0.25. In this case, the voltage change amount Δv _sd is expressed by the following formula (38). Here, each frequency of the sine wave described above is represented by ω.

In this way, the relational equation for only the change can be expressed as in equation (39) below.

When both sides of the above equation (39) are Laplace transformed, we obtain the following equation (40). Then, rearranging equation (40) gives us the following equation (41).

Assuming that the right-hand side of the above equation (41) can be decomposed as in the following equation (42), and combining the two terms in parentheses on the right-hand side, we get the following equation (43).

Rearranging the numerator of the above equation (43) with respect to the Laplace operator s gives the following equation (44).

If we consider equations (41) and (44) to be identities with respect to the Laplace operator s and compare the coefficients on the right-hand side, we obtain the following three equations: (45), (46), and (47).

A to C are found based on the above three equations. First, equation (48) below is obtained from equations (45) and (46). Then, substituting equation (48) into equation (47) gives equation (49) below. By rearranging equation (49), A can be found as in equation (50) below.

Since A is obtained in this way, B and C can be derived as shown in equations (51) and (52) below.

As described above, A to C were found, but for simplicity, we will leave them written as A to C. If we rewrite equation (41), which is the Laplace transform of equation (39), which is the relationship between voltage and current, using A to C, we get the following equations (53) and (54).

When the above equation (54) is inversely Laplace transformed, it becomes the following equation (55). Then, when A to C are substituted into equation (55), it becomes the following equation (56).

Since the first term in the parentheses in the above formula (56) is a term that decays with time, it may be ignored after a time sufficiently longer than the time constant ( _Lf / _Rtotal ) has elapsed. Taking this into consideration, the above formula (56) can be simplified as shown in the following formula (57).

Here, the above-mentioned equation for the current value is expressed by coefficients _Hs and _Hc as in the following equation (58). In this case, the coefficients _Hs and _Hc are defined as in the following equations (59), (60), and (61). According to such definitions, it can be seen that if the _{amplitude Kdvsd0} and the angular frequency ω of the applied frequency component are constant, the coefficients _Hs and _Hc are also constant values.

From the above definitions, the leakage inductance L _f (=σL _s ) and the sum R _total of the primary resistance and secondary resistance can be expressed as the following equations (62) and (63), respectively.

The coefficients _Hs and _Hc are respectively a coefficient multiplied by the sine function and a coefficient multiplied by the cosine function of the current value equation shown in equation (58). The coefficients _Hs and _Hc are obtained by integrating the value obtained by multiplying the current value flowing during N cycles of the applied AC voltage by a cosine function or a sine function, as shown in the following equations (64) and (65). Tst is the time when the integration starts. The integration time is set to a time sufficiently longer than the time constant ( _Lf / _Rtotal ). However, "T = (2π/ω) [s]".

From this, the coefficients _Hs and _Hc can be obtained as shown in the following equations (66) and (67).

By substituting the coefficients _Hs and _Hc thus obtained into the equations (62) and (63), the leakage inductance identified value _{Lf_calc} and the sum _{Rtotal_calc} of the primary resistance and the secondary resistance can be obtained as shown in the following equations (68) and (69). Here, only the leakage inductance identified value _{Lf_calc} is used.

The voltage command value and current value during leakage inductance identification performed as described above are as shown in the leakage inductance identification phase in Figure 18.

<Identification of secondary time constant and no-load current>
In the induction motor drive device 1 of the first configuration example, the output is shut off, i.e., gated and blocked, after current is applied, and the magnetic flux is calculated using the voltage detection value. That is, as shown in Fig. 2, when DC is applied to the induction motor 2 and the voltage command value becomes a constant value, the gate is blocked so that the output voltage becomes 0, and then, due to the change in the attenuating secondary magnetic flux, a voltage is generated at the motor terminals.

The voltage value appearing at the motor terminals is found, and the amount of magnetic flux that results from its integral value is calculated. This modeling is performed as follows. First, the following equation (70) is obtained from the first line of equation (3).

After the gate is blocked, the current value converges to 0 in a time that is much shorter than the secondary time constant. Therefore, if the primary frequency is set to 0, the rotor is not moving, and the Q-axis magnetic flux is set to 0, the following equation (71) is obtained.

On the other hand, if you write the third line of equation (3), it becomes equation (72) below.

Similarly, after the gate is blocked, the current value converges to 0 in a time much shorter than the quadratic time constant, so equation (72) can be expressed as equation (73) below.

When this differential equation is solved with the initial value of the secondary magnetic flux set to φ _rd0 (=Mi _sd0 ) and then subjected to Laplace transformation, the following equations (74) and (75) are obtained.

The secondary magnetic flux can be expressed as a function of time as shown in equation (76) below.

From the above equations (71) to (76), the observed voltage at the motor terminals can be calculated using the following equations (77) and (78).

Since the following equation (79) is obtained, the magnetic flux that was generated before the attenuation can be obtained by integrating the observed voltage value until it approaches zero. In other words, the magnetic flux can be obtained as shown in the following equation (80).

Furthermore, since the following equation (81) is obtained, it can be expressed as the following equations (82) and (83). This gives the following equation (84).

From this magnetic flux equation, it can be seen that when the flowing current i _sd0 is equivalent to the no-load current i _sdn of the induction motor 2, the magnetic flux amount obtained is the rated primary magnetic flux L _s i _sdn minus the leakage magnetic flux amount σL _s i _sdn .

The induction motor 2 is a motor that can change the magnetic flux by the applied voltage. Therefore, there are various methods for determining the magnetic flux reference. Here, the no-load current value that flows when the motor is operated at a rated angular frequency ω _sn and a rated voltage v _sn is defined as the rated magnetic flux current i _sn , and the value converted onto the DQ synchronous axis is defined as the rated magnetic flux current i _sdn , which is the condition for control. Here, the rated voltage V _sn is the phase voltage effective value, and the rated magnetic flux current i _sn is the phase current effective value.

In this case, the operating point is the same as in the no-load test described above, so equations (20) and (21) are obtained in the same way.Then, the sum of the squares of each component of the primary voltage expressed by equations (20) and (21) coincides with the magnitude of the rated voltage _Vsn , that is, the square value of the amplitude, so the rated primary magnetic flux _Ls i _sdn can be obtained by the following equations (85) and (86).

The amount of magnetic flux obtained by multiplying the rated primary magnetic flux _Ls i _sdn obtained in this manner by "1-σ", that is, the amount of magnetic flux expressed by the following formula (87), is the amount of magnetic flux obtained when the no-load current i _sdn is identified. At the start of identification or during identification, the no-load current i _sdn is an unknown quantity, that is, is being identified, so the target magnetic flux is determined as shown in the following formula (88) using the current value i _sd0 given at that time.

When the applied current value i _sd0 is adjusted so that the target magnetic flux and the magnetic flux value of the previous equation (84) match, the applied current value i _sd0 is adjusted to be equivalent to the no-load current i _sdn . In this manner, the no-load current i _sdn is identified. Once the no-load current i _sdn is identified, the primary inductance L _s can also be obtained from equations (85) and (86). At this time, the time until the magnetic flux value of equation (84) becomes equivalent to "1-e ^-1 " times the target magnetic flux is approximately the secondary time constant, so that the secondary time constant can also be measured at the same time.

The adjustment of the applied current value is performed, for example, according to the following rules. That is, n is the number of times the current value is applied. However, "n = 1, 2, 3...", that is, n is a natural number. When n = 1, for example, "current (1) = √2 x In/5" is given as the initial value. However, In is equivalent to the rated current effective value. When n ≥ 2, the current value, that is, the current command value, is calculated using the following formula (89). However, magnetic flux (0) = 0 and current (0) = 0.

As a result of adjusting the current (n) using equation (89) so that the target magnetic flux coincides with the magnetic flux in equation (84), which is the integral of the voltage value, the no-load current identification value i _{sdn_calc} can be made to coincide with the current (n), as shown in equation (90) below.

The above formula (89) is derived from the following idea. That is, the ratio of the change in current to the change in magnetic flux is calculated from the past two current and magnetic flux values, as shown in the following formula (91).

As shown in the following equation (92), the amount of magnetic flux that is the result of the previous integration is calculated to be how much short of the target magnetic flux.

As shown in the following formula (93), the current value required to obtain the magnetic flux deficiency is calculated by multiplying the magnetic flux deficiency by the ratio previously determined.

As shown in equation (94) below, the amount of shortfall current is added to the previous current command to obtain the next current command value.

From the relationships in equations (91) to (94) above, the following current command equation, that is, equation (89), is derived as shown in equation (95) below.

The waveforms of each part when identifying the no-load current and secondary time constant as described above are as shown in Figure 3. Note that in Figure 3, the gate block is abbreviated as GB. In the induction motor drive device 1 of the first configuration example, the operation for obtaining the identified value of the no-load current and the identified value of the secondary time constant is performed in the following manner.

A current command value generating unit 15 provides a first current command value i _{sdref_[1]} , which is about 20% of the rated current of the induction motor 2, to a current adjusting unit 10 for a predetermined first waiting time. After the output of the power converter 6 is cut off and a predetermined second waiting time has elapsed, a motor constant identifying unit 11 calculates a magnetic flux [1] shown in the following equation while a value v _sdmeas , which is obtained by converting the detected value of the phase-to-phase voltage onto the same control axis as the current command value, falls below 0.
Magnetic flux [1] = ∫ (-v _{sd m ea s} ) dt

Here, the target magnetic flux is determined to be the amount of magnetic flux obtained by subtracting the leakage magnetic flux calculated using the identified value of the leakage inductance from the rated primary magnetic flux of the induction motor 2. The current command value generating unit 15 provides the second current command value i _{sdref_[2]} , which is calculated by the following formula using the magnetic flux[1], to the current adjusting unit 10.
i _{sdref_[2]} = i _{sdref_[1]} + (target flux - flux[1]) x (i _{sdref_[1]} ÷ flux[1])

The motor constant identification unit 11 calculates the magnetic flux [2] in the same manner as when calculating the magnetic flux [1]. Here, when N is a natural number of 3 or more, the current command value generation unit 15 provides the Nth current command value i _{sdref_[N]} calculated by the following formula to the current adjustment unit 10.
i _{sdref_[N]} = i _{sdref_[N-1]} + (target flux - flux[N-1]) x (i _{sdref_[N-1]} - _{i sdref_[N-2]} ) ÷ (flux[N-1] - flux[N-2])

The motor constant identification unit 11 calculates the magnetic flux [N] in the same manner as when calculating the magnetic flux [1]. By repeating these operations to perform a convergent calculation of i _{sdref_[N]} , an identified value i _{sdn_calc} of the no-load current shown in the following equation is obtained.
i _{sdn_calc} = i _{sdref_[N]}

When the current command value generating unit 15 again provides the Nth current command value i _{sdref_ [N] to the current adjusting unit 10 as the N+1th current command value i sdref_} [N+1] and calculates the magnetic flux [N+1] in the same manner as when calculating the magnetic flux [1], a measurement operation is performed to measure the time T _{r_est} until the magnetic flux [N+1] becomes approximately "1-e ^-1 " times the target magnetic flux. Such a measurement operation is performed once or multiple times, and the value of the measured time T _{r_est} or the average value of multiple measured times T _{r_est} is set as the identified value T _{r_calc} of the secondary time constant.

<Regarding the induction motor driving device of the second configuration example>
In the identification method using the induction motor drive device 1 of the first configuration example, it is necessary to detect the voltage value observed before the magnetic flux generated by current application attenuates after the gate is blocked, so the phase-to-phase voltage detection unit 12, which is a voltage detector, and the UV,WV/UVW converter 13 and UVW/dq converter 14, which are peripheral circuits, are required. From the viewpoint of keeping the number of parts as small as possible, it is desirable to be able to identify the no-load current and the secondary time constant in an induction motor drive device having a configuration without a voltage detector, as in the first configuration example having a voltage detector. Below, an induction motor drive device of a second configuration example having no such voltage detector will be described.

The induction motor drive device 21 of the second configuration example shown in FIG. 4 is configured without a phase-to-phase voltage detection unit, which is a voltage detector. The induction motor drive device 21 of the second configuration example differs from the induction motor drive device 1 of the first configuration example in that it has a control device 22 instead of the control device 7. The control device 22 differs from the control device 7 in that it omits the phase-to-phase voltage detection unit 12, the UV,WV/UVW converter 13, and the UVW/dq converter 14, and that it has a motor constant identification unit 23 instead of the motor constant identification unit 11.

The motor constant identifying unit 23 identifies the electrical constants of the induction motor 2 based on the D-axis voltage command value v _sdref and the D-axis current command i _sdref or the D-axis current detection value i _sdmeas . Although details will be described later, in the above configuration, the electrical constants are identified as follows. That is, the current command value generating unit 15 generates two different preset current command values in a state in which the rotor of the induction motor 2 is stopped. Then, the motor constant identifying unit 23 identifies the no-load current and the secondary time constant based on the voltage command value after the current command value generating unit 15 changes the current command value from a high level to a low level.

In the induction motor driving device 21 of the second configuration example, the operation for acquiring the identified value of the no-load current and the identified value of the secondary time constant is performed as follows: The current command value generating unit 15 provides the base current command value i _{sdref_base} , which is a predetermined relatively small current value when the induction motor 2 is stopped, to the current adjusting unit 10 for a predetermined first waiting time Twait_A, and then the motor constant identifying unit 23 stores the voltage command value v _{sdref_base} , which has become a steady value.

Then, the current command value generating unit 15 provides the first current command value i _{sdref_[1]} , which is greater than the base current command value i _{sdref_base} , to the current adjusting unit 10 for the first waiting time Twait_A, and then switches back to the base current command value i _{sdref_base} . After a predetermined second waiting time Twait_B has elapsed, the motor constant identifying unit 23 calculates the magnetic flux [1] shown in the following equation while the voltage command value v _sdref falls below the voltage command value v _{sdref_base} , which is the steady-state value when the base current command value i _{sdref_base} is applied.
Magnetic flux [1] = ∫ (v _{sdref_base} - v _sdref ) dt

Here, the target magnetic flux is determined to be the amount of magnetic flux obtained by subtracting the leakage magnetic flux calculated using the identified value of the leakage inductance from the rated primary magnetic flux of the induction motor 2. The current command value generating unit 15 provides the second current command value i _{sdref_[2]} calculated by the following equation to the current adjusting unit 10.
i _{sdref_[2]} = i _{sdref_[1]} + (target flux - flux[1]) x (i _{sdref_[1]} ÷ flux[1])

The motor constant identification unit 23 calculates the magnetic flux [2] in the same manner as when calculating the magnetic flux [1]. Here, when N is a natural number of 3 or more, the current command value generation unit 15 provides the Nth current command value i _{sdref_[N]} calculated by the following formula to the current adjustment unit 10.
i _{sdref_[N]} = i _{sdref_[N-1]} + (target flux - flux[N-1]) x (i _{sdref_[N-1]} - _{i sdref_[N-2]} ) ÷ (flux[N-1] - flux[N-2])

The motor constant identification unit 23 calculates the magnetic flux [N] in the same manner as when calculating the magnetic flux [1]. By repeating these operations to perform a convergent calculation of i _{sdref_[N]} , an identified value i _{sdn_calc} of the no-load current is obtained as shown in the following equation.
i _{sdn_calc} = i _{sdref_[N]} - i _{sdref_base}

When the current command value generating unit 15 again provides the Nth current command value i _{sdref_ [N] to the current adjusting unit 10 as the N+1th current command value i sdref_} [N+1] and calculates the magnetic flux [N+1] in the same manner as when calculating the magnetic flux [1], a measurement operation is performed to measure the time T _{r_est} until the magnetic flux [N+1] becomes approximately "1-e ^-1 " times the target magnetic flux. Such a measurement operation is performed once or a plurality of times, and the measured value of the time T _{r_est} or the average value of the plurality of measured times T _{r_est} is set as the identified value T _{r_calc} of the secondary time constant.

Next, a specific method for identifying each electrical constant implemented in the induction motor driving device 21 of the second configuration example will be described. As shown in Fig. 5, the identification method of the second configuration example employs a method in which the base current command value _{i_sdref_base} is continuously applied as a minute current value, instead of the gate block in the identification method of the first configuration example.

In this case, the voltage command after quickly decreasing the current value from the "no-load current identification value + base current command value i _{sdref_base} " to the current value of the base current command value i _{sdref_base} is used in the same way as the voltage detection value. However, in order to do this, the following two points must be noted.
[1] Elimination of voltage value for maintaining small current instead of gate block [2] Avoidance of transient phenomenon when transitioning to small current instead of gate block

<Removal of voltage value to maintain minute current>
It is necessary to measure in advance the base voltage command value v _{sdref_base} , which is a voltage command value that maintains the current level of the base current command value i _{sdref_base} , and to integrate the voltage value obtained by subtracting the base voltage command value v _{sdref_base} from the voltage command value, by substituting it for the voltage detection value. The magnetic flux obtained by such integration is expressed by the following equation (96).

As a result of adjusting the current (n) by equation (89) so that the target magnetic flux coincides with the magnetic flux in equation (96), which is the integral of the voltage value, the current value equivalent to the "no-load current identified value + base current command value i _{sdref_base} " is adjusted as the current (n). Therefore, the no-load current identified value i _{sdn_calc} is expressed by the following equation (97).

<Avoiding transient phenomena in current control>
When the current command value switches from "no-load current identification value + base current command value i _{sdref_base} " to the base current command value i _{sdref_base} , the voltage command value, which is the output of the current controller while the actual current value is transitioning, also includes the compensation amount of the current controller. Therefore, in order to accurately identify the no-load current, it is necessary to remove the compensation amount corresponding to the output of the current controller during this transient phenomenon from the integral calculation of the voltage command that calculates the magnetic flux. For this purpose, it is effective not to perform the integral operation of the voltage command value for a preset time Twait_B after the start of the transition.

When the induction motor 2 is a relatively large capacity model, the secondary time constant is sufficiently long compared with the current response time constant of the current controller, so the impact on the identification result of the no-load current value is small even if the integral operation is not performed during the time Twait_B. However, when the induction motor 2 is a small capacity model with a short secondary time constant, when the difference between the secondary time constant T _r and the current response time constant T _curr becomes small, the impact on the identification accuracy cannot be ignored.

Therefore, when the difference between the secondary time constant T _r and the current response time constant T _curr is small, the voltage value due to the magnetic flux change during the waiting time Twait_B, that is, the voltage value corresponding to the formula (77), is estimated from the change in the voltage command value after the current value reaches the base current command value i _{sdref_base} , and the magnetic flux amount is compensated to improve the accuracy of the no-load current identification value. Note that the magnetic flux amount in this case is an integral value. Specifically, a linear approximation as shown in FIG. 6 is performed.

The time when the current command value is switched to the base current command value i _{sdref_base} is defined as "t=0", the time when the current value has almost completely decayed to the base current command value i _{sdref_base} is defined as "t=T0", the time when the current value is sufficiently close to the base current command value i _{sdref_base} is defined as "t=T1", and double that time is defined as T2. For example, T0 can be defined as "2×current response time constant T _curr " and T1 can be defined as "N×T _curr ". In this case, for example, N can be defined as "N=5".

The timer setting values T0, T1, and T2 are known, and V1 and V2 are the voltage command values at time T1 and time T2, respectively. The slope of the change in the voltage command between time T1 and time T2 is found, and the voltage value V0 due to the magnetic flux change among the voltage command values at time T0 is approximately calculated. Such an approximation can be performed based on the following equation (98).

The timer is set as shown in the following formulas (99) to (101), where N is an integer.

Then, the voltage value V0 is expressed as follows:

When the magnetic flux change amount, which is the integral value from T0 to T1, is calculated by trapezoidal approximation, it becomes the following formula (103).

Substituting V0 in the above formula (102) into the above formula (103) gives the integral value S of the voltage value of the magnetic flux change excluding the current control compensation amount from T0 to T1. The integration of the voltage command value, which is the calculation of the magnetic flux, starts at "t=T1", the point in time when the current value is sufficiently close to the base current command value _{isdref_base} . This is because the compensation voltage command value, which is the output of the current controller, is not integrated. Thereafter, by adding the integral value S obtained as described above to the magnetic flux calculation value at the point in time "t=T2", the magnetic flux amount equivalent to the integral value of the voltage value generated by the secondary magnetic flux change before T1 is corrected, making it possible to identify the no-load current with high accuracy.

As described above, in the second configuration example, when calculating the magnetic flux [1], magnetic flux [2], magnetic flux [N], and magnetic flux [N+1], the current command value is switched to the base current command value i _{sdref_base} , and the amount of magnetic flux change in the induction motor 2 during the second waiting time Twait_B is calculated as a magnetic flux correction amount from the amount of change over time of the voltage command value observed after the second waiting time Twait_B has elapsed, and the magnetic flux correction amount is added to the magnetic flux amounts of the magnetic flux [1], magnetic flux [2], magnetic flux [N], and magnetic flux [N+1].

More specifically, in the second configuration example, when calculating magnetic flux [1], magnetic flux [2], magnetic flux [N], and magnetic flux [N+1], the current command value is switched to the base current command value i _{sdref_base} , and the induced voltage due to the magnetic flux change in the induction motor 2 during the second waiting time Twait_B is linearly approximated from the time change amount of the voltage command value observed after the second waiting time Twait_B has elapsed. The magnetic flux correction amount is calculated as the amount obtained by integrating the linearly approximated induced voltage over time during the second waiting time Twait_B, and the magnetic flux correction amount is added to the magnetic flux amounts of magnetic flux [1], magnetic flux [2], magnetic flux [N], and magnetic flux [N+1].

<Optimization of current control gain>
At this time, it is desirable to set the gain of the current control to a gain that does not cause overshoot. This is because realizing a current response that does not cause overshoot has the effect of preventing a large compensation voltage value from being generated in the voltage command value. When the current controller is realized by a PI compensator, the proportional gain _Kp and integral gain _Ki of the current controller that does not cause overshoot in the current response can be obtained from the leakage inductance value and primary resistance value obtained in the previous stage as shown in the following equations (104) and (105). Here, the current response angular frequency is ω _curr .

The derivation of such a current control gain will be explained below. First, the first line of equation (3) is written as follows: (106).

The voltage applied here is DC, and since current is applied only to the D-axis when the induction motor 2 is stopped and the rotor is not rotating, the Q-axis current can be considered to be 0, and the Q-axis component of the secondary magnetic flux can also be considered to be 0. Taking this into account, equation (106) can be rearranged to give the following equation (107).

Although there are differential terms between the primary current and the secondary magnetic flux, the change in the secondary time constant is sufficiently slow compared to the dynamics of the primary current, so during the time it takes for a relatively fast current controller to achieve a step-like response of 50 to 100 Hz, the secondary magnetic flux can be considered to remain almost unchanged and constant. Therefore, if we consider the secondary magnetic flux to be constant in the above equation (107) during the time when the current value responds immediately after the start of current control, that is, immediately after the current command is changed, then the following equation (108) holds.

また、この場合、電流制御器はＰＩ制御により実現するようになっているため、下記（１０９）式のように表せる。

In this case, the current controller is realized by PI control, and can be expressed as in the following equation (109).

When the above formulas (108) and (109) are expressed as a block diagram, it becomes a diagram as shown in Fig. 7. When the transfer function _Gc (s) from the current command to the current value is obtained, it becomes the following formula (110).

When it is desired to set the current control response to ω _curr , that is, when it is desired to set the current response shown in the above formula (110) to "a first-order lag response with a response time constant T _curr of 1/ω _curr ", it is sufficient to determine the proportional gain K _p and the integral gain K _i so that the following formula (111) holds.

Rearranging the above equation (111), we obtain the relationship shown in the following equation (112). Then, by comparing the coefficients, we obtain the following equations (113) and (114).

Therefore, the proportional gain _Kp and the integral gain _Ki can be set as shown in the above formulas (104) and (105). Furthermore, when these formulas are expressed using an identification formula, they become the following formulas (115) and (116).

<Optimization of identification sequence (waiting time)>
8 is a chart showing the optimization of a series of operations related to the identification of four electrical constants, the primary resistance, the leakage inductance, the no-load current, and the secondary time constant. In identifying the no-load current, it is desirable to have a current response without overshoot, so it is advisable to identify the primary resistance and the leakage inductance first, and then optimize the control gain of the current controller.

In the induction motor driving device 21 of the second configuration example having no voltage detector, the base voltage command value v _{sdref_base,} which is the voltage reference when calculating the magnetic flux, is required when identifying the no-load current, but the voltage command required when applying the base current command value i _{sdref_base} , which is the initial current command value at the no-load current, is unknown. Therefore, the base voltage command value v _{sdref_base_ini} is determined as an initial value, for example, as shown in the following formula (117), using a value smaller than the previously identified primary resistance identification value.

After identifying the primary resistance and leakage inductance, the secondary time constant is roughly adjusted using the induced voltage equivalent generated in the voltage command value after the current command value is switched to the base current command value i _{sdref_base} . Specifically, the time from when the current command value is switched until when the voltage command value exceeds the base voltage command initial value v _{sdref_base_ini} is measured to obtain an approximate secondary time constant rough adjustment value T _{r_temp} .

In the case of the induction motor driving device 1 of the first configuration example having a voltage detector, the observed induced voltage ultimately becomes close to zero, so as an example of a small negative value, a negative value of the above-mentioned base voltage command initial value v _{sdref_base_ini} is set as the threshold value, and the secondary time constant coarse adjustment value T _{r_temp} is obtained by measuring the time until the detected voltage becomes higher than the threshold value.

The waiting time Twait_A is determined based on the thus obtained secondary time constant coarse adjustment value T _{r_temp} . The waiting time Twait_A may be set to, for example, about five times the secondary time constant coarse adjustment value T _{r_temp} . The identification sequence can be optimized by using the waiting time Twait_A as the current application time and the integration time for magnetic flux calculation in the subsequent identification of the no-load current value.

As described above, in induction motor driving device 21 of the second configuration example, the identification sequence is optimized as follows: That is, when transitioning from the current command value used during the operation for acquiring the identified value _{Rs_calc} of the primary resistance and the identified value _{Lf_calc} of the leakage inductance to the base current command value _{i_sdref_base} , which is the initial current command value during identification of the no-load current, the coarse adjustment value _{Tr_temp} of the secondary time constant is measured by observing the voltage command value.

In the induction motor driving device 1 of the first configuration example, after obtaining the identified value _{Rs_calc} of the primary resistance and the identified value _{Lf_calc} of the leakage inductance, the output of the power converter 6 is temporarily shut off and then the coarse adjustment value _{Tr_temp} of the secondary time constant is measured by observing the voltage detection value. In this case, when the current response angular frequency is _ωcurr , the first waiting time Twait_A is set to about 5 to 10 times the coarse adjustment value _{Tr_temp} of the secondary time constant, and the second waiting time Twait_B is set to about 2 to 5 times the reciprocal of the current response angular frequency _ωcurr .

<Acquisition of saturation characteristics in no-load current identification>
It has been shown that the target magnetic flux level in identifying the no-load current is determined as shown in equations (85) and (86). If the no-load current value that flows during no-load operation at the rated angular frequency ω _sn and the rated voltage v _sn is defined as the rated magnetic flux current i _sn , and the value converted onto the DQ synchronous axis is the rated magnetic flux current i _sdn , then the amount of magnetic flux that is generated when this current value flows on the D axis is defined as the rated magnetic flux.

As shown in Figure 9, by setting the target flux to, for example, 0.75, 1.0, 1.25, or 1.5 times the rated flux, and performing operations similar to those performed when identifying the no-load current, the D-axis current level for generating each flux amount can be identified. Specific examples of operations related to such identification include the first operation example shown in Figure 10, the second operation example shown in Figure 11, and the third operation example shown in Figure 12. If the electrical constant does not change depending on the flux amount, the flux and current value are proportional, but magnetic saturation characteristics make it difficult to generate flux, and the greater the flux amount, the greater the deviation from the proportional relationship.

As described above, in this embodiment, the motor constant identification units 11 and 23 are configured to select the target flux from at least three values: a rated flux amount obtained by subtracting the leakage flux amount calculated using the leakage inductance identification value from the rated primary flux of the induction motor 2, a flux amount greater than the rated flux amount, and a flux amount smaller than the rated flux amount. When each of the three target fluxes is selected, the no-load current is identified to obtain the magnetic saturation characteristics of the induction motor 2.

<Optimization of the method for identifying primary resistance and leakage inductance>
In this embodiment, the operation for obtaining the identified value of the primary resistance and the identified value of the leakage inductance is performed in the following manner before the operation for obtaining the identified value of the no-load current and the identified value of the secondary time constant is performed. First, the current command value generating unit 15 provides the current adjusting unit 10 with a current command value i _sdref0 of a predetermined magnitude for a predetermined time T0. Then, after the predetermined time T0 has elapsed, the motor constant identifying units 11 and 23 calculate the identified value R s_calc of the primary resistance using the voltage command value v _sdref0 , which is the voltage command value that has become a steady value, and the current command value i _sdref0 or the current detection value i _sd0 , which is the current detection value that has become a steady _value .

Next, the voltage command generating unit 16 calculates the identified value _{Rs_calc} of the primary resistance as a voltage command value obtained by adding the voltage command value _vsdref0 to a sine wave whose amplitude is a value obtained by multiplying the voltage command value _vsdref0 by a coefficient _Kd that is a value smaller than 1. Then, the coefficient _Kd is adjusted so that the amplitude of the AC component of the observed current becomes a value equivalent to a predetermined magnitude. After that, the motor constant identifying units 11, 23 calculate the identified value Lf_calc of the leakage _inductance from the AC amount of the current value observed by the current detecting units 8U, 8W, the adjusted coefficient _Kd , and the voltage command value _vsdref0 .

The identified value _{Rs_calc} of the primary resistance and the identified value _{Lf_calc} of the leakage inductance are used to calculate the set values of the proportional gain _Kp and the integral gain _Ki of the proportional and integral compensator included in the current adjusting unit 10. Specifically, when the current response angular frequency is _ωcurr , the values of the proportional gain Kp and the integral gain _Ki of the proportional and integral compensator set using the identified value _{Rs_calc} of the primary resistance and the identified value _{Lf_calc} of the leakage _inductance are set as shown in the following equations.
_Kp = ω _curr
K _i = ω _curr × R _{s_calc} ÷ L _{f_calc}

As described above, the identification method implemented by the induction motor drive devices 1 and 21 of this embodiment allows the electrical constants of the induction motor 2 to be identified with high accuracy while the rotor of the induction motor 2 is stopped. In particular, the identification method of this embodiment makes it possible to make the identification of the no-load current and secondary time constant, which is performed while the rotor of the induction motor 2 is stopped, less susceptible to the influence of identification errors in the primary resistance. Furthermore, since the accuracy of the leakage inductance affects the identification accuracy of the no-load current, the identification method of this embodiment employs a method that allows the accuracy of the leakage inductance identification to be maintained at a good level.

In addition, in this embodiment, an optimization method is adopted in which the accuracy is improved by optimizing the calculation process so that the voltage compensation due to current control is not affected when performing integral calculation of the voltage command value immediately after the current level is switched in no-load current identification. Furthermore, in this embodiment, the current control response is set to a gain setting with no overshoot, and the voltage compensation value due to current control is prevented from becoming a value higher than necessary, thereby improving the accuracy of no-load current identification.

The no-load current is also the rated magnetic flux current at the applied voltage, and it changes due to the influence of magnetic saturation characteristics depending on the applied voltage value. Therefore, in the method of identifying the no-load current of this embodiment, by identifying the no-load current according to the applied voltage, i.e., the set magnetic flux level, it is also possible to identify the magnetic saturation characteristics of the no-load current.

According to this embodiment, even if the induction motor 2 to be driven is a motor whose electrical constants are unknown, the above-mentioned identification method or the like can be used to accurately obtain electrical information about the induction motor 2, such as slippage, while the induction motor 2 is in operation using sensorless vector control technology, resulting in the excellent effect of enabling highly accurate sensorless vector control.

Other Embodiments
The present invention is not limited to the embodiments described above and shown in the drawings, and can be modified, combined, or expanded in any manner without departing from the spirit and scope of the present invention.
The numerical values and the like shown in the above embodiment are merely examples and are not intended to be limiting.

Although several embodiments of the present invention have been described above, these embodiments are presented as examples and are not intended to limit the scope of the invention. These novel embodiments can be embodied in various other forms, and various omissions, substitutions, and modifications can be made without departing from the gist of the invention. These embodiments and their modifications are included in the scope and gist of the invention, and are included in the scope of the invention and its equivalents as set forth in the claims.

In the drawing, 1 and 21 are induction motor drive devices, 2 is an induction motor, 6 is a power converter, 8U and 8W are current detection units, 10 is a current adjustment unit, 11 and 23 are motor constant identification units, 12 is a phase-to-phase voltage detection unit, 15 is a current command value generation unit, and 16 is a voltage command generation unit.

Claims (10)

a power converter that converts a DC voltage into a three-phase AC voltage with variable voltage and frequency and supplies the AC voltage to an induction motor;
A phase-to-phase voltage detection unit that detects phase-to-phase voltages between two or more phases of the induction motor;
A current detection unit that detects a current of the induction motor;
a current adjustment unit that calculates a voltage adjustment value so that a current detection value, which is a value obtained by converting the detection value of the current onto the same control axis as the current command value, coincides with the current command value;
a current command value generating unit that generates the current command value and provides the current command value to the current adjusting unit;
a voltage command generating unit that generates a voltage command value by using the voltage adjustment value obtained from the current adjusting unit;
a motor constant identification unit that identifies an electrical constant of the induction motor based on the detected value of the interphase voltage or the voltage command value and the current command value or the current detected value;
Equipped with
The electric constants identified by the motor constant identification unit include identified values of a primary resistance, a leakage inductance, a no-load current, and a secondary time constant,
the current command value generating unit generates the current command value having a preset magnitude in a state in which a rotor of the induction motor is stopped;
After a preset time has elapsed and the output of the power converter is cut off,
The induction motor drive device, wherein the motor constant identification unit identifies the no-load current and the secondary time constant based on the detected value of the phase-to-phase voltage. a power converter that converts a DC voltage into a three-phase AC voltage with variable voltage and frequency and supplies the AC voltage to an induction motor;
A current detection unit that detects a current of the induction motor;
a current adjustment unit that calculates a voltage adjustment value so that a current detection value, which is a value obtained by converting the detection value of the current onto the same control axis as the current command value, coincides with the current command value;
a current command value generating unit that generates the current command value and provides the current command value to the current adjusting unit;
a voltage command generating unit that generates a voltage command value by using the voltage adjustment value obtained from the current adjusting unit;
a motor constant identification unit that identifies an electrical constant of the induction motor based on the voltage command value, and the current command value or the current detection value;
Equipped with
The electric constants identified by the motor constant identification unit include identified values of a primary resistance, a leakage inductance, a no-load current, and a secondary time constant,
the current command value generating unit generates the current command values having two different magnitudes that are set in advance in a state in which a rotor of the induction motor is stopped;
an induction motor drive device in which the motor constant identification unit identifies the no-load current and the secondary time constant based on the voltage command value after the current command value generation unit changes the current command value from a high level to a low level. the current command value generating unit provides a first current command value i _{sdref_[1]} to the current adjusting unit for a predetermined first waiting time;
After a predetermined second waiting time has elapsed since the output of the power converter is cut off,
The motor constant identification unit calculates a magnetic flux [1] shown in the following formula while a value v _sdmeas obtained by converting the detection value of the phase-to-phase voltage onto the same control axis as the current command value is below 0,
Magnetic flux [1] = ∫ (-v _{sd m ea s} ) dt
A target magnetic flux is set to a magnetic flux amount obtained by subtracting a leakage magnetic flux amount calculated using the identified value of the leakage inductance from a rated primary magnetic flux of the induction motor,
Using the magnetic flux [1], a second current command value i _{sdref_[2]} calculated by the following formula is provided to the current adjusting unit;
i _{sdref_[2]} = i _{sdref_[1]} + (target flux - flux[1]) x (i _{sdref_[1]} ÷ flux[1])
Calculate the magnetic flux [2] in the same manner as when calculating the magnetic flux [1],
When N is a natural number greater than or equal to 3,
An N-th current command value i _{sdref_[N]} calculated by the following formula is provided to the current adjusting unit,
i _{sdref_[N]} = i _{sdref_[N-1]} + (target flux - flux[N-1]) x (i _{sdref_[N-1]} - _{i sdref_[N-2]} ) ÷ (flux[N-1] - flux[N-2])
Calculate the magnetic flux [N] in the same manner as when calculating the magnetic flux [1].
By repeating these operations to perform a convergent calculation of i _{sdref_[N]} , the identified value i _{sdn_calc} of the no-load current is obtained as shown in the following equation:
i _{sdn_calc} = i _{sdref_[N]}
The induction motor driving device of claim 1, wherein the Nth current command value i _{sdref_ [N] is again provided to the current adjustment unit as the N+1th current command value i sdref_} [N+1], and when calculating the magnetic flux [N+1] in the same manner as when the magnetic flux [1] was calculated, a measurement operation is performed to measure the time T _{r_est} until the magnetic flux [N+1] becomes approximately (1-e ^-1 ) times the target magnetic flux, and the measurement operation is performed one or more times, and the measured value of the time T _{r_est} or the average value of the multiple measured times T _{r_est} is set as the identified value T _{r_calc} of the secondary time constant. The current command value generating unit provides the current adjusting unit with a base current command value i _{sdref_base} , which is a predetermined relatively small current value in a state in which the induction motor is stopped, for a predetermined first waiting time, and then
The motor constant identification unit stores the voltage command value v _{sdref_base} that has become a steady-state value,
Thereafter, the current command value generating unit provides the current adjusting unit with a first current command value i _{sdref_[1]} that is larger than the base current command value i _{sdref_base} for only the first waiting time, and then switches back to the base current command value i _{sdref_base} . Then, after a predetermined second waiting time has elapsed,
The motor constant identification unit calculates a magnetic flux [1] shown in the following formula while the voltage command value v _sdref is below the voltage command value v _{sdref_base} , which is a steady value when the base current command value i _{sdref_base} is applied:
Magnetic flux [1] = ∫ (v _{sdref_base} - v _sdref ) dt
A target magnetic flux is set to a magnetic flux amount obtained by subtracting a leakage magnetic flux amount calculated using the identified value of the leakage inductance from a rated primary magnetic flux of the induction motor,
A second current command value i _{sdref_[2]} calculated by the following formula is provided to the current adjusting unit,
i _{sdref_[2]} = i _{sdref_[1]} + (target flux - flux[1]) x (i _{sdref_[1]} ÷ flux[1])
Calculate the magnetic flux [2] in the same manner as when calculating the magnetic flux [1],
When N is a natural number greater than or equal to 3,
An N-th current command value i _{sdref_[N]} calculated by the following formula is provided to the current adjusting unit,
i _{sdref_[N]} = i _{sdref_[N-1]} + (target flux - flux[N-1]) x (i _{sdref_[N-1]} - _{i sdref_[N-2]} ) ÷ (flux[N-1] - flux[N-2])
Calculate the magnetic flux [N] in the same manner as when calculating the magnetic flux [1].
By repeating these operations to perform a convergent calculation of i _{sdref_[N]} , the identified value i _{sdn_calc} of the no-load current is obtained as shown in the following equation:
i _{sdn_calc} = i _{sdref_[N]} - i _{sdref_base}
The induction motor driving device of claim 2, wherein the Nth current command value i _{sdref_ [N] is again provided to the current adjustment unit as the N+1th current command value i sdref_} [N+1], and when calculating the magnetic flux [N+1] in the same manner as when the magnetic flux [1] was calculated, a measurement operation is performed to measure the time T _{r_est} until the magnetic flux [N+1] becomes approximately (1-e ^-1 ) times the target magnetic flux, and the measurement operation is performed one or more times, and the measured value of the time T _{r_est} or the average value of the multiple measured times T _{r_est} is set as the identified value T _{r_calc} of the secondary time constant. The motor constant identification unit is
When calculating the magnetic flux [1], the magnetic flux [2], the magnetic flux [N], and the magnetic flux [N+1],
switching the current command value to the base current command value i _{sdref_base} , and calculating, from a time change amount of the voltage command value observed after the second waiting time has elapsed, a magnetic flux change amount in the induction motor during the second waiting time as a magnetic flux correction amount;
5. The induction motor drive device according to claim 4, wherein the magnetic flux correction amount is added to the magnetic flux amounts of the magnetic flux [1], the magnetic flux [2], the magnetic flux [N] and the magnetic flux [N+1]. The motor constant identification unit is
the target magnetic flux can be selected from at least three values: a rated magnetic flux amount obtained by subtracting a leakage magnetic flux amount calculated using the identified value of the leakage inductance from a rated primary magnetic flux of the induction motor; a magnetic flux amount larger than the rated magnetic flux amount; and a magnetic flux amount smaller than the rated magnetic flux amount;
6. The induction motor drive device according to claim 3, wherein the magnetic saturation characteristics of the induction motor are obtained by identifying the no-load current when each of the three target magnetic fluxes is selected. before performing the operation for obtaining the identified value of the no-load current and the identified value of the secondary time constant,
The current command value generating unit provides a current command value i _sdref0 having a predetermined magnitude to the current adjusting unit for a predetermined time T0,
After the predetermined time T0 has elapsed, the motor constant identification unit calculates an identified value R s_calc of a primary resistance using the voltage command value v _sdref0 , which is the voltage command value that has become a steady value, and the current command value i _sdref0 or the current detection value i _sd0 , which is the current detection value that has become a steady _value ;
the voltage command generating unit calculates the identified value R of the primary resistance, and _sets , as the voltage command value, a value obtained by adding the voltage command value _{v_sdref0} and a sine wave having an amplitude obtained by multiplying the voltage command value _{v_sdref0} by a coefficient _Kd that is a value smaller than 1;
The coefficient _Kd is adjusted so that the amplitude of the AC component of the observed current is equal to a predetermined value;
the motor constant identification unit calculates an identified value L _{f_calc} of the leakage inductance from the AC amount of the current value observed by the current detection unit, the adjusted coefficient K _d , and the voltage command value v _sdref0 ;
The current adjusting unit includes a proportional integral compensator that performs PI control,
The induction motor drive device according to claim 3 , wherein the identified value R _{s_calc} of the primary resistance and the identified value L _{f_calc} of the leakage inductance are used to calculate set values of a proportional gain and an integral gain of the proportional-integral compensator. before performing the operation for obtaining the identified value of the no-load current and the identified value of the secondary time constant,
performing an operation for acquiring an identified value R _{s_calc} of the primary resistance and an identified value L _{f_calc} of the leakage inductance;
When a current response angular frequency is ω _curr , the values of the proportional gain K _p and the integral gain K _i in the proportional-integral compensator are set using the identified value R _{s_calc} of the primary resistance and the identified value L _{f_calc} of the leakage inductance, as shown in the following equation: K _p =ω _curr
K _i = ω _curr × R _{s_calc} ÷ L _{f_calc}
8. An induction motor drive system according to claim 7. In a case where the phase-to-phase voltage detection unit is not provided, when the current command value used in an operation for acquiring the identified value _{Rs_calc} of the primary resistance and the identified value _{Lf_calc} of the leakage inductance is transitioned to the base current command value _{i_sdref_base} , which is an initial current command value in identifying the no-load current, the voltage command value is observed to measure a coarse adjustment value _{Tr_temp} of the secondary time constant;
In a case where the phase-to-phase voltage detection unit is provided, after acquiring the identified value _{Rs_calc} of the primary resistance and the identified value _{Lf_calc} of the leakage inductance, the power converter measures a coarse adjustment value _{Tr_temp} of the secondary time constant by observing the voltage detection value after temporarily cutting off an output of the power converter;
8. The induction motor drive device according to claim 7, wherein, when a current response angular frequency is ω _curr , the first waiting time is set to about 5 to 10 times the coarse adjustment T _{r_temp} of the secondary time constant, and the second waiting time is set to about 2 to 5 times the reciprocal of the current response angular frequency ω _curr . In a case where the phase-to-phase voltage detection unit is not provided, when the current command value used in an operation for acquiring the identified value _{Rs_calc} of the primary resistance and the identified value _{Lf_calc} of the leakage inductance is transitioned to the base current command value _{i_sdref_base} , which is an initial current command value in identifying the no-load current, the voltage command value is observed to measure a coarse adjustment value _{Tr_temp} of the secondary time constant;
In a case where the phase-to-phase voltage detection unit is provided, after acquiring the identified value _{Rs_calc} of the primary resistance and the identified value _{Lf_calc} of the leakage inductance, the power converter measures a coarse adjustment value _{Tr_temp} of the secondary time constant by observing the voltage detection value after temporarily cutting off an output of the power converter;
The induction motor drive device according to claim 8, wherein the first waiting time is set to about 5 to 10 times the coarse adjustment T _{r_temp} of the secondary time constant, and the second waiting time is set to about 2 to 5 times the reciprocal of the current response angular frequency ω _curr .

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