JP2024038793A - Material property measuring device - Google Patents

Abstract

[Problem] A material property measuring device that measures the material properties of a material to be measured by irradiating the material with electromagnetic waves and receiving and analyzing the transmitted electromagnetic waves, which eliminates the need for a VNA and eliminates DC offset. Eliminates the need for calibration for removal. SOLUTION: A transmitting antenna 4 and a receiving antenna 7 are arranged side by side, and a reflecting plate 5 whose piston moves linearly in front is arranged. The electromagnetic waves reflected by the reflector 5 are received and subjected to homodyne detection by the mixer 9 to obtain a Doppler signal. The relative dielectric constant and tan δ of the sample 6 are calculated from the amplitude and phase of the Doppler signal when the sample 6 is not in front of the receiving antenna 7 and when it is present. [Selection diagram] Figure 1

Description

The present invention relates to a material property measuring device that measures the material properties of a material to be measured by irradiating the material with electromagnetic waves and receiving and analyzing the electromagnetic waves that have passed through the material.

The free space method is a method for measuring material properties using electromagnetic waves in the microwave and millimeter wave bands. In the free space method, when the receiving antenna receives electromagnetic waves radiated from the transmitting antenna, a material under test (hereinafter sometimes referred to as a DUT: device under test) is placed between the transmitting antenna and the receiving antenna. This method calculates the dielectric constant and tan δ (loss angle) as material properties of the sample from the measured values of the amplitude and phase of electromagnetic waves transmitted or reflected by the sample. There are devices described in Patent Documents 1 and 2 as material property measuring devices using the free space method.

Furthermore, as a material property measuring device using the free space method, there is one shown in FIG. 5, although it is not described in the literature. This material property measuring device includes a signal generator 1, a power divider 2, a transmitting antenna 4, a receiving antenna 7, a quadrature detector 11, and DC amplifiers 12 and 13. Here, the signal generator 1, power divider 2, quadrature detector 11, and DC amplifiers 12 and 13 are basic circuits for measuring the amplitude and phase of the electromagnetic waves transmitted through the sample 6 (a frequency conversion circuit, etc. omission).

In FIG. 5, a part of the high frequency signal in the microwave/millimeter wave band generated by the signal generator 1 is distributed by the power divider 2 as a high frequency signal for transmission, and the microwave/millimeter wave band is transmitted from the transmitting antenna 4. is emitted as electromagnetic waves. This electromagnetic wave passes through the sample 6 and is received by the receiving antenna 7, becomes a high frequency received signal, and is input to the quadrature detector 11. The high frequency signal generated by the signal generator 1 and distributed by the power divider 2 is input to the quadrature detector 11 as a reference signal, and the high frequency received signal is quadrature detected and outputs an I signal and a Q signal. , the I signal and the Q signal are converted into DC by the DC amplifiers 12 and 13.

Here, a vector network analyzer (VNA) is used as a means to measure the amplitude and phase of the electromagnetic waves that have passed through the sample 6, and the amplitude and phase are determined from S21 (transmission coefficient) of the S (Scattering) parameter. However, the basic circuit of the VNA for determining S21 (frequency conversion circuit etc. is omitted) includes a quadrature detector 11 and DC amplifiers 12 and 13, as shown in FIG.

Japanese Patent Application Publication No. 2006-133048 JP 2018-13452 Publication

However, since the quadrature detector 11 and the DC amplifiers 12 and 13 in the basic circuit shown in FIG. 5 handle DC, there is a DC offset. Therefore, it is necessary to perform calibration before starting measurement to remove the DC offset. This is because the S parameter S21 determines the received signal as a vector consisting of amplitude and phase, so the DC offset must be 0. Furthermore, the quadrature detector 11 has two channels of output, the I signal and Q signal, which are orthogonal to each other, and is detected as a vector, but it is difficult to obtain accurate amplitude and 90° phase difference for the I signal and Q signal. Therefore, since errors are included, there is a problem that these errors become errors in measuring the material properties (relative permittivity, tan δ) of the sample 6.

The present invention was made to solve such problems, and its purpose is to irradiate the material to be measured with electromagnetic waves and receive and analyze the electromagnetic waves that have passed through the material. The purpose of this technology is to eliminate the need for a VNA and the need for calibration to remove DC offset in a material property measurement device that measures material properties.

The present invention provides a signal generating means for generating a high frequency signal, a transmitting means for transmitting the high frequency signal as an electromagnetic wave, a reflecting means for reflecting the electromagnetic wave transmitted by the transmitting means while moving in a straight line, and a reflecting means for reflecting the electromagnetic wave by the reflecting means. a receiving means for receiving the received electromagnetic wave and outputting a high-frequency received signal; a signal acquisition means for homodyne-detecting the high-frequency received signal with the high-frequency signal to obtain a Doppler signal; the reflecting means; and the transmitting means or the receiving means. Calculation means for calculating relative permittivity and tan δ as material properties of the material to be measured based on the amplitude and phase of the Doppler signal when the material to be measured is placed between the material and the material to be measured. It is a measuring device.

According to the present invention, a VNA is no longer necessary in a material property measuring device that measures the material properties of a material to be measured by irradiating the material to be measured with electromagnetic waves and receiving and analyzing the transmitted electromagnetic waves. , calibration to remove DC offset becomes unnecessary.

1 is a diagram showing the configuration of a material property measuring device according to a first embodiment of the present invention. FIG. 3 is a diagram showing a sinusoidal Doppler signal obtained by homodyne detection of a received reflected wave. FIG. 3 is a diagram for explaining a method for determining the dielectric constant of a sample from a sinusoidal Doppler signal. FIG. 3 is a diagram showing the configuration of a material property measuring device according to a second embodiment of the present invention. 1 is a diagram showing the configuration of a conventional material property measuring device including a basic circuit for determining S21 of an S parameter using a VNA.

Embodiments of the present invention will be described in detail below with reference to the drawings.
[First embodiment]
<Configuration and general operation of material property measuring device>

FIG. 1 is a diagram showing the configuration of a material property measuring device according to a first embodiment of the present invention (hereinafter referred to as a material property measuring device according to the first embodiment). In this figure, the same reference numerals as in FIG. 5 are given to the same or corresponding components as in FIG. 5.

The material property measuring device according to the first embodiment includes a signal generator 1, a power divider 2, an isolator 3, a transmitting antenna 4, a reflector 5, a receiving antenna 7, an isolator 8, a mixer 9, and a DC amplifier 10. There is. Here, the signal generator 1, the transmitting antenna 4, the reflecting plate 5, the receiving antenna 7, the mixer 9, and the DC amplifier 10 are respectively a signal generating means, a transmitting means, a reflecting means, a receiving means, and a signal acquiring means according to the present invention. Equivalent to.

First, the positional relationship among the transmitting antenna 4, receiving antenna 7, and reflector 5 will be explained.
The transmitting antenna 4 and the receiving antenna 7 are arranged side by side. That is, they are arranged at the same position in the front-back direction (horizontal direction in the figure) and at the same position (that is, the same height) in the up-down direction (vertical direction). However, this is just an example, and the position in the front-rear direction and the position in the up-down direction may be different.

The reflecting plate 5 has its surface (reflecting surface) arranged parallel to a vertical plane, and can be piston-moved in the front-rear direction by a reflecting plate driving mechanism (not shown).

Further, a sample 6 is placed between the reflecting plate 5 and the receiving antenna 7. The sample 6 is plate-shaped, and its surface (electromagnetic wave transmission surface) is arranged parallel to the vertical plane.

The distance R between the transmitting antenna 4 and the receiving antenna 7 and the reflecting plate 5 is a distance at which a plane wave is established. The distance R satisfies the conditions for a plane wave, that is, the conditions for a far electromagnetic field, and is determined by the aperture size of the antenna, the size of the reflector, etc.

A signal generator 1 generates a high frequency signal (hereinafter referred to as a high frequency signal) in a microwave band or a millimeter wave band, and supplies it to a power divider 2 . Power divider 2 distributes the high frequency signal supplied from signal generator 1 to isolator 3 and mixer 9. The isolator 3 supplies the high frequency signal supplied from the power divider 2 to the transmitting antenna 4 with low loss. The transmitting antenna 4 radiates the high frequency signal supplied from the isolator 3 into free space as electromagnetic waves.

The reflecting plate 5 moves linearly (moves straight at a constant speed) in the front-back direction (left-right direction in the figure) and reflects the electromagnetic waves radiated from the transmitting antenna 4. The receiving antenna 7 receives the electromagnetic waves reflected by the reflector 5 and transmitted through the sample 6 when the sample 6 is present, and directly receives the electromagnetic waves reflected by the reflector 5 when the sample 6 is not present, and receives the high-frequency received signal. Supplied to isolator 8.

Isolator 8 supplies the high frequency reception signal supplied from reception antenna 7 to mixer 9 with low loss. The mixer 9 (DBM: double balance mixer here) uses the high frequency signal supplied from the power divider 2 as a reference signal (local oscillation signal) to homodyne-detect the high frequency received signal supplied from the isolator 8, and detects the high frequency received signal supplied from the isolator 8. Output to.

The sinusoidal Doppler signal that is the output of the DC amplifier 10 is input to a calculation device (such as a personal computer) not shown, and the relative dielectric constant and tan δ are calculated as material properties of the sample 6 using a predetermined calculation formula. This calculation formula will be described later.

FIG. 2 is a diagram showing the waveform of a sinusoidal Doppler signal. Here, the sinusoidal waveform represented by the dotted line is the waveform of the Doppler signal 14 (referred to as V1) when the sample 6 is not present, and the sinusoidal waveform represented by the solid line is the waveform when the sample 6 is present. This is the waveform of Doppler signal 15 (assumed to be V2). The amplitude A of the Doppler signal 14 and the amplitude B of the Doppler signal 15 are irrelevant even if there is a DC offset. Further, the phase is relatively determined from the time difference ΔTd between the Doppler signal 14 when the sample 6 is not present and the Doppler signal 15 when the sample 6 is present. In other words, the amplitude and phase difference of the Doppler signal when the sample 6 is absent and when the sample 6 is present are both determined relatively, and are therefore unrelated to the presence of a DC offset between the mixer 9 and the DC amplifier 10. Therefore, no calibration is required to remove the DC offset. Furthermore, unlike the material property measuring device shown in FIG. 5, a quadrature detector is not required, and therefore a VNA is not required.

In this way, according to the material property measuring device according to the first embodiment, the homodyne detection can be performed using a simple mixer (DBM) 9, and therefore does not include errors caused by the quadrature detector 11. Furthermore, as shown in FIG. 2, due to the linear movement of the reflector 5, the amplitudes of the Doppler signals 14 and 15, which change in a sinusoidal manner when there is no sample 6 in front of the receiving antenna 7 and when there is a sample 6 in front of the receiving antenna 7, are changed by the amplitude ( Since the phase is determined by taking the phase difference between the sinusoidal Doppler signals 14 and 15, both the amplitude and the phase can be determined relatively. Therefore, since there is no relation to the DC offset of mixer 9 or DC amplifier 10, there is an effect that calibration is not necessary. Furthermore, since an expensive VNA is not required, the device can be realized at low cost.

<Detailed operation of material property measuring device>
Next, the detailed operation of the material property measuring device according to the first embodiment will be explained.

In FIG. 1, a high frequency signal with an angular frequency ω output from a signal generator 1 is divided into two parts by a power divider 2, one of which is supplied as a high frequency signal for transmission to a transmitting antenna 4 and radiated as an electromagnetic wave. This high-frequency signal for transmission is expressed as ν ₀ =cos(ωt+θ)...Equation [1]
First, when there is no sample 6, the electromagnetic waves radiated from the transmitting antenna 4 and reflected by the reflector 5 in front are received by the receiving antenna 7 with a delay of time τ ₁ (=2R/c), so the reception The high frequency received signal ν ₁ output from the antenna 7 is ν ₁ =Acos[ω(t-τ ₁ )+θ]=Acos(ωt-ωτ ₁ +θ)...Equation [2]
becomes. Here, A=amplitude, ω=2πf, f=frequency, θ=arbitrary phase, t=time, R=distance to the reflecting plate 5, and c=speed of light.

Therefore, the high-frequency signal distributed by the power divider 2 and the high-frequency received signal output from the receiving antenna 7 are homodyne-detected by the mixer 9, and the output V1 amplified by the DC amplifier 10 is obtained by formula [1] and formula [ 2] From,
V1=LPF[ν ₀ ×ν ₁ ]=LPF[cos(ωt+θ)×Acos(ωt-ωτ ₁ +θ)]=(A/2)cosωτ ₁
=(A/2)cos2β ₁ R...Formula [3]
becomes. Here, LPF means passing through a low-pass filter, β ₁ is a phase constant, and assuming that the wavelength of the signal from the signal generator 1 is λ, β ₁ =(2π/λ).

Therefore, from equation [3], when the distance R to the reflection plate 5 changes linearly with respect to time τ, the output V1 of the DC amplifier 10 becomes the sinusoidal Doppler signal 14 shown in FIG. 2.

Similarly, when a sample 6 with a thickness d and a relative permittivity ε _r is placed in front of the receiving antenna 7, the electromagnetic waves emitted from the transmitting antenna 4, reflected by the front reflector 5, and transmitted through the sample 6 are Since it is received with a delay of τ ₂ , the high-frequency received signal ν ₂ output from the receiving antenna 7 is ν ₂ =Bcos[ω(t-τ ₂ )+θ]...Equation [4]
becomes. Here, τ ₂ is expressed by the following formula [5]. As will be explained later, strictly speaking, ε _r is the complex dielectric constant, but in the case of tan δ ≦ 0.1, the square root of ε _r is √(ε _r ) and the square root of the real part ε _{′ r} of ε _r is √(ε ' _r ) can be considered to be approximately equal, so both ε _r and ε _{' r} are referred to as relative dielectric constants in this specification.

Therefore, the output V2 of the DC amplifier 10 is expressed by the following equation [6].

Here, B is the amplitude of the high frequency received signal ν ₂ .

Therefore, when the distance R to the reflection plate 5 changes linearly with time τ, the output V2 of the DC amplifier 10 can become the sinusoidal Doppler signal 15 shown in FIG. 2 according to equation [6]. I understand.

Therefore, the phase difference Δθ between V1 and V2 is expressed by the following equation [7] from equation [3] and equation [6].

However, β ₂ is a phase constant within the sample 6, and is expressed by the following equation [8].

The phase difference Δθ between the signal transmitted through the sample 6 having the thickness d and the signal transmitted through the free space when the sample 6 was not present was determined using equation [7].

Further, the ratio between the amplitude B of the signal transmitted through the sample 6 and the amplitude A of the signal when the sample 6 is not present is determined by Equation [3] and Equation [6], and becomes the following Equation [9].
|(V2/V1)|=B/A...Formula [9]

As described above, the reflecting plate 5 that moves linearly is placed in front of the transmitting antenna 4 and the receiving antenna 7, and from the Doppler signals 14 and 15 obtained by receiving the reflected waves of the reflecting plate 5, the sample is detected in front of the receiving antenna 7. It is possible to measure the phase difference and amplitude ratio when sample 6 is absent and when sample 6 is present. Then, the relative dielectric constant ε _r and tan δ of the sample 6 can be determined from the ratio of these phase differences and amplitudes.

<Sample transmission coefficient T>
Next, in the material property measuring device according to the first embodiment, the material property (relative dielectric constant, tan δ) of the sample 6 is determined by the phase difference between when the sample 6 is not in front of the receiving antenna 7 and when the sample 6 is present. This shows that it can be calculated from Equation [7] expressing the amplitude ratio and Equation [9] expressing the amplitude ratio.

In FIGS. 1 and 2, the level of the Doppler signal 15 (=output V2 of the DC amplifier 10) when the sample 6 is present is proportional to the transmission coefficient T of the sample 6, so
V2=kT...Formula [10]
It is expressed as Here, the proportionality constant k is proportional to the transmission output, the reflection intensity of the reflection plate 5, and the like.

On the other hand, if the electromagnetic waves transmitted from the transmitting antenna 4 are flat, the transmission coefficient T of the plate-shaped sample 6 with a thickness d and a relative dielectric constant ε _r can be expressed by the following equation [11] by theoretical calculation. I know.

Here, r is the reflection coefficient of the sample 6, and is expressed by the following equation [12].

In addition, the dielectric constant of sample 6 is
ε _r =ε _r′ (1−jtanδ) …Formula [13]
It is a complex number represented by . In this equation, ε _r' is a real number, tan δ is a loss angle, and j is an imaginary unit.

Next, the level of the Doppler signal 14 (=output V1 of the DC amplifier 10) when the sample 6 is not present is:
V1=k...Formula [14]
Therefore, by taking the ratio with Equation [11], the transmission coefficient T of sample 6 can be obtained by the following Equation [15].

Here, when tanδ≦0.1, the square root √(ε _r ) of ε _r and the square root √(ε _r′ ) of ε _r′ can be considered to be approximately equal, so
β′ ₂ = (2π/λ)√(ε _r′ ) …Formula [16]
Then, when formula [15] is rewritten, the following formula [17] is obtained.

Here, η', r', and β ₁ are expressed by the following equations [18], [19], and [20], respectively.

Furthermore, when formula [13] is applied as ε _r in formula [17], the following formula [21] is obtained.

Therefore, equation [17] becomes equation [22] below.

Here, C and D are represented by the following formulas [23] and [24], respectively.

<Calculation of relative permittivity ε _r′ of sample 6>
In the phase of the sinusoidal Doppler signals 14 and 15 obtained by receiving the reflected signals of the reflecting plate 5 that moves linearly in front of the transmitting antenna 4 and the receiving antenna 7, The phase difference Δθ when sample 6 is present is given by one period Td of the Doppler signal and the time difference ΔTd.
Δθ=2π(ΔTd/Td)...Formula [25]
is required. Since this value is equal to the phase of equation [22], it can be expressed as equation [26] below.

Here, if tan δ≦0.1, C is approximately 1, so the right side of equation [26] is a function of ε _r′ . Therefore, as shown in FIG. 3, the relative dielectric constant ε _r ' of the sample 6 can be determined by giving various ε _r' and determining the value that corresponds to the measured value on the left side.

<Calculation of tanδ of sample 6>
When the absolute value of equation [22] is taken, the following equation [27] is obtained.

Here, since D is a function of tan δ according to equation [24], the following equation [28] can be obtained by solving equation [27] for tan δ.

Here, |(V2/V1)| is the level ratio of the sinusoidal outputs V1 and V2 in FIG. 2, and is a measured value.

Note that the sample 6 placed in front of the receiving antenna 7 has the same effect even when placed in front of the transmitting antenna 4, and may be placed between the transmitting antenna 4 and the receiving antenna 7. Further, it is preferable that the sample 6 be placed vertically, but if the inclination is within 3 degrees, the error will be within 1% since cos3 degrees = 0.9986, which is within the permissible range. Further, it is preferable that the measurement environment be in an anechoic chamber or an anechoic box.

[Second embodiment]
The material property measuring device according to the first embodiment receives the reflected wave from the reflecting plate 5 that moves linearly, and measures the phase and amplitude of the transmitted wave of the sample 6 from the Doppler signal accompanying the linear movement of the reflecting plate 5. The method is characterized in that the relative dielectric constant and tan δ of the sample 6 are calculated using the method, but a plane wave is required as a measurement condition. Therefore, the distance R between the antenna and the reflector must satisfy the conditions for a plane wave, that is, the conditions for a far electromagnetic field. Furthermore, it is desirable that the measurement environment be in an anechoic chamber or an anechoic box. Therefore, when the frequency is low, it becomes difficult to secure a distance R at which a plane wave is established. The material property measuring device according to the second embodiment of the present invention can solve this problem.

FIG. 4 is a diagram showing the configuration of a material property measuring device according to a second embodiment of the present invention. In this figure, the same or corresponding components as in FIG. 1 are given the same reference numerals as in FIG. 1.

Whereas the material property measuring device according to the first embodiment propagates electromagnetic waves in free space, the material property measuring device according to the second embodiment of the present invention (hereinafter referred to as the material property measuring device according to the second embodiment) ) is fundamentally different in that electromagnetic waves are propagated within a waveguide.

As shown in the figure, the material property measuring device according to the second embodiment includes a signal generator 1, a power divider 2, a mixer 9, and a DC amplifier 10, like the material property measuring device according to the first embodiment. . These configurations and functions are the same as the components with the same names in the material property device according to the first embodiment.

Further, the material property measuring device according to the second embodiment includes a transmitting side coaxial waveguide converter 17, a transmitting side isolator 18, a circulator 19, a receiving side isolator 20, a receiving side coaxial waveguide converter 21, a first guide It includes a wave tube 22, a second wave guide 23, a third wave guide 24, a fourth wave guide 25, and a fifth wave guide 26. Further, the material property measuring device according to the second embodiment is configured such that the reflection plate 5 attached to the reflection plate support 27 of the reflection plate drive mechanism is capable of linear piston movement within the third waveguide 24. and the sample 6 is inserted into the fourth waveguide 25.

In the material property measuring device according to the second embodiment, an electromagnetic wave is injected from the coaxial waveguide converter 17 into the first waveguide 22, passes through the transmitting side isolator 18, and is transferred to the second waveguide 23 of the circulator 19. 1 port and is supplied into the third waveguide 24 connected to the second port of the circulator 19. The electromagnetic waves supplied into the third waveguide 24 are reflected by the reflection plate 5, and are supplied from the third port of the circulator 19 into the fourth waveguide 25 into which the sample 6 is inserted, and after passing through the sample 6. , the receiving side isolator 20, and the fifth waveguide 26, and is extracted as a high frequency reception signal by the receiving side coaxial waveguide converter 21. If this high frequency reception signal is subjected to homodyne detection by the mixer 9 and a Doppler signal is extracted by the DC amplifier 10, the apparatus operates equivalently to the material property measuring apparatus according to the first embodiment. Here, even if the sample 6 inserted into the fourth waveguide 25 is inserted into the first waveguide 22 which is placed in front of the isolator (transmission side isolator 18) like the fourth waveguide 25, The effect of the present invention is the same.

Note that in the material property measuring devices according to the first and second embodiments described above, the frequencies handled are microwave bands and millimeter wave bands, but the present invention can also be applied to electromagnetic waves up to the terahertz band.

DESCRIPTION OF SYMBOLS 1... Signal generator, 2... Power divider, 3, 8, 18, 20... Isolator, 4... Transmission antenna, 5... Reflection plate, 6... Sample (DUT), 7... Receiving antenna, 9... Mixer, 10... DC amplifier, 14... Sinusoidal Doppler signal when there is no sample, 15... Sinusoidal Doppler signal when sample is present, 16... Relationship curve of measured phase difference with relative dielectric constant, 17... Transmitting side coaxial waveguide Converter, 18... Transmission side isolator, 19... Circulator, 20... Receiving side isolator, 21... Receiving side coaxial waveguide converter, 22 to 26... First to fifth waveguides, 27... Reflection plate drive mechanism. Reflector support.

Claims (3)

a signal generating means for generating a high frequency signal;
Transmitting means for transmitting the high frequency signal as an electromagnetic wave;
a reflecting means that reflects the electromagnetic waves transmitted by the transmitting means while moving in a straight line;
receiving means for receiving the electromagnetic waves reflected by the reflecting means and outputting a high frequency reception signal;
signal acquisition means for homodyne-detecting the high-frequency received signal with the high-frequency signal to obtain a Doppler signal;
Based on the amplitude and phase of the Doppler signal when the material to be measured is placed and not placed between the reflecting means and the transmitting means or the receiving means, the relative dielectric constant and tan δ are determined as material properties of the material to be measured. A calculation means for calculating,
A material property measuring device with The material property measuring device according to claim 1,
The transmitting means transmits electromagnetic waves to free space, the receiving means receives electromagnetic waves from free space, and the reflecting means and the material to be measured are arranged in free space. The material property measuring device according to claim 1,
The transmitting means transmits an electromagnetic wave into a waveguide, the receiving means receives the electromagnetic wave from the waveguide, and the reflecting means and the material to be measured are arranged in the waveguide.

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