JP2023183034A - Control circuit and power conversion device - Google Patents

Abstract

To provide a control circuit and a power conversion device capable of performing soft switching by three-level voltage control with practical calculation load.SOLUTION: A control circuit, for controlling a power conversion device including a transformer, a primary side three-level bridge circuit, and a secondary side bridge circuit, includes: a feedback circuit for performing feedback control of a phase difference φ based on a given current command value and a state value of the transformer; a timing calculation circuit for calculating two phases α and β determining switching timing for each switching element of the primary side three-level bridge circuit, where 0<α<β<π, for the current command value according to each relational expression; and a PWM signal generation circuit for generating a PWM signal for controlling switching elements of the primary side three-level bridge circuit and the secondary side bridge circuit using the phase difference φ, and the phases α and β.SELECTED DRAWING: Figure 5

Description

This disclosure relates to a control circuit and a power conversion device.

As a power converter, a DAB (Dual Active Bridge) isolated converter using a full bridge circuit on the primary side of a transformer is known. It is known that in such a converter, when the step-up ratio is large or when the transmitted power is small, switching becomes so-called hard switching, resulting in large losses.

One proposal for solving these problems is disclosed in Non-Patent Document 1 listed below. The technology disclosed in Non-Patent Document 1 relates to a DAB type converter in which the voltage applied to the transformer is set to three levels instead of two levels. According to Non-Patent Document 1, by setting the voltage to three levels and appropriately determining the parameters, it is possible to avoid hard switching and realize soft switching. In the following explanation, when explaining the configuration of the converter, "3 levels" will be indicated as "3L" and "2 levels" will be indicated as "2L" for the purpose of simplifying the description. There is.

A. Filba-Martinez, S. Busquets-Monge, J. Nicolas-Apruzzese and J. Bordonau, "Operating Principle and Performance Optimization of a Three-Level NPC Dual-Active-Bridge DC-DC Converter," in IEEE Transactions on Industrial Electronics, vol. 63, no. 2, pp. 678-690, Feb. 2016.

However, the technique disclosed in Non-Patent Document 1 generates three levels of voltage to be applied to the transformer. Therefore, the number of semiconductor switching elements in the primary circuit to be controlled increases, and the number of parameters for controlling them also increases. The technique disclosed in Non-Patent Document 1 proposes feedback control of these parameters. Although that proposal performs feedback control after imposing various conditions, there is a problem in that the computational load is high due to the large number of parameters to be controlled. As a result, the required functions cannot be realized depending on the processing power and calculation speed of the microprocessor used in a typical power converter, making it impractical.

The purpose of this disclosure is to provide a control circuit and a power conversion device that can perform soft switching using three-level voltage control with a practical calculation load.

A control circuit according to a first aspect of this disclosure includes a transformer, a primary side three-level bridge circuit connected to the primary side of the transformer, and a secondary side bridge circuit connected to the secondary side of the transformer, Controls a power conversion device that performs power conversion by changing the phase difference φ between the voltage waveform that the primary side 3-level bridge circuit applies to the transformer and the voltage waveform that the secondary side bridge circuit applies to the transformer. The control circuit includes a feedback circuit that performs feedback control of the phase difference φ based on the given current command value and the state value of the transformer, and a feedback circuit that performs feedback control of the phase difference φ based on the given current command value and the state value of the transformer. Using a timing calculation circuit that calculates two phases α and β that determine the switching timing of each switching element of the side three-level bridge circuit, where 0<α<β<π, and a phase difference φ and phases α and β. It includes a PWM signal generation circuit that generates a PWM (Pulse Width Modulation) signal that controls switching elements of a primary side three-level bridge circuit and a secondary side bridge circuit.

A power conversion device according to a second aspect of this disclosure includes the control circuit, a transformer, and a semiconductor switching element whose switching is controlled by the control circuit, and includes three primary levels connected to the primary side of the transformer. It includes a bridge circuit and a secondary circuit connected to the next side of the transformer.

A power conversion device according to a third aspect of this disclosure includes the control circuit, a transformer, and a semiconductor switching element whose switching is controlled by a primary side PWM signal generation circuit of the control circuit. A secondary side 3-level bridge circuit connected to the secondary side of the transformer, including a connected primary side 3-level bridge circuit and a semiconductor switching element whose switching is controlled by a secondary side PWM signal generation circuit of the control circuit. including.

As described above, according to this disclosure, it is possible to provide a control circuit and a power conversion device that can perform soft switching using three-level voltage control with a practical calculation load.

FIG. 1 is a diagram for explaining loss due to hard switching. FIG. 2 is a circuit block diagram of a DAB type converter according to the first embodiment of this disclosure. FIG. 3 is a graph for explaining parameters related to three-level control in the first embodiment. FIG. 4 is a graph showing changes in the primary voltage, secondary voltage, and transformer current of the transformer in the first embodiment. FIG. 5 is a block diagram showing the configuration of the control circuit in the first embodiment. FIG. 6 is a graph showing the range of combinations of phase difference φ and transformation ratio m that can realize soft switching for various phases β. FIG. 7 is a diagram plotting transformation ratios at which soft switching can be realized for combinations of phase difference φ and phase β based on the relationship shown in FIG. FIG. 8 is a diagram plotting the maximum current value that allows soft switching for combinations of phase difference φ and phase β. FIG. 9 is a diagram that combines the area where soft switching can be realized at a desired transformation ratio in FIG. 7 and the graph in FIG. FIG. 4 is a diagram showing a state in which lines indicating combinations of phase β and phase β are drawn. FIG. 10 is a graph in which the line shown in FIG. 9 is graphed as the relationship between the current command value I and the phase β, and the graph is further approximated by a cubic curve. FIG. 11 is a block diagram showing the configuration of a control circuit according to the second embodiment. FIG. 12 is a diagram showing a method for determining the phase β when changing the transformation ratio based on the graph shown in FIG. 10. FIG. 13 is a block diagram showing the configuration of a control circuit according to the third embodiment. FIG. 14 is a block diagram showing the configuration of a control circuit using constant voltage control. FIG. 15 is a diagram showing simulation results comparing the range in which soft switching can be realized by the 3L-2L converter according to the present disclosure with a conventional 2L-2L converter. FIG. 16 is a diagram showing transformer voltage waveforms and transformer current waveforms in a simulation regarding a 2L-2L converter. FIG. 17 is a diagram showing gate-source voltage waveforms and current waveforms of a switching element in a simulation regarding a 2L-2L converter. FIG. 18 is a diagram showing transformer voltage waveforms and transformer current waveforms in a simulation regarding the 3L-2L converter according to this disclosure. FIG. 19 is a diagram showing gate-source voltage waveforms and current waveforms of a switching element in a simulation regarding a 3L-2L converter according to this disclosure. FIG. 20 is a circuit block diagram of a 3L-3L converter according to a modification of this disclosure. FIG. 21 is a block diagram showing the configuration of the control circuit shown in FIG. 20. FIG. 22 is a circuit block diagram of a 3L-2L converter according to another modification of this disclosure.

[Description of embodiments of the present disclosure]
In the following description and drawings, identical parts are provided with the same reference numerals. Therefore, detailed description thereof will not be repeated. Note that any features of one or more embodiments below may be combined.

(1) The control circuit according to the first aspect of this disclosure includes a transformer, a primary side three-level bridge circuit connected to the primary side of the transformer, and a secondary side bridge circuit connected to the secondary side of the transformer. A power conversion system that performs power conversion by changing the phase difference φ between the voltage waveform that the primary side three-level bridge circuit applies to the transformer and the voltage waveform that the secondary side bridge circuit applies to the transformer. A control circuit for controlling the device, which performs feedback control of the phase difference φ based on the given current command value and the state value of the transformer; , a timing calculation circuit that calculates two phases α and β that determine the switching timing of each switching element of the primary side three-level bridge circuit, where 0<α<β<π, and a timing calculation circuit that calculates the phase difference φ and the phases α and β. and a PWM signal generation circuit that generates a PWM signal that controls the switching elements of the primary side three-level bridge circuit and the secondary side bridge circuit using the PWM signal generation circuit.

Since the phases α and β are calculated by the relational expression with the current command value, the load for control is reduced. As a result, it is possible to provide a control circuit capable of soft switching by three-level voltage control with a practical calculation load.

(2) In (1) above, the calculation cycle of the two phases α and β by the timing calculation circuit is longer than the feedback cycle by the feedback circuit.

Since the calculation cycle of the phases α and β can be lengthened, the load for control is further reduced.

(3) In (1) or (2) above, the relational expression may include a first relational expression regarding the phase α and the current command value, and a second relational expression regarding the phase β and the current command value.

Phase α and phase β are calculated using separate relational expressions regarding the current command value. As a result, these phases can be calculated easily and independently of each other, reducing the control load.

(4) In (3) above, the first relational expression may be a function of the current command value that uniquely determines the phase α for a certain range of current command values.

Since the phase α is uniquely determined, control becomes simple.

(5) In the above (4), the first relational expression may be a strongly monotonically decreasing function of the current command value within a certain range, which determines the phase α.

A monotonically decreasing function in the strong sense defines a one-to-one relationship between the phase α and the current command value. As a result, the phase α can be uniquely determined by a simple calculation, which has the effect of simplifying control.

(6) In any one of (3) to (5) above, the second relational expression may be a function of the current command value that uniquely determines the phase β for a certain range of current command values. good.

Since the phase β is uniquely determined, control becomes simple.

(7) In the above (6), the second relational expression may be a strongly monotonically decreasing function of the current command value within a certain range, which determines the phase β.

By doing so, the relationship between the current command value and the phase β is determined in a one-to-one correspondence. As a result, the phase β can be uniquely determined by a simple calculation, which has the effect of simplifying the control.

(8) In any one of (1) to (7) above, each switching element of the primary side three-level bridge circuit establishes continuity between the first terminal, the second terminal, and the first terminal and the second terminal. The relational expression may be selected such that the switching by each switching element is a soft switch.

Since the switching by the switching element is soft switching, generation of power loss due to the switching element can be prevented.

(9) In (8) above, the relational expression indicates that when each switching element is not conducting, the direction of the current flowing through the body diode of each switching element is opposite to the voltage applied to the first terminal and the second terminal. The phase α and the phase β may be selected so that the switching element is conductive while the switching element is turned on.

When the switching element is turned on, the direction of current flowing through the body diode of the switching element is opposite to the voltage. Therefore, no power loss occurs when the switching element is turned on, and the efficiency of the power conversion circuit can be increased.

(10) In any one of (1) to (9) above, the primary side three-level bridge circuit applies 0, ±V1/2, and ±V1 to the primary side to the transformer. The phase α may determine the timing for changing the voltage applied to the primary side of the transformer from 0 to ±V1/2, and the phase obtained by subtracting the phase α from the phase π is the phase α of the transformer. The timing for changing the voltage applied to the primary side of the voltage from ±V1/2 to 0 may be determined.

By determining the phase α in this manner, the voltage applied to the primary side of the transformer is switched between three levels. Voltage changes due to switching of the switching element are reduced. Therefore, smaller switching elements can be used. As a result, switching by the switching element can be performed as soft switching for a wide range of transformation ratios while keeping costs low. As a result, loss in the power conversion device can be kept low while keeping costs low.

(11) In (10) above, the phase β may determine the timing at which the voltage applied to the primary side of the transformer is changed from ±V1/2 to ±V1, respectively, and the phase β is subtracted from the phase π. The phase may determine the timing at which the voltage applied to the primary side of the transformer is changed from ±V1 to ±V1/2.

By determining the phase β in this manner, the voltage applied to the primary side of the transformer is switched between three levels. Furthermore, voltage changes due to switching of each switching element are reduced. Therefore, smaller switching elements can be used. As a result, switching by the switching element can be performed as soft switching for a wide range of transformation ratios while keeping costs low. As a result, loss in the power conversion device can be kept low while keeping costs low.

(12) In any one of the above (1) to (11), the relational expression may be a polynomial of the current command value.

By employing a polynomial of the current command value as the relational expression, the phase α and the phase β can be calculated with a small amount of calculation. As a result, soft switching with three-level voltage control becomes possible with a practical calculation load.

(13) In the above (12), the relational expression may be a third-order or lower polynomial of the current command value.

By employing a polynomial of the third order or less of the current command value as the relational expression, the phase α and the phase β can be calculated with a very low amount of calculation. As a result, soft switching with three-level voltage control becomes possible with a practical calculation load.

(14) In any one of (1) to (13) above, the control circuit further generates a current command value by feedback based on the voltage command value and the voltage applied to the primary side of the transformer. The current command value generation circuit may also include a current command value generation circuit.

With this configuration, even under constant voltage control, by utilizing the constant current control configuration, soft switching with three levels of voltage control is possible with a practical calculation load.

(15) In any one of (1) to (14) above, the secondary side bridge circuit may include a secondary side 3-level bridge circuit, and the control circuit further controls each current command value. The secondary side timing calculation circuit may further include a secondary side timing calculation circuit that calculates two phases γ and δ that determine the switching timing of each switching element of the secondary side three-level bridge circuit, where 0<γ<δ<π, according to the relational expression. Typically, a PWM signal generation circuit controls the switching elements of a primary side three-level bridge circuit using a phase difference φ and phases α and β, and controls the switching elements of a secondary side three-level bridge circuit using a phase difference φ and phases γ and δ. It may also include a 3-level PWM signal generation circuit for controlling the switching elements of the level bridge circuit.

By adopting a three-level bridge circuit not only on the primary side but also on the secondary side, and determining the phases γ and δ necessary for its control from the relational expression with the current command value, it can be used over a wide range of transformation ratios. It is possible to provide a control circuit that can stably realize soft switching even when voltage fluctuations are high and that can operate with a practical calculation load.

(16) A power conversion device according to a second aspect of this disclosure includes the control circuit of any one of (1) to (15) above, a transformer, and a semiconductor switching element whose switching is controlled by the control circuit. , a primary side three-level bridge circuit connected to the primary side of the transformer, and a secondary side circuit connected to the secondary side of the transformer.

Since the phases α and β are calculated by the relational expression with the current command value, the load for control is reduced. As a result, it is possible to provide a power conversion device that enables soft switching by three-level voltage control with a practical calculation load and has reduced loss.

(17) A power conversion device according to a third aspect of this disclosure includes the control circuit of (15) above, a transformer, and a semiconductor switching element whose switching is controlled by a primary side PWM signal generation circuit of the control circuit. , connected to the secondary side of the transformer, including a primary side three-level bridge circuit connected to the primary side of the transformer, and a semiconductor switching element whose switching is controlled by a secondary side PWM signal generation circuit of the control circuit. and a secondary side three-level bridge circuit.

Phases α, β, γ, and δ are calculated using a relational expression with the current command value. Therefore, the load for control becomes smaller. As a result, it is possible to provide a power conversion device that enables soft switching by three-level voltage control with a practical calculation load and has reduced loss.

[Details of embodiments of the present disclosure]
A specific example of a converter according to an embodiment of the present disclosure will be described below with reference to the drawings. Note that the present disclosure is not limited to these examples, but is indicated by the scope of the claims, and is intended to include all changes within the meaning and scope equivalent to the scope of the claims.

1. First embodiment 1A. Background With reference to FIG. 1, the occurrence of loss due to hard switching will be explained. As shown in the voltage waveform 50 of FIG. 1, it is assumed that there is a voltage difference across the semiconductor switch when no current flows through the semiconductor switch. When the semiconductor switch is turned on in this state, a current begins to flow through the semiconductor switch as shown by a current waveform 52 before the voltage difference across the semiconductor disappears due to the capacitance component of the semiconductor switch. As a result, power loss 58 occurs in region 54 where both current and voltage are non-zero. This is also the case when the semiconductor switch is turned off, and power loss 60 occurs in the region 56 where both current and voltage are non-zero.

This embodiment is intended to prevent such power loss from occurring, and for this purpose, as described below, a 3L full bridge circuit is adopted on the primary side, and the switching timing of the semiconductor switch is adjusted. By controlling with a simple configuration, power loss is prevented within a wide range of transformation ratios and over a wide range of current values.

1B. Configuration FIG. 2 shows a schematic configuration of the 3L-2L converter 100 according to this embodiment. Referring to FIG. 2, the 3L-2L converter 100 includes a transformer 114 having a leakage inductance 116, and a primary side 3L full bridge circuit 110 connected between the storage battery 102 and the primary side terminal of the transformer 114. include. 3L-2L converter 100 further includes a secondary full-bridge circuit 112 connected to a secondary terminal of transformer 114 and DC bus 104.

The primary side 3L full bridge circuit 110 includes switching elements S1, S2, S3, S4, S5, S6, S7, and S8, all of which are MOSFETs (Metal-Oxide-Semiconductor Field Effect Transistors). Among these, switching elements S1 and S4, S5 and S8 form a full bridge circuit. Switching elements S2, S3, S6, and S7 form a switching element group 124 necessary for three-level control. A node 128 connecting switching elements S1 and S4 is connected to a first terminal on the primary side of transformer 114. A node 130 connecting switching elements S5 and S8 is connected to a second terminal on the primary side of transformer 114.

Capacitors 120 and 122 are connected in series via a node 126 between two terminals of the primary side 3L full bridge circuit 110 connected to the storage battery 102 side. Node 126 is connected to nodes 128 and 130 via separate connection lines. Switching elements S2 and S3 are connected in this order between node 126 and node 128. Switching elements S6 and 7 are connected in this order between node 126 and node 130.

Secondary full bridge circuit 112 includes switching elements S9, S10, S11, and S12. Switching elements S9 and S10 are connected via node 142. Switching elements S11 and S12 are connected via node 144. Node 142 is connected to a first terminal on the secondary side of transformer 114 . Node 144 is connected to a second terminal on the secondary side of transformer 114 .

Secondary full-bridge circuit 112 further includes a capacitor 140 connected between two DC buses 104.

The 3L-2L converter 100 further provides a primary 3L full-bridge circuit 110 and a secondary full-bridge circuit 112 based on the current IL of the transformer 114 and the voltages applied to the primary and secondary sides of the transformer 114. It includes a control circuit 250 for generating a control signal for each included switching element and outputting it to the primary side 3L full-bridge circuit 110 and the secondary side full-bridge circuit 112. The configuration of the control circuit 250 will be described later with reference to FIG.

With reference to FIG. 3, parameters for controlling the primary side 3L full-bridge circuit 110 will be described. In the upper part of FIG. 3, a voltage waveform 170 applied to the transformer 114 by the primary side 3L full-bridge circuit 110 is shown. A voltage waveform 172 applied to the secondary side of the transformer 114 by the secondary side full bridge circuit 112 is shown in the lower part of FIG.

As shown in the lower part of FIG. 3, in this embodiment, the voltage waveform 172 is a normal square wave, which alternates voltages of ±V2 with a constant period. In the following explanation, this one period is indicated by 2π. On the other hand, as shown in the upper part of FIG. 3, the voltage waveform 170 generated by the primary side 3L full-bridge circuit 110 takes voltages of ±V1, 1/2, and 0.

If the first phase of one cycle is 0, the voltage waveform 170 is zero from phase 0 to phase α (0<α<π/2). At phase α, voltage waveform 170 becomes V1/2. In the phase β (however, α<β<π/2), the voltage waveform 170 becomes V1. The voltage waveform 170 maintains V1 from phase β to phase π-β, and becomes V1/2 at phase π-β. Furthermore, the voltage waveform 170 changes to 0 at phase π-α.

In the half cycle after the phase π, the voltage waveform 170 takes a waveform in which the polarity of the half cycle from the phase 0 to π described above is reversed. Thereafter, the voltage waveform 170 repeats the waveform described above.

Here, the voltage waveform 172 lags behind the voltage waveform 170 by a phase difference φ. By storing the energy generated by the voltage difference between the primary and secondary sides during this delay in the leakage inductance 116 of the transformer 114 and then releasing it, the 3L-2L converter 100 also serves as a buck-boost converter.

FIG. 4 shows an example of the primary side voltage 200, the secondary side voltage 202, and the current waveform 204 of the transformer 114 when the 3L-2L converter 100 is operated in the form shown in FIG. In this example, the primary voltage V1 is 200V and the secondary voltage is 350V.

FIG. 5 shows the configuration of the control circuit 250. Referring to FIG. 5, control circuit 250 is for performing constant current control on 3L-2L converter 100. The control circuit 250 receives the current command value I ^* and the current I of the transformer 114, and includes a subtracter 260 for feedback controlling the phase difference φ, a PI control unit 262, and a control circuit for controlling the phase difference φ of the PI control unit 262. It includes a converter 264 and a relational expression converter 268 that receives the current command value I ^* and converts it into phases α and β according to a predetermined relational expression of the current command value I ^* . The control circuit 250 further includes a PWM signal generation section for receiving the phase difference φ from the conversion section 264 and the phases α and β from the relational expression conversion section 268 and outputting a PWM signal for controlling the switching elements S1 to S12. 266 included.

Note that the point that the phase difference φ is generated by feedback is the same as that performed by a conventional 2L-2L converter. Furthermore, the mechanism for generating the PWM signal that controls the switching element S12 from the switching element S based on the phase difference φ and the phases α and β is a well-known technology, and is clear from the description in the above-mentioned Non-Patent Document 1, so the details will be described here. Don't repeat.

1C. Method for Determining a Relational Expression Hereinafter, a method for determining a relational expression used in the relational expression conversion unit 268 for converting the current command value I ^* into phases α and β will be described. The method of determining this relational expression is basically the same for both phases α and β, and the concept is also common to all switching elements of the primary side 3L full-bridge circuit 110. Therefore, only the phase β in the control of the switching element S1 will be described here. The relational expression described below is for uniquely determining the value of the phase β from the current command value I ^* .

The power transfer equation in the 3L-2L converter 100 shown in FIG. 2 is the following equation using the phase difference φ and the phases α and β as parameters.

図２に示すスイッチング素子Ｓ１がソフトスイッチングとなる条件は、スイッチング素子Ｓ１がオンとなるタイミングにおいて、矢印１４４により示すようにスイッチング素子Ｓ１にマイナス方向（電圧と逆方向）の電流が流れていることであると考えられる。この条件を図２において考えると、トランス１１４に流れるトランス電流ｉ_Ｌの値が図２の矢印１４６により示す方向のときに正だとして、「ｉ_Ｌ≦０」が条件となる。

The condition for soft switching of the switching element S1 shown in FIG. 2 is that a current in the negative direction (in the opposite direction to the voltage) flows through the switching element S1 as shown by an arrow 144 at the timing when the switching element S1 is turned on. It is thought that. Considering this condition in FIG. 2, it is assumed that the value of the transformer current i _L flowing through the transformer 114 is positive when it is in the direction indicated by the arrow 146 in FIG. 2, and the condition is "i _L ≦0".

When this formula is further refined, it becomes the following formula.

この式を変形すると以下の不等式が得られる。

By transforming this equation, the following inequality is obtained.

ただし最後の式のｍは変圧比であり、ｍ＝（Ｖ２／ｎＶ１）である。

However, m in the last equation is the voltage transformation ratio, and m=(V2/nV1).

This equation is viewed as a relational equation between φ and the transformation ratio m, and FIG. 6 is a graph of various values of β. Although FIG. 6 is a little hard to see, it shows the range in which the above equation holds true when, for example, the area between the curve 308 and the straight line 318 is β=80 degrees. Similarly, when the ranges between the curve 306 and the straight line 316, the curve 304 and the straight line 314, the curve 302 and the straight line 312, and the curve 300 and the straight line 310 are respectively β=60 degrees, 40 degrees, 20 degrees, and 0 degrees, the above equation is Indicates the range in which this is true. Note that among the graphs shown in FIG. 6, curves 300, 302, 304, 306, and 308 continue to infinity using straight lines 310, 312, 314, 316, and 318, respectively, as asymptote lines. Therefore, only the range where the transformation ratio is 2.4 or less is shown in the graph.

FIG. 7 shows a plot of the relationship shown in FIG. 6 to show the value of the transformation ratio for the combination of phase difference φ and phase β. In FIG. 7, the screen is simply shaded, but at the time of design, this screen is drawn in different colors depending on the value of the transformation ratio. The colored portion is the area that satisfies the above relational expression. Therefore, it can be seen that soft switching is possible in the areas where the areas are drawn in color. Furthermore, by looking at the drawing color of a certain point, one can also know what transformation ratio is possible for the combination of phase difference φ and phase β corresponding to that point.

On the other hand, FIG. 8 shows a diagram in which the horizontal axis represents the phase difference φ, the vertical axis represents the phase β, and the value of the maximum possible transformer current I in the 3L-2L converter 100 is plotted for that combination. The unit of the current value indicated by the legend in FIG. 8 is ampere. Note that the value of the maximum current I is calculated assuming that α=0 in the above power transfer equation.

From these two figures, the relational expression of the phase β with respect to the transformer current is specified as follows. In FIG. 7, a region where a predetermined transformation ratio, for example transformation ratio m=2.5, can be achieved is specified. This designation can be determined by looking at the drawing color of the area as described above. Next, in FIG. 8, an area corresponding to the area specified in the diagram shown in FIG. 7 is cut out and displayed on a new screen. This region is shown as region 350 in FIG. In FIGS. 7 and 8, the horizontal and vertical axes coincide, so such operations can be easily performed. Also in FIG. 9, the horizontal axis represents the phase difference φ, and the vertical axis represents the phase β.

Referring to FIG. 9, this figure shows a region 350 that is a combination of phase difference φ and phase β that allows soft switching, and in which a desired transformation ratio can be realized, and a maximum current value that is possible in that region. It is shown along with the distribution of .

Therefore, a line 352 shown in FIG. 9 is drawn inside this area 350. Then, the line 352 provides a combination (relationship) of phase difference φ and phase β that allows current to flow in a predetermined range (for example, 0 ampere to 40 ampere) at a transformation ratio of 2.5.

Therefore, the line 352 drawn in FIG. 9 can be replotted into a graph with the current on the horizontal axis and the phase β on the vertical axis, as shown in FIG. The result is curve 380. As mentioned above, this curve 380 can be considered to represent the combination of phase difference φ and phase β that allows current to flow in a predetermined range at a predetermined transformation ratio in the form of a function of phase β against phase difference φ. can. This graph is approximated by a predetermined function, such as a polynomial. A curve 382 shown in FIG. 10 represents the curve 380 by a third-order polynomial of φ. The phase β with respect to the current command value I can be specified using this third-order polynomial. This function is a simple polynomial, and if the current command value I is given, the phase beta can be easily calculated. Further, for example, calculation of this phase β can be performed independently of calculation of phase α. Therefore, unlike the technique described in Non-Patent Document 1, even a weak microprocessor can perform sufficient processing.

However, due to the nature of control, the phase β needs to be uniquely determined for a certain value of the current command value I. Therefore, the curve 380 shown in FIG. 10 needs to be a monotonous function in the strong sense (in the case of FIG. 10, a monotonically decreasing function). This is because otherwise the phase β cannot be determined. At the design stage of the 3L-2L converter 100, if the curve drawn in FIG. 10 after drawing the line 352 shown in FIG. 9 is not a monotonic function, it is necessary to newly draw the line 352.

Note that when approximating a curve using a polynomial, there is no particular restriction on the degree. However, since it is necessary to keep the amount of calculation low, a certain degree of order, for example about 3rd order, is desirable. In the example shown in FIG. 10, there is an inflection point of the curve 380, so it is approximated by a cubic equation, but if the error range can be widened, it is approximated by a quadratic equation or a linear equation. It is not impossible. Of course, conversely, the curve may be approximated by a quartic or higher order equation.

In this embodiment, fluctuations in the current command value I are small. Therefore, the control period for the phases α and β can be made longer than the control period for the phase difference φ. As a result, the load placed on the controlling microprocessor is reduced, and three-level control of the full-bridge circuit can be realized with a simple configuration.

Note that, if the microprocessor has sufficient storage capacity, instead of using an equation that approximates the curve 380, the combinations of the current command value I and the phase β that make up the curve 380 can be prepared in a table format. In this case, if the current command value I is given, the corresponding phase β can be identified by table lookup. If the specified current command value I does not exist in the table, the phase β may be interpolated using neighboring data in the table.

Furthermore, in the above embodiment, FIG. 9 is created from FIGS. 7 and 8, a line 352 is drawn on the map of FIG. 9, and the line 352 is converted to the map of FIG. 10. However, this disclosure is not limited to such embodiments. Conversely, a desired curve (a monotonically decreasing function in a strong sense) showing the relationship between the current command value I and the phase β may be drawn on the map in FIG. 10, and it may be mapped in FIG. In this case, if the line obtained from the map (the line corresponding to the line 352 in FIG. 9) falls within the region 350, the first curve drawn is adopted as indicating the relationship between the current command value I and the phase β. can. In the following, as in the above embodiment, an approximate expression or a table lookup may be used. If the line obtained from the map does not fall within the area 350, return to FIG. 10 and redraw the curve.

Further, in the above embodiment, for example, in FIG. 7, the value of the transformation ratio m is represented by the color drawn on the map. However, this disclosure is not limited to such embodiments. Since the transformation ratio is a function of the phase difference φ and the phase β, it can be represented by a three-dimensional graph in which the x-axis is the phase difference φ, the y-axis is the phase β, and the z-axis is the transformation ratio. If the transformation ratio is represented by a three-dimensional graph, when this three-dimensional graph is cut by two planes perpendicular to the Z-axis that indicate the range of the desired transformation ratio, the area surrounded by the cutting line in the three-dimensional graph is By projecting onto the xy plane, a figure corresponding to area 350 in FIG. 9 is obtained.

2. Second Embodiment In the first embodiment described above, when the transformation ratio m is limited to a certain range and the current command value I is given, the phase α and the phase β are calculated as functions of the current command value I, respectively. are doing. However, this disclosure is not limited to such embodiments. In this second embodiment, each point in the region 350 in FIG. 9 is considered to exhibit the transformation ratio shown in FIG. 7 and the current value shown in FIG. 8. Then, the diagram obtained in this way is converted into a three-dimensional graph (three-dimensional curved surface) in which each point of the area 350 is plotted with the current command value on the x-axis, the transformation ratio m on the y-axis, and the phase β on the z-axis. can. This three-dimensional graph has the same meaning as curve 380 in FIG. Therefore, the same effect as in the first embodiment can be obtained by approximating the curved surface by a function having two variables, the current command value It and the transformation ratio m, or by having it as a two-dimensional table.

FIG. 11 shows a converter 410 that calculates the phases α and β in the control circuit 400 according to the second embodiment. Referring to FIG. 11, converter 410 receives the current command value and the primary voltage and secondary voltage of transformer 114 as input, and internally converts the current command value into a function that calculates phases α and β, respectively. By inputting and transformation ratio (=secondary side voltage/(n*primary side voltage)), phases α and β are calculated and provided to the PWM signal generation unit 412. By replacing the relational expression converter 268 of FIG. 5 with the converter 410 shown in FIG. 11, the switching elements of the 3L-2L converter 100 can be controlled with a simple configuration, as in the first embodiment. Temporal fluctuations in the current command value, primary voltage, and secondary voltage are small. Therefore, in this embodiment as well, the phases α and β can be calculated at longer intervals than the feedback control of the phase difference φ. As a result, the load placed on the microprocessor for control can be reduced, and three-level control can be achieved in a practical manner.

3. Third Embodiment In the first embodiment, the case where the transformation ratio m=2.5 was explained as an example. When the transformation ratios are different, relational expressions for determining the current command value and the phases α and β are determined based on the transformation ratios using the procedure described in the first embodiment. However, if a relational expression for determining the phases α and β is obtained for a predetermined transformation ratio, by following this third embodiment, by slightly changing the control according to the first embodiment, an arbitrary The values of phases α and β can be determined for the transformation ratio.

Referring again to FIG. 3, when the primary side is a three-level full bridge circuit, the transmitted power is reduced by the portion indicated by phases α and β. On the other hand, for example, when the transformation ratio approaches 1, it is no longer necessary to make both the phases α and β large. Therefore, by reducing the phases α and β, the transmitted power can be increased. That is, when the transformation ratio approaches 1 from 2.5, both phases α and β can be reduced.

This situation is shown in FIG. A curve 380 shown in FIG. 12 is the relationship between the current command value and the phase β when the transformation ratio is 2.5, which was determined in the first embodiment. A curve 382 is an approximation of the curve 380 using a third-order polynomial. As described above, when the transformation ratio approaches 1, the phase β can be reduced. For example, as shown in FIG. 12, when the current command value I is about 12.5 amperes, the phase β determined from the curve 382 is about 0.31. However, if the transformation ratio approaches 1 (if the transformation ratio becomes smaller), the phase β can be made smaller. In order to utilize such a result, it is sufficient to use the phase β corresponding to a point 450 at a position where the entire curve 382 is reduced downward along the vertical axis. For example, the curve 382 may be multiplied by a coefficient k determined by the following formula. However, in the following formula, m is the transformation ratio.

図１３に、この第３実施形態に係る３Ｌ－２Ｌコンバータの１次側の制御回路４６０の構成を示す。図１３を参照して、この第３実施形態に係る制御回路４６０が図５に示す制御回路２５０と異なるのは、関係式変換部２６８が出力する位相α及びβに上記係数ｋを乗算するための乗算回路４７０をさらに含む点である。それ以外の点においては、制御回路４６０は制御回路２５０と同じである。

FIG. 13 shows the configuration of a primary side control circuit 460 of a 3L-2L converter according to the third embodiment. Referring to FIG. 13, a control circuit 460 according to the third embodiment is different from the control circuit 250 shown in FIG. It further includes a multiplication circuit 470. In other respects, control circuit 460 is the same as control circuit 250.

According to the third embodiment, when the relational expression between the current command value I and the phases α and β at a specific transformation ratio is obtained, the relation also applies to a transformation ratio different from that transformation ratio. The phases α and β can be calculated by simple control using the equations. As a result, the three-level full-bridge circuit can be controlled by a simple control circuit even for different transformation ratio values. In addition, the control circuit itself only needs to be slightly changed at a specific transformation ratio, which has the effect of making it easy to design the control circuit.

4. Fourth Embodiment The first embodiment, second embodiment, and third embodiment described above all relate to a control circuit that performs constant current control on the primary side 3L full-bridge circuit 110 of a 3L-2L converter. However, this circuit is not limited to such embodiments. This disclosure can also be applied to the case where the primary side 3L full bridge circuit is subjected to constant voltage control.

FIG. 14 is a block diagram showing the configuration of a control circuit 500 for constant voltage control of the primary side 3L full-bridge circuit 110 according to the fourth embodiment. Referring to FIG. 14, control circuit 500 includes a control circuit 514 similar to control circuit 250 shown in ^FIG . A subtraction circuit 510 that subtracts the next-side voltage V is provided between the subtraction circuit 510 and the control circuit 514, and generates a current command value I ^* from the output of the subtraction circuit 510 and inputs it to the positive terminal of the subtracter 260. PI control unit 512.

The control circuit 514 as a minor loop has the same configuration as the control circuit 250 shown in FIG. Therefore, the relational expression conversion unit 268 calculates the phases α and β necessary for the PWM signal generation unit 266 to generate the switching signal for controlling the primary side 3L full-bridge circuit 110 in the same manner as in the first embodiment. can.

Also in this fourth embodiment, the phases α and β can be calculated using a simple relational expression. The amount of calculation required for this purpose is also small. Further, the period for calculating the phases α and β can be made longer than the control period for the phase difference φ. As a result, the primary side 3L full-bridge circuit 110 can be practically controlled using a simple circuit and with a small amount of calculation.

5. Simulation Results The voltage/current range in which soft switching can be realized was investigated by simulation for the conventional 2L-2L converter and the 3L-2L converter 100 according to the first embodiment. The results are shown in FIG. Of the two tables shown in FIG. 15, the upper row shows the results for the 2L-2L converter. The lower row shows the results from the 3L-2L converter 100. In the two tables shown in FIG. 15, the value shown on the leftmost side is the value of the primary side voltage. In this simulation, the secondary voltage was fixed at 350V. In FIG. 15, "OK" indicates that soft switching was achieved, and "NG" indicates that soft switching was not achieved.

As can be seen from FIG. 15, in the conventional 2L-2L converter, soft switching cannot be realized when the absolute value of the current becomes small. It can also be seen that soft switching cannot be achieved as the transformation ratio increases.

On the other hand, in the case of the 3L-2L converter 100 according to the first embodiment, such a result does not occur. It can be seen that soft switching can be achieved even if the absolute value of the current becomes small or the transformation ratio becomes large.

FIG. 16 shows a voltage waveform 550 and a current waveform 552 on the primary side of the transformer in a conventional 2L-2L converter in simulation. FIG. 17 shows a voltage waveform 560 and a current waveform 562 of the switching element S1 shown in FIG. 2. Note that the phase difference φ at this time was 24.6 degrees.

In particular, from FIG. 17, it can be seen that at time 500 microseconds, the current is almost 0 when the switching element S1 is turned on, and the current starts to increase almost at the same time as the voltage of the switching element S1 increases. Therefore, in an actual device, a loss will occur in this part.

FIG. 18 shows a transformer voltage waveform 570 and current waveform 572 on the primary side when the 3L-2L converter 100 in the first embodiment is simulated under similar conditions. FIG. 19 shows a gate-source voltage waveform 590 and a current waveform 592 of the switching element S1 before and after switching the switching element S1 from off to on. At this time, the phase difference φ was 31.4 degrees, the phase α was 16.1 degrees, and the phase β was 32.2 degrees.

It can be seen from FIG. 18 that the transformer voltage on the primary side is controlled to three levels. Further, from FIG. 19, it can be seen that the current flowing through the switching element S1 is negative at the time of switching the switching element S1 from off to on. That is, the switching of the switching element S1 is soft switching and no loss occurs.

As described above, it can be seen from the simulation that the above control circuit is effective.

6. Modifications In the above embodiments, a three-level full bridge circuit is used on the primary side, and a two-level full bridge circuit is used on the secondary side. However, this disclosure is not limited to such embodiments. A three-level full bridge circuit may be employed not only on the primary side but also on the secondary side. FIG. 20 shows an example of the configuration of such a 3L-3L converter.

Referring to FIG. 20, this 3L-3L converter 600 includes a primary side 3L full bridge circuit 110 connected between a storage battery 102 and a transformer 114, and a secondary side 3L full bridge circuit 110 connected between a storage battery 102 and a transformer 114. It includes a full bridge circuit 610, a secondary side 3L full bridge circuit 610 connected to the DC bus 104, and a control circuit 620 for controlling the primary side 3L full bridge circuit 110 and the secondary side 3L full bridge circuit 610. . The combination of the primary-side 3L full-bridge circuit 110, the transformer 114, and the secondary-side 3L full-bridge circuit 610 is known, and a method for driving them at three levels is also known, for example, from the description in Non-Patent Document 1. However, the control circuit 620 differs from the prior art in that the phases α and β for controlling the primary side 3L full-bridge circuit 110 explained in each of the above embodiments are determined by the relational expression with the current command value proposed in this disclosure. In addition to determining the phase γ and delta for driving the secondary side 3L full bridge circuit 610 at three levels, the phase γ and delta are also determined based on the relational expression with the current command value. The phase γ corresponds to the phase α in the primary side 3L full-bridge circuit 110. The phase delta corresponds to the phase β in the primary side 3L full-bridge circuit 110.

Referring to FIG. 21, control circuit 620 has a similar configuration to control circuit 250 shown in FIG. However, instead of the relational expression converter 268 shown in FIG. 5, the control circuit 620 calculates and outputs the phases α, β, γ, and δ by substituting the values of the current command value I ^* into the respective corresponding functions. It differs from the control circuit 250 in that it includes a relational expression conversion section 630 for converting the relational expression. Further, the control circuit 620 receives the phase difference φ output from the conversion unit 264 and the phases α, β, γ, and δ output from the control circuit 620, instead of the PWM signal generation unit 266 shown in FIG. The control circuit also includes a 3L-3L PWM signal generation unit 632 for outputting a PWM signal that controls each switching element of the side 3L full-bridge circuit 110 and each switching element of the secondary side 3L full-bridge circuit 610. Different from 250.

The method for determining the relational expression for calculating the phases γ and δ as a function of the current command value is the same as the method for determining the relational expression for calculating the phase β described in the first embodiment.

The basic operation of the control circuit 620 is also basically the same as that of the control circuit 250 of the first embodiment. However, in the control circuit 620, unlike the first embodiment, the relational expression converter 630 calculates not only the phases α and β but also the phases γ and δ as a function of the current command value, and 3L-3L In that the PWM signal generating section 632 outputs more PWM signals to control each switching element of the secondary side 3L full bridge circuit 610 than the PWM signal generating section 266 of the first embodiment. , the operation of control circuit 620 is different from control circuit 250.

Similar to the embodiment described above, this modification also provides the effect that the 3L-3L converter 600 can be controlled with a simple configuration and with a small amount of calculation.

The above embodiments all relate to an NPC type DAB type isolated converter as shown in FIG. However, this disclosure is not limited to such embodiments. For example, instead of the NPC type primary side 3L full bridge circuit 110 shown in FIG. 2, a diode clamp type primary side 3L full bridge circuit 660 may be employed. FIG. 22 shows a circuit block diagram of a 3L-2L converter 650 according to such a modification.

Referring to FIG. 22, in the primary 3L full-bridge circuit 110 shown in FIG. It was replaced using In other respects, each part of the 3L-2L converter 650 is similar to the primary side 3L full-bridge circuit 110. The circuit configuration of the control circuit 250 is also similar to that of the first embodiment. The method for determining the relational expression (function) for calculating the phases α and β from the current command values in the control circuit 250 is also the same as the method in the first embodiment.

Similar to the above embodiments, this 3L-2L converter 650 also has the effect of being able to practically control the 3L-2L converter 650 with a simple configuration and with a small amount of calculation.

The embodiments disclosed this time should be considered to be illustrative in all respects and not restrictive. The scope of the present disclosure is not indicated by the description of the detailed description of the disclosure, but by each claim, and all changes within the meaning and scope equivalent to the wording of the claims are intended. intended to be included.

S1, S2, S3, S4, S5, S6, S7, S8, S9, S10, S11, S12 Switching element 50, 170, 172, 550, 560 Voltage waveform 52, 204, 552, 562, 572, 592 Current waveform 54 , 56, 350 Area 58, 60 Power loss 100, 650 3L-2L converter 102 Storage battery 104 DC bus 110, 660 Primary side 3L full bridge circuit 112 Secondary side full bridge circuit 114 Transformer 116 Leakage inductance 120, 122, 140 Capacitor 124 Switching element group 126, 128, 130, 142, 144 Node 200 Primary side voltage 202 Secondary side voltage 250, 400, 460, 500, 514, 620 Control circuit 260 Subtractor 262, 512 PI control section 264, 410 Conversion Sections 266, 412 PWM signal generation sections 268, 630 Relational expression conversion sections 300, 302, 304, 306, 308, 380, 382 Curves 310, 312, 314, 316, 318 Straight line 352 Line 470 Multiplier circuit 510 Subtraction circuit 570 Transformer voltage Waveform 590 Gate-source voltage waveform 600 3L-3L converter 610 Secondary side 3L full bridge circuit 632 PWM signal generation section for 3L-3L

Claims (17)

a transformer, a primary side three-level bridge circuit connected to the primary side of the transformer, and a secondary side bridge circuit connected to the secondary side of the transformer, wherein the primary side three-level bridge circuit is connected to the transformer. A control circuit for controlling a power conversion device that performs power conversion by changing a phase difference φ between a voltage waveform applied to the transformer and a voltage waveform applied by the secondary side bridge circuit to the transformer. hand,
a feedback circuit that performs feedback control of the phase difference φ based on a given current command value and a state value of the transformer;
Timing for calculating two phases α and β, where 0<α<β<π, that determine the switching timing of each switching element of the primary side three-level bridge circuit, with respect to the current command value, according to each relational expression. a calculation circuit;
a control circuit including a PWM signal generation circuit that generates a PWM signal for controlling switching elements of the primary side three-level bridge circuit and the secondary side bridge circuit using the phase difference φ and the phases α and β; . 2. The control circuit according to claim 1, wherein a calculation cycle of the two phases α and β by the timing calculation circuit is longer than a feedback cycle by the feedback circuit. The control according to claim 1 or 2, wherein the relational expression includes a first relational expression regarding the phase α and the current command value, and a second relational expression regarding the phase β and the current command value. circuit. The control circuit according to claim 3, wherein the first relational expression is a function of the current command value that uniquely determines the phase α for the current command value within a certain range. 5. The control circuit according to claim 4, wherein the first relational expression is a strongly monotonically decreasing function of the current command value in the certain range that determines the phase α. The control circuit according to claim 3, wherein the second relational expression is a function of the current command value that uniquely determines the phase β for the current command value in the certain range. 7. The control circuit according to claim 6, wherein the second relational expression is a strongly monotonically decreasing function of the current command value in the certain range that determines the phase β. Each switching element of the primary side three-level bridge circuit has a first terminal, a second terminal, and a control terminal that controls conduction between the first terminal and the second terminal,
3. The control circuit according to claim 1, wherein the relational expression is selected such that switching by each of the switching elements is a soft switch. The above relational expression is based on the condition that the current flowing through the body diode of each switching element is in the opposite direction to the voltage applied to the first terminal and the second terminal when each switching element is not conducting. 9. The control circuit according to claim 8, wherein the phase α and the phase β are selected such that the switching element is conductive. The primary side three-level bridge circuit functions to apply 0, ±V1/2, and ±V1 to the primary side to the transformer,
The phase α determines the timing at which the voltage applied to the primary side of the transformer is changed from 0 to the ±V1/2,
The control according to claim 1 or 2, wherein the phase obtained by subtracting the phase α from the phase π determines the timing at which the voltage applied to the primary side of the transformer is changed from the ±V1/2 to 0. circuit. The phase β determines the timing at which the voltage applied to the primary side of the transformer is changed from the ±V1/2 to ±V1, respectively,
11. The control circuit according to claim 10, wherein a phase obtained by subtracting the phase β from the phase π determines the timing at which the voltage applied to the primary side of the transformer is changed from the ±V1 to the ±V1/2. The control circuit according to claim 1 or 2, wherein the relational expression is a polynomial of the current command value. 13. The control circuit according to claim 12, wherein the relational expression is a polynomial of order three or less of the current command value. The control according to claim 1 or 2, further comprising a current command value generation circuit that generates the current command value by feedback based on the voltage command value and the voltage applied to the primary side of the transformer. circuit. The secondary side bridge circuit includes a secondary side 3-level bridge circuit,
The control circuit further includes:
For the current command value, two phases γ and δ, where 0<γ<δ<π, are calculated, which determine the switching timing of each switching element of the secondary side three-level bridge circuit, according to the respective relational expressions. further including a next side timing calculation circuit,
The PWM signal generation circuit includes:
The switching elements of the primary side three-level bridge circuit are controlled using the phase difference φ and the phases α and β, and the switching elements of the secondary side three-level bridge circuit are controlled using the phase difference φ and the phases γ and δ. 3. The control circuit according to claim 1, further comprising a 3-level PWM signal generation circuit for controlling switching elements of the circuit. A control circuit according to claim 1 or claim 2,
the transformer;
the primary side three-level bridge circuit connected to the primary side of the transformer, including a semiconductor switching element whose switching is controlled by the control circuit;
and a secondary side circuit connected to the secondary side of the transformer. A control circuit according to claim 15;
the transformer;
the primary side three-level bridge circuit connected to the primary side of the transformer, including a semiconductor switching element whose switching is controlled by the primary side PWM signal generation circuit of the control circuit;
A power conversion device comprising: the secondary side three-level bridge circuit connected to the secondary side of the transformer, the secondary side three-level bridge circuit including a semiconductor switching element whose switching is controlled by the secondary side PWM signal generation circuit of the control circuit.

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