JP2023169656A - Control device - Google Patents

Abstract

To provide a control device capable of reducing torque ripples in a case where an electric motor is controlled in an overmodulation region of an inverter by vector control.SOLUTION: A control device 3 comprises: a restriction unit 14 that comprises a voltage limiter 19 restricting a d-axis voltage command value and/or a q-axis voltage command value, and a filter 18 whose time constant can be modified; an identification unit 16 that identifies the harmonic component with the largest amplitude, among harmonic components superposed on currents flowing in respective phases of an electric motor M; and a time constant modification unit 17 that modifies the time constant of the filter 18 so as to reduce a harmonic component gain corresponding to the harmonic component identified by the identification unit 16. The filter 18 has such a time constant that a gain of a harmonic component of a sixth order becomes smaller than that of a harmonic component of a predetermined order equal to or lower than fifth order, and such a time constant that the gain of the harmonic component of the predetermined order is smaller than that of the harmonic component of the sixth order.SELECTED DRAWING: Figure 1

Description

The present invention relates to a control device for an inverter of an electric motor.

Non-Patent Document 1 discloses a method of controlling an electric motor by vector control in an overmodulation region of an inverter. The control device described in Non-Patent Document 1 converts an alternating current flowing through a motor into a d-axis current and a q-axis current, and controls the d-axis current and q-axis current so that they approach the d-axis current command value and the q-axis current command value. The motor is controlled by so-called vector control, in which the voltage command value is determined at control.

When using an overmodulation region that is a nonlinear region of an inverter, the inverter has nonlinear characteristics. Therefore, in order to utilize the overmodulation region in vector control, a nonlinear element is required. The control device described in Non-Patent Document 1 includes nonlinear elements such as a voltage amplitude compensator and a voltage limiter in order to perform vector control using the overmodulation region of the inverter. A voltage limiter is installed to avoid generating a command voltage that exceeds the input voltage of the inverter under driving conditions where the induced voltage is high, such as during transient conditions where the current needs to be converted sharply or during high-output driving. ing. The voltage limiter determines the d-axis voltage command value and the q-axis voltage command value so that the command voltage is equal to or lower than the input voltage of the inverter.

Yosuke Nakayama, “Filter design method for permanent magnet synchronous motor vector control system that can operate up to the overmodulation region”, [online], published March 25, 2019, Nagoya University doctoral dissertation, [January 2020] 28th search], Internet <URL: https://nagoya.repo.nii.ac.jp/records/27749#.YfOgZOrP0UE>

In the overmodulation region of the inverter, the output voltage waveform changes from a sine wave to a rectangular wave as the voltage command value increases. The harmonic components superimposed on the output current become larger as the output voltage approaches a rectangular wave. That is, when the overmodulation region of the inverter is utilized, the harmonic components superimposed on the output current become large. In vector control, current feedback control is performed on the dq coordinate system, so harmonic components appear on the dq coordinate system when a three-phase rectangular wave voltage is applied. In a control system that utilizes the overmodulation region of an inverter, a sixth-order harmonic component (harmonic component six times the electrical angular frequency of the motor) having a large amplitude is superimposed on the dq coordinate system. Therefore, in the control device described in Non-Patent Document 1, a filter is provided in the current feedback loop to attenuate (remove) the sixth harmonic component superimposed on the dq coordinate system.

One aspect of the present invention is to provide a control device that can reduce torque ripple when controlling an electric motor by vector control in an overmodulation region of an inverter.

In the course of extensive research into the above-mentioned problem, the applicant of the present application found that in a configuration in which a voltage limiter is provided to operate the vector control system to the overmodulation region, harmonics of the 6th order or lower superimposed on the dq coordinate system It has been found that there are cases where the amplitude of the sixth-order harmonic component is the highest, and cases where the amplitude of the harmonic component of a predetermined order below the fifth-order is the highest. Therefore, the applicant identified the harmonic component with the largest amplitude among the harmonic components superimposed on the current flowing through each phase of the motor, and by attenuating the harmonic component of the specified order, the torque ripple was reduced. The present invention was completed based on the knowledge that the reduction can be achieved.

A control device according to one aspect of the present invention includes a first converting section that converts current flowing through each phase of a motor into a d-axis current and a q-axis current, and a first converting section that converts a current flowing through each phase of a motor into a d-axis current and a q-axis current converted by the first converting section. and a current control unit that generates a d-axis voltage command value and a q-axis voltage command value based on the d-axis current command value and q-axis current command value, and a current control unit that generates a d-axis voltage command value and a q-axis voltage command value as input. and a voltage limit that limits the d-axis voltage command value and/or the q-axis voltage command value so that when the d-axis voltage command value and the q-axis voltage command value exceed the input voltage of the inverter, they become equal to or less than the input voltage. and a variable filter capable of changing a time constant; and converting the d-axis voltage command value and the q-axis voltage command value output from the restriction part into voltage command values corresponding to each phase of the motor. and a second conversion unit that outputs a voltage command value, a identification unit that identifies the harmonic component with the largest amplitude among the harmonic components superimposed on the current flowing through each phase of the motor, and a A control device for overmodulating an inverter, comprising: a time constant changing unit that changes a time constant of a variable filter so as to reduce a gain of a harmonic component corresponding to a harmonic component, the variable filter comprising: 6 A time constant that makes the gain of the next harmonic component smaller than the gain of the harmonic component of a predetermined order below the fifth order, and a time constant that makes the gain of the harmonic component of the predetermined order smaller than the gain of the sixth harmonic component. It has a time constant of

In the control device according to one aspect of the present invention, the identification unit identifies a harmonic component having the largest amplitude among harmonic components superimposed on a current flowing through each phase of the motor. The time constant changing section changes the time constant of the variable filter so as to reduce the gain of the harmonic component corresponding to the harmonic component identified by the identifying section. As a result, in the control device, for example, when the 6th harmonic component is dominant in the dq coordinate system, the variable filter can attenuate the 6th harmonic component, and the dq coordinate system When harmonic components of a predetermined order of the fifth order or lower are dominant in the system, the variable filter can attenuate the harmonic components of the predetermined order of the fifth order or lower. In this way, the control device can attenuate not only the sixth harmonic component superimposed on the output current of the motor, but also the harmonic components of a predetermined order of the fifth order or lower. Therefore, the control device can reduce torque ripple when controlling the electric motor by vector control in the overmodulation region of the inverter.

Three-phase AC drive electric motors (particularly PMSM: Permanent Magnet Synchronous Motors) structurally include harmonic components in the induced voltage, so the generated torque has harmonics of an integral multiple of the electrical angular frequency, mainly 6 times the order. A wave component is generated. In the control device, harmonic components generated due to the structure of the motor can also be attenuated by the variable filter.

In one embodiment, the identification unit receives at least one of the d-axis current and the q-axis current and determines a time constant that makes the gain of a harmonic component of a predetermined order smaller than the gain of a sixth harmonic component. a first filter to which at least one of the d-axis current and the q-axis current is input and which has a time constant that makes the gain of the sixth-order harmonic component smaller than the gain of the harmonic component of a predetermined order. Two filters may be provided, and the amplitude of the output current of the first filter and the amplitude of the output current of the second filter may be compared to identify a harmonic component having a large amplitude. With this configuration, it is possible to specify a harmonic component with a larger amplitude among the harmonic components superimposed on the output current of the motor.

In one embodiment, the identification unit acquires the electrical angular frequency of the motor, and detects the current flowing in at least one phase of the current flowing in each phase of the motor or the d-axis current and the q-axis current in one period of the electrical angular frequency. The harmonic component with the largest amplitude may be identified based on the number of local maximum values or local minimum values of at least one of the currents. With this configuration as well, it is possible to specify the harmonic component with the largest amplitude among the harmonic components of the sixth order or lower that are superimposed on the output current of the motor.

In one embodiment, a d-axis voltage command value and a q-axis voltage command value may be input to the variable filter. With this configuration, it is possible to improve responsiveness in vector control that performs current feedback control on the dq coordinate system.

According to one aspect of the present invention, torque ripple can be reduced when controlling an electric motor by vector control in an overmodulation region of an inverter.

FIG. 1 is a diagram showing a control system for an electric motor including a control device according to an embodiment. FIG. 2 is a diagram showing the configuration of the specifying section. FIG. 3 is a diagram showing a control system for an electric motor including a control device according to another embodiment. FIG. 4 is a diagram showing a control system for an electric motor including a control device according to another embodiment.

Hereinafter, preferred embodiments of the present invention will be described in detail with reference to the accompanying drawings. In the description of the drawings, the same or equivalent elements are given the same reference numerals, and redundant description will be omitted.

A control system 1 shown in FIG. 1 controls the drive of an electric motor M mounted on a vehicle such as an electric forklift or a plug-in hybrid vehicle, for example. The control system 1 includes an inverter circuit (inverter) 2, a control device 3, and current sensors Se1, Se2, and Se3.

The control device 3 of this embodiment controls the electric motor M by vector control in the overmodulation region of the inverter circuit 2, and harmonic components of the 6th order and below that are superimposed on the dq coordinate system are the 6th order harmonic components. It is assumed that the configuration is such that there are cases where the amplitude of the harmonic component is the largest and cases where the amplitude of the third-order harmonic component is the largest. Note that among the harmonic components of the fifth order or lower superimposed on the dq coordinate system, the order of the harmonic component whose amplitude is larger than the amplitude of the sixth harmonic component is determined by what kind of nonlinear element is provided in the control device 3. It depends.

Inverter circuit 2 drives electric motor M with DC power supplied from DC power supply P. The inverter circuit 2 includes a capacitor C and switching elements SW1, SW2, SW3, SW4, SW5, and SW6.

The capacitor C smoothes the voltage V _in output from the DC power supply P and input to the inverter circuit 2 .

The switching elements SW1 to SW6 are, for example, IGBTs (Insulated Gate Bipolar Transistors). One end of the capacitor C is connected to the positive terminal of a DC power supply P and the collector terminals of switching elements SW1, SW3, and SW5, which are three upper arm switching elements. The other end of the capacitor C is connected to the negative terminal of the DC power supply P and each emitter terminal of switching elements SW2, SW4, and SW6, which are three lower arm switching elements.

A connection point between the emitter terminal of switching element SW1 and the collector terminal of switching element SW2 is connected to the U-phase input terminal of electric motor M via current sensor Se1. A connection point between the emitter terminal of switching element SW3 and the collector terminal of switching element SW4 is connected to the V-phase input terminal of electric motor M via current sensor Se2. A connection point between the emitter terminal of switching element SW5 and the collector terminal of switching element SW6 is connected to the W-phase input terminal of electric motor M via current sensor Se3.

The switching element SW1 is turned on or off based on the drive signal S1 output from the control device 3. The switching element SW2 is turned on or off based on the drive signal S2 output from the control device 3. The switching element SW3 is turned on or off based on the drive signal S3 output from the control device 3. The switching element SW4 is turned on or off based on the drive signal S4 output from the control device 3. Switching element SW5 is turned on or off based on drive signal S5 output from control device 3. The switching element SW6 is turned on or off based on the drive signal S6 output from the control device 3.

By turning on or off the switching elements SW1 to SW6, the DC power output from the DC power supply P is converted into three AC powers whose phases differ by 120 degrees from each other, and these AC powers are applied to the three phases of the motor M. (U phase, V phase, and W phase) are input to the input terminals, and the rotor of the electric motor M rotates.

The current sensors Se1 to Se3 are composed of Hall elements, shunt resistors, and the like. Current sensor Se1 detects alternating current Iu flowing in the U phase of electric motor M and outputs it to control device 3. Current sensor Se2 detects alternating current Iv flowing in the V phase of electric motor M and outputs it to control device 3. Current sensor Se3 detects alternating current Iw flowing in the W phase of electric motor M and outputs it to control device 3. Note that if the alternating currents Iu, Iv, and Iw are not particularly distinguished, they are simply referred to as alternating current I.

Control device 3 controls inverter circuit 2 . When the modulation rate is less than a predetermined value, the control device 3 adjusts the pulse width of the drive signals S1 to S6 by increasing or decreasing the amplitude of the voltage command values Vu*, Vv*, Vw* within a range that is equal to or less than the amplitude of the carrier wave. PWM control is performed to change the modulation rate, and when the modulation rate is greater than a predetermined value, overmodulation control is performed to make the amplitudes of voltage command values Vu*, Vv*, and Vw* larger than the amplitude of the carrier wave. Overmodulation control is a control method that utilizes the overmodulation region of the inverter circuit 2. The control device 3 includes a drive circuit 4 and a calculation section 5.

The drive circuit 4 is composed of an IC (Integrated Circuit) or the like. The drive circuit 4 compares the voltage command values Vu*, Vv*, Vw* output from the calculation unit 5 with a carrier wave (triangular wave, sawtooth wave, reverse sawtooth wave, etc.), and generates a drive signal according to the comparison result. S1 to S6 are output to respective gate terminals of switching elements SW1 to SW6.

For example, the drive circuit 4 outputs a high-level drive signal S1 and a low-level drive signal S2 when the voltage command value Vu* is greater than or equal to the carrier wave, and when the voltage command value Vu* is smaller than the carrier wave, the drive circuit 4 outputs a high-level drive signal S1 and a low-level drive signal S2. , outputs a low-level drive signal S1, and outputs a high-level drive signal S2. Further, the drive circuit 4 outputs a high-level drive signal S3 and a low-level drive signal S4 when the voltage command value Vv* is greater than or equal to the carrier wave, and when the voltage command value Vv* is smaller than the carrier wave, the drive circuit 4 outputs a high-level drive signal S3 and a low-level drive signal S4. , outputs a low-level drive signal S3, and outputs a high-level drive signal S4. Further, the drive circuit 4 outputs a high-level drive signal S5 and a low-level drive signal S6 when the voltage command value Vw* is greater than or equal to the carrier wave, and when the voltage command value Vw* is smaller than the carrier wave, the drive circuit 4 outputs a high-level drive signal S5 and a low-level drive signal S6. , outputs a low-level drive signal S5, and also outputs a high-level drive signal S6.

The calculation unit 5 includes a first coordinate conversion unit (first conversion unit) 6, an estimation unit 7, a subtraction unit 8, a speed control unit 9, a command value output unit 10, subtraction units 11 and 12, and a current It includes a control section 13, a restriction section 14, a second coordinate transformation section (second transformation section) 15, a specification section 16, and a time constant change section 17.

The calculation unit 5 is an electronic control unit including a CPU (Central Processing Unit), a ROM (Read Only Memory), a RAM (Random Access Memory), and the like. The calculation unit 5 can be configured as software, for example, in which a program stored in a ROM is loaded onto a RAM and executed by a CPU. In the calculation unit 5, by executing the program, the first coordinate conversion unit 6, the estimation unit 7, the subtraction unit 8, the speed control unit 9, the command value output unit 10, the subtraction units 11 and 12, the current control unit 13, and the limit section 14, second coordinate transformation section 15, specifying section 16, and time constant changing section 17 are realized.

The first coordinate conversion unit 6 converts alternating currents Iu, Iv, and Iw flowing through the electric motor M into a d-axis current Id and a q-axis current Iq. The first coordinate conversion unit 6 uses the position θ^ output from the estimation unit 7 to convert the alternating currents Iu, Iv, and Iw detected by the current sensors Se1 to Se3 into a d-axis current Id and a q-axis current Iq. Convert. The first coordinate conversion unit 6 converts the alternating currents Iu, Iv, and Iw into a d-axis current Id and a q-axis current Iq using, for example, a conversion matrix C1 shown in equation (1) below. The first coordinate conversion unit 6 outputs the d-axis current Id and the q-axis current Iq to the subtraction units 11 and 12 and the identification unit 16.

The estimation unit 7 calculates a d-axis voltage command value Vd* and a q-axis voltage command value Vq* output from a filter 18, which will be described later, as well as a d-axis current Id and a q-axis current Iq output from the first coordinate conversion unit 6. The rotational speed (rotational speed) ω^ and position θ^ of the rotor of the electric motor M are estimated using . The estimation unit 7 calculates the back electromotive force ed^ and the back electromotive force eq^, for example, using equations (2) and (3) below.

Ｒは、電動機Ｍに含まれる抵抗である。Ｌは、電動機Ｍに含まれるコイルのインダクタンスである。

R is a resistance included in the electric motor M. L is the inductance of a coil included in the motor M.

The estimating unit 7 calculates the error θe^ using Equation 4 below.

ＫｐはＰＩ（Proportional Integral）制御の比例項の定数である。Ｋｉは、ＰＩ制御の積分項の定数である。 The estimation unit 7 calculates the rotational speed ω^ such that the error θe^ becomes zero in the following equation 5.

Kp is a constant of the proportional term of PI (Proportional Integral) control. Ki is a constant of the integral term of PI control.

The estimation unit 7 calculates the position θ^ using the following equation (6).

The subtraction unit 8 calculates the difference Δω between the rotation speed command value ω* input from the outside and the rotation speed ω^ output from the estimation unit 7.

The speed control section 9 performs PI control. Specifically, the speed control unit 9 generates the q-axis current command value Iq* from the difference Δω output from the subtraction unit 8. The speed control unit 9 calculates the q-axis current command value Iq* such that the difference Δω becomes zero in the following equation (7), for example.

The command value output section 10 outputs a predetermined d-axis current command value Id* to the subtraction section 11.

The subtraction unit 11 calculates the difference ΔId between the d-axis current command value Id* output from the command value output unit 10 and the d-axis current Id output from the first coordinate conversion unit 6. The subtraction unit 11 outputs the difference ΔId to the current control unit 13. The subtraction unit 12 calculates the difference ΔIq between the q-axis current command value Iq* output from the speed control unit 9 and the q-axis current Iq output from the first coordinate conversion unit 6. The subtraction unit 12 outputs the difference ΔIq to the current control unit 13.

Ｌｑは、電動機Ｍに含まれるコイルのｑ軸インダクタンスである。Ｌｄは、電動機Ｍに含まれるコイルのｄ軸インダクタンスである。Ｋｅは、誘起電圧定数である。 The current control unit 13 performs PI control. Specifically, the current control unit 13 generates the d-axis voltage command value Vd* and the q-axis voltage command value Vq* from the difference ΔId output from the subtraction unit 11 and the difference ΔIq output from the subtraction unit 12. . For example, the current control unit 13 calculates the d-axis voltage command value Vd* using the following formula (8), and calculates the q-axis voltage command value Vq* using the following formula (9). The current control unit 13 outputs the d-axis voltage command value Vd* and the q-axis voltage command value Vq* to the restriction unit 14.

Lq is the q-axis inductance of a coil included in the electric motor M. Ld is the d-axis inductance of the coil included in the electric motor M. Ke is an induced voltage constant.

The limiting section 14 includes a filter 18 and a voltage limiter 19 as a voltage limiting section. The limiter 14 receives the d-axis voltage command value Vd* and the q-axis voltage command value Vq*.

The filter 18 is a band elimination filter (BEF). The filter 18 is a variable filter whose cutoff frequency (fc) can be changed by changing the time constant (τ). The time constant of the filter 18 is changed by the time constant changing section 17. The cutoff frequency of the filter 18 is changed by changing the time constant by the time constant changing section 17. The filter 18 has a time constant that makes the gain of the sixth harmonic component smaller than the gain of the harmonic component of a predetermined order below the fifth order, and a time constant that makes the gain of the harmonic component of the predetermined order below the fifth order smaller than the gain of the harmonic component of a predetermined order below the fifth order. It is possible to change the time constant to be smaller than the gain of the harmonic component of . The variable filter of the present invention has a time constant that makes the gain of at least the sixth harmonic component smaller than the gain of the harmonic component of a predetermined order of the fifth order or lower, and a time constant that makes the gain of the harmonic component of the predetermined order of the fifth order or lower. It is sufficient if it is possible to change the gain to a time constant that makes the gain of the sixth harmonic component smaller than the gain of the sixth harmonic component.

The filter 18 of this embodiment has a time constant that makes the gain of the 6th harmonic component smaller than the gain of the 3rd harmonic component, and a time constant that makes the gain of the 3rd harmonic component smaller than the gain of the 6th harmonic component. It is possible to change the time constant to be smaller than . The filter 18 outputs the filtered d-axis voltage command value Vd* and q-axis voltage command value Vq* to the voltage limiter 19.

The voltage limiter 19 sets the d-axis voltage command value Vd* so that when the d-axis voltage command value Vd* and the q-axis voltage command value Vq* exceed the input voltage of the inverter circuit 2, the d-axis voltage command value Vd* becomes equal to or less than the input voltage of the inverter circuit 2. and/or limit the q-axis voltage command value Vq*. The voltage limiter 19 of this embodiment sets the input voltage of the inverter circuit 2 or less when the filtered d-axis voltage command value Vd* and q-axis voltage command value Vq* exceed the input voltage of the inverter circuit 2. The filtered d-axis voltage command value Vd* and/or q-axis voltage command value Vq* is limited.

Here, the filtered d-axis voltage command value Vd* and q-axis voltage command value Vq* in this embodiment are the d-axis voltage command value Vd* and the q-axis voltage command value Vq output from the current control unit 13. * has been subjected to filter processing, so it can be said that they are the d-axis voltage command value Vd* and the q-axis voltage command value Vq* output from the current control unit 13.

In the voltage limiter 19, various known methods can be used to limit the d-axis voltage command value Vd* and/or the q-axis voltage command value Vq*. Examples of the limiter include a fixed phase limiter that limits only the voltage amplitude while maintaining the voltage phase, and a d-axis priority limiter that limits only the q-axis voltage while maintaining the d-axis voltage. The voltage limiter 19 controls the d-axis voltage command value when limit processing is necessary (when the filtered d-axis voltage command value Vd* and q-axis voltage command value Vq* exceed the input voltage of the inverter circuit 2). At least one of Vd* and the q-axis voltage command value Vq* is subjected to limit processing and output to the second coordinate conversion section 15. The voltage limiter 19 outputs the input d-axis voltage command value Vd* and q-axis voltage command value Vq* to the second coordinate conversion unit 15 when limit processing is not necessary.

Here, the d-axis voltage command value Vd* and q-axis voltage command value Vq* output from the current control unit 13, the filtered d-axis voltage command value Vd* and q-axis voltage command value Vq*, and the limit-processed Strictly speaking, the d-axis voltage command value Vd* and the q-axis voltage command value Vq* are different values, but in the present invention, the three d-axis voltage command values Vd* and the q-axis voltage command value Vq* are expressed as a d-axis voltage command value Vd* and a q-axis voltage command value Vq*.

The second coordinate conversion unit 15 receives the d-axis voltage command value Vd* and the q-axis voltage command value Vq* output from the restriction unit 14 as input. The second coordinate conversion unit 15 of this embodiment receives the d-axis voltage command value Vd* and the q-axis voltage command value Vq* output from the voltage limiter 19. The second coordinate conversion unit 15 converts the input d-axis voltage command value Vd* and q-axis voltage command value Vq* into the U-phase and V-phase of the motor M using the position θ^ output from the estimation unit 7. , and into voltage command values Vv*, Vv*, and Vw* corresponding to the W phase. The second coordinate conversion unit 15 outputs voltage command values Vv*, Vv*, and Vw* to the drive circuit 4.

The second coordinate conversion unit 15 converts the d-axis voltage command value Vd* and the q-axis voltage command value Vq* into voltage command values Vu*, Vv*, Convert to Vw*.

The second coordinate conversion unit 15 outputs voltage command values Vu*, Vv*, Vw* to the drive circuit 4.

The identifying unit 16 compares the amplitude of the 6th harmonic component and the amplitude of the 3rd harmonic component superimposed on the d-axis current Id and/or the q-axis current Iq, and selects the harmonic component with the larger amplitude. Identify. As shown in FIG. 2, the identification unit 16 includes a first filter 160 and a second filter 161. The first filter 160 receives the d-axis current Id and/or the q-axis current Iq and passes only the sixth harmonic component. That is, it can be said that the first filter 160 has a time constant that makes the gain of the third harmonic component smaller than the gain of the sixth harmonic component. The second filter 161 receives the d-axis current Id and/or the q-axis current Iq and passes only the third-order harmonic component. That is, it can be said that the second filter 161 has a time constant that makes the gain of the sixth harmonic component smaller than the gain of the third harmonic component.

Note that the time constant of the first filter of the present invention is at least determined by a predetermined harmonic of a predetermined order whose amplitude is larger than the amplitude of the sixth harmonic component among harmonic components of the fifth order or lower superimposed on the dq coordinate system. Any time constant may be used as long as it makes the gain of the wave component smaller than the gain of the sixth harmonic component. Further, the time constant of the second filter of the present invention is such that at least the gain of the sixth harmonic component is determined by the gain of the sixth harmonic component among the harmonic components of the fifth order or lower that are superimposed on the dq coordinate system. Any time constant may be used as long as it is smaller than the gain of a harmonic component of a predetermined order that is larger than the amplitude of .

The identifying unit 16 identifies which one is larger, the third harmonic component or the sixth harmonic component, based on the amplitude of the output signal from the first filter 160 and the amplitude of the output signal from the second filter 161. . That is, the specifying unit 16 determines when the amplitude of the 6th harmonic component is the largest and when the amplitude of the 3rd harmonic component is the largest among the harmonic components of the 6th order and below superimposed on the dq coordinate system. Harmonic components that are provided in the control device 3 and are superimposed on the d-axis current Id and q-axis current Iq that are obtained by converting the alternating currents Iu, Iv, and Iw flowing through each phase of the motor M, which may be large. Among them, the harmonic component with the largest amplitude is specified. Therefore, it can be said that the identifying unit 16 identifies the harmonic component having the largest amplitude among the harmonic components superimposed on the current flowing through each phase of the motor M.

The identifying unit 16 compares the amplitude of the output signal from the first filter 160 and the amplitude of the output signal from the second filter 161, and if the amplitude of the output signal from the first filter 160 is larger, The time constant changing unit 17, which will be described later, sets the time constant of the filter 18 to a time constant that makes the gain of the 6th harmonic component smaller than the gain of the 3rd harmonic component. Send instruction signals. Further, as a result of comparing the amplitude of the output signal from the first filter 160 and the amplitude of the output signal from the second filter 161, the identification unit 16 determines that when the amplitude of the output signal from the second filter 161 is larger, The time constant changing unit 17, which will be described later, sets the time constant of the filter 18 to a time constant that makes the gain of the third harmonic component smaller than the gain of the sixth harmonic component. send an instruction signal to.

The time constant changing section 17 changes the time constant of the filter 18 based on the instruction signal from the specifying section 16 .

As explained above, in the control device 3 of the control system 1 according to the present embodiment, the identification unit 16 determines the amplitude of the sixth harmonic component and the amplitude of the third harmonic component superimposed on the dq coordinate system. , and identify the harmonic component with larger amplitude. The time constant changing section 17 changes the time constant of the filter 18 based on the instruction signal from the specifying section 16 . As a result, in the control device 3, for example, if the sixth harmonic component is dominant in the harmonic components superimposed on the dq coordinate system, the filter 18 can attenuate the sixth harmonic component. If the harmonic components of the predetermined order below the fifth order are dominant among the harmonic components superimposed on the dq coordinate system, the filter 18 filters out the harmonic components of the predetermined order below the fifth order. It can be attenuated. In this way, the control device 3 can attenuate not only the 6th harmonic component superimposed on the dq coordinate system, but also the harmonic components of a predetermined order below the 5th order superimposed on the dq coordinate system. I can do it. Therefore, in the control device 3, when controlling the electric motor M by vector control in the overmodulation region of the inverter circuit 2, it is possible to reduce torque ripple.

In the control system 1 according to the present embodiment, the identification unit 16 of the control device 3 includes a first filter 160 that passes only the sixth harmonic component, and a second filter 161 that passes only the third harmonic component. , has. The identification unit 16 compares the amplitude of the output current of the first filter 160 and the amplitude of the output current of the second filter 161, and identifies the harmonic component with the larger amplitude. With this configuration, among the harmonic components superimposed on the current flowing through each phase of the electric motor M, the harmonic component having the largest amplitude can be specified.

In the control system 1 according to the present embodiment, the d-axis voltage command value Vd* and the q-axis voltage command value Vq* are input to the filter 18 of the control device 3. With this configuration, it is possible to improve responsiveness in vector control that performs current feedback control on the dq coordinate system.

Although the embodiments of the present invention have been described above, the present invention is not necessarily limited to the embodiments described above, and various changes can be made without departing from the gist thereof.

In the embodiment described above, the identification unit 16 compares the amplitude of the output current of the first filter 160 and the amplitude of the output current of the second filter 161, and determines which of the 6th harmonic component and the 3rd harmonic component. An example of a mode in which a harmonic component with a larger amplitude is specified has been explained. However, the identification unit may identify the harmonic component with the largest amplitude using another method.

As shown in FIG. 3, in the control system 1A, a signal from the estimation unit 7 is input to the identification unit 16A of the control device 3A. The identifying unit 16A uses the position θ^ output from the estimating unit 7 to obtain the electrical angular frequency of the electric motor M. The specifying unit 16A specifies the harmonic component having the largest amplitude based on the number of local maximum values or local minimum values of the d-axis current Id and/or the q-axis current Iq in one cycle of the electrical angular frequency. For example, when the maximum value of the d-axis current Id and/or the q-axis current Iq occurs six times in one cycle of the electrical angular frequency, the identifying unit 16A identifies that the harmonic component with the largest amplitude is the sixth order. do. For example, if the maximum value of the d-axis current Id and/or the q-axis current Iq occurs three times in one cycle of the electrical angular frequency, the identifying unit 16A identifies that the harmonic component with the largest amplitude is the third order. do. Even in this case, the harmonic component with the largest amplitude can be identified.

Further, the identifying unit 16A may identify the harmonic component having the largest amplitude in one cycle of the electrical angular frequency based on the slope of the d-axis current Id and/or the q-axis current Iq. Even in this case, the harmonic component with the largest amplitude can be identified.

In the embodiment described above, the estimator 7 calculates the position θ^ as an example. However, as shown in FIG. 4, the control system 1B may include a position detection section Sp. The position detection unit Sp detects the position θ of the rotor of the electric motor M, and outputs the position θ to the control device 3. The position detection unit Sp is, for example, a resolver.

In the control system 1B, the first coordinate conversion unit 6B of the control device 3B uses the position θ detected by the position detection unit Sp to convert the alternating currents Iu, Iv detected by the current sensors Se1 to Se3 in each control period. , Iw into a d-axis current Id and a q-axis current Iq. The first coordinate conversion unit 6B converts the alternating currents Iu, Iv, and Iw into a d-axis current Id and a q-axis current Iq using the conversion matrix C1 shown in equation (1) above. In addition, in the above equation (1), the position θ^ is replaced with the position θ.

The estimation unit 7B estimates the rotational speed ω^ of the rotor of the electric motor M using the position θ detected by the position detection unit Sp for each control period. For example, the estimation unit 7B estimates the rotational speed ω^ by dividing the position θ by the control period of the control device 3.

The second coordinate conversion unit 15B converts the d-axis voltage command value Vd* and the q-axis voltage command value Vq* into the voltage command value Vu*, using the position θ detected by the position detection unit Sp for each control cycle. The voltage command value Vu* and the voltage command value Vw* are converted. The second coordinate conversion unit 15B converts the d-axis voltage command value Vd* and the q-axis voltage command value Vq* into the voltage command value Vu* and the voltage command value Vv* using the transformation matrix C2 shown in the above formula (10). , and converted into a voltage command value Vw*. In addition, in the above equation (10), the position θ^ is replaced with the position θ.

In the embodiments described above, the identification units 16 and 16A use the d-axis current Id and the q-axis current Iq to identify the harmonic component with the largest amplitude. However, the identification units 16 and 16A may use at least one of the alternating currents Iu, Iv, and Iw flowing through each phase of the motor M to identify the harmonic component having the largest amplitude.

The harmonic components superimposed on the alternating currents Iu, Iv, and Iw flowing through each phase of the motor M are as follows: Among the harmonic components of the seventh order and below, the amplitude of the fifth harmonic component is the largest, and the case where the amplitude of the seventh harmonic component is the largest. There are cases in which the amplitude of harmonic components is the largest, and cases in which the amplitudes of harmonic components of the fourth order and below are the largest. In alternating currents Iu, Iv, Iw, when the amplitude of at least one of the fifth-order harmonic component and the seventh-order harmonic component is the highest among the seventh-order harmonic components and below, the d-q coordinate Among the harmonic components superimposed on the system, the sixth harmonic component has the largest amplitude. Therefore, the specifying units 16 and 16A use the alternating currents Iu, Iv, and Iw to specify the harmonic component with the largest amplitude among the harmonic components of the seventh order or lower that are superimposed on the alternating currents Iu, Iv, and Iw. In this case, a filter that passes only the 5th harmonic component, a filter that passes only the 7th harmonic component, and a filter that passes at least one harmonic component of the 4th harmonic component or lower. , it is sufficient to have the following. Even in this case, it is possible to specify the harmonic component with the largest amplitude among the harmonic components superimposed on the current flowing through each phase of the motor M.

In the above embodiment, the d-axis voltage command value Vd* and the q-axis voltage command value Vq* that have been filtered by the filter 18 are input to the voltage limiter 19 as an example, but the present invention is not limited thereto. For example, the d-axis voltage command value Vd* and the q-axis voltage command value Vq* generated by the current control unit 13 may be input to the voltage limiter 19. In this case, the d-axis voltage command value Vd* and the q-axis voltage command value Vq* output from the voltage limiter 19 are output to the filter 18.

In the above embodiment, the filter 18 receives the d-axis voltage command value Vd* and the q-axis voltage command value Vq* output from the current control unit 13, and filters the d-axis voltage command value Vd* and q Although the embodiment has been described using an example in which the shaft voltage command value Vq* is output to the voltage limiter 19, the present invention is not limited thereto. The filter 18 makes the amplitude of the 6th harmonic component superimposed on the dq coordinate system smaller than the amplitude of the harmonic component of a predetermined order below the 5th order superimposed on the dq coordinate system, and also It may be provided anywhere in the control device 3 as long as the amplitude of the harmonic component of a predetermined order below the fifth order superimposed on the q coordinate system can be made smaller than the amplitude of the sixth harmonic component.

In the embodiment described above, the estimation unit 7 uses the d-axis voltage command value Vd* and the q-axis voltage command value Vq* processed by the filter 18 to calculate the rotational speed (rotation speed) ω^ and position θ of the rotor of the electric motor M. Although the method for estimating ^ has been described as an example, the present invention is not limited to this. For example, when estimating the rotation speed (rotation speed) ω^ and position θ^ of the rotor of the electric motor M using the d-axis voltage command value Vd* and the q-axis voltage command value Vq* output from the voltage limiter 19. There may be.

The position estimation accuracy in the estimation unit 7 is improved by estimating the position using a value close to the voltage finally applied to the inverter circuit 2.The filter 18 processed d-axis voltage command value Vd* and q-axis voltage Preferably, it is the estimating unit 7 that estimates the rotational speed (rotation speed) ω^ and position θ^ of the rotor of the electric motor M using the command value Vq*.

In the above embodiment, the filter 18 is a band-elimination filter (BEF). However, the filter 18 may be composed of a low-pass filter and a high-pass filter.

2... Inverter circuit (inverter), 3, 3A, 3B... Control device, 6... First coordinate conversion section (first conversion section), 13... Current control section, 14... Limiting section, 15... Second coordinate conversion section ( 16, 16A... Specification unit, 17... Time constant changing unit, 18... Filter (variable filter), 19... Voltage limiter (voltage limiting unit), 160... First filter, 161... Second filter, Id...d-axis current, Id*...d-axis current command value, Iq...q-axis current, Iq*...q-axis current command value, Iu, Iv, Iw...current, M...motor, Vd*...d-axis voltage command value , Vq*...q-axis voltage command value, Vu, Vv, Vw... voltage command value.

Claims (4)

a first conversion unit that converts the current flowing through each phase of the motor into a d-axis current and a q-axis current;
A d-axis voltage command value and a q-axis voltage command value are generated based on the d-axis current and the q-axis current converted by the first converter, and the d-axis current command value and the q-axis current command value. a current control section;
The d-axis voltage command value and the q-axis voltage command value are input, and when the d-axis voltage command value and the q-axis voltage command value exceed the input voltage of the inverter, they become equal to or less than the input voltage. a limiting unit including a voltage limiting unit that limits the d-axis voltage command value and/or the q-axis voltage command value; and a variable filter whose time constant can be changed;
a second conversion unit that converts the d-axis voltage command value and the q-axis voltage command value output from the restriction unit into voltage command values corresponding to each phase of the motor, and outputs the voltage command value;
a specifying unit that specifies a harmonic component having the largest amplitude among harmonic components superimposed on the current flowing through each phase of the electric motor;
a time constant changing unit that changes the time constant of the variable filter so as to reduce the gain of a harmonic component corresponding to the harmonic component identified by the identifying unit, and control for overmodulating the inverter. A device,
The variable filter has a time constant that makes the gain of the sixth harmonic component smaller than the gain of the harmonic component of a predetermined order below the fifth order, and a time constant that makes the gain of the harmonic component of the predetermined order smaller than the gain of the harmonic component of the predetermined order. and a time constant that is smaller than the gain of the wave component. The specific part is
a first filter into which at least one of the d-axis current and the q-axis current is input and has a time constant that makes the gain of the harmonic component of the predetermined order smaller than the gain of the sixth harmonic component; a second filter to which at least one of the d-axis current and the q-axis current is input and has a time constant that makes the gain of the sixth harmonic component smaller than the gain of the harmonic component of the predetermined order; has
The control device according to claim 1, wherein the amplitude of the output current of the first filter and the amplitude of the output current of the second filter are compared, and a harmonic component with a large amplitude is identified. The identification unit acquires the electrical angular frequency of the motor, and detects the current flowing in at least one phase of the currents flowing in each phase of the motor in one period of the electrical angular frequency, or the d-axis current and the q-axis current. The control device according to claim 1, wherein the harmonic component having the largest amplitude is specified based on the number of local maximum values or local minimum values of at least one of the currents. The control device according to claim 1, wherein the variable filter receives the d-axis voltage command value and the q-axis voltage command value.

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