JP2023141038A - Radar device - Google Patents

Abstract

To provide a radar device capable of improving the detection accuracy of targets.SOLUTION: The radar device includes: multiple transmitting antennas including a first transmitting antenna that emits first linearly polarized waves, and a second transmitting antenna that is placed adjacent to the first transmitting antenna that emits second linearly polarized wave different from the first linear polarization; and a transmitting circuit that multiples and transmits transmission signals with a phase rotation amount that differs by ξ or -ξ between the first transmitting antenna and the second transmitting antenna in each transmission cycle from the multiple transmitting antennas.SELECTED DRAWING: Figure 1

Description

The present disclosure relates to a radar device.

In recent years, studies have been underway on radar devices that use radar transmission signals with short wavelengths, including microwaves or millimeter waves, which can provide high resolution. Furthermore, in order to improve outdoor safety, there is a need to develop a radar device (for example, called a wide-angle radar device) that can detect small objects such as pedestrians in a wide-angle range in addition to vehicles.

As a configuration of a radar device having a wide-angle detection range, for example, reflected waves from a target are received by an array antenna consisting of multiple antennas (also called antenna elements), and the receiving position is determined with respect to the element spacing (antenna spacing). There is a configuration that uses a method (Direction of Arrival (DOA) estimation) that estimates the direction in which a reflected wave arrives (or called the angle of arrival) using a signal processing algorithm based on phase difference. For example, angle-of-arrival estimation methods include the Fourier method, and methods that provide high resolution include the Capon method, MUSIC (Multiple Signal Classification), and ESPRIT (Estimation of Signal Parameters via Rotational Invariance Techniques).

In addition, as a radar device, for example, in addition to the receiving section, the transmitting section is also equipped with multiple antennas (array antennas), and the beam scanning is performed by signal processing using the transmitting and receiving array antenna (MIMO (Multiple Input Multiple Output) radar). ) has been proposed (for example, see Non-Patent Document 1).

Special Publication No. 2011-526371 US Patent No. 9,541,638 US Patent Publication No. 2019/0064337 US Patent Publication No. 2020/0363497 Japanese Patent Application Publication No. 2020-148754

J. Li, and P. Stoica, "MIMO Radar with Colocated Antennas", Signal Processing Magazine, IEEE Vol. 24, Issue: 5, pp. 106-114, 2007 M. Kronauge, H. Rohling, "Fast two-dimensional CFAR procedure", IEEE Trans. Aerosp. Electron. Syst., 2013, 49, (3), pp. 1817-1823 Direction-of-arrival estimation using signal subspace modeling Cadzow, J.A.; Aerospace and Electronic Systems, IEEE Transactions on Volume: 28 , Issue: 1 Publication Year: 1992 , Page(s): 64 - 79

However, a method for detecting an object (or target) in a radar device (for example, a MIMO radar) has not been sufficiently studied.

Non-limiting embodiments of the present disclosure contribute to providing a radar device that improves target detection accuracy.

A radar device according to an embodiment of the present disclosure includes a first transmitting antenna that emits a first linearly polarized wave, and a second transmitting antenna that is adjacent to the first transmitting antenna and that is different from the first linearly polarized wave. a plurality of transmitting antennas including a second transmitting antenna that radiates linearly polarized waves, and a phase rotation in which the phase between the first transmitting antenna and the second transmitting antenna in each transmission period is different by φ or -φ. and a transmitting circuit that multiplex transmits the transmission signal to which the amount of transmission is assigned from the plurality of transmitting antennas.

Note that these comprehensive or specific embodiments may be realized by a system, an apparatus, a method, an integrated circuit, a computer program, or a recording medium. It may be implemented with any combination of media.

According to an embodiment of the present disclosure, it is possible to improve the detection accuracy of a target in a radar device.

Further advantages and effects of an embodiment of the disclosure will become apparent from the description and the drawings. Such advantages and/or effects may be provided by each of the several embodiments and features described in the specification and drawings, but not necessarily all are provided in order to obtain one or more of the same features. There isn't.

Block diagram showing a configuration example of a radar device Diagram showing an example of a transmission signal when using chirped pulses Diagram showing an example of Doppler shift amount and orthogonal code assignment Diagram showing an example of Doppler shift amount and orthogonal code assignment Diagram showing an example of Doppler shift amount and orthogonal code assignment Diagram showing an example of a transmitting antenna Diagram showing an example of a transmitting antenna Diagram showing an example of a transmitting antenna Diagram showing an example of the directivity of left-handed circularly polarized waves Diagram showing an example of directivity of right-handed circularly polarized waves Diagram showing an example of a transmitted signal and a received signal when chirped pulses are used Diagram showing an example of Doppler region compression processing Diagram showing an example of Doppler shift amount and orthogonal code assignment Diagram showing an example of Doppler aliasing determination Diagram showing an example of antenna arrangement Diagram showing an example of antenna arrangement Diagram showing an example of antenna arrangement Diagram showing an example of antenna arrangement Block diagram showing a configuration example of a radar device Diagram showing an example of antenna arrangement Block diagram showing a configuration example of a radar device Diagram showing an example of transmission signal switching control Diagram showing an example of transmission signal switching control Diagram showing an example of a transmission switching control table Block diagram showing a configuration example of a radar device Diagram showing an example of a Doppler multiple assignment table Diagram showing an example of a Doppler multiple assignment table Diagram showing an example of a Doppler multiple assignment table

MIMO radar transmits multiplexed signals (radar transmission waves) using, for example, time division, frequency division, or code division from multiple transmission antennas (or called transmission array antennas), and transmits signals reflected from surrounding objects ( A plurality of reception antennas (or called reception array antennas) are used to receive the radar reflected waves), and a multiplexed transmission signal is separated from each reception signal and received. Through such processing, the MIMO radar performs array signal processing on these received signals as a virtual reception array.

Furthermore, in MIMO radar, by appropriately arranging the element spacing in the transmitting and receiving array antenna, it is possible to enlarge the antenna aperture of the virtual receiving array and improve the angular resolution. Alternatively, in MIMO radar, side lobes or grating lobes can be suppressed by arranging the antennas of the virtual receiving array closer together.

For example, Patent Document 1 describes a MIMO radar (hereinafter referred to as "time division multiplex MIMO radar") that uses time division multiplex transmission in which signals are transmitted by shifting the transmission time for each transmitting antenna as a multiplex transmission method of MIMO radar. ) is disclosed. A time division multiplexing MIMO radar outputs a transmission pulse, which is an example of a transmission signal, while sequentially switching transmission antennas at a prescribed period. A time-division multiplexing MIMO radar receives a signal in which a transmitted pulse is reflected by an object using multiple receiving antennas, and performs, for example, spatial FFT (Fast Fourier Transforma) processing (reflection wave arrival direction estimation process).

A time division multiplexing MIMO radar sequentially switches transmission antennas that transmit transmission signals (for example, transmission pulses or radar transmission waves) at a prescribed cycle. As a result, the time division multiplexing MIMO radar can extract the propagation path response represented by the product of the number of transmitting antennas Nt and the number of receiving antennas Na (=Nt × Na), and these (Nt × Na) reception Array signal processing is performed using the signal as a virtual receiving array. For example, it is difficult to use transmitting antennas that exceed the number of transmitting antennas (for example, the number of time-division multiplexing) that multiplex transmit signals by switching in time division. For example, when a radar device transmits transmission signals with a time division multiplexing number Nt using Nt transmission antennas, it is difficult to extract more than (Nt×Na) propagation path responses. Therefore, if the number of antennas is restricted due to constraints such as the cost or installation location of the radar device, the angular resolution or sidelobe suppression effect will be limited and the angle measurement performance may not be improved.

Next, as an example, we will focus on a method of simultaneously multiplexing and transmitting transmission signals from a plurality of transmission antennas.

As a method for simultaneously multiplexing and transmitting transmission signals from multiple transmitting antennas, for example, there is a method of transmitting signals such that multiple transmission signals can be separated in the Doppler frequency domain at the receiving section (hereinafter referred to as "Doppler multiplex transmission"). ) (for example, see Non-Patent Document 2).

In Doppler multiplex transmission, for example, the transmitting section applies different Doppler shift amounts to the transmission signals transmitted from the reference transmission antenna and to the transmission signals transmitted from the transmission antennas different from the reference transmission antenna. and transmit signals are simultaneously transmitted from a plurality of (for example, Nt) transmit antennas. In Doppler multiplex transmission, signals received using a plurality of (for example, Na) receiving antennas are filtered in the Doppler frequency domain, so that the transmission signals transmitted from each transmitting antenna are separated and received. As a result, a MIMO radar using Doppler multiplex transmission (hereinafter referred to as "Doppler multiplex MIMO radar") has a propagation path response represented by the product of the number of transmitting antennas Nt and the number of receiving antennas Na (=Nt×Na). These (Nt×Na) received signals are used as a virtual reception array to perform array signal processing. For example, it is difficult to use transmitting antennas that exceed the number of transmitting antennas for Doppler multiplex transmission (for example, the number of Doppler multiplexes). For example, when a radar device transmits a transmission signal with a Doppler multiplex number Nt using Nt transmission antennas, it is difficult to extract more than (Nt×Na) propagation path responses.

Further, as another method for simultaneously multiplexing and transmitting transmission signals from a plurality of transmission antennas, there is code multiplex transmission (see, for example, Patent Document 2). For example, a MIMO radar using code multiplexing (hereinafter referred to as a "code multiplexing MIMO radar") uses a different code string (hereinafter referred to as a code or (also referred to as a code sequence) is repeatedly applied to perform code multiplex transmission from a plurality of (for example, Nt) transmitting antennas. Further, the code-multiplexed MIMO radar extracts distance information of the code-multiplexed received signal, for example, by performing detection processing on signals received using a plurality of (for example, Na) receiving antennas. In addition, the code-multiplexed MIMO radar, for example, divides the distance information obtained each time the transmission signal is repeatedly transmitted into M pieces and performs Fourier transform processing in the velocity direction (M is, for example, the code length of the code string). ). The code multiplexing MIMO radar adds phase correction based on the detected velocity component to the M Fourier transform processing results in the velocity direction, and multiplies it by an inverse code string that separates the code string assigned to each transmitting antenna. Separates the code-multiplexed received signal.

With this configuration of the code-multiplexed MIMO radar, for example, even if the relative speed between the target and the code-multiplexed MIMO radar is not zero, the code-multiplexed MIMO radar can suppress mutual interference between code-multiplexed received signals. , code-multiplexed received signals can be separated. As a result, the code multiplexing MIMO radar can extract the propagation path response represented by the product of the number of transmitting antennas Nt and the number of receiving antennas Na (=Nt×Na), and these (Nt×Na) received signals Array signal processing is performed using the array as a virtual receiving array. For example, it is difficult to use transmitting antennas that exceed the number of transmitting antennas for code multiplexing transmission (for example, the number of code multiplexing). For example, when a radar device transmits a transmission signal with a code multiplex number Nt using Nt transmission antennas, it is difficult to extract more than (Nt×Na) propagation path responses.

Furthermore, there is a technique for improving radar detection performance or identification performance by using antennas that emit radio waves with different polarizations or antennas that receive radio waves with different polarizations (for example, see Patent Documents 3 and 4). ). A radar device that uses multiple polarized waves is also called, for example, a "polarimetric radar."

For example, Patent Document 3 or Patent Document 4 discloses that an antenna using vertical polarization or horizontal polarization transmits a transmission signal, and a signal received by an antenna using vertical polarization or horizontal polarization is used to detect an object. A method for identifying the information is disclosed. Further, Patent Document 4 describes, for example, that a transmission signal is transmitted using an antenna that uses left-handed circularly polarized waves or right-handed circularly polarized waves, and a signal that is received by an antenna that uses left-handed circularly polarized waves or right-handed circularly polarized waves is used. , a method for detecting and identifying objects is disclosed. Note that an antenna that uses linearly polarized waves such as vertically polarized waves or horizontally polarized waves, or circularly polarized waves such as left-handed circularly polarized waves or right-handed circularly polarized waves is also referred to as a "polarized antenna."

Such polarized radars use multiple different types of polarized antennas while improving the detection or identification performance of the radar. For example, four transmitting antennas are used to transmit four types of polarization: vertical and horizontal polarization, as well as left-handed circular polarization and right-handed circular polarization. Furthermore, when configuring a MIMO radar for each polarization, more transmitting antennas are used. For example, in order to perform MIMO multiplex transmission using Nt transmit antennas for each of four polarized waves, 4×Nt transmit antennas are used. For example, in one embodiment according to the present disclosure, a radar device (for example, also referred to as a polarized radar device or a polarized MIMO radar) generates new polarized waves (for example, circularly polarized waves) by a combination of polarized antennas. By generating this, multiplex transmission using more polarizations is performed using fewer transmit antennas. As a result, the radar device according to the embodiment of the present disclosure suppresses the increase in the number of transmitting antennas and uses more virtual receiving antennas to improve the angle measurement performance of the radar device and detect target objects. Accuracy can be improved.

Hereinafter, an embodiment according to an example of the present disclosure will be described in detail with reference to the drawings. Note that in the embodiments, the same components are given the same reference numerals, and the description thereof will be omitted since it is redundant.

In the following, a configuration (for example, a MIMO radar configuration) in which a radar device transmits multiplexed different transmission signals simultaneously from a plurality of transmission antennas in a transmission branch and separates each transmission signal and performs reception processing in a reception branch is described. I will explain about it.

Further, below, as an example, a configuration of a radar system using a frequency-modulated pulse wave such as a chirp pulse (for example, also referred to as chirp pulse transmission (fast chirp modulation)) will be described. However, the modulation method is not limited to frequency modulation. For example, an embodiment of the present disclosure is also applicable to a radar system using a pulse compression radar that transmits a pulse train after phase modulation or amplitude modulation.

Further, the radar device performs, for example, Doppler multiplex transmission. Furthermore, the radar device encodes (for example, CDM) a signal (hereinafter referred to as a "Doppler multiplex transmission signal") to which different phase rotations (e.g., phase shifts) have been added for the number of Doppler multiplexes in Doppler multiplex transmission, for example. (Code Division Multiplexing)) and multiplex transmission (hereinafter referred to as "coded Doppler multiplexing").

[Configuration of radar device]
The radar device 10 in FIG. 1 includes a radar transmitting section (transmitting branch) 100, a radar receiving section (receiving branch) 200, and a positioning output section 300.

The radar transmitting unit 100 generates a radar signal (radar transmission signal), and transmits the radar transmission signal at a specified transmission period (hereinafter referred to as a , called the "radar transmission cycle").

The radar receiving unit 200 receives a reflected wave signal, which is a radar transmission signal reflected by a target (not shown), using a receiving array antenna including a plurality of receiving antennas 202-1 to 202-Na. The radar receiving unit 200 processes the reflected wave signals received by each receiving antenna 202, and detects the presence or absence of a target or estimates the arrival distance, Doppler frequency (for example, relative velocity), and direction of arrival of the reflected wave signal. and outputs information regarding the estimation results (for example, positioning information).

The positioning output unit 300 performs positioning output processing based on information regarding the estimated direction of arrival input from the radar receiving unit 200.

Note that the radar device 10 may be mounted on a moving object such as a vehicle, and the positioning output (for example, information regarding estimation results) from the positioning output unit 300 may be used, for example, in an advanced driving support system (ADAS) that increases collision safety. It may be connected to a control device ECU (Electronic Control Unit) (not shown) such as an Advanced Driver Assistance System (Advanced Driver Assistance System) or an automatic driving system, and used for vehicle drive control or alarm call control.

Further, the radar device 10 may be attached to a relatively high structure (not shown) such as a roadside utility pole or a traffic light, for example. Further, the radar device 10 may be used, for example, as a sensor in a support system that increases the safety of passing vehicles or pedestrians, or a system for preventing intrusion by suspicious persons (not shown). Further, the positioning output of the radar receiving unit 200 may be connected to a control device (not shown) in a support system for increasing safety or a system for preventing intrusion of suspicious persons, and may be used for alarm call control or abnormality detection control. . Note that the uses of the radar device 10 are not limited to these, and may be used for other uses.

Further, the target object is an object to be detected by the radar device 10, and includes, for example, a vehicle (including four wheels and two wheels), a person, a block, or a curb.

[Configuration of radar transmitter 100]
Radar transmission section 100 includes radar transmission signal generation section 101 , phase rotation amount setting section 105 , phase rotation section 108 , and transmission antenna 109 .

Radar transmission signal generation section 101 generates a radar transmission signal. The radar transmission signal generation section 101 includes, for example, a transmission signal generation control section 102, a modulation signal generation section 103, and a VCO (Voltage Controlled Oscillator) 104. Each component in radar transmission signal generation section 101 will be explained below.

For example, the transmission signal generation control unit 102 sets the transmission signal generation timing for each radar transmission cycle, and transmits information regarding the set transmission signal generation timing to the modulation signal generation unit 103 and the phase rotation amount setting unit 105 (for example, Doppler shift output to the setting section 106). Here, let the radar transmission cycle be "Tr".

The modulated signal generation section 103 periodically generates, for example, a sawtooth-shaped modulation signal, based on information regarding the transmission signal generation timing for each radar transmission period Tr, which is input from the transmission signal generation control section 102.

Based on the modulation signal input from the modulation signal generation unit 103, the VCO 104 generates a frequency modulation signal (hereinafter, for example, a frequency chirp signal or a chirp signal) as a radar transmission signal (radar transmission wave) as shown in FIG. ) is output to the phase rotation section 108 and the radar reception section 200 (mixer section 204 described later).

The phase rotation amount setting section 105 adds the amount of rotation to the radar signal for each radar transmission period Tr in the phase rotation section 108 based on the information regarding the transmission signal generation timing for each radar transmission period Tr inputted from the transmission signal generation control section 102. A phase rotation amount (for example, a phase rotation amount corresponding to coded Doppler multiplex transmission) is set. The phase rotation amount setting section 105 includes, for example, a Doppler shift setting section 106 and an encoding section 107.

For example, the Doppler shift setting unit 106 sets the amount of phase rotation corresponding to the amount of Doppler shift to be applied to the radar transmission signal (for example, chirp signal) based on information regarding the transmission signal generation timing for each radar transmission period Tr. do.

The encoding unit 107 sets the amount of phase rotation corresponding to encoding, for example, based on information regarding the transmission signal generation timing for each radar transmission period Tr. For example, the encoding unit 107 calculates the phase rotation amount for the phase rotation unit 108 based on the phase rotation amount input from the Doppler shift setting unit 106 and the phase rotation amount corresponding to encoding, and calculates the phase rotation amount for the phase rotation unit 108. Output to. Furthermore, the encoding unit 107 outputs information regarding the code sequence used for encoding (for example, each element of the orthogonal code sequence) to the radar reception unit 200 (for example, the output switching unit 209).

Note that the encoded Doppler multiplex number for the Doppler multiplex signal set by the encoder 107 does not need to depend on the phase rotation amount (Doppler shift amount) of each transmitting antenna 109 set by the phase rotation unit 108. For example, even if the phase rotation unit 108 sets the phase rotation amount (Doppler shift amount) of a pair of adjacent transmitting antennas 109 to be the same, the encoding unit 107 may set the encoded Doppler multiplexing number to be the same, It may be a different value.

The phase rotation unit 108 applies the amount of phase rotation input from the encoding unit 107 to the chirp signal input from the VCO 104, and outputs the signal after the phase rotation to the transmission antenna 109. For example, the phase rotation unit 108 includes a phase shifter, a phase modulator, etc. (not shown).

The output signal of the phase rotation unit 108 is amplified to a specified transmission power and radiated into space from each transmission antenna 109. For example, a radar transmission signal is multiplexed from a plurality of transmitting antennas 109 by being given a phase rotation amount corresponding to a Doppler shift amount and an orthogonal code sequence.

Next, an example of a method for setting the amount of phase rotation in the amount of phase rotation setting section 105 will be described.

Doppler shift setting section 106 sets a phase rotation amount φ _ndm for providing Doppler shift amount DOP _ndm , and outputs it to encoding section 107 . Here, ndm=1,~, N _DM . N _DM is the set number of different Doppler shift amounts, and is hereinafter referred to as "Doppler multiplexing number."

In radar device 10, since encoding by encoding section 107 is also used, Doppler multiplexing number _NDM may be set to be smaller than number Nt of transmitting antennas 109 used for multiplex transmission. Note that the Doppler multiplex number _NDM is set to 2 or more.

As the Doppler shift amount DOP ₁ , DOP ₂ , ~,DOP _{N_DM} (“N_DM” is also expressed as “N _DM ”), for example, a Doppler shift amount at equal intervals may be set, or a Doppler shift amount at unequal intervals may be set. A Doppler shift amount may be set. Since each Doppler shift amount DOP ₁ , DOP ₂ , ~,DOP _{N_DM} is encoded by the encoding unit 107 described later, for example, 0≦DOP ₁ , DOP ₂ , ~,DOP _{N_DM} <1/(TrL _oc ) may be set to satisfy. Alternatively, the Doppler shift amounts DOP ₁ , DOP ₂ , to DOP _{N_DM} may be set to satisfy equation (1), for example.

Further, for example, the minimum Doppler shift interval Δf _MinInterval between the Doppler shift amounts DOP ₁ , DOP ₂ , to DOP _{N_DM} may satisfy the following equation (2). Note that the Doppler shift interval may be defined by the absolute value of the difference between any two Doppler shift amounts among the Doppler shift amounts DOP ₁ , DOP ₂ , to DOP _{N_DM} . Here, Loc represents the number of code elements. For example, Loc represents the code length of the code used in the encoding unit 107. In the following, Loc=2 is used as an example (an example will be described later).

Furthermore, the phase rotation amount φ _ndm for providing each Doppler shift amount DOP ₁ , DOP ₂ , to DOP _{N_DM} may be allocated as shown in the following equation (3), for example.

In addition, when the Doppler shift amount is set at equal intervals and equal to Δf _MinInterval (hereinafter referred to as "equally spaced Doppler shift amount setting"), the phase rotation amount φ _ndm for applying the Doppler shift amount DOP _ndm is , for example, is assigned as shown in the following equation (4).

Note that the narrower the minimum Doppler shift interval Δf _MinInterval , the more likely it is that interference between Doppler multiplexed signals will occur, and the higher the possibility that the target detection accuracy will be reduced (for example, degraded). Within the range that is satisfied, it is preferable to widen the interval between Doppler shift amounts. For example, when the equality sign holds in equation (2) (for example, Δf _MinInterval =1/(T _r N _DM L _OC )), the interval in the Doppler region between Doppler multiplexed signals can be maximized (hereinafter (referred to as "maximum equidistant Doppler shift amount setting"). In this _{case ,} the Doppler shift amounts DOP ₁ , DOP ₂ , . For example, the phase rotation amount φ _ndm for providing the Doppler shift amount DOP _ndm is assigned as shown in the following equation (5). In addition, below, angles are shown in radian units.

In equation (5), for example, when the Doppler multiplex number N _DM =2, the phase rotation amount φ ₁ =0 that gives the Doppler shift amount DOP ₁ , and the phase rotation amount φ ₂ = that gives the Doppler shift amount DOP ₂ . It becomes π. Similarly, in Equation (5), for example, when the Doppler multiplex number N _DM =4, the phase rotation amount φ ₁ that gives the Doppler shift amount DOP ₁ =0, and the phase rotation amount φ ₂ that gives the Doppler shift amount DOP 2 = ₀ . =π/2, the amount of phase rotation φ ₃ that provides the amount of Doppler shift DOP ₃ =π, and the amount of phase rotation φ ₄ that provides the amount of Doppler shift DOP ₄ =3π/2. For example, the phase rotation amount φ _ndm that provides each Doppler shift amount DOP _ndm is equally spaced.

Note that the allocation of the phase rotation amount for providing the Doppler shift amounts DOP ₁ , DOP ₂ , to DOP _{N_DM} is not limited to this allocation method. For example, the assignment of the amount of phase rotation shown in equation (5) may be shifted. For example, the amount of phase rotation may be assigned as φ _ndm =2π(ndm)/N _DM . Alternatively, using the phase rotation amount assignment table, calculate the phase rotation amount φ ₁ , φ ₂ ,～, φ _{N_DM} for the Doppler shift amount DOP ₁ , DOP ₂ ,～,DOP _NDM (however, " _{N_DM} " is ) may be randomly assigned.

In addition, when setting the equal interval Doppler shift amount, if the denominator of the phase rotation amount φ _ndm shown in equation (4) is set to an integer, and the phase rotation amount is set to an integer value in degree units, the phase rotation amount setting is becomes easier. For example, by setting Δf _MinInterval =1/(T _r (N _DM +N _int )L _OC ), the phase rotation amount φ _ndm shown in equation (4) has a denominator as shown in equation (6) below. Set to an integer value. Additionally, if N _int is set so that the denominator value (N _DM +N _int ) in equation (6) is a divisor of 360, the amount of phase rotation is set to an integer value, and the setting of the amount of phase rotation is It becomes easier.

Here, N _int takes an integer value of 0 or more. For example, in the case of N _DM =7, if N _int =1 is set, φ _ndm =2π(ndm-1)/(N _DM +N _int )=π(ndm-1)/4, and φ ₁ , φ ₂ ,.., φ _{N_DM} are integer values in degree units such as 0°, 45°, 90°, 135°, ..., 270°, respectively, making it easy to set the amount of phase rotation.

Note that when N _int =0 in equation (6), the maximum equidistant Doppler shift amount is set.

The encoding _unit 107 assigns one or less than N CM phase rotation amounts φ ₁ , to φ _{N_DM} for each of the N _DM Doppler shift amounts inputted from the Doppler shift setting unit 106. Set the amount of phase rotation based on a plurality of orthogonal code sequences. Further, the encoding unit 107 sets the amount of phase rotation based on both the amount of Doppler shift and the orthogonal code sequence, for example, the “encoded Doppler phase rotation amount” that generates the encoded Doppler multiplexed signal, and the phase rotation unit 108 Output to.

An example of the operation in encoding section 107 will be described below.

For example, the encoding unit 107 uses N _CM orthogonal code sequences having the code length Loc (eg, code multiplexing number).

In the following, N _CM orthogonal code sequences having the code length Loc will be expressed as Code _ncm = {OC _ncm (1), OC _ncm (2), ~, OC _ncm (Loc)}. OC _ncm (noc) represents the noc-th code element in the ncm-th orthogonal code sequence Code _ncm . Here, noc is the index of the code element, and is noc=1,~,Loc.

In this embodiment, for example, an orthogonal code sequence with code length Loc=2 and the number of codes (for example, code multiplexing number) N _CM =2 may be used. For example, as an orthogonal code sequence of N _CM =2 with code length Loc = 2, Code ₁ = {OC ₁ (1), OC ₁ (2)}, Code ₂ = {OC ₂ (1), OC ₂ (2 )} may be used. Each code element takes a real or complex value. Also, for example, when the noc-th code element OC ₁ (noc) in the first code is the reference, the noc-th code element OC ₂ (noc) in the second code is In the case of +90° and noc=2, code elements whose phases differ by -90° may be used. Alternatively, code elements having different phases of −90° when noc=1 and +90° when noc=2 may be used.

For example, as an orthogonal code sequence of N _CM =2 with code length Loc = 2, Code ₁ = {1, 1} and Code ₂ = {j, -j} satisfy the above relationship (or condition). These are codes that are orthogonal to each other. Examples of codes that satisfy the above-mentioned conditions include {1, 1} and {-j, j}. Alternatively, examples of codes that satisfy the above-mentioned conditions include {1, j} and {j, 1}. Here, j is an imaginary unit (j ² =-1).

Generally expressing the orthogonal code series used in this embodiment, the first code Code ₁ = {A, B} and the second code Code ₂ = {j×A, -j× B} or Code ₂ = {-j×A, j×B}. These codes are orthogonal to each other, and when the noc-th code element OC ₁ (noc) in the first code is the reference, the noc-th code element OC ₂ (noc) in the second code is Code elements with different phases of +90° when noc=1 and -90° when noc=2, or different phases of -90° when noc=1 and +90° when noc=2 Become. Here, A and B are real numbers or complex numbers, and the absolute values of A and B are equal, for example, |A|=|B|.

Note that here, when the noc-th code element OC ₁ (noc) in the first code is the reference, the noc-th code element OC ₂ (noc) in the second code is The phase difference is +90° for noc=2 and -90° for noc=2, or the code element has a different phase of -90° for noc=1 and +90° for noc=2, but the phase difference is not limited to a phase difference of ±90°.

For example, a phase difference of ±ξ may be given to the noc-th code element in the first code and the second code. ξ may be in the range of π/6 to 5π/6 radians (=30° to 150°). For example, when the phase deviation between the transmitting antennas 109 is corrected in advance, a circularly polarized wave whose main beam direction of the transmitting beam changes depending on ξ is generated by beam transmission to be described later. For example, when the beam transmission antenna spacing is λ/2 and ξ=90°, a circularly polarized wave is generated in which the main beam direction is 0° in front. Further, for example, when ξ=30°, circularly polarized waves are generated in a direction in which the main beam direction is shifted by about -15° from the front direction. Further, for example, when ξ=150°, circularly polarized waves are generated in a direction in which the main beam direction is shifted by about +15° from the front direction. Here, λ is the wavelength of the high frequency signal output from the transmitting antenna 109 (if a chirp signal is used, the wavelength at the center frequency of the chirp signal).

For example, if the main beam direction may be different from the front direction (or if the front direction is an angle with a good axial ratio as a circularly polarized wave), the noc-th code element OC ₁ (noc ), the noc-th code element OC ₂ (noc) in the second code has a different phase of +ξ when noc=1 and -ξ when noc=2, or noc= The code elements may have different phases: −ξ when noc=1 and +ξ when noc=2. Here, for example, ξ may be in the range of π/6 to 5π/6 radians (=30° to 150°).

Generally expressing such an orthogonal code sequence, for the first code Code ₁ = {A, B}, the second code is Code ₂ = {exp(jξ)×A, - exp( jξ)×B} or Code ₂ = {- exp(jξ)×A, exp(jξ)×B}. These codes are orthogonal to each other, and when the noc-th code element OC ₁ (noc) in the first code is the reference, the noc-th code element OC ₂ (noc) in the second code is The code elements have different phases in +ξ when noc=1 and -ξ when noc=2, or different radian phases of -ξ when noc=1 and +ξ when noc=2. Here, A and B are real numbers or complex numbers, and the absolute values of A and B are equal, for example, |A|=|B|.

In the following explanation, an example mainly using Code ₁ = {1, 1} and Code ₂ = {j, -j} will be shown, but the example is not limited to this, and Code ₁ = {A, B}, Code ₂ = {j×A, -j×B} or Code ₂ = {-j×A, j×B} sign, or Code ₁ = {A, B}, Code ₂ = {exp(jξ)×A, - exp(jξ)×B} or Code ₂ = {- exp(jξ)×A, exp(jξ)×B} may be used, and similar effects can be obtained.

Furthermore, in addition to this embodiment, Code ₁ = {1, 1} and Code ₂ = {j, -j} are also used in Modification 1 and Modification 2 of Embodiment 1 to be described later. Examples are mainly used, but the example is not limited to Code ₁ = {A, B}, Code ₂ = {j×A, -j×B} or Code ₂ = {-j×A, j×B } or Code ₁ = {A, B}, Code ₂ = {exp(jξ)×A, - exp(jξ)×B} or Code ₂ = {- exp(jξ)×A, exp(jξ )×B} may be used, and the same effect can be obtained. Here, for example, ξ may be in the range of π/6 to 5π/6 radians (=30° to 150°).

In the encoding unit 107, the code multiplex number (hereinafter referred to as the encoded Doppler multiplex number) when encoding the Doppler multiplexed signal using the ndm-th Doppler shift amount DOP _ndm input from the Doppler shift setting unit 106 is determined. It is written as "N _{DOP_CODE} (ndm)". Here, ndm=1,~, N _DM .

For example, the encoding unit 107 determines whether the sum of the encoded Doppler multiplex numbers N _{DOP_CODE} (1), N _{DOP_CODE} (2), ~, and N _{DOP_CODE} (N _DM ) when encoding the Doppler multiplex signal is used for multiplex transmission. The number of encoded Doppler multiplexes N _{DOP_CODE} (ndm) is set to be equal to the number Nt of transmitting antennas 109 to be used. For example, the encoding unit 107 sets the encoded Doppler multiplex number N _{DOP_CODE} (ndm) so as to satisfy the following equation (7). As a result, the radar device 10 is capable of multiplex transmission in the Doppler domain and code domain (hereinafter referred to as coded Doppler multiplex transmission) using the Nt transmitting antennas 109.

Further, the encoding unit 107 uses, for example, the evenly spaced Doppler shift amount setting including the maximum evenly spaced Doppler shift amount setting, and uses the encoded Doppler multiplex number N _{DOP_CODE} (1), N _{DOP_CODE} (2), ~, N _{DOP_CODE} (N _DM ) may be set to include different numbers of encoded Doppler multiplexes ranging from 1 to N _CM . For example, the encoding unit 107 does not set the number of codes to N _CM in all of the coded Doppler multiplex numbers, but increases the coded Doppler multiplex number N _{DOP_CODE} (ndm) corresponding to at least one Doppler shift amount DOP _ndm to N _CM . Set smaller. Therefore, in multiple combinations of Doppler shift amount DOP _ndm and orthogonal code sequences, the number of multiplexes (coded Doppler multiplex number) N _{DOP_CODE} ( _ndm ) by orthogonal code sequences associated with at least one Doppler shift amount DOP ndm is: It may be different from the encoded Doppler multiplex number associated with other Doppler shift amounts. For example, the encoding unit 107 sets the number of encoded Doppler multiplexes for the Doppler multiplex signal to be non-uniform. With this setting, the radar device 10 individually separates the encoded Doppler multiplex-transmitted signals from the plurality of transmitting antennas 109 over a Doppler range of ±1/2 Tr, for example, by aliasing determination processing in the reception processing described later. and can be received.

Alternatively, the encoding unit 107 uses, for example, the evenly spaced Doppler shift amount setting narrower than the maximum evenly spaced Doppler shift amount setting, and uses the encoded Doppler multiplex number N _{DOP_CODE} (1), N _{DOP_CODE} (2), ~ , N _{DOP_CODE} (N _DM ) may be set to include the same number of encoded Doppler multiplex numbers in the range from 1 to N _CM . For example, the encoding unit 107 may set the number of codes N _CM in all the encoded Doppler multiplex numbers. Therefore, in multiple combinations of Doppler shift amount DOP _ndm and orthogonal code sequences, the number of multiplexes (coded Doppler multiplex number) N _{DOP_CODE} ( _ndm ) by orthogonal code sequences associated with each Doppler shift amount DOP ndm may be the same. . For example, the encoding unit 107 uniformly sets the number of encoded Doppler multiplexes for Doppler multiplex signals. With this setting, the radar device 10 performs coded Doppler multiplex transmission from the plurality of transmitting antennas 109 over a Doppler range of ±1/(2×Loc×Tr), for example, by loopback determination processing in the reception processing described later. signals can be separated and received individually.

Alternatively, the encoding unit 107 uses, for example, the maximum equidistant Doppler shift amount setting to set the encoded Doppler multiplex number N _{DOP_CODE} (1), N _{DOP_CODE} (2), ~, N _{DOP_CODE} (N _DM ) to 1 or more. It may be set to include the same number of encoded Doppler multiplex numbers within a range of N _CM or less. For example, the encoding unit 107 may set the number of codes N _CM in all the encoded Doppler multiplex numbers. For example, the encoding unit 107 uniformly sets the number of encoded Doppler multiplexes for Doppler multiplex signals. In this setting, for example, loopback determination processing in reception processing, which will be described later, is not applied. Further, the radar device 10 can individually separate and receive signals that are encoded and Doppler multiplex-transmitted from the plurality of transmitting antennas 109 over a Doppler range of, for example, ±1/(2Loc×N _DM ×Tr).

The encoding unit 107 calculates the encoded Doppler phase rotation amount ψ shown in the following equation (8) for the phase rotation amount φ _ndm that gives the ndm-th Doppler shift amount DOP _ndm in the m-th transmission period Tr. _{ndop_code(ndm) and ndm} (m) are set and output to the phase rotation unit 108.

Here, the subscript "ndop_code(ndm)" represents an index below the encoded Doppler multiplex number N _{DOP_CODE} (ndm) for the phase rotation amount φ _ndm that gives the Doppler shift amount DOP _ndm . For example, ndop_code(ndm)=1,..., N _{DOP_CODE} (ndm). Also, angle[x] is an operator that outputs the radian phase of the real number x. For example, angle[1]=0, angle[-1]=π, angle[j]=π/2, angle[-j ]=-π/2. Furthermore, floor[x] is an operator that outputs the largest integer that does not exceed the real number x. j is an imaginary unit.

For example, as shown in equation (8), the encoded Doppler phase rotation amount ψ _{ndop_code(ndm), ndm} (m) is the Doppler shift amount DOP _ndm given during the transmission cycle period of code length Loc times used for encoding. OC _{ndop_code( ndm)} (1),…,OC _{ndop_code} A phase rotation amount corresponding to each of _(ndm) and (Loc) is given (second item in equation (8)).

Furthermore, the encoding unit 107 outputs the orthogonal code element index OC_INDEX to the radar receiving unit 200 (output switching unit 209 described later) for each transmission period (Tr). OC_INDEX is an orthogonal code element index that indicates the element of the orthogonal code sequence Code _{ndop_code (ndm)} , and is cyclically variable in the range from 1 to Loc for each transmission period (Tr), as shown in the following equation (9). do.

Here, mod(x, y) is a modulo operator, and is a function that outputs the remainder after dividing x by y. Also, m=1, ~,Nc. Nc is the number of transmission cycles used for radar positioning (hereinafter referred to as "number of times of radar transmission signal transmission"). Further, the number of times the radar transmission signal is transmitted Nc is set to be an integral multiple (Ncode times) of Loc. For example, Nc=Loc×Ncode.

Next, an example of a method for non-uniformly setting the encoded Doppler multiplex number N _{DOP_CODE} (ndm) for the Doppler multiplex signal in the encoding section 107 will be described.

For example, the encoding unit 107 sets the number of orthogonal code sequences (for example, the number of code multiplexing or the number of codes) N _CM that satisfies the following conditions. For example, the number N _CM of orthogonal code sequences and the number N _DM of Doppler multiplexing satisfy the following relationship with respect to the number Nt of transmitting antennas 109 used for multiplex transmission.
(Number of orthogonal code sequences N _CM ) × (Number of Doppler multiplexing N _DM ) > Number of transmitting antennas used for multiplex transmission Nt

For example, among the number N _CM of orthogonal code sequences and the number N _DM of Doppler multiplexing that satisfy the above conditions, using a combination with a smaller value of the product (N _CM ×N _DM ) is advantageous in terms of the complexity of the circuit configuration. It is also more suitable. However, among the number N _CM of orthogonal code sequences and the number N _DM of Doppler multiplexing that satisfy the above conditions, the combination is not limited to a combination in which the value of the product (N _CM ×N _DM ) is smaller, and other combinations are also applicable.

Note that in this embodiment, as an example, an orthogonal code sequence with a code length Loc=2 and the number of codes (for example, code multiplexing number) N _CM =2 is used.

For example, when Nt=5, a combination of N _DM =3 and N _CM =2 is suitable.

FIG. 3 shows, as an example, a case where Nt=5, N _DM =3, and N _CM =2. For example, the Doppler shift amounts DOP ₁ , DOP ₂ , DOP ₃ and the orthogonal codes Code ₁ and Code ₂ are assigned as N _{DOP_CODE} (1), N _{DOP_CODE} (2) and N _{DOP_CODE} (3 ) is determined according to the settings.

For example, (a) in FIG. 3 shows an example where _{N DOP_CODE} (1)=2, N _{DOP_CODE} (2)=2, N _{DOP_CODE} (3)=1, and (b) in FIG. )= ₁ , N _{DOP_CODE} (2)=2, N _{DOP_CODE} (3)= _{2 .} An example of (3)=2 is shown.

Note that in FIGS. 3A and 3B, Code ₁ is used for the Doppler shift amount corresponding to the encoded Doppler multiplex number N _{DOP_CODE} (ndm)=1, but the present invention is not limited to this. For example, when the number of encoded Doppler multiplexes is set to be smaller than _NCM , Code ₂ may be used instead of Code ₁ , as shown in FIG. 3(c).

As shown in Figure 3(a), N _{DOP_CODE} (1)=N _{DOP_CODE} (2)=2, N _{DOP_CODE} (3)=1, and N _{DOP_CODE} (1)=N _{DOP_CODE} (2)≠N _{DOP_CODE} (2 ), the number of encoded Doppler multiplexes N _{DOP_CODE} is set nonuniformly for each Doppler shift amount DOP ₁ , DOP ₂ , and DOP ₃ . In such a setting, the Doppler frequency range can be, for example, equivalent to the maximum Doppler speed when transmitting from one antenna (details will be described later).

Further, for example, when Nt=6 or 7, a combination of N _DM =4 and N _CM =2 is suitable.

FIG. 4 shows, as an example, a case where Nt=6, N _DM =4, and N _CM =2. For example, the Doppler shift amounts DOP ₁ , DOP ₂ , DOP ₃ , DOP ₄ and the orthogonal codes Code ₁ and Code ₂ are assigned as N _{DOP_CODE} (1), N _{DOP_CODE} (2), N Determined according to the settings of _{DOP_CODE} (3) and N _{DOP_CODE} (4).

For example, (a) in FIG. 4 shows an example where N _{DOP_CODE} (1)=N _{DOP_CODE} (2)=2, N _{DOP_CODE} (3)=N _{DOP_CODE} (4)=1, and (b) in FIG. An example of N _{DOP_CODE} (1)=N _{DOP_CODE} (3)=2, N _{DOP_CODE} (2)=N _{DOP_CODE} (4)=1 is shown.

Note that in FIG. 4, Code ₁ is used for the Doppler shift amount corresponding to the encoded Doppler multiplex number N _{DOP_CODE} (ndm)=1, but the code is not limited to this. For example, when the number of encoded Doppler multiplexes is set to be smaller than _NCM , Code 2 may be used instead of Code 1, as shown in FIG. 4(a), and Code ₂ may be used instead of Code ₁ , as shown in FIG. 4(b). In addition, Code ₁ and Code ₂ may be mixed.

Further, for example, as shown in FIG. 4, when Nt=6, N _DM =4, and N _CM =2, there are two Doppler shift amounts that do not use all the signs. Further, for example, among N _DM =4, for combinations of Doppler shift amounts that do not use all signs, there are 6 combinations (= ₄ C ₂ ) of selecting 2 Doppler shift amounts from 4 Doppler shift amounts, In each combination, there are four combinations of codes to be used (=N _CM ×N _CM ). Therefore, in the case of Nt=6, N _DM =4, and N _CM =2, there are a total of 24 combinations of Doppler shift amount DOP and orthogonal code Code assignment.

Similarly, for example, when Nt=8, a combination of N _DM =5 and N _CM =2 is suitable. Further, for example, when Nt=9, a combination of N _DM =5 and N _CM =2 is suitable. Further, for example, when Nt=10, a combination of N _DM =6 and N _CM =2 is suitable. Note that the number Nt of transmitting antennas 109 is not limited to the above example, and an embodiment of the present disclosure can be applied to Nt=11 or more.

Next, an example of a method for uniformly setting the encoded Doppler multiplex number N _{DOP_CODE} (ndm) for the Doppler multiplex signal in the encoding section 107 will be described.

Note that the method of uniformly setting the number of encoded Doppler multiplexes N _{DOP_CODE} (ndm) for Doppler multiplexed signals in the encoding unit 107 is based on the number of orthogonal code sequences N _CM and the number of Doppler multiplexes N _DM that satisfy the following conditions. It is more preferable to use a combination with a smaller value of the product (N _CM ×N _DM ) in terms of characteristics and complexity of the circuit configuration. However, the present invention is not limited to combinations in which the value of the product (N _CM ×N _DM ) is smaller, and other combinations are also applicable.

For example, the encoding unit 107 sets the number of orthogonal code sequences (for example, the number of code multiplexing or the number of codes) N _CM that satisfies the following conditions. For example, the number N _CM of orthogonal code sequences and the number N _DM of Doppler multiplexing satisfy the following relationship with respect to the number Nt of transmitting antennas 109 used for multiplex transmission.
(Number of orthogonal code sequences N _CM ) × (Number of Doppler multiplexing N _DM ) = Number of transmitting antennas used for multiplex transmission Nt

For example, when Nt=4, a combination of N _DM =2 and N _CM =2 is suitable. Further, for example, when Nt=6, a combination of N _DM =3 and N _CM =2 is suitable. Further, for example, when Nt=8, a combination of N _DM =4 and N _CM =2 is suitable. Further, for example, when Nt=10, a combination of N _DM =5 and N _CM =2 is suitable. Further, for example, when Nt=12, a combination of N _DM =6 and N _CM =2 is suitable.

Note that the number Nt of transmitting antennas 109 is not limited to the above example, and one embodiment of the present disclosure can be applied. In this case, a combination of integers such that the number of orthogonal code sequences N _CM > 1, the number of Doppler multiplexing N _DM > 1, and (number of orthogonal code sequences N _CM ) x (number of Doppler multiplexing N _DM ) = transmitting antenna used for multiplex transmission In order to satisfy the number Nt, the number Nt of transmitting antennas used for multiplex transmission may be set to 4 or more and Nt that satisfies the above conditions.

Next, an example of setting the encoded Doppler phase rotation amount ψ _{ndop_code(ndm), ndm} (m) will be explained.

For example, in encoding section 107, the number of transmitting antennas used for multiplex transmission is Nt=4, the number of Doppler multiplexing _NDM =2, the number of code multiplexing _NCM =2, and the orthogonal code sequence Code ₁ ={1 with code length Loc=2. , 1}, Code ₂ = {j, -j} will be explained. In this case, for example, if the number of encoded Doppler multiplexes is N _{DOP_CODE} (1)=2 and N _{DOP_CODE} (2)=2, the encoding unit 107 generates the encoded Doppler multiplex number as shown in the following equations (10) to (13). The phase rotation amounts ψ _{1, 1} (m), ψ _{2, 1} (m), ψ _{1, 2} (m), ψ _{2, 2} (m) are set and output to the phase rotation section 108.

Here, as an example, if the phase rotation amount that gives the Doppler shift amount DOP _ndm is φ _ndm =2π(ndm-1)/N _DM in equation (5), the phase rotation amount that gives the Doppler shift amount DOP ₁ is φ ₁ =0 and the phase rotation amount φ ₂ =π that gives the Doppler shift amount DOP ₂ , the encoding unit 107 uses the encoded Doppler phase rotation amount ψ ₁ as shown in the following equations (14) to (17). _{, 1} (m), ψ _{2, 1} (m), ψ _{1, 2} (m), ψ _{2, 2} (m) are set and output to the phase rotation unit 108. Here, m=1, ~, Nc. Note that, here, a modulo operation based on 2π is performed, and the range of radians is from 0 to less than 2π (the same applies to the following description).

As shown in equations (14) to (17), when the phase rotation amount is set to φ _ndm =2π(ndm-1)/N _DM , which is obtained by equally dividing 2π, the encoded Doppler phase rotation amount ψ _{1, 1} (m), ψ _{2, 1} (m), ψ _{1, 2} (m), ψ _{2, 2} (m) change with a transmission period of N _DM ×N _CM =2 × 2 = 4.

Alternatively, as another example, let the phase rotation amount that gives the Doppler shift amount DOP _ndm be φ _ndm =2π(ndm)/N _DM , and the phase rotation amount that gives the Doppler shift amount DOP ₁ = π and the _Doppler The phase rotation amount φ ₂ that provides the shift amount DOP ₂ may be set to 0. In this case, the encoding unit 107 calculates the encoded Doppler phase rotation amount ψ _{1, 1} (m), ψ _{2, 1} (m), ψ _{1, 2} (m) as shown in the following equations (18) to (21). , ψ _{2, 2} (m) are set and output to the phase rotation unit 108. Here, m=1, ~, Nc.

In addition, as shown in equations (14) to (17) or equations (18) to (21), the phase number (for example, 0 and π (2) is smaller than the number Nt=4 of transmitting antennas 109 used for multiplex transmission. For example, as shown in equations (14) to (17) or equations (18) to (21), the number of phases (for example, two, 0 and π) used for the amount of phase rotation that gives the amount of Doppler shift is The number of Doppler shift amounts used for transmission (for example, the number of Doppler multiplexes) is equal to N _DM =2.

Further, for example, in the encoding section 107, the number of transmitting antennas used for multiplex transmission Nt=6, the number of Doppler multiplexing N _DM =4, the number of code multiplexing N _CM =2, and the orthogonal code sequence Code ₁ = of code length Loc=2 is set. The case where {1, 1} and Code ₂ = {j, -j} are used will be explained. In this case, for example, if the number of encoded Doppler multiplexes is N _{DOP_CODE} (1)=1, N _{DOP_CODE} (2)=1, N _{DOP_CODE} (3)=2, and N _{DOP_CODE} (4)=2, the encoding unit 107 , the encoded Doppler phase rotation amount ψ _{1, 1} (m), ψ _{1, 2} (m), ψ _{1, 3} (m) , ψ _{2, 3} (m) as shown in the following equations (22) to (27). , ψ _{1, 4} (m) and ψ _{2, 4} (m) are set and output to the phase rotation unit 108. Here, m=1, ~, Nc.

Here, as an example, let the phase rotation amount that gives the Doppler shift amount DOP _ndm be φ _ndm =2π(ndm-1) /N _DM , and the phase rotation amount that gives the Doppler shift amount DOP ₁ = ₀ , Doppler shift The phase rotation amount φ ₂ = π/2 to give the amount DOP ₂ , the phase rotation amount φ ₃ = π to give the Doppler shift amount DOP ₃ , the phase rotation amount φ ₄ = 3π/2 to give the Doppler shift amount DOP ₄ . When used, the encoding unit 107 calculates the encoded Doppler phase rotation amount ψ _{1, 1} (m), ψ _{1, 2} (m), ψ _{1, 3} (m) as shown in the following equations (28) to (33). , ψ _{2, 3} (m), ψ _{1, 4} (m), ψ _{2, 4} (m) are set and output to the phase rotation unit 108. Here, m=1, ~, Nc.

As shown in equations (28) to (33), when the amount of phase rotation is set to φ _ndm =2π(ndm-1)/N _DM , which is equal division of 2π, the amount of encoded Doppler phase rotation ψ _{1 , 1} (m), ψ _{1, 2} (m), ψ _{1, 3} (m) , ψ _{2, 3} (m) , ψ _{1, 4} (m) , ψ _{2, 4} (m) is N _DM × It changes with a transmission cycle of N _CM =4×2=8.

Furthermore, as shown in equations (28) to (33), the phase number (for example, 0, π/2, π, and 3π/ 2) is smaller than the number Nt=6 of transmitting antennas 109 used for multiplex transmission. For example, as shown in equations (28) to (33), the number of phases (for example, four, 0, π/2, π, and 3π/2) used for the amount of phase rotation that gives the amount of Doppler shift is: The number of Doppler shift amounts used for multiplex transmission (for example, Doppler multiplex number) is equal to N _DM =4.

Here, as an example, the phase rotation in the case where the number of transmitting antennas 109 Nt=4, the number of Doppler multiplexing N _DM =2, and the case where the number of transmitting antennas 109 Nt=6, the number of Doppler multiplexing N _DM =4 Although the number of transmitting antennas 109 Nt and the Doppler multiplexing number _NDM are not limited to these values. For example, regardless of the value of the number Nt of transmitting antennas 109, the number of phases used for the amount of phase rotation may be set to be smaller than the number Nt of transmitting antennas 109 used for multiplex transmission. Further, the number of phases used for the amount of phase rotation to provide the amount of Doppler shift may be set equal to the number _NDM of the amount of Doppler shift used for multiplex transmission.

In addition, as in the example above, the phase rotation amount setting shown in the maximum equidistant Doppler shift amount setting may be used to set the phase rotation amount, or the phase rotation amount setting shown in the equidistant Doppler shift amount setting may be used. The rotation amount may be set using, for example, equation (6).

The method for setting the phase rotation amount in the phase rotation amount setting section 105 has been described above.

In FIG. 1, the phase rotation unit 108 receives input from the radar transmission signal generation unit 101 based on the encoded Doppler phase rotation amount ψ _{ndop_code(ndm), ndm} (m) set in the phase rotation amount setting unit 105. A phase rotation amount is given to the chirp signal for each transmission period Tr. Here, ndm=1,~, N _DM , and ndop_code(ndm)=1,~, N _{DOP_CODE} (ndm).

The sum of the encoded Doppler multiplex numbers N _{DOP_CODE} (1), N _{DOP_CODE} (2), ~, N _{DOP_CODE} (N _DM ) is set equal to the number Nt of transmitting antennas 109, and the Nt encoded Doppler phase rotation amounts are The signals are respectively input to Nt phase rotation units 108.

The Nt phase rotation units 108 convert the input encoded Doppler phase rotation amount ψ _{ndop_code(ndm), ndm} (m) to the chirp signal input from the radar transmission signal generation unit 101 for each transmission period Tr. are given respectively. The outputs from the Nt phase rotation sections 108 (for example, referred to as coded Doppler multiplex signals) are amplified to a specified transmission power and then radiated into space from the Nt transmission antennas 109 of the transmission array antenna section.

Note that, hereinafter, the phase rotation unit 108 that provides the encoded Doppler phase rotation amount ψ _{ndop_code(ndm), ndm} (m) will be referred to as “phase rotation unit PROT#[ndop_code(ndm), ndm]”. Similarly, the transmitting antenna 109 that radiates the output of the phase rotation unit PROT#[ndop_code(ndm), ndm] into space is expressed as "transmitting antenna Tx#[ndop_code(ndm), ndm]." Here, ndm=1,~, N _DM , and ndop_code(ndm)=1,~, N _{DOP_CODE} (ndm).

In this embodiment, an orthogonal code sequence with code length Loc=2 and the number of codes (for example, code multiplexing number) N _CM =2 is used. Therefore, N _{DOP_CODE} (ndm)=1 or 2, and ndop_code(ndm)≦2.

For example, when the number of transmitting antennas used for multiplex transmission is Nt = 4, the number of Doppler multiplexing N _DM = 2, the number of code multiplexing N _CM = 2, and the orthogonal code sequence Code ₁ = {1, 1} with code length Loc = 2. , Code ₂ = {j, -j}, and the number of encoded Doppler multiplexes is N _{DOP_CODE} (1)=2, N _{DOP_CODE} (2)=2. In this case, the encoded Doppler phase rotation amount ψ _{1, 1} (m), ψ _{2, 1} (m), ψ _{1, 2} (m), ψ _{2, 2} is sent from the encoding unit 107 to the phase rotation unit 108. (m) is input every transmission cycle.

For example, the phase rotation unit PROT#[1, 1] calculates the phase rotation amount ψ for each transmission period as shown in the following equation (34) with respect to the chirp signal generated in each transmission period by the radar transmission signal generation unit 101. Grant _{1, 1} (m). Furthermore, the output of the phase rotation unit PROT#[1, 1] is output from the transmitting antenna Tx#[1, 1]. Here, cp(t) represents a chirp signal for each transmission period.

Similarly, the phase rotation unit PROT#[2, 1] calculates the amount of phase rotation for each transmission cycle as shown in the following equation (35) with respect to the chirp signal generated in each transmission cycle by the radar transmission signal generation unit 101. Give ψ _{2, 1} (m). Furthermore, the output of the phase rotation unit PROT#[2, 1] is output from the transmitting antenna Tx#[2, 1].

Similarly, the phase rotation unit PROT#[1, 2] calculates the phase rotation amount for each transmission period as shown in the following equation (36) with respect to the chirp signal generated in each transmission period by the radar transmission signal generation unit 101. Give ψ _{1, 2} (m). Furthermore, the output of the phase rotation unit PROT#[1, 2] is output from the transmitting antenna Tx#[1, 2].

Similarly, the phase rotation unit PROT#[2, 2] calculates the phase rotation amount for each transmission period as shown in the following equation (37) with respect to the chirp signal generated in each transmission period by the radar transmission signal generation unit 101. Give ψ _{2, 2} (m). Furthermore, the output of the phase rotation unit PROT#[2, 2] is output from the transmitting antenna Tx#[2, 2].

An example of setting the encoded Doppler phase rotation amount ψ _{ndop_code(ndm), ndm} (m) has been described above.

Further, in this embodiment, for example, the polarization of the transmitting antenna 109 and the arrangement of the transmitting antenna 109 are associated with the assignment of the encoded Doppler phase rotation amount as follows. Due to this association, in radar processing, the radar device 10 uses not only the transmitting antenna 109 for multiplex transmission, but also the transmitting antenna (for example, , circularly polarized waves) (examples will be described later).

For example, at least one set of adjacent transmitting antennas 109 are antennas that emit (or radiate) different orthogonal polarizations (e.g., horizontal polarization and vertical polarization), and have the same Doppler multiplexing (e.g., Doppler shift). transmit a radar transmission signal using

In this embodiment, N _{DOP_CODE} (ndm _{_BF} )=2 because an orthogonal code sequence with code length Loc=2 and the number of codes (for example, code multiplexing number) N _CM =2 is used.

For example, adjacent N _{DOP_CODE} (ndm _{_BF} )=2 transmitting antennas 109 include a transmitting antenna Tx to which a phase rotation unit PROT#[1, ndm _{_BF} ] and a phase rotation unit PROT#[2, ndm _{_BF} ] are assigned. #[1, ndm _{_BF} ], transmit antenna Tx#[2, ndm _{_BF} ]. Here, _{ndm_BF} can be any value from 1 to N _DM . For example, among multiple combinations of the Doppler shift amount DOP _ndm and orthogonal code sequences, adjacent transmitting antennas 109 are antennas that emit mutually different orthogonal polarized waves, and are associated with each antenna. The Doppler shift amounts are the same (for example, ndm= _{ndm_BF} ) in the combinations shown in FIG.

For example, one or more combinations (for example, pairs) that satisfy the association between the above-mentioned assignment of the amount of encoded Doppler phase rotation and the polarization and arrangement of the transmitting antenna 109 may be included.

As an example, when the number of transmitting antennas used for multiplex transmission is Nt = 4, the number of Doppler multiplexing N _DM = 2, the number of code multiplexing N _CM = 2, and the orthogonal code sequence Code ₁ = {1, 1 with code length Loc = 2. }, Code ₂ = {j, -j}, and the case where the number of encoded Doppler multiplexes is N _{DOP_CODE} (1)=2, N _{DOP_CODE} (2)=2 will be explained. Note that the number of beam transmitting antennas N _BF =2, and _{ndm_BF} =1 and 2 are used as indices of Doppler multiplexed signals used for the beam transmitting antennas.

In FIG. 6, for example, Nt=4 transmitting antennas 109 adjacent in the horizontal direction are transmitting antenna Tx#[1, 1], transmitting antenna Tx#[2, 1], transmitting antenna Tx# from the left antenna. [1, 2] and transmitting antenna Tx#[2, 2].

In FIG. 6, 2 (=N _{DOP_CODE} (1)) adjacent transmitting antennas Tx#[1, 1] and Tx#[2, 1] from the left are the same Doppler multiplexed (Doppler shift amount = DOP ₁ ) to transmit radar transmission signals. Furthermore, transmitting antennas Tx#[1, 1] and Tx#[2, 1] that transmit radar transmission signals using the same Doppler multiplexing (Doppler shift amount = DOP ₁ ) are linearly polarized waves that are orthogonal to each other. Uses an antenna that emits radio waves. For example, in the example shown in FIG. 6, an antenna that emits horizontally polarized radio waves (for example, a "horizontal polarized antenna") may be used as the transmitting antenna Tx#[1, 1], and the transmitting antenna Tx#[ 2, 1], an antenna that emits vertically polarized radio waves (for example, a "vertically polarized antenna") may be used. Note that the present invention is not limited to this, and a vertically polarized antenna may be used as the transmitting antenna Tx#[1, 1], and a horizontally polarized antenna may be used as the transmitting antenna Tx#[2, 1]. Note that the black circles (●) in FIG. 6 indicate the phase center of each transmitting antenna.

In addition, in FIG. 6, 2 (=N _{DOP_CODE} (2)) adjacent transmitting antennas Tx#[1, 2] and Tx#[2, 2] from the right side have the same Doppler multiplexing (Doppler shift amount=DOP ₂ ) to transmit the radar transmission signal. Furthermore, transmitting antennas Tx#[1, 2] and Tx#[2, 2] that transmit radar transmission signals using the same Doppler multiplexing (Doppler shift amount = DOP ₂ ) are linearly polarized waves that are orthogonal to each other. Uses an antenna that emits radio waves. For example, in the example shown in FIG. 6, a horizontally polarized antenna may be used as the transmitting antenna Tx#[1, 2], and a vertically polarized antenna may be used as the transmitting antenna Tx#[2, 2]. Note that the present invention is not limited to this, and a vertically polarized antenna may be used for the transmitting antenna Tx#[1, 2], and a horizontally polarized antenna may be used for the transmitting antenna Tx#[2, 2].

As another example, when the number of transmitting antennas used for multiplex transmission Nt = 5, the number of Doppler multiplexing N _DM = 3, the number of code multiplexing N _CM = 2, and the orthogonal code sequence Code ₁ = code length Loc = 2. A case will be described in which {1, 1}, Code ₂ = {j, -j}, and the number of encoded Doppler multiplexes are N _{DOP_CODE} (1)=2, N _{DOP_CODE} (2)=2. Note that the number of beam transmitting antennas N _BF =2, and _{ndm_BF1} =1 and _{ndm_BF2} =2 are used as indices of Doppler multiplexed signals used for the beam transmitting antennas.

In FIG. 7, for example, Nt=5 transmitting antennas 109 adjacent in the horizontal direction are transmitting antenna Tx#[1, 1], transmitting antenna Tx#[2, 1], transmitting antenna Tx# from the left antenna. [1, 2], transmitting antenna Tx#[2, 2], and transmitting antenna Tx#[1, 3]. Note that the black circles (●) in FIG. 7 indicate the phase center of each transmitting antenna.

In FIG. 7, 2 (=N _{DOP_CODE} (1)) adjacent transmitting antennas Tx#[1, 1] and Tx#[2, 1] from the left side are the same Doppler multiplexed (Doppler shift amount=DOP ₁ ) to transmit radar transmission signals. Furthermore, transmitting antennas Tx#[1, 1] and Tx#[2, 1] that transmit radar transmission signals using the same Doppler multiplexing (Doppler shift amount = DOP ₁ ) are linearly polarized waves that are orthogonal to each other. Uses an antenna that emits radio waves. For example, in the example shown in FIG. 7, a horizontally polarized antenna may be used for transmitting antenna Tx#[1, 1], and a vertically polarized antenna may be used for transmitting antenna Tx#[2, 1]. Note that the present invention is not limited to this, and a vertically polarized antenna may be used as the transmitting antenna Tx#[1, 1], and a horizontally polarized antenna may be used as the transmitting antenna Tx#[2, 1].

In addition, 2 (=N _{DOP_CODE} (2)) adjacent transmit antennas Tx#[1, 2] and Tx#[2] on the right side of transmit antennas Tx#[1, 1] and Tx#[2, 1] , 2] transmits radar transmission signals using the same Doppler multiplexing (Doppler shift amount = DOP ₂ ). Furthermore, transmitting antennas Tx#[1, 2] and Tx#[2, 2] that transmit radar transmission signals using the same Doppler multiplexing (Doppler shift amount = DOP ₂ ) are linearly polarized waves that are orthogonal to each other. Uses an antenna that emits radio waves. For example, in the example shown in FIG. 7, a horizontally polarized antenna may be used for transmitting antenna Tx#[1, 2], and a vertically polarized antenna may be used for transmitting antenna Tx#[2, 2]. Note that the present invention is not limited to this, and a vertically polarized antenna may be used for the transmitting antenna Tx#[1, 2], and a horizontally polarized antenna may be used for the transmitting antenna Tx#[2, 2]. Note that any polarized antenna may be used as the transmitting antenna Tx#[1, 3].

Further, for example, when a plurality of sets (or pairs) of the above-mentioned adjacent horizontally polarized antennas and vertically polarized antennas are included, the radar device 10 may set a different Doppler shift amount for each of the plurality of sets. . For example, in FIGS. 6 and 7, a set of transmitting antennas Tx#[1, 1] and Tx#[2, 1] and a set of transmitting antennas Tx#[1, 2] and Tx#[2, 2] The amount of Doppler shift to be set may be different.

As such, at least one set of adjacent transmitting antennas 109 (e.g., corresponding to the first transmitting antenna) includes a transmitting antenna 109 that radiates horizontally polarized waves and a transmitting antenna 109 that radiates vertically polarized waves (e.g., corresponding to the first transmitting antenna). , corresponding to the second transmit antenna). Furthermore, the same amount of Doppler shift is set for at least one set of adjacent transmitting antennas 109, and radar transmission signals are transmitted using the same Doppler multiplexing. For example, at least one set of adjacent transmit antennas 109 perform code multiplexed transmission using the same Doppler multiplexing and radiate radar transmission signals into space using different polarization antennas.

Here, the received signal for each transmission period corresponding to the radar transmission signal code-multiplexed using the same Doppler multiplexing can be regarded as the received signal corresponding to beam transmission by the plurality of transmitting antennas 109. For example, the above-mentioned transmission by at least one set of adjacent transmitting antennas 109 is equivalent to beam transmission by a sub-array constituted by the adjacent transmitting antennas 109. For example, when transmitting radar transmission signals with equal power from at least one set of adjacent transmitting antennas 109 described above, the radar device 10 performs the transmission with the midpoint position of the adjacent transmitting antennas 109 as the phase center of the sub-array. It can be handled as transmission by a new transmitting antenna (hereinafter referred to as a "beam transmitting antenna") (details will be described later in reception processing). Note that the white circles (◯) in FIGS. 6 and 7 indicate the phase center of each beam transmission antenna.

Moreover, for the at least one set of adjacent transmitting antennas 109 described above, polarized antennas that emit mutually orthogonal linearly polarized radio waves, such as a horizontally polarized antenna and a vertically polarized antenna, may be used, for example. As a result, the received signal for each transmission period corresponding to the radar transmission signal code-multiplexed using the same Doppler multiplexing is a circularly polarized radio wave radiated from at least one set of adjacent transmission antennas 109. It can be regarded as the received signal corresponding to the transmission from the transmitted antenna. This principle will be explained below using computer simulation.

A computer simulation in which two types of orthogonal linearly polarized antennas (for example, a horizontally polarized antenna and a vertically polarized antenna) are combined to generate different polarized waves (right-handed circularly polarized waves or left-handed circularly polarized waves) will be described.

For example, as shown in FIG. 8, the feeding phase to the horizontally polarized antenna (ANT#1) and vertically polarized antenna (ANT#2) is set to ANT#1 with ANT#1 as the reference (for example, 0 degrees). We will explain the results of a computer simulation when the phase of the signal is set to 90 degrees or -90 degrees and signals are transmitted simultaneously. Here, as an example, the element spacing of the horizontally polarized antenna (ANT#1) and the vertically polarized antenna (ANT#2) is set to 0.5 wavelength.

Note that the element spacing is not limited to 0.5 wavelength, and may be wider. When the element spacing is increased, the beam width of the main beam formed by the two antennas becomes narrower, and the viewing angle at which the axial ratio can be ensured becomes narrower. In addition, if the element spacing is increased by about one wavelength or more, a grating lobe will occur in a direction different from the main beam direction composed of two antennas, and radio waves that are circularly polarized in the main beam direction and grating lobe direction will be emitted. be done.

Although FIG. 8 shows an example of a linearly polarized antenna using a single-point feeding planar patch antenna, the present invention is not limited to this, and an antenna element of a two-point feeding type may also be used for a planar patch antenna. good. Alternatively, an antenna element configured into an array using a plurality of antenna elements may be used.

For example, when the phase of ANT#2 is shifted by 90 degrees with respect to ANT#1, a directional left-handed circularly polarized wave shown in FIG. 9 is emitted. 9(a) shows the directivity in the horizontal plane, and FIG. 9(b) shows the directivity in the vertical plane. Furthermore, when the phase of ANT#2 is shifted by -90 degrees with respect to ANT#1, a right-handed circularly polarized wave with the directivity shown in FIG. 10 is emitted. FIG. 10(a) shows the directivity in the horizontal plane, and FIG. 10(b) shows the directivity in the vertical plane.

Here, the two types of phases [0°, 90°] and [0°, -90°] that generate left-handed circularly polarized waves or right-handed circularly polarized waves using ANT#1 and ANT#2 are encoded. This corresponds to the case where Code ₁ = {1, 1} and Code ₂ = {j, -j} are used in section 107 as an orthogonal code sequence of N _CM =2 with code length Loc = 2. For example, they are orthogonal to each other, and if the noc-th code element OC _ncm (noc) of the first code is the reference, the noc-th code element OC _ncm (noc) of the second code is +90° Alternatively, the -90° phase becomes a different code element. Here noc=1 or 2. For example, among multiple orthogonal codes, the first code Code ₁ ={1, 1} for the radar transmission signal transmitted from ANT#1, and the second code for the radar transmission signal transmitted from ANT#2. Between the code Code ₂ ={j, -j}, the phase of the code element OC corresponding to each transmission period differs by 90°.

Radar transmission signals, which are code-multiplexed and transmitted using the same Doppler multiplexing, are transmitted from multiple transmitting antennas 109 with a phase difference of +90° or -90°. The relationship in which the signals are transmitted with different phases is maintained. Therefore, the received signals corresponding to these radar transmission signals can be regarded as the received signals corresponding to the transmission from the transmitting antenna from which circularly polarized radio waves are radiated.

For example, when using Code ₁ = {1, 1} and Code ₂ = {j, -j}, in the first transmission cycle, the first transmission signal of Code ₁ is The amount of phase rotation corresponding to the first code element “1” of Code 2 is given, and the amount of phase rotation corresponding to the first code element “j” of Code ₂ is given to the radar transmission signal transmitted from ANT#2. May be granted. As a result, in the first transmission cycle, left-handed circularly polarized radio waves are emitted by ANT#1 and ANT#2. Also, for example, in the second transmission period following the first transmission period, the phase rotation amount corresponding to the second code element "1" of Code ₁ with respect to the radar transmission signal transmitted from ANT#1 is may be added, and a phase rotation amount corresponding to the second code element "-j" of Code ₂ may be added to the radar transmission signal transmitted from ANT#2. As a result, in the second transmission cycle, ANT#1 and ANT#2 radiate right-handed circularly polarized radio waves.

As described above, for example, the radar device 10 (radar transmitting unit 100) provides a transmission signal with a phase rotation amount in which the phase between ANT#1 and ANT#2 differs by 90 degrees in each transmission cycle, Radar transmission signals are multiplexed from ANT#1 and ANT#2. With such a transmission method, the radar device 10 can use transmitting antennas exceeding the number Nt of transmitting antennas for multiplex transmission, and can transmit polarized waves (for example, horizontally polarized waves and vertically polarized waves) different from the polarized waves of the transmitting antenna 109 (for example, horizontally polarized waves and vertically polarized waves). For example, it becomes possible to use a circularly polarized transmitting antenna.

In addition, although FIG.6 and FIG.7 demonstrated the transmitting antenna 109 arranged in the horizontal direction as an example, the arrangement method of the transmitting antenna 109 is not limited to this. For example, the transmitting antenna 109 may be arranged in a vertical direction, or may be arranged in a plane in a horizontal direction and a vertical direction. Further, the antenna constituting the transmitting antenna 109 may be composed of a plurality of sub-array elements arranged in the horizontal direction, a plurality of sub-array elements arranged in the vertical direction, or a plurality of sub-array elements arranged horizontally and vertically. may be configured. Further, the antennas shown in FIGS. 6 and 7 may be part of the plurality of antennas included in the radar device 10.

As described above, in the present embodiment, a combination of the Doppler shift amount DOP _ndm and the orthogonal code sequence Code _ncm , in which at least one of the Doppler shift amount DOP _ndm and the orthogonal code sequence Code _ncm is different, is provided to the plurality of transmitting antennas 109. (for example, assignment) are associated with each other.

In addition, in this embodiment, when the encoded Doppler multiplex number N _{DOP_CODE} ( _ndm ) for the Doppler multiplexed signal is set non-uniformly, each Doppler shift _amount is The number of multiplexes of orthogonal code sequences Code _ncm corresponding to DOP _ndm (for example, coded Doppler multiplex number N _{DOP_CODE} (ndm)) may be different. As an example, as shown in FIG. 3, the Nt transmitting antennas 109 include at least a plurality of (for example, two) transmitting antennas 109 each transmitting a transmitting signal that is code-multiplexed using different orthogonal code sequences; At least one transmit antenna 109 through which a non-code multiplexed transmission signal is transmitted may be included. For example, the radar transmission signal transmitted from the radar transmitter 100 includes at least a coded Doppler multiplex signal in which the coded Doppler multiplex number N _{DOP_CODE} (ndm) is set to the code number N CM, and a coded Doppler multiplex signal in which the coded Doppler multiplex number N DOP_CODE (ndm) is set to the code _number N _CM . (ndm) is set to be smaller than the number of codes _NCM .

In addition, in this embodiment, when setting the encoded Doppler multiplex number N _{DOP_CODE} (ndm) for the Doppler multiplexed signal uniformly, in the combination of the Doppler shift amount DOP _ndm and the orthogonal code sequence Code _ncm , the Doppler shift amount DOP _ndm The number of multiplexes of the orthogonal code series Code _ncm (for example, the number of encoded Doppler multiplexes N _{DOP_CODE} (ndm)) corresponding to each may be the same.

[Configuration of radar receiving section 200]
In FIG. 1, a radar receiving unit 200 includes Na receiving antennas 202, forming an array antenna. The radar receiving unit 200 also includes Na antenna system processing units 201-1 to 201-Na, a CFAR (Constant False Alarm Rate) unit 211, a coded Doppler multiplexing/demultiplexing unit 212, and a Doppler multiplexing/demultiplexing unit 213. , and a direction estimation unit 214.

Each receiving antenna 202 receives a reflected wave signal that is a radar transmission signal reflected by a target, and outputs the received reflected wave signal to the corresponding antenna system processing unit 201 as a received signal.

Each antenna system processing section 201 includes a reception radio section 203 and a signal processing section 206.

The reception radio section 203 includes a mixer section 204 and an LPF (low pass filter) 205. In the reception radio section 203 , the mixer section 204 mixes the received reflected wave signal with a chirp signal, which is a transmission signal inputted from the radar transmission signal generation section 101 , and passes the mixed signal through the LPF 205 . As a result, a beat signal having a frequency corresponding to the delay time of the reflected wave signal is extracted. For example, as shown in FIG. 11, the difference frequency between the frequency of the transmission chirp signal (transmission frequency modulated wave) and the frequency of the reception chirp signal (reception frequency modulated wave) is obtained as the beat frequency.

The signal processing unit 206 of each antenna system processing unit 201-z (where z=1 to Na) includes an AD conversion unit 207, a beat frequency analysis unit 208, an output switching unit 209, and a Doppler analysis unit 210. and has.

A signal (for example, a beat signal) output from the LPF 205 is converted into discrete sample data that is sampled discretely by an AD conversion unit 207 in a signal processing unit 206 .

The beat frequency analysis unit 208 performs FFT processing on N _data discrete sample data obtained in a specified time range (range gate) for each transmission period Tr. As a result, the signal processing unit 206 outputs a frequency spectrum in which a peak appears at a beat frequency corresponding to the delay time of the reflected wave signal (radar reflected wave). Note that during FFT processing, the beat frequency analysis unit 208 may, for example, multiply by a window function coefficient such as a Han window or a Hamming window. By using the window function coefficient, side lobes occurring around the beat frequency peak can be suppressed.

Here, the beat frequency response output from the beat frequency analysis section 208 in the z-th signal processing section 206 obtained by the m-th chirp pulse transmission is expressed as RFT _z (f _b , m). Here, f _b represents the beat frequency index and corresponds to the FFT index (bin number). For example, f _b =0,~,N _data /2-1, z=1,~,Na, and m=1,~, _NC . The smaller the beat frequency index f _b is, the smaller the delay time of the reflected wave signal is (for example, the shorter the distance to the target).

Furthermore, the beat frequency index f _b can be converted into distance information R(f _b ) using the following equation (38). Therefore, below, the beat frequency index f _b will be referred to as a "distance index f _b ".

Here, B _w represents the frequency modulation bandwidth within the range gate in the chirp signal, and C ₀ represents the speed of light.

The output switching unit 209 switches the output of the beat frequency analysis unit 208 for each transmission period to the Loc Doppler analysis units 210 based on the orthogonal code element index OC_INDEX input from the encoding unit 107 of the phase rotation amount setting unit 105. Among them, the OC_INDEX-th Doppler analysis unit 210 is selectively switched and outputted. For example, the output switching unit 209 selects the OC_INDEX-th Doppler analysis unit 210 obtained by equation (9) in the m-th transmission period Tr.

The signal processing unit 206 has Loc (here, as an example, Loc=2) Doppler analysis units 210-1 to 210-Loc. For example, the output switching unit 209 inputs data to the noc-th Doppler analysis unit 210 every Loc transmission period (Loc×Tr). For this reason, the noc-th Doppler analysis unit 210 uses the data of the Ncode transmission period among the Nc transmission periods (for example, the beat frequency response RFT _z (f _b , m )), Doppler analysis is performed for each distance index f _b . Here, noc is the index of the code element, noc=1, ~, Loc.

For example, when Ncode is a power of 2, FFT processing can be applied in Doppler analysis. In this case, the FFT size is Ncode, and the maximum Doppler frequency at which aliasing does not occur, which is derived from the sampling theorem, is ±1/(2Loc×Tr). Also, the Doppler frequency interval of the Doppler frequency index f _s is 1/(Ncode × Loc × Tr), and the range of the Doppler frequency index f _s is f _s = -Ncode/2, ~, 0, ~, Ncode/2− It is 1.

In the following, a case where Ncode is a power of 2 will be described as an example. Note that if the Ncode is not a power of 2, FFT processing can be performed as a data size (FFT size) of a power of 2 by including zero-padded data, for example. Further, the Doppler analysis unit 210 may multiply by a window function coefficient such as a Han window or a Hamming window during FFT processing. By applying a window function, side lobes that occur around the Doppler frequency peak can be suppressed.

For example, the output VFT _z ^noc (f _b , f _s ) of the Doppler analysis unit 210 of the z-th signal processing unit 206 is expressed by the following equation (39). Note that j is an imaginary unit, and z=1 to Na.

The processing in each component of the signal processing unit 206 has been described above.

In FIG. 1, the CFAR unit 211 performs CFAR processing (for example, adaptive threshold determination) using the outputs of the Loc Doppler analysis units 210 of the first to Nath signal processing units 206, and Extract the distance index f _{b_cfar} and Doppler frequency index f _{s_cfar} that give the signal.

The CFAR unit 211 adds the power of the output VFT _z ^noc (f _b , f _s ) of the Doppler analysis unit 210 of the first to Na-th signal processing units 206 as shown in the following equation (40), and calculates the distance. Performs two-dimensional CFAR processing consisting of an axis and Doppler frequency axis (corresponding to relative velocity), or CFAR processing that combines one-dimensional CFAR processing. For CFAR processing that is a combination of two-dimensional CFAR processing or one-dimensional CFAR processing, for example, the processing disclosed in Non-Patent Document 2 may be applied.

The CFAR unit 211 adaptively sets a threshold and performs encoded Doppler multiplexing on the distance index f _{b_cfar} , the Doppler frequency index f _{s_cfar} , and the received power information PowerFT(f _{b_cfar} , f _{s_cfar} ) that result in received power greater than the threshold. It is output to the separation section 212.

Note that when, for example, equation (5) is used as the phase rotation amount φ _ndm for giving the Doppler shift amount DOP _ndm , the intervals of the Doppler shift amount in the Doppler frequency domain in the output of the Doppler analysis unit 210 are equal intervals, If the Doppler shift amount interval ΔFD is expressed by the Doppler frequency index interval, then ΔFD=Ncode/ _NDM . Therefore, in the output of the Doppler analysis unit 210, peaks are detected at intervals of ΔFD for each signal subjected to Doppler shift multiplexing in the Doppler frequency domain. Note that when using equation (5) as the phase rotation amount _φndm , ΔFD may not be an integer depending on Ncode and _NDM . In such a case, ΔFD can be set to an integer value by using equation (59), which will be described later. In the following, the reception processing operation will be explained assuming that ΔFD is an integer value.

FIG. 12A shows an example of the output of the Doppler analysis unit 210 at a distance where reflected waves from three targets exist when N _DM =2. For example, as shown in FIG. 12(a), when the reflected waves of three targets are observed at Doppler frequency indexes f1, f2, and f3, the reflected waves have ΔFD for each of f1, f2, and f3. (e.g., f1-ΔFD, f2-ΔFD, f3-ΔFD+Ncode).

Therefore, the CFAR unit 211 divides each output of the Doppler analysis unit 210 into Doppler shift amount intervals ΔFD, and calculates the Doppler shift for each divided range as shown in the following equation (41). After combining the multiplexed signal peak positions and adding up the power (for example, referred to as "Doppler region compression"), CFAR processing (for example, referred to as "Doppler region compression CFAR processing") may be performed. Here, f _{s_comp} =-ΔFD/2,~,- ΔFD/2-1. For example, if ΔFD=Ncode/N _DM , f _{s_comp} =Ncode/(2N _DM ),~,Ncode/(2N _DM )-1.

の場合は、Ncodeを加えたドップラ周波数インデックスを用いる。 However, in equation (41),

In this case, use Doppler frequency index plus Ncode.

の場合は、更に、Ncodeを減算したドップラ周波数インデックスを用いる。 Similarly, in equation (41),

In this case, use the Doppler frequency index obtained by subtracting the Ncode.

As a result, the Doppler frequency range for CFAR processing can be compressed to 1/N _DM , the amount of CFAR processing can be reduced, and the circuit configuration can be simplified. In addition, since the CFAR unit 211 can add the power of each N _DM Doppler shift multiplexed signal, the SNR (Signal to Noise Ratio) can be improved by about (N _DM ) ^1/2 , and the radar detection performance of the radar device 10 can be improved. You can improve.

FIG. 12(b) shows an example of the output after applying the Doppler region compression process shown in equation (41) to the output of the Doppler analysis unit 210 shown in FIG. 12(a). As shown in FIG. 12(b), when N _DM =2, the CFAR unit 211 adds the power component of the Doppler frequency index f1 and the power component of f1-ΔFD by Doppler region compression processing and outputs the result. do. Similarly, as shown in FIG. 12(b), the CFAR unit 211 adds the power component of the Doppler frequency index f2 and the power component of f2-ΔFD and outputs the sum. Further, regarding the power component of the Doppler frequency index f3, since f3-ΔFD is smaller than -Ncode/2, the CFAR unit 211 combines the power component of the Doppler frequency index f3 with f3-ΔFD+Ncode (for example, N _DM =2 In the case of , the power component of f3+ΔFD) is added and output.

As a result of Doppler domain compression, the range of the Doppler frequency index fs_comp in the Doppler frequency domain is -∆FD/2 or more, ~, ∆FD/2-1 or less (if ∆FD = Ncode/N _DM , -Ncode/(2N _DM ) or more) , ~, Ncode/(2N _DM )-1 or less), and the range of CFAR processing is compressed, so the amount of calculations for CFAR processing can be reduced. Further, in FIG. 12, for example, the powers of reflected waves from three targets are added together, so that the SNR of the signal component is improved. Note that since the noise components are also power-combined, the SNR improvement effect is, for example, about (N _DM ) ^1/2 .

The CFAR unit 211 using Doppler region compression CFAR processing, for example, adaptively sets a threshold, and selects a distance index f _{b_cfar} , a Doppler frequency index f _{s_comp_cfar} , and N _DM Doppler multiplexing for which the received power is larger than the threshold. Received power information PowerFT(f _{b_cfar} , f _{s_comp_cfar} +(nfd-ceil(N _DM /2)-1) at the Doppler frequency index of the signal (f _{s_comp_cfar} + (nfd-ceil(N _DM /2)-1) × ΔFD) ×ΔFD), nfd=1,...,N _DM is output to the encoded Doppler demultiplexer 212.

Further, the CFAR section 211 outputs, for example, the distance index f _{b_cfar} and the Doppler frequency index f _{s_comp_cfar} to the Doppler multiplexing/demultiplexing section 213 .

Note that the phase rotation amount φ _ndm for providing the Doppler shift amount DOP _ndm is not limited to equation (5). For example, if each signal subjected to Doppler shift multiplexing has a phase rotation amount φ _ndm in which peaks are detected at regular intervals in the Doppler frequency domain output from the Doppler analysis unit 210, the CFAR unit 211 performs Doppler domain compression CFAR Processing can be applied.

For example, when Δf _MinInterval =1/(Tr(N _DM +N _int )L _OC ) is set using the equally spaced Doppler shift amount setting, the phase rotation amount φ _ndm is set according to equation (6), Each signal subjected to Doppler shift multiplexing is detected as a peak at an interval of ΔFD=Ncode/(N _DM +N _int ) in the Doppler frequency domain output from the Doppler analysis unit 210. Even in such a case, the CFAR unit 211 can apply Doppler region compression CFAR processing.

Next, an example of the operation of the encoded Doppler demultiplexer 212 shown in FIG. 1 will be described. The encoded Doppler demultiplexer 212 uses, for example, both the output of the first Doppler analyzer 210 and the output of the second Doppler analyzer 210 to generate linearly polarized waves (for example, horizontally polarized waves and vertically polarized waves). Separate the signals.

Note that, below, an example of the processing of the encoded Doppler multiplexer/demultiplexer 212 when the CFAR unit 211 uses Doppler region compression CFAR processing will be described.

The encoded Doppler demultiplexer 212 uses the distance index f _{b_cfar} , the Doppler frequency index f _{s_comp_cfar} , which are the outputs of _the CFAR unit 211, and the Doppler frequency index (f _{s_comp_cfar} +(nfd-ceil(N Based on the received power information PowerFT ₍ f _{b_cfar} , f _{s_comp_cfar} +(nfd-ceil(N _DM /2)-1)×ΔFD), nfd=1,~,N _DM , the output of the Doppler analysis unit 210 is used to separate the encoded Doppler multiplex-transmitted signals, determine the transmitting antenna 109 (for example, also referred to as determination or identification), and determine the Doppler frequency (for example, Doppler velocity or relative velocity). ).

As described above, when using the evenly spaced Doppler shift amount setting including the maximum equally spaced Doppler shift amount setting, the encoding section 107 of the phase rotation amount setting section 105 uses N _DM encoded Doppler multiplex numbers N, for example. _{DOP_CODE} (1), N _{DOP_CODE} (2),…, N _{DOP_CODE} (N _DM ) may not all be set to N _CM , but at least one encoded Doppler multiplex number may be set to a value smaller than N _CM . . For example, the coded Doppler demultiplexer 212 (1) performs code demultiplexing processing, detects a coded Doppler multiplex signal in which the number of coded Doppler multiplexes is set to be smaller than N _CM (for example, an unused signal that is not used for multiplex transmission); (detects the encoded Doppler multiplex signal used) and performs aliasing determination. Thereafter, the coded Doppler demultiplexer 212 performs Doppler code demultiplexing processing on the coded Doppler multiplex signal used for multiplex transmission, based on (2) the aliasing determination result.

Processes (1) and (2) in the above-mentioned encoded Doppler multiplexer/demultiplexer 212 will be explained below.

<(1) Return determination processing (detection processing of unused encoded Doppler multiplex signal)>
The encoded Doppler multiplexer/demultiplexer 212 performs Doppler aliasing determination processing, for example, with the Doppler range of the assumed target set to ±1/(2Tr).

Here, for example, if Ncode is a power of 2, the Doppler analysis unit 210 applies FFT processing to each code element, so the Doppler analysis unit 210 uses the output from the beat frequency analysis unit 208 at a period of (Loc×Tr). Perform FFT processing. Therefore, the Doppler range in which aliasing does not occur in the Doppler analysis unit 210 according to the sampling theorem is ±1/(2Loc×Tr). This Doppler range ±1/(2Loc×Tr) is further subjected to Doppler multiplexing using the Doppler multiplexing number N _DM . Therefore, the encoded Doppler demultiplexer 212 divides the Doppler range ±1/(2Tr), which is multiplied by Loc×N _DM , from the Doppler range ±1/(2Loc×N _DM ×Tr) in which aliasing due to Doppler multiplexing does not occur. The loopback determination process is performed assuming the above.

Here, as an example, a case will be described in which Nt=3, Doppler multiplexing number N _DM =2, and code multiplexing number N _CM =2. Here, the phase rotation amount φ _ndm for providing the Doppler shift amount DOP _ndm is assigned as shown in equation (5) based on the maximum equidistant Doppler shift amount setting, as an example. In this case, the phase rotation amount φ ₁ that provides the Doppler shift amount DOP ₁ = 0, and the phase rotation amount φ ₂ = π that provides the Doppler shift amount DOP ₂ . Furthermore, the encoding unit 107 uses two orthogonal codes Code ₁ ={1, 1} and Code ₂ ={j, -j} with code length Loc=2. Further, as shown in FIG. 13(a), N _{DOP_CODE} (1)=2 and N _{DOP_CODE} (2)=1 are used.

In this case, the coded Doppler demultiplexer 212 divides the Doppler range ±1/(2Loc×N _DM The aliasing determination process is performed assuming a Doppler range of ±1/(2Tr) of N _DM times).

Here, the Doppler component VFT _z ^noc (f _{b_cfar} , f _{s_comp_cfar} ), which is the output of the Doppler analysis unit 210 corresponding to the distance index f _{b_cfar} and the Doppler frequency index f _{s_comp_cfar} extracted in the CFAR unit 211, has ±1, for example. In the Doppler range of /(2Tr), Doppler components including aliasing as shown in (a) and (b) in FIG. 14 may be included.

For example, as shown in (a) in FIG. 14, when f _{s_comp_cfar} <0, in the Doppler range of ±1/(2Tr), f _{s_comp_cfar} -Ncode/N _DM , f _{s_comp_cfar} , f _{s_comp_cfar} +Ncode/N _DM , and f _{s_comp_cfar} +2Ncode/N _DM , there are 4 (=Loc×N _DM ) possible Doppler components (using ΔFD=Ncode/N _DM , f _{s_comp_cfar} -ΔFD, f _{s_comp_cfar} , f _{s_comp_cfar} +ΔFD, respectively) , and f _{s_comp_cfar} +2ΔFD).

For example, as shown in (b) in FIG. 14, when f _{s_comp_cfar} >0, in the Doppler range of ±1/(2Tr), f _{s_comp_cfar} -2Ncode/N _DM , f _{s_comp_cfar} -Ncode/N _DM , f There are four possible Doppler components (=Loc×N _DM ) of _{s_comp_cfar} and f _{s_comp_cfar} +Ncode/N _DM (using ΔFD=Ncode/N _DM , f _{s_comp_cfar} -2ΔFD and f _{s_comp_cfar} -, respectively). It can also be expressed as ΔFD, f _{s_comp_cfar} , and f _{s_comp_cfar} +ΔFD). These possible Doppler components (4 (=Loc×N _DM ) ways) for f _{s_comp_cfar} are called “Doppler component candidates” for f _{s_comp_cfar} . In the following, each Doppler region in which four (=Loc×N _DM ) Doppler component candidates exist will be expressed using an index “D _r ” indicating the Doppler folding range as shown in FIG. 14 . D _r is an index indicating the Doppler folding range, for example, an integer value in the range of D _r ∈{-ceil(Loc×N _DM /2),…, ceil(Loc×N _DM /2)-1} is used. . In FIG. 14, D _r =-2,~,1. Note that an area where D _r =0 is an area where Doppler aliasing does not occur, and an area where D _r ≠0 is an area where Doppler aliasing occurs. Further, the larger the absolute value of D _r is, the farther the Doppler region is from the Doppler region indicated by D _r =0.

The encoded Doppler demultiplexer 212 corrects phase changes corresponding to 4 (=Loc×N _DM ) Doppler components including aliasing in a Doppler range of ±1/(2Tr) as shown in FIG. , performs coded Doppler multiplex demultiplexing processing on a coded Doppler multiplex signal (for example, an unused coded Doppler multiplex signal) with the number of coded Doppler multiplexes set to be smaller than N _CM .

Then, the coded Doppler demultiplexer 212 determines whether each Doppler component candidate is a true Doppler component based on the received power of the component obtained by performing coded Doppler demultiplex processing on the unused coded Doppler multiplex signal. Determine whether

For example, the coded Doppler demultiplexer 212 selects the Doppler component whose received power is the smallest, obtained by performing coded Doppler demultiplexing processing based on the unused coded Doppler multiplex signal, among the Doppler component candidates for f _{s_comp_cfar} . The detected Doppler component may be determined to be a true Doppler component. For example, the encoded Doppler demultiplexer 212 may determine, among the Doppler component candidates for f _{s_comp_cfar} , a Doppler component with a received power different from the minimum received power as a false Doppler component.

This aliasing determination process can resolve ambiguity in the Doppler range of ±1/(2Tr). In addition, through this aliasing determination process, the range in which the Doppler frequency can be detected without ambiguity can be determined by comparing it with the Doppler range ±1/(2Loc×N _DM ×Tr)=±1/(8Tr) in which aliasing due to Doppler multiplexing does not occur. , -1/(2Tr) or more and less than 1/(2Tr).

By demultiplexing coded Doppler multiplexed signals based on unused coded Doppler multiplex signals, for example, for true Doppler components, the phase change of the Doppler component is correctly corrected, and the coded Doppler signals used for multiplex transmission are Orthogonality between the Doppler multiplex and the unused encoded Doppler multiplex is maintained. Therefore, the encoded Doppler multiplex signal code used for multiplex transmission and the unused encoded Doppler multiplex signal have no correlation, and the received power becomes approximately the noise level.

On the other hand, for example, for a false Doppler component, the phase change of the Doppler component is incorrectly corrected, and the orthogonality between the coded Doppler multiplex signal used for multiplex transmission and the unused coded Doppler multiplex signal is maintained. Not done. Therefore, a correlation component (interference component) occurs between the coded Doppler multiplex signal code used for multiplex transmission and the unused coded Doppler multiplex signal, and for example, received power greater than the noise level may be detected. Therefore, as described above, the coded Doppler demultiplexer 212 selects the Doppler component with the minimum received power among the Doppler component candidates for f _{s_comp_cfar} that has been subjected to coded Doppler demultiplexing based on the unused coded Doppler multiplex signal. may be determined to be a true Doppler component, and other Doppler components having received power different from the minimum received power may be determined to be false Doppler components.

For example, the encoded Doppler demultiplexer 212 corrects the phase change according to each Doppler component of the Doppler component candidates for f _{s_comp_cfar} based on the output of the Doppler analyzer 210 in each antenna system processor 201, and The received power P _DAR (f _{b_cfar} , f _{s_comp_cfar} , _Dr , nuc,nud) after code separation using the coded Doppler multiplexed signal is calculated according to equation (42).

Here, nuc,nud represent the index of the orthogonal code and the index of the Doppler multiplex signal, which are unused coded Doppler multiplex signals. For example, in the case of (b) in FIG. 13, the unused coded Doppler multiplex signal is indicated by an x in the figure, and is assigned a code of Code ₂ and a Doppler shift amount of DOP ₁ . There is. Therefore, the index of the orthogonal code to which the unused coded Doppler multiplex signal is assigned is nuc=2, nud=1.

In the following, a set of an orthogonal code index and a Doppler multiplex signal index used for a coded Doppler multiplex signal will be described as "DCI (orthogonal code index, Doppler multiplex signal index)." DCI(nuc, nud) represents, for example, an index of an orthogonal code to which an unused coded Doppler multiplex signal is assigned and an index of the Doppler multiplex signal. For example, in the case of FIG. 13(b), the unused encoded Doppler multiplex signal is assigned to DCI (2,1). Similarly, for example, in the case of (a) in FIG. 4, unused encoded Doppler multiplex signals are assigned to DCI(2, 3) and DCI(2, 4).

Here, Y _z (f _{b_cfar} , f _{s_comp_cfar} , _Dr , nuc,nud) is calculated based on the output of the Doppler analysis unit 210 in the z- _th antenna system processing unit 201 as shown in the following equation (43). This is the received signal after correcting the phase change according to each Doppler component of the Doppler component candidates for and separating the unused encoded Doppler multiplexed signal to which DCI (nuc, nud) is assigned.

In equations (42) and (43), in order to separate the unused coded Doppler multiplex signal to which DCI(nuc, nud) is assigned, the output VFTALL of the Doppler analysis unit 210 in the z-th antenna system processing unit 201 For _z (f _{b_cfar} , f _{s_comp_cfar} , _Dr , nud), the received power after code separation using the unused orthogonal code Code _nuc is calculated, and the received power is calculated for all antenna system processing units 201. The sum is calculated. Thereby, even when the received signal level is low, the accuracy of aliasing determination can be improved. However, instead of formula (42), the received power after demultiplexing the unused encoded Doppler multiplexed signal may be calculated for the output of the Doppler analysis unit 210 in some of the antenna system processing units 201. Even in this case, for example, in a range where the received signal level is sufficiently high, the amount of calculation processing can be reduced while maintaining the accuracy of the aliasing determination.

Note that in equations (42) and (43), D _r is an index indicating the Doppler folding range; for example, D _r ∈{-ceil(Loc×N _DM /2),~, ceil(Loc×N _DM Takes an integer value in the range /2)-1}.

は、要素数が等しいベクトル同士の要素毎の積を表す。例えば、n次ベクトルA=[a₁,..,a_n]及びB=[b₁,..,b_n]に対して、要素毎の積は以下の式(44)で表される。

Also, in equation (43),

represents the element-wise product of vectors with the same number of elements. For example, for n-dimensional vectors A=[a ₁ ,..,a _n ] and B=[b ₁ ,..,b _n ], the product of each element is expressed by the following equation (44).

Furthermore, in equation (43), the superscript T represents vector transposition, and the superscript * (asterisk) represents a complex conjugate operator.

In Equation (43), α(f _{s_comp_cfar} , _Dr ) represents a “Doppler phase correction vector”. For example, the Doppler phase correction vector α(f _{s_comp_cfar} , _Dr ) is set such that the Doppler frequency index f _{s_comp_cfar} extracted in the CFAR unit 211 is the output range of the Doppler analysis unit 210 (for example, Doppler range) that does not include Doppler folding. In this case, the Doppler phase rotation caused by the time difference in Doppler analysis between the Loc Doppler analysis units 210 in the Doppler return range _Dr is corrected.

For example, the Doppler phase correction vector α(f _{s_comp_cfar} , _Dr ) is expressed as in the following equation (45). The Doppler phase correction vector α (f _{s_comp_cfar} , _Dr ) shown in equation (45) is, for example, based on the Doppler analysis time of the output VFT _z ¹ (f _{b_cfar} , f _{s_comp_cfar} ) of the first Doppler analysis unit 210. _{Tr ,} ^2Tr _, ^～ _{, ( Loc-} 1) This is a vector whose elements are Doppler phase correction coefficients that correct the phase rotation in the Doppler component in the Doppler folding range _Dr of the Doppler frequency index f _{s_comp_cfar} caused by the time delay of the Tr. Note that the term D _r N _code /N _DM in Equation (45) can also be written as D _r ΔFD using ΔFD=N code/N _DM . Therefore, it is applicable not only to ΔFD=Ncode/ _NDM .

Such phase correction using the Doppler phase correction vector α (f _{s_comp_cfar} , _Dr ) corresponds to correcting a phase change according to each Doppler component in the Doppler component candidates for f _{s_comp_cfar} .

In addition, in equation (43), VFTALL _z (f _{b_cfar} , f _{s_comp_cfar} , D _r , nud) is the Loc number of Doppler analyzes in the z-th antenna system processing unit 201, for example, as shown in the following equation (46). ^DCI _{( nuc , _ _ _} This is a vector representation of the extracted components of the unused coded Doppler multiplex signal to which the coded Doppler multiplex signal (ndu) is assigned. However, noc=1,~,Loc, and takes an integer value in the range of D _r ={-ceil(Loc×N _DM /2),~, ceil(Loc×N _DM /2)-1}.

In Equation (46), N _{code FR} (D _r , nud)/N _DM represents the offset value of the Doppler index of the nud-th Doppler multiplexed signal with respect to f _{s_comp_cfar} in the Doppler folding range D _r . Note that the term N _{code FR} (D _r , nud)/N _DM in equation (46) can also be written as _FR (D _r , nud) ΔFD using ΔFD=N code/N _DM . Therefore, it is applicable not only to ΔFD=Ncode/ _NDM . Here, ndm=1,~, N _DM .

F _R (D _r , nud) is calculated once the Doppler folding range D _r and the phase rotation amount φ ₁ , φ ₂ , ~, φ _{N_DM} that gives the Doppler shift amount DOP ₁ , DOP ₂ , ~, DOP _{N_DM} are determined. Can be set in advance. Therefore, for example, the encoded Doppler demultiplexer 212 creates a table of the correspondence between the Doppler aliasing range _Dr and the amount of phase rotation, and F _R (D _r , nud), and calculates the correspondence between the Doppler aliasing range _Dr and the amount of phase rotation. Based on this, F _R (D _r , nud) may be read. Also, for example, the phase rotation amounts φ ₁ , φ ₂ , ~, φ _{N_DM} that give the Doppler shift amounts DOP ₁ , DOP ₂ , ~, DOP _{N_DM} satisfy -π≦φ ₁ <φ ₂ <...<φ _{N_DM} <π If satisfied, F _R (D _r , nud) can be expressed as in the following equation (47).

For example, the coded Doppler demultiplexer 212 uses the unused coded Doppler multiplexed signal to which DCI(nuc, nud) is assigned according to equations (42) and (43) to generate the received power P _DAR after code separation. (f _{b_cfar} , f _{s_comp_cfar} , _Dr , nuc,nud) respectively in the range of each _Dr ∈{-ceil(Loc×N _DM /2),…, ceil(Loc×N _DM /2)-1} calculate.

Then, the encoded Doppler multiplexer/demultiplexer 212 detects the _Dr whose received power P _DAR (f _{b_cfar} , f _{s_comp_cfar} , _Dr , nuc, nud) is the minimum among the ranges of each _Dr. In the following, as shown in the following equation (48), the D _r for which the received power P _DAR (f _{b_cfar} , f _{s_comp_cfar} , D _r , nuc, nud) is the minimum within the range of each D _r is referred to as "D _rmin ". Expressed as

Note that when there are multiple unused coded Doppler multiplex signals, the coded Doppler _{demultiplexer} 212 uses _{the following} formula ( ₄₉ ), the received power Pall _DAR (f _{b_cfar} , f _{s_comp_cfar} , _Dr ) after code separation using all unused orthogonal codes may be used.

By determining the received power after code separation using all unused orthogonal codes, the accuracy of the folding process can be improved even when the received signal level is low.

For _example , the encoded Doppler _{demultiplexer} 212 performs Pall _DAR ( _{f b_cfar} , f _{s_comp_cfar} , _Dr ) is calculated, and _Dr (for example, _Drmin ) with which Pall _DAR (f _{b_cfar} , f _{s_comp_cfar} , _Dr ) is minimized is detected. For example, when using equation (49), below, as shown in equation (50) below, _Dr that provides the minimum received power in the range of each _Dr is expressed as "D _rmin ".

Further, the coded Doppler demultiplexer 212 generates the minimum received power Pall _DAR (f _{b_cfar} , f _{s_comp_cfar} , D _rmin ) and the received power PowerFT_comp(f _{b_cfar} , f _{s_comp_cfar} ) of equation (41), which is obtained by adding the power of each signal peak position Doppler shift-multiplexed in the CFAR unit 211, the certainty of the aliasing judgment is determined. (for example, measurement) may be performed. In this case, the encoded Doppler multiplexer/demultiplexer 212 may determine the probability of aliasing determination, for example, according to the following equations (51) and (52).

For example, the _{encoded Doppler multiplexer} / _{demultiplexer} 212 determines that _DCI When the minimum received power Pall _DAR (f _{b_cfar} , f _{s_comp_cfar} , _Drmin ) after code separation using an unused coded Doppler multiplexed signal to which (nuc, nud) is assigned is small (for example, Equation (51) ), it is determined that the loopback determination is sufficiently probable. In this case, the radar device 10 may perform subsequent processing (for example, code separation processing), for example.

On the other hand, for example, the coded Doppler demultiplexer 212 selects the unused coded Doppler multiplexer to which DCI(nuc, nud) is assigned, rather than the value obtained by multiplying PowerFT_comp(f _b , f _{s_comp_cfar} ) by Threshold _DR . If the minimum received power Pall _DAR (f _{b_cfar} , f _{s_comp_cfar} , _Drmin ) after code separation using a signal is equal or large (e.g., equation (52)), the accuracy of the aliasing determination is not sufficient and the reliability of the aliasing determination is It is determined that the quality is low (for example, noise component). In this case, the radar device 10 does not need to perform subsequent processing (for example, code separation processing), for example.

Through such processing, errors in aliasing determination can be reduced and noise components can be removed. Note that the predetermined value Threshold _DR may be set, for example, in a range from 0 to less than 1. As an example, considering that noise components are included, Threshold _DR may be set in a range of about 0.1 to 0.5.

The operation example of the return process has been described above.

<(2) Doppler code separation processing of coded Doppler multiplex signal used for multiplex transmission>
The coded Doppler demultiplexer 212 performs coded Doppler demultiplexer processing on the coded Doppler multiplex signal used for multiplex transmission based on the return determination result.

For example, the encoded Doppler demultiplexer 212 applies Equation (43) based on D _rmin , which is the return judgment result in the return judgment process, as shown in Equation (53) below. Separate and receive coded Doppler multiplexed signals to which DCI (ncm, ndm) is assigned. For example, the coded Doppler multiplex demultiplexer 212 performs separation processing using the following equation (53) to separate and receive the coded Doppler multiplexed signal to which the DCI (ncm, ndm) used for multiplex transmission is assigned. conduct. In the aliasing determination process, the index (D _rtrue ), which is the true Doppler aliasing range, can be determined within a Doppler range of -1/(2Tr) or more and less than 1/(2Tr) (for example, D _rmin = D _rtrue ), the encoded Doppler demultiplexer 212 uses a Doppler range of -1/(2Tr) or more and less than 1/(2Tr) between orthogonal codes used for code multiplexing. It is possible to set the correlation value to zero, and it is possible to perform separation processing that suppresses interference between code-multiplexed signals.

Here, Y _z (f _{b_cfar} , f _{s_comp_cfar} , Dr _min , ncm,ndm) is the output of the distance index f _{b_cfar} and the Doppler frequency index f _{s_comp_cfar} of the Doppler analysis unit 210 in the z-th antenna system processing unit 201. For the _ndm -th coded Doppler multiplex signal VFTALL _z (f _{b_cfar} , f _{s_comp_cfar} , _Drmin , ndm) in the Doppler range _Drmin , the code-separated output (for example, code It is possible to separate the coded Doppler multiplexed signal to which the DCI (ncm, ndm) used for multiplex transmission is assigned. Note that z=1,~,Na, and ncm=1,~,N _CM .

Through the above-described code separation processing, the radar device 10 performs aliasing assuming a Doppler range ±1/(2Tr) that is Loc times the Doppler range ±1/(2Loc×Tr) in which aliasing does not occur in the Doppler analysis unit 210. Based on the determination result, the encoded Doppler multiplex signal to which the DCI (ncm, ndm) used for multiplex transmission is assigned can be separated and received.

Further, since the encoded Doppler multiplex signal to which DCI (ncm, ndm) is assigned is transmitted from the transmit antenna Tx#[ncm, ndm], the transmit antenna 109 can also be determined. For example, the radar device 10 can separate and receive a coded Doppler multiplex signal to which DCI (ncm, ndm) is assigned and transmitted from the transmitting antenna Tx#[ncm, ndm].

In addition, for example, during coded Doppler demultiplexing processing, the radar device 10 performs Doppler phase correction including Doppler folding (for example, Doppler phase correction vector α (f _{s_comp_cfar} , _Dr ).These phase corrections correspond to correcting the phase change according to each Doppler component in the Doppler component candidates for f _{s_comp_cfar.For} this reason, the mutual difference between code multiplexed signals is Interference can be reduced, for example, to about the noise level. For example, in the radar device 10, intersymbol interference can be reduced, and the influence on deterioration of detection performance in the radar device 10 can be suppressed.

An example of the operation of the encoded Doppler demultiplexer 212 has been described above.

In FIG. 1, the Doppler demultiplexer 213 includes, for example, Doppler demultiplexers 213-1 to 213-Loc corresponding to the outputs of the Loc Doppler analysis units 210, respectively. In the example shown in FIG. 1, Loc=2, and the first Doppler demultiplexer 213 (or Doppler demultiplexer 213-1) and the second Doppler demultiplexer 213 (or Doppler demultiplexer 213-1) 213-2).

The first Doppler demultiplexer 213 converts the output of the first Doppler analyzer 210 (or referred to as Doppler analyzer 210-1) to the distance index f _{b_cfar} and Doppler frequency index f _{s_comp_cfar} input from the CFAR unit 211. , is output to the direction estimation section 214. At this time, the first Doppler demultiplexer 213 may use, for example, _Drmin , which is the Doppler return determination result inputted from the encoded Doppler demultiplexer 212.

For example, in the example shown in FIG. 1, the first Doppler demultiplexer 213 uses the output VFT _z ¹ (f _{b_cfar} , f _{s_comp_cfar} +( _N F _R (D _rmin , ndm_BF)/N _DM )) is output to the direction estimation unit 214. Here, ndm_BF is one of 1, to N _DM , and the plurality of transmitting antennas 109 to which the ndm_BFth Doppler multiplexed signal is assigned are, for example, the transmitting antennas 109 that satisfy the above-mentioned conditions for adjacent arrangement. It's a combination.

The first Doppler demultiplexer 213 uses, for example, the output of the first Doppler analyzer 210 to separate circularly polarized signals. As a result, the first Doppler multiplexer/demultiplexer 213 obtains a received signal of the reflected wave of the transmitted signal which becomes a circularly polarized wave. Note that z=1 to Na.

In FIG. 1, the second Doppler demultiplexer 213 uses the second Doppler analyzer 210 (or Doppler analyzer 210-2) for the distance index f _{b_cfar} and Doppler frequency index f _{s_comp_cfar} input from the CFAR unit 211. ) is output to the direction estimation unit 214. At this time, the second Doppler demultiplexer 213 may use, for example, _Drmin , which is the Doppler return determination result inputted from the encoded Doppler demultiplexer 212.

_{For example} , in the _example ^shown in FIG _. F _R (D _rmin , ndm_BF)/N _DM )) is output to the direction estimation unit 214. Here, ndm_BF is one of 1,..., N _DM , and the plurality of transmitting antennas 109 to which the ndm_BFth Doppler multiplexed signal is assigned are, for example, the transmitting antennas 109 that satisfy the above-mentioned conditions for adjacent arrangement. It's a combination.

The second Doppler demultiplexer 213 uses, for example, the output of the second Doppler analyzer 210 to separate the circularly polarized signal. As a result, the second Doppler multiplexer/demultiplexer 213 obtains a received signal of the reflected wave of the transmitted signal which becomes a circularly polarized wave. Note that z=1 to Na.

In addition, the second Doppler demultiplexer 213 receives the received signal of the reflected wave from the circularly polarized transmission signal, which is the output of the first Doppler demultiplexer 213. The received signal is a reflected wave corresponding to the transmitted signal, which becomes a circularly polarized wave in a rotation direction different from the rotation direction of the circularly polarized wave corresponding to the output. For example, if the circularly polarized wave corresponding to the output of the first Doppler demultiplexer 213 is a left-handed circularly polarized wave, the circularly polarized wave corresponding to the output of the second Doppler demultiplexer 213 is a right-handed circularly polarized wave. It becomes a wave. Furthermore, for example, if the circularly polarized wave corresponding to the output of the first Doppler demultiplexer 213 is a right-handed circularly polarized wave, the circularly polarized wave corresponding to the output of the second Doppler demultiplexer 213 is a left-handed circularly polarized wave. It becomes circularly polarized.

Hereinafter, based on the rotation of the circularly polarized wave corresponding to the output of the first Doppler multiplexer/demultiplexer 213, the circularly polarized wave will be referred to as "normal circularly polarized wave." The circularly polarized wave corresponding to the output of the second Doppler multiplexer/demultiplexer 213 is a counter-rotating circularly polarized wave of normal rotation, and is therefore referred to as a "counter-rotating circularly polarized wave."

In FIG. 1, the direction estimation unit 214 uses the transmitting antenna Tx#[ _{ncm , ndm} Target direction estimation processing is performed based on the separated received signal Y _z (f _{b_cfar} , f _{s_comp_cfar} , _Drmin , ncm, ndm) of the coded Doppler multiplex signal to which DCI (ncm, ndm) is assigned and transmitted from ] (hereinafter referred to as "direction estimation processing for linearly polarized waves").

The direction estimation unit 214 also estimates the direction of the target based on the output from the first Doppler analysis unit 210 (Doppler analysis unit 210-1 in FIG. 1) that is input from the first Doppler demultiplexing unit 213. processing (hereinafter referred to as "direction estimation processing for normal circularly polarized waves").

The direction estimation unit 214 also estimates the direction of the target based on the output from the second Doppler analysis unit 210 (Doppler analysis unit 210-2 in FIG. 1) that is input from the second Doppler demultiplexing unit 213. processing (hereinafter referred to as "direction estimation processing for anti-circularly polarized waves").

The direction estimation process in the direction estimation unit 214 may include, for example, a direction estimation process for linearly polarized waves, a direction estimation process for normal circularly polarized waves, and a direction estimation process for anticircularly polarized waves. Each direction estimation process will be explained below.

<Direction estimation processing for linearly polarized waves>
For example, the direction estimation unit 214 generates a virtual reception array correlation vector h (f _{b_cfar} , f _{s_comp_cfar} ) as shown in the following equation (54) based on the output of the encoded Doppler demultiplexer 212, and performs the direction estimation process. I do.

The virtual receiving array correlation vector h (f _{b_cfar} , f _{s_comp_cfar} ) includes Nt×Na elements that are the product of the number of transmitting antennas Nt and the number of receiving antennas Na.

The direction estimation unit 214 performs a process of estimating the direction of the reflected wave signal from the target based on the phase difference between each receiving antenna 202 using the virtual receiving array correlation vector h (f _{b_cfar} , f _{s_comp_cfar} ).

Here, the virtual receive array correlation vector h(f _{b_cfar} , f _{s_comp_cfar} ) includes reflected received signals of signals transmitted from different linearly polarized antennas (eg, a vertically polarized antenna and a horizontally polarized antenna). Therefore, the direction estimation unit 214 extracts an element that is a combination of the polarization of the predetermined transmitting antenna 109 and the polarization of the receiving antenna 202 from the virtual receiving array correlation vector h (f _{b_cfar} , f _{s_comp_cfar} ). The direction estimation process may be performed using a virtual receiving array correlation vector consisting of elements. This makes it possible to obtain direction estimation results for each polarization of the predetermined transmitting antenna 109 and receiving antenna 202.

<Direction estimation processing for normal circular polarization>
The direction estimation unit 214 generates a normal circular polarization virtual receiving array correlation vector as shown in the following equation (55) based on the output of the first Doppler analysis unit 210 input from the first Doppler demultiplexing unit 213. h _c1 (f _{b_cfar} , f _{s_comp_cfar} ) is generated, and direction estimation processing for the target's normal circularly polarized wave is performed.

Here, the normal circularly polarized virtual reception array correlation vector h _c1 (f _{b_cfar} , f _{s_comp_cfar} ) is the output of the first Doppler analysis unit 210 (for example, VFT _z ¹ (f _{b_cfar} , f _{s_comp_cfar} +(N _code F _R (D _rmin , ndm_BF)/N _DM ))), the signals are code-multiplexed using the same Doppler multiplexing, beam-transmitted by adjacent transmitting antennas 109 with different linear polarizations, and circularly polarized signals. includes the reflected wave received signal. For example, when there are N _BF beam transmitting antennas (for example, when ndm_BF=1,~, N _BF ), the normal circularly polarized virtual receiver array correlation vector h _c1 (f _{b_cfar} , f _{s_comp_cfar} ) is N _BF × Contains Na elements. The direction estimation unit 214 uses the right circularly polarized virtual receiving array correlation vector h _c1 (f _{b_cfar} , f _{s_comp_cfar} ) to estimate the direction of the reflected wave signal from the target based on the phase difference between each receiving antenna 202. You may perform processing to do so.

<Direction estimation processing for anti-circularly polarized waves>
The direction estimation section 214 performs direction estimation processing for the anti-circularly polarized wave of the target based on the output of the second Doppler analysis section 210 inputted from the second Doppler demultiplexing section 213. The direction estimation unit 214 generates, for example, a counter-circularly polarized wave virtual receiving array correlation vector h _c2 (f _{b_cfar} , f _{s_comp_cfar} ) as shown in the following equation (56), and estimates the direction for the target's counter-circularly polarized wave. Perform processing.

Here, the counter-rotating circularly polarized virtual reception array correlation vector h _c2 (f _{b_cfar} , f _{s_comp_cfar} ) is the output of the second Doppler analysis unit 210 (for example, VFT _z ² (f _{b_cfar} , f _{s_comp_cfar} +(N _code F _R (D _rmin , ndm_BF)/N _DM ))), the signals are code-multiplexed using the same Doppler multiplexing, beam-transmitted by adjacent transmitting antennas 109 with different linear polarizations, and circularly polarized signals. includes the reflected wave received signal. For example, when there are N _BF beam transmitting antennas (for example, when ndm_BF=1,~, N _BF ), the anti-rotating virtual receiving array correlation vector h _c2 (f _{b_cfar} , f _{s_comp_cfar} ) is N _BF ×Na Contains elements.

The direction estimation unit 214 uses the counter-rotating circularly polarized virtual receiving array correlation vector h _c2 (f _{b_cfar} , f _{s_comp_cfar} ) to estimate the direction of the reflected wave signal from the target based on the phase difference between each receiving antenna 202. Used for processing.

[Antenna arrangement example 1]
Hereinafter, examples of direction estimation processing for linearly polarized waves, direction estimation processing for normal circularly polarized waves, and direction estimation processing for anticircularly polarized waves in the direction estimation unit 214 will be described using antenna arrangement examples.

For example, when the number of transmitting antennas used for multiplex transmission is Nt = 4, the number of Doppler multiplexing N _DM = 2, the number of code multiplexing N _CM = 2, and the orthogonal code sequence Code ₁ = {1, 1} with code length Loc = 2. , Code ₂ = {j, -j}, and the number of encoded Doppler multiplexes is N _{DOP_CODE} (1)=2, N _{DOP_CODE} (2)=2. Note that the number of beam transmitting antennas is N _BF =2, and _{ndm_BF} =1 and _{ndm_BF} =2 are used as indices of Doppler multiplexed signals used for the beam transmitting antennas.

In FIG. 15, for example, in the radar device 10, four transmitting antennas 109 (Tx#1, Tx#2, Tx#3, and Tx#4) arranged horizontally transmit horizontally polarized waves from the left antenna. (denoted as “H”) transmitting antenna Tx#[1, 1], vertically polarized (denoted as “V”) transmitting antenna Tx#[2, 1], horizontally polarized transmitting antenna Tx#[1, 2], and vertically polarized transmitting antenna Tx#[2, 2].

In FIG. 15, two adjacent transmitting antennas Tx#1 (Tx#[1, 1]) and Tx#2 (Tx#[2, 1]) from the left side have the same Doppler multiplexing (Doppler shift amount = DOP ₁₎ . ) to code-multiplex the radar transmission signal. Therefore, in FIG. 15, a beam transmission antenna is formed by Tx#1 and Tx#2, and circularly polarized waves are transmitted. In addition, for transmitting antennas Tx#1 (Tx#[1, 1]) and Tx#2 (Tx#[2, 1]), each transmission cycle is The normal circularly polarized wave or the counterclockwise circularly polarized wave may be switched and transmitted.

Similarly, in FIG. 15, two adjacent transmitting antennas Tx#3 (Tx#[1, 2]) and Tx#4 (Tx#[2, 2]) from the right side have the same Doppler multiplexing (Doppler shift amount). =DOP ₂ ) to code-multiplex the radar transmission signal. Therefore, in FIG. 15, a beam transmission antenna is formed by Tx#3 and Tx#4, and circularly polarized waves are transmitted. In addition, for transmitting antennas Tx#1 (Tx#[1, 2]) and Tx#2 (Tx#[2, 2]), each transmission cycle is The normal circularly polarized wave or the counterclockwise circularly polarized wave may be switched and transmitted.

In FIG. 15, the number of beam transmitting antennas N _BF =2. Below, the beam transmitting antenna for Tx#1 and Tx#2 in FIG. 15 may be referred to as "Tx#5." Further, the beam transmitting antenna for Tx#3 and Tx#4 in FIG. 15 may be referred to as "Tx#6". For example, Tx#5 and Tx#6 can be treated equivalently as antennas that transmit transmission by switching between normal circularly polarized waves and counterclockwise circularly polarized waves every transmission period.

In addition, here, Tx#5 and Tx#6 will be explained using codes that transmit circularly polarized waves in the same rotation direction, normal or counterclockwise, for each transmission period, but the Tx#5 and Tx#6 are not limited to this. #5 and Tx#6 may use circularly polarized waves with different rotation directions.

Further, as shown in FIG. 15, the number of receiving antennas Na is eight (for example, Rx#1 to Rx#8), including four types of antennas with different polarizations. In the example shown in FIG. 15, Rx#1 and Rx#2 are horizontally polarized (H) receiving antennas, Rx#3 and Rx#4 are vertically polarized (V) receiving antennas, and Rx#5 is a horizontally polarized (H) receiving antenna. and Rx#6 are receiving antennas for positive circularly polarized waves (C), and Rx#7 and Rx#8 are receiving antennas for reverse circularly polarized waves (R).

Note that the number of receiving antennas Na is not limited to eight, and may be any other number, for example. Further, the types of polarized waves used in the receiving antenna 202 are not limited to four types, but may be three types, two types, or one type. Furthermore, in addition to horizontally polarized waves and vertically polarized waves, linearly polarized waves tilted in an oblique direction, for example, polarized waves tilted in the ±45° direction may be used.

For example, if radar transmission signals are transmitted with equal power from adjacent Tx#1 (Tx#[1, 1]) and Tx#2 (Tx#[2, 1]), Tx#1 and Tx#2 The midpoint position becomes the phase center of beam transmitting antenna Tx#5 (x mark shown in (a) of FIG. 15). Similarly, for example, if radar transmission signals are transmitted with equal power from adjacent Tx#3 (Tx#[1, 2]) and Tx#4 (Tx#[2, 2]), Tx#3 and The midpoint position of Tx#4 becomes the phase center of beam transmitting antenna Tx#6 (x mark shown in (a) of FIG. 15).

Note that when radar transmission signals are not transmitted with equal power from the transmitting antennas 109 that constitute the beam transmitting antenna, the position (from each transmitting antenna It can be treated as transmission by a beam transmitting antenna with the phase center of the sub-array being at the center of gravity of the transmit power.

Transmitting antennas Tx#1 to Tx#4 (for example, Nt transmitting antennas 109) and receiving antennas Rx#1 to Rx#8 (for example, Na receiving antennas 202) as shown in FIG. 15(a). From the arrangement, virtual receiving antenna (or MIMO virtual antenna) arrangements VA#1 to VA#32 as shown in FIG. 15(b) are constructed.

In addition, in (a) of FIG. 15, from the arrangement of right circularly polarized beam transmitting antennas Tx#5 and Tx#6 (for example, N _BF transmitting antennas) and receiving antennas Rx#1 to Rx#8, , arrangements CA#1 to CA#16 of right-rotating circularly polarized virtual receiving antennas are configured as shown in FIG. 15(c). Arrangements CA#1 to CA#16 of the right circularly polarized virtual receiving antennas are also expressed as VA#33 to VA#48, for example.

In addition, in (a) of FIG. 15, from the arrangement of anti-rotating circularly polarized beam transmitting antennas Tx#5 and Tx#6 (for example, N _BF transmitting antennas) and receiving antennas Rx#1 to Rx#8, , The arrangement RA#1 to RA#16 of the counter-rotating circularly polarized virtual receiving antenna is composed of the arrangement RA#1 to RA#16 of the counter-rotating circularly polarized virtual receiving antenna as shown in FIG. 15(d). , for example, also expressed as VA#33 to VA#48. Arrangements RA#1 to RA#16 of the counter-rotating circularly polarized virtual receiving antennas are the same as the arrangements CA#1 to CA#16 of the right-rotating circularly polarized virtual receiving antennas, respectively.

Here, the arrangement of the virtual receiving antenna (virtual receiving array) is, for example, the position of the transmitting antenna 109 forming the transmitting array antenna (e.g., the position of the feeding point) and the position of the receiving antenna 202 forming the receiving array antenna (e.g. , the position of the feeding point), it may be expressed as the following equation (57).

Here, the position coordinates of the transmitting antenna 109 (for example, Tx#n) constituting the transmitting array antenna are expressed as (X _{T_#n} , Y _{T_#n} ) (for example, n=1,~, Nt+N _BF ). , the position coordinates of the receiving antenna 202 (for example, Rx#m) constituting the receiving array antenna are expressed as (X _{R_#m} ,Y _{R_#m} ) (for example, m=1, ~, Na), and the virtual receiving array antenna is The position coordinates of the virtual antenna VA#k constituting the are expressed as (X _{V_#k} , Y _{V_#k} ) (for example, k=1, ~, (Nt+N _BF )×Na).

Note that in equation (57), for example, VA#1 is expressed as the position reference (0,0) of the virtual reception array.

FIG. 15(b) shows an example of virtual receiving antenna arrangement when using the arrangement of Tx #1 to #4 and Rx #1 to #8 shown in FIG. 15(a). The virtual receiving antenna includes 32 antenna elements, each arrangement shown as VA#1 to VA#32.

Note that the combination of the transmitting antenna polarization (for example, H or V) and the receiving antenna polarization (for example, H, V, C, or R) in each virtual receiving antenna is different for each antenna. /receiving antenna polarization). For example, a virtual receiving antenna for horizontally polarized wave transmission and horizontally polarized wave reception is written as "H/H". (c) and (d) in FIG. 15 are also written in the same manner. In addition, the following notation also follows this.

Since the arrangement of Tx #1 to #4 and Rx #1 to #8 in Fig. 15 (a) is a one-dimensional arrangement on the X axis (horizontal direction in Fig. 15), the virtual receiving antennas are also placed on top. In addition, since the arrangement includes overlapping arrangements on the X-axis, in FIG. 15(b), they are shown shifted in the vertical direction, but they are arranged one-dimensionally at the respective positions on the X-axis. (c) and (d) in FIG. 15 are also written in the same manner. In addition, the following notation also follows this.

FIG. 15(c) shows a virtual receiving antenna arrangement when using beam transmitting antennas Tx #5 and #6 and Rx #1 to #8 that transmit normal circularly polarized waves. The virtual reception antenna has an array arrangement of 16 antenna elements (eg, CA#1 to CA#16). Note that CA#1 to CA#16 correspond to VA#33 to VA#48, respectively.

Further, (d) in FIG. 15 shows a virtual receiving antenna arrangement when using beam transmitting antennas Tx #5 and #6 and Rx #1 to #8 that transmit anti-rotating circularly polarized waves. The virtual reception antenna has an antenna arrangement of 16 antenna elements (for example, RA#1 to RA#16) similar to FIG. 15(c). Note that RA#1 to RA#16 correspond to VA#33 to VA#48, respectively.

In this way, the virtual receiving antenna arrangement using the beam transmitting antenna enables the radar device 10 to transmit using more polarized waves. Furthermore, by combining the polarization of the transmitting antenna 109 and the polarization of the receiving antenna 202, in addition to the combination of transmitting and receiving antennas with the same polarization, the combination of orthogonal polarized waves (also referred to as "cross polarized waves") (for example, A received signal can be obtained using a combination of horizontally polarized waves and vertically polarized waves, or a combination of left-handed circularly polarized waves and right-handed circularly polarized waves. In the radar device 10, the detection performance or identification performance can be improved by utilizing the fact that the receiving characteristics of the reflected waves of the target change depending on the combination of transmitted and received polarized waves.

<Direction estimation processing for linearly polarized waves>
For example, when performing direction estimation processing using a horizontally polarized antenna (H) for both transmission and reception, the direction estimation unit 214 uses four virtual reception antennas consisting of VA#1, VA#2, VA#17, and VA#18. Extract and perform direction estimation processing.

The virtual receiving array correlation vector h (f _{b_cfar} , f _{s_comp_cfar} ) is a column vector, and the first to Nt×Na elements included in this vector are the virtual antennas VA#1 to VA#(Nt×Na ) represents the received signal. _For example, the direction estimation unit 214 calculates the first, _second , and 17. The direction of arrival is estimated using the partial array correlation vector extracted from the 18th element. The extracted partial array correlation vector is expressed as h _sub (f _{b_cfar} , f _{s_comp_cfar} ). Further, the number of dimensions of h _sub (f _{b_cfar} , f _{s_comp_cfar} ) is assumed to be N _sub .

The direction estimation unit 214 calculates a spatial profile by making the azimuth direction θ in the direction estimation evaluation function value P _H (θ, f _{b_cfar} , f _{s_comp_cfar} ) variable within a defined angular range, for example. The direction estimation unit 214 extracts a predetermined number of maximum peaks of the calculated spatial profile in descending order of magnitude, and outputs the azimuth direction of the maximum peak as an estimated direction of arrival value (eg, positioning output).

Note that there are various methods for determining the direction estimation evaluation function value P _H (θ, f _{b_cfar} , f _{s_comp_cfar} ) depending on the direction of arrival estimation algorithm. For example, an estimation method using an array antenna disclosed in Non-Patent Document 3 may be used.

For example, the beamformer method can be expressed as the following equation (58). Other methods such as Capon and MUSIC are also applicable.

Here, in equation (58), the superscript H is a Hermitian transposition operator. Further, a _sub (θ _u ) indicates the direction vector of the extracted partial virtual receiving arrays VA#1, VA#2, VA#17, and VA#18 with respect to the arriving wave in the azimuth direction θ _u . For example, if virtual receiving antennas consisting of VA#1, #2, #17, #18 are arranged in a straight line at equal intervals _DH , the virtual receiving array VA#1, VA#2, VA#17, VA# The 18 direction vectors can be expressed as in the following equation (59).

Further, the azimuth direction θ _u is a vector that is changed by an azimuth interval β ₁ within the azimuth range in which direction of arrival is estimated. For example, θ _u is set as follows.
θ _u = θmin + uβ ₁ , u=0,…, NU
NU=floor[(θmax-θmin)/β ₁ ]+1
Here, floor(x) is a function that returns the largest integer value that does not exceed the real number x.

In addition, in equation (58), D _cal is an array correction coefficient that corrects the phase deviation and amplitude deviation between the antennas of the partial virtual reception array VA#1, #2, #17, #18, and is an N _sub- order matrix containing coefficients that reduce the effects of inter-element coupling. If the coupling between the antennas of the virtual receiving array can be ignored, D _cal becomes a diagonal matrix, and the diagonal components include array correction coefficients that correct phase deviations and amplitude deviations between the transmitting array antennas and between the receiving array antennas. .

For example, when performing direction estimation processing when vertically polarized antennas are used for both transmission and reception, the direction estimation unit 214 uses four virtual reception antennas consisting of VA#11, VA#12, VA#27, and VA#28. Then, direction estimation processing similar to the processing described above is performed.

Further, for example, the direction estimation unit 214 may perform direction estimation processing using cross-polarized waves. For example, when performing direction estimation processing using a vertically polarized antenna for transmission and a horizontally polarized antenna for reception, the direction estimation unit 214 includes VA#9, VA#10, VA#25, VA Four virtual receiving antennas consisting of #26 are extracted and direction estimation processing similar to the processing described above is performed.

The present invention is not limited to the combination of a transmitting polarized antenna and a receiving polarized antenna as described above, but may also be a combination of different combinations of transmitting and receiving polarized antennas.

For example, the direction estimation unit 214 is not limited to the direction estimation process using a virtual receiving array in which the combination of transmitting and receiving polarized antennas is the same, but also in the direction estimation process using a virtual receiving array having a different combination of transmitting and receiving polarized antennas. Estimation processing may also be performed. For example, the direction estimation unit 214 extracts all virtual reception antennas obtained from the output of the encoded Doppler multiplexing unit 212 consisting of VA#1 to VA#32, and performs direction estimation processing similar to the processing described above. Good too. In this case, the direction estimation unit 214 performs direction estimation processing using a received signal that includes a combination of different types of polarization (for example, H/V, H/C, etc.), thereby eliminating polarization dependence. A low direction estimation result is obtained. Furthermore, since the direction estimation unit 214 performs direction estimation processing using all virtual reception antennas, the reception SNR is improved and the detection performance of the radar device 10 can be improved. Furthermore, since direction estimation processing is performed using the maximum aperture length of the available virtual reception antenna, angular resolution is also improved.

For example, the direction estimation unit 214 performs direction estimation processing using the same type of polarized antenna for both transmission and reception as described above, and direction estimation processing using a combination of different types of polarization, and calculates the direction estimation results of both of these. may be used as the direction estimation processing result. As a result, a direction estimation processing result with a high polarization dependence and a direction estimation processing result with a low polarization dependence can be obtained. The results of such direction estimation processing may be input to a processing unit for each object mark (not shown) to perform target object identification processing.

<Direction estimation processing for normal circular polarization>
For example, when performing direction estimation processing using a right circularly polarized antenna (C) for both transmission and reception, the direction estimation unit 214 uses four virtual reception antennas consisting of CA#5, CA#6, CA#13, and CA#14. is extracted and direction estimation processing is performed.

The normal circularly polarized virtual receiver array correlation vector h _c1 (f _{b_cfar} , f _{s_comp_cfar} ) is a column vector, and the first to N _BF ×Na elements included in this are the virtual antenna VA#( Represents the received signal from Nt×Na+1) to VA#(Nt+N _BF )×Na, or the received signal from CA#1 to CA#(N _BF ×Na). For example, the direction estimation unit 214 calculates the order of the normal circularly polarized virtual receiving array correlation vectors h _c1 (f _{b_cfar} , f _{s_comp_cfar} ) corresponding to the received signals of CA#5, CA#6, CA#13, and CA#14. Direction of arrival estimation is performed using partial array correlation vectors obtained by extracting the 5th, 6th, 13th, and 14th elements. The extracted partial array correlation vector is expressed as h _c1sub (f _{b_cfar} , f _{s_comp_cfar} ). Further, the number of dimensions of h _c1sub (f _{b_cfar} , f _{s_comp_cfar} ) is assumed to be N _c1sub .

The direction estimation unit 214 calculates a spatial profile by making the azimuth direction θ in the direction estimation evaluation function value P _Hc1 (θ, f _{b_cfar} , f _{s_comp_cfar} ) variable within a defined angular range, for example. The direction estimation unit 214 extracts a predetermined number of maximum peaks of the calculated spatial profile in descending order of magnitude, and outputs the azimuth direction of the maximum peak as an estimated direction of arrival value (eg, positioning output).

Note that there are various methods for determining the direction estimation evaluation function value P _Hc1 (θ, f _{b_cfar} , f _{s_comp_cfar} ) depending on the direction of arrival estimation algorithm. For example, an estimation method using an array antenna disclosed in Non-Patent Document 3 may be used.

For example, the beamformer method can be expressed as the following equation (60). Other methods such as Capon and MUSIC are also applicable.

Further, a _c1sub (θ _u ) indicates the direction vector of the extracted partial virtual reception arrays CA#5, CA#6, CA#13, and CA#14 with respect to the arriving wave in the azimuth direction θ _u .

In addition, D _c1cal is the array correction coefficient that corrects the phase deviation and amplitude deviation between the antennas of the partial virtual reception array CA#5, CA#6, CA#13, CA#14, and the inter-element coupling between the antennas. It is a matrix of order N _c1sub containing coefficients that reduce the influence. If the coupling between the antennas of the virtual receiving array can be ignored, D _c1cal becomes a diagonal matrix, and the diagonal components include array correction coefficients that correct the phase deviation and amplitude deviation between the transmitting array antennas and between the receiving array antennas. .

Further, for example, the direction estimation unit 214 may perform direction estimation processing using cross-polarized waves. For example, when performing direction estimation processing when a normal circularly polarized antenna is used for transmission and a counterclockwise circularly polarized antenna is used for reception, the direction estimation unit 214 uses CA#7, CA#8, CA Four virtual reception antennas consisting of #15 and CA#16 are extracted, and direction estimation processing similar to the processing described above is performed.

For example, the direction estimation unit 214 is not limited to the direction estimation process using a virtual receiving array in which the combination of transmitting and receiving polarized antennas is the same, but also in the direction estimation process using a virtual receiving array having a different combination of transmitting and receiving polarized antennas. Estimation processing may also be performed. For example, the direction estimation unit 214 extracts all virtual reception antennas obtained from the output of the first Doppler multiplexing unit 213 consisting of CA#1 to CA#16, and performs direction estimation processing similar to the processing described above. It's okay. In this case, the direction estimation unit 214 performs direction estimation processing using a received signal that includes a combination of different types of polarization (for example, C/H, C/V, etc.), thereby eliminating polarization dependence. A low direction estimation result is obtained. Furthermore, since the direction estimation unit 214 performs direction estimation processing using all virtual reception antennas, the reception SNR is improved and the detection performance of the radar device 10 can be improved. Furthermore, since direction estimation processing is performed using the maximum aperture length of the available virtual reception antenna, angular resolution is also improved.

For example, the direction estimation unit 214 performs direction estimation processing using the same type of polarized antenna for both transmission and reception as described above, and direction estimation processing that combines different types of polarization, and uses the direction estimation results of both of these. It may also be a result of direction estimation processing. As a result, a direction estimation processing result with a high polarization dependence and a direction estimation processing result with a low polarization dependence can be obtained. The results of such direction estimation processing may be input to a processing unit for each object mark (not shown) to perform target object identification processing.

<Direction estimation processing for anti-circularly polarized waves>
For example, when performing direction estimation processing using counter-rotating circularly polarized antennas for both transmission and reception, the direction estimation unit 214 extracts four virtual reception antennas consisting of RA#7, RA#8, RA#15, and RA#16. , performs direction estimation processing.

The counter-rotating circularly polarized virtual reception array correlation vector h _c2 (f _{b_cfar} , f _{s_comp_cfar} ) is a column vector, and the first to N _BF ×Na elements included in this are the virtual antenna VA#( Represents the received signal from Nt×Na+1) to VA#(Nt+N _BF )×Na, or the received signal from RA#1 to RA#(N _BF ×Na). For example, the direction estimation unit 214 calculates the direction of the anti-circularly polarized virtual receiving array correlation vector h _c2 (f _{b_cfar} , f _{s_comp_cfar} ) corresponding to the received signals of RA#7, RA#8, RA#15, and RA#16. Direction of arrival is estimated using the partial array correlation vector obtained by extracting the 7th, 8th, 15th, and 16th elements. The extracted partial array correlation vector is expressed as h _c2sub (f _{b_cfar} , f _{s_comp_cfar} ). Further, the number of dimensions of h _c2sub (f _{b_cfar} , f _{s_comp_cfar} ) is assumed to be N _c2sub .

The direction estimation unit 214 calculates a spatial profile by making the azimuth direction θ in the direction estimation evaluation function value P _Hc2 (θ, f _{b_cfar} , f _{s_comp_cfar} ) variable within a defined angular range, for example. The direction estimation unit 214 extracts a predetermined number of maximum peaks of the calculated spatial profile in descending order of magnitude, and outputs the azimuth direction of the maximum peak as an estimated direction of arrival value (eg, positioning output).

Note that there are various methods for determining the direction estimation evaluation function value P _Hc2 (θ, f _{b_cfar} , f _{s_comp_cfar} ) depending on the direction of arrival estimation algorithm. For example, an estimation method using an array antenna disclosed in Non-Patent Document 3 may be used.

For example, the beamformer method can be expressed as the following equation (61). Other methods such as Capon and MUSIC are also applicable.

Further, a _c2sub (θ _u ) indicates the direction vector of the extracted partial virtual reception arrays RA#7, RA#8, RA#15, and RA#16 with respect to the arriving wave in the azimuth direction θ _u .

In addition, D _c2cal is the array correction coefficient that corrects the phase deviation and amplitude deviation between the antennas of the partial virtual reception arrays RA#7, RA#8, RA#15, and RA#16, and the inter-element coupling between the antennas. It is a matrix of order N _c2sub containing coefficients that reduce the influence. If the coupling between the antennas of the virtual receiving array can be ignored, D _c2cal becomes a diagonal matrix, and the diagonal components include array correction coefficients that correct the phase deviation and amplitude deviation between the transmitting array antennas and between the receiving array antennas. .

Further, for example, the direction estimation unit 214 may perform direction estimation processing using cross-polarized waves. For example, when performing direction estimation processing when a counter-rotating circularly polarized antenna is used for transmission and a right-rotating circularly polarized antenna is used for reception, the direction estimation unit 214 uses RA#5, RA#6, RA Four virtual receiving antennas consisting of #13 and RA #14 are extracted, and direction estimation processing similar to the processing described above is performed.

For example, the direction estimation unit 214 is not limited to the direction estimation process using a virtual receiving array in which the combination of transmitting and receiving polarized antennas is the same, but also in the direction estimation process using a virtual receiving array having a different combination of transmitting and receiving polarized antennas. Estimation processing may also be performed. For example, the direction estimation unit 214 extracts all virtual receiving antennas obtained from the output of the second Doppler demultiplexer 213 consisting of RA#1 to RA#16, and performs direction estimation processing similar to the processing described above. It's okay. In this case, the direction estimation unit 214 performs direction estimation processing using a received signal that includes a combination of different types of polarization (for example, R/H, R/V, etc.), thereby eliminating polarization dependence. A low direction estimation result is obtained. Furthermore, since the direction estimation unit 214 performs direction estimation processing using all virtual reception antennas, the reception SNR is improved and the detection performance of the radar device 10 can be improved. Furthermore, since direction estimation processing is performed using the maximum aperture length of the available virtual reception antenna, angular resolution is also improved.

For example, the direction estimation unit 214 performs direction estimation processing using the same type of polarized antenna for both transmission and reception as described above, and direction estimation processing that combines different types of polarization, and uses the direction estimation results of both of these. It may also be a result of direction estimation processing. As a result, a direction estimation processing result with a high polarization dependence and a direction estimation processing result with a low polarization dependence can be obtained. The results of such direction estimation processing may be input to a processing unit for each object mark (not shown) to perform target object identification processing.

An example of direction estimation processing for anti-circularly polarized waves has been described above.

Further, the direction estimation unit 214 may perform direction estimation processing using, for example, the outputs of the first and second Doppler multiplexing/demultiplexing units 213.

For example, when using a circularly polarized antenna (e.g., C/C) for both transmission and reception, and a counter-rotating circularly polarized antenna (e.g., R/R) for both transmission and reception, the direction estimation unit 214 performs first Doppler demultiplexing. Four virtual reception antennas consisting of CA#5, CA#6, CA#13, and CA#14 are extracted from the output of the second Doppler demultiplexer 213, and RA#7, RA#8, Four virtual receiving antennas consisting of RA#15 and RA#16 are extracted, and the extracted partial array correlation vectors are calculated using h _c1sub (f _{b_cfar} , f _{s_comp_cfar} ) and h _c2sub (f _{b_cfar} , f _{s_comp_cfar} ). direction estimation process.

The direction estimation unit 214 calculates a spatial profile by making the azimuth direction θ in the direction estimation evaluation function value P _Hc12 (θ, f _{b_cfar} , f _{s_comp_cfar} ) variable within a defined angular range, for example. The direction estimation unit 214 extracts a predetermined number of maximum peaks of the calculated spatial profile in descending order of magnitude, and outputs the azimuth direction of the maximum peak as an estimated direction of arrival value (eg, positioning output).

Note that there are various methods for determining the direction estimation evaluation function value P _Hc12 (θ, f _{b_cfar} , f _{s_comp_cfar} ) depending on the direction of arrival estimation algorithm. For example, an estimation method using an array antenna disclosed in Non-Patent Document 3 may be used.

For example, the beam former method can be expressed as the following equation (62). Other methods such as Capon and MUSIC are also applicable.

For example, when using a combination of circularly polarized antennas (for example, C/R and R/C) in which transmission and reception are cross-polarized waves, the direction estimation unit 214 calculates the CA Extract four virtual receiving antennas consisting of #7, CA#8, CA#15, and CA#16, and RA#5, RA#6, RA#13, RA# from the output of the second Doppler demultiplexer 213. _A similar _{direction estimation is} performed by extracting four _virtual receive antennas _consisting of Perform processing.

In this way, when the direction estimation unit 214 performs direction estimation processing using the outputs of the first and second Doppler multiplexing/demultiplexing units 213, more virtual antennas can be used, the reception SNR is improved, The target detection performance of the radar device 10 can be improved.

Further, the direction estimation unit 214 may perform the direction estimation process using the output of the encoded Doppler demultiplexer 212 and the output of either or both of the first and second Doppler demultiplexers 213. .

For example, the direction estimation unit 214 extracts all virtual receiving antennas obtained from the output of the encoded Doppler demultiplexer 212 consisting of VA#1 to #32, and selects the first Doppler antenna consisting of CA#1 to CA#16. All virtual reception antennas obtained from the output of the demultiplexer are extracted, and all virtual reception antennas obtained from the output of the second Doppler demultiplexer 213 consisting of RA#1 to RA#16 are extracted. Direction estimation processing is performed using the respective partial array correlation vectors h _sub (f _{b_cfar} , f _{s_comp_cfar} ), h _c1sub (f _{b_cfar} , f _{s_comp_cfar} ) and h _c2sub (f _{b_cfar} , f _{s_comp_cfar} ).

The direction estimation unit 214 calculates a spatial profile by making the azimuth direction θ in the direction estimation evaluation function value P _Hall (θ, f _{b_cfar} , f _{s_comp_cfar} ) variable within a defined angular range, for example. The direction estimation unit 214 extracts a predetermined number of maximum peaks of the calculated spatial profile in descending order of magnitude, and outputs the azimuth direction of the maximum peak as an estimated direction of arrival value (eg, positioning output).

Note that there are various methods for determining the direction estimation evaluation function value P _Hall (θ, f _{b_cfar} , f _{s_comp_cfar} ) depending on the direction of arrival estimation algorithm. For example, an estimation method using an array antenna disclosed in Non-Patent Document 3 may be used.

For example, the beamformer method can be expressed as in the following equation (63). Other methods such as Capon and MUSIC are also applicable.

In the above case, the direction estimation unit 214 can obtain a direction estimation result that is less dependent on polarization by performing direction estimation processing using a received signal that includes a combination of different types of polarization. Furthermore, since the direction estimation unit 214 performs direction estimation processing using all virtual reception antennas, the reception SNR is improved and the detection performance of the radar device 10 can be improved. Furthermore, since direction estimation processing is performed using the maximum aperture length of the available virtual reception antenna, angular resolution is also improved. More virtual antennas can be used, the reception SNR can be improved, and target detection performance can be improved.

For example, in order to configure 2 transmission MIMO for each of the four polarizations: horizontal polarization, vertical polarization, normal circular polarization, and anti-rotation circular polarization, the existing method requires 8 transmissions. Use an antenna. On the other hand, in this embodiment, configuring 2 transmission MIMO for each of the four polarizations can be achieved by using four transmission antennas 109, and the effect of reducing the number of transmission antennas can be obtained. . Furthermore, in this embodiment, code multiplexing transmission is performed on the Doppler multiplexed signal, so the transmission time can be shortened. For example, compared to the case where four transmitting antennas are switched in time division, the effect of halving the transmission time can also be obtained.

Further, the direction estimation unit 214 may perform direction estimation processing using a received signal that includes a combination of different types of polarization. In this case, a direction estimation result that is less dependent on polarization is obtained, and the reception SNR is improved by performing direction estimation processing using more virtual antennas, so that the detection performance of the radar device 10 can be improved. Furthermore, since direction estimation processing is performed using the maximum aperture length of the available virtual reception antenna, angular resolution is also improved. For example, when performing direction estimation processing using all virtual reception antennas, a maximum of (Nt × Na + 2 × N _BF × Na) virtual reception antennas can be used, and more than Nt × Na virtual reception antennas can be used. A virtual receiving antenna can be used.

Further, the direction estimation unit 214 performs, for example, direction estimation processing using the same type of polarized antenna for both transmission and reception as described above, and direction estimation processing using a combination of different types of polarization, and The estimation result may be the direction estimation processing result. As a result, a direction estimation processing result with a high polarization dependence and a direction estimation processing result with a low polarization dependence can be obtained. The results of such direction estimation processing may be input to a processing unit for each object mark (not shown) to perform target object identification processing.

For example, the direction estimation unit 214 includes the direction estimation result for each polarization combination, distance information based on the distance index f _{b_cfar} , and the target Doppler frequency determination result (in the encoded Doppler demultiplexer 212) as the positioning result. Doppler velocity information of the target based on the Doppler aliasing determination processing result) may be output.

Note that when, for example, equation (5) is used as the amount of phase rotation, the Doppler frequency information is calculated as shown in equation (64) using D _rmin , which is the result of Doppler aliasing determination processing in the encoded Doppler demultiplexer 212. Calculations can be made within the expanded range.

Furthermore, when using Equation (6) as the amount of phase rotation, for example, Doppler frequency information can be calculated in an expanded range as shown in Equation (65) below using D _rmin , which is the result of Doppler aliasing determination processing. Become.

Note that the Doppler frequency information may be converted into a relative velocity component and output. To convert the Doppler frequency index f _out into a relative velocity component v _d (f _out ) using D _rmin , which is the Doppler aliasing determination result of the target, the following equation (66) may be used. Here, λ is the wavelength of the carrier frequency of the RF signal output from the transmitter radio section (not shown) (when using a chirp signal, the wavelength at the center frequency of the chirp signal is used). Further, Δ _f is a Doppler frequency interval in FFT processing in the Doppler analysis unit 210. For example, in this embodiment, Δ _f =1/{N _code ×Loc×T _r }.

In addition, in the example of antenna arrangement mentioned above, the example of arrangement|positioning using four types of polarized-wave receiving antenna was shown, However, It is not limited to this, For example, you may use two types of polarized-wave receiving antenna. In this case, two types of polarization receiving antennas are included, for example, circularly polarized waves (right-handed polarized waves, left-handed polarized waves) or different linearly polarized waves (for example, horizontally or vertically polarized waves).

Alternatively, one type of polarized receiving antenna may be used. In this case, one type of polarization receiving antenna is included, for example, circularly polarized waves (right-handed polarized waves, left-handed polarized waves) or different linearly polarized waves (for example, horizontally or vertically polarized waves).

Hereinafter, examples of arrangement when the reception antenna 202 has two types of polarization and one type of polarization will be described.

[Antenna arrangement example 2]
FIG. 16 shows an example of antenna arrangement when the receiving antenna 202 has two types of polarization (for example, C and R).

As shown in FIG. 16(a), transmitting antennas Tx#1 to Tx#4 (for example, Nt transmitting antennas 109) and receiving antennas Rx#1 to Rx#8 (for example, Na receiving antennas 109), ), virtual receiving antenna (or MIMO virtual antenna) arrangements VA#1 to VA#32 as shown in FIG. 16(b) are constructed.

In addition, in (a) of FIG. 16, from the arrangement of right circularly polarized beam transmitting antennas Tx#5 and Tx#6 (for example, N _BF transmitting antennas) and receiving antennas Rx#1 to Rx#8, , arrangement CA#1 to CA#16 of right-rotating circularly polarized virtual reception antennas as shown in FIG. 16(c) are configured. Arrangements CA#1 to CA#16 of the right circularly polarized virtual receiving antennas are also expressed as VA#33 to VA#48, for example.

In addition, in (a) of FIG. 16, from the arrangement of anti-rotating circularly polarized beam transmitting antennas Tx#5 and Tx#6 (for example, N _BF transmitting antennas) and receiving antennas Rx#1 to Rx#8, , Arrangements RA#1 to RA#16 of counter-rotating circularly polarized virtual reception antennas are configured as shown in FIG. 16(d). Arrangements RA#1 to RA#16 of the counter-rotating circularly polarized virtual receiving antennas are also expressed as VA#33 to VA#48, for example. Arrangements RA#1 to RA#16 of the counter-rotating circularly polarized virtual receiving antennas are the same as the arrangements CA#1 to CA#16 of the right-rotating circularly polarized virtual receiving antennas, respectively.

Note that (b) to (d) in Fig. 16 include overlapping arrangements on the X axis (horizontal direction in Fig. 16), so in Fig. 16 (b) to (d), the illustrations are shifted in the vertical direction. However, each is placed one-dimensionally at a position on the X axis. Furthermore, since the combination of transmitting antenna polarization and receiving antenna polarization in each virtual receiving antenna is different for each antenna, it is written as (transmitting antenna polarization/receiving antenna polarization). For example, a virtual reception antenna for transmitting horizontally polarized waves and receiving right circularly polarized waves is written as "H/C."

Furthermore, in (a) of FIG. 16, as an example, the case where the receiving antenna 202 includes a normal circularly polarized wave antenna and a counterclockwise circularly polarized wave antenna is explained, but the types of polarized wave antennas included in the receiving antenna 202 are , and combinations are not limited to these, and other polarized antenna types and combinations may be applied.

[Antenna arrangement example 3]
FIG. 17 shows an example of antenna arrangement when the receiving antenna 202 has one type of polarization (for example, C).

As shown in FIG. 17(a), transmitting antennas Tx#1 to Tx#4 (for example, Nt transmitting antennas 109) and receiving antennas Rx#1 to Rx#8 (for example, Na receiving antennas 109), ), virtual receiving antenna (or MIMO virtual antenna) arrangements VA#1 to VA#32 as shown in FIG. 17(b) are constructed.

In addition, in (a) of FIG. 17, from the arrangement of right circularly polarized beam transmitting antennas Tx#5 and Tx#6 (for example, N _BF transmitting antennas) and receiving antennas Rx#1 to Rx#8, , arrangement CA#1 to CA#16 of right-rotating circularly polarized virtual reception antennas as shown in FIG. 17(c) are configured. Arrangements CA#1 to CA#16 of the right circularly polarized virtual receiving antennas are also expressed as VA#33 to VA#48, for example.

In addition, in (a) of FIG. 17, from the arrangement of anti-rotating circularly polarized beam transmitting antennas Tx#5 and Tx#6 (for example, N _BF transmitting antennas) and receiving antennas Rx#1 to Rx#8, , Arrangements RA#1 to RA#16 of counter-rotating circularly polarized virtual reception antennas are configured as shown in FIG. 17(d). Arrangements RA#1 to RA#16 of the counter-rotating circularly polarized virtual receiving antennas are also expressed as VA#33 to VA#48, for example. Arrangements RA#1 to RA#16 of the counter-rotating circularly polarized virtual receiving antennas are the same as the arrangements CA#1 to CA#16 of the right-rotating circularly polarized virtual receiving antennas, respectively.

Note that (b) in FIG. 17 includes overlapping arrangements on the X-axis (horizontal direction in FIG. 17), so although the illustrations are shifted in the vertical direction in (b) of FIG. 17, the respective positions on the X-axis are arranged one-dimensionally. Furthermore, since the combination of transmitting antenna polarization and receiving antenna polarization in each virtual receiving antenna is different for each antenna, it is written as (transmitting antenna polarization/receiving antenna polarization). For example, a virtual receiving antenna for horizontally polarized wave transmission and vertically polarized wave reception is written as "H/V."

Note that in FIG. 17(a), as an example, the case where the receiving antenna 202 includes a right circularly polarized antenna is described, but the type of polarized antenna included in the receiving antenna 202 is not limited to this. , other polarized antenna types may be applied.

Here, in the antenna arrangement examples 1 to 3 described above, examples have been described in which virtual receiving antennas whose transmitting and receiving antennas have the same polarization are arranged at uniform intervals D _H (for example, D _H =0.5 wavelength interval). However, the arrangement is not limited to such an arrangement, and the arrangement may be such that the interval D _H (for example, D _H =0.5 wavelength interval) is included in a part of the arrangement of virtual receiving antennas in which the polarization of the transmitting and receiving antennas is the same. Good too. An example of antenna arrangement having such an arrangement will be described below.

[Antenna arrangement example 4]
FIG. 18 shows an arrangement in which a part of the arrangement of virtual receiving antennas in which the polarization of the transmitting and receiving antennas is the same includes an interval D _H (for example, D _H =0.5 wavelength interval).

As shown in FIG. 18(a), transmitting antennas Tx#1 to Tx#4 (for example, Nt transmitting antennas 109) and receiving antennas Rx#1 to Rx#4 (for example, Na receiving antennas 109), ), virtual receiving antenna (or MIMO virtual antenna) arrangements VA#1 to VA#16 are constructed as shown in FIG. 18(b).

In addition, in (a) of FIG. 18, from the arrangement of right circularly polarized beam transmitting antennas Tx#5 and Tx#6 (for example, N _BF transmitting antennas) and receiving antennas Rx#1 to Rx#8, , arrangements CA#1 to CA#8 of right-rotating circularly polarized virtual receiving antennas are configured as shown in FIG. 18(c). Arrangements CA#1 to CA8# of the right circularly polarized virtual receiving antennas are also expressed as VA#17 to VA#24, for example.

In addition, in (a) of FIG. 18, from the arrangement of anti-rotating circularly polarized beam transmitting antennas Tx#5 and Tx#6 (for example, N _BF transmitting antennas) and receiving antennas Rx#1 to Rx#8, , Arrangements RA#1 to RA#8 of counter-rotating circularly polarized virtual reception antennas are configured as shown in FIG. 18(d). Arrangements RA#1 to RA#8 of the counter-rotating circularly polarized virtual receiving antennas are also expressed as VA#17 to VA#24, for example. Arrangements RA#1 to RA#8 of the anti-rotating circularly polarized virtual receiving antennas are the same as the arrangements CA#1 to CA#8 of the right-rotating circularly polarized virtual receiving antennas, respectively.

Note that (b) in FIG. 18 includes overlapping arrangements on the X-axis (horizontal direction in FIG. 18), so although the illustrations are shifted in the vertical direction in (b) of FIG. 18, the respective positions on the X-axis are arranged one-dimensionally. Furthermore, since the combination of transmitting antenna polarization and receiving antenna polarization in each virtual receiving antenna is different for each antenna, it is written as (transmitting antenna polarization/receiving antenna polarization). For example, a virtual receiving antenna for horizontally polarized wave transmission and vertically polarized wave reception is written as "H/V."

As shown in FIG. 18(a), the interval between beam transmitting antennas Tx#5 and Tx#6 is set to 2D _H (for example, 1 wavelength when D _H = 0.5 wavelength interval), and the receiving antennas Rx#1 to The interval of Rx#4 is set to an interval of 3D _H (for example, 1.5 wavelengths when D _H =0.5 wavelength interval). As a result, the regular circularly polarized virtual receiving antenna arrangement shown in FIG. 18(c) and the anti-rotating circularly polarized virtual receiving antenna arrangement shown in _{FIG .} A virtual receiving antenna arrangement including As a result, the direction estimation for normal circularly polarized waves and anti-rotating circularly polarized waves becomes a virtual receiving antenna arrangement including a spacing of 2D, which increases the aperture length and includes a spacing _DH , so that the angular resolution is improved and the grating lobe is This arrangement also makes it possible to suppress

In antenna arrangement example 4, beam transmitting antennas Tx#5 and Tx#6 transmit beams using a plurality of transmitting antennas 109, and therefore are spaced apart by about one wavelength or more. Therefore, with an antenna arrangement in which the difference between the spacing of the receiving antennas 202 with respect to the spacing of the beam transmitting antennas is the spacing _D A virtual receive antenna arrangement is obtained that includes spacing D _H and includes spacing wider than spacing D _H. As a result, in direction estimation for normal circularly polarized waves and anti-rotating circularly polarized waves, the aperture length is expanded and the interval _DH is included, so angular resolution can be improved and grating lobes can be suppressed.

In addition, in the above-mentioned arrangement example, the virtual reception antennas are arranged one-dimensionally in the horizontal direction, but the arrangement is not limited to this arrangement, and the virtual receiving antennas may be arranged on a two-dimensional plane including the horizontal and vertical directions. A transmitting/receiving antenna may also be used. By using such an arrangement, it is possible to obtain, for example, two-dimensional direction estimation results in the horizontal and vertical directions as a target direction estimation process.

Further, D _H is not limited to 0.5 wavelength, but may be set to, for example, about 0.45 wavelength to 0.8 wavelength (for example, any value in the range of 0.45λ to 0.8λ).

The antenna arrangement example has been described above.

As described above, in the present embodiment, the radar device 10 is configured to connect at least one set of adjacent transmitting antennas 109, a horizontally polarized antenna and a vertically polarized antenna, among the plurality of transmitting antennas 109 in each transmission cycle. Radar transmission signals given phase rotation amounts (for example, phase rotation amounts corresponding to orthogonal code sequences) whose phases differ by 90° or -90° are multiplexed from a plurality of transmitting antennas 109.

Thereby, the received signal for each transmission period corresponding to the signals transmitted from at least one set of adjacent transmitting antennas 109 is a linearly polarized wave (for example, horizontally polarized wave or vertically polarized wave) corresponding to each transmitting antenna 109. It can be regarded as a received signal corresponding to a radar transmission signal transmitted with a different circular polarization (for example, positive circular polarization or anti-rotating circular polarization). Therefore, the radar device 10 is capable of multiplex transmission using circularly polarized waves in addition to linearly polarized waves, for example, using a plurality of transmitting antennas 109 that are linearly polarized antennas.

Therefore, according to the present embodiment, even when transmitting four types of polarized waves, for example, left-handed circularly polarized waves and right-handed circularly polarized waves in addition to vertically polarized waves and horizontally polarized waves, the number of transmitting antennas 109 to be used can be increased. The detection accuracy of the target object in the radar device 10 can be improved by suppressing this.

Note that codes Code ₁ and Code ₂ used in this embodiment represent codes when the phase deviation between the transmitting antennas 109 is corrected in advance. Therefore, the phase difference at the feeding point of each transmitting antenna 109 is the phase difference between code elements of codes Code ₁ and Code ₂ assigned to each transmitting antenna 109. Here, when the radar device 10 transmits using Code ₁ = {OC ₁ (1), OC ₁ (2)} and Code ₂ = {OC ₂ (1), OC ₂ (2)}, each code The phase difference between elements is given by angle[OC ₂ (1)]-angle[OC ₁ (1)] and angle[OC ₂ (2)]-angle[OC ₁ (2)]. Therefore, in the present embodiment, the radar device 10 uses the same Doppler multiplexed signal from the two transmitting antennas 109 to generate an orthogonal code sequence of N _CM =2 with code length Loc = 2, for example, Code ₁ . ={1, 1} and Code ₂ ={j, -j}, the phase difference between the feeding points of the two transmitting antennas 109 is angle[OC ₂ (1)]-angle[OC ₁ (1)] = 90° and angle[OC ₂ (2)] - angle [OC ₁ (2)] = -90°, and the phase difference between the feeding points of the two transmitting antennas 109 for each transmission period. is 90° or -90°.

Similarly, in the present embodiment, the radar device 10 uses the same Doppler multiplexed signal from two transmitting antennas 109 to generate an orthogonal code sequence of N _CM =2 with code length Loc = 2, for example, Code When transmitting using ₁ = {A, B} and Code ₂ = {-j×A, j×B}, the phase difference between the feeding points of the two transmitting antennas 109 is angle[OC ₂ (1) ]-angle[OC ₁ (1)] = -90° and angle[OC ₂ (2)]-angle[OC ₁ (2)] = 90°, and two transmitting antennas 109 are fed every transmission period. The phase difference between the points is 90° or -90°.

Similarly, in the present embodiment, the radar device 10 uses the same Doppler multiplexed signal from two transmitting antennas 109 to generate an orthogonal code sequence of N _CM =2 with code length Loc = 2, for example, Code When transmitting using ₁ = {A, B}, Code ₂ = {exp(jξ)×A, - exp(jξ)×B}, the phase difference between the feeding points of the two transmitting antennas 109 is angle [OC ₂ (1)]-angle[OC ₁ (1)]=ξ and angle[OC ₂ (2)]-angle[OC ₁ (2)]=-ξ, and two transmissions are performed in each transmission period. The phase difference between the feeding points of the antenna is ξ or -ξ. Here, for example, ξ may be in the range of π/6 to 5π/6 radians (=30° to 150°). The same applies to Modification 1 and Modification 2 below.

(Modification 1 of Embodiment 1)
In the first embodiment, the number N _DM of Doppler multiplexing may be set to 1, and MIMO multiplex transmission may be performed using code multiplexing without using Doppler multiplexing.

FIG. 19 shows a configuration example of a radar device 10a according to a first modification of the first embodiment. Modification 1 of Embodiment 1 has a configuration that does not use Doppler multiplexing, so the configuration related to Doppler multiplexing transmission and reception operations (for example, Doppler shift setting unit 106, code The configuration includes a code demultiplexer 215 in place of the encoded Doppler demultiplexer 212.

Further, in the example of FIG. 19, in the radar device 10a, the Doppler multiplexing number N _DM is set to 1, and code multiplexing is performed using a code with a code length of 2. Therefore, in FIG. 19, the configuration is such that the number of transmitting antennas is Nt=2. The transmitting antenna 109 may perform code multiplex transmission as in the first embodiment, for example, by using different linearly polarized antennas (eg, horizontally polarized antennas and vertically polarized antennas). As a result, in addition to receiving signals from different linearly polarized antennas (for example, horizontally polarized antennas and vertically polarized antennas), the radar device 10a receives signals from normal circularly polarized waves and counterclockwise circularly polarized waves when transmitting a beam during code multiplexing. This makes it possible to transmit signals with positive circular polarization and anti-circular polarization.

In this case, the radar device 10a has beam transmitting antenna configurations of different linearly polarized antennas (for example, horizontally polarized antennas and vertically polarized antennas), normal circularly polarized waves, and anti-rotating circularly polarized waves, and each polarized wave is When a plurality of Na receiving antennas 202 are provided, a SIMO (Single in multiple output) configuration is obtained in which 1×Na virtual receiving antennas are obtained for each polarized wave.

Hereinafter, an example of the operation of each component of the radar device 10a shown in FIG. 19 will be described.

In the radar device 10a shown in FIG. 19, the operation of the radar transmitter 100a is as follows: Doppler multiplex number N _DM =1, phase rotation amount φ ₁ that provides the first Doppler shift amount DOP ₁ =0, N _{DOP_CODE} (1) =N _CM =2, and the number of transmitting antennas Nt = 2. Since the operation is similar to that of the first embodiment, a description of the operation will be omitted.

Next, an example of the operation of the radar receiving section 200a of the radar device 10a shown in FIG. 19 will be described.

The operation of each component from the receiving antenna 202 to the antenna system processing section 201 is the same as that in the first embodiment, and therefore the description of the operation will be omitted.

Since Doppler multiplex transmission is not performed (for example, Doppler multiplex number N _DM =1), the CFAR unit 211 performs an operation when Doppler region compression CFAR processing is not used.

The code demultiplexer 215 uses the distance index f _{b_cfar} which is the output of the CFAR unit 211 (output when Doppler region compression CFAR processing is not used), the Doppler frequency index f _{s_cfar} , and the output of the Doppler analysis unit 210, Separate the code multiplexed signals. Code multiplexing/demultiplexing section 215 outputs the separated received signal of the code multiplexed signal to direction estimation section 214 .

For example, the code demultiplexer 215 performs demultiplexing and reception of the code multiplexed signal by multiplying by the complex conjugate of the code used for multiplex transmission, as shown in the following equation (67). The separated received signal Y _z (f _{b_cfar} , f _{s_cfar} , ncm) of the code multiplexed signal is output to the direction estimation section 214 .

Here, Y _{z (} f _{b_cfar ,} f _{s_cfar} , ncm) is VFTALL _z (f _{b_cfar} , f _{s_cfar} ), the code multiplexed signal is separated using the orthogonal code Code _ncm during transmission. Note that z=1,~,Na, and ncm=1,~,N _CM .

Here, the Doppler phase correction vector α(f _{s_cfar} ) is expressed as in the following equation (68). For example, the Doppler phase correction vector α (f _{s_cfar} ) shown in equation (68) is based on the Doppler analysis time of the output VFT _z ¹ (f _{b_cfar} , f _{s_comp_cfar} ) of the first Doppler analysis unit 210, and This is a vector whose elements are Doppler phase correction coefficients that correct the phase rotation in the Doppler component of the Doppler frequency index f _{s_cfar} caused by the time delay of the Tr in the output VFT _z ² (f _{b_cfar} , f _{s_comp_cfar} ) of the Doppler analysis unit 210 of .

Further, VFTALL _z (f _{b_cfar} , f _{s_cfar} ) shown in equation (67) is the output VFT of the two Doppler analysis units 210 in the z-th antenna system processing unit 201, for example, as shown in the following equation (69). Of _z ^noc (f _b , f _s ), it is a value expressed in a vector format as a component extracted based on the distance index f _{b_cfar} and the Doppler frequency index f _{s_cfar} extracted by the CFAR unit 211. However, noc=1,2.

In FIG. 19, the direction estimation unit 214 calculates the distance index f _{b_cfar} input from the code demultiplexing unit 215 based on the separated received signal Y _z (f _{b_cfar} , f _{s_cfar} , ncm) of the code multiplexed signal with respect to the Doppler frequency index f _{s_cfar} . Then, target direction estimation processing (hereinafter referred to as direction estimation processing for linearly polarized waves) is performed.

Further, the direction estimation unit 214 performs a direction estimation process of the target (hereinafter, direction estimation for normal circular polarization) based on the output from the first Doppler analysis unit 210 (Doppler analysis unit 210-1 in FIG. 19). processing).

The direction estimation unit 214 also performs direction estimation processing of the target (hereinafter, direction estimation for anti-circular polarization) based on the output from the second Doppler analysis unit 210 (Doppler analysis unit 210-2 in FIG. 19). processing).

The direction estimation process in the direction estimation unit 214 may include, for example, a direction estimation process for linearly polarized waves, a direction estimation process for normal circularly polarized waves, and a direction estimation process for anticircularly polarized waves. Each direction estimation process will be explained below.

<Direction estimation processing for linearly polarized waves>
For example, the direction estimation unit 214 generates a virtual receiving array correlation vector h (f _{b_cfar} , f _{s_cfar} ) as shown in the following equation (70) based on the output of the code demultiplexing unit 215 and performs direction estimation processing. .

The virtual receiving array correlation vector h (f _{b_cfar} , f _{s_cfar} ) includes Nt×Na elements that are the product of the number Nt of transmitting antennas and the number Na of receiving antennas.

The direction estimation unit 214 uses the virtual receiving array correlation vector h (f _{b_cfar} , f _{s_cfar} ) to perform direction estimation processing on the reflected wave signal from the target based on the phase difference between each receiving antenna 202 .

Here, the virtual receiver array correlation vector h(f _{b_cfar} , f _{s_cfar} ) includes reflected wave reception signals of signals transmitted from different linearly polarized antennas (eg, a vertically polarized antenna and a horizontally polarized antenna). Therefore, the direction estimation unit 214 extracts an element that is a combination of the polarization of the predetermined transmitting antenna 109 and the polarization of the receiving antenna 202 from the virtual receiving array correlation vector h (f _{b_cfar} , f _{s_cfar} ). The direction estimation process may be performed using a virtual receiving array correlation vector consisting of elements. This makes it possible to obtain direction estimation results for each polarization of the predetermined transmitting antenna 109 and receiving antenna 202.

<Direction estimation processing for normal circular polarization>
The direction estimation unit 214 generates a normal circular polarization virtual reception array correlation vector h _c1 (f _{b_cfar} , f _{s_cfar} ) as shown in the following equation (71) based on the output of the first Doppler analysis unit 210. , performs direction estimation processing for the target's normal circularly polarized wave.

Here, the normal circularly polarized virtual reception array correlation vector h _c1 (f _{b_cfar} , f _{s_cfar} ) is a code based on the output of the first Doppler analysis unit 210 (for example, VFT _z ¹ (f _{b_cfar} , f _{s_cfar} ) It is multiplexed and beam-transmitted by adjacent transmitting antennas 109 with different linear polarizations, and includes a reflected wave received signal of a circularly polarized signal.For example, if there is one beam transmitting antenna, a right circularly polarized wave is transmitted. The virtual reception _array correlation vector _{h c1} (f _{b_cfar} , f _{s_cfar} ) includes ₁ ×Na elements. is used for processing to estimate the direction of the reflected wave signal from the target based on the phase difference between each receiving antenna 202.

<Direction estimation processing for anti-circularly polarized waves>
The direction estimation unit 214 performs direction estimation processing for the anti-circularly polarized wave of the target based on the output of the second Doppler analysis unit 210. The direction estimation unit 214 generates, for example, a counter-circularly polarized wave virtual receiving array correlation vector h _c2 (f _{b_cfar} , f _{s_cfar} ) as shown in the following equation (72), and estimates the direction for the target's counter-circularly polarized wave. Perform processing.

Here, the counter-rotating circularly polarized virtual receiving array correlation vector h _c2 (f _{b_cfar} , f _{s_cfar} ) is based on the output of the second Doppler analysis unit 210 (e.g., VFT _z ² (f _{b_cfar} , f _{s_ cfar} ), Code-multiplexed transmission, beam transmission by adjacent transmitting antennas 109 of different linear polarizations, and a reflected wave reception signal of a circularly polarized signal is included.For example, if there is one beam transmission antenna, anti-rotation virtual reception is performed. The array correlation vector h _c2 (f _{b_cfar} , f _{s_cfar} ) includes 1×Na elements.

The direction estimation unit 214 uses the counter-rotating circularly polarized virtual receiving array correlation vector h _c2 (f _{b_cfar} , f _{s_cfar} ) to estimate the direction of the reflected wave signal from the target based on the phase difference between each receiving antenna 202. I do.

Hereinafter, the direction estimation process for linearly polarized waves using the virtual reception array correlation vector h (f _{b_cfar} , f _{s_cfar} ) in the direction estimation unit 214, and the direction estimation process for the linearly polarized wave using the virtual reception array correlation vector h _c1 (f _{b_cfar} , f _{s_cfar} ) ) and the direction estimation process for anti-circular polarization using the anti-circular polarization virtual reception array correlation vector h _c2 (f _{b_cfar} , f _{s_cfar} ). Since this is the same as in Embodiment 1, a description of its operation will be omitted.

Note that the direction estimation unit 214 may perform direction estimation processing using a received signal that includes a combination of different types of polarization. In this case, a direction estimation result that is less dependent on polarization can be obtained. Furthermore, by performing direction estimation processing using more virtual antennas, the reception SNR can be improved and the detection performance of the radar device 10a can be improved. Furthermore, since direction estimation processing is performed using the maximum aperture length of the available virtual reception antenna, angular resolution is also improved. For example, when performing direction estimation processing using all virtual receiving antennas, here, since Nt=2,N _BF =1, the maximum number of virtual receiving antennas is (Nt×Na+2×N _BF ×Na)=4Na. can be used, and more than 2Na (=Nt×Na) virtual reception antennas can be used.

The direction estimation unit 214 also performs direction estimation processing using the same type of polarized antenna for both transmission and reception as described above, and direction estimation processing that combines different types of polarization for transmission and reception. Both direction estimation results may be used as the direction estimation processing results. As a result, a direction estimation processing result with a high polarization dependence and a direction estimation processing result with a low polarization dependence can be obtained. The results of such direction estimation processing may be input to a processing unit for each object mark (not shown) to perform target object identification processing.

Here, to configure one transmission SIMO for each of the four types of polarization (different linearly polarized antennas (e.g., vertically polarized antennas and horizontally polarized antennas) and normal/anti-rotating circularly polarized waves), , existing methods use four transmit antennas. On the other hand, in the configuration of FIG. 19, configuring one transmission SIMO for each of the four polarized waves can be achieved by using two transmission antennas 109, and the effect of reducing the number of transmission antennas can be obtained. . Furthermore, since the radar device 10a performs code multiplex transmission, the transmission time can be shortened. For example, compared to the case where four transmitting antennas 109 are switched in a time-division manner, the effect of halving the transmission time can also be obtained.

Furthermore, although the number of transmitting antennas is smaller than in the first embodiment, the same effects as in the first embodiment can be obtained. A specific example of antenna arrangement will be shown below.

[Antenna arrangement example 5]
FIG. 20 shows an example of antenna arrangement when the receiving antenna 202 has one type of polarization (for example, V). As shown in FIG. 20(a), transmitting antennas Tx#1 to Tx#2 (for example, Nt transmitting antennas 109) and receiving antennas Rx#1 to Rx#3 (for example, Na receiving antennas 109), ), virtual receiving antenna (or MIMO virtual antenna) arrangements VA#1 to VA#6 as shown in FIG. 20(b) are constructed. In addition, in (a) of FIG. 20, from the arrangement of the right circularly polarized beam transmitting antenna Tx#3 (for example, N _BF transmitting antennas) and the receiving antennas Rx#1 to Rx#3, Arrangements CA#1 to CA#3 of right-rotating circularly polarized virtual receiving antennas are configured as shown in (c). Arrangements CA#1 to CA#3 of the right circularly polarized virtual receiving antennas are also expressed as VA#7 to VA#9, for example.

In addition, in (a) of FIG. 20, from the arrangement of the anti-rotating circularly polarized beam transmitting antenna Tx#3 (for example, N _BF transmitting antennas) and the receiving antennas Rx#1 to Rx#3, Arrangements RA#1 to RA#3 of counter-rotating circularly polarized virtual receiving antennas are configured as shown in d). Arrangements RA#1 to RA#3 of the counter-rotating circularly polarized virtual receiving antennas are also expressed as VA#7 to VA#9. Arrangements RA#1 to RA#3 of the anti-rotating circularly polarized virtual receiving antennas are the same as the arrangements CA#1 to CA#3 of the right-rotating circularly polarized virtual receiving antennas, respectively.

Furthermore, since the combination of transmitting antenna polarization and receiving antenna polarization in each virtual receiving antenna is different for each antenna, it is written as (transmitting antenna polarization/receiving antenna polarization). For example, a virtual receiving antenna for horizontally polarized wave transmission and vertically polarized wave reception is written as "H/V."

Note that in FIG. 20(a), as an example, the case where the receiving antenna 202 includes a vertically polarized antenna has been described, but the type of polarized antenna included in the receiving antenna 202 is not limited to this, and other types may be used. types of polarized antennas may be applied. For example, a horizontally polarized antenna or a circularly polarized antenna may be applied. Alternatively, multiple types of polarized antennas may be applied to any of Rx#1 to Rx#3.

Here, in the arrangement shown in (a) of FIG. 20, the receiving antenna Rx#1 _is , Rx#3, Rx#2 (adjacent receiving antennas 202) are arranged at an antenna interval of 2D _H in the X-axis direction (for example, horizontal direction), and one of the receiving antennas Rx#1, Rx#3, Rx#2 The antennas of the antennas are arranged offset in the Y-axis direction (eg, vertically) by an antenna spacing _Dv with respect to the other antennas. With this arrangement, the following effects can be obtained.

For example, as shown in FIG. 20(a), the receiving antenna 202 is arranged on the Y axis orthogonal to the receiving antennas Rx#1 and Rx#2 that are arranged linearly in the X axis direction (for example, horizontal direction). Since Rx#3 is arranged offset by the interval D _V in the direction (e.g., vertical direction), two-dimensional direction estimation in the X-axis direction and Y-axis direction (e.g., vertical and horizontal directions) is possible. . For example, by setting the _DV at half-wavelength intervals, it is possible to suppress the generation of grating lobes in angle measurement processing in the range of ±90° in the vertical direction.

Note that _DV is not limited to half-wavelength intervals, and may be set at intervals shorter than the wavelength (λ) of the radar transmission signal. For example, _DV may be set to approximately 0.45λ to 0.8λ (for example, any value in the range of 0.45λ to 0.8λ). Note that λ represents the wavelength of the carrier frequency of the radar transmission signal. For example, when using a chirp signal as a radar transmission signal, λ is the wavelength of the center frequency in the frequency sweep band of the chirp signal.

Furthermore, as shown in FIG. 20(a), the transmitting antennas Tx#1 and Tx#2 are lined up in the X-axis direction with an antenna spacing D _H , so the virtual receiving antenna shown in FIG. 20(b) is As shown in the arrangement, the antenna spacing between VA#1 and VA#4, VA#2 and VA#5, and VA#3 and VA#6 is _DH .

Furthermore, as shown in FIG. 20(a), the receiving antennas Rx#1, Rx#3, and Rx#2 are arranged in the X-axis direction (for example, in the horizontal direction) with an antenna spacing of 2D _H , so that the transmitting antenna The difference in the spacing between the receiving antennas 202 with respect to the spacing between Tx#1 and Tx#2 (for example, D _H ) is the spacing D _H , and the virtual receiving antenna arrangement includes the spacing D _H . For example, as shown in the virtual receiving antenna arrangement shown in FIG. _H (however, VA#3 and VA#6 are arranged offset in the Y-axis direction (for example, vertical direction) by an interval D _V ).

For example, by setting D _H at half-wavelength intervals, it is possible to suppress the generation of grating lobes in angle measurement processing in the horizontal direction ±90° range. Furthermore, since the antennas are arranged at equal antenna intervals D _H , side lobes can be suppressed in angle measurement processing, and the detection performance of multiple targets can be improved.

Note that D _H is not limited to half-wavelength intervals, and may be set at intervals shorter than the wavelength (λ) of the radar transmission signal. For example, D _H may be set to approximately 0.45λ to 0.8λ (for example, any value in the range of 0.45λ to 0.8λ). Note that λ represents the wavelength of the carrier frequency of the radar transmission signal. For example, when using a chirp signal as a radar transmission signal, λ is the wavelength of the center frequency in the frequency sweep band of the chirp signal. The same applies to the following arrangement examples.

In addition, in the direction estimation process for the linearly polarized wave by the direction estimation unit 214, the radar device 10a extracts an element that is a combination of the polarized wave of a predetermined transmitting antenna 109 and the polarized wave of the receiving antenna 202, and from the extracted elements. When performing direction estimation processing using a virtual receiving array correlation vector, for example, when combining a horizontally polarized transmitting antenna and a vertically polarized antenna as a cross-polarized combination, virtual receiving antenna VA#1 , VA#2 and VA3 are used to perform the direction estimation process, and a direction estimation result that depends on the transmission and reception characteristics between the cross-polarized antennas can be obtained.

For example, when using a combination of a vertically polarized transmitting antenna and a vertically polarized antenna as a combination of the same type of polarization, the radar device 10a uses virtual receiving antennas VA#4, VA#5, and VA#6. By using this method, direction estimation processing can be performed, and a direction estimation result that depends on the transmission and reception characteristics of the same type of polarized waves can be obtained. By using the direction estimation results of the former combination of transmitting antennas and receiving antennas and the latter combination, it is possible to improve detection and identification performance.

Note that the radar device 10a extracts elements that are a combination of the polarization of the predetermined transmitting antenna 109 and the polarization of the receiving antenna 202, and performs direction estimation processing using a virtual receiving array correlation vector consisting of the extracted elements. When doing this, if the antenna spacing D _H is half a wavelength, the element spacing in the X-axis direction (for example, horizontal direction) is the antenna spacing 2D _H , which is one wavelength. Lobes occur.

For this reason, if the detection angle range assumed by the radar device 10a is wider than the angle at which the grating lobe occurs, the radar device may mistakenly detect a false peak due to the grating lobe within the detection angle range ( The probability of detection as a target object increases, and the detection performance of the radar device 10a may deteriorate.

However, the direction estimation unit 214 can suppress grating lobes by performing direction estimation processing using a received signal that includes a combination of different types of polarization. In this case, the direction estimation unit 214 performs direction estimation processing using, for example, virtual receiving antennas VA#1 to VA#6 as received signals that include combinations of different types of polarized waves. As a result, the element spacing in the X-axis direction (for example, horizontal direction) is arranged to include the antenna spacing D _H , and grating lobes can be suppressed in the angle measurement process in the horizontal direction ±90° range.

Furthermore, since the direction estimation unit 214 performs direction estimation processing using the maximum aperture length of the available virtual reception antenna, the angular resolution is also improved. Furthermore, since the direction estimation unit 214 performs direction estimation processing using all available virtual reception antennas, the reception SNR is also improved.

In addition, the direction estimation unit 214 performs direction estimation processing using CA#1, CA#2, and CA#3 in the direction estimation processing for normal circularly polarized waves. Similarly, the direction estimation unit 214 performs direction estimation processing using RA#1, RA#2, and RA#3 in the direction estimation processing for anti-rotating circularly polarized waves. As a result, it is possible to use a combination of a circularly polarized wave transmitting antenna with normal or counter-rotating rotation and a vertically polarized antenna, and it is possible to obtain direction estimation results that depend on the transmission and reception characteristics of polarized waves, which improves detection and identification performance. You can improve your performance.

In addition, in the direction estimation process using CA#1, CA#2, CA#3 or the direction estimation process using RA#1, RA#2, RA#3, if the antenna spacing D _H is a half wavelength, Since the element spacing in the X-axis direction (for example, horizontal direction) is the antenna spacing 2D _H , which is one wavelength, a grating lobe will occur in the angle measurement process in the horizontal direction ±90° range, but the direction estimation unit 214 , it is also possible to suppress grating lobes by performing direction estimation processing using received signals that include combinations of different types of polarization.

In this case, the direction estimating unit 214 uses the virtual receiving antennas VA#1 to VA#6, CA#1, CA#2, CA#3, RA#1 as received signals including combinations of different types of polarization, for example. , RA#2 and RA#3 are used to perform direction estimation processing. As a result, the element spacing in the X-axis direction (for example, horizontal direction) is arranged to include the antenna spacing D _H , and grating lobes can be suppressed in the angle measurement process in the horizontal direction ±90° range. Furthermore, since the direction estimation unit 214 performs direction estimation processing using the maximum aperture length of the available virtual reception antenna, the angular resolution is also improved. Furthermore, since the direction estimation process is performed using all available virtual reception antennas, the reception SNR is also improved.

(Modification 2 of Embodiment 1)
The configuration of FIG. 19 in Modification 1 of Embodiment 1 has the number of transmitting antennas Nt=2, and four types of polarization (different linearly polarized antennas (e.g., vertically polarized antennas and horizontally polarized antennas) and normal rotation antennas). Although the configuration has been shown in which one transmission SIMO is performed for each of the two types of polarized waves (/reverse circularly polarized waves), four types of polarized waves can be transmitted by combining time division multiplexing and switching the code multiplexing transmitting antenna 109 in a time division manner. It is also possible to have a MIMO configuration for each of them.

For example, FIG. 21 shows a configuration example of a radar device 10b according to a second modification of the first embodiment. In the radar device 10b, for example, the number of transmitting antennas Nt = 4, the pair of two code-multiplexed transmitting antennas 109 is switched in a time division manner, and 2 transmitting MIMO is performed for each of the four types of polarized waves. A configuration example is shown.

In the radar device 10b shown in FIG. 21, a transmission switching control section 110 and a transmission switching section 111 are added to the configuration of the radar device 10a shown in FIG.

Hereinafter, the operation of the portions of the radar device 10b shown in FIG. 21 that differ from the configuration of the radar device 10a shown in FIG. 19 will be mainly described.

The operations of the radar transmission signal generation section 101, phase rotation amount setting section 105, and phase rotation section 108 in the radar transmission section 100b of the radar apparatus 10b shown in _FIG . The operation is the same as that in Embodiment ₁ when the phase rotation amount φ ₁ =0, N _{DOP_CODE} (1)=N _CM =2, and the number of transmitting antennas Nt = 2, giving the shift amount DOP 1. The explanation of is omitted.

The transmission switching control unit 110 performs control to switch between the two transmission antennas 109 output from the transmission switching unit 111 at every predetermined transmission cycle. For example, as shown in FIG. 22, the transmission switching control unit 110 transmits two signals output from the transmission switching unit 111 every two transmission cycles, which is a transmission cycle of code length Loc=2 given by the encoding unit 107. Control is performed to switch the transmitting antenna 109. Note that FIG. 22 shows an example in which codes Code ₁ =[1 1] and Code ₂ =[j - j] are assigned in the encoding unit 107.

Here, the two transmitting antennas 109 serving as the unit of switching are a pair of transmitting antennas 109 that are arranged adjacent to each other and have different linearly polarized waves (for example, horizontally polarized waves and vertically polarized waves). The transmission switching control unit 110 performs an operation of switching these pairs every two transmission cycles.

Hereinafter, the number of pairs of adjacently arranged transmitting antennas 109 that provide different linearly polarized waves will be expressed as "Nsw". In addition, among the Nt transmitting antennas 109, the pair of transmitting antennas 109 that transmit first is called a "first transmitting pair", and the pair of transmitting antennas 109 that are switched in the next two transmission cycles is called a "second transmitting pair". It is called. Hereinafter, when Nt/2 transmission pairs are included for Nt transmission antennas 109, each transmission pair will be expressed as a transmission pair index "nsw". Here, nsw=1,~,Nt/2. Furthermore, the transmission switching control unit 110 outputs the index nsw of the transmission pair to the output switching unit 209 for each transmission cycle.

In addition, in FIGS. 21 and 22, among the nsw-th transmission pair, the transmitting antenna 109 that radiates the output of the phase rotation unit PROT#[1, 1] into space is referred to as "transmitting antenna Tx#[1, nsw]". It is written as. In addition, the transmitting antenna 109 that radiates the output of the phase rotation unit PROT#[ndop_code(1), 1]= PROT#[2, 1] into space is called "transmitting antenna Tx#[ndop_code(ndm), nsw]= Tx# [2, nsw]”. In FIGS. 21 and 22, Nt=4, and for the four transmitting antennas 109, Tx#[1, 1], Tx#[ndop_code(1), 1](=Tx#[2, 1] ), Tx#[1, 2], Tx#[ndop_code(1), 2] (=Tx#[2, 2]) are assigned. Also, ndop_code(1)=2.

Based on the control by the transmission switching control section 110 described above, the transmission switching section 111 performs an operation of pairing adjacently arranged transmitting antennas with different linear polarizations and switching these pairs every two transmission periods. For example, as shown in FIG. 22, the radar device 10b time-division multiplex transmits radar transmission signals from each of a plurality of pairs of transmitting antennas 109.

Next, an example of the operation of the radar receiving section 200b of the radar apparatus 10b shown in FIG. 21 will be described.

The operation of each component from the receiving antenna 202 to the antenna system processing section 201 is the same as that in the first embodiment, and therefore the description of the operation will be omitted.

The output switching unit 209 performs an output switching process every transmission period based on the orthogonal code element index OC_INDEX input from the encoding unit 107 of the phase rotation amount setting unit 105 and the transmission pair index nsw input from the transmission switching control unit 110. The output of the beat frequency analysis unit 208 of Switch and output. For example, the output switching unit 209 selects the Doppler analysis unit 210 obtained by the following equation (73) in the m-th transmission period Tr. Here, Loc=2.

The signal processing unit 206 has Loc×Nsw Doppler analysis units 210-1 to 210-Loc×Nsw. For example, the output switching unit 209 inputs data to the ncsw-th Doppler analysis unit 210 every Loc×Nsw transmission cycles (Loc×Nsw×Tr). Here, ncsw=1,~,Loc×Nsw. Therefore, the ncsw-th Doppler analysis unit 210 uses data of Ncsub=Nc/(Loc×Nsw) transmission cycles among the Nc transmission cycles (for example, the beat frequency input from the beat frequency analysis unit 208). Doppler analysis is performed for each distance index f _b using the response RFT _z (f _b , m)). Note that Nc is set to an integral multiple of (Loc×Nsw).

For example, if Ncsub is a power of 2, FFT processing can be applied in Doppler analysis. In this case, the FFT size is Ncsub, and the maximum Doppler frequency at which aliasing does not occur, which is derived from the sampling theorem, is ±1/(2 Loc×Nsw×Tr). Also, the Doppler frequency interval of the Doppler frequency index f _s is 1/(Loc × Nsw × Ncsub × Tr), and the range of the Doppler frequency index f _s is f _s = -Ncsub/2, ~, 0, ~, Ncsub/ It is 2-1. ncsw=1,～, (Loc×Nsw)

For example, the output VFT _z ^ncsw (f _b , f _s ) of the Doppler analysis unit 210 of the z-th signal processing unit 206 is obtained by replacing Ncode with Ncsub and replacing Loc with (Loc×Nsw) in equation (39). , noc is replaced with ncsw, and other than these, the operation is the same as in the first embodiment.

In FIG. 21, the CFAR unit 211 performs CFAR processing (for example, adaptive threshold determination) using the outputs of the (Loc×Nsw) Doppler analysis units 210 of the first to Nath signal processing units 206. and extracts the distance index f _{b_cfar} and Doppler frequency index f _{s_cfar} that give the peak signal.

The CFAR unit 211 calculates the output VFT _z ¹ (f _b , f _s ) ~ VFT _z ^Nsw×Loc (f _b , f _s ) is added using PowerFT (f _b , f _s ) to generate a two-dimensional CFAR consisting of a distance axis and a Doppler frequency axis (corresponding to relative velocity). processing or CFAR processing that is a combination of one-dimensional CFAR processing.

The CFAR unit 211 adaptively sets a threshold, and sends the distance index f _{b_cfar} , Doppler frequency index f _{s_cfar} , and received power information PowerFT(f _{b_cfar} , f _{s_cfar} ), which result in received power greater than the threshold, to the code demultiplexer. 215. Note that since Doppler multiplex transmission is not performed (Doppler multiplex number N _DM =1), the CFAR unit 211 performs an operation when Doppler region compression CFAR processing is not used.

The code demultiplexer 215 uses the distance index f _{b_cfar} which is the output of the CFAR unit 211 (output when Doppler region compression CFAR processing is not used), the Doppler frequency index f _{s_cfar} , and the output of the Doppler analysis unit 210, Separate the code multiplexed signals. Code multiplexing/demultiplexing section 215 outputs the separated received signal of the code multiplexed signal to direction estimation section 214 .

For example, the code demultiplexer 215 performs demultiplexing and reception of the code multiplexed signal by multiplying by the complex conjugate of the code used for multiplex transmission, as shown in the following equation (75). The separated received signal Y _z (f _{b_cfar} , f _{s_cfar} , ncm, nsw) of the code multiplexed signal is output to the direction estimation section 214 .

Here, Y _z (f _{b_cfar} , f _{s_cfar} , ncm, nsw) is the result of the Doppler analysis unit 210 in the z-th antenna system processing unit 201 when transmitted by the transmitting antenna 109 that is the nsw-th transmission pair. This is an output obtained by separating a code multiplexed signal using an orthogonal code Code _ncm at the time of transmission with respect to VFTALL _z (f _{b_cfar} , f _{s_cfar} , nsw) which is the output of the distance index f _{b_cfar} and the Doppler frequency index f _{s_cfar} . Note that z=1,~,Na, ncm=1,~,N _CM , and nsw=1,~,N _SW .

Here, the Doppler phase correction vector α(f _{s_cfar} ) is expressed as in the following equation (76). For example, the Doppler phase correction vector α (f _{s_cfar} ) shown in equation (76) is based on the Doppler analysis time of the output VFT _z ¹ (f _{b_cfar} , f _{s_cfar} ) of the first Doppler analysis unit 210, and The phase rotation in the Doppler component of the Doppler frequency index f _{s_cfar} caused by the transmission time delay (ncsw-1) × Tr / (Loc × N _SW ) in the output VFT _z ^ncsw (f _{b_cfar} , f _{s_cfar} ) of the Doppler analysis unit 210 of This is a vector whose elements are Doppler phase correction coefficients to be corrected.

In addition, in equation (75), VFTALL _z (f _{b_cfar} , f _{s_cfar} , nsw) is calculated by the two Doppler analysis units 210 in the z-th antenna system processing unit 201, for example, as in the following equation (77). This is a value expressed in a vector format of a component extracted from the output VFT _z ^ncsw (f _b , f _s ) based on the distance index f _{b_cfar} and the Doppler frequency index f _{s_cfar} extracted by the CFAR unit 211. However, noc=1,2.

In FIG. 21, the direction estimation unit 214 separates the received signal Y _z (f _{b_cfar} , f _{s_cfar} , ncm,nsw) of the code multiplexed signal with respect to the distance index f _{b_cfar} and the Doppler frequency index f _{s_cfar} input from the code multiplexing and demultiplexing unit 215 . Target direction estimation processing (hereinafter referred to as direction estimation processing for linearly polarized waves) is performed based on the following.

In addition, the direction estimation unit 214 outputs the output that becomes the received signal of the transmission that becomes the normal circularly polarized wave among the outputs of the first to 2Nsw Doppler analysis units 210 (in FIG. 21, the Doppler analysis units 210-1 to 2Nsw). is used to perform target direction estimation processing (hereinafter referred to as direction estimation processing for normal circular polarization).

Further, the direction estimation unit 214 outputs the output that becomes the received signal of the transmission that becomes the anti-circularly polarized wave among the outputs of the first to 2Nsw Doppler analysis units 210 (in FIG. 21, the Doppler analysis units 210-1 to 2Nsw). is used to perform target direction estimation processing (hereinafter referred to as direction estimation processing for anti-rotating circularly polarized waves).

The direction estimation process in the direction estimation unit 214 may include a direction estimation process for linearly polarized waves, a direction estimation process for normal circularly polarized waves, and a direction estimation process for anti-rotating circularly polarized waves. The operation of each direction estimation process will be explained below.

<Direction estimation processing for linearly polarized waves>
For example, the direction estimation unit 214 generates a virtual reception array correlation vector h (f _{b_cfar} , f _{s_cfar} ) as shown in the following equation (78) based on the output of the code demultiplexing unit 215, and performs direction estimation processing. .

The virtual receiving array correlation vector h (f _{b_cfar} , f _{s_cfar} ) includes Nt×Na elements that are the product of the number Nt of transmitting antennas and the number Na of receiving antennas.

Here, hsw _nsw (f _{b_cfar} , f _{s_cfar} ) is the reception vector of the reception signal that is code-separated when transmitted by the transmission antenna 109 that is the nswth transmission pair, and is expressed as the following equation (79). contains 2Na elements, which is the product of the number of transmitting antennas, 2, and the number of receiving antennas, Na.

The direction estimation unit 214 uses the virtual receiving array correlation vector h (f _{b_cfar} , f _{s_cfar} ) to perform direction estimation processing on the reflected wave signal from the target based on the phase difference between each receiving antenna 202 .

Here, the virtual receiver array correlation vector h(f _{b_cfar} , f _{s_cfar} ) includes reflected wave reception signals of signals transmitted from different linearly polarized antennas (eg, a vertically polarized antenna and a horizontally polarized antenna). Therefore, the direction estimation unit 214 extracts an element that is a combination of the polarization of the predetermined transmitting antenna 109 and the polarization of the receiving antenna 202 from the virtual receiving array correlation vector h (f _{b_cfar} , f _{s_cfar} ). The direction estimation process may be performed using a virtual receiving array correlation vector consisting of elements. This makes it possible to obtain direction estimation results for each polarization of the predetermined transmitting antenna 109 and receiving antenna 202.

<Direction estimation processing for normal circular polarization>
The direction estimating unit 214 uses the output of the first to second Nsw Doppler analysis units 210, which is the received signal of the transmission which is the normal circular polarization, to calculate the normal circular polarization virtual reception array correlation vector h _c1 (f _{b_cfar} , f _{s_cfar} ) and performs direction estimation processing for the target's normal circular polarization.

Here, the normal circularly polarized virtual receiving array correlation vector h _c1 (f _{b_cfar} , f _{s_cfar} ) includes Nsw×Na elements. The direction estimation unit 214 uses the right circularly polarized virtual receiving array correlation vector h _c1 (f _{b_cfar} , f _{s_cfar} ) to estimate the direction of the reflected wave signal from the target based on the phase difference between each receiving antenna 202. Perform processing.

For example, as shown in Figure 22, Tx#[1,1] is a horizontally polarized antenna, Tx#[2,1] is a vertically polarized antenna, Tx#[1,2] is a horizontally polarized antenna, and Tx#[ 2,2] is a vertically polarized antenna, and when Code1=[1, 1], Code2=[j -j] is used in code multiplexing, the normal circularly polarized virtual receiving array correlation vector h _c1 (f _{b_cfar} , f _{s_cfar} ) can be expressed as in the following equation (80).

<Direction estimation processing for anti-circularly polarized waves>
The direction estimation unit 214 uses the output of the first to second Nsw Doppler analysis units 210 that becomes the reception signal of the transmission that is the anti-circular polarization to calculate the anti-circular polarization virtual reception array correlation vector h _c2 (f _{b_cfar} , f _{s_cfar} ) and performs direction estimation processing for the anti-circularly polarized wave of the target.

Here, the counter-rotating circularly polarized virtual receiving array correlation vector h _c2 (f _{b_cfar} , f _{s_cfar} ) includes Nsw×Na elements. The direction estimation unit 214 uses the counter-rotating circularly polarized virtual receiving array correlation vector h _c2 (f _{b_cfar} , f _{s_cfar} ) to estimate the direction of the reflected wave signal from the target based on the phase difference between each receiving antenna 202. Perform processing.

For example, as shown in Figure 22, Tx#[1,1] is a horizontally polarized antenna, Tx#[2,1] is a vertically polarized antenna, Tx#[1,2] is a horizontally polarized antenna, and Tx#[ 2,2] is a vertically polarized antenna and Code1=[1, 1], Code2=[j -j] is used in code multiplexing, the counter-rotating circularly polarized virtual receiving array correlation vector h _c2 (f _{b_cfar} , f _{s_cfar} ) can be expressed as in the following equation (81).

Hereinafter, the direction estimation process for linearly polarized waves using the virtual reception array correlation vector h (f _{b_cfar} , f _{s_cfar} ) in the direction estimation unit 214, and the direction estimation process for the linearly polarized wave using the virtual reception array correlation vector h _c1 (f _{b_cfar} , f _{s_cfar} ) ) and the direction estimation process for anti-circular polarization using the anti-circular polarization virtual reception array correlation vector h _c2 (f _{b_cfar} , f _{s_cfar} ). Since this is the same as in Embodiment 1, a description of its operation will be omitted.

Note that the direction estimation unit 214 may perform direction estimation processing using a received signal that includes a combination of different types of polarization. In this case, a direction estimation result that is less dependent on polarization can be obtained. Furthermore, by performing direction estimation processing using more virtual antennas, the reception SNR can be improved and the detection performance of the radar device 10b can be improved. Furthermore, since direction estimation processing is performed using the maximum aperture length of the available virtual reception antenna, angular resolution is also improved. For example, when performing direction estimation processing using all virtual reception antennas, a maximum of (Nt × Na + 2 × N _BF × Na) virtual reception antennas can be used, and more virtual reception antennas than Nt × Na can be used. Antenna available.

Further, the direction estimation unit 214 performs direction estimation processing using the same type of polarized antenna for both transmission and reception as described above, and direction estimation processing using a combination of different types of polarization, and calculates the direction estimation results of both of these. may be used as the direction estimation processing result. As a result, a direction estimation processing result with a high polarization dependence and a direction estimation processing result with a low polarization dependence can be obtained. The results of such direction estimation processing may be input to a processing unit for each object mark (not shown) to perform target object identification processing.

As described above, in the second modification of the first embodiment, a pair of adjacent different linearly polarized antennas (for example, horizontal (H), vertical (V) polarized antennas) has a 90° phase difference and an orthogonal relationship. Code multiplexing is performed using orthogonal codes (for example, orthogonal codes [0°, 90°] and [0°, -90°]). Furthermore, when there are a plurality of such transmitting antennas 109 that form a pair, the radar device 10b performs time division multiplex transmission by switching the transmission time for each pair.

Through this operation, in addition to transmission from different linearly polarized antennas, reflected waves from transmission by two types of circularly polarized waves (for example, right-handed and left-handed circularly polarized waves in mutually opposite directions) are reflected. A received signal can be obtained, and target detection or identification performance can be improved.

Note that some of the transmitting antennas 109 may overlap between the transmitting pairs. For example, as shown in FIG. 23, the transmission switching control unit 110 outputs two transmissions from the transmission switching unit 111 every two transmission cycles, which is a transmission cycle of code length Loc=2 given in the encoding unit 107. Performs control to switch antennas. Further, in the encoding unit 107, an example is shown in which codes Code ₁ = [1 1] and Code ₂ = [j - j] are given.

Here, the transmission switching control unit 110 may hold information regarding transmission switching control (eg, "transmission switching control table") shown in FIG. 24. Transmission switching control section 110 may perform code assignment and switching control for transmitting antennas #1 to #4 based on the transmission switching control table. For example, when the transmission pair index nsw=1, the transmission switching control unit 110 uses transmission antennas Tx#1 and Tx#2 and assigns codes Code ₁ = [1 1] and Code ₂ = [j - j], respectively. do. When the transmission pair index nsw=2, the transmission switching control unit 110 uses the transmission antennas Tx#2 and Tx#3 and assigns codes Code ₁ = [1 1] and Code ₂ = [j - j], respectively. Furthermore, when the transmission pair index nsw=3, the transmission switching control unit 110 uses the transmission antennas Tx#3 and Tx#4 and assigns codes Code ₁ = [1 1] and Code ₂ = [j - j], respectively. do. In this example, the transmitting pair indexes nsw=1 and 2, and the antennas (Tx#2) partially overlap between the transmitting pairs. Furthermore, when the transmission pair indexes nsw=3 and 4, some antennas (Tx#3) overlap between the transmission pairs.

Based on such control by the transmission switching control section 110, the encoding section 107, the phase rotation section 108, and the transmission switching section 111 may be similarly controlled.

Further, the radar receiving section 200b also performs the same operation as that described in FIG. 21.

By partially overlapping the antennas between the transmitting pairs as described above, the direction estimation unit 214 calculates the normal circularly polarized virtual receiving array correlation vector h _c1 (f _{b_cfar} , f _{s_cfar} ) and the counterclockwise circularly polarized virtual receiving array. The number of elements of the array correlation vector h _c2 (f _{b_cfar} , f _{s_cfar} ) can be increased, and the reception SNR of the direction estimation process for normal circularly polarized waves and the direction estimation process for reverse circularly polarized waves can be improved. Further, for example, by devising the antenna arrangement, grating lobes or side lobes during direction estimation can be reduced. Furthermore, by devising the antenna arrangement, the aperture length can be increased and the angular resolution can be improved.

(Embodiment 2)
In the first embodiment, the operation using MIMO multiplex transmission, which is a combination of Doppler multiplexing and code multiplexing, has been described. In Embodiment 1, in a pair where different linearly polarized antennas are arranged adjacent to each other (for example, a pair of a horizontally polarized antenna and a vertically polarized antenna), a common Doppler multiplexed signal (for example, the same amount of Doppler shift) is transmitted. code multiplex transmission is performed using Further, as codes used for code multiplexing, orthogonal codes (for example, orthogonal codes [1 1], [j - j]) are used in which code elements have a phase difference of 90° and are orthogonal. Further, when there are a plurality of pairs in which different linearly polarized antennas are arranged adjacent to each other, code multiplex transmission is performed in these pairs using different Doppler multiplexed signals (for example, different Doppler shift amounts).

Through these operations, in addition to transmitting from different linearly polarized antennas, it is possible to receive reflected waves by transmitting two types of circularly polarized waves (for example, right-handed and left-handed circularly polarized waves in mutually opposite directions). A signal can be obtained, and transmission can be performed using more types of polarized waves than the types of polarized waves of the transmitting antenna 109, and transmission using many polarized waves can be performed using fewer transmitting antennas. Furthermore, by combining the polarized waves of the transmitting antenna 109 and the polarized waves of the receiving antenna 202, more combinations of transmitting and receiving polarized waves can be obtained. In addition, by performing direction estimation processing in the direction estimation unit 214 for each combination of polarization of each transmitting and receiving antenna, for example, when a reflected wave with a characteristic polarization is obtained for each target, target detection performance is improved. Or the identification performance can be improved. Furthermore, by using MIMO multiplex transmission, which is a combination of Doppler multiplexing and code multiplexing, it is possible to achieve the effect of shortening the transmission time compared to transmitting by sequentially switching antennas in a time-division manner.

The present invention is not limited to the operation using MIMO multiplex transmission shown in Embodiment 1, but similar effects can be obtained in operations using other MIMO multiplex transmission.

For example, in this embodiment, in MIMO multiplex transmission that performs Doppler multiplex transmission, transmission processing that performs Doppler multiplexing using a plurality of linearly polarized antennas (for example, processing that performs multiplex transmission by applying different Doppler shift charges) and The operation of transmitting by temporally switching the transmission process of Doppler multiplexing beam transmission using linearly polarized antennas with different antenna arrangements will be described.

Even in such an operation, in addition to transmission from different linearly polarized antennas, at least one type of circularly polarized wave (for example, one type of circularly polarized wave in opposite directions of right-hand rotation and left-hand rotation) is transmitted. A received signal is obtained as a reflected wave transmitted by a polarized wave. For example, it is possible to transmit with more types of polarized waves than the types of polarized waves of the transmitting antenna, and with fewer transmitting antennas, it is possible to transmit with many polarized waves, and an effect similar to that shown in Embodiment 1 can be obtained. It will be done.

FIG. 25 shows a configuration example of the radar device 10c in this embodiment.

In contrast to the configuration of Embodiment 1 (FIG. 1), in FIG. 25, Doppler multiplexing transmission using a plurality of linearly polarized antennas and beam transmission using linearly polarized antennas with different adjacent antenna arrangements is performed. A transmission switching control unit 112 is added that temporally switches and controls multiplexed transmissions. In addition, in FIG. 25, instead of the encoding unit 107, weight coefficient multiplication is performed to obtain circularly polarized waves for adjacent linearly polarized antennas with different antenna arrangements, or zero amplitude is used to turn off transmission for antennas that do not transmit. It has a transmission weight generation section 113 and a transmission weight multiplication section 114 that set and multiply the weight (operation equivalent to transmission off control).

Hereinafter, parts that perform operations different from those in Embodiment 1 will be mainly described.

[Configuration of radar transmitter 100c]
The operation of radar transmission signal generation section 101 is the same as that in Embodiment 1, so a description thereof will be omitted.

The _transmission switching control unit 112 transmits information regarding Doppler shift amounts (hereinafter, an example (referred to as the "Doppler multiple assignment table"). The transmission switching control unit 112 controls each component of the phase rotation amount setting unit 105 based on, for example, a Doppler multiplex assignment table. Further, the transmission switching control unit 112 outputs information regarding the Doppler shift variable setting period (for example, a transmission switching index nsw described later) to the output switching unit 209.

FIG. 26 shows an example of a Doppler multiple assignment table. The table shown in FIG. 26 has the antenna arrangement shown in FIG. 25, Tx #1 to #4 (number of transmitting antennas Nt = 4), number of beam transmitting antennas _NBF = 2, and Doppler shift variable setting period Nsw = 2. This is the Doppler multiple assignment table for the case.

In the example shown in FIG. 26, in the transmission period of nsw=1, two beam transmitting antennas Tx#5 (beam transmission with normal circular polarization using Tx#1 and Tx#2), Tx#6 (Tx #3 and Tx #4 are used to transmit a beam with normal circular polarization), Doppler shift amounts (DOP ₁ and DOP ₂ ) are given with a Doppler multiplex number N _DM (1)=2.

Further, in the example shown in FIG. 26, Doppler shift is applied from four transmitting antennas Tx#1 to Tx#4 with a Doppler multiplex number N _DM (2)=4 in a transmission period of nsw=2.

Here, the transmission cycle of nsw=1 is a transmission cycle that satisfies mod(m-1,Nsw)+1=1. Further, the transmission cycle of nsw=2 is a transmission cycle that satisfies mod(m-1,Nsw)+1=2. Here, m is an index indicating the number of transmissions in the transmission cycle, and m=1,~,Nc. Hereinafter, nsw will be referred to as a "transmission switching index."

Note that, hereinafter, the number of Doppler multiplexing in the transmission cycle of nsw will be expressed as N _DM (nsw). Here, N _DM (1)=2 and N _DM (2)=4 are set, but the setting is not limited to this, and any setting that satisfies the following conditions may be used.

<Setting conditions for Doppler multiplexing number in Doppler shift variable setting period Nsw>
The sum of the Doppler multiplexing numbers in the Doppler shift variable setting cycle Nsw is set to be greater than or equal to the sum (Nt+N _BF ) of the number of transmitting antennas Nt and the number of beam transmissions N _BF , as shown in the following equation (82).

Furthermore, the setting may be such that transmission is performed at least once from each of the transmitting antennas of the number of transmitting antennas Nt and the number of beam transmissions _NBF . Further, in each Doppler shift variable setting cycle, the transmitting antenna 109 and the transmitting antenna used for beam transmission may be set so as not to overlap.

Note that the Doppler multiplex number N _DM (nsw) in each nsw transmission cycle may be set so that Nt≧N _DM (nsw)≧1. Here, nsw=1,~,Nsw.

For example, by setting Nsw=3, the Doppler shift variable setting period may be set to 3 transmission periods. For example, when the transmission switching index nsw=1, the beam transmitting antenna transmits a normal circularly polarized wave, when nsw=2, the beam transmitting antenna transmits a counterclockwise circularly polarized wave, and when the transmission switching index nsw=3, each transmitting antenna 109 (horizontal or vertical polarization).

FIG. 27 shows an example of a Doppler multiple assignment table. The table shown in FIG. 27 has the antenna arrangement shown in FIG. 25, Tx #1 to #4 (number of transmitting antennas Nt = 4), number of beam transmitting antennas _NBF = 2, and Doppler shift variable setting period Nsw = 3. This is the Doppler multiple assignment table for the case. In the example shown in FIG. 27, in the transmission period of nsw=1, two beam transmitting antennas Tx#5 (beam transmission with normal circular polarization using Tx#1 and Tx#2), Tx#6 (Tx #3 and Tx #4 are used to transmit a beam with normal circular polarization), Doppler shift amounts (DOP ₁ and DOP ₂ ) are given with a Doppler multiplex number N _DM (1)=2.

In addition, in the example shown in FIG. 27, in the transmission period of nsw=2, two beam transmission antennas Tx#5 (beam transmission with anti-circular polarization using Tx#1 and Tx#2), Tx#6 (Beam transmission with anti-circular polarization using Tx#3 and Tx#4), Doppler shift amounts (DOP ₁ and DOP ₂ ) are given with Doppler multiplex number N _DM (1)=2. Further, in the example shown in FIG. 27, Doppler shift is applied from four transmitting antennas with a Doppler multiplex number N _DM (3)=4 in a transmission period of nsw=3.

In addition, for example, at each Doppler shift variable setting period, transmission of normal or anti-circular polarization by a beam transmission antenna, and transmission from a transmission antenna (horizontal polarization or vertical polarization) different from the beam transmission antenna. It may be configured to transmit a mixture of circularly polarized waves and linearly polarized waves. For example, the radar transmitting unit 100c multiplex-transmits radar transmission signals by applying the same amount of Doppler shift and transmission weights whose phases differ by 90 degrees when transmitting from at least one set of adjacent transmitting antennas 109. The transmission process and the transmission process of multiplexing radar transmission signals by applying different Doppler shift amounts may be performed in the same transmission cycle.

FIG. 28 shows an example of a Doppler multiple assignment table. The table shown in FIG. 28 has the antenna arrangement shown in FIG. 25, Tx#1 to #4 (number of transmitting antennas Nt=4), number of beam transmitting antennas N _BF =2, and Doppler shift variable setting period Nsw=2. This is the Doppler multiple assignment table for the case. In the example shown in FIG. 28, in the transmission period of nsw=1, one beam transmitting antenna Tx#5 (from Tx#1 and Tx#2 with a Doppler shift amount DOP ₁ is used to transmit a beam with normal circular polarization). ) and two transmitting antennas Tx#3 and Tx#4 (transmission using Doppler shift amounts DOP ₂ and DOP ₃ , respectively), a Doppler shift is given with a Doppler multiplex number N _DM (2)=3.

In addition, in the example shown in FIG. 28, in the transmission period of nsw=2, using the Doppler shift amount DOP ₃ from the other beam transmitting antenna Tx#6 (Tx#3 and Tx#4), the wave becomes a right circularly polarized wave. Doppler shift is given by the Doppler multiplex number N _DM (2) = 3 from the two transmitting antennas Tx#1 and Tx#2 (transmission using Doppler shift amounts DOP ₁ and DOP ₂ , respectively). .

For example, the transmission switching control unit 112 performs the following control on the Doppler shift setting unit 106 based on the above-mentioned Doppler multiple assignment table.

Based on the control from the transmission switching control unit 112, the Doppler shift setting unit 106 uses the Doppler multiplex number N _DM (nsw) in the transmission period of nsw to set a phase rotation for applying the Doppler shift amount DOP _ndm (nsw). The amount φ _ndm (nsw) is set and output to the transmission weight generation section 113. Note that an antenna that does not transmit in the nsw transmission cycle (transmission-off antenna) may be included. Here, ndm=1,~, N _DM (nsw), and nsw=1,~,Nsw.

The setting of the phase rotation amount φ _ndm for applying the Doppler shift amount DOP _ndm (nsw) using the Doppler multiplex number N _DM (nsw) in the phase rotation amount setting unit 105 is the same operation as in the first embodiment. , equally spaced Doppler shift amounts may be set, or unequally spaced Doppler shift amounts may be set.

For example, the following equation (83) may be used as the equally spaced Doppler shift amount.

Further, for example, the following equation (84) may be used as the amount of Doppler shift at irregular intervals.

Here, N _int (nsw) takes an integer value of 0 or more.

Under the control of the transmission switching control unit 112, the transmission weight generation unit 113 sets the transmission weight W _ntx (nsw) consisting of the amplitude and phase for the transmission antennas Tx #1 to #Nt (the number of transmission antennas Nt) as follows. do. Here, ntx=1 to Nt.

(1) When the beam transmission antenna is on, the transmission weight generation unit 113 generates a weight coefficient for circularly polarized waves using different linearly polarized antennas for the transmission antenna 109 that constitutes the beam transmission antenna whose transmission is turned on. Set.

The transmission weight coefficient for circularly polarized waves using different linearly polarized antennas is, for example, if a transmission weight of Amp×exp[jη] is set for one transmitting antenna 109, for the other transmitting antenna 109, The transmission weight is Amp×exp[j(η+π/2)] or Amp×exp[j(η-π/2)]. For example, the phase difference between circularly polarized transmission weights using different linearly polarized antennas is +90° or -90°. By multiplying the transmission weights in this manner, a transmission signal that becomes a right-handed or left-handed circularly polarized wave is transmitted. Hereinafter, when one of the directions of circular polarization is defined as normal rotation, the other direction of circular polarization may be referred to as reverse rotation. Here ξ is an arbitrary phase.

(2) A transmitting antenna that is not included in the beam transmitting antennas that are turned on and whose transmission is turned on is given a predetermined amplitude value Amp that is not zero.

For transmitting antennas that are not included in (1) or (2) and whose transmission is turned off, zero amplitude weights (operation equivalent to transmission off control) are set to turn off transmission.

For example, when the transmission switching control section 112 performs control based on the Doppler multiplex allocation table shown in FIG. 26, the transmission weight generation section 113 may set the transmission weight as described below.

If nsw=1,
W ₁ (1)=Amp×exp[jη], W ₂ (1)=Amp×exp[j(η+π/2)],
W ₃ (1)=Amp×exp[jη], W ₄ (1)= Amp×exp[j(η+π/2)]
or
W ₁ (1)=Amp×exp[jη], W ₂ (1)=Amp×exp[j(η-π/2)],
W ₃ (1)=Amp×exp[jη], W ₄ (1)= Amp×exp[j(η-π/2)]
If nsw=2, W ₁ (2)=W ₂ (2)=W ₃ (2)=W ₄ (2)= Amp

Note that when the beam transmitting antenna is on, the transmitting weight generation unit 113 generates a transmitting weight coefficient for circularly polarized waves using a different linearly polarized antenna for the transmitting antenna 109 that constitutes the beam transmitting antenna whose transmitting is on. may be set. For example, when setting a transmission weight of Amp×exp[jη] for one transmitting antenna 109, the transmitting weight generating unit 113 sets a transmitting weight of Amp×exp[j(η+ξ)] for the other transmitting antenna 109. Alternatively, the transmission weight may be set as Amp×exp[j(η-ξ)].

Here, ξ may be in the range of π/6 to 5π/6 radians (=30° to 150°). For example, when the phase deviation between the transmitting antennas 109 is corrected in advance, beam transmission generates circularly polarized waves in which the main beam direction of the transmitting beam changes depending on ξ. For example, when the beam transmission antenna spacing is λ/2 and ξ=90°, a circularly polarized wave is generated in which the main beam direction is 0° in front.

Further, for example, when ξ=30°, circularly polarized waves are generated in a direction in which the main beam direction is shifted by about -15° from the front direction. Further, for example, when ξ=150°, circularly polarized waves are generated in a direction in which the main beam direction is shifted by about +15° from the front direction. Here, λ is the wavelength of the high frequency signal output from the transmitting antenna (if a chirp signal is used, the wavelength at the center frequency of the chirp signal is used).

Transmission weight generation section 113 sets a transmission weight (hereinafter referred to as "Doppler multiplex transmission weight") WD _ntx that includes the set transmission weight and the amount of Doppler phase rotation, and outputs it to transmission weight multiplication section 114 .

The following equation (85) indicates the Doppler multiplex transmission weight WD _ntx for the ntx-th transmitting antenna in the m-th transmission period Tr. Here, ntx=1,~,Nt.

Note that floor[x] is an operator that outputs the largest integer that does not exceed the real number x. j is an imaginary unit.

The Nt transmission weight multipliers 114 calculate Doppler multiplex transmission weights WD _ntx for each of the Nt transmission antennas 109 for the chirp signal cp(t) input from the radar transmission signal generation section 101 for each transmission period Tr. Multiply each by (m). Note that ntx=1,~,Nt. The outputs from the Nt transmission weight multipliers 114 are amplified to a specified transmission power and then radiated into space from the Nt transmission antennas 109 of the transmission array antenna section.

For example, the ntx-th transmission weight multiplication unit 114 multiplies the m-th chirp signal cp(t) generated by the radar transmission signal generation unit 101 by the Doppler multiplex transmission weight WD _ntx (m), It is output to the ntx-th transmitting antenna 109. For example, the ntx-th transmission weight multiplier 114 applies the Doppler multiplex transmission weight WD _ntx (nsw) to the Nc chirp signals cp(t) generated in each transmission cycle by the radar transmission signal generation section 101. , and output to the ntx-th transmitting antenna.

In this way, the radar transmitting unit 100c transmits data from at least one set of adjacent transmitting antennas 109 (for example, a pair of a horizontally polarized antenna and a vertically polarized antenna) using the same Doppler shift amount and phase. A transmission process in which radar transmission signals are multiplexed by applying transmission weights that differ by 90° and a transmission process in which radar transmission signals are multiplexed by applying different Doppler shift amounts are temporally switched. For example, by using transmission weights whose phases differ by 90 degrees, radar device 10c can use transmission antennas exceeding the number Nt of transmission antennas for multiplex transmission, as in the first embodiment, and the polarization of transmission antenna 109 (for example, It becomes possible to use a transmitting antenna with a polarization different from the horizontal polarization (horizontal polarization and vertical polarization) (for example, circular polarization).

[Configuration of radar receiving unit 200c]
In FIG. 25, the operation of radar receiving section 200c from mixer section 204 to beat frequency analysis section 208 is the same as in the first embodiment.

Based on the transmission switching index nsw input from the transmission switching control unit 112, the output switching unit 209 converts the output of the beat frequency analysis unit 208 for each transmission period into the nsw-th Doppler among the Nsw Doppler analysis units 210. It is selectively switched to and output to the analysis section 210. For example, the output switching unit 209 selects the mod(m−1, Nsw)+1th Doppler analysis unit 210 in the mth transmission period Tr. Here, nsw=1,~,Nsw.

The signal processing unit 206 includes Nsw Doppler analysis units 210-1 to 210-Nsw (or referred to as first to Nsw Doppler analysis units 210). For example, the output switching unit 209 inputs data to the nsw-th Doppler analysis unit 210 every Nsw transmission cycles (Nsw×Tr). The nsw-th Doppler analysis unit 210 generates data of Ncsub=Nc/Nsw transmission cycles among the Nc transmission cycles (for example, the beat frequency response RFT _z (f _b , Doppler analysis is performed for each distance index f _b using m)). Note that Nc is set to an integral multiple of Nsw.

For example, when Ncode is a power of 2, FFT processing can be applied in Doppler analysis. In this case, the FFT size is Ncsub, and the maximum Doppler frequency at which aliasing does not occur, which is derived from the sampling theorem, is ±1/(2Nsw×Tr). Also, the Doppler frequency interval of the Doppler frequency index f _s is 1/(Nsw×Ncsub×Tr), and the range of the Doppler frequency index f _s is f _s = -Ncsub/2,~, 0,~, Ncsub/2− It is 1.

For example, the output VFT _z ^nsw (f _b , f _s ) of the Doppler analysis unit 210 of the z-th signal processing unit 206 is determined by replacing Ncode with Ncsub, replacing Loc with Nsw, and replacing noc with nsw in equation (39). Other than these, the operation is the same as in the first embodiment.

In FIG. 25, the CFAR unit 211 performs CFAR processing (for example, adaptive threshold determination) using the outputs of the Nsw Doppler analysis units 210 of the first to Nath signal processing units 206, and Extract the distance index f _{b_cfar} and Doppler frequency index f _{s_cfar} that give the signal.

For example, the CFAR unit 211 calculates, for each output VFT _z ^nsw (f _b , f _s ) of the nsw-th Doppler analysis unit 210 in the first to Na-th signal processing units 206, as shown in the following equation (87). Two-dimensional CFAR processing consisting of distance axis and Doppler frequency axis (corresponding to relative velocity) or CFAR processing that combines one-dimensional CFAR processing using PowerFT _nsw (f _b , f _s ) with power addition I do. The CFAR unit 211 adaptively sets a threshold value, and calculates the distance index f _{b_cfar} (nsw), Doppler frequency index f _{s_cfar} (nsw), and received power information PowerFT _nsw (f _{b_cfar} (nsw ), f _{s_cfar} (nsw)) to the Doppler demultiplexer 216.

Alternatively, if the number of Doppler multiplexing in each nsw is the same (N _DM (1)=...=N _DM (Nsw)) and the amount of Doppler shift in each nsw is also the same, the CFAR unit 211 calculates the following equation (88). CFAR processing may be performed in common by outputting PowerFT(f _b , f _s ) obtained by adding up the power of all the outputs of the Doppler analysis unit 210 of each nsw. The CFAR unit 211 adaptively sets a threshold, and sends the distance index f _{b_cfar} , Doppler frequency index f _{s_cfar} , and received power information PowerFT (f _{b_cfar} , f _{s_cfar} ), which result in received power greater than the threshold, to the Doppler demultiplexer. Output to.

Note that when, for example, equation (83) is used as the phase rotation amount φ _ndm for giving the Doppler shift amount DOP _ndm , the intervals of the Doppler shift amount in the Doppler frequency domain in the output of the Doppler analysis unit 210 are equal intervals, If the Doppler shift amount interval ΔFD(nsw) is expressed by the Doppler frequency index interval, then ΔFD(nsw)=Ncsub/N _DM (nsw). Therefore, in the Doppler frequency domain, in the output of Doppler analysis section 210, peaks are detected at intervals of ΔFD for each signal subjected to Doppler shift multiplexing, and the Doppler Area compression CFAR processing can be applied. In this case, f _{s_comp} =-ΔFD(nsw)/2,~,- ΔFD(nsw)/2-1. Further, the CFAR unit 211 outputs, for example, a distance index f _{b_cfar} and a Doppler frequency index f _{s_comp_cfar} (nsw) to the Doppler multiplexing/demultiplexing unit 216.

Furthermore, when using, for example, equation (84) as the phase rotation amount φ _ndm for giving the Doppler shift amount DOP _ndm , the intervals of the Doppler shift amount in the Doppler frequency domain in the output of the Doppler analysis unit 210 are unequal intervals. , the Doppler frequency index interval is an integer multiple of the Doppler shift amount interval ΔFD(nsw). Here, ΔFD(nsw)=Ncsub/(N _DM (nsw)+N _int (nsw)). Therefore, in the output of the Doppler analysis unit 210, in the Doppler frequency domain, peaks are detected for each Doppler shift multiplexed signal at intervals of ΔFD(nsw) or at intervals of an integer multiple of ΔFD(nsw), and The Doppler region compression CFAR processing described as the processing of the CFAR unit 211 in the first embodiment can be applied. In this case, f _{s_comp} =-ΔFD(nsw)/2,~,- ΔFD(nsw)/2-1. Further, the CFAR unit 211 outputs, for example, a distance index f _{b_cfar} and a Doppler frequency index f _{s_comp_cfar} (nsw) to the Doppler multiplexing/demultiplexing unit 216.

Below, the operation when applying Doppler region compression CFAR processing will be explained.

In FIG. 25, the nsw-th Doppler demultiplexer 216 receives the distance index f _{b_cfar} (nsw) and Doppler frequency index f _{s_comp_cfar} (nsw) input from the CFAR unit 211, and the output of the nsw-th Doppler analyzer 210. Separate Doppler multiplexed signals based on .

For example, separation of non-uniformly spaced Doppler multiplexed signals using equation (84) is described in Patent Document 5, and detailed explanation will be omitted. Further, the Doppler demultiplexer 216 outputs the output VFT _z ^nsw (f _{b_cfar} , f _{s_comp_cfar} (nsw)) of the nsw-th Doppler analysis unit 210 at the distance index f _{b_cfar} (nsw) and the Doppler frequency index f _{s_comp_cfar} (nsw) and the VFT _The Doppler multiplex signal DOP _ndm (nsw) is specified by using the power information of the Doppler frequency index that is ^z nsw (f _{b_cfar} , f _{s_comp_cfar} (nsw)+ΔFD(nsw)×(integer multiple)). Further, the Doppler multiplexing/demultiplexing section 216 can detect the transmitting antenna 109 assigned to the Doppler multiplexed signal based on the Doppler multiplexing assignment table. Further, the Doppler demultiplexer 216 can detect the Doppler frequency of the target object within a range of ±1/(2Nsw×Tr).

Further, for example, in separating equally spaced Doppler multiplexed signals using equation (83), the Doppler demultiplexer 216 sets the Doppler frequency of the target in the range of ±1/(2Nsw×N _DM (nsw)×Tr). , _{the output VFT z nsw (f b_cfar , f s_comp_cfar (} ^nsw _{) )} and VFT _z ^nsw (f _{b_cfar} , f _{s_comp_cfar} (nsw)+ΔFD(nsw)×(integer multiple)) Doppler multiplexed signal DOP _ndm (nsw) is specified using the Doppler frequency index. Further, the Doppler multiplexing/demultiplexing section 216 can detect the transmitting antenna 109 assigned to the Doppler multiplexed signal based on the Doppler multiplexing assignment table.

In the following, the Doppler frequency of the ndm-th Doppler _multiplex signal of the detected target is calculated as FDP ₍ ndm, f It is written as _{s_comp_cfar} (nsw)).

Through the above operations, the nsw-th Doppler demultiplexer 216 performs Doppler demultiplexing based on the distance index f _{b_cfar} and Doppler frequency index f _{s_comp_cfar} input from the CFAR unit 211 and the output of the nsw-th Doppler analysis unit 210. Perform demultiplexing. Then, the nsw-th Doppler demultiplexer 216 uses the distance index f _{b_cfar} and the output VFT _z ^nsw ( _{f b_cfar} , FDP(1,f _{s_comp_cfar} )), VFT _z ^nsw (f _{b_cfar} , FDP(2,f _{s_comp_cfar} )), ~, VFT _z ^nsw (f _{b_cfar} , FDP(N _DM (nsw),f _{s_comp_cfar} ) as a Doppler multiplexed signal It is output to the direction estimating section 214 as a separated received signal of z=1,~,Na, and nsw=1,~,Nsw.

In FIG. 25, the direction estimation unit 214 performs target direction estimation processing based on the separated received signal of the Doppler multiplexed signal for the distance index f _{b_cfar} and Doppler frequency index f _{s_comp_cfar} input from the Doppler demultiplexer 216.

The direction estimation unit 214 uses the outputs of each nsw-th Doppler analysis unit 210, which are separated received signals of the Doppler multiplexed signal, VFT _z ^nsw (f _{b_cfar} , FDP(1, f _{s_comp_cfar} )), VFT _z ^nsw (f _{b_cfar} , FDP( 2, f _{s_comp_cfar} )),～, For each VFT _z ^nsw (f _{b_cfar} , FDP(N _DM (nsw), f _{s_comp_cfar} ), generate each virtual receiving array correlation vector hs _nsw ( _{b_cfar} , f _{s_comp_cfar} ) and perform direction estimation. As shown in Equation (89), each virtual receiving array correlation vector hs _nsw ( _{b_cfar} , f _{s_comp_cfar} ) includes N _DM (nsw)×Na elements. Using the virtual receiving array correlation vector hs ₁ (f _{b_cfar} , f _{s_comp_cfar} ) to hs _Nsw (f _{b_cfar} , f _{s_comp_cfar} ), direction estimation is performed based on the phase difference between each receiving antenna 202 for the reflected wave signal from the target. Perform processing.

The direction estimation unit 214 calculates the polarization between the predetermined transmitting antenna polarization and the receiving antenna polarization from the Nsw virtual receiving array correlation vectors hs ₁ (f _{b_cfar} , f _{s_comp_cfar} ) to hs _Nsw (f _{b_cfar} , f _{s_comp_cfar} ). Elements to be combined may be extracted and direction estimation processing may be performed using a virtual reception array correlation vector made up of the extracted elements. Thereby, it is possible to obtain direction estimation results for each polarization of a predetermined transmitting antenna and a predetermined receiving antenna.

For example, when the transmission switching control unit ₁₁₂ performs control based on the above _- described Doppler multiplex _{assignment table of} FIG. , f _{s_comp_cfar} ) is obtained.

Here, the virtual receiving array correlation vector hs ₁ (f _{b_cfar} , f _{s_comp_cfar} ) with nsw=1 includes an element as shown in the following equation (90), and the Doppler multiplex number N _DM (1)=2. In addition, since the Doppler multiplexed signal using DOP ₁ and DOP ₂ is a reflected reception signal for the transmission signal of normal circular polarization, the direction estimation unit 214 uses virtual reception in the direction estimation process for normal circular polarization. Direction estimation processing is performed using the array correlation vector hs ₁ (f _{b_cfar} , f _{s_comp_cfar} ).

Further, the virtual reception array correlation vector hs ₂ (f _{b_cfar} , f _{s_comp_cfar} ) with nsw=2 includes an element as shown in the following equation (91), and the Doppler multiplex number N _DM (1)=4. Furthermore, since the Doppler multiplexed signal using DOP ₁ , DOP ₂ , DOP ₃ , and DOP ₄ is a reflected reception signal for a horizontally polarized or vertically polarized transmitted signal, the direction estimation unit 214 determines the direction for linearly polarized waves. In the estimation process, the direction estimation process is performed using the virtual receiving array correlation vector hs ₂ (f _{b_cfar} , f _{s_comp_cfar} ).

Hereinafter, the direction estimation process for linearly polarized waves using the virtual reception array correlation vector h _s2 (f _{b_cfar} , f _{s_cfar} ) in the direction estimation unit 214, and the direction estimation process for linearly polarized waves using the virtual reception array correlation vector h _s2 (f _{b_cfar} , f The direction estimation process for normal circularly polarized waves using _{(s_cfar} ) is the same as in Embodiment 1, and the explanation of its operation will be omitted.

Note that the direction estimation unit 214 may perform direction estimation processing using a received signal that includes a combination of different types of polarized waves. In this case, a direction estimation result that is less dependent on polarization is obtained, and by performing direction estimation processing using more virtual antennas, the reception SNR is improved, and therefore the detection performance of the radar device 10c is improved. Furthermore, since direction estimation processing is performed using the maximum aperture length of the available virtual reception antenna, angular resolution is also improved.

Further, the direction estimation unit 214 performs direction estimation processing using the same type of polarized antenna for both transmission and reception as described above, and direction estimation processing using a combination of different types of polarized waves, and uses the direction estimation results of both of these as the direction estimation process. It may also be an estimation processing result. As a result, a direction estimation processing result with a high polarization dependence and a direction estimation processing result with a low polarization dependence can be obtained. The results of such direction estimation processing may be input to a processing unit for each object mark (not shown) to perform target object identification processing.

Note that the transmission weight W _ntx (nsw) used in the second embodiment represents a code when the phase deviation between the transmitting antennas 109 is corrected in advance. Therefore, when the beam transmission antenna is on, the transmission weight generation unit 113 calculates that the phase difference at each feeding point of the two transmission antennas forming the beam transmission antenna is the phase difference between the transmission weights given to each transmission antenna. becomes.

Here, as the transmission weight given to the two transmitting antennas constituting the beam transmitting antenna, a transmitting weight of Amp×exp[jη] is set for one transmitting antenna 109, and a transmitting weight of Amp×exp[jη] is set for the other transmitting antenna 109. When assigning a transmission weight of Amp×exp[j(η+π/2)] or Amp×exp[j(η-π/2)], the weight between the feeding points of two transmitting antennas is The phase difference is 90° or -90°.

Similarly, as the transmission weight given to the two transmitting antennas constituting the beam transmitting antenna, a transmitting weight of Amp×exp[jη] is set for one transmitting antenna 109, and a transmitting weight of Amp×exp[jη] is set for the other transmitting antenna 109. When assigning a transmission weight of Amp×exp[j(η+ξ)] or Amp×exp[j(η-ξ)], the phase difference between the feeding points of two transmitting antennas in each transmission period is ξ or -ξ. Here, for example, ξ may be in the range of π/6 to 5π/6 radians (=30° to 150°).

Each embodiment of the present disclosure has been described above.

[Other embodiments]
(1) In the antenna arrangement examples shown in FIGS. 15 to 18, as an example, the types of polarization of the transmitting antenna 109 are arranged as "H", "V", "H", and "V" from the left. The arrangement of the transmitting antennas 109 is not limited to this, but may be any arrangement in which adjacent transmitting antennas 109 form a pair of a horizontally polarized antenna (H) and a vertically polarized antenna (V).

For example, in the antenna arrangements shown in FIGS. 15 to 18, the polarization types of the transmitting antenna 109 may be arranged in the order of "H", "V", "V", and "H" from the left. In this arrangement, for example, isolation is high. Furthermore, for example, in the antenna arrangements shown in FIGS. 15 to 18, the polarization types of the transmitting antenna 109 may be arranged in the order of "V", "H", "H", and "V" from the left.

(2) In the radar device according to an embodiment of the present disclosure, the radar transmitting section and the radar receiving section may be individually arranged at physically separate locations. Furthermore, in the radar receiving section according to an embodiment of the present disclosure, the direction estimating section and other components may be individually arranged at physically distant locations.

(3) Number of transmitting antennas Nt, number of receiving antennas Na, number of Doppler multiplexing N _DM , number of codes N _CM , number of beam transmitting antennas N _BF , number of pairs of transmitting antennas, and variable Doppler shift used in an embodiment of the present disclosure The numerical values of parameters such as the setting period are just examples, and the present invention is not limited to these values.

Although not shown, the radar device according to an embodiment of the present disclosure includes, for example, a CPU (Central Processing Unit), a storage medium such as a ROM (Read Only Memory) storing a control program, and a RAM (Random Access Memory). Has working memory. In this case, the functions of each section described above are realized by the CPU executing a control program. However, the hardware configuration of the radar device is not limited to this example. For example, each functional unit of the radar device may be realized as an integrated circuit (IC). Each functional unit may be individually integrated into one chip, or may include a part or all of the functional units into one chip.

Although various embodiments have been described above with reference to the drawings, it goes without saying that the present disclosure is not limited to such examples. It is clear that those skilled in the art can come up with various changes or modifications within the scope of the claims, and these naturally fall within the technical scope of the present disclosure. Understood. Further, each of the constituent elements in the above embodiments may be arbitrarily combined without departing from the spirit of the disclosure.

Furthermore, in the embodiments described above, the expression "...unit" means "...circuitry", "...assembly", "...device", "...unit", or , "...module" may be substituted with other notations.

In each of the above embodiments, the present disclosure has been explained using an example configured using hardware, but the present disclosure can also be realized by software in cooperation with hardware.

Furthermore, each functional block used in the description of each of the above embodiments is typically realized as an LSI, which is an integrated circuit. The integrated circuit may control each functional block used in the description of the above embodiments, and may include an input terminal and an output terminal. These may be integrated into one chip individually, or may be integrated into one chip including some or all of them. Although it is referred to as an LSI here, it may also be called an IC, system LSI, super LSI, or ultra LSI depending on the degree of integration.

Moreover, the method of circuit integration is not limited to LSI, and may be implemented using a dedicated circuit or a general-purpose processor. After the LSI is manufactured, an FPGA (Field Programmable Gate Array) that can be programmed or a reconfigurable processor that can reconfigure the connections or settings of circuit cells inside the LSI may be used.

Furthermore, if an integrated circuit technology that replaces LSI emerges due to advancements in semiconductor technology or other derivative technologies, it is natural that functional blocks may be integrated using that technology. Possibilities include the application of biotechnology.

<Summary of this disclosure>
A radar device according to an embodiment of the present disclosure includes a first transmitting antenna that emits a first linearly polarized wave, and a second transmitting antenna that is adjacent to the first transmitting antenna and that is different from the first linearly polarized wave. a plurality of transmitting antennas including a second transmitting antenna that radiates linearly polarized waves; and a phase rotation in which the phase between the first transmitting antenna and the second transmitting antenna in each transmission period is different by ξ or −ξ. and a transmitting circuit that multiplex transmits the transmission signal to which the amount of transmission is assigned from the plurality of transmitting antennas.

In one embodiment of the present disclosure, the transmission circuit code-multiplexes the transmission signal using a plurality of orthogonal codes, and the transmission circuit transmits the transmission signal from the first transmission antenna among the plurality of orthogonal codes. Between the first orthogonal code for the signal and the second orthogonal code for the transmission signal transmitted from the second transmission antenna, the phase of the code element corresponding to each transmission period differs by ξ or −ξ.

In one embodiment of the present disclosure, ξ has a value in the range of 30° to 150°.

In one embodiment of the present disclosure, the transmitting circuit sets the same amount of Doppler shift to the first transmitting antenna and the second transmitting antenna.

In one embodiment of the present disclosure, when the plurality of transmitting antennas include a plurality of pairs of the first transmitting antenna and the second transmitting antenna, the transmitting circuit is configured to provide a different transmitting circuit for each of the plurality of pairs. Set the amount of Doppler shift.

In an embodiment of the present disclosure, in one of an odd-numbered transmission period and an even-numbered transmission period, a reflected wave signal obtained by reflecting the transmission signal transmitted from the first transmission antenna and the second transmission antenna on a target. a first Doppler analysis circuit that performs Doppler analysis using the reflected wave signal; a second Doppler analysis circuit that performs Doppler analysis using the reflected wave signal in the other of the odd-numbered and even-numbered transmission cycles; a first separation circuit that separates a circularly polarized wave signal using either the output of the first Doppler analysis circuit and the output of the second Doppler analysis circuit; the output of the first Doppler analysis circuit; The apparatus further includes a second separation circuit that separates at least one signal of the first linearly polarized wave and the second linearly polarized wave using both outputs of the second Doppler analysis circuit.

An embodiment of the present disclosure further includes a plurality of receiving antennas, the plurality of receiving antennas including the first linearly polarized wave, the second linearly polarized wave, the circularly polarized wave in the first direction, and an antenna that receives at least one circularly polarized wave in a second direction opposite to the first direction.

In an embodiment of the present disclosure, when the plurality of transmitting antennas include a plurality of pairs of the first transmitting antenna and the second transmitting antenna, the transmitting circuit selects the transmitter from each of the plurality of pairs. Time division multiplex transmission of transmission signals.

In one embodiment of the present disclosure, the transmitting circuit transmits signals from the first transmitting antenna and the second transmitting antenna using the same amount of Doppler shift and different phases by ξ or −ξ. A first transmission process in which the transmission signal is multiplexed by applying a transmission weight and a second transmission process in which the transmission signal is multiplexed by applying a different amount of Doppler shift are temporally switched.

In one embodiment of the present disclosure, the transmitting circuit transmits signals from the first transmitting antenna and the second transmitting antenna using the same amount of Doppler shift and different phases by ξ or −ξ. A first transmission process in which the transmission signal is multiplexed by applying a transmission weight, and a second transmission process in which the transmission signal is multiplexed by applying a different amount of Doppler shift are performed in the same transmission cycle.

In an embodiment of the present disclosure, when the plurality of transmitting antennas include a plurality of pairs of the first transmitting antenna and the second transmitting antenna, the transmitting circuit may perform the A different amount of Doppler shift is set for each of the plurality of pairs.

An embodiment of the present disclosure further includes a plurality of receiving antennas, the plurality of transmitting antennas include a plurality of pairs of the first transmitting antenna and the second transmitting antenna, and the plurality of pairs of transmitting antennas include a plurality of pairs of the first transmitting antenna and the second transmitting antenna. The difference between the spacing between the phase centers of each of the plurality of receiving antennas and the spacing between adjacent receiving antennas among the plurality of receiving antennas is a specified value based on the wavelength of the transmission signal.

In one embodiment of the present disclosure, the specified value is a value in a range of 0.45 times to 0.8 times the wavelength.

In an embodiment of the present disclosure, the embodiment further includes a plurality of receiving antennas, and in the first direction, the distance between the first transmitting antenna and the second transmitting antenna, and the distance between the first transmitting antenna and the second transmitting antenna, The interval between adjacent receiving antennas is a first specified value based on the wavelength of the transmission signal, and in a second direction orthogonal to the first direction, some of the plurality of receiving antennas The distance between one receiving antenna and another receiving antenna is a second specified value based on the wavelength of the transmitted signal.

In one embodiment of the present disclosure, each of the first specified value and the second specified value is a value in a range of 0.45 times to 0.8 times the wavelength.

The present disclosure is suitable as a radar device that detects a wide angle range.

10, 10a, 10b, 10c radar device 100, 100a, 100b, 100c radar transmitter 101 radar transmission signal generator 102 transmitter signal generation controller 103 modulation signal generator 104 VCO
105 Phase rotation amount setting section 106 Doppler shift setting section 107 Encoding section 108 Phase rotation section 109 Transmission antenna 110, 112 Transmission switching control section 111 Transmission switching section 113 Transmission weight generation section 114 Transmission weight multiplication section 200 Radar reception section 201 Antenna system Processing section 202 Receiving antenna 203 Receiving radio section 204 Mixer section 205 LPF
206 Signal processing section 207 AD conversion section 208 Beat frequency analysis section 209 Output switching section 210 Doppler analysis section 211 CFAR section 212 Coded Doppler demultiplexing section 213, 216 Doppler demultiplexing section 214 Direction estimation section 215 Code demultiplexing section 300 Positioning output Department

Claims (15)

a first transmitting antenna that radiates a first linearly polarized wave; and a second transmitting antenna that is adjacent to the first transmitting antenna and radiates a second linearly polarized wave that is different from the first linearly polarized wave. multiple transmitting antennas including;
a transmitting circuit that multiplex transmits, from the plurality of transmitting antennas, transmitting signals to which a phase rotation amount that differs by ξ or -ξ between the first transmitting antenna and the second transmitting antenna in each transmission cycle; ,
A radar device equipped with. The transmission circuit code-multiplexes the transmission signal using a plurality of orthogonal codes,
Among the plurality of orthogonal codes, a first orthogonal code for the transmission signal transmitted from the first transmission antenna and a second orthogonal code for the transmission signal transmitted from the second transmission antenna. between, the phases of the code elements corresponding to each transmission period differ by ξ or -ξ,
The radar device according to claim 1. The ξ is any value in the range of 30° to 150°,
The radar device according to claim 1. The transmitting circuit sets the same amount of Doppler shift to the first transmitting antenna and the second transmitting antenna,
The radar device according to claim 2. When the plurality of transmitting antennas include a plurality of pairs of the first transmitting antenna and the second transmitting antenna, the transmitting circuit sets a different amount of Doppler shift for each of the plurality of pairs,
The radar device according to claim 2. A Doppler analysis is performed using a reflected wave signal of the transmission signal transmitted from the first transmission antenna and the second transmission antenna reflected by the target in either an odd-numbered transmission period or an even-numbered transmission period. 1 Doppler analysis circuit,
a second Doppler analysis circuit that performs Doppler analysis using the reflected wave signal in the other of the odd-numbered and even-numbered transmission cycles;
a first separation circuit that separates circularly polarized signals using either the output of the first Doppler analysis circuit or the output of the second Doppler analysis circuit;
A second method for separating at least one signal of the first linearly polarized wave and the second linearly polarized wave using both the output of the first Doppler analysis circuit and the output of the second Doppler analysis circuit. further comprising: a separation circuit;
The radar device according to claim 1. further comprising a plurality of receiving antennas,
The plurality of receiving antennas receive the first linearly polarized wave, the second linearly polarized wave, the circularly polarized wave in a first direction, and the circularly polarized wave in a second direction opposite to the first direction. an antenna for receiving at least one polarization of
The radar device according to claim 1. When the plurality of transmitting antennas include a plurality of pairs of the first transmitting antenna and the second transmitting antenna, the transmitting circuit time-division multiplex transmits the transmitting signal from each of the plurality of pairs. ,
The radar device according to claim 1. The transmitting circuit, upon transmission from each of the first transmitting antenna and the second transmitting antenna,
a first transmission process of multiplex transmitting the transmission signals by applying transmission weights having the same Doppler shift amount and different phases by ξ or −ξ;
a second transmission process of multiplexing the transmission signal by applying different Doppler shift amounts;
change over time,
The radar device according to claim 1. The transmitting circuit, upon transmission from each of the first transmitting antenna and the second transmitting antenna,
a first transmission process of multiplex transmitting the transmission signals by applying transmission weights having the same Doppler shift amount and different phases by ξ or −ξ;
a second transmission process of multiplexing the transmission signal by applying different Doppler shift amounts;
in the same transmission cycle,
The radar device according to claim 1. When the plurality of transmitting antennas include a plurality of pairs of the first transmitting antenna and the second transmitting antenna, the transmitting circuit performs a different Doppler operation on each of the plurality of pairs in the first transmitting process. Set the shift amount,
The radar device according to claim 8. further comprising a plurality of receiving antennas,
The plurality of transmitting antennas include a plurality of pairs of the first transmitting antenna and the second transmitting antenna,
The difference between the spacing between the phase centers of each of the plurality of pairs and the spacing between adjacent receiving antennas among the plurality of receiving antennas is a specified value based on the wavelength of the transmission signal,
The radar device according to claim 1. The specified value is any value in the range of 0.45 times to 0.8 times the wavelength,
The radar device according to claim 12. further comprising a plurality of receiving antennas,
In the first direction, the distance between the first transmitting antenna and the second transmitting antenna and the distance between adjacent receiving antennas among the plurality of receiving antennas are determined based on the wavelength of the transmitted signal. The default value is 1,
In a second direction orthogonal to the first direction, the distance between some of the plurality of reception antennas and other reception antennas is a second specified value based on the wavelength of the transmission signal. is,
The radar device according to claim 1. Each of the first specified value and the second specified value is a value in a range of 0.45 times to 0.8 times the wavelength,
The radar device according to claim 14.

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